Interleaved 3 Phase DC/DC Converter for Automotive Applications

O. Cornea, N. Muntean, M. Gavris “Politehnica” University of Timisoara/Department of Electrical Engineering, Timisoara, Romania

octavian.cornea@et.upt.ro; nic_muntean@gmail.com; mihaela.gavris@et.upt.ro Abstract- The paper presents an interleaved 3-phase step-down converter that can be used to charge the 12 V battery and to feed the corresponding loads from the 42 V bus, in automotive dual voltage system. The converter has a maximum 3 kW output power and efficiency above 93% at full load. The simulation results show good responses under various input and output conditions.

I. INTRODUCTION

In 1994 the automotive industry realized that the voltage level of electrical systems at that time (12-14V) would not be sufficient for the future vehicles and a new voltage level (42V) was soon proposed [1,11]. Now the 42V power bus is internationally accepted as the automotive electrical system of the future. At this moment, however, the electrical system must be dual voltage architecture because the standardized (cheap) components such as small power electronics and motors, lamps etc. are 14V electrical devices. Thus a dual voltage method is used in the transition period between the existent and the future electrical system. The 42V bus supplies high-power loads and the 14V bus supplies low-power devices.

The dual voltage architecture can have one or two batteries. The structure shown in Fig. 1 has one 42V battery and one 14V battery.

The dc/dc converter is used to charge the 14V battery. The

configuration presented in Fig. 2 has only one 42V battery and the low-power devices are feeded through the 42V/14V dc/dc converter. The dc/dc converter, unidirectional or bidirectional, that interfaces the two voltage buses, is the most important element of the dual-voltage electrical system.

Reference [1] presents a table with the required main features of a bidirectional dc/dc converter for a dual-battery, dual-voltage electrical system of a C segment car. When the

converter is operating in step-down mode, the continuous output power required is 600W and the peak output power is 1 kW. The efficiency of the converter should be 96% at 100% output power and 93% at 30% output power. Obtaining this very high efficiency is a real challenge for the design engineer.

The required peak output power when the converter is operating in step-up mode is only 40% of the value in step-down mode and the efficiency is not very important because this mode of operation is activated only in emergencies and works only for limited periods of time

Due to the tendency of the automotive systems electrification, it was foreseen that the peak output power of the buck converter will rise to 2-3kW or even 5kW.

II. EXISTING SOLUTIONS

Both inductors and capacitors have been used as energy storage elements in the interface between 42V and 14V buses.

The most simple and first used topology was the classical buck converter, which can be used for low power levels. However, when the power increases the buck converter presents several disadvantages which have been discussed in the literature.

For high power levels a multiphase interleaved buck converter can be used [1]. Each converter has its inductor and the output capacitive filter is common. To improve the efficiency the diodes which are normally used to recirculate the current through the inductors can be replaced by power MOSFETs. This is called synchronous rectification. The advantage of the synchronous rectification is the lower voltage drop on the transistors when they are in conduction.

In the interleaved buck converters the output current is the sum of the phase currents. One of the advantages of using a multiphase configuration is the reduction of the total current

Fig. 1. A dual voltage electrical system that has a 36 V battery and a 12 V battery.

Fig. 2. A dual voltage electrical system that has only one 36 V battery.

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ripple which is filtered by the output capacitors; the value of the output filter capacitance is smaller. There is also possible a shift in the technology of the output capacitors. Ceramic capacitors can be used for filtering instead of bulky electrolytic capacitors. This fact has the consequence of reducing the power loss because the ceramic capacitors have much lower values of the equivalent series resistance.

In the interleaved buck converters the input current ripple is reduced and the efficiency is always better compared with classic converter.

If the synchronous rectification is used the same converter can be utilized in step-down mode or in step-up mode, only by changing the control signals.

The interleaved converter can be implemented using off-the-shelf inductors but there are other possibilities to build them. One possibility is to use coupled inductors, where the current slope of one inductor is affected by the voltage across the other inductors [2, 7]. The main advantage in using coupled inductors is that the current ripple in steady-state is reduced in comparison with the case of the uncoupled inductors. If the current ripple is lower, the commutation frequency of the MOSFETs control can be reduced thus lowering the switching losses in power transistors.

Reference [3] presents a 4 phase interleaved buck converter design, in which not all the inductors are coupled. A coupled inductor prototype is compared with an uncoupled inductor prototype. Regarding the coupled design, only the inductors of two phases, that are driven by 180° shifted signals, are coupled together. The windings are integrated in the printed circuit board. Because of the higher value of steady-state inductance the current ripple is reduced and the power losses of the switches and the copper windings are also reduced for the prototype with coupled inductors. The power losses being smaller the temperatures at different measurement points are several degrees lower for the coupled inductor prototype. The dynamic responses of the two prototypes are the same due to the fact that the transient inductance is unchanged.

Two interleaved buck converters design with high number of phases are presented in [4]. Both have the nominal output power of 1 kW; one design has 16 and the other has 32 interleaved phases, in a modular construction. The inductors are integrated and the magnetic cores are added at the moment when the other components are placed on the PCB. The maximum, full load (70A), efficiency of the prototype with 16 phases, is 95%. For the other converter the maximum efficiency is about 93%. The high number of phases brings problems in the control part of the system. The authors used a FPGA custom design to implement the control signals for the 16 and 32 phases.

A converter with nominal output power of 1kW without magnetic components was presented in [5,6]. The converter

efficiency can be higher than 98% but is dependent on the switching frequency and state of charge of the batteries. The structure of the converter, the driver circuits and the control signals tend to be quite complicated but its efficiency cannot be reached by the converters with magnetic components. The converter can be operated at a frequency between 1-10 kHz but the efficiency for 1 kHz is below 90%. It is most efficient at 100% battery voltage and a frequency of 10 kHz.

The output voltage of this type of converter can’t be controlled as in the case of the converter with magnetic components. This can be a disadvantage because the ratio between the voltages of the two batteries is constant and if one battery is overcharged or empty the voltage on the other side of the converter is different than the expected value. The voltage control can be implemented by changing the switching frequency, boosting the complexity of the converter.

The efficiency improvement for this type of converters is obtained mainly by reducing the switching looses because the switching frequency is in the range of 1–10 kHz in comparison to the converters with magnetic components where it is in general 100–150 kHz.

This paper is focused on a 3kW unidirectional dc/dc buck converter that is used to charge the 14V battery from the 42V bus.

III. THE PROPOSED SOLUTION

The main specifications of the proposed interleaved dc-dc converters are the following:

Input Voltage: 40-57 V; Output Voltage: 14 V; Nominal Power (1Q): 3 kW; No of Phases: 3.

The block diagram of the converter is presented in Fig. 3. There are three buck converters in parallel. Each of them has its own synchronization voltage that is shifted from the other two and an output “Power Good”, which indicates if the converter gives the right voltage at the output. The three “Power Good” outputs are combined to give a global indicator of the output voltage which is send to the hierarchical superior system (driven by a microcontroller or a DSP). There is also a global shutdown.

The buck converters are implemented using LTC3810 [8]. It is a current mode synchronous switching regulator controller IC from Linear Technology that can be used for step-down converters. LTC3810 allows an input voltage up to 100V which makes this IC well suited for automotive applications. The main features of this IC, that are important regarding the overall system efficiency, are summarized below:

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- Freewheeling of the inductor current is made through a power MOSFET (the so called “synchronous rectifier” or “active freewhiling”);

- There is no need of a sense resistor. The drain-source voltage of the inferior MOSFET is measured to estimate the inductor current;

- The MOSFET drivers included in LT3810 have very low output impedance for fast switching of the power MOSFET to reduce the switching losses;

- The IC supply is derived from the output through an internal low dropout regulator.

The schematic of one branch of the interleaved converter used for simulation is presented in Fig. 4. There are three identical branches in the interleaved converter, as Figure 3 shows. The schematic presented in Fig. 4 contains also the components necessary for simulation of a step load. The voltage controlled switch SW3 is used to connect or disconnect a resistive load Rload1 = 0.065 Ω.

IV. CONVERTER EFFICIENCY

The converter efficiency is determined using the basic equation (1). In the following is presented the efficiency

calculation at full load for input voltage Vin=57V and output voltage Vout=14V.

lossesout

out

PPP+

=η (1)

The output power is:

outoutout IVP .= (2) If the converter is running at full power, the output current

is 214A. This current is divided between the three branches of the converter; each phase has an average current of 72A at full load.

The main losses of the converter are given below:

1. High side transistor looses (S1) The conduction losses in the high-side transistor are given

by equation (3).

( )

.3.143

2max,

1,,

W

RI

VVP onDST

out

in

outSlosscond

=

=⋅⋅⎟⎟⎠

⎞⎜⎜⎝

⎛⋅= ρ (3)

In equation (3) the drain to source resistance is the typical value RDS(on)=3.3mΩ, the RDS(on) temperature coefficient is ρT=2, which is the maximum value in the transistor datasheet and the output current is Iout,max=214A.

Fig. 3. Block diagram of the proposed converter.

Fig. 4. The schematic of one buck converter.

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The switching looses in the high side transistor at turn-on are given by equation (4).

W

fVVCR

IIV

ft

IIV

P

th

inmillerdr

outin

on

outin

onSlosssw

58.72

23

223

max,

max,

)(1,,

=

⋅⋅⋅⋅⎟⎟⎠

⎞⎜⎜⎝

⎛ Δ−⋅=

⋅⋅⎟⎟⎠

⎞⎜⎜⎝

⎛ Δ−⋅=

(4)

In equation (4) the internal resistance of the MOSFET driver

is RDR=2.5 Ω the miller capacitor is Cmiller = 1 nF, the threshold voltage is Vth=4.5 V and the switching frequency is f = 200kHz.

The switching looses in the high side transistor at turn-off are given by equation (5) where VCC=10V.

W

fVV

VCR

IIV

ft

IIV

P

thcc

inmillerdr

outin

on

outin

offSlosssw

06.152

23

223

max,

max,

)(1,,

=

⋅−

⋅⋅⋅⎟⎟⎠

⎞⎜⎜⎝

⎛ Δ+⋅=

⋅⋅⎟⎟⎠

⎞⎜⎜⎝

⎛ Δ+⋅=

(5)

The total losses in all high side transistors are:

WP sidehighloss 82.11006.15358.733.143, =⋅+⋅+⋅=− (6)

2. Low-side transistor losses (S2) The low-side switch doesn’t have switching losses because

it is in parallel with the Schottky diode D2; it is turned-on and turned-off under zero voltage condition. The low-side switch is formed by two MOSFET transistors in parallel, so the current through one transistor is half the current of the switch. The losses in the low-side switch are given by equation (12).

W

RIV

VVP onDSout

in

outinSloss T

98.21

)(32

2 )(

2

2,

=

⋅⋅⎟⎠⎞

⎜⎝⎛

⋅⋅−⋅= =ρ

(7)

The total losses of the low side transistors are:

WP sidelowloss 89.6598.213, =⋅=− (8)

3. Inductors The inductor has conduction losses and core losses. The

conduction losses are given by equation (9) and the core losses can be taken from the inductor datasheet [8]. We use PA1294.450 low-loss inductors from Pulse Engineering. The typical dc resistance of PA1294.450 is 0.38mΩ, and the maximum value is 0.48mΩ. Two inductors in series were used to realize 0.9μH so the losses must be multiplied by 2.

W

RIR Lout

inductorlosscond

8.53

2max,

,,

=

=⋅⎟⎠⎞

⎜⎝⎛= (9)

The maximum core losses in an inductor at a switching frequency of 200 kHz are 0.7 W. The total losses in all inductors are given by equation (10).

WWWP inductorloss 397.068.56, =⋅+⋅= (10)

Neglecting the capacitor and the driver looses, the total looses are given by equation (11).

6.215,,, =++= −− inductorlosssidelowlosssidehighlosstot PPPP (11)

The efficiency of the converter related to equation (1) is:

%3.9367.2153000

3000, =

+=inductorlossη

(12)

The converter was simulated in LTSpice [10]. The efficiency of the converter was determined from the simulation data, dividing the output power by the average input power of the converter, which was obtained integrating the instantaneous input power over a number of switching periods. The result was 93% at full output power.

V. SIMULATION WORK AND RESULTS

Fig. 5 – 8 present simulation results obtained using the complete simulation model of the interleaved converter.

Fig. 5 shows the start-up of the converter and a load step response. The output voltage is settled at 14V in less than 0.5ms. In this time the 3 phase currents are limited at about 150A.

The load resistance is modified from Rload1 || Rload2 to Rload2, opening the voltage controlled switch SW3 at t = 4 ms. The controller responds very fast at this load step and continues to operate in steady-state at 14W output power. After 1ms the switch SW3 is closed and the operation is settled again at 3kW output power.

Fig. 6 (a detail of Fig. 5) presents the output voltage, the phase currents and the synchronization voltages at steady-state. After 4 ms the phase currents are completely synchronized with the synchronization voltages and the converter operates in steady state. In the first 4 ms the synchronization is not complete and the time intervals between the phase currents are not equal.

Fig. 7 presents the waveforms of the phase currents when the input voltage contains an ac component of 7V/10kHz superposed on the dc component of 50V. The converter is operating at 25% full power. The amplitude of the phase currents is automatically modified to compensate for the variation in the input voltage.

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Fig. 5. Load step response of the interleaved converter.

Fig. 6. Output voltage, phase currents and synchronization voltages

at steady-state.

Figure 7. The waveforms of the phase currents when the input

voltage has an ac component.

Figure 6 presents the output voltage, the phase currents and

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Figure 8. The converter response to the input voltage steps

The converter response to a load step from 100% to 25% output power when the input voltage contains an ac component in presented in Fig. 8. This simulation was carried out to test the response of the interleaved converter when the input condition and the output condition are changed simultaneously.

VI. CONCLUSIONS The paper presents an interleaved buck converter for the

automotive dual-voltage electrical system. The maximum output power of 3 kW is considered to meet the future specification of the automotive industry. The converter uses synchronous rectification and a commercial specialized IC to boost the efficiency.

The calculated efficiency is above 93% at full power, which is close to the value obtained from the simulation data. The simulation results show that the converter responds well to input voltage and load variations. The prototype is under construction and the experimental results are expected soon.

ACKNOWLEDGMENT

The authors gratefully acknowledge the support received from the FP7 “EE-VERT” Grant SCS7-GA-2008-218598.

This work was partially supported by the strategic grant POSDRU 2009 project ID 50783 of the Ministry of Labor, Family and Social Protection, Romania, co-financed by the European Social Fond – Investing in People.

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