multi-band rejection emi shielding

A Project report on

Multi-Band Rejection EMI Shield Using

Frequency Selective Surface

Submitted by

Sourav Rakshit [Roll: 25300311094 Reg. No. : 112530110157]

Souradeep Sinha [Roll: 25300311092 Reg. No. : 112530110155]

Souradip Mukherjee [Roll: 25300311093 Reg. No. : 112530110156]

Sumanta Chakraborty [Roll: 25300311107 Reg. No. : 112530110170]

As a partial fulfillment for the award of the degree of Bachelor of

Technology in Electronics and Communication Engineering of

West Bengal University of Technology

Under supervision of

Mr. Syed Majdur Rahim

Department of Electronics and Communication Engineering

Supreme Knowledge Foundation Group of Institutions

Mankundu, Hooghly, W.B.-712139, India

(Affiliated to West Bengal University of Technology)

May, 2015

CERTIFICATE OF APPROVAL

This is to certify that the final year project report entitled “Multi-Band Rejection EMI Shield

Using Frequency Selective Surface”, submitted by Souradeep Sinha (25300311092),

Souradip Mukherjee (25300311093), Sourav Rakhshit (25300311094) and Sumanta

Chakraborty (25300311107) to the Department of Electronics and Communication

Engineering, Supreme Knowledge Foundation Group of Institutions, Mankundu,

Hooghly, West Bengal 712139, India, for the partial fulfillment of the B. Tech. course is

bonafide record of work carried out by his/her under my guidance and supervision.

SIGNATURE OF HEAD OF THE DEPARTMENT

SIGNATURE OF THE SUPERVISOR

Mr. Soumen Khatua

Head of the Department,

Department of ECE,

Supreme Knowledge Foundation Group of

Institutions, Mankundu, Hooghly,

West Bengal 712139, India.

Mr. Syed Majdur Rahim

Teaching and Technical Assistant,

Department of ECE,

Supreme Knowledge Foundation Group of

Institutions, Mankundu, Hooghly,

West Bengal 712139, India.

Declaration

We, Souradeep Sinha, Souradip Mukherjee, Sourav Rakshit and Sumanta Chakraborty of Fourth Year

(8th Semester), B.Tech in Electronics and Communication Engineering, Supreme Knowledge

Foundation Group of Institutions (SKFGI), hereby declare that the project entitled “Multi-Band

Rejection EMI Shield Using Frequency Selective Surface” has been carried out independently by

us at SKFGI Campus, Mankundu during our fourth year under the valuable guidance of Mr. Syed

Majdur Rahim (Teaching and Technical Assistant, Electronics and Communication Engineering,

SKFGI).

No part of this has been submitted for the award of degree or diploma in any way of the university or

institutions previously.

Date:

Place: Mankundu

__________________ __________________ __________________ __________________

SOURADEEP SINHA SOURADIP MUKHERJEE SOURAV RAKSHIT SUMANTA CHAKRABORTY

Department of Electronics and Communication Engineering

Supreme Knowledge Foundation Group of Institutions (SKFGI)

Mankundu

Acknowledgement

First of all, we express our gratitude to Prof. (Dr.) Abhijit Lahiri, Campus Director,

Supreme Knowledge Foundation Group of Institutions (SKFGI), Prof. (Dr.) T. K. Sengupta,

Chief Technical Director, Supreme Knowledge Foundation Group of Institutions (SKFGI) And

Prof. (Dr.) B. N. Biswas, Chairman (Education Division) SKFGI, for giving us the opportunity to

carry out our B.Tech final year project work at SKFGI.

There are number of people we owe deeply for this work Foremost in our mind is our

project advisor Mr. Syed Masdur Rahim (Teaching and Technical Assistant, Electronics and

Communication Engineering, SKFGI) who set an exceptional example of institution, perseverance

and inspiration.

We are personally thankful to Mr. Soumen Khatua (HOD, Electronics and

Communication Engineering, SKFGI), for giving us the opportunity to undergo our B.Tech final

year project.

Dedicated to our Parents

Abstract

For almost a century, the subject of electromagnetic shielding was limited to extremely low

frequency and very low frequency. The interest originated from the necessity to protect circuits

and radio-receiving apparatus from disturbing effects of radiated fields. Electromagnetic

interference (EMI) shielding research is highly consolidated nowadays due to the emerging

mobile phone and satellite technologies. The radio devices directly or indirectly impose

potential hazards to human health and there exists a growing concern in malfunctioning of

high-speed communication systems due to radiated interference.

This has led to find out the solutions for effective isolation from the interference signals.

Although the main objective is to obtain all stop filter characteristics, the ventilation

requirements force the enclosure to allow certain band of EMI to pass.

This paper presents the design and fabrication of an ultra-thin and flexible electromagnetic

interference (EMI) shield that is capable of rejecting multiple unwanted frequencies. The

design starts with the idea of unit cell model that leaded us to determine the initial geometrical

dimensions of the rings efficiently. Then it followed by full-wave electromagnetic simulation

to fine-tune the final dimensions for the desired frequency response. Impacts of various

geometrical designs on the EMI shielding performance of the concentric ring design are

analyzed and discussed. With these results, an ultra-thin and flexible EMI shield is fabricated

using the screen printing technique. Good correlation between measurement and simulation is

demonstrated in this paper.

Contents

Page No.

List of Figures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .i-iii

1. Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1-3

1.1 Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .1

1.2 Brief History of EMI Regulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

1.3 Standards . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .2

1.4 Example Applications of EMI Shielding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

1.5 Materials used for EMI Shielding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .3

2. Electromagnetic Interference (EMI) and EMI Shielding . . . . . . . . . . . . . . . . 4-14

2.1 Types of Electromagnetic Interference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

2.1.1 Type of EMI as per the way it was created . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

2.1.2 Type of EMI as per its duration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5

2.1.3 Type of EMI as per bandwidth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .5

2.2 Electromagnetic Interference (EMI) Shielding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

2.2.1 What is EMI Shielding? . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6

2.2.2 How Does Electromagnetic Shielding Work? . . . . . . . . . . . . . . . . . . . . . . . . . 6

2.2.3 Need of EMI Shielding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .6

2.2.4 Materials Used for Shielding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

2.3 Shielding Theory . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .8

2.3.1 EM Field Generation and Shielding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

2.3.2 Generation and Propagation of EM Fields . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9

2.4 Introduction to Shielding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .11

2.4.1 Suppression (Shielding) of EM Fields . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

2.4.2 Field Strength Through Shield . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

2.4.3 Shielding and Gasketing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

3. Introduction to Frequency Selective Surface (FSS) . . . . . . . . . . . . . . . . . . . . 15-36

3.1 Background . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16

3.2 History . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

3.3 Advantages and disadvantages . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .18

3.4 Floquet's Principle . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

3.5 Field equations for 2D PEC frequency selective surfaces . . . . . . . . . . . . . . . . . . . . .18

3.6 Plane wave expansion of the fields in source-free media . . . . . . . . . . . . . . . . . . . . . 19

3.7 Elements of Design in Traditional FSS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

.20

3.7.1 Element Geometries . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20

3.7.2 Element Dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

3.8 Traditional FSS Design, Characterization, and Applications . . . . . . . . . . . . . . . . . . .

23

3.8.1 Brief Design Concept . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

3.8.2 Overview of the Elements of Traditional FSS . . . . . . . . . . . . . . . . . . . . . . . . 24

3.8.3 Group II- loop type structures . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .27

3.8.4 Most Significant Traditional Applications of FSS . . . . . . . . . . . . . . . . . . . . . 33

3.8.5 Chapter Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36

4. Some Significant Works on Frequency Selective Surface . . . . . . . . . . . . . . . 37-42

4.1 A Single-Layer FSS Surface for Ultra-Wideband EM Shielding . . . . . . . . . . . . . . . 37

4.2 An EMI Shielding FSS for Ku-Band Applications . . . . . . . . . . . . . . . . . . . . . . . . . . .39

4.3 A novel shield for GSM 1800MHz band using frequency selective surface . . . . . . . 40

4.3.1 Proposed FSS Geometry . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .41

4.4 Electromagnetic Shielding Glass of Frequency Selective Surface . . . . . . . . . . . . . . .42

4.4.1 Outline of FSS Glass . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .42

5. Design and Simulation of Multi-Band EMI Rejection Shield . . . . . . . . . . . .43-52

5.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .43

5.2 Design of Multiple Band Stop EMI Shield . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .43

5.3 Parameters and Geometrical Dimensions for Triple Band Stop EMI Shield . . . . . . . 46

5.4 Setup of Full Wave Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47

5.5 Result and Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .48

5.5.1 Analysis of Full Wave Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .48

5.5.2 Single Ring with Different Radii . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .52

6. Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

7. References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .54-56

i

List of Figures

Figure No. Title Page No.

1.1 Cross-section through a coaxial cable showing shielding . . . . . . . . . . . . . . . . .2

And other layers

1.2 Military electronic equipment . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

1.3 Medical electronic device . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

2.1 Cables with the electromagnetic shielding to Separate . . . . . . . . . . . . . . . . . . . 6

the wires from outside environments

2.2 Sheet metal used to make electromagnetic shielding . . . . . . . . . . . . . . . . . . . . 7

2.3 Microwave oven having electromagnetic shielding . . . . . . . . . . . . . . . . . . . . . 7

2.4 Current through a receiver circuit on a PC card . . . . . . . . . . . . . . . . . . . . . . . . 8

2.5 Field penetrating metallic shield barrier causing current . . . . . . . . . . . . . . . . . 9

to get attenuated

2.6 Applying AC voltage source into a pair of parallel plates . . . . . . . . . . . . . . . . 9

2.7 Lines of flux with respect to E and H field . . . . . . . . . . . . . . . . . . . . . . . . . . . .9

2.8 Placing a shielding barrier in the path of the EM field . . . . . . . . . . . . . . . . . . 11

2.9 Field penetrating barrier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

2.10 The flow of current through a shield including a gasket interface . . . . . . . . . .12

2.11 The current flowing across a gasketed maintenance cover . . . . . . . . . . . . . . . .14

3.1 FSS types and response . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .15

3.2 Radomes at the Cryptologic Operations Center, Misawa, Japan . . . . . . . . . . . 16

3.3 Two-dimensional periodic array of patch elements . . . . . . . . . . . . . . . . . . . . . 17

3.4 Periodic structures comprising of complimentary elements, . . . . . . . . . . . . . 20

patches and slots (wire-grid), and their surface impedance

3.5 A variety of FSS elements over past decades . . . . . . . . . . . . . . . . . . . . . . . . . . . . .21

3.6 A periodic array of infinitely long metallic strips (top) . . . . . . . . . . . . . . . . . . 24

And an infinite array of closely spaced dipoles (bottom)

3.7 The first and the second resonant modes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

3.8 Typical FSS Elements classified in four major groups. . . . . . . . . . . . . . . . 26

based on their shapes

ii

3.9 Reflection coefficient curve for closely packed FSS of. . . . . . . . . . . . . . . . . .27

three-legged loaded elements

3.10 Same as in Fig. 3.9 but frequency range 0 to 40 GHz. . . . . . . . . . . . . . . . . . . . 28

3.11 Same FSS as in Fig. 3.9 but normal angle of incidence and. . . . . . . . . . . .29

frequency range 0 to 40 GHz

3.12 Current distribution on a three-legged loaded element. . . . . . . . . . . . . . . . . . 30

at the (a) first even resonance at about 11 GHz; (b) second

even resonance from about 30 to 40 GHz; (c) first odd

resonance at about 25 GHz

3.13 By reducing the transmission line spacing we obtain a. . . . . . . . . . . . . . . . .31

significant reduction in bandwidth compared to Fig. 3.8

3.14 Reflection coefficient curves for an FSS of closely packed. . . . . . . . . . . .32

hexagon elements

3.15 Application of FSS as radome covers in the aircraft technology. . . . . . . . . . . 33

for reducing the antenna RCS

3.16 Dual-frequency reflector antenna using an FSS as the sub reflector. . . . . . . .34

3.17 An active L-C array comprising metallic strips interrupted. . . . . . . . . . . . 35

by gaps in a periodic fashion.

3.18 Modes of operation for an active L-C array (a) Transmission mode. . . . . . 36

for adjusting the wave intensity (b) Reflection mode for changing

the phase

4.1 (a) Top surface: cross dipole element (b) Bottom . . . . . . . . . . . . . . . . . . . . . 37

surface: ring patch Element (c) Perspective view: both

4.2 Measurement setup showing the FSS prototype, two ridge. . . . . . . . . . . . . . . .38

horn antennas and network analyzer connected via 50-Ω SMA

cables

4.3 Comparison of the numerically computed transmission through . . . . . . . . . . .38

the FSS, with only the cross dipoles, with only the rings and with

both cross dipoles and rings

4.4 An element of the FSS coated on a flat glass . . . . . . . . . . . . . . . . . . . . . . . .39

4.5 Measurement data of reflection (S11) for the square flat Glass . . . . . . . . . . . .40

coated with and without the proposed FSS

4.6 Unit cell geometry. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41

iii

4.7 (A): TE and TM mode characteristics (B): Comparison of the. . . . . . . . . . . . 41

transmission characteristics

4.8 Attenuation peak. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .42

5.1 Geometry of a unit cell and periodic array of single-ring and multiple. . . . . 43

concentric rings

5.2 Geometry of a unit cell of multiple concentric rings. . . . . . . . . . . . . . . . . . . . .45

5.3 Equivalent circuit and two-port ABCD network representations. . . . . . . . . . . .45

of single ring

5.4 3D Model of a unit cell of multiple concentric rings. . . . . . . . . . . . . . . . . . . . .46

5.5 Parameters and geometrical dimensions concentric ring. . . . . . . . . . . . . . . 47

5.6 Experimental and simulation result of tri-band rejection EMI shield. . . . . . . .48

5.7 First stop band measured at -3 dB. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48

5.8 First stop band measured at -10 dB. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

5.9 Second stop band measured at -3 dB. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

5.10 Second stop band measured at -10 dB. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50

5.11 Third stop band measured at -3 dB. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50

5.12 Third stop band measured at -10 dB. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51

5.13 Full-wave simulated and estimated transmission coefficients. . . . . . . . . . . . . .52

for different radii

4

CHAPTER – 1

Introduction

Multi-Band Rejection EMI Shield Using FSS Dept. of ECE, SKFGI

1

Introduction 1__

1.1 Overview

With the exponential growth of wireless communications, the number of base stations is

expected to increase for better service coverage. As a result, there is a strong likelihood of

electromagnetic interference (EMI) from these wireless communications to other sensitive

electronic devices. EMI mitigation, such as shielding, has been gaining attention. To protect

sensitive equipment from potential EMI threat, there is an increasing demand for architectural

shielding. The conventional shielded enclosures are heavy and add structural loadings to

existing buildings hence, light weight EMI shields that are ultra-thin, highly flexible and can

be applied to existing walls of a building will be an attractive solution. Besides the weight

issue, conventional metallic enclosures have no frequency selective shielding feature and

practically block out electromagnetic waves of all frequencies. For EMI shield that only block

out several undesirable frequencies, frequency selective surface (FSS) design maybe applied

to offer such capability. The FSS design also allows void areas to be implemented on the shield,

leading to another desirable property, the optical transparency. It is to be noted that in order to

achieve multi-band frequency selective surfaces, cascaded FSSs are usually used. The cascaded

nature of the design leads to relatively thick shield. In order to implement the design onto a

single layer, loop design is a good choice. By taking advantage of printed periodic elements to

provide the frequency selective feature, the EMI shield can reject specific unwanted

frequencies without affecting other wireless services. Here describes the design procedure of

an ultra-thin and flexible multiple-band rejection EMI shield. Screen-printing technique is

adopted for the fabrication of the EMI shield because of its roll-by-roll mass production

capability. A prototype based on screen-printing is fabricated and its multi-band rejection

capability is demonstrated experimentally.

1.2 Brief History of EMI Regulation

Since the advent of radio communications, EMI’s negative effects have been observed from

both intentional and unintentional sources. The International Electrotechnical Commission

(IEC) met in 1933 in order to recommend that the International Special Committee on Radio

Interference (CISPR) be created in order to help deal with the growing EMI problem. The

committee then created technical documentation that produced the first measurement and

testing techniques to be used in industry along with emission limitations. These regulations

have since evolved into the basic electromagnetic transmission regulations that are in place

Introduction [2]


today. In the United States, the FCC put legal limitations on electromagnetic emissions

throughout the country. Today, most developed countries have some level of EMI regulation

in place to help ensure a higher level of performance across all industries.

1.3 Standards

The International Special Committee for Radio Interference or CISPR (French acronym for

"Comité International Spécial des Perturbations Radioélectriques"), which is a committee of

the International Electrotechnical Commission (IEC) sets international standards for radiated

and conducted electromagnetic interference. These are civilian standards for domestic,

commercial, Industrial and Automotive sectors. These standards form the basis of other

regional and national standards most notably the European Norms (EN) written by CENELEC

(European committee for electrotechnical standardization).

1.4 Example Applications of EMI Shielding

One example is a shielded cable, which has

electromagnetic shielding in the form of a wire mesh

surrounding an inner core conductor. The shielding

impedes the escape of any signal from the core

conductor, and also prevents signals from being added

to the core conductor. Some cables have two

separate coaxial screens, one connected at both ends,

the other at one end only, to maximize shielding of both

electromagnetic and electrostatic fields.

The door of a microwave oven has a screen built into the window. From the perspective

of microwaves (with wavelengths of 12 cm) this screen finishes a Faraday cage formed

by the oven's metal housing. Visible light, with wavelengths ranging between 400 nm

and 700 nm, passes easily through the screen holes.

RF shielding is also used to prevent access to data stored on RFID chips embedded in

various devices, such as biometric passports.

NATO specifies electromagnetic shielding for computers and keyboards to prevent

passive monitoring of keyboard emissions that would allow passwords to be captured;

consumer keyboards do not offer this protection primarily because of the prohibitive

cost.

Figure 1.1: Cross-section through a coaxial cable showing shielding and other layers

Introduction [3]


RF shielding is also used to protect medical and

laboratory equipment to provide protection against

interfering signals, including AM, FM, TV, emergency

services, dispatch, pagers, ESMR, cellular, and PCS. It

can also be used to protect the equipment at the AM,

FM or TV broadcast facilities.

Shielding medical electronic devices is significant design

issue due to the pervasiveness of electronic equipment in

the hospital. Autocatalytic selective plating and

conductive paint are applied onto many medical

electronic device enclosures to provide EMI shielding.

1.5 Materials used for EMI Shielding

Typical materials used for electromagnetic shielding include sheet metal, metal screen,

and metal foam. Any holes in the shield or mesh must be significantly smaller than

the wavelength of the radiation that is being kept out, or the enclosure will not effectively

approximate an unbroken conducting surface.

Another commonly used shielding method, especially with electronic goods housed in plastic

enclosures, is to coat the inside of the enclosure with a metallic ink or similar material. The ink

consists of a carrier material loaded with a suitable metal, typically copper or nickel, in the

form of very small particulates. It is sprayed on to the enclosure and, once dry, produces a

continuous conductive layer of metal, which can be electrically connected to the chassis ground

of the equipment, thus providing effective shielding.

RF shielding enclosures filter a range of frequencies for specific conditions. Copper is used for

radio frequency (RF) shielding because it absorbs radio and magnetic waves. Properly designed

and constructed copper RF shielding enclosures satisfy most RF shielding needs, from

computer and electrical switching rooms to hospital CAT-scan and MRI facilities.

Figure 1.2: Military Electronic Equipment

Figure 1.3: Medical Electronic Device

http://en.wikipedia.org/wiki/Sheet_metal

http://en.wikipedia.org/wiki/Metal_foam

http://en.wikipedia.org/wiki/Wavelength

http://en.wikipedia.org/wiki/Copper

http://en.wikipedia.org/wiki/Nickel

http://en.wikipedia.org/wiki/Copper_in_architecture#Radio_frequency_shielding


http://en.wikipedia.org/wiki/Radio_wave

http://en.wikipedia.org/wiki/Magnetic_wave


http://en.wikipedia.org/wiki/CAT-scan

http://en.wikipedia.org/wiki/MRI

CHAPTER – 2

Electromagnetic Interference (EMI) and

EMI Shielding

4

Electromagnetic Interference (EMI) 2__

And EMI Shielding

Electromagnetic interference (EMI, also called radio-frequency interference or RFI

when in radio frequency) is disturbance that affects an electrical circuit due to either

electromagnetic induction or electromagnetic radiation emitted from an external source. The

disturbance may interrupt, obstruct, or otherwise degrade or limit the effective performance of

the circuit. These effects can range from a simple degradation of data to a total loss of data.

The source may be any object, artificial or natural, that carries rapidly changing electrical

currents, such as an electrical circuit, the Sun or the Northern Lights.

Electromagnetic Interference (EMI) most commonly occurs in the 104 to 1012 Hertz frequency

range of the electromagnetic spectrum. A number of sources create this interference, including

radio transmitters, electric motors, power lines, fluorescent lights, and computer circuits. If

electrical equipment do not have suitable EMI shielding in place, device failure may result

from the interference due to the number of sensitive electronic components in most electronic

equipment produced today. Although there are many national regulations that restrict products

emissions today, taking into account EMI shielding for organically and non-organically created

EMI is still a fundamental part of the electronic design process.

Electromagnetic interference negatively impacts an electrical circuit due to direct interference

from RF transmissions or electromagnetic induction. This interference may degrade, interrupt,

obstruct, or otherwise limit the electronic circuit’s performance. Interference may occur

naturally or other electronic equipment may generate it. EMI can be generated purposely in

order to jam radios or radars as a form of electronic warfare.

2.1 Types of Electromagnetic Interference

EMI - Electromagnetic Interference can arise in many ways and from a number of sources. The

different types of EMI can be categorized in a number of ways.

2.1.1 Categorizing the type of EMI by the way it was created:

Man-made EMI: This type of EMI generally arises from other electronics circuits,

although some EMI can arise from switching of large currents, etc.

Naturally occurring EMI: This type of EMI can arise from many sources - cosmic

noise as well as lightning and other atmospheric types of noise all contribute.

Electromagnetic Interference (EMI) and EMI Shielding [5]


2.1.2 Categorizing the type of EMI is by its duration:

Continuous interference: This type of EMI generally arises from a source such as a

circuit that is emitting a continuous signal. However background noise, which is

continuous may be created in a number of ways, either manmade or naturally occurring.

Impulse noise: Again, this type of EMI may be man-made or naturally occurring.

Lightning, ESD, and switching systems all contribute to impulse noise which is a form

of EMI.

2.1.3 Categorizing the different types of EMI by their bandwidth:

Narrowband: Typically this form of EMI is likely to be a single carrier source -

possibly generated by an oscillator of some form. Another form of narrowband EMI is

the spurious signals caused by intermodulation and other forms of distortion in a

transmitter such as a mobile phone of Wi-Fi router. These spurious signals will appear

at different points in the spectrum and may cause interference to another user of the

radio spectrum. As such these spurious signals must be kept within tight limits.

Broadband: There are many forms of broadband noise which can be experienced. It

can arise from a great variety of sources. Man-made broadband interference can arise

from sources such as arc welders where a spark is continuously generated. Naturally

occurring broadband noise can be experienced from the Sun - it can cause sun-outs for

satellite television systems when the Sun appears behind the satellite and noise can

mask the wanted satellite signal. Fortunately these episodes only last for a few minutes.



2.2 Electromagnetic Interference (EMI) Shielding

2.2.1 What is EMI Shielding?

Electromagnetic shielding is designed to limit the influence of electromagnetic fields and

radiation on a device or object. The process uses a barrier made from conductive material

containing electric charges of either positive or negative properties at the subatomic particle

level. Usually, this material is used to separate the electrical components on the inside of the

device from the outside world. Cables also utilize the concept to separate wires from outside

environments. When used to block radio frequencies, it is known as RF shielding.

The exact purpose of this shielding is to protect devices from the coupling effect, the transfer

of one form of energy to a device that uses a different form. This is commonly caused by radio

waves, electrostatic fields, and the full spectrum of electromagnetic radiation. The full level of

protection is based on the amount of reduction to the electric and magnetic fields. This depends

on the size, shape and orientation of the shielding. No matter the standards in place, however,

shielding cannot protect against low-frequency magnetic fields.

2.2.2 How Does Electromagnetic Shielding Work?

Electromagnetic shielding provides “immunity” for electronic components that are susceptible

to EMI and prevents the same components from transmitting excessive interference to their

surrounding environment. The main method for doing this now entails circuit grounding and

design and placement of the critical components in the device architecture. EMI shielding

compounds and Faraday cages are also used in a housing technique to further shield a device’s

transmissions and protect against outside interference.

2.2.3 Need of EMI Shielding

Today’s electrical and electronic devices are

subject to mandatory EMC requirements

throughout the world. Many devices operate

at high frequencies and are very small. They

are placed in nonconductive plastic cases

providing no shielding. Essentially, all these

devices cannot meet these mandatory

requirements or they may cause interference

to other devices or receive interference

causing susceptibility problems without a

proper program of EMI control. This program consists of identifying the “suspect” components

Figure 2.1: Cables with the electromagnetic shielding to separate the wires from outside environments

http://www.wisegeek.com/what-are-radio-waves.htm

http://www.wisegeek.com/what-are-radio-waves.htm

http://www.wisegeek.org/what-is-electromagnetic-radiation.htm



and circuits that may cause or be susceptible to EMI. This is completed early on in the program

to allow for an efficient design in keeping the cost of dealing with EMI as low as possible. A

complete EMC program consists of proper filtering, grounding and shielding.

2.2.4 Materials Used for Shielding

A variety of materials can be used as electromagnetic

shielding to protect an electrical device. Examples

include ionized gas in the form of plasma, metal foam

with gas-filled pores, or simply sheet metal. In order

for holes within the shielding to be present, they must

be considerably smaller than any wavelength from

the electromagnetic field. If the shielding contains

any openings larger than the wavelength, it cannot

effectively prevent the device from becoming

compromised.

Household devices often use a different shielding

method due to the likelihood of exposure to

electromagnetic fields. Plastic enclosures usually

use some sort of metallic ink consisting of copper or

nickel in a small particular state. This material can

be sprayed onto the enclosure, producing a

conductive layer of metal that acts as protection.

The main reason this layer works is due to its close

proximity to the grounding of the device.

Many common day-to-day items contain electromagnetic shielding. One of the most common

examples of this is the microwave oven found within most kitchens. With the metal housing

working in unison with the screen on the window, a Faraday cage is created. While some visible

light is able to pass through the window screen, waves of other frequencies cannot.

Figure 2.2: Sheet metal used to make electromagnetic shielding

Figure 2.3: Microwave oven having electromagnetic shielding.

http://www.wisegeek.com/what-is-sheet-metal.htm

http://www.wisegeek.com/what-is-a-wavelength.htm

http://www.wisegeek.com/what-is-an-electromagnetic-field.htm

http://www.wisegeek.org/what-is-copper.htm

http://www.wisegeek.org/what-is-a-faraday-cage.htm

http://www.wisegeek.org/what-is-visible-light.htm

http://www.wisegeek.org/what-is-visible-light.htm



2.3 Shielding Theory

There are two ways of approaching the theory of shielding. These are by the use of circuit

theory and by the use of field theory. The EMC industry uses a field theory approach to

shielding theory using abstract mathematical modeling techniques to yield a value of merit

classified as "shielding effectiveness". Shielding effectiveness is then used as a measurement

to gauge the attenuation of an EM field through shielding barrier material.

The problem with the use of shielding effectiveness is that there can be a significant differential

between the attenuation of the electric E fields, magnetic H fields, and power, where the

difference can exceed 100 dB. The actual difference will vary as a function of variables

associated with specific applications where the literature on the shielding of radiated EM fields

does not address these conditions. The result is a significant confusion factor in the selection

of shielding barrier material, facing design engineers who are required to meet EMC radiated

emission and susceptibility requirements.

The circuit theory approach (included herein) employs mathematical modeling techniques

consistent with college course work and yields a predicted field strength at any given distance

from the shielding barrier material. The results can also be used to predict the shielding of a

seam or gasketed joint in the barrier material (or enclosure). The circuit theory approach given

below examines the field as it penetrates a barrier and yields a value of the field as it exits the

barrier.

2.3.1 EM Field Generation and Shielding

A radiated electromagnetic (EM) force field is generated by the action of driving a current

through a wire. An example is shown in Figure 2.4.

The wire (or PC card trace) acts as a transmitting antenna

as an emitter of EM interference and as a receptor with

regard to EM susceptibility. A common method of

reducing (or eliminating) the possibility of the PC trace

being an emitter or receptor is by the use of a shielding

barrier.

When an EM force field is impinged on a metallic

(conductive) shielding barrier, currents are caused to

flow in the barrier. As the field penetrates the barrier, the

current is attenuated (i.e., reduced in amplitude as illustrated in Figure 2.5) by a force called

skin effect.

The power of the field as it leaves the barrier is approximately equal to the current squared

times the impedance of the barrier, and is in watts per meter squared.

Figure 2.4: Current through a receiver circuit on a PC card.



As we learned above, currents flow

in the shielding barrier as a function

of the radiated field being impinged

on the barrier. When the current

crosses a seam in the barrier (created

by maintenance covers, etc.), a

voltage is created across the seam,

where the value of the voltage is

equal to the current times the

impedance of the seam. The seam

then becomes a radiating antenna

where the impedance and pattern is

similar to that of a slot antenna. EMI gaskets are used to reduce the impedance of the seam and

subsequent power radiating from the seam.

2.3.2 Generation and Propagation of EM Fields

The undergraduate courses on EM theory introduce the concept of an EM field by driving a

pair of parallel plates with an AC voltage source as illustrated in Figure 26. The current that

flows through the wire comes from the top plate and

is stored in the bottom plate. The over-presence of the

electrons on the bottom plate is illustrated by + and

the absence of electrons on the top plate and is

illustrated by +. This creates an electromagnetic field

which is illustrated in Figure 4. The field consisting

of the straight lines is classified as a displacement

field and is in amperes per meter squared. The

magnitude of the E field is equal to the voltage

differential between the plates divided by the distance

between the plates in meters. The resultant E field is

in volts/meter

As is illustrated in Figure 2.7, the lines of flux in

the center of the plates are straight and flow from

the bottom to the top plate. At the edges they bow

out, where the fields or lines of flux repel each

other, forcing the bowing. The field that bows out

represents a radiated EM field. The radiated EM

field emanating from the trace of Figure 1 is similar

to the radiated EM field illustrated in Figure 4. The

electric "E" field is tangent to the lines of force as

illustrated in Figure 2.7. The magnetic "H" field is a

Figure 2.5: Field penetrating metallic shield barrier causing current to get attenuated.

Figure 2.6: Applying AC voltage source into a pair of parallel plates

Figure 2.7: Lines of flux with respect to E and H field.



field perpendicular to the lines of force and points out of the paper.

The set of plates as illustrated in Figure 2.7 produce a field similar to that of the PC card trace

of Figure 2.4 (and of an electric dipole antenna). If the transmitted power is known, the field

strength can be calculated using the dipole antenna equation, i.e.,

PR ≈ 1.6Pt / 4𝜋𝑅2

Where, PR = Field strength at distance R (𝜔/𝑚2)

Pt = Transmitted power (𝜔/2𝑚2)

R = Distance from radiating source (m)

And power equation (pointing vector):

𝐸 x H = PR

E / H = 377λ/ 2𝜋R R <λ/2𝜋 (ohms)

= 377 R ≥λ/2𝜋

And, λ = 3x108/f (m)

If the power is not known, the value of the electric field can be approximated using the

following equation:

E ≈ 𝑒/𝜋𝑅

E = Electric field strength (v/m)

e = Voltage across plates

H ≈ 2𝜋𝑅𝐸/377𝜆 R <λ/ 2𝜋(A / m)

= E / 377 R ≥ 𝜆/ 2𝜋



2.4 Introduction to Shielding

2.4.1 Suppression (Shielding) of EM Fields

When we place a shielding barrier in the path of the EM field, the force of the field causes

current to flow in the barrier. As is illustrated in Figure 2.8, the excess electrons in the bottom

plate create a force on the electrons to flow away from the point of contact. In a similar manner,

the lack of electrons on the upper plate will create an excess of electrons on the barrier at the

upper point of contact. This current flow in the barrier is called the “surface current density”

(Js) in amperes/meter, and is approximately equal to twice the H field incident on the barrier

when the field is perpendicular to the barrier. The current flowing in the barrier is attenuated

by the skin effect.

The current on the transmitted side is equal to Jsi 𝑒−𝑑/𝛿 (i.e, the current on the incident side

attenuated by skin effect). The impedance of the field emanating from the barrier is equal to

the impedance of the barrier. The values of Et and Ht are as illustrated in Figure5 and are as

follows.

Figure 2.8: Placing a shielding barrier in the path of the EM field.



2.4.2 Field Strength Through Shield

From antenna theory we know that the

power from an antenna is reduced as the

square of the distance from its source.

Shielding theory purposes that the field as

it passes through a barrier is attenuated but

not changed with regard to direction. As

such, the loss of power is a function of the

distance from the original source of the

field as illustrated in Figure 2.9.

2.4.3 Shielding and Gasketing

If the source contains a large current flow compared to its potential, such as may be generated

by a loop, a transformer, or power lines, it is called a current, magnetic, or low impedance

source.

The latter definition is derived from the fact that the ratio of E to H has a small value conversely,

if the source operates at high voltage, and only a small amount of current flows, the source

impedance is said to be high, and the wave is commonly referred to as an electric field. At very

large distances from the source, the ratio of E to H is equal for either wave regardless of its

origination. When this occurs, the wave is said to be a plane wave, and the wave impedance is

equal to 377 ohms, which is the intrinsic impedance of free space. Beyond this point all waves

essentially lose their curvature, and the surface

containing the two components becomes a plane

instead of a section of a sphere in the case of a point

source of radiation.

If the gasket is made of a material identical to the walls

of the shielded enclosure, the current distribution in

the gasket will also be the same assuming it could

perfectly fill the slot. (This is not possible due to

mechanical considerations.) The flow of current

through a shield including a gasket interface is

illustrated in Figure 2.10. Electromagnetic leakage

through the seam can occur in two ways. First, the

energy can leak through the material directly. The

gasket material shown in Figure 2.10 is assumed to have

lower conductivity than the material in the shield. The

rate of current decay, therefore, is also less in the gasket.

It is apparent that more current will appear on the far side of the shield.

Figure 2.9: Field penetrating barrier

Figure 2.10: The flow of current through a shield including a gasket interface



This increased flow causes a larger leakage field to appear on the far side of the shield. Second,

leakage can occur at the interface between the gasket and the shield.

If an air gap exists in the seam, the flow of current will be diverted to those points or areas

which are in contact. A change in the direction of the flow of current alters the current

distribution in the shield as well as in the gasket. A high resistance joint does not behave much

differently than open seams. It simply alters the distribution of current somewhat. A current

distribution for a typical seam is shown in Figure 210. Lines of constant current flow spaced at

larger intervals indicate less flow of current. It is important in gasket design to make the

electrical properties of the gasket as similar to the shield as possible, maintain a high degree of

electrical conductivity at the interface, and avoid air, or high resistance gaps.

RF Gaskets

A gasket is a mechanical seal which fills the space between two or more mating surfaces,

generally to prevent leakage from or into the joined objects while under compression.

Although there are hundreds of gasket varieties based upon geometry and materials, there are

four principle categories of shielding gaskets: beryllium copper and other metal spring fingers,

knitted wire mesh, conductive particle filled elastomers and conductive fabric-over-foam. Each

of these materials has distinct advantages and disadvantages, depending upon the application.

Regardless of the gasket type, the important factors to be considered when choosing a gasket

are RF impedance (R + jX, where R = resistance, jX = inductive reactance), shielding

effectiveness, material compatibility corrosion control, compression forces, compressibility,

compression range, compression set, and environmental sealing.

Below is a comprehensive list of selection factors.

http://en.wikipedia.org/wiki/Seal_%28mechanical%29

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Gasketed joint shielding

When a radiated EM force field is impinged on a metallic shielding barrier, a current (surface

current density in amperes per meter) is generated in the material. When the current flows

across a gasketed maintenance cover as illustrated in Figure 2.11, a voltage e is generated

across the gasket. The value of e is equal to the current in amperes/meter times the impedance

of the joint (transfer impedance in ohm-meters).

The EM force field illustrated in Figure 2.11 is generated by the voltage across the gap and has

the characteristics of a low impedance slot antenna.

The radiated field can be estimated from the example of Figure 2.11 as follows:

ET ≈ 2e / R = 2JS ZT /R (V/ m)

HT ≈ ET λ / 2 R (377) R< λ / 2 (A/ m)

HT = ET / 377 R > λ/2

Figure 2.11: The current flowing across a gasketed maintenance cover

JS = Current due to Field Striking Barrier

e = Voltage across Gasket = JS ZT

ZT = Transfer Impedance of Gasketed Joint (ohm-m)

CHAPTER – 3

Introduction to Frequency Selective Surface (FSS)


15

Introduction to 3__

Frequency Selective Surface (FSS)

FSS is a periodic structure of conductive elements or apertures in either one or two dimensions

that provide a filter operation when they are illuminated with EM wave. When illuminated by

an electromagnetic wave, FSS exhibits total transmission/reflection around the resonance

frequency. This spatial filter behavior of FSS is used in designing the FSS shield. The filter

behavior (low-pass, high-pass, band-pass and band-stop) of the FSS depends on the shape of

the element.

Figure 3.1: FSS types and response, (a) solid patch array — low pass, (b) slot array — high pass,

(c) patch looped array — band stop and (d) slot looped array — band pass.

A frequency-selective surface (FSS) is any thin, repetitive surface (such as the screen on a

microwave oven) designed to reflect, transmit or absorb electromagnetic fields based on

frequency. In this sense, an FSS is a type of optical filter or metal-mesh optical filters in which

the filtering is accomplished by virtue of the regular, periodic (usually metallic, but sometimes

dielectric) pattern on the surface of the FSS. Frequency-selective surfaces have been most

commonly used in the radio frequency region of the electromagnetic spectrum and find use in

applications as diverse as the aforementioned microwave oven, antenna radomes and

modern metamaterials. Sometimes frequency selective surfaces are referred to simply as

periodic surfaces and are a 2-dimensional analog of the new periodic volumes known as

photonic crystals.

http://en.wikipedia.org/wiki/Optical_filter

http://en.wikipedia.org/wiki/Metal-mesh_optical_filter

http://en.wikipedia.org/wiki/Microwave_oven

http://en.wikipedia.org/wiki/Antenna_(radio)

http://en.wikipedia.org/wiki/Radome

http://en.wikipedia.org/wiki/Metamaterial

http://en.wikipedia.org/wiki/Photonic_crystals

Introduction to Frequency Selective Surface (FSS) [16]


3.1 Background

Filters play a fundamental role in almost electronic or RF circuit. Once being incorporated into

a design, the filter acts as a device that controls the frequency content of the signal for

mitigating noise and unwanted interference. Filters are categorized, based on their function,

into three major groups: lowpass, bandpass, and highpass filter. A lowpass filter, for example,

allows for the lower frequencies to pass through the circuitry and blocks higher frequencies.

Frequency-selective surface (or dichroic) structures to space waves are the counterparts of filter

in transmission lines.

Once exposed to the electromagnetic radiation, a frequency- selective surface (FSS) acts like

a spatial filter; some frequency bands are transmitted and some are reflected. In a way, an FSS

can be a cover for hiding communication facilities.

This is probably the first potential application of FSS structures, as they have actually been

used as covers named radomes. Radomes are bandpass FSS filter that are used to reduce the

radar cross-section (RCS) of an antenna system outside its frequency band of operation.

Fig. 3.1 shows the radomes at the Cryptologic Operation Center in Japan.

Since the early 1960's, because of potential military applications, FSS structures have been the

subject of intensive study. Marconi and Franklin, however, are believed, to be the early

pioneers in this area for their contribution of a parabolic reflector made using half-wavelength

wire sections in 1919. FSSs as frequency-selective materials have been used traditionally in

stealth technology for reducing the RCS of communications systems.

The concept of stealth or being able to operate without the knowledge of the enemy has always

been a goal of military technology. In order to minimize the detection, FSS layers cover the

facilities to reduce the RCS.

Figure 3.2: Radomes at the Cryptologic Operations Center, Misawa, Japan (photo courtesy of en. Wikipedia)



FSSs most commonly take the form of planar, periodic metal-dielectric arrays in two-

dimensional space. Frequency behavior of an FSS is entirely determined by the geometry of

the surface in one period (unit cell) provided that the surface size is infinite. A periodic array

of patch elements is shown in Fig. 3.2. This array is shown to have a capacitive frequency

characteristic.

Although taking different shapes, conventional FSSs have similar operation mechanisms that

can be explained by the phenomenon of resonance. Consider an array of elements on a planar

surface. Upon contact with a plane-wave, the elements of the periodic surface resonate at

frequencies where the effective length of the elements is a multiple of the resonance length.

Corresponding to the phase front of the wave, these elements have a certain phase delay. As a

result, the scattered radiations of individual elements add up coherently. An example of such

arrangement of elements is Marconi and Franklin's reflector. This reflector is very much similar

to the most famous FSS design, an array of half-wave dipoles. A large reflector antenna

constructed using wire-grids is shown in Fig. 3.2.

The resonance characteristics of a resonance-length based FSS usually depend on the way the

surface is exposed to the electromagnetic wave. This includes the effective aperture size of the

FSS and the incidence angle of the wave. The dependence of the FSS frequency response with

respect to these factors could be a major drawback for some applications. As a result, over the

years new ideas have been sought to overcome the issue of the dependence on the size and

angle. Besides the two major problems mentioned above, harmonics are another effect that

influence the performance of an FSS. The situation becomes more involved given that the

harmonics of the intended frequency are themselves dependent upon the incidence angle.

3.2 History

Historically, the first approach to solving for fields reflected and transmitted by FSS was the

spectral domain method (SDM), and it's still a valuable tool even today. The spectral domain

method is known at Ohio State University as the periodic method of moments (PMM). The

SDM starts out with an assumed Floquet/Fourier series solution for all fields, currents and

potentials whereas the PMM starts out with a single scatterer, then adds in all of the scatterers

Figure 3.3: Two-dimensional periodic array of patch elements



in the infinite plane (in the spatial domain), then uses a transformation to yield the spectral

domain representation of the fields. Both approaches are effectively the same approach, in the

sense that they both assume an infinite planar structure which gives rise to a discrete Fourier

series representation for the fields.

3.3 Advantages and disadvantages

The spectral domain method has one very important advantage over other – strictly numerical

- solutions to Maxwell's equations for FSS. And that is that it yields a matrix equation of very

small dimensionality, so it is amenable to solution on virtually any type of computer. The

dimension of the matrix is determined by the number of current basis functions on each

individual scatterer and can be as small as 1×1 for a dipole at or below resonance. The matrix

elements however take longer to compute than with volumetric approaches such as FEM.

Volumetric approaches require that a volume surrounding the unit cell be gridded accurately,

and can require many thousands of elements for an accurate solution, though the matrices are

usually sparse.

3.4 Floquet's principle

The spectral domain method is based on Floquet's principle, which says that if an infinite,

planar, periodic structure is illuminated by an infinite plane wave, then every unit cell in the

periodic plane will contain exactly the same currents and fields, except for a phase shift,

corresponding to the incident field phase. This principle allows all currents, fields and

potentials to be written in terms of a modified Fourier series, which consists of an ordinary

Fourier series multiplied by the incident field phase. If the periodic plane occupies the x-y

plane, then the Fourier series is a 2-dimensional Fourier series in x, y.

3.5 Field equations for 2D PEC frequency selective surfaces

Perfectly electrically conducting (PEC) periodic surfaces are not only the most common but

also the easiest to understand mathematically, as they admit only electric current sources J.

This section presents the spectral domain method for analyzing a free-standing (no substrate)

PEC FSS. The electric field E is related to the vector magnetic potential A via the well-known

relation:

------------------ (3.1)



And the vector magnetic potential is in turn related to the source currents via:

Where,

3.6 Plane wave expansion of the fields in source-free media

Frequency-selective surfaces are frequently stratified in the direction normal to the plane of the

surface. That is, all dielectrics are stratified and all metallic conductors are considered stratified

as well, and they will be regarded as perfectly planar. As a result, we are excluding metallic

vias (wires perpendicular to the plane of the FSS) which could potentially connect currents

from different strata of the FSS structure. With this type of a stratified structure in mind, we

can then use a plane wave expansion for the fields in and around the FSS, since plane waves

are the eigenfunction solution to the vector wave equations in source-free media.

To solve equations (3.1) and (3.2) for a free-standing, doubly-periodic surface, we consider an

infinite 2D periodic surface occupying the entire x-y plane, and assume a discrete plane wave

expansion for all currents, fields and potentials .

where for mathematical simplicity, we assume a rectangular lattice in which α only depends

on m and β only depends on n. In the equations above,

and,

where lx, ly are the dimensions of the unit cell in the x,y directions respectively, λ is the free

space wavelength and θ0, φ0 are the directions of an assumed incident plane wave, with the

FSS regarded as lying in the x-y plane. In (3.5c), the root is taken which has a positive real part

and non-positive imaginary part).

------------------ (3.2)

------------------ (3.3)

------------------ (3.4a)

------------------ (3.4c)

------------------ (3.4b)

------------------ (3.5a)

------------------ (3.5c)

------------------ (3.5b)

------------------ (3.6)



3.7 Elements of Design in Traditional FSS

As mentioned previously, FSS structures are periodic arrays of special elements printed on a

substrate. For numerical analysis, such arrays are assumed to be infinite in dimension as FSSs

usually consist of many elements. The infinite array approximation reduces the whole problem

of analysis to calculating the frequency response of a single element in the array given the

periodic nature of the FSSs. A brief overview of the available FSS elements is provided in this

section.

3.7.1 Element Geometries

In general, the FSS structures can be categorized into two major groups: patch-type elements

and aperture-type elements. As an introduction to FSS structures two complementary planar

arrays, array of patches and array of slots, are usually considered. A simple structure consisting

of periodic array of metallic patches (Fig. 3.3) has been shown to have a low-pass characteristic.

Once hit by a plane-wave, this surface transmits low-frequency content of the wave and reflects

the higher frequencies. Another observation is to consider such an array as a capacitive surface

given its frequency response provided in Fig. 3.3. The complementary structure (see Fig. 3.3)

has an inductive response, hence acting as a highpass filter.

Figure 3.4: Periodic structures comprising of complimentary elements, patches and slots (wire-grid), and their surface impedance- The patch-array produces a capacitive response, whereas the array of slots is inductive



As it will be discussed later, inductive and capacitive surfaces can be put together to produce a

desired filter response. Over the years, a variety of FSS elements were introduced for bandpass

and band-stop applications.

Depending on the application requirements, different FSS designs can be chosen to fulfill the

demands. These requirements usually include level of dependence on the incidence angle of

the incoming wave; level of cross-polarization; bandwidth; and level of band separation.

A comparison between some of the most famous FSS designs is provided in Table. 3.1 based

on the above criteria. As shown, for instance, the dipole array is very sensitive to the angle.

3.7.2 Element Dimensions

As mentioned above, FSSs are traditionally designed based on the resonant elements. A planar

array of strip dipoles, for example, produces a frequency response consisting of multiple

notches at frequencies where the length of the dipoles is a multiple of half a wavelength.

A similar effect can explain the operation of other elements. The square loop element, for

example, can be imagined as two dipoles that are connected to one another at each end.

Using the same argument as that of the dipole, a loop resonates when the length of the two

sides equals the length of a resonant dipole, λ/2. In other words, each side of the loop is about

λ/4. Although the shape of the elements has the utmost importance effect in the frequency

response, the way these elements are arranged in the array format is also part of the design

work. Moreover, the response also depends on the characteristics of the substrate used. This in

Figure 3.5: A variety of FSS elements over past decades



fact becomes very important as we will present a square loop element whose sides are as small

as ¸ λ /12. This is where the miniaturized-element FSS design comes to picture.

Element Angle

independence

Cross-

polarization

level

Larger

bandwidth

Smaller band

separation

Loaded dipole 1 2 1 1

Jerusalem cross 2 3 2 2

Rings 1 2 1 1

Tripole 3 3 3 2

Crossed dipole 3 3 3 3

Square loop 1 1 1 1

Dipole 4 1 4 1

Table 3.1: Comparison Between the Performance of the Common FSS Elements

This is where the miniaturized-element FSS design comes to picture. Elements that are much

smaller than the wavelength are designed to create capacitive gaps and inductive traces. By

thinning and miniaturizing the unit cell, capacitive junctions in the form of shunt or series

capacitors are achieved. Inductive traces are also held very close to one another to produce a

larger inductive effect as a result of mutual magnetic coupling.



3.8 Traditional Frequency-Selective Surfaces: Design,

Characterization, and Applications

Periodic structures, or arrangements of equally spaced, identical elements, have been of interest

in many areas of physics and engineering. Although being very old from mathematical stand

view, periodic structures have also a long history of development from more practical

perspectives. In fact, these arrangements were considered in an engineering design problem

over 200 years ago by the physicist David Rittenhouse. He invented a grating with equally-

spaced hairs. A periodic structure can be optimized for a particular application which requires

certain characteristics. This process includes designing the elements as well as the way they

are placed in the periodic array.

3.8.1 Brief Design Concept

A frequency-selective surface (FSS) is a periodic, planar assembly of generally metallic

elements on a dielectric layer. It is built in conjunction with the electromagnetic waves in order

to \tailor" an electromagnetic link in the free-space environment. Acting as a barrier for the

waves propagating along the link, the FSS controls the flow of the electromagnetic energy. The

transfer function of the FSS manipulates the spectral content of the wave. As a result, some of

the frequency constituents of the wave are blocked, and some pass through the

FSS fence. In another perspective, an FSS is analogous to a filter in circuit theory. For their

filtering effects, FSS structures are also called spatial filters in electromagnetic engineering.

There are many examples in real world which clearly confirm the importance of having

knowledge about periodic structures. Consider a reflective surface which is to be built by least

amount of metallization at a given frequency. Intuitively, the first observation is that the less

the metal is, the weaker the reflection becomes. Now, consider two possibilities: using an array

of long strips or an array of short dipoles. Having a larger metalized area, the long strip is

expected to produce a stronger reflection, according to our very first observation.

Rigorous analysis of the two candidates, however, reveals that the dipole array can actually

produce a total reflectivity at a certain frequency, whereas the array of strips never becomes

totally reflective. This problem, which was studied by Marconi and Franklin when they

proposed their reflector antenna, can be explained through modeling the two arrays using

circuit theory. This is shown in Fig. 3.5 where the model for the array of long strips is just an

inductor, while the array of dipoles is an L-C circuit. The dipole array, therefore, becomes total

reflective at the frequency of resonance of the L-C circuit.



Following the brief discussions highlighted in Chapter 1, this chapter overviews some of the

past, famous FSS structures,

including the physical interpretation

of their operation.

The overview section is followed by

a discussion on the methods of

characterization and modeling of

FSS. Finally, later in this chapter, a

more detailed overview on the

applications of FSS is presented.

3.8.2 Overview of the Elements of Traditional FSS

The operation mechanism of traditional FSSs, as mentioned previously, is based on the

resonant elements. Simply, the idea is that a plane-wave illuminates an array of metallic

elements, thus exciting electric current on the elements. The amplitude of the generated current

depends on the strength of the coupling of energy between the wave and the elements.

The coupling reaches its highest level at the fundamental frequency where the length of

elements is a λ/2. As a result, the elements are shaped so that they are resonant near the

frequency of operation. Depending on its distribution, the current itself acts as an

electromagnetic source, thus producing a scattered field. The scattered field added to the

incident field constitutes the total field in the space surrounding the FSS. By controlling the

scattered field (designing elements), therefore, the required filter response is produced which

can be seen in the spectrum of the total field. As mentioned above, the distribution of the current

on the elements determines the frequency behavior of the FSS. The current itself depends on

the shape of the elements.

Given the dependence of the traditional FSS on the length, excitation of the higher-order

modes, in addition to the first fundamental mode, becomes inevitable. As a result, the frequency

response of the traditional FSS usually has a high harmonic content. The first and the second

possible modes are shown in Fig. 3.6. The issue of harmonics not only affects the frequency

characteristics of the FSS but also degrades its scan performance because some of the

Figure 3.6: A periodic array of infinitely long metallic strips (top) produces an inductive characteristic, whereas an infinite array of closely-spaced dipoles (bottom) produces a notch frequency behavior which in turn results in a total reflectivity at the resonance frequency of the notch. In this way, although the dipole array uses less metallic area, it produces a complete reflective state



harmonics may be excited only when the incidence angle changes from normal to the FSS

plane.

Although the geometry of the elements has a critical influence on the filtering behavior, there

are other parameters that can affect the frequency response. This could be the choice of the

parameters of the substrate supporting the elements of the FSS and also its inter- element

spacing. The substrate is shown to affect both the frequency of operation and the bandwidth of

the response. The spacing between the elements, on the other hand, points out the issue of the

grating lobe which is inherent in any array radiator; the larger the spacing, the earlier the onset

of the grating lobes. As a result, a smaller inter-element spacing is usually preferred. However,

the spacing can also change the bandwidth; a larger spacing in general produces a narrower

bandwidth.

To develop a better understanding of operation of FSS structures, a comparative study over

different types of FSS is provided here. But, first, we need to somehow categorize FSS

structures. There are a number of constraints based upon which FSSs are classified. Here, we

use Munk's approach in classifying frequency-selective surfaces which is based on the shape

of the elements.

In Fig. 3.7, four major categories of FSS arrangements are demonstrated which are:

The center connected or N-poles, such as dipole, three-legged element, the Jerusalem

cross, and the square spiral.

The loop types such as the three and four-legged loaded elements, the circular loops,

and the square and hexagonal loops.

Solid interiors or plate types.

Combinations.

Figure 3.7: The first and the second resonant modes-(top) shows the fundamental mode which is excited for any element shape irrespective as the incidence angle. (Bottom) shows the first odd mode at about 2ff which may be excited only at oblique incidence. The frequency of this mode may change slightly depending on the element shape



Each class along with simulated examples are presented next. All the simulation results belong

to the book by Munk on FSS, and use a substrate with dielectric constant of ϵr= 2.2 with the

thickness of 0.5 mm. The examples were intended for operation at 10 GHz.

The simulated results show the performance for both TE and TM polarizations at 0° and

45°.

Figure 3.8: Typical FSS Elements classified in four major groups based on their shapes

Our design constructions includes group II shape structures, and here is the detail description

about those shapes.

Group I: The center connected or N-poles

Group II: The loop types

Group III: Solid interiors or plate types

Group IV: Combinations



3.8.3 Group II- loop type structures

I. Three-legged Loaded Element:

A surface of three-legged loaded elements is shown in Figs. 3.8 to 3.10. It was developed as a

direct consequence of the four-legged loaded element, although it is not as easy to explain.

However, the fact remains that both of these elements resonate when their circumferences are

approximately one full wavelength, and they show a load null at the frequency where the legs

are approximately h/4 long.

The bandwidth variation with polarization is suited quite well for use in a hybrid radome (better

than the three-legged unloaded element). It is considerably more broadbanded than the

Cm

Cm

Frequency (GHz)

Ref

lect

ion

(d

B)

Dx=0 .3457 cm

Dz=0 .3792cm

Monopole Length = 0.2784 cm

Monopole Width = 12 cm

Wire Width = 0.0171 cm

Surface profile-

Ɛr= 2.2 D=20mil

Figure 3.9: Reflection coefficient curve for closely packed FSS of three-legged loaded elements. Angle of incidence equals 45", orthogonal and parallel polarization. Note the load null at about 19 GHz for both polarizations.

Ɛr= 2.2 D=20mil



Four-legged cases, the primary reason being that the inter-element spacings Dx,Dz are

considerably smaller; in other words, an increase in Dx,Dz could reduce the bandwidth

considerably. We finally show in Fig. 3.9 the reflection curves of the same three-legged loaded

element in the larger frequency range 0 to 40 GHz. Here the plane of incidence is along the

vertical leg, α = 90°. It is worth mentioning that we have also run the reflection curves in the

horizontal plane (α = 0°).

However, since these two curves are practically indistinguishable, we do not show the latter

case.

Ref

lect

ion

(d

B)

Frequency (GHz)

Cm

Cm

Dx=0 .3457 cm

Dz=0 .3792cm




Surface profile-

Ɛr= 2.2 D=20mil

Ɛr= 2.2 D=20mil

Figure 3.10: Same as in Fig. 3.7 but frequency range 0 to 40 GHz.



R

efle

ctio

n (

dB

)

Frequency (GHz)

Cm

Cm

Dx=0 .3457 cm

Dz=0 .3792cm




Surface profile-

Ɛr= 2.2 D=20mil

Ɛr= 2.2 D=20mil

Figure 3.11: Same FSS as in Fig. 3.7 but normal angle of incidence and frequency range 0 to 40 GHz.

Angles of incidence and parallel polarization. Inspection of Fig. 3.8 shows this to happen at 25

GHz, while the second even resonance is now being pushed up to 40 GHz (as usual with a null

between). Finally note that grating lobes are not excited before around 40 GHz for 45"

incidence angles. If this feature is important, the three-legged loaded element is hard to beat (a

competitor would be a Gangbuster of higher order and an interlaced four-legged loaded

element).



II. Hexagon Element

The physical appearance of the hexagon array is shown in Fig. 3.11 for close inter-element

spacing and in Fig. 3.12 for wider spacing, where we also show the reflection curves at 45°

angles of incidence for orthogonal and parallel polarization, respectively. Note that the

frequency range in both cases is 0 to 40 GHz. not just 0 to 20 GHz. This is done to fully show

the superior behavior of the hexagon array, namely the location of the first modal

Interaction null at about 29 GHz; this is about twice as high as anything seen earlier with the

exception of the square spiral element. Also the difference in inter-element spacing is seen to

lead to a rather significant change of bandwidth and also a difference in element circumference:

1.5 cm compared to 1.98 cm.

Note how the voltage distribution for the fundamental mode leads to a strong voltage difference

across the “capacitor,” that is, the fundamental frequency will be greatly pulled downward.

However, inspection of the voltage distribution at the second harmonic shows no voltage

difference across the capacitor, that is, no downward pulling of that frequency. This explains

First Even Resonance ~ 11 GHz

(a)

Second Even Resonance ~ 30-40 GHz

(b)

First Odd Resonance ~ 25 GHz

(c)

Figure 3.12: Current distribution on a three-legged loaded element at the (a) first even resonance at about 11 GHz; (b) second even resonance from about 30 to 40 GHz; (c) first odd resonance at about 25 GHz.



how the second resonance of the hexagon element may be approximately three times the

fundamental.

It should be noted that other loop elements act similar, but none are superior to the hexagon.

For example, a circular loop has the second resonance about twice the fundamental, unless the

elements are almost touching each other. One may finally wonder: Why is it that the end

Ref

lect

ion

(d

B)

Frequency (GHz)

Cm

Cm

Dx=0 .3457 cm

Dz=0 .3792cm


Monopole Width = 0.06 cm


Surface profile-

Ɛr= 2.2 D=20mil

Ɛr= 2.2 D=20mil

Figure 3.13: By reducing the transmission line spacing we obtain a significant reduction in bandwidth compared to Fig. 3.8. With the same Dx and Dz, we preserve the same onset frequency of the grating lobes.



capacity of the anchor element does not similarly produce a large difference in frequency

pulling of the fundamental and the second harmonic as we just saw for the hexagon element

(see Figs. 3.8 and 3.9) The answer to this question is obtained by inspection of the current and

voltage modes for the anchor element. Note that a significant voltage difference between the

element tips and the center of the adjacent elements are present for both the first and second

harmonics; that is, they are both pulled significantly downward.

Frequency (GHz)

Ref

lect

ion

(d

B)

Cm

Cm

Dx=0 .4171 cm

Dz=0 .4751cm

Gap= .025cm

Hexagon perimeter= 1.5cm

Line Width = 0.0171 cm

Surface profile-

Ɛr= 2.2 D=20mil

Ɛr= 2.2 D=20mil

Figure 3.14: Reflection coefficient curves for an FSS of closely packed hexagon elements. Angle of incidence equals 45", orthogonal and parallel polarization. Note that the location of first null is unusually high (around 28 GHz).



3.8.4 Most Significant Traditional Applications of FSS

Common electromagnetic design problems that can benefit from frequency-selective surfaces

is presented here. As briefly discussed in Chapter 1, conventional applications include radomes

(bandpass spatial filters) and multi-frequency reflector antennas. Because of their frequency-

dependent behavior, FSSs have also been looked at as reactive surfaces and been used in a

variety of beam forming applications as well as high-performance antenna designs.

In this section, some of the current applications are presented.

I. Radomes:

Frequency-selective surfaces have been employed primarily in design of special covers called

radomes that are capable of reducing the radar cross-section (RCS) of antennas under cover. In

this application, FSS is transparent to a desired frequency band and reflecting at other

frequencies.

In practice, bandpass, spatial covers of this type are used in certain applications, for example,

in the case of an aircraft. A typical assembly is shown in Fig. 3.13. As shown, the radar

(antenna) mounted at the front tip of the airplane is covered by a shaped radome. Radome has

two modes of operation: In the transparent mode the signal passes through the radome and is

collected by the antenna. In the reflecting mode, however, the radome behaves like a metallic

surface which reflects the signal in the specular direction. The advantage of this method is that

the radome can be shaped particularly in order to reach the lowest levels of RCS. This can be

done by designing the shape of the radome so that the direction of the strong reflection

(specular) in reflecting mode is out of the sight of the transmitter of the signal, i.e. it is not

directed toward the incoming signal.

Figure 3.15: Application of frequency-selective surfaces as radome covers in the aircraft technology for reducing the antenna RCS.



II. Multi-Frequency Reflectors

As mentioned earlier, frequency-selective surfaces have been utilized in multiband reflector

antenna applications. A common approach uses an FSS as a subreflector in addition to the large

reflecting dish in a reflector antenna, as shown in Fig. 3.14. The subreflector, which is an FSS,

is designed to be reflective at a frequency band and be simultaneously transparent at another

desired band. The frequency-dependent subreflector allows for application of multiple feeds in

this systems, thus enabling the multi- frequency state of the antenna. Different frequency feeds

are optimized and positioned at the real and virtual foci of the subreflector. Hence, only one

main reflector can act as a multiband antenna.

A practical application of this idea is the high-gain antenna (HGA) of the Voyager space- craft

which was designed, to diplex S and X bands. An FSS forms the subreflector which is reflecting

at X-band whereas is transmitting at S-band. In this antenna, the S-band feed is placed at the

prime focus of the reflector, whereas the X-band feed is located at the Cassegrain focal point

(see Fig. 3.14). As a result, loaded with an FSS, a single reflector acts as a dual-band antenna,

thus reducing the overall mass, volume, and most importantly the fabrication cost. A similar

approach was proposed later by Wu to build a four-band reflector system. In this work, a four-

frequency FSS was utilized as the subreflector.

III. Beam Control Arrays

Frequency-selective surfaces have also been used in creation of impedance surfaces with tuning

capability. This problem is believed, that first appeared in where scattering properties of a

corrugated surface loaded with microwave solid-state amplifiers were studied. In this surface,

connecting each slot region to an amplifier, the authors have been able to not only control the

phase of the reflected wave but also to manipulate its power gain. It was envisioned that this

loaded array would be able to change the phase front (shape) and intensity of an incident plane-

wave.

Figure 3.16: Dual-frequency reflector antenna using an FSS as the subreflector.



FSSs considered for the beam-shaping application are commonly of the type series resonators,

L-C. A possible way for constructing an L-C resonating surface is to use an array of metallic

strips that are cut at periodic intervals in order to create capacitive junctions along the strips.

This array, which is simply an array of small strips, is shown in

Fig. 3.15. The basic idea can be explained using a quasi-optical approach in which the beam

is assumed to be a plane-wave over an infinitely long, periodic array in two dimensions on the

surface.

The metallic traces are inductive which together with the gap capacitances create a series

combination of inductors with capacitors. By mounting lumped reactive elements such as

varactors diodes, one can tune the resonance characteristics of the array. It should be pointed

out that this is only an approximate method since if the elements of the array are long, the wave

variations along the elements must be taken into account. Nevertheless, this approach builds

up a good understanding on the behavior of active arrays. Two modes of operation can be

specified: 1) Transmission mode (shown in Fig. 3.16(a)) for controlling the beam amplitude,

and 2) reflection mode (shown in Fig. 3.16(b)) for changing the beam phase. As shown in Fig.

3.16(b), in the reflection mode, the array is backed by a metallic surface to assure the total

reflection.

Figure 3.17: An active L-C array comprising metallic strips interrupted by gaps in a periodic fashion. The gaps further are loaded with varactor diodes to alter the gap capacitance.



The first experimental version of L-C active arrays in reflection mode was demonstrated

previously. This reflective array was designed to work at 93 GHz and was able to produce a

70° phase shift with about 6.5 dB loss. This research was further expanded and lead to

demonstration of more involved active arrays such as multiplier arrays, oscillator arrays, and

amplifier arrays. Transmission mode operation has also been tested and demonstrated.

3.8.5 Chapter Conclusions

A numerical, comparative study over the conventional elements in design of frequency-

selective surfaces is presented in this chapter. Traditional surfaces are categorized based on

their geometries, into four major groups: I) the center connected or N-poles, such as dipole,

three-legged element, the Jerusalem cross, and the square spiral. II) The loop types such as the

three- and four-legged loaded elements, the circular loops, and the square and hexagonal loops.

III) Solid interiors or plate types. IV) Combinations which are a mixer of previous groups.

Through a physical discussion, it is shown that the operation mechanism of the traditional FSS

is dependent on the structural-based resonances. The overview study of this section is followed

by introducing some of the conventional applications involving frequency-selective surfaces,

demonstrating the versatility of these surfaces in microwave engineering.

(a) (b)

Figure 3.18: Modes of operation for an active L-C array. (a) Transmission mode for adjusting the wave intensity. (b) Reflection mode for changing the phase

CHAPTER – 4

Some Significant Works on



37

Some Significant Works on 4__


Frequency Selective Surface (FSS) has find lot of applications in Electromagnetic Shielding

and it has become the base for many works on Electromagnetic Shielding. Some of that works

are as follows:

4.1 A Single-Layer Frequency Selective Surface for Ultra-Wideband Electromagnetic

Shielding:-

An efficient approach to achieve the shielding effectiveness (SE) by using a frequency-

selective surface (FSS) is presented. This FSS, which consists of cross dipoles and rings printed

on the opposite sides of a single-layer FR-4 substrate, exhibits a wide, 7.5-GHz stop band to

provide simultaneous shielding in both X- and Ka-bands. Experimental results confirm SE of

the prototype over an ultra-wide band with more than 20-dB measured attenuation. The design

is compact and suitable to provide shielding against the radiation interference caused by

license-free and other radio systems.

Figure 4.1: (a) Top surface: cross dipole element. (b) Bottom surface: ring patch Element. (c) Perspective view: both (all dimensions are in mm).

Some Significant Works on Frequency Selective Surface [38]


This paper presents a single-layer FSS that exhibits an attenuation of more than 20 dB with an

ultra-wide 7.5-GHz stop band from 6.5 to 14 GHz range.

Figure 4.2: Measurement setup showing the FSS prototype, two ridge horn antennas and network analyzer connected via 50-Ω SMA cables.

Figure 4.3: Comparison of the numerically computed transmission through the FSS, with only the cross dipoles, with only the rings and with both cross dipoles and rings.



The FSS uses cross dipoles and rings printed on the opposite surfaces of a thin dielectric

substrate. This design, which provides a wide 7.5-GHz stopband from 6.5 to 14 GHz, has been

successfully tested to provide effective shielding with a minimum attenuation from around 20–

35dB. Due to symmetry, the designed FSS is stable to both TE and TM linear waves

polarizations as well as all other polarizations of normally incident waves and its oblique

incidence performance is satisfactory for smaller angles of incidence. The FSS can be easily

tuned to a desirable stopband by properly choosing the dimensions of resonators and other key

parameters. A complete parametric study is presented to demonstrate the possible control of

stopband.

4.2 An EMI Shielding FSS for Ku-Band Applications:-

A FSS consisting of cross-slot elements coated on a flat glass was used to study its impact on

transmission and reflection at Ku-band frequencies of 12.2~12.7 GHz. Based on the simulation

experience of the first partial study, the FSS element with geometrical parameters W=1.0 mm,

L=7.2 mm, P =9 mm, and T=5 mm was proposed for signals to transmit the flat glass coated

with the FSS at Ku-band frequencies of 12.2~12.7 GHz and to be blocked by the flat glass

coated with the FSS outside the Ku-band frequencies. From simulation results and

measurement data, it is shown that the resonant frequency of the flat glass coated with the

proposed FSS is found to be around 12.5 GHz which is in the Ku-band (12.2~12.7 GHz) for

the reflection with better than -10 dB. It is also shown that high transmission in the useful

frequency band (12.2~12.7 GHz) can be achieved by using the proposed FSS coated on the flat

glass.

Figure 4.4: An element of the FSS coated on a flat glass



Dimensions. Parameters shown in the figure are W=0.4~1.6 mm L=6~7.4 mm, P=9~12 mm,

and T=2~7 mm.

Based on simulation studies, a cross-slot FSS element with geometrical parameters W=1.0 mm,

L=7.2 mm, P =9 mm, and T=5 mm was proposed for signals to transmit the flat glass coated

with the FSS at Ku-band frequencies of 12.2~12.7 GHz and to be blocked by the flat glass

coated with the FSS outside the Ku-band frequencies. From simulation results and

measurement data, it is demonstrated that the flat glass coated with the FSS having geometrical

parameters W=1.0 mm, L=7.2 mm, P =9 mm, and T=5 mm can successfully be used to finish

the above purpose.

4.3 A novel shield for GSM 1800MHz band using frequency selective surface

This paper describes a novel FSS which functions as band stop filter to shield the GSM

1800MHz downlink band. The FSS is designed to operate with the resonant frequency of

1820MHz which is the center frequency for the GSM 1800MHz downlink band. The novelty

is attributed to its unique geometry and the circular apertures endowed with it. The proposed

geometry provides shielding effectiveness of 20 dB alongside with 133MHz bandwidth.

The structure holds identical response for both TE and TM Modes of polarization. In addition,

the geometry with its circular apertures, a hitherto unexplored feasibility serves the purpose of

ventilation and heat dissipation. The simulated results are validated using experimental

measurements.

Figure 4.5: Measurement data of reflection (S11) for the square flat Glass coated with and without the proposed FSS.



4.3.1 Proposed FSS geometry:

The unit cell contour of the proposed FSS is portrayed in Fig. 4.6. The structure is obtained by

rotating the alphabet V one to the other by 90± to get a cross like design (CLD) and is printed

on either side of the dielectric substrate. The circular slots penetrating deep into the substrate

in each of its notches cater to the ventilation needs.

Figure 4.6: Unit cell geometry.

Figure 4.7 (A): TE and TM mode characteristics Figure 4.7 (B): Comparison of the transmission characteristics



4.4 Electromagnetic Shielding Glass of Frequency Selective Surface:-

A Frequency Selective Surface (FSS) consists of a cluster of thin wires that act as dipole

antennas. The cross-section of the reflecting aperture is much wider than that of the wire dipole

antenna. Since an FSS provides a similar response to a dipole antenna, its reflection

corresponds with the dipole resonant frequency. We have developed FSS Glass, which has star

elements on its surface. For 0.5mm diameter silver wire, the typical peak attenuation of FSS

operating at 1.9GHz is 35dB or more. Since FSS Glass has a wide reflecting aperture, the

window glass provides a reflector panel, like a metal plate. There are noncontact gaps between

the frame and the FSS element, but the Electro-magnetic wave cannot penetrate the window

glass. The FSS Glass enables the PHS resources to be utilized efficiently.

4.4.1 Outline of FSS Glass

FSS Glass is an Electro-magnetic wave reflector, and it has a cluster of thin antennas printed

on it the two antennas are short circuited (resistance Rt=O), making half wavelength dipole

antenna. This antenna thus makes an area called a “Reflecting aperture”, which corresponds to

the dipole resonant frequency. The reflecting aperture is calculated as hX 1/2 h. When these

apertures are arranged correctly on the window glass, they form a frequency selective shield.

As shown in Fg-5 the FSS Glass has a particular attenuation peak at 1.9GHz. It attenuates 35dB

or more, and the FSS Glass has a 35MHz period of band around l.9GHz, where it attenuates

more than 30dB. The result shows that the FSS Glass has satisfactory qualities as a PHS shield

Figure 4.8: Attenuation Peak

Frequency (GHz)

Att

enu

ati

on

(d

B)

CHAPTER – 5

Design and Simulation of Multi-Band

EMI Rejection Shield


43

Design and Simulation of 5__

Multi-Band EMI Rejection Shield

5.1 Introduction

In this ultra-thin and multi-band EMI shield, the design of the FSS structure is made circular

loop type. To achieve multi-band rejection capability with a single layer structure, a ring

structure is proposed, as multiple resonances can be created using concentric rings of different

radii. The simulation is done on a unit cell and result is obtained based on that cell.

5.2 Design of Multiple Band Stop EMI Shield

We have inspired from a journal of Electromagnetic analysis and applications and try to work

on that type of structural arrangements mentioned in that journal.

The objective of the screen is to act as a spatial band stop filter. Hence, the appropriate

configuration would be a patch design. This configuration allows the screen to act as a band-

stop filter at specific intended resonant frequency. To meet the other objectives of providing

optical transmission, concentric rings would be the most appropriate design. The use of hollow

rings rather than patches makes it almost ideal for application where high degree of optical

transparency is needed.

Figure 5.1: Geometry of a unit cell and periodic array of single-ring and multiple concentric rings.

Design and Simulation of Multi-Band EMI Rejection Shield [44]


As mentioned earlier, the ring structure is selected as it allows multiple rings to be placed on

the same layer. Figure 5.1 shows the geometry of a unit.

The fundamental resonance for a concentric ring can be estimated with the following equation:

λ = c/f = 2𝜋𝑟

where, λ is the resonant wavelength

c is the speed of light in vacuum,

f is the resonant frequency and

r is the radius of the ring.

For a single ring, its inductance is given by:

L = 39.37 (𝑟2

8𝑟+11𝑤) k

Where, K = 0.57-0.145 ln(𝜔/ℎ)

k is a correction factor to account for the effect of the ground plane

h is the substrate thickness and

w is the width of the ring.

With the known inductance and the resonant frequency of the ring structure, its equivalent

capacitance can be obtained by:

C = (1/𝐿) × (1/2𝜋𝑓)2

With the calculated equivalent inductance and capacitance of the ring, the ABCD matrix

representation of a ring is shown in Figure 5.2

The reflection and transmission coefficients can be determined as follows:

𝑆11 =(𝐴+𝐵−𝐶−𝐷)

(𝐴+𝐵+𝐶+𝐷)

𝑆21 = 2(𝐴𝐷−𝐵𝐶)

(𝐴+𝐵+𝐶+𝐷)



Figure 5.2: Geometry of a unit cell of multiple concentric rings

Figure 5.3: Equivalent circuit and two-port ABCD network representations of single ring



Where A = D = 1, for a shunt network, B = 0 and C is the normalized admittance Y. For a

series network, B is the normalized impedance and C = 0. The values of 𝑆11 and 𝑆12 allow

efficient computation of the transmission and reflection coefficients.

5.3 Parameters and Geometrical Dimensions for Triple Band Stop EMI Shield

Here, Teflon is used as the dielectric substrate. The thickness of the substrate is 0.1 mm. around

the substrate two air boxes are created. The height of those air boxes is 20 mm.

Now, on the substrate the FSS structure is created. The ring structures are made of PEC

material.

There are three concentric rings on the substrate. The radius of the innermost circle is 19.5 mm.

and the thickness is 3 mm. The radius of the second circle is 25.5 mm which has a width of

25.5 mm. And the outermost circle’s radius is 50 mm which has a thickness of 3 mm.

Figure 5.4: 3D Model of a unit cell of multiple concentric rings



5.4 Setup of Full Wave Model

Once the initial geometrical dimensions of the ring structure is obtained from the equivalent

circuit model, 3D full wave simulation of the structure can be carried out using a commercial

3D EM solver software with the necessary boundary conditions being considered.

To perform the simulation for a unit cell using finite difference time domain (FDTD), special

boundary conditions have to be applied. A unit cell is constructed with two wave-guide ports,

placed in the -z and +z direction, being used as the excitation and receiving source. For

simulating periodic boundary conditions in time domain, a perfect electric wall is placed in the

-x and +x direction while a perfect magnetic wall is placed in the -y and +y direction.

Figure 5.5: Parameters and geometrical dimensions concentric ring



5.5 Result and Analysis

5.5.1 Analysis of Full Wave Model

Figure 5.6: Experimental and simulation result of tri-band rejection EMI shield

Here reflection coefficient S(1,1) and transmission coefficient S(1,2) are taken in the range of

0 GHz – 3.50 GHz. There are three bands shielded.

Figure 5.7: First stop band measured at -3 dB



Stop Band Frequency Range

(−3 𝑑𝐵)

Bandwidth

(−3 𝑑𝐵)

Centre Frequency

1nd stop band 0.52 GHz – 1.32 GHz 0.8080 GHz 0.917 GHz


(−10 𝑑𝐵)

Bandwidth

(−10 𝑑𝐵)

Centre Frequency

1st Stop Band 0.76 GHz – 1.07 GHz 0.3185 GHz 0.917 GHz

Figure 5.8: First stop band measured at -10 dB

Figure 5.9: Second stop band measured at -3 dB




(−3 𝑑𝐵)

Bandwidth

(−3 𝑑𝐵)

Centre Frequency

2nd Stop Band 1.74 GHz – 2.04 GHz 0.3026 GHz 1.84 GHz


(−10 𝑑𝐵)

Bandwidth

(−10 𝑑𝐵)

Centre Frequency

2nd stop band 1.79 GHz – 1.90 GHz 0.1079 GHz 1.84 GHz

Figure 5.11: Third stop band measured at -3 dB

Figure 5.10: Second stop band measured at -10 dB




(−3 𝑑𝐵)

Bandwidth

(−3 𝑑𝐵)

Centre Frequency

3rd Stop band 2.36 GHz – 2.93

GHz

0.5717 GHz 2.47 GHz


(−10 𝑑𝐵)

Bandwidth

(−10 𝑑𝐵)

Centre Frequency

3rd Stop band 2.42 GHz – 2.56 GHz 0.1425 GHz 2.47 GHz

A triple band-stop EMI shield is designed using the results obtained from the earlier study.

Based on the previous study, it was observed that in order to achieve a shield with resonant

frequencies at 917 MHz, 1.846 GHz, and 2.47 GHz, the radii of the rings are approximately

19.5 mm, 25.5 mm, and 50 mm, respectively. Using the initial findings, further fine-tuning is

carried out to achieve the desired dimensions for all resonances to be met on a single design.

Figure 5.12: Third stop band measured at -10 dB



5.5.2 Single Ring with Different Radii

Figure 5.13 shows the full-wave simulated and estimated transmission coefficients for different

radii, respectively. It can be seen that the results estimated based on the equivalent circuit model

(ECM) resemble well with those simulated using full-wave simulation tool except very slight

shift in resonant frequency. However, the ECM approach is highly efficient to synthesize the

design quickly without heavy computational effort. This allows parametric analysis to be

carried out efficiently as compared to full-wave simulation.

Figure 5.13: full-wave simulated and estimated transmission coefficients for different radii


53

CONCLUSION

The design of a frequency selective EMI shield, with the ability to reject different resonant

frequencies discussed. The design is implemented using screen-printing technology, which is a

suitable for low cost and large volume production.

This paper has shown that multi-band rejection EMI shield can be realized based on screen-

printing of conductive concentric rings on a single layer, which makes it ultra-thin and highly

flexible. The initial design of the EMI shield can be easily established using an efficient equivalent

circuit approach and further fine tuning can be achieved with full-wave simulation. Using a tri-

band rejection EMI shield as an example, good agreement has been demonstrated experimentally.

Teflon was used as the dielectric substrate. The implementation of the designs on different

substrates makes it suitable for application of the EMI shield onto walls and windows.

The performance is able to provide adequate reduction in the RF signal to minimize its impact on

EMI sensitive devices, thus demonstrating it to be a feasible solution for light weight and flexible

architectural shielding.

This kind of EMI shields can be easily applied as wall-paper to existing building walls for rejection

of unwanted frequency bands without affecting the desirable frequency bands.


54

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