ta2022 stereo 90w (4Ω) class-t™ digital audio amplifier ... · stereo 90w (4Ω) class-t™...

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Tripath Technology, Inc. - Technical Information 1 TA2022 – KLI/1.2/07-04 TA2022 STEREO 90W (4 ) CLASS-T™ DIGITAL AUDIO AMPLIFIER DRIVER USING DIGITAL POWER PROCESSING (DPP™) TECHNOLOGY Technical Information Revision 1.2 – JULY 2004 GENERAL DESCRIPTION The TA2022 is a 90W (4) continuous average per channel Class-T Digital Audio Power Amplifier IC using Tripath’s proprietary Digital Power Processing (DPP TM ) technology. Class-T amplifiers offer both the audio fidelity of Class-AB and the power efficiency of Class-D amplifiers. APPLICATIONS ¾ DVD Players ¾ Mini/Micro Component Systems ¾ Home Theater ¾ Powered Speakers BENEFITS ¾ Fully integrated solution with internal FETs ¾ Dramatically improves efficiency versus Class-AB amplifiers ¾ Signal fidelity equal to high quality linear amplifiers ¾ High dynamic range compatible with digital media such as CD and DVD FEATURES ¾ Class-T architecture ¾ High Power ¾ 100W @ 4, 1.0% THD+N ¾ 90W @ 4, 0.1% THD+N ¾ 60W @ 8, 0.1% THD+N ¾ “Audiophile” Sound Quality ¾ 0.015% THD+N @ 70W 4¾ 0.015% THD+N @ 45W 8¾ 0.10% IHF-IM @ 25W 4¾ High Efficiency ¾ 92% @ 88W 8¾ 87% @ 125W 4¾ Dynamic Range = 102 dB ¾ Mute Input ¾ Integrated Gate Drive Supply ¾ Over-current protection ¾ Over and under-voltage protection ¾ Single ended outputs ¾ Outputs can be operated in bridged mode ¾ 32-pin SSIP package

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  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    1 TA2022 KLI/1.2/07-04

    TA2022

    STEREO 90W (4) CLASS-T DIGITAL AUDIO AMPLIFIER DRIVER USING DIGITAL POWER PROCESSING (DPP) TECHNOLOGY T e c h n i c a l I n f o r m a t i o n R e v i s i o n 1 . 2 J U L Y 2 0 0 4

    G E N E R A L D E S C R I P T I O N

    The TA2022 is a 90W (4) continuous average per channel Class-T Digital Audio Power Amplifier IC using Tripaths proprietary Digital Power Processing (DPPTM) technology. Class-T amplifiers offer both the audio fidelity of Class-AB and the power efficiency of Class-D amplifiers.

    A P P L I C A T I O N S DVD Players Mini/Micro Component Systems Home Theater Powered Speakers

    B E N E F I T S Fully integrated solution with internal

    FETs Dramatically improves efficiency versus

    Class-AB amplifiers Signal fidelity equal to high quality linear

    amplifiers High dynamic range compatible with

    digital media such as CD and DVD

    F E A T U R E S Class-T architecture High Power 100W @ 4, 1.0% THD+N 90W @ 4, 0.1% THD+N 60W @ 8, 0.1% THD+N

    Audiophile Sound Quality 0.015% THD+N @ 70W 4 0.015% THD+N @ 45W 8 0.10% IHF-IM @ 25W 4

    High Efficiency 92% @ 88W 8 87% @ 125W 4

    Dynamic Range = 102 dB Mute Input Integrated Gate Drive Supply Over-current protection Over and under-voltage protection Single ended outputs Outputs can be operated in bridged mode 32-pin SSIP package

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    2 TA2022 KLI/1.2/07-04

    A B S O L U T E M A X I M U M R A T I N G S (Note 1) SYMBOL PARAMETER Value UNITS

    VPP, VNN Supply Voltage (VPP1, VPP2, VNN1, VNN2) +/- 40 V V5 Positive 5V Bias Supply

    Voltage at Input Pins (pins 18, 19, 23, 24, 26, 28-32) 6

    -0.3V to (V5+0.3V) V V

    VN10 Voltage for low-side FET drive VNN+13 V TSTORE Storage Temperature Range -55 to 150 C TA Operating Free-air Temperature Range (Note 2) -40 to 85 C TJ Junction Temperature 150 C ESDHB ESD Susceptibility Human Body Model (Note 3)

    All pins (except pin 27) Pin 27

    4000 1500

    V V

    ESDMM ESD Susceptibility Machine Model (Note 4) All pins

    200

    V

    Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur.

    See the table below for Operating Conditions. Note 2: This is a target specification. Characterization is still needed to validate this temperature range. Note 3: Human body model, 100pF discharged through a 1.5K resistor. Note 4: Machine model, 220pF 240pF discharged through all pins. O P E R A T I N G C O N D I T I O N S (Note 5)

    SYMBOL PARAMETER MIN. TYP. MAX. UNITS VPP, VNN Supply Voltage (VPP1, VPP2, VNN1, VNN2) +/- 12 +/-31 +/- 36 V V5 Positive 5 V Bias Supply 4.5 5 5.5 V VN10 Voltage for FET drive (Volts above VNN) 9 11 12 V Note 5: Recommended Operating Conditions indicate conditions for which the device is functional.

    See Electrical Characteristics for guaranteed specific performance limits. T H E R M A L C H A R A C T E R I S T I C S

    SYMBOL PARAMETER VALUE UNITS JC Junction-to-case Thermal Resistance 1.0 C/W

    JA Junction-to-ambient Thermal Resistance (still air) 20 C/W

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    3 TA2022 KLI/1.2/07-04

    E L E C T R I C A L C H A R A C T E R I S T I C S (Notes 6, 7)

    TA = 25 C. See Application/Test Circuit on page 8. Unless otherwise noted, the supply voltage is VPP=|VNN|=31V.

    SYMBOL PARAMETER CONDITIONS MIN. TYP. MAX. UNITS

    Iq Quiescent Current (No load, Mute = 0V)

    VPP = +31V VNN = -31V (Note 8) V5 = 5V (Note 9) VN10 = 11V (Note 10)

    20 55 45 65

    60 80

    mA mA mA mA

    IMUTE Mute Supply Current (No load, Mute = 5V)

    VPP = +31V VNN = -31V (Note 8) V5 = 5V (Note 9)

    0.5 2 20

    25

    mA mA mA

    VIH High-level input voltage (MUTE) IIH = See Mute Control Section 3.5 V VIL Low-level input voltage (MUTE) IIL = See Mute Control Section 1.0 V VOH High-level output voltage (HMUTE) IOH = 3mA 4.0 V VOL Low-level output voltage (HMUTE) IOL = 3mA 0.5 V VOFFSET Output Offset Voltage No Load, MUTE = Logic low

    0.1% RFBA, RFBB, RFBC resistors -750 750 mV

    IOC Over Current Sense Threshold TBD 7 8 A IVPPSENSE VPPSENSE Threshold Currents

    Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted)

    138

    62

    162 154 79 72

    178

    87

    A A A A

    VVPPSENSE Threshold Voltages with RVPPSENSE = 249K (Note 11, Note 12)

    Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted)

    36.5

    17.8

    42.8 40.9 22.2 20.4

    47.3

    24.4

    V V V V

    IVNNSENSE VNNSENSE Threshold Currents

    Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted)

    152

    65

    174 169 86 77

    191

    95

    A A A A

    VVNNSENSE Threshold Voltages with RVNNSENSE = 249K (Note 11, Note 12)

    Over-voltage turn on (muted) Over-voltage turn off (mute off) Under-voltage turn off (mute off) Under-voltage turn on (muted)

    -36.2

    -14.8

    -42.1 -40.8 -20.2 -17.9

    -46.8

    -22.6

    V V V V

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    4 TA2022 KLI/1.2/07-04

    P E R F O R M A N C E C H A R A C T E R I S T I C S S I N G L E E N D E D (Notes 6, 7)

    TA = 25 C. Unless otherwise noted, the supply voltage is VPP=|VNN|=31V, the input frequency is 1kHz and the measurement bandwidth is 20kHz. See Application/Test Circuit on Page 8.

    SYMBOL PARAMETER CONDITIONS MIN. TYP. MAX. UNITS POUT Output Power

    (Continuous Average/Channel) (Note 13)

    VPP = |VNN| = +/-31V, RL = 4 THD+N = 0.1% THD+N = 1.0% THD+N = 10%

    VPP = |VNN| = +/-35V, RL = 8 THD+N = 0.1% THD+N = 10%

    80

    90

    100 125

    60 88

    W W W

    W W

    THD + N Total Harmonic Distortion Plus

    Noise POUT = 70W/Channel, RL = 4 VPP = |VNN| = +/-31V POUT = 45W/Channel, RL = 8 VPP = |VNN| = +/-35V

    0.015

    0.015

    %

    %

    IHF-IM IHF Intermodulation Distortion 19kHz, 20kHz, 1:1 (IHF), RL = 4 POUT = 25W/Channel

    0.1 %

    SNR Signal-to-Noise Ratio A-Weighted 0dB = 90W/Channel, RL = 4

    102 dB

    CS Channel Separation 0dB = 25W, RL = 4 83 dB AV Amplifier Gain POUT = 10W/Channel, RL = 4,

    See Application / Test Circuit 18.1 V/V

    AVERROR Channel to Channel Gain Error POUT = 10W/Channel, RL = 4 See Application / Test Circuit

    0.5 dB

    Power Efficiency POUT = 88W/Channel, RL = 8 POUT = 125W/Channel, RL = 4

    92 87

    % %

    ISLOAD Source Current POUT = 125W/Channel, RL = 4 VPP = +31V VNN = -31V V5 = 5V

    4.59 4.61 45

    A A

    mA eNOUT Output Noise Voltage A-Weighted, input AC grounded 150 V

    P E R F O R M A N C E C H A R A C T E R I S T I C S B R I D G E D T I E D L O A D (Notes 6, 7)

    TA = 25 C. Unless otherwise noted, the supply voltage is VPP=|VNN|=30V, the input frequency is 1kHz and the measurement bandwidth is 20kHz.

    SYMBOL PARAMETER CONDITIONS MIN. TYP. MAX. UNITS POUT Output Power

    (Continuous Average) (Note 13)

    VPP = |VNN| = +/-30V, RL = 8 THD+N = 0.1% THD+N = 10%

    150 235

    W W

    THD + N Total Harmonic Distortion Plus Noise

    POUT = 100W, RL = 8 0.05 %

    IHF-IM IHF Intermodulation Distortion 19kHz, 20kHz, 1:1 (IHF), RL = 8 POUT = 25W

    0.10 %

    Power Efficiency POUT = 225W, RL = 8 87 % SNR Signal-to-Noise Ratio A-Weighted, RL = 8

    0dB = 150W 104 dB

    eNOUT Output Noise Voltage A-Weighted, input AC grounded 220 V

    Note 6: Minimum and maximum limits are guaranteed but may not be 100% tested. Note 7: For operation in ambient temperatures greater than 25C, the device must be derated

    based on the maximum junction temperature and the thermal resistance determined by the mounting technique.

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    5 TA2022 KLI/1.2/07-04

    Note 8: This specification includes the current draw from the internal buck regulator. If an external floating supply is used, instead of the internal buck regulator, the quiescent current draw of the VNN supply will be approximately 20mA.

    Note 9: This specification includes the current draw from both the TA2022 and the external feedback biasing.

    Note 10: This is the current draw of the VN10 pin if an external floating 11V supply is used instead of the internal buck regulator

    Note 11: These supply voltages are calculated using the IVPPSENSE AND IVNNSENSE values shown in the Electrical Characteristics table. The typical voltage values shown are calculated using a RVPPSENSE and RVNNSENSE value of 249kohm without any tolerance variation. The minimum and maximum voltage limits shown include either a +1% or 1% (+1% for Over-voltage turn on and Under-voltage turn off, -1% for Over-voltage turn off and Under-voltage turn on) variation of RVPPSENSE or RVNNSENSE off the nominal 249kohm value. These voltage specifications are examples to show both typical and worst case voltage ranges for a given RVPPSENSE and RVNNSENSE resistor value of 249kohm. Please refer to the Application Information section for a more detailed description of how to calculate the over and under voltage trip voltages for a given resistor value.

    Note 12: The fact that the over-voltage turn on and over-voltage turn off specifications exceed the absolute maximum of +/-40V for the TA2022 does not imply that the part will work at these elevated supply voltages. It also does not imply that TA2022 is tested or guaranteed at these supply voltages. The supply voltages are simply a calculation based on the process spread of the IVPPSENSE and IVNNSENSE currents (see Note 11). The supply voltage must be maintained below the absolute maximum of +/-40V or permanent damage to the TA2022 may occur.

    Note 13: The supply voltage limitation for 4 ohm single ended (+/-31V), or 8 ohm bridged (+/-30V), is based on the current limit protection circuitry. The current limit circuitry may be activated during large output excursions if the recommended supply voltage ranges are exceeded. This will result in the amplifier being muted.

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    6 TA2022 KLI/1.2/07-04

    TA2022 PINOUT

    P I N D E S C R I P T I O N Pin Function Description

    1, 13 VBOOT2, VBOOT1 Bootstrap voltages for gate drive of high side MOSFETs 2 VN10 Floating supply input. Normally connected to the output of onboard VN10 buck

    converter. This voltage must be stable and referenced to VNN. 3 VN10GND Power ground for onboard VN10 generator. Electrically tied to the TA2022 case.

    4, 12 VPP2, VPP1 Positive power supply input pins. 5 VN10SW Switching output voltage for onboard VN10 generator (buck converter). 6 NC Not connected internally. May be connected to pin 7 without any loss of

    functionality or performance. 7,10 OUT2, OUT1 Power amplifier outputs. 8, 9 VNN2, VNN1 Negative power supply inputs. 11 NC Not connected internally. May be connected to pin 10 without any loss of

    functionality or performance. 14 VN10FDBK Feedback for onboard VN10 generator (nominally 11V above VNN)

    15, 20 AGND Analog Ground. 16, 21 V5 5V power supply input.

    17 REF Used to set internal bias currents. The pin voltage is typically 1.1V. 18 VNNSENSE Negative supply voltage sense input. This pin is used for both over and under

    voltage sensing for the VNN supply. 19 VPPSENSE Positive supply voltage sense input. This pin is used for both over and under

    voltage sensing for the VPP supply. 22, 25 OAOUT1, OAOUT2 Outputs of Input Stage op amps. 23, 26 INV1, INV2 Inverting inputs of Input Stage op amps.

    24 MUTE Logic input. A logic high puts the amplifier in mute mode. Ground pin if not used. Please refer to the section, Mute Control, in the Application Information.

    27 BIASCAP Bandgap reference times two (typically 2.5VDC). Used to set the common mode voltage for the input op amps. This pin is not capable of driving external circuitry.

    28, 29 FBKGND2, FBKOUT2 Output voltage differential feedback for channel 2. 30,31 FBKGND1, FBKOUT1 Output voltage differential feedback for channel 1.

    32 HMUTE Logic Output. A logic high indicates both amplifiers are muted, due to the mute pin state, or a fault such as an overcurrent, undervoltage, or overvoltage condition.

    FBKGND1

    AGND

    VNNSENSE VPPSENSE

    AGND V5

    OAOUT1 INV1

    MUTE OAOUT2

    INV2

    VBOOT2

    BIASCAP FBKGND2 FBKOUT2

    FBKOUT1

    V5

    VN10FDBK VBOOT1

    VPP1 NC

    OUT1 VNN1 VNN2 OUT2

    NC

    VPP2 VN10GND

    VN10

    HMUTE

    30

    16 17 18 19 20 21 22 23 24 25 26 27 28 29

    1

    15 14 13

    11 10

    12

    9 8 7 6 5 4 3 2

    32-pin SSIP Package (Front View)

    32 31

    REF

    VN10SW

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    7 TA2022 KLI/1.2/07-04

    A P P L I C A T I O N / T E S T C I R C U I T

    TA2022

    RL4 or 8

    HMUTE

    OAOUT1

    INV1OUT1

    CI3.3uF

    22

    23

    32

    10+

    VPP1

    VNN1

    Processing&

    ModulationCO

    0.22uF

    LO10uH, 10A

    Analog Ground

    Power Ground

    (Pin 9)

    RF20K

    RI20K

    CZ0.22uF

    RZ6.2, 2W

    CB0.1uF

    CBAUX47uF

    VBOOT113 RB 250

    VNN1

    VN10

    DB11DQ09 VN10

    FBKOUT131FBKGND130

    RFBA1K

    *RFBB1.1K

    *RFBB1.1K

    RFBA1K

    DOMUR120

    *RFBC9.1K

    *RFBC9.1K

    V5 (Pin 21)

    AGND (Pin 20)

    +

    V5

    AGND

    CFB390pF

    CS0.1uF

    CS0.1uF

    21

    20

    V5

    15

    AGND

    AGND

    V516

    5V

    OAOUT2

    INV2 OUT2

    CI3.3uF

    25

    26 7+

    VPP2

    VNN2

    Processing&

    Modulation CO0.22uF

    LO10uH, 10A

    (Pin 8)

    RF20K

    RI20K

    CZ0.22uF

    RZ6.2, 2W

    CB0.1uF

    CBAUX47uF

    VBOOT21 RB 250

    VNN2

    VN10

    DB11DQ09

    VN10

    FBKOUT229FBKGND228

    RFBA1K

    *RFBB1.1K

    *RFBB1.1K

    RFBA1K

    DOMUR120

    *RFBC9.1K

    *RFBC9.1K

    V5 (Pin 21)

    +

    -+

    V5

    AGND

    CFB560pF

    MUTE 245V

    V5

    BIASCAP

    CA0.1uF 27

    17REF

    RREF8.25K, 1%

    12VPP1

    9VNN1+CHBR

    0.1uF,100V Cs220uF, 50V

    CS0.1uF, 50V

    +

    Cs220uF, 50V

    CS0.1uF, 50V

    8VNN2

    4VPP2CHBR0.1uF,100V

    VPP

    VNN

    VN10GND

    *RVPPSENSE(VPP1)VPP

    VNN

    19 VPPSENSE

    18 VNNSENSE

    3

    VN10Generator

    VNN

    AGND (Pin 20)

    VN10SW5

    CSWFB0.1uF,50V

    VN10FBK14

    VNN

    CSW0.1uF,35V

    VN102

    VNN

    DSW11DQ09

    LSW100uH, 1A

    RSWFB1K

    +CSW100uF, 35V

    VN10

    RL4 or 8

    2.5V

    200K

    (Pin 20)

    (Pin 20)

    (Pin 20)

    ROFA50K

    ROFB10K

    to FBKGND1(Pin30)

    Offset TrimCircuit

    V5 (Pin 21)

    AGND (Pin 20)

    VNN

    249K, 1%

    ROFA50K

    ROFB10K

    to FBKGND2(Pin28)

    Offset TrimCircuit

    V5 (Pin 21)

    AGND (Pin 20)

    -+

    *RVNNSENSE(VNN1)249K, 1%

    6 NC11 NC

    *RVPP2V5

    V5*RVNN2

    Optional Components -See Application Information

    * The values of these components must beadjusted based on supply voltage range.See Application Information.

    VPP2(Pin 4) DO

    MUR120

    VPP1(Pin 12) DO

    MUR120

    Application / Test Diagram

    CS0.1uF, 50VCS0.1uF, 50VCS0.1uF, 50VCS0.1uF, 50V

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    8 TA2022 KLI/1.2/07-04

    E X T E R N A L C O M P O N E N T S D E S C R I P T I O N (Refer to the Application/Test Circuit)

    Components Description RI Inverting input resistance to provide AC gain in conjunction with RF. This input is biased

    at the BIASCAP voltage (approximately 2.5VDC). RF Feedback resistor to set AC gain in conjunction with RI. Please refer to the Amplifier

    Gain paragraph, in the Application Information section. CI AC input coupling capacitor which, in conjunction with RI, forms a high pass filter at

    )CR2(1f IIC = . RFBA Feedback divider resistor connected to V5. This resistor is normally set at 1k. RFBB Feedback divider resistor connected to AGND. This value of this resistor depends on

    the supply voltage setting and helps set the TA2022 gain in conjunction with RI, RF, RFBA, and RFBC. Please see the Modulator Feedback Design paragraphs in the Application Information Section.

    RFBC Feedback resistor connected from either the OUT1(OUT2) to FBKOUT1(FBKOUT2) or speaker ground to FBKGND1(FBKGND2). The value of this resistor depends on the supply voltage setting and helps set the TA2022 gain in conjunction with RI, RF, RFBA,, and RFBB. It should be noted that the resistor from OUT1(OUT2) to FBKOUT1(FBKOUT2) must have a power rating of greater than )(2RVPPP FBC2DISS = . Please see the Modulator

    Feedback Design paragraphs in the Application Information Section. CFB Feedback delay capacitor that both lowers the idle switching frequency and filters very

    high frequency noise from the feedback signal, which improves amplifier performance. The value of CFB should be offset between channel 1 and channel 2 so that the idle switching difference is greater than 40kHz. Please refer to the Application / Test Circuit.

    ROFA Potentiometer used to manually trim the DC offset on the output of the TA2022. ROFB Resistor that limits the manual DC offset trim range and allows for more precise

    adjustment. RREF Bias resistor. Locate close to pin 17 and ground at pin 20. CA BIASCAP decoupling capacitor. Should be located close to pin 27 and grounded at pin 20. DB Bootstrap diode. This diode charges up the bootstrap capacitors when the output is low

    (at VNN) to drive the high side gate circuitry. Schottky or fast recovery diode rated at least 200mA, 90V, 50nS is recommended for the bootstrap circuitry. In addition, the bootstrap diode must be able to sustain the entire VPP-VNN voltage. Thus, for most applications, a 90V (or greater) diode should be used.

    CB High frequency bootstrap capacitor, which filters the high side gate drive supply. This capacitor must be located as close to pin 13 (VBOOT1) or pin1n (VBOOT2) for reliable operation. The other side of CB should be connected directly to the OUT1 (pin 10) or OUT2 (pin 7). Please refer to the Application / Test Circuit.

    CBAUX Bulk bootstrap capacitor that supplements CB during clipping events, which result in a reduction in the average switching frequency.

    RB Bootstrap resistor that limits CBAUX charging current during TA2022 power up (bootstrap supply charging).

    CSW VN10 generator filter capacitors. The high frequency capacitor (0.1uF) must be located close to pin 2 (VN10) to maximize device performance. The value of the bulk capacitor should be sized appropriately such that the VN10 voltage does not overshoot with respect to VNN during TA2022 turn on. Tripath recommends using a value of 100F for the bulk capacitor.

    LSW VN10 generator filter inductor. This inductor sized appropriately so that LSW does not

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    9 TA2022 KLI/1.2/07-04

    saturate. If the recommended inductor value of 100H is not used, the VN10 may overshoot with respect to VNN during TA2022 turn on.

    DSW Flywheel diode for the internal VN10 buck converter. This diode also prevents VN10SW from going more than one diode drop negative with respect to VNN. This Diode can be a Fast Recovery, Switching or Shottky, but must be rated at least 200mA, 30V, 50nS.

    CSWFB VN10 generator feedback capacitor. This capacitor, in conjunction with RSWFB, filters the VN10 feedback signal such that the loop is unconditionally stable.

    RSWFB VN10 generator feedback resistor. This resistor sets the nominal VN10 voltage. With RSWFB equal to 1k, the internally VN10 voltage will typically be 11V above VNN.

    CS Supply decoupling for the power supply pins. For optimum performance, these components should be located close to the TA2022 and returned to their respective ground as shown in the Application/Test Circuit.

    RVNNSNESE Overvoltage and undervoltage sense resistor for the negative supply (VNN). Please refer to the Electrical Characteristics Section for the trip points as well as the hysteresis band. Also, please refer to the Over / Under-voltage Protection section in the Application Information for a detailed discussion of the internal circuit operation and external component selection.

    RVPPSENSE Overvoltage and undervoltage sense resistor for the positive supply (VPP). Please refer to the Electrical Characteristics Section for the trip points as well as the hysteresis band. Also, please refer to the Over / Under-voltage Protection section in the Application Information for a detailed discussion of the internal circuit operation and external component selection.

    CHBR Supply decoupling for the high current Half-bridge supply pins. These components must be located as close to the device as possible to minimize supply overshoot and maximize device reliability. These capacitors should have good high frequency performance including low ESR and low ESL. In addition, the capacitor voltage rating must be twice the maximum VPP voltage.

    CZ Zobel capacitor, which in conjunction with RZ, terminates the output filter at high frequencies. Use a high quality film capacitor capable of sustaining the ripple current caused by the switching outputs.

    RZ Zobel resistor, which in conjunction with CZ, terminates the output filter at high frequencies. The combination of RZ and CZ minimizes peaking of the output filter under both no load conditions or with real world loads, including loudspeakers which usually exhibit a rising impedance with increasing frequency. The recommended power rating is 2 watts.

    DO Fast Recovery diodes that minimize overshoots and undershoots of the outputs with respect to power ground during switching transitions as well as output shorts to ground. For maximum effectiveness, these diodes must be located close to the output pins and returned to their respective VPP and VNN. Also, they should be rated with a maximum Forward Voltage of 1V at 10A. Please see Application/Test Circuit for VPP and VNN return pins.

    LO Output inductor, which in conjunction with CO, demodulates (filters) the switching waveform into an audio signal. Forms a second order filter with a cutoff frequency of )CL2(1f OOC = and a quality factor of OOOL CLCRQ = . These inductors must be rated at least 10A with high linearity. Please see Output Filter Design section for details.

    CO

    Output capacitor, which, in conjunction with LO, demodulates (filters) the switching waveform into an audio signal. Forms a second order low-pass filter with a cutoff frequency of )CL2(1f OOC = and a quality factor of OOOL CLCRQ = . Use a

    high quality film capacitor capable of sustaining the ripple current caused by the switching outputs. Electrolytic capacitors should not be used.

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    10 TA2022 KLI/1.2/07-04

    T Y P I C A L P E R F O R M A N C E C H A R A C T E R I S T I C S S I N G L E E N D E D

    0.005

    10

    0.01

    0.02

    0.05

    0.1

    0.2

    0.5

    1

    2

    5

    2 1005 10 20 50Output Power (W)

    THD

    +N (%

    )

    THD+N vs Output Powerf = 1kHzRL= 8VPP=|VNN|=35VAES 17 Filter

    1

    0.005

    10

    0.01

    0.02

    0.05

    0.1

    0.2

    0.5

    1

    2

    5

    20 20k50 100 200 500 1k 2k 5k 10kFrequency (Hz)

    THD

    +N (%

    )THD+N vs Frequency

    Po = 25W/chRL = 8VPP=|VNN|=35V

    BW = 22kHz

    BW = 30kHz

    0.005

    10

    0.01

    0.02

    0.05

    0.1

    0.2

    0.5

    1

    2

    5

    20 20k50 100 200 500 1k 2k 5k 10kFrequency (Hz)

    THD

    +N (%

    )

    THD+N vs FrequencyPo = 50W/chRL = 4VPP=|VNN|=31V

    BW = 22kHz

    BW = 30kHz

    0.005

    10

    0.01

    0.02

    0.05

    0.1

    0.2

    0.5

    1

    2

    5

    1 2002 5 10 20 50 100Output Power (W)

    THD

    +N (%

    )

    THD+N vs Output Powerf = 1kHzRL= 4VPP=|VNN|=31VAES 17 Filter

    -140

    +0

    -130

    -120

    -110

    -100

    -90

    -80

    -70

    -60

    -50

    -40

    -30

    -20

    -10

    20 20k50 100 200 500 1k 2k 5k 10kFrequency (Hz)

    Ampl

    itude

    (dBr

    )

    Intermodulation Distortion19kHz, 20kHz 1:1Po = 25W/ch, 40dBr = 10.0VrmsVPP=|VNN|=31VBW = 22Hz - 30kHz

    -140

    +0

    -130

    -120

    -110

    -100

    -90

    -80

    -70

    -60

    -50

    -40

    -30

    -20

    -10

    20 20k50 100 200 500 1k 2k 5k 10kFrequency (Hz)

    Ampl

    itude

    (dBr

    )

    Intermodulation Distortion19kHz, 20kHz 1:1Po = 12.5W/ch, 80dBr = 10.0VrmsVPP=|VNN|=31VBW = 22Hz - 30kHz

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    11 TA2022 KLI/1.2/07-04

    T Y P I C A L P E R F O R M A N C E C H A R A C T E R I S T I C S S I N G L E E N D E D

    T Y

    0.005

    10

    0.01

    0.02

    0.05

    0.1

    0.2

    0.5

    1

    2

    5

    2 1005 10 20 50Output Power (W)

    THD

    +N (%

    )THD+N vs Output Power

    f = 1kHzRL= 8AES 17 Filter

    1

    +/-25V

    +/-30V

    +/-35V

    -100

    -40

    -95

    -90

    -85

    -80

    -75

    -70

    -65

    -60

    -55

    -50

    -45

    20 20k50 100 200 500 1k 2k 5k 10k

    Frequency (Hz)

    Cha

    nnel

    Sep

    arat

    ion

    (dBr

    )

    Channel Separation

    VPP=|VNN|=31VPo = 25W/ch, 4Po = 12.5W/ch, 8BW = 22Hz - 22kHz

    RL = 8

    RL = 4

    -120

    -70

    -115

    -110

    -105

    -100

    -95

    -90

    -85

    -80

    -75

    20 20k50 100 200 500 1k 2k 5k 10kFrequency (Hz)

    Ampl

    itude

    (dBr

    )

    Noise FloorVPP=|VNN|=31VRF=RI=20k16kFFTAES 17 Filter

    0.005

    10

    0.01

    0.02

    0.05

    0.1

    0.2

    0.5

    1

    2

    5

    1 2002 5 10 20 50 100

    Output Power (W)

    THD

    +N (%

    )

    THD+N vs Output Powerf = 1kHzRL= 4AES 17 Filter

    +/-23V

    +/-27V

    +/-31V

    0

    10

    20

    30

    40

    50

    60

    70

    80

    90

    100

    0 10 20 30 40 50 60 70 80 90 100 110 120

    Output Power (W)

    Effic

    ienc

    y (%

    )

    VPP=|VNN|=31VRL = 4AES 17 FilterTHD+N < 10%

    Efficiency vs Output Power

    0

    10

    20

    30

    40

    50

    60

    70

    80

    90

    100

    0 10 20 30 40 50 60 70 80 90

    Output Power (W)

    Effic

    ienc

    y (%

    )

    VPP=|VNN|=35VRL = 8AES 17 FilterTHD+N < 10%

    Efficiency vs Output Power

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    12 TA2022 KLI/1.2/07-04

    T Y P I C A L P E R F O R M A N C E C H A R A C T E R I S T I C S B R I D G E D

    -140

    +0

    -130

    -120

    -110

    -100

    -90

    -80

    -70

    -60

    -50

    -40

    -30

    -20

    -10

    20 20k50 100 200 500 1k 2k 5k 10kFrequency (Hz)

    Ampl

    itude

    (dBr

    )

    Intermodulation Distortion19kHz, 20kHz 1:1Po = 50W/ch, 8 BRIDGED0dBr = 20.0VrmsVPP=|VNN|=30VBW = 22Hz - 30kHz

    2 3005 10 20 50 100 2000.005

    10

    0.01

    0.02

    0.05

    0.1

    0.2

    0.5

    1

    2

    5

    Output Power (W)

    THD

    +N (%

    )

    THD+N vs Output Powerf = 1kHzRL= 8 BRIDGEDAES 17 Filter

    1

    +/-23V

    +/-27V

    +/-30V

    0.005

    10

    0.01

    0.02

    0.05

    0.1

    0.2

    0.5

    1

    2

    5

    20 20k50 100 200 500 1k 2k 5k 10kFrequency (Hz)

    THD

    +N (%

    )

    THD+N vs FrequencyPo = 100W/chRL = 8 BRIDGEDVPP=|VNN|=30V

    BW = 22kHz

    BW = 30kHz

    -120

    -70

    -115

    -110

    -105

    -100

    -95

    -90

    -85

    -80

    -75

    20 20k50 100 200 500 1k 2k 5k 10k

    Frequency (Hz)

    Ampl

    itude

    (dBr

    )

    Noise FloorVPP=|VNN|=30VRF=RI=20k8 BRIDGED16kFFTAES 17 Filter

    0

    10

    20

    30

    40

    50

    60

    70

    80

    90

    100

    0 20 40 60 80 100 120 140 160 180 200 220 240

    Output Power (W)

    Effic

    ienc

    y (%

    )

    VPP=|VNN|=30VRL = 8 BRIDGEDAES 17 FilterTHD+N < 10%

    Efficiency vs Output Power

    0.005

    10

    0.01

    0.02

    0.05

    0.1

    0.2

    0.5

    1

    2

    5

    2 3005 10 20 50 100 2001

    Output Power (W)

    THD

    +N (%

    )

    f = 1kHzVPP=|VNN|=30VRL= 8 BRIDGEDAES 17 Filter

    THD+N vs Output Power

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    13 TA2022 KLI/1.2/07-04

    A P P L I C A T I O N I N F O R M A T I O N

    T A 2 0 2 2 B a s i c A m p l i f i e r O p e r a t i o n

    The TA2022 has three major operational blocks: the signal processor, the MOSFET driver, and the power MOSFETs. The signal processor is a 5V CMOS block that amplifies the audio input signal and converts the audio signal to a switching pattern. This switching pattern is spread spectrum with a typical idle switching frequency of about 650kHz. The switching patterns for the two channels are not synchronized and the idle switching frequencies should differ by at least 40kHz to avoid increasing the audio band noise floor. The idle frequency difference can be accomplished by offsetting the value of CFB for each channel. Typical values of CFB are 390pF for channel 1 and 560pF for channel 2. The MOSFET driver level-shifts the signal processors 5V switching patterns to the power supply voltages and drives the power MOSFETs. The MOSFET driver includes a switching power supply integrated to generate the VN10 supply. The VN10 supply powers the low side gate drivers as well provides the charging current need for the bootstrapped supplies (VBOOT1 and VBOOT2) that power the high side MOSFET drivers. VN10 must be stable (regulated) at 10V to 12V above VNN. The VN10 circuitry shown in the Application / Test Circuit typically produces 11V above VNN. The power MOSFETs are N-channel devices configured in half-bridges and are used to supply power to the output load. The outputs of the power MOSFETs (OUT1 and OUT2) must be low pass filtered to remove the high frequency switching pattern. A residual voltage from the switching pattern will remain on the speaker outputs when the recommended output LC filter is used, but this signal is outside of the audio band and will not affect audio performance.

    C i r c u i t B o a r d L a y o u t

    The TA2022 is a power (high current) amplifier that operates at relatively high switching frequencies. The output of the amplifier switches between VPP and VNN at high speeds while driving large currents. This high-frequency digital signal is passed through an LC low-pass filter to recover the amplified audio signal. Since the amplifier must drive the inductive LC output filter and speaker loads, the amplifier outputs can be pulled above the supply voltage and below ground by the energy in the output inductance. To avoid subjecting the TA2022 to potentially damaging voltage stress, it is critical to have a good printed circuit board layout. It is recommended that Tripaths layout and application circuit be used for all applications and only be deviated from after careful analysis of the effects of any changes. Please refer to the TA2022 evaluation board document, EB-TA2022, available on the Tripath website, at www.tripath.com. The following components are important to place near their associated TA2022 pins and are ranked in order of layout importance, either for proper device operation or performance considerations.

    - The capacitors CHBR provide high frequency bypassing of the amplifier power supplies and will serve to reduce spikes across the supply rails. CHBR should be kept within 1/8 (3mm) of the VNN(8,9) and VPP(4,12) pins. Please note that both VNN1 and VPP1 as well as VNN2 and VPP2 must be decoupled separately. In addition, the voltage rating for CHBR should be 100V as this capacitor is exposed to the full supply range, VPP-VNN.

    - DO, fast recovery PN junction diodes minimize undershoots of the outputs with respect to

    power ground during switching transitions and abnormal load conditions such as output shorts to ground. For maximum effectiveness, these diodes must be located close to the output pins and returned to their respective VNN1(2). Please see Application/Test Circuit for ground return pin.

    - CFB removes very high frequency components from the amplifier feedback signals and

    lowers the output switching frequency by delaying the feedback signals. In addition, the value of CFB is different for channel 1 and channel 2 to keep the average switching frequency difference greater than 40kHz. This minimizes in-band audio noise.

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    14 TA2022 KLI/1.2/07-04

    - To minimize noise pickup and minimize THD+N, RFBC should be located as close to the TA2022 as possible. Make sure that the routing of the high voltage feedback lines is kept far away from the input op amps or significant noise coupling may occur. It is best to shield the high voltage feedback lines by using a ground plane around these traces as well as the input section.

    - CB provides high frequency bypassing for the bootstrap supplies. Very high currents are

    present on these supplies.

    - CSW provides high frequency bypassing for the VN10 generator circuit. Very high currents are present on these supplies.

    - CSWFB filters the feedback signal (VN10FDBK) for the hysteretic VN10 buck converter. The

    feedback signal is noise sensitive and the trace from CSWFB to VNN should be kept short.

    - DSW is the flywheel diode for the VN10 buck converter and prevents VN10SW(pin 5) from going more than one diode drop below VNN.

    In general, to enable placement as close to the TA2022, and minimize PCB parasitics, the capacitors listed above should be surface mount types, located on the solder side of the board. Some components are not sensitive to location but are very sensitive to layout and trace routing.

    - To maximize the damping factor and reduce distortion and noise, the modulator feedback connections should be routed directly to the pins of the output inductors, LO. This was done on the EB-TA2022 board.

    - The output filter capacitor, CO, and zobel capacitor, CZ, should be star connected with the

    load return. The output ground feedback signal should be taken from this star point. This is suggested by the routing on the Application/Test schematic, but, for space/layout reasons, this was not fully implemented on the EB-2022.

    - The modulator feedback resistors, RFBA and RFBB should all be grounded and attached to

    5V together. These connections will serve to minimize common mode noise via the differential feedback. Please refer to the EB-TA2022 evaluation board for more information.

    T A 2 0 2 2 G r o u n d i n g

    Proper grounding techniques are required to maximize TA2022 functionality and performance. Parametric parameters such as THD+N, Noise Floor and crosstalk can be adversely affected if proper grounding techniques are not implemented on the PCB layout. The following discussion highlights some recommendations about grounding both with respect to the TA2022 as well as general audio system design rules. The TA2022 is divided into two sections: the input section, which spans pin 15 through pin 32, and the output (high power) section, which spans pin 1 through pin 14. On the TA2022 evaluation board, the ground is also divided into distinct sections, one for the input and one for the output. To minimize ground loops and keep the audio noise floor as low as possible, the input and output ground must be only connected at a single point. Depending on the system design, the single point connection may be in the form of a ferrite bead or a PCB trace. The analog grounds, pin 15 and pin 20 must be connected locally at the TA2022 for proper device functionality. On the TA2022 evaluation board, Tripath has used an analog ground plane to minimize the impedances between pin 15 and pin 20 as well as the other analog ground connections, such as V5 supply bypassing, and feedback divider networks. The ground for the V5 power supply should connect directly to pin 20. Additionally, any external input circuitry such as preamps, or active filters, should be referenced to pin 20.

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    15 TA2022 KLI/1.2/07-04

    For the power section, Tripath has traditionally used a star grounding scheme. Thus, the load ground returns and the power supply decoupling traces are routed separately back to the power supply. In addition, any type of shield or chassis connection would be connected directly to the ground star located at the power supply. These precautions will both minimize audible noise and enhance the crosstalk performance of the TA2022. The TA2022 incorporates a differential feedback system to minimize the effects of ground bounce and cancel out common mode ground noise. As such, the feedback from the output ground for each channel needs to be properly sensed. This can be accomplished by connecting the output ground sensing trace directly to the star formed by the output ground return, output capacitor, CO, and the zobel capacitor, CZ. Refer to the Application / Test Circuit for a schematic description. Pin 3, VN10GND, is used for the VN10 buck converter. Pin 3 can be connected to the main power supply decoupling ground trace (or plane) without any loss in functionality or reduction of performance. This pin is electrically shorted to the copper heat sink (case) of the TA2022. Even if the internal VN10 regulator is not being used, VN10GND should still be connected to PGND.

    T A 2 0 2 2 A m p l i f i e r G a i n

    The gain of the TA2022 is the product of the input stage gain and the modulator gain. Please refer to the sections, Input Stage Design, and Modulator Feedback Design, for a complete explanation of how to determine the external component values.

    MODULATORV EVINPUTSTAG VTA2022 A* AA =

    +

    + 1

    R*R)R(R*R

    RRA

    FBBFBA

    FBBFBAFBC

    I

    F VTA2022

    For example, using a TA2022 with the following external components, RI = 20k

    RF = 20k RFBA = 1k RFBB = 1.13k

    RFBC = 9.09k

    VV18.13 1

    1.13k *1.0k )1.13k(1.0k *9.09k

    20k 20k AVTA2022 =

    +

    +

    I n p u t S t a g e D e s i g n

    The TA2022 input stage is configured as an inverting amplifier, allowing the system designer flexibility in setting the input stage gain and frequency response. Figure 1 shows a typical application where the input stage is a constant gain inverting amplifier. The input stage gain should be set so that the maximum input signal level will drive the input stage output to 4Vpp.

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    16 TA2022 KLI/1.2/07-04

    TA20222

    INPUT2OAOUT2

    V5

    OAOUT1

    +

    -CI

    +

    -

    INV1INPUT1

    BIASCAP

    AGND

    RF

    RICIRF

    AGND

    INV2

    V5

    RI

    F i g u r e 1 : I n p u t S t a g e

    The gain of the input stage, above the low frequency high pass filter point, is that of a simple inverting amplifier: It should be noted that the input opamps are biased at approximately 2.5VDC. Thus, the polarity of CI must be followed as shown in Figure 1 for a standard ground referenced input signal

    I

    F EVINPUTSTAG

    RRA =

    I n p u t C a p a c i t o r S e l e c t i o n

    CI can be calculated once a value for RI has been determined. CI and RI determine the input low frequency pole. Typically this pole is set below 10Hz. CI is calculated according to:

    IP

    IRf2

    1C

    =

    where: IR = Input resistor value in ohms.

    Pf = Input low frequency pole (typically 10Hz or below).

    M o d u l a t o r F e e d b a c k D e s i g n

    The modulator converts the signal from the input stage to the high-voltage output signal. The optimum gain of the modulator is determined from the maximum allowable feedback level for the modulator and maximum supply voltages for the power stage. Depending on the maximum supply voltage, the feedback ratio will need to be adjusted to maximize performance. The values of RFBA, RFBB and RFBC (see explanation below) define the gain of the modulator. Once these values are chosen, based on the maximum supply voltage, the gain of the modulator will be fixed even with as the supply voltage fluctuates due to current draw. For the best signal-to-noise ratio and lowest distortion, the maximum modulator feedback voltage should be approximately 4Vpp. This will keep the gain of the modulator as low as possible and still allow headroom so that the feedback signal does not clip the modulator feedback stage. Figure 2 shows how the feedback from the output of the amplifier is returned to the input of the modulator. The input to the modulator (FBKOUT1/FBKGND1 for channel 1) can be viewed as inputs to an inverting differential amplifier. RFBA and RFBB bias the feedback signal to approximately 2.5V and RFBC scales the large OUT1/OUT2 signal to down to 4Vpp.

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    17 TA2022 KLI/1.2/07-04

    OUT 1 GROUND

    RFBB

    FBKGND1

    RFBA

    OUT1

    RFBC

    V5

    RFBCFBKOUT1

    RFBB

    RFBA

    AGND

    1/2 TA2022

    Processing

    Modulation

    &

    F i g u r e 2 : M o d u l a t o r F e e d b a c k

    The modulator feedback resistors are: = 1Ktypically specified, UserRFBA

    4)-(VPPVPP*RR FBA FBB =

    4VPP*RR FBA FBC =

    1R*R

    )R(R*RAFBBFBA

    FBBFBAFBC MODULATOR-V +

    +

    The above equations assume that VPP=|VNN|. For example, in a system with VPPMAX=36V and VNNMAX=-36V, RFBA = 1k, 1% RFBB = 1.125k, use 1.13k, 1% RFBC = 9.0k, use 9.09k, 1% The resultant modulator gain is:

    18.13V/V 11.13k *1.0k

    )1.13k(1.0k *9.09k A MODULATOR-V =++

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    18 TA2022 KLI/1.2/07-04

    M u t e

    The mute pin must be driven to a logic low or logic high state for proper operation. The state of the mute pin is latched in to minimize the effects of noise on this pin, which could cause the TA2022 to switch state unintentionally. Controlling the mute pin with a push-pull output from a microcontroller, or a physical switch between V5 and AGND, works well as both solutions have low impedance drive capability. In some cases, it may be desirable to drive the mute pin with an alternative approach. When the device is in mute, the pin must be pulled low via approximately 1kohm to overcome the internal latch and change the TA2022 state (i.e. out of mute). When the device is not in mute, the mute pin must be pulled high via approximately 2kohm to overcome the internal latch and change the TA2022 state (i.e. into mute). Figure 3 shows a simple control circuit that buffers a Mute Control signal that is not capable of driving the Mute pin of the TA2022 directly. When the Mute Control signal is high, the Mute pin will be driven low and the TA2022 will be on. If the Mute Control signal is low, the 2k resistor will pull the Mute pin high and the TA2022 will be muted.

    F i g u r e 3 : L o w i m p e d a n c e d r i v e f o r M u t e P i n

    To ensure proper device operation, including minimization of turn on/off transients that can result in undesirable audio artifacts, Tripath recommends that the TA2022 device be muted prior to power up or power down of the 5V supply. The sensing of the V5 supply can be easily accomplished by using a microcontroller supervisor or equivalent to drive the TA2022 mute pin high when the V5 voltage is below 4.5V. This will ensure proper operation of the TA2022 input circuitry. A micro-controller supervisor such as the MCP101-450 from Microchip Corporation has been used by Tripath to implement clean power up/down operation. If turn-on and/or turn-off noise is still present with a TA2022 amplifier, the cause may be other circuitry external to the TA2022. While the TA2022 has circuitry to suppress turn-on and turn-off transients, the combination of power supply and other audio circuitry with the TA2022 in a particular application may exhibit audible transients. One solution that will completely eliminate turn-on and turn-off pops and clicks is to use a relay to connect/disconnect that amplifier from the speakers with the appropriate timing during power on/off.

    T A 2 0 2 2 O u t p u t C a p a b i l i t y The TA2022 can output two channels at 100 watts each into a 4ohm load at 1% THD+N. The maximum amplifier output power is determined by a number of factors including the TA2022 junction temperature, the load impedance and the power supply voltage. Tripath does not recommend driving loads below 4 ohm single ended as the amplifier efficiency will be seriously reduced and the amplifier may prematurely current limit.

    MUTE

    AGND

    TO MUTE(Pin 24)

    2k

    CONTROL

    10k

    V5

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    19 TA2022 KLI/1.2/07-04

    B r i d g i n g t h e T A 2 0 2 2

    The TA2022 can be bridged by returning the signal from OAOUT1 to the input resistor at INV2. OUT1 will then be a gained version of OAOUT1, and OUT2 will be a gained and inverted version of OAOUT1 (see Figure 3). When the two amplifier outputs are bridged, the apparent load impedance seen by each output is halved, so the minimum recommended impedance for bridged operation is 8 ohms. Due to the internal current limit setting, the maximum supply voltage recommended for bridged operation is +/-30V. Bridged operation into loads below 8ohms is possible, but, as mentioned above, the amplifier efficiency will be reduced and the amplifier may prematurely current limit. The TA2022 is capable of 150W into 8 ohms bridged at 0.1% THD+N.

    AGND

    OAOUT2

    V5

    OAOUT1

    AGND

    BIASCAP

    V5

    RI

    20k

    CIINV1

    RF

    TA2022

    INPUT

    INV2

    +

    -

    +

    -

    20k

    F i g u r e 4 : I n p u t S t a g e S e t u p f o r B r i d g i n g

    The switching outputs, OUT1 and OUT2, are not synchronized, so a common inductor may not be used with a bridged TA2022. For this same reason, individual zobel networks must be applied to each output to load each output and lower the Q of each common mode differential LC filter.

    O u t p u t V o l t a g e O f f s e t

    The output offset voltage of the TA2022 is largely determined by the matching of the respective RFBA, RFBB, and RFBC networks for FBKOUT1(FBKOUT2) and FBKGND1 (FBKOUT2). Thus, the intrinsic offset of the TA2022 can be altered by the external feedback network resistor matching. To minimize the nominal untrimmed offset voltage, 1% tolerance resistors are recommended. In most applications, the output offset voltage will need to be trimmed via an external circuit (either passive or active). The output offset voltage of the TA2022 can be nulled by modifying the modulator feedback as shown in Figure 4. Potentiometer ROFA is used to trim the effective resistance seen by the output ground, and therefore the output offset. ROFB limits the trim range.

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    20 TA2022 KLI/1.2/07-04

    F i g u r e 5 : M a n u a l O u t p u t O f f s e t T r i m C i r c u i t

    A DC servo can also be used to automatically null any offset voltage. The TA2022 evaluation board incorporates a DC servo. Please refer to the TA2022 evaluation board document, EB-TA2022, available on the Tripath website, at www.tripath.com.

    C u r r e n t P r o t e c t i o n D e s i g n

    Although the over-current (Ioc) trip point is internally fixed, there are external components that can affect output current levels during a short circuit event. Referring to the Application/Test diagram these include the output inductor (Lo), output diodes (Do) and supply bypassing (Cs). The two output inductors, Lo, directly impact the peak output current levels reached during short circuit events. For this reason they must be rated at least 10A regardless of the systems maximum operating current, ensuring the inductor does not saturate which could lead to switching currents passing unimpeded to the load. When this occurs, the Ioc may be forced to trip on average current levels instead of peak current levels, directly impacting the current levels seen by the load and potentially causing damage to the TA2022. Please refer to the Output Filter Design section for more details on the inductor requirements and figure 6 for comparison of load currents with an unsaturated and saturated inductor. The four output diodes, Do, minimize overshoots and undershoots with respect to the supply rails VPP and VNN. In order for these diodes to work properly they must have a low Forward Voltage rating at 10A. We recommend a Fast Recovery diode or Ultra-Fast PN Rectifier diode with a Vf rating of 1V at 10A, or better. Also, These should be placed close to their respective output pins to maximum effectiveness. The two bulk supply voltage capacitors, Cs, will absorb the current generated when the diodes (Do) conduct. We recommend that a high frequency capacitor designed for low impedance and high ripple currents be used. Tripath recommends an impedance of .1 or better at 100kHz and a ripple current rating of 1Arms at 100kHz. For maximum effectiveness, Tripath recommends that these capacitors be placed close to their respective supply pins.

    RFBC

    10K

    Processing

    RFBB

    &

    OUT1

    OUT 1 GROUND

    RFBB

    RFBC

    V5

    FBKOUT1

    AGND

    RFBA

    ROFB

    ROFA50K

    1/2 TA2022

    RFBA

    FBKGND1Modulation

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    21 TA2022 KLI/1.2/07-04

    F i g u r e 6 : S h o r t c i r c u i t l o a d c u r r e n t w i t h u n s a t u r a t e d t o r o i d a l i n d u c t o r a n d s a t u r a t e d b o b b i n I n d u c t o r ( s h i e l d e d )

    O u t p u t F i l t e r D e s i g n

    Tripath amplifiers generally have a higher switching frequency than PWM implementations allowing the use of higher cutoff frequency filters, reducing the load dependent peaking/drooping in the 20kHz audio band. This is especially important for applications where the end customer may attach any speaker to the amplifier (as opposed to a system where speakers are shipped with the amplifier), since speakers are not purely resistive loads and the impedance they present changes over frequency and from speaker model to speaker model. An RC network, or zobel (RZ, CZ) must be placed at the filter output to control the impedance seen by the TA2022. The TA2022 works well with a 2nd order, 107kHz LC filter with LO = 10uH and CO = 0.22uF and RZ = 6.2ohm/2W and CZ = 0.22uF. Some applications may require a more aggressive filter to reduce out of band noise. Below are some proven filter combinations: - 49.5kHz 2nd order filter - 33.6kHz 2nd order filter LO = 22uH LO = 33uH CO = .47uF CO = .68uF RZ = 8 RZ = 6.2 CZ = .47uF CZ = .68uF

    - 65kHz 4th order filter LO1 = 15uH

    CO1 = 1uF LO2 = 10uH CO2 = .22uF RZ = 10 CZ = .47uF Output inductor selection is a critical design step. The core material and geometry of the output filter inductor affects the TA2022 distortion levels, efficiency, over-current protection, power dissipation and EMI output. The inductor should have low loss at 700kHz with 80Vpp. It should be reiterated that regardless of the systems maximum operating current, a 10A rating is required to ensure that peak current conditions will not cause the inductor to saturate. During a short circuit event the inductor current increases very quickly in a saturated core (see figure 6), compromising the current protection scheme. A 10A rating is sufficient to ensure that current increases through the inductor are linear, and provides a safety margin for the TA2022. There are two types of inductors available in the 10A range that offers some EMI containment: they are the toroidal type and the bobbin (shielded) type inductor. In bobbin construction, a ferrite shield is placed around the core of a bobbin inductor to help contain radiated emissions. This shield can reduce the amount of energy the inductor can store in the core by reducing the air gap, which can lower the peak current capability of the inductor. Typically, a 7-10A shielded bobbin inductor will not have the peak current capability necessary to ensure that the core

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    22 TA2022 KLI/1.2/07-04

    will not saturate during short circuit events; this is why they are not recommended for use with the TA2022. Also it should be noted that shielded bobbin construction is not as effective as toroidal construction for EMI containment.

    Tripath recommends that the customer use a toroidal inductor with a Carbonyl-E core for all applications of the TA2022. This core has a high peak current capability due to its low- Carbonyl-E metal powder. A distributed air gap increases its energy storage capability, which allows for a small footprint and high current capability. Carbonyl-E toroidal iron powder cores have low loss and good linearity. The toroidal shape is ideal for EMI containment. Also, EMI can be further contained by sizing the toroid to accept a full layer of windings. This aids in shielding the electric field. Tripath recommends:

    - Micrometals (www.micrometals.com) Type-2 (Carbonyl-E) toroidal iron powder cores. The specific core Tripath initially verified and used on the EB-TA2022 was a T94-2 (23.9mm outer diameter) wound to 11uH with 19AWG wire. Since then Tripath has determined that much smaller Carbonyl-E toroids will not saturate during high current events. Tripath has also used T68-2 (17.5mm outer diameter) and the T60-2B/60(15.2mm outer diameter) cores wound to 11uH with 22AWG with good success. If a smaller core is required, core outer diameters as small as 15.2mm (T60-2) work well, but core temperature effects should be tested. The T60-2 core did not saturate during short circuit testing, but maximum core temperatures must be considered and multiple layer winding must be used to achieve 11uH. Multiple winding can increase winding capacitance, which may cause ringing and increased radiated emissions. Bank winding techniques can minimize this effect. It should be noted that at core temperatures above 130C the single build wire used by most inductor manufacturers should be replaced with a heavy build wire. Micrometals does not provide winding services, but many companies purchase directly from them and provide completely finished inductors. Pulse Engineering has assigned a part number for the T68-2 wound with 44 turns of 22 AWG single build wire. The part number is PA0291.

    - Amidon Inc./American Cores type-06 (Carbonyl-E) toroidal iron powder cores. Tripath has

    used T690-06 (17.5mm outer diameter) cores wound to 11uH with good success. Amidon carries type-06 cores in the 23.9mm to 15.2mm outer diameter range. They have assigned a part number for the T690-06 wound with 44 turns of 22 AWG single build wire. This part is approved by Tripath and is 690064422.

    P o w e r S u p p l i e s

    The TA2022 requires the split supply rails VPP1(VPP2) and VNN1(VNN2), and V5. It also uses some additional voltages, VN10, VBOOT1 and VBOOT2 that are generated internally. The selection of components for the switching regulator is shown in the Application / Test Diagram.

    M i n i m u m a n d M a x i m u m S u p p l y V o l t a g e O p e r a t i n g R a n g e

    The TA2022 can operate over a wide range of power supply voltages from +/-12V to +/-36V. In order to optimize operation for either the low or high range, the user must select the proper values for RVNNSENSE, RVPPSENSE, RFBA, RFBB, and RFBC. Please refer to the Modulator Feedback Design and Over/Under-voltage Protection sections for more additional information.

    V N 1 0 S u p p l y

    The TA2022 has an internal hysteretic buck converter, which, in conjunction with a few passive components, generates the necessary floating power supply for the MOSFET driver stage (nominally 11V with the external components shown in Application / Test Circuit). The performance curves shown in the data sheet as well as efficiency measurements were done using the internal VN10 generator. Tripath recommends that the internal VN10 generator be used. In some cases, though, a designer may wish to use an external VN10 generator. The specification for VN10 quiescent current (65mA typical, 80mA maximum) in the Electrical Characteristics section states the amount of current needed when an external floating supply is used. If the internal VN10

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    23 TA2022 KLI/1.2/07-04

    generator is not used, Tripath recommends shorting VN10SW(pin 5) to VN10GND(pin 3) and VN10FDBK(pin 14) to VN10GND(pin 3). VN10GND should still be connected to the system power (high current) ground star for noise reasons. The external VN10 supply must be able to source a maximum of 80mA into the VN10 pin. Thus, a positive supply must be used. In addition, this supply must be referenced to the VNN rail. If the external VN10 supply does not track fluctuations in the VNN supply or is not able to source current into the VN10 pin, the TA2022 will, at the very least, not work, but more likely, be permanently damaged. Figure 7 shows a simple circuit for an external VN10 supply. Though simple, there is one problem with this circuit; the maximum input voltage of the 7810. If the maximum input voltage of the 7810 is exceeded (typically this voltage is 35V), then the 7810 will be damaged which will likely cause damage to the TA2022. Thus, this circuit should only be used where the VNN power supply is well regulated even under heavy load conditions (including the effects of power supply pumping).

    F i g u r e 7 : S i m p l e E x t e r n a l V N 1 0 S u p p l y

    Figure 8 shows a much more robust VN10 supply. In this case, the maximum supply differential the LM317 experiences is the input voltage minus the output voltage. The maximum differential specification is 40V for the LM317. When used as the VN10 supply for the TA2022, the maximum differential the LM317 will experience is 25V, even at maximum operating voltage of 36V for the TA2022. As configured, VOUT will be about 10.63V above VNN.

    F i g u r e 8 : R o b u s t E x t e r n a l V N 1 0 S u p p l y

    It should be noted that the maximum power dissipation for either Figure 6 or Figure 7 is:

    1.6W80mA(max.)11V)(31VIOUTVOUT)(VINP DMAX == Thus, the LM7810 or LM317 must be sufficiently heat sinked to sustain 80mA in the system ambient temperature. In the case where multiple TA2022s are run off the same VN10 generator, the power dissipation may be prohibitively large for the linear regulator in conjunction with allowable heat sink. In these cases, a more sophisticated scheme using an additional transformer secondary winding referenced to VNN may be necessary to minimize the linear regulator power dissipation.

    VIN TO VN10 (PIN 2)VOUT

    0.1uF 0.1uF10uF7810

    "GND"

    TO POWER SUPPLY"STAR" GND 10uF

    TO VNN (PIN 8,9)

    ADJ0.1uF

    VIN

    240

    VOUT

    10uF

    1.8k

    10uF

    TO POWER SUPPLY"STAR" GND

    TO VNN (PIN 8,9)

    0.1uF

    TO VN10 (PIN 2)

    10uF

    LM317

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    24 TA2022 KLI/1.2/07-04

    P r o t e c t i o n C i r c u i t s

    The TA2022 is guarded against over-current, over / under-voltage and over-temperature conditions. If the device goes into an over-current or over / under-voltage condition, the HMUTE goes to a logic HIGH indicating a fault condition. When this occurs, the amplifier is muted, all outputs are TRI-STATED, and will float to approximately 2.5VDC.

    O v e r - C u r r e n t P r o t e c t i o n

    An over-current fault occurs if more than approximately 8 amps of current flows from any of the amplifier output pins. This can occur if the speaker wires are shorted together or if one side of the speaker is shorted to ground. An over-current fault sets an internal latch that can only be cleared if the MUTE pin is toggled or if the part is powered down. See the over-current curves in the Typical Characteristics section for more information.

    O v e r / U n d e r V o l t a g e P r o t e c t i o n

    The TA2022 has built-in over and under voltage protection for both the VPP and VNN supply rails. The nominal operating voltage will typically be chosen as the supply center point. This allows the supply voltage to fluctuate, both above and below, the nominal supply voltage. VPPSENSE (pin 19) performs the over and undervoltage sensing for the positive supply, VPP. VNNSENSE (pin 18) performs the same function for the negative rail, VNN. In the simplest implementation, the supply is done via a single, external resistor per sense pin. This scheme is referred to as the single resistor sense circuit. Figure 9 shows the single resistor sense circuit.

    F i g u r e 9 : S i n g l e R e s i s t o r S e n s e C i r c u i t

    When the current through RVPPSENSE (or RVNNSENSE) goes below or above the values shown in the Electrical Characteristics section (caused by changing the power supply voltage), the TA2022 will be muted. VPPSENSE is internally biased at 2.5V and VNNSENSE is biased at 1.25V. For the single resistor sense case (as shown in the Application / Test Diagram), these bias points must be taken into consideration when calculating the RVPPSENSE or RVNNSENSE resistor. Once the supply comes back into the supply voltage operating range (as defined by the supply sense resistors), the TA2022 will automatically be unmuted and will begin to amplify. There is a hysteresis range on both the VPPSENSE and VNNSENSE pins. If the amplifier is powered up in the hysteresis band the TA2022 will be muted. Thus, the usable supply range is the difference between the over-voltage turn-off and under-voltage turn-off for both the VPP and VNN supplies. It should be noted that there is a timer of approximately 200mS with respect to the over and under voltage sensing circuit.

    VPP

    19

    TA2022

    18

    VNN

    RVPPSENSE

    VNNSENSE

    VPPSENSE

    RVNNSENSE

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    25 TA2022 KLI/1.2/07-04

    Thus, the supply voltage must be outside of the user defined supply range for greater than 200mS for the TA2022 to be muted. The equation for calculating RVPPSENSE is as follows:

    VPPSENSE VPPSENSE

    I2.5V - VPPR =

    The equation for calculating RVNNSENSE is as follows:

    VNNSENSE VNNSENSE

    I VNN-1.25V R =

    where IVPPSENSE or IVNNSENSE can be any of the currents shown in the Electrical Characteristics table for VPPSENSE and VNNSENSE, respectively.

    Example: Nominal supply voltage +/-32.5V +/-10% From this information, a value of RVPPSENSE and RVNNSENSE can be calculated using the above formulas.

    36V use35.75V1.132.5VVPP MAX ==

    36V- use35.75V1.132.5VVNN MAX ==

    %1,k243use242.75A138

    2.5V -36V R VPPSENSE == k

    where IVPPSENSE is the minimum over-voltage turn off current for VPPSENSE.

    %1,k249use245.1A15236V- -1.25V R VNNSENSE == k

    where IVNNSENSE is the minimum over-voltage turn off current for VNNSENSE.

    Using the resistor values from above, the actual minimum over voltage turn off points will be:

    36.03V2.5V138k243VPP N_OFFMIN_OV_TUR =+= A 36.60V-A152k4921.25VVNN N_OFFMIN_OV_TUR ==

    The other three trip points can be calculated using the same formula but inserting the appropriate IVPPSENSE (or IVNNSENSE) current value. As stated earlier, the usable supply range is the difference between the minimum overvoltage turn off and maximum under voltage turn-off for both the VPP and VNN supplies.

    N_OFFMAX_UV_TURN_OFFMIN_OV_TUR RANGE VPP-VPPVPP = N_OFFMAX_UV_TURN_OFFMIN_OV_TUR RANGE VNN-VNNVNN =

    Using the resistor values from above, and the maximum under voltage trip currents shown in the Electrical Characteristics table, the maximum under voltage turn off points will be:

    23.64V2.5V87k243VPP N_OFFMAX_UV_TUR =+= A 22.41V-A95k4921.25VVNN N_OFFMAX_UV_TUR ==

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    26 TA2022 KLI/1.2/07-04

    and the resultant supply ranges will be:

    12.39V23.64V03.36VPP RANGE == 14.19V-22.41V60.36VNN RANGE ==

    It should also be noted that the tolerance of the RVPPSENSE (or RVNNSENSE) resistors will effect the trip voltages and thus, the usable supply range. To minimize the additional variance Tripath recommends 1% tolerance resistors. As a matter of completeness, the formulas below include the effect of resistor tolerance assuming a known value of RVPPSENSE or RVNNSENSE.

    2.5V)100/TOL1()IR(VPP TRIPVPPSENSE N_OFFMIN_OV_TUR ++= 2.5V)100/TOL1()IR(VPP TRIPVPPSENSE N_OFFMAX_UV_TUR ++=

    )100/TOL1()IR(25.1VNN TRIPVNNSENSE N_OFFMIN_OV_TUR += )100/TOL1()IR(25.1VNN TRIPVNNSENSE N_OFFMAX_UV_TUR +=

    Using a value of 243k for RVPPSENSE and a value of 249k for RVNNSENSE, assuming 5% tolerance, along with the appropriate value of ITRIP, the trip voltages and supply ranges can be calculated.

    34.44V 2.5V)100/51()138243k (VPP N_OFFMIN_OV_TUR =++= A 24.70V 2.5V)100/51()A87243k (VPP N_OFFMAX_UV_TUR =++=

    9.74V24.70V44.34VPP RANGE ==

    34.80V)100/51()A152k492(25.1VNN N_OFFMIN_OV_TUR =+= 23.59V)100/51()A95k492(25.1VNN N_OFFMAX_UV_TUR =+=

    11.21V-23.59V80.34VNN RANGE ==

    Thus, by using 5% resistors, the supply range for the VPP has been reduced by 2.65V while the VNN range has been reduced by approximately 3.0V (as compared to resistors with no tolerance variation). In actuality, if a 5% resistor was to be used, then the initial value of RVPPSENSE and RVNNSENSE would have had to be adjusted such that the minimum over voltage turn off points would have never been less than +/-36V as defined by the supply voltage and tolerance specification. It should be noted that the values for VVPPSENSE and VVNNSENSE shown in the Electrical Characteristics table were calculated using a value of 249k for both RVPPSENSE and RVNNSENSE. In addition, for the maximum and minimum values, as opposed to the typical ones, a 1% tolerance resistor value around 249k was chosen to show the effect on supply range. Thus, the minimum and maximum values would be worst case assuming a supply voltage of 5V for the input section of the TA2022. The entire discussion thus far has been for the one resistor sense circuit. This configuration requires a single resistor from either VPPSENSE or VNNSENSE to the respective power supply. While the simplest configuration, in terms of external components, there are some drawbacks to this configuration. The first drawback is that the range for VPPRANGE and VNNRANGE are asymmetric due to the different internal bias voltages of VPPSENSE and VNNSENSE. A second issue is that current through RVPPSENSE or RVNNSENSE will change if the V5 voltage is not exactly 5V, since the bias voltages of pin 18 and pin 19 are set by resistor dividers between V5 and AGND. With an additional resistor per supply sense pin (2 resistors per VPPSENSE or VNNSENSE), the drawbacks of the one resistor sense circuit can be eliminated. In addition, the calculations of the sense resistors are actually more straightforward for the two resistor sense circuit as opposed to the one resistor scheme. Figure 10 shows the proper connection for the two resistor sense circuit for both the VPPSENSE and VNNSENSE pins.

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    27 TA2022 KLI/1.2/07-04

    F i g u r e 1 0 : T w o R e s i s t o r S e n s e C i r c u i t

    The equation for calculating RVPP1 is as follows:

    VPPSENSE VPP1

    I VPPR =

    Set VPP1 VPP2 RR = .

    The equation for calculating RVNNSENSE is as follows:

    VNNSENSE VNN1

    IVNNR =

    Set VNN1 VNN2 R3R = .

    IVPPSENSE or IVNNSENSE can be any of the currents shown in the Electrical Characteristics table for VPPSENSE and VNNSENSE, respectively.

    Example: Nominal supply voltage +/-32.5V +/-10% From this information, a value of RVPP1, RVPP2, RVNN1, and RVNN2 can be calculated using the above formulas.

    36V use35.75V1.132.5VVPP MAX ==

    36V- use35.75V1.132.5VVNN MAX ==

    %1,k261use260.87A138

    36VR VPP1 == k

    Set %1,261 RVPP2 = k . where IVPPSENSE is the minimum over-voltage turn off current for VPPSENSE.

    %1,k237use236.84A152

    36VR VNN2 == k

    Set %1,k715 RVNN2 = . where IVNNSENSE is the minimum over-voltage turn off current for VNNSENSE.

    RVNN2

    TA2022RVNN1

    RVPP1

    VPP

    18 VNNSENSE

    VPPSENSE

    RVNN2

    V5 VNN

    V5

    19

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    28 TA2022 KLI/1.2/07-04

    The two additional resistors, RVPP2 and RVNN2 compensate for the internal bias points. Thus, RVPP1 and RVNN1 can be used for the direct calculation of the actual VPP and VNN trip voltages without considering the effect of RVPP2 and RVNN2. Using the resistor values from above, the actual minimum over voltage turn off points will be:

    36.02V138k261VPP N_OFFMIN_OV_TUR == A 36.24V-A152k372VNN N_OFFMIN_OV_TUR ==

    The other three trip points can be calculated using the same formula but inserting the appropriate IVPPSENSE (or IVNNSENSE) current value. As stated earlier, the usable supply range is the difference between the minimum overvoltage turn off and maximum under voltage turn-off for both the VPP and VNN supplies.

    N_OFFMAX_UV_TURN_OFFMIN_OV_TUR RANGE VPP-VPPVPP = N_OFFMAX_UV_TURN_OFFMIN_OV_TUR RANGE VNN-VNNVNN =

    Using the resistor values from above, and the maximum under voltage trip currents shown in the Electrical Characteristics table, the maximum under voltage turn off points will be:

    22.71V87k261VPP N_OFFMAX_UV_TUR == A 22..51V-A95k372 VNN N_OFFMAX_UV_TUR ==

    and the resultant minimum supply ranges will be:

    13.31V22.71V02.36VPP RANGE == 13.73V-22.51V24.36VNN RANGE ==

    By adding a total of 2 additional resistors (1 for VPPSENSE and 1 for VNNSENSE), the minimum supply range is now about 3% different as opposed to 13% for the single resistor sense case. In addition, the VPP range has been increased by nearly one volt. This represents a 7% improvement in supply range for VPP. As in the single resistor case, the tolerance of the RVPP1 and RVPP2 (or RVNN1 and RVNN2) resistors will affect the trip voltages and thus, the usable supply range. As the nominal supply voltage is decreased, the effect of RVPP2 and RVNN2 becomes more pronounced. To minimize the additional variance Tripath recommends 1% tolerance resistors. It is possible to calculate the effect of resistor tolerances for the two resistor sense circuit, but ultimately, 1% resistors should be used for the sense circuit in all, but the tightest regulated supply schemes.

    O v e r T e m p e r a t u r e P r o t e c t i o n An over-temperature fault occurs if the junction temperature of the part exceeds approximately 165C. The thermal hysteresis of the part is approximately 30C, therefore the fault will automatically clear when the junction temperature drops below 135C.

    H M U T E ( p i n 3 2 )

    The HMUTE pin is a 5V logic output that indicates various fault conditions within the device. These conditions include: over-current, overvoltage and undervoltage. The HMUTE output is capable of directly driving an LED through a series 2k resistor.

    H e a t S i n k R e q u i r e m e n t s

    most applications it will be necessary to fasten the TA2022 to a heat sink. The determining factor is that the 150C maximum junction temperature, TJ(max) cannot be exceeded, as specified by the following equation:

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    29 TA2022 KLI/1.2/07-04

    PDISS = ( )

    JA

    A)MAX(J TT

    where: PDISS = maximum power dissipation TJMAX = maximum junction temperature of TA2022 TA = operating ambient temperature JA = junction-to-ambient thermal resistance JA = JC + CS + SA

    Example: What size heat sink is required to operate the TA2022 at 80W per channel continuously in a 70C ambient temperature? PDISS is determined by:

    Efficiency = = IN

    OUT

    PP

    = DISSOUT

    OUT

    PPP

    PDISS (per channel) = W88.519085.0

    90== OUT

    OUT PP

    Thus, PDISS for two channels = 31.76W

    JA = ( )

    DISS

    A)MAX(J

    PTT =

    76.3170150 = 2.52C/W

    The JC of the TA2022 is 1.0C/W, so a heat sink of 1.32C/W is required for this example (assuming a CS = 0.2C/W). In actual applications, other factors such as the average PDISS with a music source (as opposed to a continuous sine wave) and regulatory agency testing requirements will help determine the size of the heat sink required.

    P e r f o r m a n c e M e a s u r e m e n t s o f t h e T A 2 0 2 2

    The TA2022 operates by generating a high frequency switching signal based on the audio input. This signal is sent through a low-pass filter (external to the Tripath amplifier) that recovers an amplified version of the audio input. The frequency of the switching pattern is spread spectrum in nature and typically varies between 100kHz and 1MHz, which is well above the 20Hz 20kHz audio band. The pattern itself does not alter or distort the audio input signal, but it does introduce some inaudible components. The measurements of certain performance parameters, particularly noise related specifications such as THD+N, are significantly affected by the design of the low-pass filter used on the output as well as the bandwidth setting of the measurement instrument used. Unless the filter has a very sharp roll-off just beyond the audio band or the bandwidth of the measurement instrument is limited, some of the inaudible noise components introduced by the TA2022 amplifier switching pattern will degrade the measurement. One feature of the TA2022 is that it does not require large multi-pole filters to achieve excellent performance in listening tests, usually a more critical factor than performance measurements. Though using a multi-pole filter may remove high-frequency noise and improve THD+N type measurements (when they are made with wide-bandwidth measuring equipment), these same filters degrade frequency response. The TA2022 Evaluation Board uses the Application/Test Circuit of this data sheet, which has a simple two-pole output filter and excellent performance in listening tests. Measurements in this data sheet were taken using this same circuit with a limited bandwidth setting in the measurement instrument.

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    30 TA2022 KLI/1.2/07-04

    P A C K A G E I N F O R M A T I O N 3 2 - P i n S S I P P a c k a g e

  • T r i p a t h T e c h n o l o g y, I n c . - T e c h n i c a l I n f o r m a t i o n

    31 TA2022 KLI/1.2/07-04

    Tripath Technology, Inc. reserves the right to make changes without further notice to any products herein to improve reliability, function or design. Tripath does not assume any liability arising out of the application of use of any product or circuit described herein; neither does it convey any license under its patent rights nor the rights of others. Tripath and Digital Power Processing are trademarks of Tripath Technology. Other trademarks referenced in this document are owned by their respective companies. TRIPATHS PRODUCT ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN CONSENT OF THE PRESIDENT OF TRIPATH TECHONOLOGY, INC. As used herein:

    1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, and whose failure to perform, when properly used in accordance with instructions for use provided in this labeling, can be reasonably expected to result in significant injury of the user.

    2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness.

    Contact In format ion T R I P A T H T E C H N O L O G Y , I N C

    2560 Orchard Parkway, San Jose, CA 95131 408.750.3000 - P 408.750.3001 - F

    For more Sales Information, please visit us @ www.tripath.com/cont_s.htm For more Technical Information, please visit us @ www.tripath.com/data.htm