eldo rf user's manual - bme eet

Eldo RF User’s Manual

Release AMS 2009.2

© 1999 - 2009 Mentor Graphics CorporationAll rights reserved.

This document contains information that is proprietary to Mentor Graphics Corporation. The original recipient of thisdocument may duplicate this document in whole or in part for internal business purposes only, provided that this entirenotice appears in all copies. In duplicating any part of this document, the recipient agrees to make every reasonableeffort to prevent the unauthorized use and distribution of the proprietary information.

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Eldo RF User’s Manual, AMS 2009.2 3

Table of Contents

Chapter 1Introduction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

Steady-State Analysis Types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17Steady-State Analysis for Non-Autonomous Circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17Steady-State Analysis for Autonomous Circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

Analyses Based on Steady-State Analysis. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18Parametric Steady-State Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18Local Stability Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19Steady-State AC Analysis (SSTAC). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19Steady-State Transfer Function Analysis (SSTXF) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19Steady-State Noise Analysis (SSTNOISE). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19

Modulated Steady-State Analysis Types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20Modulated Steady-State Analysis for Non-Autonomous Circuits . . . . . . . . . . . . . . . . . . . 20Modulated Steady-State Analysis for Autonomous Circuits . . . . . . . . . . . . . . . . . . . . . . . 20

Eldo RF Algorithms. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21Algorithm for Modulated Steady-State Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21

Optimization Capability. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22Verilog-A . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22Questa ADMS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23CommLib RF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23Eldo RF Multi-Threading . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23Invoking Eldo RF. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23

64-bit Mode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24Related Documentation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25

Chapter 2Analysis Command Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27Analysis Command Descriptions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

.SST. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29DC Operating Point Calculation for RF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31.SST OSCIL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33Steady-State Oscillator Algorithm Details . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35Digital Block Analysis During Steady-State Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37Pre-Transient Phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37.SST PLL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39.SST STABIL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43.SSTAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44.SSTXF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46.SSTNLCONTRIB . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47.SSTSENSRLC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50

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4 Eldo RF User’s Manual, AMS 2009.2

.SSTNOISE. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53

.SNF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57

.WCASE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 60

.MC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61

.MODSST . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

.PART MODSST . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 64

.RFBLOCK . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67

.CHRSIM . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70

.AGE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73

.OP RF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74

Chapter 3Display Command Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75

Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75Display Command Descriptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

.PLOT/.PRINT FSST/TSST . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76

.PLOT/.PRINT SSTAC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 78

.PLOT/.PRINT SSTXF. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80

.PLOT/.PRINT SSTNOISE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81

.PLOT/.PRINT SSTJITTER . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 88

.PLOT/.PRINT SSTSTABIL . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89

.PLOT/.PRINT FMODSST/TMODSST. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 92

.PLOT CONTOUR . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99

.EXTRACT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 101Specific RF Pre-Defined Functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102Adjacent Channel Power Ratio (ACPR). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103Two-Port Gain Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 104Two-Port Stability Factors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 107Two-Port Noise Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 108Two-Port Noise Circles. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111Two-Port Constant Gain Circles. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111Two-Port Stability Circles . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 112Local Stability Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113Steady-State Jitter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113

Chapter 4Sources . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115Steady-State Sources . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 116

Multi-Tone Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 116Probe Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 119Phase Noise Sources . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 121Correlation Between Noise Sources . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 123Source Usage in Pre-Transient Phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124Voltage Controlled Oscillator Source. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 125

Digitally Modulated Sources . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 127Baseband Sources . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 128GMSK (Gaussian Minimum Shift Keying) Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 130

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GFSK (Gaussian Frequency Shift Keying) Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 131OQPSK (Offset Quaternary Phase Shift Keying) Source. . . . . . . . . . . . . . . . . . . . . . . . . . 132PI4QPSK (pi/4 Quaternary Phase Shift Keying) Source . . . . . . . . . . . . . . . . . . . . . . . . . . 133MPSK (M-ary Phase Shift Keying) Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 134MFSK (M-ary Frequency Shift Keying) Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 135MQAM (M-ary Quadrature Amplitude Modulator) Source. . . . . . . . . . . . . . . . . . . . . . . . 136IQMOD (I-Q Modulator) Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 137EDGE (8PSK Modulated Source with 3pi/8 SymbolRotation for EDGE Standard) Source. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 139OFDM (Orthogonal Frequency Division Multiplexing) Source . . . . . . . . . . . . . . . . . . . . 140HPSK (Hybrid Phase Shift Keying) Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 142CCK (Complementary Code Keying) Source. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 146ZigBee (IEEE 802.15.4 standard for WPAN) Source . . . . . . . . . . . . . . . . . . . . . . . . . . . . 147Modulation Signal (PATTERN). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 148Filtering Information. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 150

Chapter 5Command Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 153

Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 153Eldo RF Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 156

Time Domain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 156Accuracy . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 157Convergence . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 160Frequency Tolerance Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 165Noise Result Presentation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 166Multi-Threaded Simulation Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 168MODSST Analysis Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 168Miscellaneous . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 171

Chapter 6Additional Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175

Special Devices . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175DC Cut . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175DC Feed . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175

RF to Digital and Digital to RF Converters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 175RF to Analog and Analog to RF Converters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 177RF Envelope Detector . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 178Flexible Frequency Divider Macromodel . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 179Save and Restart Capabilities. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 181

Pre-Transient Phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 181Steady-State Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 182Steady-State Noise Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 182Modulated Steady-State Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 183

Automated Sweeps. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 183

Chapter 7Convergence Troubleshooting . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 185

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 185

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Divergence Types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 185Type_1A . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 185Type_1B . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 186Type_2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 186

Circuit Types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 186Troubleshooting Forced Circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 187

Single-Tone Circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 187Multi-Tone Circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 189

Troubleshooting Autonomous Circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 190High Q Oscillators . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 190Very Non-Linear Oscillators. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 190

Chapter 8Working with S, Y, Z Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193Simulation Setup for S, Y, Z Parameter Extraction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193S, Y, Z Parameter Extraction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 195

Multitone and Autonomous Large-Signal S-Parameter Extraction . . . . . . . . . . . . . . . . . . 195For S-Parameter Extraction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 195For Mixed Mode S-Parameter Extraction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 196For Y-Parameter Extraction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 197For Z-Parameter Extraction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 197

Matrix Parameter Extraction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 197For G-Parameter Extraction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 197For H-Parameter Extraction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 198For T-Parameter Extraction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 198For A-Parameter Extraction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 198

Output File Specification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 198Simulating a Block Defined by its S-Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 200Instantiating a Block Defined by S-Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 201

Touchstone Data Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 205Mixed Mode S-Parameter Extraction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 206

Chapter 9Transmission Lines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 209

Lossy Transmission Line . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 209Level 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 209Level 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 211Level 3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 213Level 4 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 214LDTL Model Error Message Treatment . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 216Technical Precision. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 217Lossy Transmission Line: W Model. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 225RLGC File Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 227RLGC Model Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 230Tabular RLGC Model Syntax. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 232RLGC Model Error Message Treatment. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 235Lossy Transmission Line: U Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 236

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Model Syntax . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 237U Model Error Message Treatment . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 240

Chapter 10Microstrip and Stripline Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 241

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 241Microstrip Discontinuity Models. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 242

MTEE—Microstrip T Junction. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 242MBEND—Microstrip Bend (Arbitrary Angle, Optimally Mitered). . . . . . . . . . . . . . . . . . 245MBEND2—90-degree Microstrip Bend (Mitered) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 248MBEND3—90-degree Microstrip Bend (Optimally Mitered) . . . . . . . . . . . . . . . . . . . . . . 251MCORN—90-degree Microstrip Bend (Unmitered) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 254MSTEP—Microstrip Step in Width . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 257VIA2—Cylindrical Via Hole in Microstrip . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260

Stripline Discontinuity Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 263SBEND—Unmitered Stripline Bend . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 263STEE—Stripline T Junction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 266SSTEP—Stripline Step in Width . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 269

Chapter 11Behavioral Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 271

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 271Using CommLib RF Verilog-AMS with ADMS RF. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 272Using CommLib RF Verilog-AMS with Eldo RF Verilog-A . . . . . . . . . . . . . . . . . . . . . . . . 272

Chapter 12Eldo RF and Verilog-A. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 275

Eldo RF and the Verilog-A Compiler . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 275Including Verilog-A Modules in an Eldo RF Netlist. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 275

Compiling Verilog-A Source Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 276Instantiating Verilog-A Modules . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 277Referencing Verilog-A Compiled Files . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 278Example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 280

Restrictions Related to RF Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 281RF Modeling Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 282

Chapter 13Eldo RF and Questa ADMS. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 283

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 283Restrictions when using Eldo RF with Questa ADMS. . . . . . . . . . . . . . . . . . . . . . . . . . . . 284Circuit Partitioning . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 284RF Modeling Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 285

Using Eldo RF from the Questa ADMS GUI . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 285Eldo Commands Dialog . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 285

VHDL-AMS RF Subset. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 288

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Chapter 14Eldo RF Tutorials . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 293

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 293Tutorial #1—Single-Tone Steady-State Analysis and Two-Port Noise Extraction for an

Amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 294Tutorial #2—Power Efficiency and 1dB Compression Point Extraction for an Amplifier. . 301Tutorial #3—Multi-Tone Analysis, and IM3 and IP3 Extraction for an Amplifier . . . . . . . 306Tutorial #4—S-Parameter Extraction for an Amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 310Tutorial #5—Mixer Steady-State and Noise Analysis for a Gilbert Cell . . . . . . . . . . . . . . . 317Tutorial #6—Mixer Steady-State AC Analysis for a Gilbert Cell. . . . . . . . . . . . . . . . . . . . . 322Tutorial #7—Mixer Steady-State TF Analysis for a Gilbert Cell . . . . . . . . . . . . . . . . . . . . . 325Tutorial #8—Steady-State Analysis for a Frequency Divider. . . . . . . . . . . . . . . . . . . . . . . . 331Tutorial #9—Steady-State Analysis of Autonomous Circuit for an Oscillator . . . . . . . . . . . 335Tutorial #10—Sweeping the Oscillator Frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 340Tutorial #11—Phase Noise Extraction for an Oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . . 343Tutorial #12—Digitally Modulated Sources . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 346Tutorial #13—ACPR Computations for an Amplifier. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 353Tutorial #14—Verilog-A Usage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 357Tutorial #15—NPR Computation with Steady-State Analysis for an Amplifier . . . . . . . . . 362Tutorial #16—NPR Computation with Modulated Steady-State Analysis for an Amplifier 367Tutorial #17—EVM and BER Computations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 372Tutorial #18—Load Pull Contours . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 379Tutorial #19—SST Simulation using the .RFBLOCK Command . . . . . . . . . . . . . . . . . . . . 386Tutorial #20—SST Simulation using the .SSTNLCONTRIB Command . . . . . . . . . . . . . . . 391Tutorial #21—Multitone Large Signal S Parameters Extraction. . . . . . . . . . . . . . . . . . . . . . 399

Chapter 15Eldo RF Tutorial—Mixer Simulations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 403

Objective . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 403Part I. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 403

Architecture. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 403The Input Signals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 405Steady-State Analysis—A bit of Theory . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 406Intermodulation Products . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 406Truncation Models . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 407Steady State Analysis Parameters. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 408Conclusion of Part I . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 412

Part II . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 413Simulation of Third-Order Intermodulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 413Three-Tone Steady-State Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 413Input Signals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 414Automatic Measurement of the IM3. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 417Automatic Measurement of the Third-Order Intercept Point (IP3) . . . . . . . . . . . . . . . . . . 418Parametric (Sweep) Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 423Conclusion of Part II. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 424

Part III . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 425Simulation of the Mixer Noise—Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 425Simulation of the Mixer Output Noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 425

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Sorted Contributions of the Devices . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 426Noise Figure and Noise Factor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 428Computing the Sensitivity . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 431Computing the Spurious Free Dynamic Range . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 432Conclusion of Part III . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 435References. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 435

Chapter 16Eldo RF Tutorial—Port Impedance and Admittance Modeling . . . . . . . . . . . . . . . . . . . . 437

Objective . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4371—Converting Port Impedance into

Thévenin Equivalent Circuit (AC Analysis) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4372—Converting Port Impedance into

Thévenin Equivalent Circuit (SST Analysis) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4383—Converting Port Admittance into

Norton Equivalent Circuit (ac-domain) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4394—Converting Port Admittance into

Norton Equivalent Circuit (SST Analysis) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4405—Input Impedance and Admittance of a LNA . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4406—How to Save S-Parameters into a File? . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4427—How to use a Saved S-Parameter File as Part of the Circuit? . . . . . . . . . . . . . . . . . . . . . 443Further Information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 443

Chapter 17ADMS RF Tutorial—AGC Loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 445

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 445Differential Logarithmic Mixers. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 446Analog Processing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 446Digital Processing . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 447

Netlist Explanation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 447Simulation with ADMS RF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 448Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 449

Interactive Mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 449Batch Mode. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 452

Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 452Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 455

Chapter 18ADMS RF Tutorial—ZigBee Chain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 457

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 457Overview . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 457ZigBee Modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 458

Principles of the 802.15.4 PHY Level 2.4 GHz Band . . . . . . . . . . . . . . . . . . . . . . . . . . . . 458Description of ZigBee Modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 459

Power Amplification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 461Transmission . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 462

Phase Noise Addition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 462Channel and Multipath Modelization . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 462

Table of Contents


ZigBee Demodulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 463OQPSK Demodulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 463Chips Bufferization. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 463Chip to Symbol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 464Symbol to Serial Bit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 464

BER Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 464Detailed Netlist Explanation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 465Running the Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 467Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 470

Constellation Plots . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 473BER Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 475

Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 476

Glossary

Index

End-User License Agreement


List of Figures

Figure 3-1. Spectral density . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 83Figure 3-2. Example Plot for SSTSTABIL NET_POLE. . . . . . . . . . . . . . . . . . . . . . . . . . . . 90Figure 3-3. Example Plot for SSTSTABIL Q_FACTOR . . . . . . . . . . . . . . . . . . . . . . . . . . . 91Figure 3-4. Eye Diagram Tool dialog box . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94Figure 3-5. EZwave window—eye diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 95Figure 3-6. EZwave window—trajectory diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 96Figure 3-7. EZwave window—constellation diagram. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 97Figure 3-8. Adjacent Channel Power Ratio . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 104Figure 4-1. Phase Noise . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122Figure 9-1. Microstrip Line Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220Figure 9-2. Covered Pair Microstrip Line Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220Figure 9-3. Stripline Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220Figure 10-1. Microstrip T Junction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 242Figure 10-2. Microstrip T Junction Symbol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 242Figure 10-3. Equivalent circuit Microstrip T Junction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 243Figure 10-4. Microstrip Bend (Arbitrary Angle, Optimally Mitered) . . . . . . . . . . . . . . . . . . 245Figure 10-5. Microstrip Bend (Arbitrary Angle, Optimally Mitered) Symbol . . . . . . . . . . . 245Figure 10-6. Equivalent circuit Microstrip Bend . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .(Arbitrary Angle, Optimally Mitered) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 246Figure 10-7. 90-degree Microstrip Bend (Mitered) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 248Figure 10-8. 90-degree Microstrip Bend (Mitered) Symbol . . . . . . . . . . . . . . . . . . . . . . . . . 248Figure 10-9. Equivalent circuit MBEND2. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 249Figure 10-10. 90-degree Microstrip Bend (Optimally Mitered) . . . . . . . . . . . . . . . . . . . . . . 251Figure 10-11. 90-degree Microstrip Bend (Optimally Mitered) Symbol . . . . . . . . . . . . . . . 251Figure 10-12. Equivalent Circuit MBEND3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 252Figure 10-13. 90-degree Microstrip Bend (Unmitered). . . . . . . . . . . . . . . . . . . . . . . . . . . . . 254Figure 10-14. 90-degree Microstrip Bend (Unmitered) Symbol . . . . . . . . . . . . . . . . . . . . . . 254Figure 10-15. Equivalent circuit Microstrip corner . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 255Figure 10-16. Microstrip Step in Width. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 257Figure 10-17. Microstrip Step in Width Symbol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 257Figure 10-18. Equivalent circuit of a microstrip step in width . . . . . . . . . . . . . . . . . . . . . . . 258Figure 10-19. Cylindrical Via Hole in Microstrip . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260Figure 10-20. Cylindrical Via Hole in Microstrip Symbol . . . . . . . . . . . . . . . . . . . . . . . . . . 260Figure 10-21. Equivalent circuit Cylindrical Via Hole in Microstrip . . . . . . . . . . . . . . . . . . 262Figure 10-22. Unmitered Stripline Bend . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 263Figure 10-23. Unmitered Stripline Bend Symbol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 263Figure 10-24. Equivalent circuit of an unmitered stripline bend . . . . . . . . . . . . . . . . . . . . . . 264Figure 10-25. Stripline T Junction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 266Figure 10-26. Stripline T Junction Symbol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 266Figure 10-27. Stripline Step in Width . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 269

List of Figures


Figure 10-28. Stripline Step in Width Symbol . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 269Figure 10-29. Equivalent circuit for a Stripline Step in Width . . . . . . . . . . . . . . . . . . . . . . . 270Figure 13-1. Selecting MODSST Analysis in the Questa ADMS GUI. . . . . . . . . . . . . . . . . 286Figure 13-2. Modulated Steady State Dialog in the Questa ADMS GUI . . . . . . . . . . . . . . . 287Figure 13-3. Fundamental Frequency Specification Dialog in the Questa ADMS GUI . . . . 287Figure 14-1. Amplifier Circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 294Figure 14-2. Single-tone Steady-State Analysis and Two-port NoiseExtraction Simulation Results 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 297Figure 14-3. Single-tone Steady-State Analysis and Two-port NoiseExtraction Simulation Results 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 298Figure 14-4. Single-tone Steady-State Analysis and Two-port NoiseExtraction Simulation Results 3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 299Figure 14-5. Single-tone Steady-State Analysis and Two-port NoiseExtraction Simulation Results 4 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 300Figure 14-6. Amplifier Circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 301Figure 14-7. Power Efficiency and 1 dB Compression Point ExtractionSimulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 305Figure 14-8. Amplifier Circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 306Figure 14-9. Multi-tone Analysis—IM3 and IP3 Extraction Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 309Figure 14-10. Amplifier Circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 310Figure 14-11. Two Port Constant Gain Circle—GAC. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 313Figure 14-12. Two Port Constant Gain Circle—GPC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 314Figure 14-13. Two Port Stability Circle—LSC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 315Figure 14-14. Two Port Stability Circle—SSC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 316Figure 14-15. Gilbert Cell Circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 317Figure 14-16. Mixer Steady-State and Noise Analysis Simulation Results 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 320Figure 14-17. Mixer Steady-State and Noise Analysis Simulation Results 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 321Figure 14-18. Gilbert Cell Circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 322Figure 14-19. Mixer Steady-State AC Analysis Simulation Results. . . . . . . . . . . . . . . . . . . 324Figure 14-20. Mixer Steady-State TF Analysis Simulation Results 1. . . . . . . . . . . . . . . . . . 328Figure 14-21. Mixer Steady-State TF Analysis Simulation Results 2. . . . . . . . . . . . . . . . . . 329Figure 14-22. Mixer Steady-State TF Analysis Simulation Results 3. . . . . . . . . . . . . . . . . . 330Figure 14-23. Frequency Divider Circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 331Figure 14-24. Steady-State Analysis Simulation Results 1 . . . . . . . . . . . . . . . . . . . . . . . . . . 333Figure 14-25. Steady-State Analysis Simulation Results 2 . . . . . . . . . . . . . . . . . . . . . . . . . . 334Figure 14-26. Amplifier Circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 335Figure 14-27. Steady-State Analysis of Autonomous Circuit Simulation Results 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 338Figure 14-28. Steady-State Analysis of Autonomous Circuit Simulation Results 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 339Figure 14-29. Amplifier Circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 340Figure 14-30. Sweeping the Oscillator Frequency Simulation Results . . . . . . . . . . . . . . . . . 342

List of Figures


Figure 14-31. Amplifier Circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 343Figure 14-32. Phase Noise Extraction Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . 345Figure 14-33. Plotting in the Complex Plane . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 350Figure 14-34. IQ Trajectory Diagram for a PI/4QPSK Source . . . . . . . . . . . . . . . . . . . . . . . 351Figure 14-35. Constellation Diagram for a PI/4QPSK Source . . . . . . . . . . . . . . . . . . . . . . . 352Figure 14-36. ACPR Computations Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . 356Figure 14-37. Verilog-A Usage Simulation Results 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 359Figure 14-38. Verilog-A Usage Simulation Results 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 360Figure 14-39. Verilog-A Usage Simulation Results 3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 361Figure 14-40. NPR Computation with Steady-State Simulation Results 1 . . . . . . . . . . . . . . 365Figure 14-41. NPR Computation with Steady-State Simulation Results 2 . . . . . . . . . . . . . . 366Figure 14-42. NPR Computation with Modulated Steady-State Simulation Results 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 370Figure 14-43. NPR Computation with Modulated Steady-State Simulation Results 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 371Figure 14-44. EVM and BER Computations Complex Function . . . . . . . . . . . . . . . . . . . . . 375Figure 14-45. EVM and BER Computations Complex Plane Plots . . . . . . . . . . . . . . . . . . . 376Figure 14-46. EVM and BER Computations Constellation Diagram Setup . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 377Figure 14-47. EVM and BER Computations Constellation Diagram Plots . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 377Figure 14-48. EVM and BER Computations EVM and BER Setup . . . . . . . . . . . . . . . . . . . 378Figure 14-49. Zout_R values Pull Load Contours Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 382Figure 14-50. Load Pull Contours Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 384Figure 14-51. Final Waveforms—Load Pull Contours Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 385Figure 14-52. Schematic diagram of the circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 386Figure 14-53. Simulation Results — TSST . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 389Figure 14-54. Simulation Results — FSST . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 390Figure 14-55. Schematic of the circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 393Figure 14-56. Schematic of the LNA. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 394Figure 14-57. Schematic of the Mixer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 394Figure 14-58. Synoptic of Oscillator and Down-Converting Mixer . . . . . . . . . . . . . . . . . . . 399Figure 14-59. S21 Value in dB Versus Frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 401Figure 14-60. S11 Plot on Smith Chart . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 401Figure 14-61. S11 Plot on Polar Chart. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 402Figure 16-1. Schematic of LNA. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 441Figure 17-1. Synoptic of the AGC Loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 445Figure 17-2. Load Design Dialog . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 450Figure 17-3. Questa ADMS User Interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 451Figure 17-4. Wave Window. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 453Figure 17-5. Overlapped Input and Output Waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . 454Figure 18-1. The ZigBee Chain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 457

List of Figures


Figure 18-2. Synoptic of the Complete ZigBee Chain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 458Figure 18-3. Mapping, Spreading and Modulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 459Figure 18-4. O-QPSK Chip Offsets (Tc=period_chip=0.5µs). . . . . . . . . . . . . . . . . . . . . . . . 460Figure 18-5. Sample Baseband Chip Sequences with Pulse Shaping . . . . . . . . . . . . . . . . . . 461Figure 18-6. IQ Modulation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 461Figure 18-7. Multipath Effects. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 462Figure 18-8. Questa ADMS User Interface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 469Figure 18-9. Simulation Results — First Set of Signals . . . . . . . . . . . . . . . . . . . . . . . . . . . . 471Figure 18-10. More Simulation Results. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 472Figure 18-11. Filtering Effects. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 473Figure 18-12. Constellation Diagram Dialog Box. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 474Figure 18-13. Constellation Diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 475Figure 18-14. BER Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 476


List of Tables

Table 2-1. Analysis Commands . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27Table 3-1. Display Commands . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75Table 4-1. Complex Number Format . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 117Table 5-1. Eldo RF Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 153Table 5-2. Time Domain Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 154Table 5-3. Accuracy Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 154Table 5-4. Convergence Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 154Table 5-5. Frequency Tolerance Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 155Table 5-6. Noise Result Presentation Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 155Table 5-7. Multi-threaded Simulation Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 155Table 5-8. MODSST Analysis Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 155Table 5-9. Miscellaneous Analysis Options . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 155Table 5-10. SST_ACCURACY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 159Table 5-11. SST_ESTIM_ACCURACY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 159Table 5-12. Convergence Option Restrictions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 164Table 5-13. SST_CIRCUIT_TYPE Descriptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 172Table 8-1. Default Rule . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 196Table 9-1. LDTL Level 3 Parameter Combinations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 219Table 9-2. RLGC Separator Characters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 229Table 9-3. Lossy Transmission Line: U Model Parameter Combinations . . . . . . . . . . . . . . 240Table 10-1. Microstrip T Junction Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 242Table 10-2. Microstrip Bend (Arbitrary Angle, Optimally Mitered) Parameters . . . . . . . . . 245Table 10-3. 90-degree Microstrip Bend (Mitered) Parameters . . . . . . . . . . . . . . . . . . . . . . . 248Table 10-4. 90-degree Microstrip Bend (Optimally Mitered) Parameters . . . . . . . . . . . . . . 251Table 10-5. 90-degree Microstrip Bend (Unmitered) Parameters . . . . . . . . . . . . . . . . . . . . 254Table 10-6. Microstrip Step in Width Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 257Table 10-7. Cylindrical Via Hole in Microstrip Parameters . . . . . . . . . . . . . . . . . . . . . . . . . 261Table 10-8. Unmitered Stripline Bend Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 263Table 10-9. Stripline T Junction Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 266Table 10-10. Stripline Step in Width Parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 269Table 12-1. Verilog-A Restrictions Related to RF Analysis . . . . . . . . . . . . . . . . . . . . . . . . 281Table 13-1. VHDL-AMS RF Subset Support . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 288Table 14-1. Tutorials Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 293Table 14-2. Simulation Times for Each Netlist . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 390Table 18-1. Symbol-to-Chip Mapping . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 459Table 18-2. Decision Regions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 463

List of Tables



Chapter 1Introduction

Steady-State Analysis TypesThis document describes the functionality of Eldo RF™, which is the Eldo® simulator withadded extensions for RF simulation in order to allow the fast large-signal Steady-State analysis(SST analysis) of high frequency electronic circuits.

Steady-State Analysis for Non-Autonomous CircuitsThis analysis determines the spectral content and waveform of all circuit nodes at Steady-Statefor non-autonomous circuits excited with periodic (single-tone) or quasi-periodic (multi-tone)signals. Single-tone Steady-State analysis can be used to compute Total Harmonic Distortion,Compression Point, Power Efficiency and Power Added Efficiency of amplifiers and frequencymultipliers. Multi-tone Steady-State analysis can be used to compute Intermodulation (IMx),compression point or Intercept Point (IPx) of amplifiers, mixers, filters, and rejecters. Steady-State analysis can also be used to compute large-signal S, Y or Z parameters. Additionally,Steady-State analysis can be used to analyze frequency dividers.

For more information see “Steady-State Analysis of Non-Autonomous Circuits” onpage 29.

Steady-State Analysis for Autonomous CircuitsThis analysis determines the spectral content and waveform of all circuit nodes at Steady-Statefor autonomous circuits.

Steady-State analysis for autonomous circuits requires the insertion of a probe in the circuit.Probes are used to compute frequency and oscillation level at insertion points. DuringSteady-State analysis for autonomous circuits, they behave like a pure sinusoidal generator atthe oscillation fundamental frequency, and an open circuit for all other frequencies. No morethan one probe should be inserted per oscillation frequency.

During a Steady-State analysis for autonomous circuits, Eldo RF uses an optimizationprocedure to compute the oscillation frequency and the magnitude of the probe. Theoptimization stops when the admittance of the probe is below a given tolerance. The circuitmust then be an oscillator at this frequency. A good estimation of oscillation frequency isrequired to guarantee rapid convergence. This estimation can be provided by the user, orobtained using the local stability analysis (see “Local Stability Analysis” on page 19).

Eldo RF User’s Manual, AMS 2009.218

IntroductionAnalyses Based on Steady-State Analysis

Efficiency of analysis depends also on where the probe is placed in the circuit. Insert a voltageprobe in parallel with the resonator, or in parallel with the load. The probe should have someeffect on the oscillation; in particular, it should not be put after the buffer, or in the biasingcircuitry. In case of convergence problems, you should manually change the place where theprobe is connected.

For analyses other than Steady-State analysis:

• Voltage probes behave as an open circuit and won’t have any influence on thesimulation results.

Eldo RF can also analyze multi-tone autonomous circuits. It can be one autonomous frequencyplus another forced one (for instance a self-oscillating mixer or circuits including a free-runningoscillator and a mixer). However, it can also be a multi-autonomous circuit (a circuit containingdifferent free-running oscillations).

For more information see “Steady-State Analysis of Autonomous Circuits” on page 33.

Analyses Based on Steady-State Analysis

Parametric Steady-State AnalysisParametric simulation allows you to compute the Steady-State response of circuits whilesweeping one circuit’s parameter. The type of parameter that can be swept is one of thefollowing:

• Generator parameter (magnitude or phase)

• Fundamental frequency

• Parameter of any device

• Any .param parameter

For further information on parameter sweeps, please refer to .STEP of the Eldo User’sManual.

Results are output as a family of waveforms (each waveform specified by the output commandfor each value of the swept parameter) and as a parametric curve (specified with the extractioncommand).

The simulation is performed using solution prediction from preceding parameter values, whichyields significant performance improvement over a succession of independent analyses.

IntroductionAnalyses Based on Steady-State Analysis


Local Stability AnalysisNumeric modeling of non-linear circuits may yield several equilibrium points, only a few ofwhich are physically observable, i.e. stable. All the others are unstable. In order to know if a DCsolution is stable or unstable a local stability analysis can be performed.

Local stability analysis consists of applying a small perturbation to the system, and to observe ifit comes back to its equilibrium point. If it doesn’t, it is unstable.

A local stability analysis will indicate if it finds the circuit stable or not, and if the circuit isunstable it will provide an estimation of the oscillation frequency(ies). This analysis isparticularly useful with oscillators (to provide an initial guess for the oscillation frequency, or totell if the circuit has multiple oscillation frequencies). When more than one potential oscillationfrequency is computed, they are displayed in the most probable order.

Steady-State AC Analysis (SSTAC)Steady-State AC analysis can follow a Steady-State or Modulated Steady-State analysis. It isused with circuits that exhibit frequency conversion, such as mixers. The Steady-State analysisis first computed, then the circuit is linearized around the quasi-periodic time-varying operatingpoint, and a small signal is applied to compute the small signal response. This analysis can beused as a simplified multi-tone Steady-State analysis for the cases where one signal can beconsidered as small compared to the others, and is treated as a small perturbation around theSteady-State obtained with the other large signals.

SSTAC is much faster than multi-tone SST. SSTAC is particularly useful to compute the small-signal conversion gain/loss.

Steady-State Transfer Function Analysis (SSTXF)Steady-State Transfer Function (TF) analysis can follow a Steady-State or Modulated Steady-State analysis. SSTXF computes transfer functions (TF) from any source at any frequency in thecircuit to a single output at a single frequency. It can be used to analyze circuit parameters suchas conversion efficiency (transfer function from input to output at desired frequency), imageand sideband rejection, (transfer function from input to output at undesired frequency), LOfeedthrough and power supply rejection (undesired input to output at all frequencies).

Steady-State Noise Analysis (SSTNOISE)Steady-State Noise analysis can follow a Steady-State or Modulated Steady-State analysis and itis based on SSTAC or SSTXF analysis. This means that the circuit under consideration islinearized around its quasi-periodic time-varying operating point and the noise spectrum iscomputed at a specified output. The output noise spectrum is generated by frequency up/downconversion of all the circuit noise sources. This analysis can be applied to amplifiers, mixers, as


IntroductionModulated Steady-State Analysis Types

well as oscillators to compute phase noise. It computes the output noise spectrum, thecontribution of any noisy device, as well as the Noise Figure.

An alternative algorithm is available in order to speed-up SSTNOISE analysis since Eldo RFv6.2. This can be activated with the IMPROVED_SSTNOISE_PERF option. Theseimprovements lead to a 2× to 5× speed ratio compared to the previous solution for most circuits.

In addition, a different algorithm is automatically used to solve the Steady-State Phase Noise ofPLL circuits. Compared to the previous solution for phase noise computations of PLL, thisapproach is much more efficient (2× to 50× faster) and more accurate (especially at lowfrequency offset).

Modulated Steady-State Analysis Types

Modulated Steady-State Analysis for Non-AutonomousCircuits

In comparison with IP3 or 1dB gain compression, characterizations such as ACPR (AdjacentChannel Power Ratio), and NPR (Noise Power Ratio) of Power Amplifiers are difficult to beextracted from a simple SST analysis. The Modulated Steady-State analysis (MODSSTanalysis) is a simulation technique that can handle realistic digitally modulated RF sources.

This analysis determines the time-varying spectral content and waveform of all circuit nodes fornon-autonomous circuits excited with periodic (single-tone) or quasi-periodic (multi-tone)signals and digitally modulated signals. Since the output from MODSST analysis is a time-varying spectrum, it is possible to get the instantaneous amplitude and phase modulationinformation of each harmonic. Also, it is possible to plot the time-varying spectral content ofany nodes and currents.

All sources used for a SST analysis are compatible with MODSST analysis. Standard Eldoperiodic and non-periodic sources are also accepted whenever MODSST analysis is activated.

Modulated Steady-State Analysis for AutonomousCircuits

MODSST can also be applied to autonomous circuits. It is used to study oscillator start-uptransient or also to analyze modulated oscillators. It is possible to display time-varyingspectrum information as for standard MODSST and it is also possible to display theinstantaneous oscillation frequency versus time.

Run the examples named crystal_modsst.cir and crystal_tran.cir for more information.

IntroductionEldo RF Algorithms


Eldo RF AlgorithmsDuring a .SST analysis Eldo RF computes the large signal steady-state of circuits stimulated byN-tone periodic signals. The algorithm used in Eldo RF is an advanced version of HarmonicBalance based on a matrix-free Krylov Subspace solver. The basics of this algorithm can befound in reference [1].

Multi-tone steady-state is supported thanks to an N-dimension FFT algorithm or using anArtificial Frequency Mapping (AFM), N dimensions are mapped to one artificial dimension.The AFM algorithm is described in reference [2].

For the simulation of very non-linear circuits such as frequency dividers a special purposepreconditioner is used as described in reference [3].

Local DC stability analysis is computed using the algorithm described in reference [4]. Eldo RFsolves the steady-state of autonomous circuits (oscillators) using extensions and combinationsof the algorithms described in references [5] and [6].

Noise and Phase Noise analysis around steady-state are performed using extended versions ofthe algorithms described in references [7] and [8]. It should be noted that the algorithmdescribed in reference [9] is not used for the analysis of oscillators because it is a Quasi-staticmethod and may lead to incorrect phase noise results.

The background theory of the Modulated Steady-State (MODSST) analysis is described inreferences [10], [11] and in reference [12] for oscillators.

Algorithm for Modulated Steady-State AnalysisSST analysis directly computes the state reached by a circuit submitted to periodic or quasi-periodic large signal excitation when all transients have died out. However, not all relevantquantities can be analyzed using SST analysis and derived analyses. Also for simulating modernwireless circuits with complex digitally modulated RF signals such as CDMA (Code DivisionMultiple Access), TDMA (Time Division Multiple Access) and FDMA (Frequency DivisionMultiple Access), it is better to analyze in the time domain. Time integration, however, is stillinefficient due to the high frequencies present in the circuit that force the use of very small timesteps.

Modulated Steady-State analysis (MODSST analysis) is based on a mixed time-frequencymethod. The simulation technique does not have the above limitations.

Consider the circuit equations in time domain:

i v t( )( ) dq v t( )( )dt

--------------------- is t( ) 0=+ +


IntroductionOptimization Capability

where node voltage

and is(t) is the current source, , and = the fundamental frequency.

The time-varying Fourier coefficients of the currents are:

The first two terms are the steady-state terms as computed by SST analysis and they do notdepend on the past history or derivative information. The last two terms represent the envelopecurrent at frequency k caused by a time-varying change in the envelope voltage at frequency k.MODSST analysis applies a time-domain technique on top of the frequency-domain harmonicbalance solution during simulation. The derivative term can be computed using any order ofintegration (backward-euler, trapezoidal, gear). Backward-euler is used by default. The time-domain is handled by Eldo in terms of time-step control (LTE, accuracy options) as in a regulartransient analysis (see Eldo User’s Manual). At each timepoint, a SST analysis is performed infrequency-domain.

MODSST analysis accepts the input stimulus as RF carriers with time-varying complexenvelopes (digital modulation). This method has a fundamental advantage over time-domainsimulators in that the size of the timestep only has to be small enough to catch the bandwidth ofthe modulation envelope (~10 kHz - 1 MHz), instead of the RF carrier (~1-5 GHz). The outputsolution is represented by the sum of the RF carriers and their harmonics, each with a time-varying complex envelope.

Optimization CapabilityThe optimization capability of Eldo can be used for Eldo RF analyses. Optimization can beused, for example, to optimize gains, power dissipation, intermodulation, matching networks,etc.

Please refer to the chapter Optimizer in Eldo of the Eldo User’s Manual.

Verilog-AVerilog-A models can be simulated with RF analysis (.SST, .SSTAC, .SSTXF, .SSTNOISE and.MODSST). Verilog-A models are directly available from a “Spice” netlist through a Yxxelement instantiation.

Please refer to Chapter 12, “Eldo RF and Verilog-A” of this manual.

v t( ) Re V k t( )ejωk t

k 0=

N

∑

=

ωk 2π f 0= f 0

I k t( ) jωkQk t( )dQk t( )

dt----------------- I k

st( ) 0=+ + +

IntroductionQuesta ADMS


Questa ADMSQuesta ADMS can be used to launch a MODSST analysis with RF/Analog/Digital circuits.

Please refer to Chapter 13, “Eldo RF and Questa ADMS” of this manual.

CommLib RFCommLib RF is a library of Verilog-A behavioral models and supporting packages targetingtop-down design of radio frequency integrated circuits (RF ICs). The library is supplied assource code in the Verilog-A analog hardware description language. Each model describes theessential characteristic functionality of the corresponding device class using abstract modelingtechniques.

CommLib RF models allow you to get up and running quickly. The models as supplied can beused for system level design and architectural exploration. The model sources provide avaluable lesson in behavioral modeling techniques. The sources also serve as prototypes thatcan be customized and extended to fit the designers special requirements.

CommLib RF can be used in conjunction with Eldo RF and ADMS RF. Additionally, it iscompatible with the Verilog-A subset supported for SST and MODSST analyses.

For further information, see the CommLib RF Library.

Eldo RF Multi-ThreadingEldo RF multi-threading is automatically activated when a design is suitably large enough andthe simulation is being run on a multi-processor machine. For the single simulation, Eldo RFwill share computer resources on the multi-processor machine.

The following message will be displayed when multi-threading is used for a simulation:

Launching Multithreading: 4 processors potentially used

For information on the multi-threading options please refer to “Multi-Threaded SimulationOptions” on page 168.

Invoking Eldo RFEldo RF can be invoked in the same way as Eldo from the command line:

eldo <circuit_name>

Additional arguments may be specified as required.


IntroductionInvoking Eldo RF

64-bit ModeEldo RF is also available as a 64-bit version for Solaris and Linux platforms. This enablessimulation of circuits which would require more than 2GB of memory, and which wouldtherefore not work on 32-bit machines.

Eldo RF 64-bit mode can be invoked using the command:

eldo <circuit_name> -64b <other_arguments>

To run Eldo RF with the -64b flag on a Solaris 64-bit machine you must first set theenvironment variable AMS_VCO_MODE to 64, for example:

setenv AMS_VCO_MODE 64

IntroductionRelated Documentation


Related DocumentationOther documents and manuals that are referenced in this manual, and that you may need to referto are:

• Eldo User’s Manual

• Eldo Verilog-A User’s Manual

• EZwave User’s and Reference Manual

• Questa ADMS User’s Manual

References1. H. G. Brachtendorf, G. Welsch and R. Laur, Fast simulation of the steady-state of

circuits by the harmonic balance technique Circuits and Systems, 1995. ISCAS '95.,1995 IEEE International Symposium on Volume 2, 28 April-3 May 1995 Page(s):1388- 1391 vol.2.

2. D. Hente and R.H. Jansen, Frequency Domain Continuation Method for the Analysisand Stability Investigation of Nonlinear Microwave Circuits, IEEE Proceedings, part H,vol. 133, no.5, pp. 351-362, Oct. 1986.

3. F. Veerse, Efficient iterative time preconditioners for harmonic balance RF circuitsimulation ICCAD-2003. International Conference on Computer Aided Design, 2003.pp251 - 254.

4. P. Bolcato, J. C. Nallatamby, C. Rumolo, R. Larcheveque, M. Prigent and J. Obregon,Efficient algorithm for steady-state stability analysis of large analog/RF circuitsMicrowave Symposium Digest, 2001 IEEE MTT-S International, Volume 1, 20-25May 2001 Page(s):451 - 454 vol.1

5. E. Ngoya, A. Suarez, R. Sommet and R. Quere, Steady state analysis of free or forcedoscillators by harmonic balance and stability investigation of periodic and quasi-periodic regimes, International Journal of Microwave and Millimeter-Wave Computer-Aided Engineering, vol. 5, no. 3, pp. 210-233, Mar. 1995.

6. V. Rizzoli, A. Costanzo and A. Neri, Harmonic-balance analysis of microwaveoscillators with automatic suppression of degenerate solution Electronics LettersVolume 28, Issue 3, 30 Jan. 1992 Page(s):256 - 257

7. J. Roychowdhury, D. Long and P. Feldmann, Cyclostationary noise analysis of large RFcircuits with multitone excitations, Solid-State Circuits, IEEE Journal of Volume 33,Issue 3, March 1998 Page(s):324 - 336

8. P. Bolcato, J. C. Nallatamby, R. Larcheveque, M. Prigent and J. Obregon, A unifiedapproach of PM noise calculation in large RF multitone autonomous circuits


IntroductionReferences

Microwave Symposium Digest., 2000 IEEE MTT-S International, Volume 1, 11-16June 2000 Page(s):417 - 420 vol.1

9. V. Rizzoli, F. Mastri and D. Masotti, General noise analysis of nonlinear microwavecircuits by the piecewise harmonic-balance technique Microwave Theory andTechniques, IEEE Transactions on Volume 42, Issue 5, May 1994 Page(s):807 - 819

10. E. Ngoya and R. Larcheveque, Envelope Transient Analysis: A New Method for theTransient and Steady State Analysis of Microwave Communication Circuits and SystemsIEEE MTT Symposium Digest, 1996, pp. 1365-1368.

11. P. Feldmann and J. Roychowdhury, Computation of Waveform Envelopes Using anEfficient, Matrix-decomposed Harmonic Balance Algorithm Proc. of the IEEE/ ACMInternational Conference on Computer-Aided Design, pp. 295-300, Nov. 1996.

12. H. G. Brachtendorf, G. Welsch and R. Laur, A time-frequency algorithm for thesimulation of the initial transient response of oscillators, in Proc. of the IEEEInternational Symposium on Circuits and Systems, 1998. ISCAS '98. Volume 6, 31May-3 June 1998 Page(s):236 - 239 vol.6


Chapter 2Analysis Command Syntax

SummaryThe following table shows the available analyses in Eldo RF.

Table 2-1. Analysis Commands

Description Command

Steady-State Analysis .SST

Steady-State Analysis of Autonomous Circuits .SST OSCIL

Steady-State Analysis of PLLs .SST PLL

Local Stability Analysis .SST STABIL

Steady-State AC Analysis .SSTAC

Steady-State TF Analysis .SSTXF

Steady-State Non-linear Contributors Analysis .SSTNLCONTRIB

Steady-State RLC Sensitivity Analysis .SSTSENSRLC

Steady-State Noise Analysis .SSTNOISE

Steady-State Noise Analysis, Spot Noise Figure .SNF

Steady-State Worst Case Analysis .WCASE

Steady-State Monte Carlo Analysis .MC

Modulated Steady-State Analysis .MODSST

Modulated Steady-State Circuit Partitioning .PART MODSST

Steady-State Circuit Partitioning .RFBLOCK

Drives a node with Simulation Data .CHRSIM

Age Analysis .AGE

DC Operating Point Calculation for RF .OP RF


Analysis Command SyntaxAnalysis Command Descriptions

Analysis Command Descriptions

Steady-State AnalysisThe .SST command activates Steady-State analysis. By specifying the correct parameters, it isable to perform the following analyses:

• Steady-State of amplifiers, mixers, frequency dividers, and PLLs.

• Steady-State of free-running oscillators and self oscillating mixers.

• Local stability analysis (stability of DC operating points).

• Steady-State of the PFD/CP block of a PLL in synchronized conditions with automatictuning.

• Steady-State of free-running oscillators with automatic tuning.

The same command is used to activate the different kinds of analyses depending on thespecified parameters.

Analysis Command Syntax.SST


.SSTSteady-State Analysis of Non-Autonomous Circuits

Command Syntax

.SST+ FUND1=real_value NHARM1=INTEGER_VALUE+ [FUNDxx=real_value NHARMxx=INTEGER_VALUE]+ [PHASE_CONTROL_SOURCE=source_name] [OUTCP_NODE=node_name]+ [VOUTCP_TARGET=targeted_voltage_value]

NoteIt is not possible to specify multiple .SST commands in the same netlist.

Parameters

• FUNDxx

Specifies the fundamental frequencies of the circuit. The term ‘fundamental frequencies’ isunderstood as non-harmonically related frequencies, or frequencies that are very highharmonics of a common frequency (e.g. 100 MHz and 101 MHz are harmonics of 1 MHz,but considered as fundamental frequencies). There should be at least one fundamentalfrequency. The values should be non-negative, otherwise the analysis will abort with anerror message. See also the Notes below.

• NHARMxx

Specifies the maximum number of harmonics corresponding to each fundamentalfrequency. They must be non-negative integer values, otherwise the analysis will abort withan error message. All waveforms in the steady-state solution are represented using Fourierseries, where NHARMxx corresponds to the last term in the truncated series. Having moreharmonics and/or more time-points in the FFT (see “SST_OVRSMP” on page 158)minimizes aliasing and improves accuracy (at the expense of longer run time and highermemory requirement).

The accuracy of the SST analysis results is directly related to the specified number ofharmonics. This number has to be large enough to represent the signals by thecorresponding truncated Fourier Series.

This NHARM parameter also greatly influences the accuracy of the SSTNOISE results(especially in circuits having digital-like signals such as frequency dividers).

• xx

Specifies the number of the fundamental frequencies and corresponding harmonics. Thenumber should range from 1 to the total number of fundamental frequencies.

• PHASE_CONTROL_SOURCE

Specifies the input source for PFD/CP tuning. The voltage at the output node (OUTCP_NODE)is dependent upon the phase of this input source. During the SST analysis, the phase of this



input source is tuned until the voltage at OUTCP_NODE is equal to the value specified byVOUTCP_TARGET.

• OUTCP_NODE

Specifies the output node whose voltage is dependent upon the phase of the input sourcePHASE_CONTROL_SOURCE.

• VOUTCP_TARGET

Target output-node voltage. Specifies the value against which to measure the voltage ofOUTCP_NODE.

Notes

• The FUNDxx keywords can be reused for the definition of multi-tone sources (see Multi-Tone Source).

• In a mixer, FUND1 should be the LO frequency, and FUND2 the RF or the IF frequency.

• Frequency division is only allowed for FUND1. Frequency divisions for anything otherthan FUND1 will be ignored and a Warning will be issued as follows:

Frequency division detected (and ignored) on FUNDxx direction,because division is only handled for FUND1 direction!

• In netlists containing a .SST command, FUNDxx can be used as a keyword inparameter definition of plots, e.g.

.SST FUND1=1.9Giga NHARM1=5

.extract fsst yval(vdb(out), FUND1)

• There is no limit on the number of fundamental frequencies (except CPU time andmemory).

• The actual frequencies used in the computed spectrum are listed in the intermodulationtable in the .chi file.

• real_value and integer_value can be expressed as parameters.

• In order to independently analyze the PFD/CP block as if it was inside a PLL undersynchronized conditions, the optimal pulse source delay corresponding to the tunedphase of the control source must be found. This means that the specification ofPHASE_CONTROL_SOURCE, OUTCP_NODE and VOUTCP_TARGET parameters of the .SSTcommand are required in order to achieve these synchronized conditions.

Examples

.SST FUND1=500meg NHARM1=10

.SST FUND1=1giga NHARM1=3+ FUND2=920meg NHARM2=3+ FUND3=930meg NHARM3=3



The following example shows steady-state analysis for PFD/CP with automatic tuning:

.SST FUND1=45meg NHARM1=200+ PHASE_CONTROL_SOURCE=vcontrol+ OUTCP_NODE=in_filter+ VOUTCP_TARGET=1.5

For more information run the file pfd_tuning.cir, located in the directory:$MGC_AMS_HOME/examples/rfic/

In the example, the phase of the source vcontrol (PHASE_CONTROL_SOURCE) is tuned so that itproduces a 1.5 V voltage (VOUTCP_TARGET) on node in_filter (OUTCP_NODE).

Of interest is the optimal pulse source delay Td=22.133 ns corresponding to the tuned phase ofthe control source. This is the value of the td parameter that is used (instead of the 0ns specifiedin the netlist) in order to analyze independently the PFD/CP block as if it was inside a PLLunder synchronized conditions producing a voltage v(in_filter)=1.5V.

DC Operating Point Calculation for RFIt is possible to write out the exact DC obtained prior to the SST analysis. This feature has beendeveloped because some sources are not handled in the same way for the operating point.

For .OP the value of the source at time=0 is taken into account if no DC value is specified, butfor .OP RF (see .OP RF) the average value of the source is taken into acount, even if a DCvalue is specified.

If .OP and .OP RF are specified in the same netlist, .OP RF is ignored.

For instance, in the folowing simple example:

V1 1 0 PULSE 0 2 0 0 0 0.5n 1nR1 1 2 1kR2 2 0 1k

.PARAM FUND1=1g

.OP RF

.SST FUND1=1g NHARM1=1

.PLOT TSST v(1)

.END

Then .OP RF gives:

1 1.0020E+002 5.0100E-01

whereas .OP, because the source at time=0 is 0, gives:



DC:1 iterations FOR DC analysis1 0.02 0.0

Analysis Command Syntax.SST OSCIL


.SST OSCILSteady-State Analysis of Autonomous Circuits

Command Syntax

.SST OSCIL+ [FUND_OSC_GUESS1=real_value] NHARM_OSC1=INTEGER_VALUE+ [FUND_OSC_GUESSxx=real_value NHARM_OSCxx=INTEGER_VALUE]+ [FUNDyy=real_value NHARMyy=INTEGER_VALUE]+ [FUND_OSC_TARGET=real_value] [VCONTROL=voltage_source_name]

Parameters

• OSCIL

Keyword specifying that the circuit is autonomous, and that self-oscillating fundamentalfrequency(ies) have to be calculated.

• FUND_OSC_GUESSxx

Estimated oscillation fundamental frequency. If not specified for FUND_OSC_GUESS1, theLocal Stability Analysis result is used. The values should be positive.

• NHARM_OSCxx

Number of harmonics (of the oscillation frequency) corresponding to FUND_OSC_GUESSxx.The value should be a non-negative integer.

The accuracy of the SST analysis results is directly related to the specified number ofharmonics. This number has to be large enough to represent the signals by thecorresponding truncated Fourier Series.

This NHARM parameter also greatly influences the accuracy of the SSTNOISE results(especially in circuits having digital-like signals such as frequency dividers).

• xx


• FUNDyy

Fundamental frequencies (different from oscillation frequency). These can be used forinstance in the case of self-oscillating mixers. The values should be non-negative.

• NHARMyy

Number of harmonics corresponding to FUNDyy. The values should be non-negative integervalues. All waveforms in the steady-state solution are represented using Fourier series,where NHARMyy corresponds to the last term in the truncated series. Having moreharmonics and/or more time-points in the FFT (see “SST_OVRSMP” on page 158)minimizes aliasing and improves accuracy (at the expense of longer run time and highermemory requirement).



• yy


• FUND_OSC_TARGET

If specified, along with VCONTROL, Eldo RF finds the steady-state regime corresponding to acalculated fundamental oscillation frequency, fund_osc=fund_osc_target, for a specificcalculated value of the VCONTROL source voltage.

• VCONTROL

Specifies the voltage source controlling the fundamental oscillation frequency, fund_osc,of the oscillator circuit.

Notes

• real_value and integer_value can be expressed as parameters.

Constraints

There should be only one probe per oscillation frequency in the circuit.

See the “Probe Source” on page 119 for further information.

Example

This example specifies a steady-state analysis with an initial guess of 1.8GHz for the oscillationfrequency with ten harmonics.

.SST OSCIL FUND_OSC_GUESS1=1.8G NHARM_OSC1=10

The next example shows a setup for a combination oscillator+mixer.

.SST OSCIL FUND_OSC_GUESS1=1.8G NHARM_OSC1=10+ FUND2=1.9G NHARM2=5

The next example specifies a steady-state analysis with an initial estimation of 1.8GHz for theoscillation frequency with five harmonics, followed by an estimation of 1.1GHz for theoscillation frequency with five harmonics.

.SST OSCIL FUND_OSC_GUESS1=1.8G NHARM_OSC1=5+ FUND_OSC_GUESS2=1.1G NHARM_OSC2=5

The next example specifies a steady-state analysis for an autonomous circuit with automatictuning. The VCO fundamental oscillation frequency is defined as follows:

.SST OSCIL FUND_OSC_TARGET=2.05g VCONTROL=vcontrol NHARM=10



The VCONTROL parameter is used to specify which voltage source controls the fundamentaloscillation frequency of the oscillator circuit.

For more information run the file vco_fosc_target.cir, located in the directory:$MGC_AMS_HOME/examples/rfic/

The expected result is a steady-state regime where fund_osc = FUND_OSC_TARGET for a specificVCONTROL source voltage value. The output display for the vco_fosc_target example is:

Starting VCO Optimization for F0 = 2.0500e+09 (initial conditions:Vctrl = 1.2000e+00 / F0 = 1.8362e+09) ...

Vctrl = 1.893204e+00 (dVctrl = 6.932039e-01) F0 = 1.995778e+09Vctrl = 2.128771e+00 (dVctrl = 2.355672e-01) F0 = 2.034410e+09Vctrl = 2.223834e+00 (dVctrl = 9.506297e-02) F0 = 2.048581e+09Vctrl = 2.233352e+00 (dVctrl = 9.517642e-03) F0 = 2.049960e+09Vctrl = 2.233626e+00 (dVctrl = 2.746513e-04) F0 = 2.050000e+09

----- Optimization suceeded : ----- Voltage control (Vctrl=2.2336e+00) for Oscillation frequency(F0=2.0500e+09) in 5 iterations

Steady-State Oscillator Algorithm DetailsBy default, the algorithm used to simulate an oscillator is divided into three main phases:

Phase 1 (P1)

The initialization phase, performed in order to find the minimum value of the curve Y(Vprobe).During this phase the oscillation frequency is kept constant (at FUND_OSC_GUESS1 or at thevalue found by the stability analysis) and the probe voltage is swept until the objective isreached.

Phase 2 (P2)

The optimization phase, performed in order to find the optimal oscillation conditions. Forexample, a sequence of iterations will be performed on the variables Vprobe and fund_osc untilY(Vprobe, fund_osc) < internal tolerance. The iterations can be seen in the terminal window.

Phase 3 (P3)

The improvement phase, performed to reach an accurate solution with respect to the oscillationfrequency.



When a frequency divider is present in the circuit and/or the optionSST_CONVERGENCE_HELP=transient is used in the netlist, a short transient analysis (Tr) isperformed after P1.

The initial conditions for the transient analysis are the results from P1. Hence the completealgorithm sequence is:

P1 ⇒ [Tr] ⇒ P2 ⇒ P3

When a parameter sweep analysis (.STEP) is specified in the netlist, only the first step willbegin with phase one (and the Tr phase if this is required), the remaining steps will begin withphase two.

With recalcitrant oscillators (oscillator with convergence problems), the default algorithm canfail to converge during one of the phases. The option SST_OSC_PHASE_SEQUENCE should beused to prevent this failure and you can specify the different combinations of the four phases.By using this option for a parameter sweep analysis, only the first step point is analyzed with thespecified sequence. To use the specified sequence applied for all the point in the sweep theoption SST_OSC_KEEP_PHASE SEQUENCE must be used.

SST_OSC_PHASE_SEQUENCE

This option is not used by default, it must be specified with the .OPTION command:

.OPTION SST_OSC_PHASE_SEQUENCE= SEQ_1|SEQ_2|SEQ_3|SEQ_4|SEQ_5

• SEQ_1

The default sequence will be used: P1 ⇒ [Tr] ⇒ P2 ⇒ P3

• SEQ_2

Phase two (P2) is removed: P1 ⇒ [Tr] ⇒ P3

• SEQ_3

Phase one (P1) is removed and replaced by a short transient (Tr) phase: Tr ⇒ P2 ⇒ P3

• SEQ_4

Phase one (P1) and two (P2) are removed and replaced by a short transient (Tr) phase:Tr ⇒ P3

• SEQ_5

The continuation method will be used.Standard simulation methods might fail if the initial estimate of the fundamental frequencyis not close enough to the exact ones as is often the case with high-Q oscillators. A possiblesolution is to use the continuation method for the simulation of the steady-state of anoscillator. This method, based on a 2-dimensional homotopy, reaches the steady-state of anoscillator by following a curve that emulates the start-up of the circuit involved.



The sequences SEQ_3 and SEQ_4 are recommended for oscillators that are strongly non-linearbut with a short transient phase. In case of non convergence during the P3, the Tr phase can beimproved with the option SST_TRAN_NPER (see “SST_TRAN_NPER=val” on page 165). Besure that the initial conditions are correct to start oscillations during the transient phase (usingthe .IC command).

For more information run the file colpitts_phase_sequence.cir, located in the directory:$MGC_AMS_HOME/examples/rfic/

Limitations

• The option SST_OSC_PHASE_SEQUENCE can only be used on a single tone analysis.

• If you use .IC to start the oscillator, you must specify FUND_OSC_GUESS, otherwise thestability analysis will provide this estimated oscillation fundamental frequency.

Digital Block Analysis During Steady-State AnalysisA digital part of a circuit can be handled during steady-state analysis. During the steady-stateanalysis the digital part will remain constant and hold the DC value, except if a pre-transientphase is activated. In this case, during the pre-transient phase, the digital blocks will act as anormal .TRAN analysis and will hold the value they have at the last time point for the steady-state analysis.

Pre-Transient PhaseIn order to help the convergence of steady-state analysis, a pre-transient phase can beperformed. The last period of this pre-transient is then used to initialise the steady-state analysis.The pre-transient phase can be activated in the following ways:

• By default when the circuit contains a frequency divider.

• Forced by the option:

.OPTION SST_CONVERGENCE_HELP=transient

• Activated by the option:

.OPTION SST_AT_TIME=val

This option will set the transient duration.

The length of the pre-transient is, by default, 10 periods of the fundamental frequency, but canbe specified with the following option:

.OPTION SST_TRAN_NPER=val



The pre-transient phase is similar to the Eldo analysis .TRAN in the way they can both beconfigured by the same options or commands. This means that pre-transient phase is affected bythe Eldo command .IC and, for example, the options EPS, VNTOL and RETOL.

See Simulator Commands in the Eldo User’s Manual for more information on the .TRAN

and .IC commands and options.

Pre-Transient Phase and Sources

Periodic and non-periodic sources can be used during the pre-transient phase. For moreinformation please refer to “Source Usage in Pre-Transient Phase” on page 124.

Analysis Command Syntax.SST PLL


.SST PLLSteady-State PLL Analysis

Command Syntax

.SST PLL+ FUND1=real_value NHARM1=integer_value+ [FUND2=real_value NHARM2=integer_value]+ RANK=real_value

This command will perform a steady-state analysis of a PLL circuit, based on circuitpartitioning (one partition for the VCO, another partition for the frequency divider and onepartition for the rest (PFD, charge pump and loop filter)) and dedicated algorithms for thespecific partitions. The rank of the frequency divider can be specified. It is not necessary todefine the partition interfaces such as VCO control, VCO output and frequency divider output.The algorithm reconstructs them from the circuit and partition descriptions.

To use this command you must first specify each partition, using .RFBLOCK, and then use.SST PLL.

Note that the .SST PLL command supports the Steady-State (SST) and Phase Noise(SSTNOISE) analysis in the following particular cases:

• For PLLs with the reference input signal not specified at a fundamental frequency, but ata given harmonic of the fundamental frequency.

This may happen with rank N/M PLLs. In this case a divider by N is integrated into thePFD partition and the reference input signal is applied to harmonic N of the fundamentalfrequency.

• For PLLs with the reference input signal and the signal at the output of the dividerpartition not being at the same frequency (that is, not at the same harmonic of thefundamental frequency).

This may happen when the reference input signal is at the fundamental frequency, andthe signal at the output of the divider partition is at the second harmonic of thefundamental frequency.

Parameters

• PLL

Keyword specifying that a steady-state analysis of a PLL circuit is required.

• RANK

Specifies the rank of the frequency divider. Fractional PLLs are supported for this analysisby using the RF_FREQUENCY_DIVIDER macromodel. For the particular case of RANK=1PLLs it is not necessary to define a frequency divider partition, only two partitions areneeded: one partition for the VCO, and one partition for the rest (PFD, charge pump andloop filter).



• FUND1

Specifies the fundamental frequency of the circuit. See the parameter description in “.SST”on page 29.

• NHARM1

Specifies the maximum number of harmonics corresponding to the fundamental frequency,FUND1. See the parameter description in “.SST” on page 29.

• FUND2

Defines the output frequency. This functionality, and the use of the Flexible FrequencyDivider Macromodel, makes the .SST and .SSTNOISE analyses of fractional PLLs possible,see “SST PLL Analysis of a Fractional (non-integer rank) PLL” on page 42.

• NHARM2

Specifies the maximum number of harmonics corresponding to the output frequency,FUND2.

Notes

You can perform a steady-state noise analysis in the same simulation session by specifying a.SSTNOISE command after .SST PLL.

Example

.RFBLOCK NAME=VCO INST=XVCO

.RFBLOCK NAME=DIVIDER INST=XDIV

.SST PLL+ FUND1=240e6 NHARM1=160+ RANK=10

.SST PLL Run ProcedureThe following procedure describes how to run a dedicated steady-state PLL analysis withEldo RF from a netlist that is already running a transient analysis.

1. You must first partition the circuit into three (unless RANK=1, see RANK). Two of thepartitions are defined using the .RFBLOCK command. One partition for the VCO andanother for the divider. The Eldo RF algorithm will automatically detect the thirdpartition with the rest of the circuit (phase comparator, charge pump, and loop filter).See .RFBLOCK for details.

2. Specify a dedicated .SSTPROBE inside the VCO partition. See Probe Source for details.

3. Add a Voltage Controlled Voltage Source E (vcvs) between the output of the loop filterand the control voltage pin of the VCO. This source must be included in the VCOpartition. For example:

.RFBLOCK name=vco inst=(XVCO,EVCO)



4. Configure the .SST PLL analysis with fund1 equal to the reference frequency, rank

equal to the rank of the PLL, and Nharm1 the number of harmonics. (You must adjustNharm1 accordingly to take account of the rank.)

5. You can now run the simulation.

Notes

• If the simulation does not converge in the VCO partition, add optionSST_OSC_PHASE_SEQUENCE=seq_4. If necessary, adjust initial conditions to kick theVCO.

• If the algorithm does not converge in the first relaxation in the PFD or in the dividerpartition, you can:

o tune the pre-transient phase (increase tran accuracy and length)

o increase the number of harmonics in the .SST PLL command line

.SSTNOISE Run Procedure for PLLThe following procedures describe how to run a steady-state noise analysis for a PLL circuit.There are two ways to run an SSTNOISE analysis for PLL:

• compute the SSTNOISE with the PLL seen as a global partition

o Perform steps 1-4 in the .SST PLL Run Procedure.

o Add option SSTNOISE_GLOBPART=1 in the netlist. Specify the .SSTNOISEcommand with the harmonic parameter (HARM) set as a function of the SST PLLRANK parameter. For example:

.SSTNOISE v(out) harm(16) dec 3 1k 1Meg

o You can now run the simulation.

• compute the noise in each partition before computing the global noise in the completePLL

o Perform steps 1-4 in the .SST PLL Run Procedure.

o Add a Voltage Controlled Voltage Source E (vcvs) between the VCO and thedivider, and between the divider and the input of the phase comparator. The sourcebetween the VCO and the divider must be in the divider partition. The E sourcebetween the divider and the PFD must be in the PFD partition (default partition).

o Specify the .SSTNOISE command with the harmonic parameter (HARM) set as afunction of the SST PLL RANK parameter. For example:

.SSTNOISE v(out) harm(16) dec 3 1k 1Meg

o You can now run the simulation.



SST PLL Analysis of a Fractional (non-integer rank) PLLFor a PLL where RANK is non-integer (fractional), FUND1 specifies the input frequency andFUND2 defines the output frequency. For example:

.SST PLL+ RANK=19.9+ FUND1=45meg NHARM1=200+ FUND2='19.9*45meg' NHARM2=50

SST PLL analysis requires the addition of a Voltage Controlled Voltage Source (VCVS) Ebetween the input of a Flexible Frequency Divider Macromodel(RF_FREQUENCY_DIVIDER) and the output of the VCO, for example:

E_FREQDIV OUT_VCO_P 0 OUT_VCO 0 1Y_FREQDIV RF_FREQUENCY_DIVIDER PIN: OUT_VCO_P 0 NET036 0+ PARAM:+ INPUT_H_FUND1=0+ INPUT_H_FUND2=1+ OUTPUT_H_FUND1=1+ GAIN=1+ DIV_FACTOR=19.9+ DC_OFFSET=1.65

The complete example is given in the non-integer_PLL.cir file, located in the directory:$MGC_AMS_HOME/examples/rfic/RFBLOCK/

Analysis Command Syntax.SST STABIL


.SST STABILLocal Stability Analysis

Eldo RF performs a local stability analysis of the DC operating point. The analysis computesthe poles of the circuit and identifies whether some poles have a positive real part (leading toinstability).

It is possible to access the right most pole of the complex plane through plot and extractcommands, see “.PLOT/.PRINT SSTSTABIL” on page 89 and “Local Stability Analysis” onpage 113.

It is also possible to get an estimation of the quality factor for any autonomous circuit, see“.PLOT/.PRINT SSTSTABIL” on page 89.

Command Syntax

.SST STABIL

Parameters

• STABIL

Keyword specifying that a local stability analysis is required.

Notes

• This analysis can be specified together with a steady-state analysis, please refer to“.SST” on page 29.

• The result of this analysis is a report on the circuit stability: STABLE or UNSTABLE(together with an estimation of the oscillation frequency(ies)). This information is outputto the screen and also in the .chi file. When more than one potential oscillationfrequency is computed, they are displayed in the most probable order.

• Iterative techniques (Krylov subspace methods) allow the efficiency, capacity, androbustness of the DC stability analysis algorithm to handle large size circuits (>1000nodes). The CPU and memory cost is linear versus the circuit size.

• During a stability analysis, the frequency dependence of a device is not supported, onlythe value at a frequency of 0Hz is considered.


Analysis Command Syntax.SSTAC

.SSTACSteady-State AC Analysis

Command Syntax

.SSTAC DEC|OCT|LIN NP FSTART FSTOP+ [XAXIS=FREQ_COMMAND|FREQ_SPECTRUM].SSTAC LIST real_value real_value+ [XAXIS=FREQ_COMMAND|FREQ_SPECTRUM]

Parameters

• DEC|OCT|LIN

Keyword selecting a logarithmic, octave, or linear frequency sweep respectively.

• NP

Parameter specifying the number of frequency points per decade, per octave, or in thefrequency range depending on the specified keyword (DEC, OCT or LIN). This can beexpressed as a parameter.

• FSTART, FSTOP

Parameters that specify the minimum and maximum values of the frequency range. FSTARTshould be less than FSTOP, otherwise a warning will be issued and values of FSTART andFSTOP will be exchanged. These can be expressed as parameters.

• LIST

Means that the frequency points will be given explicitly in the command line following thatkeyword. real_value can be expressed as a parameter.

• XAXIS

Parameter that allows you to select whether the output should be plotted against thefrequencies specified on the command line (FREQ_COMMAND), or in the actual outputspectrum (FREQ_SPECTRUM). Default is FREQ_SPECTRUM.

Notes

• .SSTAC can follow a .SST or a .MODSST.

• Display of the following is supported during .SSTAC analysis, see .PLOT/.PRINTSSTAC:

o Extraction of large-signal/small-signal S-parameters

o Display of loop gain

o Group delay of outputs

Example

.SSTAC LIN 100 880MEG 909MEG XAXIS=FREQ_COMMNAND

Analysis Command Syntax.SSTAC


Please refer to Tutorial #6—Mixer Steady-State AC Analysis for a Gilbert Cell.


Analysis Command Syntax.SSTXF

.SSTXFSteady-State TF Analysis

Command Syntax

.SSTXF OIV DEC|OCT|LIN NP FSTART FSTOP+ [XAXIS=FREQ_COMMAND|FREQ_SPECTRUM].SSTXF OIV LIST real_val real_val+ [XAXIS=FREQ_COMMAND|FREQ_SPECTRUM]

Parameters

• OIV

Specifies the name of the output. It can be voltage node(s) or current through a voltagesource. The syntax is as follows: V(N1[,N2]) or I(VXX).

• DEC|OCT|LIN


• NP


• FSTART, FSTOP


• LIST

Means that the frequency points will be given explicitly in the command line following thatkeyword. real_value can be expressed as a parameter.

• XAXIS

Parameter that allows you to select whether the output should be plotted against thefrequencies specified on the command line (FREQ_COMMAND), or in the actual outputspectrum (FREQ_SPECTRUM). Default is FREQ_SPECTRUM.

Notes

• .SSTXF can follow a .SST or a .MODSST.

Please refer to Tutorial #7—Mixer Steady-State TF Analysis for a Gilbert Cell.

Analysis Command Syntax.SSTNLCONTRIB


.SSTNLCONTRIBSteady-State Non-Linear Contributors Analysis

The .SSTNLCONTRIB analysis identifies and ranks the non-linear devices that contribute toeither a specified voltage output or a specified current through a voltage source. The.SSTNLCONTRIB analysis can follow a Steady-State or Modulated Steady-State analysis.

By default the .SSTNLCONTRIB analysis results are printed in ASCII form in the .chi file. Youcan use the command option “SSTNLCONTRIB_FILE[=filename]” on page 174 to printthe results to a separate file.

Command Syntax

.SSTNLCONTRIB OUT [HARM(i[,j[,k]])]+ [SORT_REL=val]+ [SORT_ABS=val]+ [SORT_NBMAX=val]+ [SORT_NBHARM=val]+ [INCLUDE_DEVICES=device_list | EXCLUDE_DEVICES=device_list]+ [RESULTS=HIERARCHICAL | FLAT]+ [VIEW=SUMMARY | DETAILED]

Parameters

• OUT

Name of the output voltage node(s) or current through a voltage source, for whichnon-linear device contributions are computed. The syntax is as follows: V(N1 [, N2]) orI(Vxx).

• HARM(i [,j[,k]])

Specifies the intermodulation of the output for which non-linear device contributions arecomputed. Default value is HARM(0) corresponding to the output DC component.

• SORT_REL=val

Option used to limit the number of non-linear devices contributions that are printed.Contributions with absolute value below the maximum absolute non-linear devicecontributions times the SORT_REL value are not printed. Default value is 0.

• SORT_ABS=val

Option used to limit the number of non-linear devices contributions that are printed.Contributions with absolute value below the SORT_ABS value are not printed. Default valueis 0.01%.

• SORT_NBMAX=val

Option used to limit the number of non-linear devices contributions that are printed. At mosta number of contributions equal to the specified SORT_NBMAX value are printed. Thisparameter is applied to each level of the hierarchy when RESULTS=HIERARCHICAL. Bydefault there is no limit.



• SORT_NBHARM=val

Option used to limit the number of harmonic sub-contributions to be printed. This parameteris ignored when VIEW=SUMMARY. Default value is 3.

• INCLUDE_DEVICES=device_list

Contributions of non-linear devices belonging to the device families specified in thedevice_list are computed.Where:

device_list:= nonlinear_device , nonlinear_device

nonlinear_device:= RES|CAP|IND|DIODE|BJT|JFET|MOS|BHVL|SWITCH

These correspond to a non-linear resistor, capacitor, inductor, diode, BJT model, JFETmodel, MOS model, behavioural model, and switch capacitor respectively.

• EXCLUDE_DEVICES=device_list

Contributions of non-linear devices belonging to the device families specified in thedevice_list are not computed.

• RESULTS=HIERARCHICAL|FLAT

HIERARCHICAL

The non-linear subckt and non-linear device contributions are printed using the samehierarchy as the circuit. The results are written in decreasing order of magnitudeinside a given hierarchy level. Default.

FLAT

The non-linear device contributions are printed in decreasing order of magnitude.

• VIEW=SUMMARY|DETAILED

The contributions are given relative to the global contribution of all the non-linear devices tothe output, they are displayed in decreasing order of magnitude. A value of 100%corresponds to this global contribution. Positive (respectively negative) values indicate thatthe corresponding device tends to enhance (respectively to counteract) the global effect ofthe non-linear devices to the output.

SUMMARY

Only non-linear device contributions (and non-linear subckt contributions whenRESULTS=HIERARCHICAL) are printed. Default.

DETAILED

The non-linear device contributions and non-linear device pin contributions areprinted. This enables you to identify the non-linear device pins that effect the output.In addition, each pin contribution is expanded into harmonic contributions to allowthe identification of the harmonic components at which these non-linear contributionseffect the output.



NoteThe sorting parameters SORT_REL, SORT_ABS and SORT_NBMAX have cumulativeeffects.

Example

.sst fund1=flo nharm1=5 fund2=frf nharm2=5

.plot tsst v(out_p) v(out_n)

.sstnlcontrib v(out_p,out_n) harm(1,-1) sort_nbmax=4

In the above netlist the computation of the contributions of non-linear devices to the harmoniccomponent of voltage v(out_p, out_n) at frequency flo-frf at the output of a differentialmixer circuit is requested. The output is limited to the four most contributing non-linear devicesby the sort_nbmax parameter.

The corresponding results are displayed below:

**************************************************************

*** Eldo Steady-State Nonlinear Contributors Analysis ***

.SSTNLCONTRIB command # 1, Output: V(OUT_P, OUT_N) HARM(1,-1)

**************************************************************

subckt instance X1 : 100.00% device instance MRF1 : 37.26% device instance MRF2 : 37.26% device instance MCML4 : -12.55% device instance MCML2 : -12.55%

Contributions are given relative to the global contribution of all the non-linear devices to theoutput, they are displayed in decreasing order of magnitude. A value of 100% corresponds tothis global contribution. Positive (respectively negative) values indicate that the correspondingdevice tends to enhance (respectively to counteract) the global effect of the non-linear devicesto the output. In this example the instance MCML4 counteracts the global contribution of thenon-linear devices by 12.55%. Without instance MCML4 the global contribution would be12.55% greater.


Analysis Command Syntax.SSTSENSRLC

.SSTSENSRLCSteady-State RLC Sensitivity Analysis

The .SSTSENSRLC analysis computes the sensitivities of a specified voltage output or aspecified current through a voltage source to variations in the circuit linear passive elements(R,L,C). The .SSTSENSRLC analysis must follow a Steady-State or Modulated Steady-Stateanalysis.

By default, the .SSTSENSRLC analysis results are printed in ASCII format in the .chi file. Toprint the results to a separate file the option SSTSENSRLC_FILE can be used. A positive ornegative magnitude or phase sensitivity indicates that increasing the value of the correspondingR, L, or C element would increase or decrease the magnitude or phase of the specified steady-state output. Sensitivities are printed with MKSA units by default. Control over the units usedfor computing and printing the sensitivities is available.

By default, magnitude sensitivities are printed with MKSA units:

• Volt / Ohm for resistive load sensitivities (or Ampere / Ohm for a current output)

• Volt / Henry for inductive load sensitivities (or Ampere / Henry)

• Volt / Farad for capacitive load sensitivities (or Ampere / Farad)

• Volt or Ampere for mutual resistive or inductive coupling load sensitivities.

By default, phase sensitivities are printed with MKSA units:

• Degree / Ohm for resistive load sensitivities

• Degree / Henry for inductive load sensitivities

• Degree / Farad for capacitive load sensitivities

• Degree for mutual resistive or inductive coupling load sensitivities.

Command Syntax

.SSTSENSRLC OUT [HARM(I[,j[,k]])]+ [SORT_REL=value] [SORT_ABS=value] [SORT_NBMAX=value]+ [OUTPUT_TYPE=ABSOLUTE|SEMINORMALIZED|NORMALIZED]

Parameters

• OUT

Name of the output voltage node(s) or current through a voltage source, for which thesensitivities are computed. The syntax is: V(N1[, N2]) or I(Vxx).

• HARM(i[,j[,k]]])

Specifies the intermodulation of the output for which the sensitivities are computed. Defaultis HARM(0) corresponding to the output DC component.



• SORT_REL=val

Limits the number of sensitivities that are printed. Magnitude sensitivities with anabsolute value < maximum absolute sensitivity × SORT_REL are not printed. Default is 0, nolimit. Optional.

• SORT_ABS=val

Limits the number of sensitivities that are printed. The value specified is dependant on boththe circuit being simulated and the specific output voltage or current component for whichsensitivities are required. Magnitude sensitivities with an absolute value < SORT_ABS arenot printed. Default is 0, no limit. Optional.

• SORT_NBMAX=val

Specifies the maximum number of sensitivities that are printed. Default is 0, no limit.Optional.

NoteThe sorting parameters SORT_REL, SORT_ABS, and, SORT_NBMAX have cumulativeeffects.

• OUTPUT_TYPE==ABSOLUTE|SEMINORMALIZED|NORMALIZED

Specifies the units used for computing and printing the sensitivities. Default is ABSOLUTE.Optional.

ABSOLUTE

Sensitivities are printed with MKSA units. Default. Sensitivities are printed bycategory (resistive, inductive, capacitive, mutual resistive, and mutual inductive) andordered by decreasing absolute magnitude sensitivities inside each category.

SEMINORMALIZED

Sensitivities are printed in Volt or Ampere per percent of the corresponding R, L, C,or K parameter value. If a passive element has a zero parameter value, thecorresponding semi-normalized sensitivity cannot be computed. A warning messageis displayed.

NORMALIZED

Sensitivities are printed in percent per percent. Whenever the output magnitude orphase is zero, the normalized sensitivities cannot be computed. A warning message isthen displayed to inform that semi-normalized sensitivities are output instead.

Example

.sst fund1=100MegaHz nharm1=10

.plot tsst v(100)

.sstsensrlc v(100) harm(1) sort_abs=1.0e-10

.option sstsensrlc_file=amplifier.sens

In this example, the .SSTSENSRLC command is used to analyze how sensitivity variations inthe R, L, and C values will effect the steady-state output of an amplifier at the fundamental



frequency. The parameter SORT_ABS is set to a value that ensures that only the most importantsensitivities are printed to the output file, the value chosen is specific to this example. Thecorresponding results are written to file amplifier.sens as defined on the optionSSTSENSRLC_FILE. The output results are shown below:

************ SSTSENSRLC RESULTS ************

TEMPERATURE = 2.7000E+01 Celsius

*************************************************************************

*** Eldo Steady-State RLC Sensitivity Analysis ***

.SSTSENSRLC command # 1, Output: V(100) HARM(1)

*** Display format: Magnitude sensitivity (Phase sensitivity)

**************************************************************************

Resistive Load Sensitivity Resistor Name

-5.744268e-07 (-3.081122e-01) REE2 7.618738e-08 ( 1.011905e-02) REE1 -4.905675e-08 (-2.369411e-02) REE3 -1.201759e-09 (-2.861953e-03) L1 impedance resistor -1.201759e-09 (-2.861953e-03) RL 7.752949e-10 ( 6.345923e-04) L2 impedance resistor -3.387182e-10 (-3.866532e-05) R2

Resistive Coupling Load Sensitivity Mutual Coefficient Name

-2.048043e-04 ( 6.027684e+01) K12 impedance resistive coupling

Inductive Load Sensitivity Inductor Name

1.229433e+01 (-1.364167e+05) L1 impedance inductor 1.100682e+00 (-5.455549e+04) L2 impedance inductor

Inductive Coupling Load Sensitivity Mutual Coefficient Name

2.619691e-06 (-9.296004e-03) K12 impedance inductive coupling

Capacitive Load Sensitivity Capacitor Name

-6.191644e+06 (-3.688471e+10) C1 2.282739e+04 (-3.614700e+10) C2 2.950268e-08 (-4.083738e-03) CEX

Analysis Command Syntax.SSTNOISE


.SSTNOISESteady-State Noise Analysis

The .SSTNOISE analysis computes the noise generated by the circuit at one output around oneharmonic. This noise is the sum of all the contributions from all the noisy devices in the circuit.The .SSTNOISE analysis is different from a .NOISE analysis in the sense that it includesfrequency conversion effects. In a conventional .NOISE analysis, the circuit is in a DCoperating point and all the noise sources as well as their contributions to the output noise are atthe same frequency. In a .SSTNOISE analysis, the circuit is in a periodically time-varyingoperating point. This means that, compared to the .NOISE analysis, two new effects willimpact noise results. First, the noise sources that depend on bias (thermal and flicker noise inMOS transistors, shot noise in diodes or BJTs... for instance) are modulated by the time varyingoperating point. Second, the transfer functions from the noise sources to the output are alsoperiodically time-varying. Therefore the contributions of the noise sources to the output are alsomodulated. The consequence of these modulations is a transposition or a conversion of the noisespectra around all the harmonics of the large signal steady-state.

A different algorithm is used to solve the Steady-State Phase Noise of PLL circuits. This isbased on circuit partitioning (same as the one used in the .SST PLL analysis). Compared to theprevious solution for phase noise computations of PLL (on the complete circuit) this approach ismuch more efficient (2× to 50× faster) and more accurate (especially at low frequency offset).This algorithm requires a VCVS between the VCO and the divider partitions, and anotherVCVS between the divider and the PFD partitions. When an .SSTNOISE is specified and thenetlist contains these two VCVS then this algorithm is automatically activated. It can bedeactivated by using the option SSTNOISE_GLOBPART=1.

Command Syntax

.SSTNOISE OUT [Input_src [INPUT_HARM (i[,j[,k]])]] [HARM (i[,j[,k]])]+ DEC|OCT|LIN NP FSTART FSTOP+ [XAXIS=FREQ_COMMAND|FREQ_SPECTRUM]+ [CONTRIB_LIST real_val real_val].SSTNOISE OUT [HARM (i[,j[,k]])] [Input_src] [INPUT_HARM (i[,j[,k]])]+ LIST real_val real_val+ [XAXIS=FREQ_COMMAND|FREQ_SPECTRUM]+ [CONTRIB_LIST real_val real_val]

NoteIt is possible to specify multiple .SSTNOISE commands. These different command linesmay consider different noise outputs or different noise frequencies or different noisesidebands.For more information run the example named vco_div2.cir, located in the directory:$MGC_AMS_HOME/examples/rfic/



Parameters

• OUT

Name of the output voltage node(s) or current through a voltage source where the noise iscomputed. The syntax is as follows:V(N1[,N2]) orI(Vxx) specifies the argument (Vxx) as a voltage source.

• Input_src

Name of the input voltage or current source for which the equivalent input noise is to becalculated. Optional.

• INPUT_HARM

Specifies the intermodulation of the harmonic at which the equivalent input noise iscomputed. Default value is HARM(0).

• HARM

Keyword specifying that the noise frequencies are defined relative to a harmonic of theSteady-State analysis (offset from the carrier).

• (i[,j[,k]])

Specifies the intermodulation of the corresponding harmonic (origin of the noisefrequencies). Default value is HARM(0) corresponding to DC.

The noise is computed at the specified output for the specified offset frequencies relative tothe large signal harmonic (specified by HARM).

In the case of phase noise computation, HARM has to be different from (0) as phase noise isnull around DC. We suggest you specify HARM at the harmonic where there is maximumpower.

For example, a circuit is composed of a VCO followed by a frequency divider by N. If thespecified output of the .SSTNOISE analysis command is the output of the VCO then specifyHARM (N). But if the specified output of the .SSTNOISE analysis command is the output ofthe divider then specify HARM (1).

• DEC|OCT|LIN


• NP


• FSTART, FSTOP




• CONTRIB_LIST

Keyword specifying that the SSTNOISE result printing will be done only for a subset offrequencies that are defined by the real_val values. The ASCII information printedconcerns the individual contributions of noisy devices.

• LIST

Keyword specifying that the frequency points will be specified explicitly.

• real_val

List of frequency point values, specified together with the CONTRIB_LIST or LISTkeywords. The values can be expressed as parameters.

• XAXIS

Parameter that allows you to select whether the output should be plotted against thefrequencies specified on the command line (FREQ_COMMAND), or in the actual outputspectrum (FREQ_SPECTRUM). Default is FREQ_SPECTRUM however, if a phnoise output isrequested (Sphi or lf) the default value is FREQ_COMMAND.

The .SSTNOISE analysis results can be plotted or printed with the .PLOT or .PRINTcommands (see “Steady-State Noise Analysis Results” on page 81). Eldo RF also prints inASCII form the individual contributions of the noisy devices in the .chi file (see “ASCII OutputFile (.chi) Results” on page 87) or in a separate file (specified with an option, see“SSTNOISE_FILE[=filename]” on page 167). You can define whether individualcontributions of SPHI and AMNOISE are printed as DSB results or SSB results by specifying“SSTNOISE_CONTRIB_TYPE=SSB|DSB” on page 167. Default is DSB. It is also possible todefine the contributions from device families using options“SSTNOISE_INCLUDE_DEVICES=device_type” on page 167 and“SSTNOISE_EXCLUDE_DEVICES=device_type” on page 167.

Oscillator jitter information can be computed and displayed for .SSTNOISE analysis. Twotypes of jitter can be computed: long time jitter and the period jitter. They are computed fromthe phase noise spectrum. Long time jitter is available through the keyword LT_JITTER for the.PLOT and .PRINT commands (see “Steady-State Jitter Results” on page 88). The period jitteris available through the keyword PERIOD_JITTER for the .EXTRACT command (see “Steady-State Jitter” on page 113).

Notes

• .SSTNOISE can follow a .SST or a .MODSST.See “Tutorial #11—Phase Noise Extraction for an Oscillator” on page 343(vco_phnoise.cir) for an example of .SSTNOISE after .SST. Run examplevco_phnoise_modsst.cir for an example of .SSTNOISE after .MODSST.

• LDTL devices can be handled as noise contributors in .SSTNOISE analysis. The noisecontribution is calculated using the Twiss formula, see “Twiss Formula” on page 204.

• F block devices (those defined by S, Y or Z parameters in a “Touchstone” file format)can be handled as noise contributions in .SSTNOISE analysis. The noise contribution is



calculated using the noise parameters when defined in the Touchstone file format,otherwise the Twiss formula is used.

Related Options

An alternative algorithm is available in order to speed-up SSTNOISE analysis. This can beactivated with the IMPROVED_SSTNOISE_PERF option, see “IMPROVED_SSTNOISE_PERF”on page 167.

Analysis Command Syntax.SNF


.SNFSteady-State Noise Analysis—Spot Noise Figure

Command Syntax

.SNF INPUT=(List_of_devices) OUTPUT=(List_of_devices)+ [INPUT_TEMP=val] [INPUT_SIDEBAND=(List_of_sidebands)]

The Noise Figure (SNF) is calculated in the following way:

See .SNF of the Eldo User’s Manual for more information.

In Eldo RF, the Output_Noise_due_to_Source contains the contributions from the sidebandsspecified with the INPUT_SIDEBAND parameter.

NoteThe List_of_sidebands are only considered in the .SSTNOISE analysis and ignoredfor .NOISE analysis.

Therefore, the default Noise Figure .SNF in Eldo RF is defined as:

Please refer to Tutorial #5—Mixer Steady-State and Noise Analysis for a Gilbert Cell forfull calculations based on a working example, and Chapter 15, “Eldo RF Tutorial—MixerSimulations” for a special case of mixer noise and noise figure (NF) simulations.

It is also possible to compute Noise Figures using the IEEE definition. This is activated bydefining the INPUT_TEMP parameter. The noise figure is calculated in different ways accordingto whether or not this parameter is defined (see below).

Parameters

• List_of_devices

As defined in the Eldo User’s Manual. This can contain wildcard ‘*’ characters anywhere inthe name. Each item must be separated with a space or a comma. Brackets are optional.However, if more than one name is provided, then commas and brackets must be used.

SNF Total_Output_Noise Output_Noise_due_to_Load–Output_Noise_due_to_Source

---------------------------------------------------------------------------------------------------------------------------=

SNF Total_Output_Noise Output_Noise_due_to_Load–Output_Noise_due_to_Source_at_specified_Sidebands------------------------------------------------------------------------------------------------------------------------------------=



• INPUT_TEMP=val

As defined in the Eldo User’s Manual. Parameter used to define the temperature (in degreesCelsius) of the source devices (INPUT). This parameter activates the IEEE definition in thecalculation of the Noise Figure. See IEEE Noise Figure below for details. If INPUT_TEMP isnot specified then the temperature value of the circuit is used.

• List_of_sidebands

Contains the list of the side bands used in the computation ofOutput_Noise_due_to_Source. Each side band is defined by the corresponding harmonicindexes inside brackets. The different side bands are separated with a space or a comma.Harmonic indexes may contain wildcard characters, in which case all matching indexes willbe contributing. Default value of the input_sideband parameter is (*,*).

IEEE Noise FigureThe IEEE Noise Figure is defined as:

By providing a parameter (INPUT_TEMP) we allow the possibility to define the temperature ofthe source to use the IEEE definition if INPUT_TEMP is specified to be 27°C.

• When INPUT_TEMP is defined:

o Total_Output_Noise=

Total output noise minus the contribution of the source devices, plus the noisegenerated by the source devices at the frequencies defined by INPUT_SIDEBAND andat temperature defined by INPUT_TEMP.

o Output_Noise_due_to_Load=

Contribution to the output noise of the noise generated by the Load devices at theoutput frequency.

o Output_Noise_due_to_Source=

Contribution to the output noise of the noise generated by the source devices at thefrequencies defined by INPUT_SIDEBAND and at temperature defined byINPUT_TEMP.

• When INPUT_TEMP is not defined:

o Total_Output_Noise=

Total output noise.

o Output_Noise_due_to_Load=

Contribution to the output noise of the noise generated by the Load devices at theoutput frequency.

SN F IEEETotal Noise Load Noise–

Source Noise at RF Frequency at 27C-------------------------------------------------------------------------------------------=



o Output_Noise_due_to_Source=

Contribution to the output noise of the noise generated by the source devices at thefrequencies defined by INPUT_SIDEBAND.

Example

.SNF INPUT=Rin OUTPUT=Rout INPUT_SIDEBAND=((1,0) (-1,*))

.SNF input=(VBB1)+ output=(ROUT1N, ROUT1P)+ input_temp=27+ input_sideband=( (-1, 0) (1, 0) )


Analysis Command Syntax.WCASE

.WCASESteady-State Worst Case Analysis

The syntax is similar to the standard Eldo syntax, except that the analysis type is replaced bySST.

.WCASE SST [OUTPUT=MIN|MAX|BOTH]+ [VARY=LOT|DEV|BOTH] [TOL=VAL] [ALL]

Parameters

• SST

Specifies a steady state analysis. An analysis type MUST be specified.

Worst Case Analysis computes worst case values for waveform data extracted using the.EXTRACT command.

See .WCASE of the Eldo User’s Manual for more information.

Analysis Command Syntax.MC


.MCSteady-State Monte Carlo Analysis

The syntax is used in exactly the same way as for the general Eldo Monte Carlo analysis.

.MC RUNNO [OUTER] [OV] [SEED=integer_value] [NONOM] [ALL]+ [VARY=LOT|DEV] [IRUN=val] [NBBINS=val] [ORDMCS] [MCLIMIT]

The Monte Carlo system may be implemented for Steady-State analysis and is useful to obtainstatistical information derived from estimates of the random variability of all circuitcomponents. The Monte Carlo analysis system carries out multiple simulation runs, each runusing model and device values differing from the nominal one within the specified tolerancelimit, the variation being a simulated random variable satisfying a specified distribution(uniform, Gaussian, or user-defined).

See .MC of the Eldo User’s Manual for more information.


Analysis Command Syntax.MODSST

.MODSSTModulated Steady-State Analysis

The .MODSST command activates Modulated Steady-State analysis (MODSST analysis).

Command Syntax

.MODSST+ MODSST_TPRINT MODSST_TSTOP [SSTUIC] [UIC]

Parameters

• MODSST_TPRINT

Printing or plotting increment for the printer output (in seconds). Also used to compute adefault HMAX value. This can be expressed as a parameter.

• MODSST_TSTOP

The Modulated Steady-State analysis duration in seconds. This can be expressed as aparameter.

• SSTUIC

Specifies that a SST is computed prior to the MODSST analysis.

• UIC

Specfies you do not want Eldo RF to solve for the quiescent operating point beforebeginning the transient analysis. Eldo automatically initializes all the node voltages itself aswell as any user-defined initial node voltages included in a .IC command. The UIC optionis recommended for the simulation of astable or very large digital circuits.

The first time point of the MODSST analysis (corresponding to time=0) is by default thesolution of the DC analysis. However, it is also possible to start from another circuit state. Thiscan be a steady-state saved in a file and reinjected with the .RESTART command. It is alsopossible to specify Eldo RF to perform a steady-state and to start the MODSST from this point.This is invoked by specifying the SSTUIC keyword.

For information on the MODSST analysis options available please refer to the sectionMODSST Analysis Options.

Constraints

• .MODSST analysis must be specified in conjunction with a .SST command, otherwise,the MODSST analysis will be aborted.

• Modulated steady-state analysis with multi-oscillation frequency circuits is onlypossible when starting from a steady-state solution a t=0. A steady-state solution isprovided through either a .RESTART SST command or by performing a steady-stateanalysis prior to the modulated steady-state analysis (using the SST_UIC flag with the

Analysis Command Syntax.MODSST


.MODSST command). If a .RESTART is not specified, Eldo RF will perform a steady-state analysis to initialize the modulated steady-state simulation.

• When running a modulated multi-oscillation simulation, the oscillation frequencies aretime-dependent variables. Hence Eldo RF enables you to choose reference frequenciesto represent the results from FFT computations (i.e. results from the use of .OPTFOURcommand).

Example

.sst oscil fund_osc_guess1=1.9G nharm_osc1=10

.modsst 1n 1u

.plot tmodsst fund_osc

The example above runs an oscillator analysis defined by the .SST OSCIL command, tenharmonics will be computed in the analysis. The estimated oscillation fundamental frequency is1.9GHz. The .MODSST command activates a Modulated Steady-State analysis (MODSST),with a printing or plotting increment for the printer output of 1n second. The MODSST analysiswill stop after 1µs. The command .PLOT TMODSST will generate the curves of fund_osc inthe time domain.


Analysis Command Syntax.PART MODSST

.PART MODSSTMODSST Circuit Partitioning

Eldo RF and ADMS RF have the capability to use simultaneously the transient algorithm andthe MODSST (MODulated Steady-State) algorithm. Transient and MODSST algorithms areused selectively for specified subcircuit instances.

In order to use this feature, you have to tell Eldo which instances have to be simulated using theMODSST algorithm. The .PART MODSST command instructs Eldo to use the MODSSTalgorithm in place of the regular transient algorithm, for a certain selection of instances.

If the command .PART MODSST is specified multiple times, Eldo RF will group all instancesand subckts into the same partition.

Command Syntax

.PART MODSST+ INST=<instance_name> SUBCKT=<subcircuit_name>

The instances to be partitioned may be listed explicitly, using the <instance_name>. They mayalso be implicitly designated, using the <subcircuit_name>.

Parameters

• MODSST

Instructs Eldo to use the MODSST algorithm. Other algorithms may be specified, see.PART of the Eldo User’s Manual.

• instance_name

Name of the instance to be partitioned. If multiple instances are specified they must beseparated using a comma and the list must be enclosed in parenthesis (see Example).Instance names may also contain wildcard characters (* and ?).

• subcircuit_name

Name of the subcircuit to be partitioned. If multiple subcircuits are specified they must beseparated using a comma and the list must be enclosed in parenthesis (see Example).Subcircuit names may also contain wildcard characters (* and ?).

Example

.PART MODSST INST=(XTOP.XLNA, XTOP.XM.XMIXER?) SUBCKT=(VCO*)

In this example instance XTOP.LNA, all instances whose name matches XTOP.XM.XMIXER?, andall instances whose type name (the name of their associated subcircuit) matches VCO*, will besimulated using the MODSST algorithm. The rest of the circuit will be simulated using thetransient algorithm.

The Eldo kernel takes care of the “connections” between the partitions simulated withMODSST and those simulated with transient. No “converters” are required. Sources such as the



FOUR sources and the Eldo RF digitally modulated sources are automatically handled by theMODSST algorithm, even if they do not belong to a MODSST instance (as selected by the.PART command).

The MODSST algorithm is much more effective than the transient algorithm for high speed“RF” signals. Typically the signals running in the first stages of an RF receiver, such as LowNoise Amplifiers, Mixers and VCOs, are better handled by the MODSST algorithm, whichmanages to abstract the high-speed RF carriers that cause a major slowdown in transientsimulation. See the description of the “Algorithm for Modulated Steady-State Analysis” onpage 21 and its potential benefits.

However, for low speed blocks (typically in the blocks which process down-converted signals),it may be more efficient to still use the transient algorithm, particularly if the number of activedevices and/or nodes is large.

The partitioning scheme allowed by the .PART command allows using the best algorithm on aper-block basis, to optimize the simulation efficiency.

For example, let’s take the case of the simulation of an RF receiver chain with LNA, SAWfilter, mixer and VCO, Low-Pass filter, A-to-D converter, and digital demodulation. The mosteffective combination might be to select the MODSST algorithm for the LNA, SAW filter,Mixer and VCO, and to leave the rest (LP filter, A-D converter and digital demodulation) to thetransient algorithm. Note that handling the digital demodulation (in Verilog or VHDL) wouldrequire to run an ADMS RF simulation actually (for more information, see the “ADMS RFTutorial—AGC Loop” on page 445).

In order to use this feature, the command file must contain a .MODSST command to specify thesimulation duration, and its associated .SST command to define the fundamentals (see “.SST”on page 29 and “.MODSST” on page 62). It is not possible to use this feature if a .TRANcommand is used.

To visualize the simulation results, both .PLOT TRAN... and .PLOT FMODSST|TMODSST...commands may be used. Of course .PLOT TMODSST|FMODSST commands may only be used forquantities that belong to a MODSST partition.

Using the .wdb output, all results (TRAN and MODSST) are written to the same output file,circuit.wdb. The JWDB format allows different time bases for the waveforms.

Notes

• As this feature uses the MODSST or SST algorithm, it requires the Eldo RF option(license). If used inside Questa ADMS, it requires the Eldo RF option and the ADMSRF option.

• If using the .cou output format, Eldo cannot dump transient and MODSST informationto the same file, because the .cou format allows a unique common time base only. Forthis reason the TRAN outputs and the TMODSST|FMODSST outputs are written to



separate output files. The TRAN outputs are written to circuit.cou, whereas theMODSST outputs are written to circuit_sst.cou. It is always possible to overlay theseresults in the waveform viewer.

Related Options and Command

A true RF-Analog separation mode can be implemented using the RF_PARTITIONING_MODE

option, see “RF_PARTITIONING_MODE=STANDARD|FAST” on page 169.

A threshold for the mode is set using the RF_PARTITIONING_THRESHOLD option, see“RF_PARTITIONING_THRESHOLD” on page 170.

Steady-State partitioning can be performed using the command .RFBLOCK.

Analysis Command Syntax.RFBLOCK


.RFBLOCKSteady-State Circuit Partitioning

Eldo RF and ADMS RF have the capability to partition a netlist into individual partitions,compute the steady-state for each partition, then perform the steady-state analysis for thecomplete netlist to find the steady-state solution. You should list all devices that have the samefunctionality and frequency spectrum in the same SST partition, this will help convergence andreduce the simulation time.

The .RFBLOCK command can be used in conjunction with the .SST and .SSTNOISEcommands and is required when using the .SST PLL command.

If the command .RFBLOCK is specified multiple times, Eldo RF will create separate partitionseach with the name block_name as defined on the .RFBLOCK command.

Command Syntax

.RFBLOCK+ NAME=block_name TYPE=circuit_type+ INST=instance_name | SUBCKT=subcircuit_name+ OPTIONS=(option1 option2 ...)

The instances to be included in the SST partition block_name may be listed explicitly usingthe instance_name. They may also be implicitly designated, using the subcircuit_name.

NoteYou must specify at least one instance or subcircuit to be partitioned, either aninstance_name or a subcircuit_name. You can specify both an instance_name anda subcircuit_name on the same command using the syntax above.You can also specify a name to cover the parts of the circuit which remain unpartitionedat the beginning of simulation (named OTHER by the simulator when it provides the listof the partitions). For this particular usage of the command, it must be defined after all theother .RFBLOCK command definitions, and the only other parameters you can specify areTYPE and OPTIONS, that is, INST and SUBCKT must remain unspecified.

Parameters

• NAME=block_name

Name of the Steady-State partition. See also note above.

• TYPE=circuit_type

One of the circuit types listed in Table 5-13 on page 172 which, if specified, causes Eldo RFto automatically select a set of options appropriate for the partition.

• INST=instance_name

Name of the instance to be included in the SST partition. If multiple instances are specifiedthey must be separated using a comma and the list must be enclosed in parenthesis (see



Example). Instance names may also contain wildcard characters (* and ?). Optional,however, you must specify at least one instance or subcircuit to define an SST partition.

• SUBCKT=subcircuit_name

Name of the subcircuit to be included in the SST partition. If multiple subcircuits arespecified they must be separated using a comma and the list must be enclosed in parenthesis(see Examples). Subcircuit names may also contain wildcard characters (* and ?). Optional,however, you must specify at least one instance or subcircuit to define an SST partition.

• OPTIONS=(option1 option2 ...)

One or more options from the following list:

o SST_CONVERGENCE_HELP=TRANSIENTSST_CONVERGENCE_HELP=ADVANCED_NEWTON, seeSST_CONVERGENCE_HELP=TRANSIENT|NO_TRANSIENT|CONTINUATION|ADVANCED_NEWTON|PSEUDO_MODSST

o SST_TRAN_NPER=val

o SST_AT_TIME=val

o SST_NDIM_FFT

o SST_USE_NTONE_PROCEDURE

o SST_OVRSMP

o SST_MAX_LINITER=VAL

o SST_TRAN_NPER=val

o SST_PRECONDITION=ADAPTIVE|TIME|TIME_MODERATE|TIME_ACCURATE|TIME_FOR_NOISE_ONLY|TIME_MODERATE_FOR_NOISE_ONLY|TIME_ACCURATE_FOR_NOISE_ONLY

o HMAX (from the Eldo User’s Manual)

o EPS (from the Eldo User’s Manual)

Note that when multiple options are specified they must be separated by a space, not by acomma.

Examples

.RFBLOCK NAME=VCO INST=(XHB1.XPOL, XHB1.XVCO)

.RFBLOCK NAME=DIVIDER2 INST=XHB1.XDIV2

.RFBLOCK NAME=LNA INST=XHB1.LNA7

In this example a VCO, XVCO, a frequency divider, XDIV2, and an LNA, LNA7, are defined asindividual partitions to be simulated with the SST algorithm.

.RFBLOCK NAME=VCO INST=(XHB1.XPOL, XHB1.XVCO)+ OPTIONS=(SST_CONVERGENCE_HELP=ADVANCED_NEWTON).RFBLOCK NAME=other OPTIONS=(SST_MAX_LINITER=100)



In this example a VCO, XVCO, is defined as an individual partition to be simulated with the SSTalgorithm using Advanced Newton enhancement. The rest of the circuit (which has been giventhe name “other”) is to be simulated with the SST algorithm with up to 100 iterations.


Analysis Command Syntax.CHRSIM

.CHRSIMInput from a Prior Simulation

The .CHRSIM command enables the output of a previous simulation to drive a node in thecurrent simulation. The previous simulation data can be generated from either a Steady-Stateanalysis in the frequency domain or a Modulated Steady-State analysis in the time-frequencydomain. An output file that was generated from a transient analysis can be used to drive a nodein the current simulation if either a Modulated Steady-State analysis or an SST analysis with apre-transient phase specified. The previous simulation data can either be in the form of a .wdbfile or a .cou file.

Command Syntax

.CHRSIM FSST|FMODSST IN V(OUT)[.h(i[,j,...])] FILE [FORMAT=WDB|COU]+ [output_h=(i[,j,...])] [TSIM=val]

Parameters

• FSST

Specifies the format to be read was generated in a Steady-State analysis in the frequencydomain.

• FMODSST

Specifies the format to be read was generated in a Modulated Steady-State analysis in thetime-frequency domain.

• in

Name of the input node for the current simulation. Mandatory.

• v(out)

out is the name of the node previously simulated in file. Mandatory.

• .h(i[,j,...])

Defines the harmonic(s) that will be read from the file file. Only to be used with theFMODSST format. If omitted all the saved harmonics will be read.

• FILE

Name of the file that contains the simulation data. The file extension is omitted. Mandatory.

• FORMAT=WDB|COU

Specifies the format of the file that contains the simulation data. Default is WDB. Optional.

• output_h=(i[,j,..])

Defines the mapping between the harmonic that are read in from the output file and theharmonics in the current design. Only to be used with the FMODSST format. If output_h

is omitted and a .MODSST analysis is specified, each harmonic read will be mapped to theclosest corresponding harmonic for the current simulation. When a .SST analysis isspecified output_h will be ignored. Optional.



• TSIM

Defines the time instant in the .wdb file that the corresponding steady-state simulation willstart from. Can only be used when the format is FMODSST and the current analysis is.SST. Optional.

Use of .CHRSIM with FSST Data in a .SST Analysis

The following netlist is used to generate the .wdb file that will be used to drive a node in asubsequent simulation:

vin 1 0 four fund1 ma (1) 1 0rin 1 0 1k.sst fund1=1g nharm1=2.plot fsst v(1).end

The file tap1.wdb that was generated by the previous netlist is used on the .chrsim commandto drive node 2 in the following netlist:

.chrsim FSST 2 v(1) tap1 format=wdbrin 2 0 1k.sst fund1=1g nharm1=2.plot fsst v(2).plot tsst v(2).end

Use of .CHRSIM with FMODSST Data in a .SST Analysis


vin 1 0 four fund1 ma (1) 1 0rin 1 0 1k.modsst 0 10n.sst fund1=1g nharm1=2.plot fmodsst v(1).end

The file rap1.wdb that was generated by the previous netlist is used on the .chrsim commandto drive node 2 in the following netlist:

.chrsim FMODSST 2 v(1) rap1 format=wdb tsim=4.3nrin 2 0 1k.sst fund1=1g nharm1=2.plot fsst v(2).plot tsst v(2).end

The data will be extracted from the timepoint 4.3ns in the .wdb file.

Use of .CHRSIM with FSST Data in a .MODSST Analysis




vin 1 0 four fund1 ma (1) 1 0rin 1 0 1k.sst fund1=1g nharm1=2.plot fsst v(1).end

The file zap1.wdb that was generated by the previous netlist is used on the .chrsim commandto drive node 2 in the following netlist:

.chrsim FSST 2 v(1) zap1 format=wdbrin 2 0 1k.modsst 0 10n.sst fund1=1g nharm1=2.plot fmodsst v(2).option modsst_full_display=2.plot tmodsst v(2).end

Use of .CHRSIM with FMODSST Data in a .MODSST Analysis


vin 1 0 four fund1 ma (1) 1 0rin 1 0 1k.modsst 0 10n.sst fund1=1g nharm1=2.plot fmodsst v(1).end

The file map1.wdb that was generated by the previous netlist is used on the .chrsim commandto drive node 2 in the following netlist:

.chrsim 2 v(1).h(1) map1 format=wdbrin 2 0 1k.modsst 0 10n.sst fund1=1g nharm1=2.plot fmodsst v(2).option modsst_full_display=1.plot tmodsst v(2).end

Only the first harmonic will be read in from the .wdb file.

Analysis Command Syntax.AGE


.AGEAge Analysis

The .AGE reliability analysis command is supported with .SST (forced and autonomouscircuits), .SSTAC, .SSTNOISE and .SSTXF analyses, but it is not compatible with modulatedsteady-state analyses, .MODSST and .PART MODSST.

The computation of stresses is performed over the reference period corresponding to the firstfundamental frequency for non-autonomous circuits, and over the oscillation period forautonomous circuits.

Note.AGE steady-state reliability simulation is incompatible with RFBLOCK usage, both forpartitioned steady-state and for PLL steady-state simulations.

Command Syntax

Refer to .AGE in the Eldo User’s Manual, but note that the following parameters, which areonly relevant to transient reliability simulation, are ignored:

• [TSTART=value]

• [TSTOP=value]

• [TWINDOW=(a1,b1) (an,bn)]

References

For further information about the User Defined Reliability Model (UDRM), refer to the EldoUDRM User’s Manual.


Analysis Command Syntax.OP RF

.OP RFDC Operating Point Calculation for RF

This command determines the DC operating point. It differs from the Eldo .OP command byusing an average value of the source, even if a DC value is specified.

If .OP and .OP RF are specified in the same netlist, .OP RF is ignored.

Command Syntax

.OP RF [[KEYWORD] T1 [KEYWORD] TN]

.OP RF TIME=VAL|END [STEP=VAL] [TEMP=VAL]

.OP RF DC=VAL [DC2=VAL] [STEP=VAL] [TEMP=VAL]

Parameters

Refer to the .OP command description in the Simulator Commands chapter of the Eldo User’sManual.

Use of .OP RF in a .SST Analysis

See “DC Operating Point Calculation for RF” on page 31.


Chapter 3Display Command Syntax

SummaryThe following table shows the display commands available in Eldo RF.

Separate folders are created in the JWDB database for RF analysis results. For example, FSSTanalysis results will be stored in a folder named “FSST”. Disabled this functionality byspecifying the flag -jwdb_norffolder when invoking Eldo RF (see the Eldo User’s Manual). RFanalysis results will then be stored in the AC and TRAN folders.

Table 3-1. Display Commands

Description Command

Steady-State Analysis Results .PLOT/.PRINT FSST/TSST

Steady-State AC Analysis Results .PLOT/.PRINT SSTAC

Steady-State TF Analysis Results .PLOT/.PRINT SSTXF

Steady-State Noise Analysis Results .PLOT/.PRINT SSTNOISE

Jitter Results From a Steady-State Noise Analysis .PLOT/.PRINT SSTJITTER

Local Stability Analysis Results .PLOT/.PRINT SSTSTABIL

Modulated Steady-State Analysis Results .PLOT/.PRINT FMODSST/TMODSST

Contours .PLOT CONTOUR

Extract Command .EXTRACT


Display Command SyntaxDisplay Command Descriptions

Display Command Descriptions.PLOT/.PRINT FSST/TSSTSteady-State Analysis Results

Results from a .SST analysis (see “.SST” on page 29) can be plotted or printed with the .PLOT

or .PRINT commands.

Frequency Domain

The syntax is similar to the syntax for AC analysis output, except that the AC keyword isreplaced by FSST. For further details, see the .PLOT command or the .PRINT command ofthe Eldo User’s Manual.

.PLOT FSST OVN OVN

.PRINT FSST OVN OVN

The curves generated by the FSST mode of the .PLOT command in the .wdb file are displayedwith EZwave in a spectral representation.

It is also possible to display Power dissipated in dipoles and any circuit device. The syntax is asfollows:

.PLOT FSST PM(dev_name)|PDB(dev_name)|PDBM(dev_name)

.PRINT FSST PM(dev_name)|PDB(dev_name)|PDBM(dev_name)

• PM

Represents the power in Watts.

• PDB

Represents the power in dB (0dB corresponds to 1W).

• PDBM

Represents the power in dBm (0dBm corresponds to 1mW).

Time Domain

The syntax is similar to the syntax for TRAN analysis output, except that the TRAN keyword isreplaced by TSST. For further details, see the .PLOT command or the .PRINT command ofthe Eldo User’s Manual.

.PLOT TSST OVN OVN

.PRINT TSST OVN OVN

For single-tone signals, the default values of parameters of the time domain waveform is twoperiods, with a minimum of 64 points per period. For more information please refer to TimeDomain.

Display Command Syntax.PLOT/.PRINT FSST/TSST


For multi-tone signals, the default value is two periods of the lowest fundamental frequency,with a minimum of 64 points per period. For more information please refer to Time Domain.

NotePlotting or printing TSST outputs may require a large CPU time in the case of multi-tonesimulation when the lowest frequency is much lower than the high frequency tones.

The number of periods and number of points can be changed with the SST_NPER and SST_NPT

options described in “SST_NPER” on page 156.


Display Command Syntax.PLOT/.PRINT SSTAC

.PLOT/.PRINT SSTACSteady-State AC Analysis Results

Results from a .SSTAC analysis (see “.SSTAC” on page 44) can be plotted or printed with the.PLOT or .PRINT commands.

The syntax is similar to the syntax for AC analysis output, except that the AC keyword isreplaced by SSTAC. For further details,see the .PLOT command or the .PRINT command ofthe Eldo User’s Manual.

.PLOT SSTAC OVN OVN

.PRINT SSTAC OVN OVN

For each output specified in the .PLOT or .PRINT command, Eldo RF generates one curve perharmonic of the Steady-State analysis. The x-axis of all these curves is specified in the .SSTAC

command line (XAXIS flag). Each curve will have its name augmented with the extension.H(i) where i is the corresponding harmonic. A positive i corresponds to an upper sideband and a negative i corresponds to a lower side band.

When two or more harmonic indices are specified, the extension is .H(i,j,k,...) where i,j, k, ... are the corresponding harmonics according to the different fundamental frequencies.Indices specifying a positive frequency correspond to upper side bands, and negative indicescorrespond to lower side bands.

It is also possible to address each specific curve by specifying the corresponding extension.When a curve name contains an extension .H(I), it can be manipulated by the .DEFWAVE or.EXTRACT functions (see .DEFWAVE and .EXTRACT in Simulator Commands chapter of theEldo User’s Manual). This is not the case for a curve name without extension.

Example

.SST fund1=1.2giga nharm1=5

.SSTAC LIN 10 100meg 200meg

.PLOT SSTAC vdb(2).h(0) vdb(2).h(2)

Eldo RF will generate the following curves: vdb(2).H(0) and vdb(2).H(2).

Extraction of Large-Signal/Small-Signal S-Parameters using .SSTAC

Incident and reflected waves are considered as small (AC) signals, computed around the largesteady-state signal. This means that this analysis takes into account frequency conversion oftransmitted and reflected waves in the computation of the S-parameters.

The port specification is the same as for AC analysis. Ports has to be rank from 1 to themaximum number of ports used. These ports do not need to have large signal specifications.

The display syntax is:

.PLOT SSTAC Sxx(i,j)[.h(k)]

Display Command Syntax.PLOT/.PRINT SSTAC


(Y-, Z-, G-, H-, T- and A-parameters are also available.)

This provides the same kind of results as a multitone large-signal S-parameter extraction duringan .SST analysis (see “S, Y, Z Parameter Extraction” on page 195), but this method is probablymore intuitive, faster and more compact.

Display of Loop Gain

First you will have to invoke Loop Stability Analysis using the .LSTB command:

.LSTB <source_name> [HARM(i,j,k)]

Where

• <source_name> is the voltage source that is placed in series in the loop to compute theloop gain (using the Middlebrook Technique)

• HARM(i,j,k) specifies the harmonics of the large signal steady-state around which thesource stimulates the loop in order to compute the loop gain. The default is DC, that is,HARM(0)

See also .LSTB in the Simulator Commands chapter of the Eldo User’s Manual.

The syntax to display the loop gain in different formats is:

.PLOT SSTAC LSTB_xx[.h(i,j,k)]

Where

• xx can be:

• DB for magnitude in dB

• M for magnitude

• P for phase

• R for real part

• I for imaginary part

• .h(i,j,k) specifies the harmonic of the large signal steady-state around which the loopgain is computed. The default is all harmonics.

Group Delay (GD) of Outputs

Group Delay (the derivative of the phase with respect to the frequency) of outputs is supportedwith SSTAC results, for example:

.PLOT SSTAC vgd(n1)[.h(i,j,k)]


Display Command Syntax.PLOT/.PRINT SSTXF

.PLOT/.PRINT SSTXFSteady-State TF Analysis Results

Results from a .SSTXF analysis (see “.SSTXF” on page 46) can be plotted or printed with the.PLOT or .PRINT commands.

.PLOT SSTXF OVN OVN

.PRINT SSTXF OVN OVN

For each output specified in the .PLOT or .PRINT command, Eldo RF generates one curve perharmonic of the Steady-State analysis. Each curve will have its name augmented with theextension .H(i) where i is the corresponding harmonic. A positive i corresponds to anupper side band and a negative i corresponds to a lower side band.

It is also possible to address each specific curve by specifying the corresponding extension.When a curve name contains an extension .H(I), it can be manipulated by the .DEFWAVE or.EXTRACT functions. This is not the case for a curve name without extension.

• OVN

Specifies the source (voltage or current) from which the transfer function will be displayed,and the format of the results. Possible formats:XFM(dev_name), XFP(dev_name), XFDB(dev_name), XFR(dev_name), XFI(dev_name).

Example

.SST fund1=1.2giga nharm1=5

.SSTXF v(2) LIN 10 100meg 200meg

.PLOT SSTXF xfdb(Vdd).h(-1)

Eldo RF will compute the transfer function between the voltage source Vdd and node 2, anddisplay the results in dB.

The simulator prints all the transfer functions (between all the circuit sources and the output) inthe netlist.chi file in ASCII form.

Group Delay (GD) of Outputs

Group Delay (the derivative of the phase with respect to the frequency) of outputs is supportedwith SSTXF results, for example:

.PLOT SSTXF XFGD(Vin)[.h(i,j,k)]

Display Command Syntax.PLOT/.PRINT SSTNOISE


.PLOT/.PRINT SSTNOISESteady-State Noise Analysis Results

Results from a .SSTNOISE analysis (see “.SSTNOISE” on page 53) can be plotted or printedwith the .PLOT or .PRINT commands.

The syntax is similar to the syntax for NOISE analysis output, except that the NOISE keywordis replaced by SSTNOISE. For further details, see the .PLOT command or the .PRINTcommand of the Eldo User’s Manual.

Two kinds of ASCII information can be printed. One through the .PRINT command, and theother that is done by default (.chi file) which concerns the individual contributions of noisydevices. These contributions can also be printed to a separate file, specified with the optionSSTNOISE_FILE (see “SSTNOISE_FILE[=filename]” on page 167). You can definewhether individual contributions of SPHI and AMNOISE are printed as DSB results or SSBresults by specifying the option SSTNOISE_CONTRIB_TYPE (see“SSTNOISE_CONTRIB_TYPE=SSB|DSB” on page 167). Default is DSB. It is also possibleto define the contributions from device families using the options SSTNOISE_INCLUDE_DEVICESand SSTNOISE_EXCLUDE_DEVICES (see“SSTNOISE_INCLUDE_DEVICES=device_type” on page 167).

During an .SSTNOISE analysis, Eldo RF computes the noise spectrum generated at thespecified outputs of the circuit. It is also able to predict the Phase noise spectrum ( ). Thiscorresponds to the spectral density of the phase deviation due to noise. This result isasymptotically equal to 2L (f) for frequencies greater than Fc (see Figure 3-1); where L(f) isusually measured by spectrum analyzers and corresponds to the noise spectral density (in onesideband) normalized to the carrier signal power.

Syntax

.PLOT SSTNOISE ONOISE|INOISE|NOISE(elem_name)|DB(ONOISE)|DB(INOISE)

.PLOT SSTNOISE SNF|+ SPHI[(elem_name)]|SPHI_SSB[(elem_name)]|LF[(elem_name)]|+ SPHI_VCO|SPHI_PFD|SPHI_DIV|+ AMNOISE[(elem_name)]|AMNOISE_SSB[(elem_name)]|+ DB(SPHI[(elem_name)])|DB(SPHI_SSB[(elem_name)])|+ DB(SPHI_VCO)|DB(SPHI_PFD)|DB(SPHI_DIV)|+ DB(LF[(elem_name)])|DB(AMNOISE[(elem_name)])|+ DB(AMNOISE_SSB[(elem_name)]).PRINT SSTNOISE ONOISE|INOISE|NOISE(elem_name)|DB(ONOISE)|DB(INOISE).PRINT SSTNOISE SNF|+ SPHI[(elem_name)]|SPHI_SSB[(elem_name)]|LF[(elem_name)]|+ SPHI_VCO|SPHI_PFD|SPHI_DIV|+ AMNOISE[(elem_name)]|AMNOISE_SSB[(elem_name)]|+ DB(SPHI[(elem_name)])|DB(SPHI_SSB[(elem_name)])|+ DB(SPHI_VCO)|DB(SPHI_PFD)|DB(SPHI_DIV)|+ DB(LF[(elem_name)])|DB(AMNOISE[(elem_name)])|+ DB(AMNOISE_SSB[(elem_name)])

Sφ



For plotting thermal and flicker noise, see “Thermal and Flicker Noise Contribution” onpage 86.

Compared to the NOISE syntax, there are additional curves that can be plotted:

• ONOISE

Output noise spectrum (in ) at the OUT node specified in the .SSTNOISEcommand line. See Noise Equations for further details.

• INOISE

Input noise spectrum (in ) at the input voltage source of the circuit (Vin nodespecified in the .SSTNOISE command line). INOISE is computed as ONOISE divided by thetransfer function between the specified input source at the specified harmonic and the circuitoutput at the specified harmonic.

• DB(ONOISE)

Provides output noise results in dB/Hz.

• DB(INOISE)

Provides input noise results in dB/Hz.

• SNF

Spot Noise Figure results (if a .SNF command is specified).

• SPHI

Double Side Band (DSB) Phase Noise Spectrum ( ), units . See NoiseEquations for further details.

• SPHI_SSB

Single Side Band (SSB) Phase Noise Spectrum ( ) asymptotically corresponding to. It is also possible to access the individual contribution of any noisy device to the

total phase noise spectrum with the syntax: SPHI_SSB(elem_name).

• LF

Corresponds to the (single side band) noise power spectral density normalized to the signalpower. This quantity is flat at small offset noise frequency and is asymptotic to SPHI_SSBat larger offset frequencies (after the phnoise cutoff frequency).

• SPHI_VCO

The contribution to the total output phase noise of the devices of the VCO partition of apartitioned PLL (.SST PLL and .RFBLOCK).

• SPHI_PFD

The contribution to the total output phase noise of the devices of the PFD partition of apartitioned PLL (.SST PLL and .RFBLOCK).

V Hz⁄

V Hz⁄

Sφ 1 Hz⁄

Sφ 2⁄L f( )



• SPHI_DIV

The contribution to the total output phase noise of the devices of the divider partition of apartitioned PLL (.SST PLL and .RFBLOCK).

• AMNOISE

DSB amplitude noise power spectral density, with units . See Noise Equations forfurther details.

• AMNOISE_SSB

SSB amplitude noise power spectral density, with units .

During an .SSTNOISE analysis, the Eldo RF algorithm computes the Phase Noise Spectrum (or) corresponding to the “spectral density” of the phase deviation normalized to the carrier

power. This spectral density blows up at zero frequency offset, which reflects the fact that thephase deviation has no bound. This phenomenon is often confusing because it is not directlymeasurable by spectrum analyzers and also because it is mistaken with normalized noisespectral density L(f). This normalized noise spectral density (L(f)) is flat at small offsetfrequencies (below Fc, see diagram below) and is asymptotic to at larger offsetfrequencies (above Fc, see Figure 3-1).

Figure 3-1. Spectral density

• DB(SPHI)

Provides DSB phase noise results in dBc/Hz.

• DB(SPHI_SSB)

Provides SSB phase noise results in dBc/Hz.

• DB(SPHI_VCO)

Provides the contribution to the total output phase noise of the devices of the VCO partitionof a partitioned PLL in dBc/Hz.

• DB(SPHI_PFD)

Provides the contribution to the total output phase noise of the devices of the PFD partitionof a partitioned PLL in dBc/Hz.

1 Hz⁄

1 Hz⁄

Sφ

Sφ 2⁄

Frequency(Hz)

2L(f)

Sφ

Fc



• DB(SPHI_DIV)

Provides the contribution to the total output phase noise of the devices of the dividerpartition of a partitioned PLL in dBc/Hz.

• DB(LF)

Provides L(f) in dBc/Hz.

• DB(AMNOISE)

Provides DSB amplitude noise power results in dBc/Hz.

• DB(AMNOISE_SSB)

Provides SSB amplitude noise power results in dBc/Hz.

The simulator also prints ASCII results in the netlist.chi file. It prints the contribution of eachnoisy device (with internal noise sources) sorted by device types. Contributions are also sortedby harmonics.

NoteIn the .chi file, the phase noise contributions use DSB values.

• elem_name

Can be a device element or a subcircuit instance.

• SPHI(elem_name)

Contribution of elem_name to the total DSB phase noise spectrum (in ).

• SPHI_SSB(elem_name)

Contribution of elem_name to the total SSB phase noise spectrum (in ).

• LF(elem_name)

Contribution of elem_name to LF (in ).

• AMNOISE(elem_name)

Contribution of elem_name to the total DSB amplitude noise power (in ).

• AMNOISE_SSB(elem_name)

Contribution of elem_name to the total SSB amplitude noise power (in ).

• DB(NOISE(elem_name))

Provides the contribution of noise(elem_name) in dBc/Hz.

• DB(SPHI(elem_name))

Provides the contribution of SPHI(elem_name) in dBc/Hz.

• DB(SPHI_SSB(elem_name))

Provides the contribution of SPHI_SSB(elem_name) in dBc/Hz.

1 Hz⁄

1 Hz⁄

1 Hz⁄

1 Hz⁄

1 Hz⁄



• DB(LF(elem_name))

Provides the contribution of LF(elem_name) in dBc/Hz.

• DB(AMNOISE(elem_name))

Provides the contribution of AMNOISE(elem_name) in dBc/Hz.

• DB(AMNOISE_SSB(elem_name))

Provides the contribution of AMNOISE_SSB(elem_name) in dBc/Hz.

• NOISE(elem_name)

Contribution of elem_name to the total output noise spectrum (in ).

Noise Equations

where PSD is the Power Spectral Density

where Sv is the Amplitude Noise Spectrum

where Sφ is the Phase Noise Spectrum

where:

V Hz⁄

ONOISE PSD=

AMNOISE Sv=

SPHI Sφ=

fo fo+fpfo-fp

SφS11 S 1– 1– 2–+ Re S1 1– e

2 j Φss⋅ ⋅⋅( )⋅

Vss2

----------------------------------------------------------------------------------------------=

SvS11 S 1– 1– 2+ + Re S1 1– e

2 j Φss⋅ ⋅⋅( )⋅

Vss2

----------------------------------------------------------------------------------------------=

S11 PSD at fo fp+= O( NOISE2 at fo f p )+

S 1– 1– PSD at fo fp–( )= O( NOISE2 at fo f p )–



S 1 -1 is the cross correlation between fo + fp and fo - fp

fo is the large signal reference frequency for phase noise

fp is the noise frequency (offset from fo)

Vss and are the magnitude and phase respectively of the large signal Steady-State analysis(SST analysis) output (the output is defined in the .SSTNOISE command).

If and Sv are added, and assume S11 and S -1 -1 to be equal, then:

Examples

.PLOT SSTNOISE SPHI(M3) SPHI_SSB SPHI

This example shows how it is possible to access the individual contribution of the noisy device(transistor M3) to the total Double Side Band phase noise spectrum, and also to have access toboth Single and Double Side Band Phase Noise Spectrums through two different keywordsSPHI_SSB and SPHI.

.SSTNOISE V(out) Vin INPUT_HARM(i) [HARM(j)] ...

.PLOT SSTNOISE INOISE

.PLOT SSTNOISE db(INOISE)

This example shows how to plot the equivalent noise at the input of the circuit.

Thermal and Flicker Noise ContributionThermal and flicker contributions can be added to all the plotted SSTNOISE results. This isavailable though the .PLOT SSTNOISE and .PRINT SSTNOISE commands using the followingkeywords:

.PLOT SSTNOISE+ ONOISE_THERMAL[(dev_name)]) | DB(ONOISE_THERMAL[(dev_name)]) |+ SPHI_THERMAL[(dev_name)] | DB(SPHI_THERMAL[(dev_name)])|+ SPHI_SSB_THERMAL[(dev_name)] | DB(SPHI_SSB_THERMAL[(dev_name)]) |+ LF_THERMAL[(dev_name)] | DB(LF_THERMAL[(dev_name)]) |+ AMNOISE_THERMAL[(dev_name)] | DB(AMNOISE_THERMAL[(dev_name)]) |+ AMNOISE_SSB_THERMAL[(dev_name)] | DB(AMNOISE_SSB_THERMAL[(dev_name)])

.PLOT SSTNOISE+ ONOISE_FLICKER[(dev_name)]) | DB(ONOISE_FLICKER[(dev_name)]) |+ SPHI_FLICKER[(dev_name)] | DB(SPHI_FLICKER[(dev_name)])|+ SPHI_SSB_FLICKER[(dev_name)] | DB(SPHI_SSB_FLICKER[(dev_name)]) |+ LF_FLICKER[(dev_name)] | DB(LF_FLICKER[(dev_name)]) |+ AMNOISE_FLICKER[(dev_name)] | DB(AMNOISE_FLICKER[(dev_name)]) |+ AMNOISE_SSB_FLICKER[(dev_name)] | DB(AMNOISE_SSB_FLICKER[(dev_name)])

Φss

Sφ

ONOISE2 Vss2

4----------- SPHI2 AMNOISE2

+( )⋅=



ASCII Output File (.chi) ResultsEldo RF analyzes the noise generated at the output at the frequency Fcontrib, whereFcontrib =|Fout + k.Fund1|, k is the harmonic and Fout is the frequency respectivelydefined in the .SSTNOISE command.

In the case of a non-linear noise analysis, the noise at the output at the frequency Fnoise, is a asum of the contributions coming from all the harmonics. This means that a noise source ispresent somewhere in the circuit at the frequency Fcontrib = |Fout + j.Fund1| and willcontribute to the output noise. This is the case for all noise sources and all the harmonics j, thisphenomenon is called noise conversion and is a result of non-linearities in the circuit. Theintegral of the NOISE results (ONOISE, PHNOISE and AMNOISE) in the frequency band arecomputed for each device, the results are printed at the end of the computation i.e. after the lastpoint of the frequency band. The contributions from all the noise sources and all the harmonicsj are computed separately. The section MOST IMPORTANT HARMONICCONTRIBUTIONS in the chi file are a list of the most important contributions for the differentharmonics j.

The following results have been extracted from a chi file generated by a noise analysis.

CONTRIBUTION FROM THE HARMONICS:( -5) : 1.20664E-24( -4) : 4.77852E-24( -3) : 5.25354E-23( -2) : 9.91715E-22( -1) : 2.48496E-19( 0) : 5.52853E-16 (66.2% of ONOISE)( 1) : 2.80732E-16 (33.6% of ONOISE)( 2) : 6.46849E-19( 3) : 3.86016E-20( 4) : 4.16943E-22( 5) : 1.43915E-23

For harmonic 0, the noise spectrum contribution is 66%. For harmonic 1, the noise contributionis 33%. This is shown in the following equations:

Fcontrib 1Meg 0( )Fund1+ 1Meg= =

Fcontrib 1Meg 1( )Fund1+ 1851Meg= =


Display Command Syntax.PLOT/.PRINT SSTJITTER

.PLOT/.PRINT SSTJITTERSteady-State Jitter Results

.PLOT SSTJITTER LT_JITTER

Eldo RF will print the long time jitter results from a .SSTNOISE analysis, see “.SSTNOISE” onpage 53. A jitter is a small movement of a signal in either phase or time that can cause asynchronization error.

• LT_JITTER

Long time oscillation jitter information, computed from the phase noise spectrum. Theequation used to compute the jitter is shown below:

For more information, the application note titled Analysis of Phase Noise in Phase-Locked Loops with Eldo RF is available upon request.

lt_jitter(t)2

π2f 0

2⋅-------------------- Sphi_ssb f m( ) 2 π f m t⋅ ⋅( )sin⋅ f md

0

+∞∫⋅=

Display Command Syntax.PLOT/.PRINT SSTSTABIL


.PLOT/.PRINT SSTSTABILLocal Stability Analysis Results

Results from a .SST STABIL analysis (see “.SST STABIL” on page 43) can be plotted orprinted with the .PLOT or .PRINT commands.

.PLOT SSTSTABIL NET_POLE | Q_FACTOR

• NET_POLE

Eldo RF performs a local stability analysis of the DC operating point. The stability analysiscomputes the poles of the circuit and identifies whether some poles have a positive real part(leading to instability). The right most pole of the complex plane can be obtained using the.PLOT and .PRINT commands. For further details, see the .PLOT command or the.PRINT command of the Eldo User’s Manual.

NoteCan be specified together with a steady-state analysis, please refer to “.SST” on page 29.

• Q_FACTOR

Quality Factor Estimation for Autonomous Circuits. It is possible to get an estimation of thequality factor for any autonomous circuit. This value is calculated with .SST STABILanalysis, using the theory of the linear system:

Q_factor = ω0/2σ

where ω0 and σ are derived from the calculated pole that exhibits the largest positive realpart as follows:

p = σ + j ω0

Example

.MODEL TS2 NPN+ BF=10 br=1 xtb=3 is=10f eg=1.11 RB=100+ rc=10 vaf=50 tr=6n mjc=0.33 vjc=0.75 mje=0.33 vje=0.75Vdd dd 0 3VR1 1 0 8.2kR2 dd 1 12kR3 2 3 3R4 4 0 1.5kL1 dd 2 0.01C1 dd 4 0.1uC2 3 4 pcQ 3 1 4 TS2.PARAM PC=47n.STEP PARAM PC 40n 60n 10n.SST STABIL.PLOT SSTSTABIL NET_POLE.PLOT SSTSTABIL Q_FACTOR.END



The results of running this example are shown in Figure 3-2 and Figure 3-3.

Figure 3-2. Example Plot for SSTSTABIL NET_POLE



Figure 3-3. Example Plot for SSTSTABIL Q_FACTOR


Display Command Syntax.PLOT/.PRINT FMODSST/TMODSST

.PLOT/.PRINT FMODSST/TMODSSTModulated Steady-State Analysis Results

Results from a .MODSST analysis (see “.MODSST” on page 62) can be plotted or printed withthe .PLOT or .PRINT commands.

Time-Frequency DomainThese kind of results are useful to display I and Q channels for trajectory and constellationdiagrams.

The syntax is a mixture between the syntax for TRAN and AC analysis output, except that thekeyword is replaced by FMODSST. For further details, see the .PLOT command or the.PRINT command of the Eldo User’s Manual.

.PLOT FMODSST OVN OVN

.PRINT FMODSST OVN OVN

For each output specified in the .PLOT or .PRINT command Eldo RF generates one curve perharmonic. The x-axis of all these curves is specified in the .MODSST command line. Thedisplayed name of each curve contains the extension .H(i) where i is the correspondingharmonic. By default, without the extension syntax, Eldo RF will generate all the harmonics.

The keyword FUND_OSC can be used to obtain the actual oscillation frequency in the case ofoscillator analysis. It can be handled in .PLOT/.PRINT commands.

It is also possible to compute and display the Cumulative Complementary Density Function(CCDF) from FMODSST results, see syntax below. CCDF displays the probability of a curve toexceed its average power value by a given amount of power function of this amount of power.This display is extremely important to characterize digitally modulated signals. It providesinformation concerning power characteristics and peak-to-average power data.

.DSP FMODSST label=label_name dsp=CCDF waveform_name

.PLOT dsp dsp(label_name)

• waveform_name

The harmonic of a time-dependent spectrum representation of a waveform (FMODSST basedformat).

It is also possible to address each specific curve by specifying the corresponding extension.When a curve name contains an extension .H(I), it can be manipulated by the .DEFWAVE or.EXTRACT functions. This is not the case for a curve name without extension. If you omit the.H(i) extension in a .DEFWAVE or .EXTRACT command, an error will be output, for example:

ERROR 481: COMMAND .DEFWAVE NEW_WAVE: expects .H extensions;



Examples

.SST FUND1=900MEG NHARM1=10

.MODSST 1U 100U

.PLOT FMODSST VDB(2).H(0)

.PLOT FMODSST VM(2).H(1)

.PLOT FMODSST VR(3)

Here, Eldo RF will generate in time domain the following curves: VDB(2).H(0), VM(2).H(1)

and VR(3).

The VDB(2).H(0) waveform represents the time-varying Fourier component of the DC in dBof node 2. The VM(2).H(1) waveform represents the magnitude of the time-varying Fouriercomponent of the first harmonic of node 2. The VR(3) waveform will be generated as the realpart of the time-varying Fourier component of all harmonics .H(i) of node 3.

.DSP FMODSST label=CCDF_Vout dsp=CCDF v(out).H(1)

.PLOT dsp dsp(CCDF_Vout)

The .DSP command with the dsp=CCDF option, will display the Complementary CumulativeDensity Function of the first harmonic (H(1)) of the V(out) spectrum component.

The curves generated by the FMODSST mode of the .PLOT command in the .wdb file aredisplayed with EZwave in a time-varying representation.

Eye Diagram, Trajectory Diagram, and Constellation Diagram Example

The purpose of this example is to show how to generate an eye diagram, trajectory diagram, anda constellation diagram.

This example uses a 16 QAM modulated source with an initial delay of 100ns and an RF macromodel (frequency divider) to provide gain and phase rotation. The netlist is shown below:

*VMOD 1 0 MQAM M=16 LPF=ROOT_RAISED_COSINE BETA=0.05 IASC=1 QASC=1+ FOUR FUND1 MA (1) 1 -90 PATTERN DELAY=DELAY TSYMB=1u RANDOM

YDIV RF_FREQUENCY_DIVIDER PIN: 1 0 2 0+ PARAM: GAIN=2 DIV_FACTOR=1 INPUT_H1=1 OUTPUT_H1=1 OUT_PHASE=PHASE

.SST FUND1=5G NHARM1=1

.PARAM DELAY=100n PHASE=30

.MODSST 0 250u

.OPTION MODSST_HMAX=10n

.PLOT FMODSST VR(1).H(1) VI(1).H(1)

.PLOT FMODSST VR(2).H(1) VI(2).H(1)

Copy the netlist above into a text file and name the file etc_diagrams.cir. To run theexample, execute the following:

eldo etc_diagrams.cir

Now load the simulation results in EZwave by executing the following command:



ezwave etc_diagrams.swd &

Now that the simulation results have been generated the Eye diagram, Trajectory diagram, andConstellation diagram can be generated, as described below:

• Eye diagram

To generate an eye diagram load the Eye Diagram Tool dialog box. To do this select thewave (VR(1).H(1)) in the window Wave:1, then select Tools > Eye Diagram, the EyeDiagram Tool box will be displayed as shown in Figure 3-4.

Figure 3-4. Eye Diagram Tool dialog box

Enter “2u” in the Eye Width field and click Apply, close the dialog box by clickingCancel. The eye diagram will be displayed in the EZwave window as shown inFigure 3-5. The eye diagram shows the initial delay required to generate the correctconstellation diagram (i.e. 600ns). A period of the signal (time symbol, 1µs) can also beseen.



Figure 3-5. EZwave window—eye diagram

• Trajectory diagram

To generate the trajectory diagram, with the waveforms VR(1).H(1) and VI(1).H(1)plotted in the wave window, select the waveform VR(1).H(1) using the right mousebutton, a pop-up menu will be displayed. In the pop-up menu select Set as Xaxis, thetrajectory diagram will be displayed in the EZwave window, as shown in Figure 3-6.



Figure 3-6. EZwave window—trajectory diagram

• Constellation diagram

The Waveform Calculator is used to generate the constellation diagram. Drag and dropthe waveforms V(1).H(1) and V(2).H(1) from the Waveform List Panel into the windowWave:2. Load the Waveform Calculator by selecting Tools > Waveform Calculator. Inthe Waveform Calculator select RF from the toolbar dropdown list, the buttons in theWaveform Calculator will now change. Select the Constellation Diagram button cd, theConstellation Diagram dialog box will be displayed. Select the waveform V(1).H(1)from the Source Waveform dropdown list, enter “600n” in the Delay field, and “1u” inthe Symbol Period field, and click Apply. Now select the waveform V(2).H(1) from theSource Waveform dropdown list, ensure that the value in the Delay and Symbol Periodfields are set to “600n” and “1u” respectively, and click OK. To plot the results, openthe Results Waveforms hierarchy in the Rslts tab of the Waveform Calculator. Select theresults under “Results Waveforms” using the right mouse button, and select PlotWaveform in the pop-up list. Repeat this for both waveforms. Finally drag and drop theplotted waveforms into the same window. Figure 3-7 shows the constellation diagramsof the signal provided by the modulated source (Constellation_V_1_H_1) and theconstellation diagram of the signal provided by the RF macro model which introducesgain and phase rotation (Constellation_V_2_H_1).



Figure 3-7. EZwave window—constellation diagram

Time DomainThese kind of results are useful to see the settling time of circuits.

The syntax is similar to the syntax for TRAN analysis output, except that the TRAN keyword isreplaced by TMODSST. For further details, see the .PLOT command or the .PRINTcommand of the Eldo User’s Manual.

.PLOT TMODSST OVN OVN

.PRINT TMODSST OVN OVN

MODSST analysis has the advantage that the step size should be small enough to capture themodulation. It would be possible to only plot the calculated point. Nevertheless, it is difficult toanalyze such waveforms. One may prefer to see, at each timepoint, the time-varying ofamplitudes of voltage and current showing the high frequency signal. Furthermore, between twotimepoints, the duration of the plot will depend on the period of the high frequencies. For single-tone signals, the default value of this duration is two periods. For multi-tone signals, the defaultvalue is two periods of the lowest fundamental frequency.

Example

.PLOT TMODSST V(IN1) V(OUT1,OUT2)



Eldo RF will generate in time domain two curves V(IN1) and V(OUT1,OUT2) on the samegraph.

CautionRequesting TMODSST outputs may quickly generate huge output files, and considerablyslow down the simulation.

Frequency DomainThese kind of results are useful to compute ACPR (Adjacent Channel Power Ratio) or NPR(Noise Power Ratio).

By doing an FFT on the time-varying Fourier coefficient, it is possible to get the frequencydomain results.

.PLOT FOURMODSST OVN OVN

The modulation spectrum around each harmonic is therefore obtained by performing an FFT onits complex time-varying value. This is done by using the .OPTFOUR command (see SimulatorCommands in the Eldo User’s Manual) and the plotting will be referred to FOURMODSST mode.

Example

.PLOT FOURMODSST FOURDB(v(100).H(1))

Eldo RF will generate in frequency domain the spectrum around the first harmonic in dB of thenode 100.

Please refer to Tutorial #13—ACPR Computations for an Amplifier for more informationon how to compute the ACPR.

Display Command Syntax.PLOT CONTOUR


.PLOT CONTOURContours

When designing RF blocks such as PA, the key point is the determination of the conditions forthe bias, input & output impedances in order to optimize power and power efficiency. A LoadPull contour is a useful plot that may help in the design of optimal functioning conditions. ALoad Pull contour is a set of points (on a Smith Chart) representing all the loads dissipating agiven amount of power. Thus it can help to determine which load will dissipate the most power;it can also help to match a load to a PA for maximum power transfer.

Load Pull contours are obtained by putting a source at the output of the analyzed block and bysweeping the magnitude and the phase of this source in order to reach all the possiblefunctioning conditions. Once this has been performed the contours are the collection of pointscorresponding to specified situation (same dissipated power, same input reflection coefficient,etc.).

Methodology and syntax to plot contours:

1. Sweep power and phase of output source, for example:

Vout LOAD 0 RPORT=Rout iport=2 FOUR fund1 PdBm (1)+ p2 ph_out.step param p2 -30 -10 5.step param ph_out 0 360 50

2. Specify what will define the contour (output power, input reflection coefficient, etc.), forexample:

.defwave Zin=V(in) / I(Vin)

.defwave Zout=-V(load) / I(Vout)

.defmac gamma(Zx) = (Zx - Z0) / (Zx + Z0)

.defwave gamma_in = $gamma(w(Zin))

.extract fsst label=Zout_r yval(wr(Zout), valfund_1)

.extract fsst label=Zout_i yval(wi(Zout), valfund_1)

.extract fsst label=gamma_in yval(wm(gamma_in), fund1)

.extract fsst label=pm_out yval(pm(Vout), fund1)

3. Extract the contours, for example:

.extract catvect sweep label=G81r xycond(meas(Zout_r),+ meas(gamma_in) == 0.81).extract catvect sweep label=G81i xycond(meas(Zout_i),+ meas(gamma_in) == 0.81).plot contour meas(G81r) meas(G81i) (smith).extract catvect sweep label=P103r xycond(meas(Zout_r),+ meas(pm_out) == 0.0103).extract catvect sweep label=P103i xycond(meas(Zout_i),+ meas(pm_out) == 0.0103).plot contour meas(P103r) meas(P103i) (smith)


Display Command Syntax.PLOT CONTOUR

• CATVECT

By default, .EXTRACT returns the first value which matches the expression. If keywordCATVECT is set on the .EXTRACT statement, then all values will be returned. In addition allmeasurements corresponding to all analyses (.STEP/.TEMP) will be combined. Thisfunctionality is usually used in conjunction with the CONTOUR function in the .PLOTcommand.

• CONTOUR

Eldo will plot the second measurement value with respect to the first. The wave will bedisplayed in a Smith chart if keyword SMITH is specified. CONTOUR information is writtento the .ext file.

NoteThe keyword VECT or CATVEC must have been specified on the .EXTRACT which isreferred to in the .PLOT CONTOUR, otherwise the corresponding data will be reduced to asingle print.The two measurements in the CONTOUR must have the same dimension. If not, the .PLOT

CONTOUR will be ignored.

Please refer to the tutorial Tutorial #18—Load Pull Contours for further information.

Display Command Syntax.EXTRACT


.EXTRACTExtract Command

The syntax of the .EXTRACT command is enhanced with the following options:

• FSST

Results of Steady-State analysis in the frequency domain.

• TSST

Results of Steady-State analysis in the time domain.

• SSTAC

Results of Steady-State AC analysis.

• SSTXF

Results of Steady-State TF analysis.

• SSTNOISE

Results of Steady-State Noise analysis.

• SSTJITTER

Jitter results of Steady-State Noise analysis

• SSTSTABIL

Results of Steady-State Stability analysis.

• TMODSST

Results of modulated Steady-State analysis in the time domain.

• FMODSST

Results of modulated Steady-State analysis in the time-frequency domain.

• FOURMODSST

Results of modulated Steady-State analysis in the frequency domain.

NoteThe function pow() is not supported during a MODSST analysis.

The general syntax of .EXTRACT for Steady-State analysis and Steady-State based analysis isthus the following:

.EXTRACT [FSST|TSST|SSTAC|SSTXF|SSTNOISE] $MACRO|FUNCTION

.EXTRACT SSTSTABIL NET_POLE_I|NET_POLE_R

.EXTRACT SSTJITTER PERIOD_JITTER | JITTER

For more information, see the .EXTRACT command of the Eldo User’s Manual.



The keyword FUND_OSCxx can be used to obtain the actual oscillation frequency in the case ofoscillator analysis. It can be handled in .EXTRACT functions. Additionally, in netlistscontaining a .SST command, FUNDxx can be used as a keyword in parameter definition ofplots.

Examples

This example will provide the oscillation frequency.

.EXTRACT FSST FUND_OSC1

This next example will provide the value of the spectrum vdb(osc_out) at the 3rd harmonic ofthe oscillation frequency.

.EXTRACT FSST yval(vdb(osc_out), 3*FUND_OSC1)

Specific RF Pre-Defined Functions• OIPX(Wave, Freq1, Freq2)

Returns the output referred intercept point of order n from the value of the circuit output(Wave). Wave must be in dB or dBm. The intercept order is calculated using the specifiedfundamental frequency Freq1 and intermodulation frequency Freq2.

• IIPX(Wave_In, Wave_Out, Freq1, Freq2)

Returns the input referred intercept point of order n from the value of the circuit input andoutput (Wave_In and Wave_Out respectively). Wave_In and Wave_Out must be in dB ordBm. The intercept order is calculated using the specified fundamental frequency Freq1and intermodulation frequency Freq2.

The last lines in the following example show OIP3 and IIP3 extraction.

Example

Vin in 0 iport=1 rport=50 FOUR fund1 fund2 pdbm+ (1,0) Pin -90+ (0,1) Pin 90.sst fund1=f1_val nharm1=4+ fund2=f2_val nharm2=4

.param Pin=-20

.param f1_val=1g f2_val=1.001g

.extract fsst OIPx(vdb(out), f2_val, 2*f2_val-f1_val)

.extract fsst IIPx(Pdb(Vin), vdb(out), f2_val,+ 2*f2_val-f1_val)

Please refer to Tutorial #3—Multi-Tone Analysis, and IM3 and IP3 Extraction for anAmplifier for a fully worked example on Multi-tone analysis—IM3 and IP3 extraction.



• COMPRESS(wave,value)

Extracts the Y-axis value of the wave at the point where the difference between the actualvalue of wave and the linear extrapolation of wave based on the computed slope valuebecomes greater than value.

• XCOMPRESS(wave,value)

Extracts the X-axis value of the wave at the point where the difference between the actualvalue of wave and the linear extrapolation of wave based on the computed slope valuebecomes greater than value.

• DISTO(WAVE, FUND_FREQ, FMIN, FMAX)

Returns the total harmonic distortion of a signal. FUND_FREQ is the fundamental frequencyand FMIN and FMAX specify the window in which you require the harmonic distortion to becalculated.

Please refer to the descriptions in the .EXTRACT command of the Eldo User’s Manualfor the previous three functions listed.

• PIB(wave, fmin, fmax)

V(f) is a voltage output in the frequency domain.

PIB returns the power in a bandwidth [fmin, fmax] of wave for a load of 0.5Ω. wave mustbe a frequency output (FSST or FOURMODSST) corresponding to a node.

The following example computes the power in a bandwidth [-30k,30k] and returns the valuein dB.

Example

.PARAM fmin=-30k

.PARAM fmax=30k

.EXTRACT FOURMODSST LABEL=power_in_bandwidth10.*log10(PIB(fourm(v(out).h(1)), fmin, fmax))

Adjacent Channel Power Ratio (ACPR)In digital modulation, the channel leakage is an important aspect characterizing the corruptionby the adjacent channel to the signal channel. This phenomenon can be quantified by comparingthe power in the adjacent channel to the power in the main channel.

PIB V f( ) 2

f f min=

f max

∑=

ACPRPAdjacent Channel

PMain Channel------------------------------------=



By using a combination of the .EXTRACT command and PIB function, it is possible to obtainthe ACPR value.

Example

.PARAM fmain=2.5Meg

.PARAM offset=5Meg

.PARAM fb=2Meg

.EXTRACT FOURMODSST LABEL=acpr_up 10.*log10(PIB(fourm(v(out).h(1)),-fb+offset, fb+offset) / PIB(fourm(v(out).h(1)), -fmain, fmain))

Extracts the power ratio of V(out).h(1) in a bandwidth [-fb+offset, fb+offset] and [-fmain, fmain] in dBc as shown in the diagram below.

Figure 3-8. Adjacent Channel Power Ratio

For more information on ACPR see Tutorial #13—ACPR Computations for anAmplifier.

Two-Port Gain ParametersIn two-port circuits, different gains can be computed. Eldo RF provides the following sixquantities: GA, GAM, GASM, GP, TGP and GAUM.

Power

fmain-fmain

f (MHz)(2)(2)

fbfb2.50



Consider a two-port circuit where impedances and reflection coefficients are defined:

• ZS

Source impedance

• ZL

Load impedance

• Γ S

Source reflection coefficient

• Γ L

Load reflection coefficient

• Z0

Characteristic impedance (by default 50 Ω; to modify this value see the ZCHAR option ofthe Eldo User’s Manual).

• GA

Available power Gain. This is the ratio of the power available from the two-port circuit tothe power available from the source when the load is conjugately matched to the output portof the circuit (ΓL = conj(ΓOUT)).

Two-portcircuit [S]

ZSZL

VLVS

ΓS ΓIN ΓOUT ΓL

Z0



• GAM

Maximum Available power Gain. When the input port of the circuit is conjugately matchedto the source impedance and the output port of the circuit is conjugately matched to the loadimpedance (ΓS = conj(ΓIN) and ΓL = conj(ΓOUT)).

For a bilateral case:

and:

Refer to “KFACTOR” on page 107.

For a unilateral case, see the “GAUM” on page 107.

• GASM

Maximum Available Stable Gain. See the GAM definition above.

• GP

Power Gain. This is the ratio of the power delivered to the load to the power input to thetwo-port circuit when the input port of the circuit is conjugately matched to the sourceimpedance (ΓS = conj(ΓIN)).

GA1 Γ s

2–

1 S11Γ s–2

---------------------------- S212 1

1 ΓOUT2

–----------------------------=

where

Γ s

Zs Z0–

Zs Z0+------------------=

Γout S22

S12S21Γ s

1 S11Γ s–-----------------------+=

GAMS21

S12------- KFACTOR KFACTOR

2 1––( )=

if KFACTOR 1>

GAMS21

S12-------=

if KFACTOR 1≤

GASMS21

S12-------=



• TGP

Transducer Power Gain. This is the ratio of the power delivered to the load to the poweravailable from the source.

• GAUM

Maximum Unilateral transducer power Gain. This is the transducer gain when the circuitports are both optimally matched (ΓS = conj(ΓIN) and ΓL = conj(ΓOUT) and S12 = 0).

Example

(see Tutorial #1—Single-Tone Steady-State Analysis and Two-Port Noise Extraction for anAmplifier): Available Gain of an amplifier

.param fund1=900Meg

.extract fsst label=Available_Gain yval(GA, fund1)

Two-Port Stability FactorsIt is also possible to plot different stability factors for the analysis of two ports by specifyingKfactor, B_factor, Mu_factor, Mufactor_L, and Mufactor_S. They are calculated from the Sparameters:

• KFACTOR

Computes the stability factor for 2-ports. Available with AC and FSST results.

GP1 ΓL

2–

1 S22ΓL–2

----------------------------- S211

1 Γ IN2

–-----------------------=

where

ΓL

Z L Z0–

Z L Z0+-------------------=

Γ IN S11

S12S21ΓL

1 S22ΓL–-----------------------+=

TGP1 Γ s

2–

1 S11Γ s–2

---------------------------- S21

1 ΓL2

–

1 Γ– OUT ΓL2

-----------------------------------=

GAUM1

1 S112

–---------------------- S21

2 1

1 S222

–----------------------=

KFACTOR1 S11 2

– S22 2– S11 S22⋅ S12 S21⋅( )–

2+

2 S12 S21⋅×( )-------------------------------------------------------------------------------------------------------------------=



• BFACTOR

Rollett stability factor.

• MUFACTOR

Rollett stability factor.

where

• MUFACTOR_L

Load stability factor.

where

• MUFACTOR_S

Source stability factor.

where

Two-Port Noise ParametersAny noisy two-ports can be represented by the equivalent noiseless two-port with the twoequivalent noise sources (en and in) or the corresponding noise correlation matrix CA, see figurebelow:

BFACTOR1 S22

2–

S11 S22∗∆– S22 S11⋅+

-------------------------------------------------------------=

MUFACTOR1 S11

2S22

2– ∆ 2

+ +

S11 S22∗∆–

---------------------------------------------------------=

∆ = S11 S22 S12 S21⋅( )–⋅

MUFACTOR_L1 S11

2–

S22 S11( )* ∆⋅– S12 S21⋅+----------------------------------------------------------------------=

∆ = S11 S22 S12 S21⋅( )–⋅

MUFACTOR_S1 S22

2–

S11 S22( )* ∆⋅– S21 S12⋅+----------------------------------------------------------------------=

∆ = S11 S22 S12 S21⋅( )–⋅



Eldo RF calculates the six equivalent two-port noise parameters: NFMIN, RNEQ, GOPT, BOPT,GAMMA_OPT_MAG and PHI_OPT.

• NFMIN

Minimal noise figure of the two-port.

• GOPT

Real part of the optimal source admittance.

• BOPT

Imaginary part of the optimal source admittance. The sign convention of the BOPTparameter can be changed using the NOISE_SGNCONV option.

• RNEQ

Equivalent noise resistance.

• YOPT=GOPT + j · BOPT

Source admittance that produces the minimal noise figure.

These four parameters are computed from the two-port noise correlation matrix CA:

Noisy 2Ports

Noiseless2 Ports

en

in

CA

( )en

2en in

∗⋅

en∗ in⋅ in

2

C11A

C12A

C21A

C22A

= =

RNEQC11

A

4 k T⋅ ⋅------------------=

BOPTIm C12

A

C11A

----------------------=

GOPT1

C11A

-------- C11A

C22A

Im C12A ( )

2–⋅=



• GAMMA_OPT_MAG

Magnitude of the optimal reflection coefficient associated with the minimum noise figure.(GAMMA_OPT is the complex quantity of this.)

• PHI_OPT

Angle of the optimal reflection coefficient associated with the minimum noise figure. Thesign convention of the PHI_OPT parameter can be changed using the NOISE_SGNCONVoption.

These two parameters are calculated as follows.

All six of the above parameters are available after a .NOISE as well as a .SSTNOISE analysisthrough .PLOT and .EXTRACT commands.

NoteThe NOISETEMP parameter of the port sources will be taken into account for thecalculation of two-port noise parameters (NFMIN, GOPT, BOPT, RNEQ) for NOISE andSSTNOISE analyses.

Example

.PLOT NOISE NFMIN

.EXTRACT NOISE YVAL(RNEQ, 10k)

.PLOT SSTNOISE GOPT

.EXTRACT SSTNOISE YVAL(BOPT, 1Meg)

Run Tutorial #1—Single-Tone Steady-State Analysis and Two-Port Noise Extraction foran Amplifier for more information.

NFMIN 1 2Re C12

A GOPT C11A⋅+

4 k T⋅ ⋅-----------------------------------------------------------⋅+=

Γopt1 Z0 Gopt j Bopt⋅+( )⋅( )–

1 Z0 Gopt j Bopt⋅+( )⋅+----------------------------------------------------------------=

GAMMA_OPT_MAG Γopt=

PHI_OPT Γopt∠=



Two-Port Noise Circles• NC

A noise circle is plotted for a given value of a Noise Figure (SNF_val). When this value isnot specified, the circle is plotted for a Noise Figure value corresponding to the actualcircuit.

Example

.PLOT NOISE NC (<SNF_val>) (SMITH)

.PLOT NOISE NC (SMITH)

.PLOT SSTNOISE NC (<SNF_val>) (SMITH)

.PLOT SSTNOISE NC (SMITH)

Run Tutorial #1—Single-Tone Steady-State Analysis and Two-Port Noise Extraction foran Amplifier for more information.

Two-Port Constant Gain CirclesEldo RF proposes two constant gain circles named GAC and GPC. GAC determines constantgain contour at the input port and GPC determines constant gain contour at the output port. Withrespect to the definition of GA and GP, GAC represents an optimum match at the output portand GPC represents an optimum match at the input port.

• GAC

Available Gain Circle.

centerΓopt

1 N i+---------------=

radiusN i

2N i 1 Γopt

2–( )⋅+

1 N i+-----------------------------------------------------=

N i

SNF_val NFMIN–( ) 1 Γopt+2⋅

4 RNEQ⋅----------------------------------------------------------------------------------=

centerGA S11

*S22∆*

–( )

S212

GA S112 ∆ 2

–( )+-------------------------------------------------------------=

where ∆ S11S22 S21S12–( )=



• GPC

Power Gain Circle.

Two-Port Stability CirclesEldo RF proposes two stability circles named SSC and LSC. SSC determines the locus of ΓSwhich produce |ΓOUT| = 1, and LSC determines the locus of ΓL which produce |ΓIN| = 1.

• SSC

Source Stability Circle.

• LSC

Load Stability Circle.

radius

1 2KFACTOR S21S12GA

S212

-------------– S21S12GA

S212

------------- 2

+

1GA

S212

------------- S112 ∆ 2

–( )+------------------------------------------------------------------------------------------------------------------------------=

centerGP S22

*S11∆*

–( )

S212

GP S222 ∆ 2

–( )+-------------------------------------------------------------=

radius

1 2KFACTOR S21S12GP

S212

-------------– S21S12GP

S212

------------- 2

+

1GP

S212

------------- S222 ∆ 2

–( )+------------------------------------------------------------------------------------------------------------------------------=

centerS22∆*

S11*

–

∆ 2S11

2–

----------------------------=

radius =S12S21

∆ 2S11

2–

----------------------------

centerS11∆*

S22*

–

∆ 2S11

2–

----------------------------=

radius =S12S21

∆ 2S22

2–

----------------------------



NoteThese circles can be directly plotted into a Smith Chart Diagram by using the command:.plot fsst xxC (smith)

For more information on plotting stability circles, please refer to the Circles subsectionof Smith Chart in the Creating Special Diagrams and Charts chapter of the EZwaveUser’s Manual.

Example

(see Tutorial #4—S-Parameter Extraction for an Amplifier): Available Gain circle, Power Gaincircle and Stability circles (LSC and SSC) of an amplifier

Local Stability AnalysisEldo RF performs a stability analysis of the DC operating point and can consecutively performa steady-state analysis. The stability analysis computes the poles of the circuit and identifieswhether some poles have a positive real part (leading to instability). The right most pole of thecomplex plane can be obtained using the .EXTRACT command.

• NET_POLE_I

Extracts the imaginary part of the right most pole.

• NET_POLE_R

Extracts the real part of the right most pole.

Steady-State JitterOscillator and forced circuit jitter information can be computed and displayed for .SSTNOISEanalysis.

Oscillator Jitter Information• PERIOD_JITTER

Defines the period jitter (uncertainty on the period) information, computed from the phasenoise spectrum. Only the white contributions to the phase noise are handled in thiscomputation. The equation used to compute period jitter is shown below:

J T c T⋅ cf 0------= =



For more information, the application note titled Analysis of Phase Noise inPhase-Locked Loops with Eldo RF is available upon request.

Forced Circuit Jitter Information• JITTER

Defines the synchronous jitter information, computed from the phase noise spectrum. Theequation used to compute the synchronous jitter is shown below:

where T0 is the period (inverse of the fundamental frequency), fmin and fmax are the minimumand maximum frequencies of the .SSTNOISE analysis.

J SyncT 02π------- SPHI

fmin

fmax

∫=


Chapter 4Sources

IntroductionThree different sources may be instantiated for Steady-State analysis (.SSTPROBE, FOUR andPHNOISE) see “Steady-State Sources” on page 116. The instance is the same as for other Eldosources; .SSTPROBE, FOUR and PHNOISE are the new source attributes.

NoteStandard Eldo periodic sources such as SIN and PULSE can be used to generate signalsfor a Steady-State analysis. The only restriction is that the source periods must becompatible with the fundamental frequency(ies) defined in the .SST command line. It isadvisable to use FOUR sources instead of standard SIN sources as FOUR sources are agreat deal more powerful and flexible.

All sources used for a SST analysis are compatible with MODSST analysis. Standard Eldoperiodic and non-periodic sources are also accepted whenever MODSST analysis is activated.See “Digitally Modulated Sources” on page 127.


SourcesSteady-State Sources

Steady-State SourcesMulti-Tone SourceSyntax

FOUR [DELAY=val] FUND1 [FUND2 [FUND3]] MA|RI|DB|PMA|PDB|PDBM+ (int_val1[,int_val2[,int_val3]]) real_val1 real_val2+(int_val1[,int_val2[,int_val3]]) real_val1 real_val2

NoteDC Offset:In versions v5.5 and earlier of Eldo RF, if a source contained a DC component and aFOUR component, then the DC value was added to the FOUR value, such that if:Vxx in 0 DC 5 FOUR fund1 MA (1) 1 -90

then the voltage would be calculated as:v = 5.0+1.0*sin(2*pi*fund1*t)

From version v5.6 the DC component only acts in DC and the FOUR component acts inTRAN and SST. Therefore if you need a DC component in the FOUR source you mustspecify the DC component in the DC part and the FOUR part (harmonic(0)) as follows:Vxx in 0 DC 5 FOUR fund1 MA (0) 5 0 (1) 1 -90

Parameters

• DELAY=val

Specifies the time (seconds) that the output is delayed by. This parameter is only effectiveduring Transient analysis (.TRAN), the Pre-Transient Phase, or Modulated Steady-Stateanalysis (.MODSST). Optional. Default value is zero. If used, this parameter must bespecified before the FUND1, FUND2 and FUND3 parameters. See also “Source Usage in Pre-Transient Phase” on page 124.

• FUND1, FUND2 or FUND3

Keyword or value specifying the fundamental frequency of the source. Only threefundamental frequencies are allowed but with any “i” index specified. If FUNDi is akeyword it is a reference to the FUNDi value defined in the .SST command line. If it is anexplicit value it must correspond to a value of FUNDx defined in the .SST command line,e.g.

V1 m1 m2 FOUR 1meg MA (1) 1.0 0.0.SST FUND1=1meg NHARM1=5

• MA|RI|DB|PMA|PDB|PDBM

Keyword defining the format (see table below). Power formats (PMA, PDB and PDBM) are onlyallowed on port sources (voltage or current sources where IPORT and RPORT are specified).The format is used in conjunction with real_val1 and real_val2.

SourcesMulti-Tone Source


• real_val1 and real_val2

Define the complex value of the source for the corresponding frequency in the specifiedformat (MA, RI, DB, PMA, PDB, or PDBM). When one of the above formats are specified (exceptRI), real_val2 can be specified with the keyword RANDOM (which means the value willbe randomly chosen between 0 and 2π).

The table below provides a summary of the meanings of real_val1 and real_val2 for eachformat specified:

NoteThe MA, PDB and PDBM formats use an ANGLE (in degrees) to specify the initial phase ofthe signal. Eldo RF internally uses cosine waveforms, whereas sine waveforms are morefamiliar to many users. Sine also corresponds to the SIN signal in SPICE language. Toobtain a FOUR signal whose initial phase is zero (i.e. a sine signal), an ANGLE of -90must be specified (a cosine shifted by -90 deg. is a sine...) for the FOUR source, e.g.VCOS IN 0 FOUR MA (1) 10mV 0

this is: 10e-3.cos(2.pi.f.t)VSIN IN 0 FOUR MA (1) 10mV -90

this is: 10e-3.cos(2.pi.f.t-90) = 10e-3.sin(2.pi.f.t)

• int_val1

Defines the index of the harmonic corresponding to the first fundamental frequency.

• int_val2

Defines the index of the harmonic corresponding to the second fundamental frequency.

• int_vali

Defines the index of the harmonic corresponding to the ith fundamental frequency.

The group of 1 to i index values define a frequency. For example, a source with threefundamental frequencies, (int_val1, int_val2, int_val3) specifies the frequency:

int_val1 × FUND1 + int_val2 × FUND2 + int_val3 × FUND3

Table 4-1. Complex Number Format

Format real_val1 meaning real_val2 meaning

MA Magnitude Angle

RI Real part Imaginary part

DB Magnitude in dB Angle

PMA Power in Watts Angle

PDB Power in dB Angle

PDBM Power in dBm Angle


SourcesMulti-Tone Source

NoteIn port sources, the specified power corresponds to the power dissipated outside the portwhen it is loaded with a resistor equal to RPORT. It then corresponds to the maximumavailable power and not necessarily to the power that is delivered by the source. Thismeans that .

For further information on using single-tone and multi-tone sources, please refer to thechapter Sources of the Eldo User’s Manual.

Example

In this example, the MA format is used, and we specify a 5 mV Magnitude with a -90 Angle.

Vxx inv FOUR FUND1 MA (1) 5mv -90

In this example, the PDBM format is used, and we specify a -30 dBm Power with a -90 Angle.

Vxx inv RPORT=50 IPORT=1 FOUR FUND1 PDBM (1) -30 -90

V 8 PMA× RPORT×=

SourcesProbe Source


Probe SourceSyntax

.SSTPROBE np nn [FUND_OSCxx [RANK=int_val]]

A probe is designed specifically for oscillator analysis. Steady-State analysis of autonomouscircuits require the insertion of a “probe” into the circuit. Probes are used to compute levels offrequency and oscillation at the insertion point.

During steady-state analysis of autonomous circuits, the probe behaves as a pure sinusoidalgenerator at the fundamental oscillation frequency and as an open circuit for all otherfrequencies. During the analysis, Eldo RF uses an optimization procedure to compute theoscillation frequency and the magnitude of the probe. The optimization stops when theadmittance of the probe falls below a given tolerance. At this frequency, the circuit is anoscillator.

Efficiency of analysis depends on where the probe is placed in the circuit. A probe should beinserted in parallel with the resonator or in parallel with the load. The probe should have someeffect on the oscillation; in particular it should not be put after the buffer or in the biasingcircuitry.

You should manually change the place where the probe is connected. When a probe is insertedcorrectly but causes a voltage loop (for instance it is in parallel with an inductor) it is suggestedthat you put a 1 ohm resistor in series with the probe. This will solve the problem of the voltageloop without affecting the simulator convergence or the steady-state solution.

Parameters

• np

Name of the positive node.

• nn

Name of the negative node.

• FUND_OSCxx

Keyword used when there is more than one fundamental frequency or for the case offrequency dividers. Default is FUND_OSC1. When a frequency divider is following anoscillator, and you wish to insert a .SSTPROBE between the two, this keyword must bespecified together with the int_val value which specifies the order of the division. Thesstprobe frequency will be int_val×FUND_OSCxx.

• RANK=int_val

Specifies the order of the division for frequency dividers. Must be specified together withFUND_OSCxx, see above.


SourcesProbe Source

Constraints

A .SSTPROBE is a component that must be inserted into the circuit for oscillator analysis. Whenno oscillator analysis is specified the probe is ignored and has no effect.

In the case of multi-oscillators analysis, there must be one .SSTPROBE FUND_OSCxx for eachFUND_OSC_GUESSxx specified in the .SST OSCIL command.

Example

For an oscillator followed by a frequency divider by 2:

.SSTPROBE osc_out 0 FUND_OSC1 RANK=2

SourcesPhase Noise Sources


Phase Noise SourcesIt is possible to define independent voltage and current sources of phase noise and to use themduring a noise/phase noise simulation (.SSTNOISE analysis). The phase noise current sourcemust also have a periodic signal specified. By omitting the periodic source specification fromthe voltage source, the phase noise is associated with the large signal SST voltage of the nodewhere the source is connected to.

Eldo RF assumes a zero current source or zero voltage source in any mode other thanSSTNOISE.

To generate output for this analysis use the .PLOT SSTNOISE or .PRINT SSTNOISEcommands, see .PLOT/.PRINT SSTNOISE.

Independent Phase Noise Source Syntax

Vxx np nn [PERIOD_SIG_SPEC] PHNOISE phnoise_param | value=exprIxx np nn PERIOD_SIG_SPEC PHNOISE phnoise_param | value=expr

Tabular Phase Noise Source Syntax

Vxx np nn [PERIOD_SIG_SPEC] PHNOISE TABLE [INTERP=HARM_DEC|HARM_OCT]+ (f1 val1) (f2 val2) ...Ixx np nn PERIOD_SIG_SPEC PHNOISE TABLE [INTERP=HARM_DEC|HARM_OCT]+ (f1 val1) (f2 val2) ...

Parameters

• PERIOD_SIG_SPEC

Can be:

FOUR, see “Multi-Tone Source” on page 116

SIN, see Sine Function in the Sources chapter of the Eldo User’s Manual

PULSE, see Pulse Function in the Sources chapter of the Eldo User’s Manual

• phnoise_param

Specified with the following parameters:

[PHN_FLOOR=val] [PHN_CORNER=val] [PHN_LEVEL=val] [PHN_SLOPE=val]

PHN_FLOOR

Specifies the level of phase noise floor. Units: 1/Hz. Default value is 0.0.

PHN_CORNER

Specifies the corner frequency between -30dBc/dec and -20dBc/dec zones. Units: Hz.

PHN_LEVEL

Specifies the phase noise level at the corner frequency. Units: 1/Hz. Default valueis 0.0.



PHN_SLOPE

Specifies the phase noise frequency dependence below the corner frequency. Defaultvalue is 1.0, meaning a 1/f3 dependence or a -30dBc/dec slope.

Figure 4-1. Phase Noise

NoteThe frequency of the phase noise sources are relative to (offset) the fundamentalfrequency of the periodic signal specified in the source.

Example:

Vphn n1 n2 FOUR fund1 MA (1) 1 -90+ PHNOISE PHN_FLOOR=1e-15 PHN_CORNER=1k PHN_LEVEL=1e-6

• value=expr

Defines an expression function of (noise) frequency. For example:

.param _pi=3.1415 p_fref=13e6

VN1 VDD 0 5 noise+ value = 2*_PI*sin(FREQ/p_fref)

.subckt DSQPHN outp outn ref param: fref=26meg ds_order=3 nscale=1.0

.param p_kds = '2*pi*2*pi/(12*fref)'

.param tom1='2*(ds_order-1)'Vn outp outn PHNOISE+ value = nscale * p_kds * pwr(2*abs(sin(pi * FREQ / fref)), tom1).ends

• INTERP=HARM_DEC|HARM_OCT

The method used to interpolate between the data points:

HARM_DECSpecifies a logarithmic interpolation around each harmonic of the .SSTNOISEanalysis.

PHN_CORNER

-20dBc/dec

-(2 + PHN_SLOPE) × 10dBc/dec

PHN_LEVEL

PHN_FLOOR

dBc

Frequency (Hz)



HARM_OCTSpecifies an octal interpolation around each harmonic of the .SSTNOISE analysis.

• PHNOISE TABLE

Keyword indicating that the phase noise has a tabular description. PHNOISE TABLE has thesame syntax as NOISE TABLE, see Noise Table Function in the Sources chapter of the EldoUser’s Manual.

val1, val2 ...

Corresponds to the double-sided Power Spectral Density of the source phasevariations at frequencies f1, f2, ... (in Hz-1).

Example:

V0 CKIN2 0 PHNOISE TABLE INTERP=HARM_DEC (10K 1E-12) (100K 1E-14)+ (1MEG 1E-16) (3.999G 1E-16) (3.9999G 1E-14) (3.99999G 1E-12)+ (4.00001G 1E-12) (4.0001G 1E-14) (4.001G 1E-16)

Correlation Between Noise SourcesIt is possible to define correlation coefficients between two independent noise sources with thecommand .NOISE_CORREL.

Syntax

.NOISE_CORREL VN1 VN2+ f k_r k_i

Parameters

• VN1, VN2

The two noise sources to be correlated.

• f, k_r, k_i

k_r and k_i define the real and imaginary parts respectively of the correlation coefficientbetween the two noise sources VN1 and VN2 at frequency f.

Example

V1 1 0 four fund1 ma (1) 1 -90+ noise table+ 100 1e-5+ 1k 1e-5+ 10k 1e-5

R1 1 3 1k

V2 2 0 dc 2+ noise table+ 100 1e-5+ 1k 1e-5+ 10k 1e-5



.noise_correl V1 V2*+ <f> <k_r> <k_i>+ 100 -1 0+ 1k -1 0+ 10k -1 0

R2 2 3 1kR3 3 0 1k

.sst fund1=1meg nharm1=2

.plot tsst v(3)

.sstnoise v(3) list 1k

.plot sstnoise onoise

Source Usage in Pre-Transient PhasePeriodic or non-periodic sources can be used during the pre-transient phase, but the non-periodic sources (for example PWL) will be considered as constant (DC source) during the SSTanalysis. This constant value will correspond to the value of the source at the end of the pre-transient phase.

Modulated sources and periodic FOUR sources can be deactivated (fully or partially) during thepre-transient phase. This is specified with the help of a global option:

.OPTION four_source_delay=val

Or by the parameter delay=val that can be locally specified to the different sources, forexample:

V1 n1 n2 FOUR delay=val ...

NoteThis delay parameter is only effective during Transient analysis (.TRAN), the Pre-Transient Phase, or Modulated Steady-State analysis (.MODSST). Its effect is the same asthe delay parameter of the SIN source. For more information on SIN sources see SineFunction in the Sources chapter of the Eldo User’s Manual

SourcesVoltage Controlled Oscillator Source


Voltage Controlled Oscillator SourceExx NP NN VCO HARM=(K) A=expr PHI=val VALUE=expr

This VCO source is available through the E element (VCVS) and is only supported in Eldo RF,not in Eldo. Multiple E-VCO devices are supported (during SST and MODSST analyses) withthe following constraints:

• During SST analysis: two E-VCO devices cannot affect the same fundamentalfrequency. An error is issued in this configuration.

• During MODSST analysis: two (or more) E-VCO devices can be specified affecting thesame fundamental frequency, but only the first one (in the netlist) will actually change it.The other(s) will be considered as modulation of the first one.

Parameters

• xx

Voltage controlled voltage source name.

• NP


• NN


• VCO

Keyword to specify a Voltage Controlled Oscillator source.

• HARM=(k)

Intermodulation specifying the output frequency is active on the harmonic k.

• A=expr PHI=val

Coefficients for the frequency domain. A and PHI are the amplitude and the phase indegrees respectively. A can be specified as an expression.

• VALUE=expr

Defines the frequency value of the harmonic k as a function of the voltage. Can be specifiedas an expression.

Examples

E1 1 0 VCO harm=(1) A=1.0 PHI=-90.0 VALUE=v(11)*1e7+1.0e08

VALUE defines the frequency value of the harmonic k, as a function of the voltage (v(11) inthis example).

E1 OSCOUT 0 VCO harm=(2) A=2*v(CTRL) PHI=-90.0+ VALUE=v(CTRL)*2.0e08+2.0e08VCTRL CTRL 0 dc 0.5 sin(0.5 0.25 1meg)


SourcesVoltage Controlled Oscillator Source

VALUE defines the frequency value of the harmonic k, as a function of the voltage (v(ctrl) inthis example). The parameter A defines the voltage of the output as twice the voltage acrossnode CTRL and ground.

NoteThis source cannot be used with a .SST OSCIL setup, only with a standard .SST.

The following example shows the syntax for multiple VCO sources in a .SST analysis:

E1 n1 n2 VCO harm(1,0) A=1.0 PHI=0.0 VALUE=1.0e9 + V(mod1) * 10MegHzE1 n3 0 VCO harm(0,1) A=1.0 PHI=-90 VALUE=(1.45e9 + V(mod2* 5MegHz

.SST fund1=1e9 nharm1=5 fund2=1.45e9 nharm2=3

A VCO example is supplied, run the file vco_source_step.cir, located in the directory:$MGC_AMS_HOME/examples/rfic/

SourcesDigitally Modulated Sources


Digitally Modulated SourcesDigitally modulated sources are mostly used with the .MODSST command. However, thesesources are also accepted with the transient analysis of Eldo (.TRAN).

It is also possible to handle frequency dependent devices such as LDTL (transmission lines) andFBLOCK (blocks defined by tabulated S, Y or Z parameters) with Modulated Steady-Stateanalysis (.MODSST).

Thirteen modulated sources are available. The syntax to instantiate a modulated source isalways the same whatever the source is:

Ixx np nn MSRC_TYP src_param [SAVE_xFILE=""file_name""]+ CAR_TYP car_param MOD_SIGNAL mod_paramVxx np nn MSRC_TYP src_param [SAVE_xFILE=""file_name""]+ CAR_TYP car_param MOD_SIGNAL mod_param

This manual includes a tutorial on using digitally modulated sources, see “Tutorial #12—Digitally Modulated Sources” on page 346. This tutorial is based on the supplied example,$MGC_AMS_HOME/examples/rfic/dig_mod_sources.cir.

NoteIn addition to the digitally modulated sources described here, you can use the EldoAmplitude Modulation function, AM, as a source. For details, see Amplitude ModulationFunction in the Sources chapter of the Eldo User’s Manual. However, you cannot use theSingle Frequency FM function, SFFM.

Parameters

• MSRC_TYP src_param

Keyword, specifying the modulated source type (MSRC_TYP), followed by parameters andvalues for the specified modulated source (src_param). The allowed keywords are:

GMSK GMSK (Gaussian Minimum Shift Keying) Source

GFSK GFSK (Gaussian Frequency Shift Keying) Source

OQPSK OQPSK (Offset Quaternary Phase Shift Keying) Source

PI4QPSK PI4QPSK (pi/4 Quaternary Phase Shift Keying) Source

MPSK MPSK (M-ary Phase Shift Keying) Source

MFSK MFSK (M-ary Frequency Shift Keying) Source

MQAM MQAM (M-ary Quadrature Amplitude Modulator) Source

IQMOD IQMOD (I-Q Modulator) Source

EDGE EDGE (8PSK Modulated Source with 3pi/8 Symbol Rotation forEDGE Standard) Source



• SAVE_xFILE

I and Q signals of any modulated source can be saved and stored into files using thefollowing parameters (in the corresponding modulated parameter specification):

SAVE_IFILE=""Ifile_name""

SAVE_QFILE=""Qfile_name""

SAVE_IQFILE=""IQfile_name""

• CAR_TYP car_param

Keyword, specifying the carrier type (CAR_TYP), followed by parameters and valuesdefining the carrier source (car_param). Allowed keywords to replace CAR_TYP are:

NoteThe parameters VO, VA and FR are required for specifying an Eldo RF SIN source (FRis optional in Eldo), and THETA is ignored. If FR is not specified, Eldo RF will return anerror stating that not enough parameters have been specified.

• MOD_SIGNAL mod_param

Keyword, specifying the modulation type (MOD_SIGNAL), followed by parameters definingthe modulation (mod_param). Allowed keyword to replace MOD_SIGNAL is:

Baseband SourcesAll the modulated source formats can be described as Baseband sources. The syntax to generatea Baseband source is the same as for a modulated source except that the carrier is put to DC(harmonic 0). Distinction between an I and a Q channel baseband signal is done through the useof the phase of the carrier. A zero degree phase corresponds to an I signal and a -90 degreephase corresponds to a Q signal.

Example

32QAM I & Q baseband signals:

VbbI 1 0 mqam m=32

OFDM OFDM (Orthogonal Frequency Division Multiplexing) Source

HPSK HPSK (Hybrid Phase Shift Keying) Source

CCK CCK (Complementary Code Keying) Source

ZIGBEE ZigBee (IEEE 802.15.4 standard for WPAN) Source

SIN Sine Function in the Sources chapter of the Eldo User’s Manual

FOUR “Multi-Tone Source” on page 116

PATTERN “Modulation Signal (PATTERN)” on page 148



+ four fund1 ma (0) 1 0+ pattern tsymb=1m prbs19

VbbQ 2 0 mqam m=32+ four fund1 ma (0) 1 -90+ pattern tsymb=1m prbs19


SourcesGMSK (Gaussian Minimum Shift Keying) Source

GMSK (Gaussian Minimum Shift Keying) SourceThis source format is used in communication standards such as GSM, DECT and CT2.

GMSK [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]

Parameters

• LPF=keyword

Specifies the baseband low-pass filter (see Filtering Information). Allowed keywords are:

• BETA=val

Normalized bandwidth for the Gaussian filter or Roll-Off factor for Raised Cosine andSquare Root Raised Cosine filters. Default value is 0.5.

• IASC=val

I channel amplitude scale. Multiplier of I to model amplitude imbalance. Default valueis 1.0.

• QASC=val

Q channel amplitude scale. Multiplier of Q to model amplitude imbalance. Default valueis 1.0.

Example

A simple example using default values for GMSK:

Vgmsk gmsk 0 RPORT=50 GMSK+ four fund1 MA (1) 2.0 -90+ pattern tsymb=Ts RANDOM

no_filter No filter

gaussian Gaussian filter (the default)

root_raised_cosine Square Root Raised Cosine filter

raised_cosine Raised Cosine filter

SourcesGFSK (Gaussian Frequency Shift Keying) Source


GFSK (Gaussian Frequency Shift Keying) SourceThis source format is used in communication standards such as 802.11 and bluetooth.

GFSK FDEV=val [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]

Parameters

• FDEV=val

Peak frequency deviation. No default value.

• LPF=keyword


• BETA=val


• IASC=val


• QASC=val


Example

Vgfsk gfsk 0 RPORT=50 GFSK fdev=1meg beta=.5+ four fund1 MA (1) 2.0 -90+ pattern tsymb=Ts PRBS12 INIT_PRBS="011100101100"

no_filter No filter

gaussian Gaussian filter (the default)




SourcesOQPSK (Offset Quaternary Phase Shift Keying) Source

OQPSK (Offset Quaternary Phase Shift Keying) SourceThis source format is used in communication standards such as cdmaOne.

OQPSK [LPF=keyword] [BETA=val] [IASC=val] [QASC=val] [FILT= CONV|NOCONV]

Parameters

• LPF=keyword


• BETA=val


• IASC=val


• QASC=val


• FILT= CONV|NOCONV

Specifies the algorithm used for the baseband signal filtering.

CONV

Real filtering is used, the input symbols are convoluted by the impulse response of thefilter.

NOCONV

Default. Pulse shaping is used, the input symbols are multiplied by the impulseresponse of the filter.

Example

Voqpsk oqpsk 0 RPORT=50 OQPSK lpf=raised_cosine beta=0.3+ four valfund1 MA (1) 2.0 -90.0+ pattern tsymb=Ts PRBS25 FEEDBACK=(1, 5, 8, 18)

no_filter No filter

gaussian Gaussian filter

root_raised_cosine Square Root Raised Cosine filter (the default)


SourcesPI4QPSK (pi/4 Quaternary Phase Shift Keying) Source


PI4QPSK (pi/4 Quaternary Phase Shift Keying) SourceThis source format is used in communication standards such as IS95.

PI4QPSK [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]+ [FILT= CONV|NOCONV]

Parameters

• LPF


• BETA=val


• IASC=val


• QASC=val




CONV


NOCONV


Example

Vpi4qpsk pi4qpsk 0 RPORT=50 PI4QPSK lpf=raised_cosine beta=0.5+ four valfund1 MA (1) 2.0 -90.0+ pattern tsymb=Ts PRBS12 INIT_PRBS="101010010111"

no_filter No filter





SourcesMPSK (M-ary Phase Shift Keying) Source

MPSK (M-ary Phase Shift Keying) SourceThis source format is used in communication standards such as cdmaOne, cdma2000.

MPSK M=val [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]+ [FILT= CONV|NOCONV]

Parameters

• M=val

Order of the signal space. M=2 corresponds to BPSK (Binary Phase Shift Keying) and M=4corresponds to QPSK. Constraint: M must be a power of 2.

• LPF=keyword


• BETA=val


• IASC=val


• QASC=val




CONV


NOCONV


Example

Vmpsk mpsk 0 RPORT=50 MPSK m=8 lpf=root_raised_cosine+ four fund1 MA (1) 2.0 -90+ pattern tsymb=Ts RANDOM

no_filter No filter




SourcesMFSK (M-ary Frequency Shift Keying) Source


MFSK (M-ary Frequency Shift Keying) SourceThis source format is used in communication standards such as 802.11 and bluetooth.

MFSK M=val FDEV=val [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]

Parameters

• M=val

Order of the signal space. M=2 corresponds to BFSK (Binary Frequency Shift Keying) andM=4 corresponds to C4FM. Constraint: M must be a power of 2.

• FDEV=val

Peak frequency deviation. No default value.

• LPF=keyword


• BETA=val


• IASC=val


• QASC=val


Example

Vxx n1 n2 MFSK M=4 FDEV=1.8kHz LPF=raised_cosine+ BETA=0.5+ FOUR FUND1 MA (1) 1 -90+ PATTERN tsymb=200u RANDOM

no_filter No filter





SourcesMQAM (M-ary Quadrature Amplitude Modulator) Source

MQAM (M-ary Quadrature Amplitude Modulator) SourceThis source format is used in communication standards such as WLAN transmission (802.11a).

MQAM M=val [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]+ [FILT= CONV|NOCONV]

Parameters

• M=val

Order of the signal space. Constraint: M must be a power of 2. Allowed values are 4, 8, 16,32, 64 and all the values under the form M= 22k where k is a positive integer.

• LPF=keyword


• BETA=val


• IASC=val


• QASC=val




CONV


NOCONV


Example

Vqam mqam 0 RPORT=50 MQAM m=16 lpf=gaussian beta=0.5

no_filter No filter




SourcesIQMOD (I-Q Modulator) Source


IQMOD (I-Q Modulator) SourceIQMOD [IASC=val] [QASC=val] IQ_SPEC

NoteMOD_SIGNAL cannot be specified with this source

This modulated source allows the possibility to define any arbitrary I-Q modulator by definingthe carrier and the I and Q signals using a PWL-like representation.

Parameters

• IASC=val


• QASC=val


• IQ_SPEC

Specifies the I and Q signals that are used for the modulation:

IQFILE=""<filename>"" | IFILE=""<filename>"" |+ QFILE=""<filename>""

NoteIQ_SPEC must be specified last

There are two ways to define the I and Q signals. They can be specified in the same file usingthe keyword IQFILE or in separate files, using the keywords IFILE and QFILE to specify thefilenames. These files are ASCII files and contain tabulated I and/or Q data in Cartesian format.

IQFILE

...tval1 Ival1 Qval1tval2 Ival2 Qval2tval3 Ival3 Qval3...

IFILE or QFILE

...tval1 IorQ_val1tval2 IorQ_val2tval3 IorQ_val3...


SourcesIQMOD (I-Q Modulator) Source

Example

V1 n1 n2 IQMOD four fund1 ma (1) 1 -90 IQFILE=""fileIQ.data""

SourcesEDGE (8PSK Modulated Source with 3pi/8 Symbol Rotation for EDGE Standard) Source


EDGE (8PSK Modulated Source with 3pi/8 SymbolRotation for EDGE Standard) Source

EDGE [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]

Parameters

• LPF=keyword


• BETA=val


• IASC=val


• QASC=val


Example

Vedge edge 0 rport=50 EDGE+ four valfund1 MA (1) 2.0 -90.0+ pattern tsymb=Ts RANDOM

no_filter No filter




edge Edge-Specific filter (the default)


SourcesOFDM (Orthogonal Frequency Division Multiplexing) Source

OFDM (Orthogonal Frequency Division Multiplexing)Source

This source format is used in communication standards such as 802.11 and DVB.

OFDM NB_CAR=val M=val MOD_TYP= MQAM|MPSK FFT_SIZE=val FDEV=val+ [CP_LEN=val] [ZP_LEN=val] [LPF=keyword] [BETA=val]+ [IASC=val] [QASC=val]

• NB_CAR=val

Number of sub-carriers composing the source.

• M=val

Order of the signal space of each sub-carrier. The value must be a power of 2, that is, 4, 8,16, 32, 64, and so on.

• MOD_TYP= MQAM|MPSK

Keyword defining the kind of modulation used for each sub-carrier. Allowed keywords areMQAM and MPSK.

• FFT_SIZE=val

Defines the number of points of the FFT used to compose the I and Q signals from the sub-carrier information. FFT_SIZE must be greater than NB_CAR.

• FDEV=val

Peak Frequency deviation. Sub-carrier frequency deviations are from -FDEV/2 to FDEV/2(and no signal at DC).

• CP_LEN=val

Length of the cyclic prefix referred to the signal period. Allowed values are between 0 and1. Default value is 0.

• ZP_LEN=val

Length of the zero padding prefix referred to the signal period. Allowed values are between0 and 1. Default value is 0.

• LPF=keyword


• BETA=val

Normalized bandwidth for the Gaussian filter or Roll-Off factor for Square Root RaisedCosine and Raised Cosine filters. Default value is 0.35.

no_filter No filter (the default)




SourcesOFDM (Orthogonal Frequency Division Multiplexing) Source


• IASC=val

I channel amplitude scale. Multiplier of I to model amplitude imbalance. Default is 1.0.

• QASC=val

Q channel amplitude scale. Multiplier of Q to model amplitude imbalance. Default is 1.0.

Note that the TSYMB parameter is not required in the PATTERN definition (refer to ModulationSignal (PATTERN)) because, for the OFDM source, the symbol period is defined from theFDEV, NB_CAR, CP_LEN and ZP_LEN values using the formula:

Example

An OFDM example is supplied, $MGC_AMS_HOME/examples/rfic/ofdm.cir. The following isthe contents of that file.

* OFDM source *

.param valfund1=900meg

Vofdm ofdm 0 RPORT=50 OFDM NB_CAR=nb_car M=8 MOD_TYP=MPSK FFT_SIZE=48+ FDEV=FDEV_val CP_LEN=CP_LEN ZP_LEN=ZP_LEN+ four valfund1 MA (1) 2.0 -90.0+ PATTERN randomRofdm ofdm 0 50

.param CP_LEN=0.

.param ZP_LEN=0.5

.param FDEV_val = 10Meg

.param nb_car = 10

.param T0= nb_car / FDEV_val

.param Ts=1+ZP_LEN+CP_LEN*T0

.param nbs=10

.param modsst_tstop=Ts*nbs

.param modsst_fft_tstart=4*Ts+(ZP_LEN+CP_LEN)*T0

.param modsst_fft_tstop=modsst_fft_tstart+T0

.sst fund1=valfund1 nharm1=2

.modsst 0 modsst_tstop

.plot fmodsst vi(ofdm).h(1) vr(ofdm).h(1)

.optfour tstart=modsst_fft_tstart tstop=modsst_fft_tstop nbpt=48normalized=1 interpolate=0.plot fourmodsst fourm(v(ofdm).h(1))

.end

TSYMB 1 ZP_LEN CP_LEN+ +( ) NB_CARFDEV

----------------------⋅=


SourcesHPSK (Hybrid Phase Shift Keying) Source

HPSK (Hybrid Phase Shift Keying) SourceThis source format is used in 3G communication standards such as WCDMA or CDMA2000, aswell as supporting the 3.5G standard HSDPA modulation format.

HPSK [SPREAD_FACTOR_CTRL=int_val] [GAIN_CTRL=val]+ [SPREAD_CODE_CTRL=keyword] [SPREAD_FACTOR_HSDPC=int_val]+ [GAIN_HSDPC=val] [SPREAD_CODE_HSDPC=keyword] [NB_CH_DATA=int_val]+ [SPREAD_FACTOR_DATA=int_val] [GAIN_DATAxx=val]+ [SPREAD_CODE_DATAxx=keyword]+ [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]+ [SCRAMBLING=WCDMA_UPLINK|WCDMA_DOWNLINK]+ [SCR_CODE=int_val] | [SCR_PRIM_CODE=int_val SCR_SEC_CODE=int_val]+ [FILT=CONV|NOCONV]

Parameters

• SPREAD_FACTOR_CTRL=int_val

Spreading factor for the control channel. Default value is SPREAD_FACTOR_DATA.

• GAIN_CTRL=val

Gain of the control channel. Default value is 1.0.

• SPREAD_CODE_CTRL=keyword

Keyword defining the spreading code for the control channel. Allowed keywords are:

WALSHyy

OVSFyy

where yy is an integer specifying the code number. The length of the code is equal to thespreading factor (of the control channel). Default code is WALSH0.

• SPREAD_FACTOR_HSDPC=int_val

Spreading factor for the High-Speed Dedicated Physical Control Channel (HS-DPCCH). Nodefault value. If not defined then no HS-DPCCH is considered.

• GAIN_HSDPC=val

Gain of the control channel. Default value is 1.0.

• SPREAD_CODE_HSDPC=keyword

Keyword defining the spreading code for the High Speed-Dedicated Physical ControlChannel (HS-DPCCH). Allowed keywords are:

WALSHyy

OVSFyy

where yy is an integer specifying the code number). The length of the code is equal to thespreading factor (of the HS-DPCCH). Default code is WALSH0.



• NB_CH_DATA =int_val

Defines the number of Data channels. int_val is an integer between 1 and 9.

• SPREAD_FACTOR_DATA=int_val

Spreading factor for the data channels. Default value is 4.

• GAIN_DATAxx=val

Gain for the data channel number xx. xx must be in the range of 1 and NB_CH_DATA.Default value is 1.0.

• SPREAD_CODE_DATAxx=keyword

Keyword defining the spreading code for the data channel number xx. xx must be in therange of 1 and NB_CH_DATA. Allowed keywords are:

WALSHyy

OVSFyy

Where yy is an integer specifying the code number. The length of the code is equal to thespreading factor (of the data channels). Default keyword is WALSH0.

• LPF=keyword


• BETA=val

Normalized bandwidth for the Gaussian filter or Roll-Off factor for Square Root RaisedCosine and Raised Cosine filters. Default value is 0.22.

• IASC=val

I channel amplitude scale. Multiplier of I to model amplitude imbalance. Default is 1.0.

• QASC=val

Q channel amplitude scale. Multiplier of Q to model amplitude imbalance. Default is 1.0.

• SCRAMBLING= WCDMA_UPLINK|WCDMA_DOWNLINK

Keyword specifying the kind of code used for the scrambling:

WCDMA_UPLINK

Default. Corresponds to the long uplink scrambling code (Gold sequence from twogenerator polynomials of degree 25).

no_filter No filter






WCDMA_DOWNLINK

Corresponds to the downlink scrambling code (Gold sequence from two generatorpolynomials of degree 18).

The description of the scrambling sequence generation can be found in the followingdocument: “3GPP Technical specification group radio access network; Spreading andModulation (FDD) v4.1.0 (2001-06)”.

• SCR_CODE=int_val

Defines the number of the scrambling code used as defined in the WCDMA standard.int_val is an integer between 0 and 8191. Default is 0.

• SCR_PRIM_CODE=int_val1 SCR_SEC_CODE=int_val2

Defines the number of the scrambling code used as defined in the WCDMA standard.

int_val1

An integer between 0 and 511 defining the primary scrambling code. Default is 0.

int_val2

An integer between 0 and 15 defining the secondary scrambling code. In thisconfiguration the scrambling code number is the following: 16×int_val1 +int_val2. Default is 0.


Specifies the algorithm used for the baseband signal filtering:

CONV


NOCONV


CAR_TYP and MOD_SIGNAL parameters can be specified, see “Digitally Modulated Sources” onpage 127 for the main syntax of CAR_TYP car_param and MOD_SIGNAL mod_param.

For the mod_param parameters defining the modulation (PATTERN), the following note applies.

NoteThe TSYMB parameter in the PATTERN definition defines the symbol period and the chipperiod TCHIP is computed from TSYMB and SPREAD_FACTOR:TCHIP = TSYMB/SPREAD_FACTOR

See “Modulation Signal (PATTERN)” on page 148 for more information on mod_param.



Examples

Vrf in 0 HPSK spread_factor_ctrl=4 spr_code_ctrl=walsh2+ gain_c=1.1 spread_factor_data=16 nb_ch_d=6 gain_d1=0.9+ scrambling=WCDMA_UPLINK+ lpf=gaussian beta=0.65 iasc=0.9 qasc=1.1+ four valfund1 MA (1) 1.0 -90+ pattern tsymb=1.042u prbs15

The following example is of a 3.5G standard HSDPA modulation format source:

Vrf in 0 HPSK spread_factor_ctrl=4 spr_code_ctrl=walsh2 gain_c=1.1+ spread_factor_hsdpc=4 spr_code_hsdpc=walsh4 gain_hsdpc=0.5+ spread_factor_data=16 nb_ch_d=6 gain_d1=0.9+ scrambling=WCDMA_UPLINK+ lpf=gaussian beta=0.65 iasc=0.9 qasc=1.1+ four valfund1 MA (1) 1.0 -90+ pattern tsymb=1.042u prbs15


SourcesCCK (Complementary Code Keying) Source

CCK (Complementary Code Keying) SourceUsed to used to encode data for 5.5 and 11Mbps data rates in the 2.4GHz band of 802.11bwireless networking.

CCK [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]

Parameters

• LPF=keyword


• BETA=val


• IASC=val


• QASC=val


no_filter No filter




SourcesZigBee (IEEE 802.15.4 standard for WPAN) Source


ZigBee (IEEE 802.15.4 standard for WPAN) SourceZigBee is the specification for a suite of high-level communication protocols based on the IEEE802.15.4 standard for Wireless Personal Area Networks (WPANs).

ZIGBEE [BAND=2450|900] [LPF=keyword] [BETA=val] [IASC=val] [QASC=val]

Parameters

• BAND=2450|900

Two PHYs are specified by the ZigBee standard, a 2450MHz PHY and a 868/915MHzPHY. Each having its own modulation format.

2450

Default. The 2450MHz PHY is used, this activates the spreading with 16 PN codesand OQPSK modulation.

900

The 868/915MHz PHY is used, this activates a differential encoding plus 15chip PNsequence spreading and BPSK modulation.

• LPF=keyword


When BAND is set to 2450 the default value for LPF is half_sine, otherwise it israised_cosine.

• BETA=val

Normalized bandwidth for the Gaussian filter or Roll-Off factor for Square Root RaisedCosine and Raised Cosine filters. Default is 1.0. When LPF is set to half_sine thisparameter is ignored.

• IASC=val

I channel amplitude scale. Multiplier of I signal to model imbalance. Default is 1.0.

• QASC=val

Q channel amplitude scale. Multiplier of Q signal to model imbalance. Default is 1.0.

no_filter No filter




half_sine Half-Sine Pulse Shaping filter


SourcesModulation Signal (PATTERN)

Modulation Signal (PATTERN)The only modulation signal allowed to be used in digitally modulated sources is PATTERN. ThePATTERN syntax inside Eldo RF is as follows:

PATTERN [DELAY=val] TSYMB=val [TTRANS=val]+ RANDOM|SYMBS|PRBSxx [INIT_PRBS=Bit_Stream]+ [FEEDBACK=(i1,...,in)]

Parameters

• DELAY=val

Delay before the PATTERN series is started. The value assigned during this time is the firststring value of the series. Default value is 0.

• TSYMB=val

Symbol period in seconds.

• TTRANS=val

Transition time (in seconds) between two consecutive symbols of the PATTERN. Thisparameter is ignored when filtering is activated. Default value is 0.0 in this case, otherwise itis set to HMIN (when no filtering).

• SYMBS

String of 1 and 0 values representing the symbols. For binary sources a symbol is composedof 1 bit, for M-ary signals a symbol is composed of log2M bits.

• RANDOM

For a random sequence of symbols, the keyword RANDOM can be specified instead of astring of 1’s and 0’s.

• PRBSxx

Specifies that the Pattern is generated with a Pseudo-Random Bit Sequence by a feedbackshift register. xx indicates the length of the generating shift register; the maximum lengthof the sequence will be 2xx−1 bits. xx must not be larger than 99.

• INIT_PRBS=Bit_Stream

Optional parameter used to initialize the shift register (can be used as a seed). The constraintis that Bit_Stream has to be xx bits long. The default value is 0 for all the bits except bitxx which is 1.

• FEEDBACK=(i1,...,in)

Optional parameter used to specify the registers that produce the feedbacks. The defaultvalues are dependent on the length of the shift register and such that they produce asequence of maximal length.

Example of Modulated Sources

Irf1 n1 n2 MQAM M=16 LPF=no_filter+ FOUR fund1 MA (1) 1.0 -90

SourcesModulation Signal (PATTERN)


+ PATTERN DELAY=10n TTRANS=10n TSYMB=800n RANDOMVrf2 n3 n4 GMSK FOUR fund1 MA (1) 1.0 0.0+ PATTERN DELAY=0 TSYMB=3.7u 110100011101101110010Vrf3 n5 n6 MFSK M=4 FDEV=1.4kHz LPF=root_raised_cosine+ BETA=0.4+ FOUR Fund1 MA (1) 1.0 45+ PATTERN TSYMB=50u PRBS12 INIT_PRBS="000110111010"+ FEEDBACK=(1,4,6)


SourcesFiltering Information

Filtering InformationModulated sources are usually filtered by a baseband low-pass filter to limit their spectralbandwidth. The baseband signal filtering is achieved by pulse shaping. This means that eachsymbol is shaped in order to be similar to the impulse response of the specified filter. This isobtained by multiplying each symbol by the corresponding impulse response. It is possible toactivate a real filter (rather than a pulse shaping filter) by specifying FILT=CONV on themodulated source instance.

There are five available filters:

• Gaussian

• Raised Cosine

• Square Root Raised Cosine

• Edge-Specific

• Half-Sine Pulse Shaping

Their transfer functions as described in the following subsections.

Filter Transfer Functions

Gaussian Filter

Raised Cosine Filter

H f( ) 2ln2

--------T symb f⋅

BT----------------------

2

– exp=

H f( )

1

12--- 1 π

T symb

BT-------------- f 1 BT–( )

2T symb---------------------–

cos+

0

=

0 f1 BT–( )2T symb

---------------------≤ ≤

1 BT–( )2T symb

--------------------- f1 BT+( )2T symb

----------------------≤ ≤

1 BT+( )2T symb

---------------------- f≤



Square Root Raised Cosine Filter

Reference: B. Sklar, “Digital Communications, fundamentals & applications” PrenticeHall 1988.

Edge-Specific Filter

Each phasor i corresponding to an ith symbol is multiplied by the following shaping function:

where

See EDGE specifications, GSM 05.04 v8.1.0 (1999-12) ETSI.

H f( )

1

12--- 1 π

T symb

BT-------------- f 1 BT–( )

2T symb---------------------–

cos+

0

=

0 f1 BT–( )2T symb

---------------------≤ ≤

1 BT–( )2T symb

--------------------- f1 BT+( )2T symb

----------------------≤ ≤

1 BT+( )2T symb

---------------------- f≤

c0 t i– Tsymb⋅ 2.5Tsymb+( )

co t( )S t i Tsymb⋅+( )

i 0=

3

∏ for 0 t 5 Tsymb⋅≤ ≤

0 else

=

S t( )

π g τ( ) τd

0

t

∫

sin for 0 t 4 Tsymb⋅≤ ≤

π2--- π g τ( ) τd

0

t 4 Tsymb⋅–

∫–

sin for 4 Tsymb⋅ t 8 Tsymb⋅≤ ≤

0 else

=

g t( ) 12 Tsymb⋅------------------------ Q 2π BT⋅ 1 2.5Tsymb–

Tsymb 2ln---------------------------------

Q 2π BT⋅ 1 1.5Tsymb–

Tsymb 2ln---------------------------------

– =

Q t( ) 1

2π---------- e

τ2

2----–

τd

t

∞

∫=



Half-Sine Pulse Shaping Filter

For 0 ≤ t ≤ 2Tc

Otherwise, p(t) = 0

Reference: IEEE Standard 802.15.4™-2003, Part 15.4: Wireless Medium Access Control(MAC) and Physical Layer (PHY) Specifications for Low-Rate Wireless Personal AreaNetworks (LR-WPANs)

p t( ) π t2T c---------⋅

sin=


Chapter 5Command Options

SummaryThis section relates to the .OPTION command in Eldo RF. The first table lists all the options inalphabetical order. The following tables list the options divided into categories.

Table 5-1. Eldo RF Options

AUTOSTOP FOUR_SOURCE_DELAY

IMPROVED_SSTNOISE_PERF MODSST_EPS

MODSST_CENTRAL_FUND_OSCxx MODSST_FULL_DISPLAY

MODSST_FFT_FUND_FREQ MODSST_FFT_NHARM

MODSST_FFT_TSTART MODSST_FULL_DISPLAY_FORCED

MODSST_HMAX MODSST_HMIN

MODSST_USE_AVERAGE_FUND_OSC NO_SST

RF_PARTITIONING_MODE RF_PARTITIONING_THRESHOLD

SST_ABSTOL SST_ACCURACY

SST_AT_TIME SST_CIRCUIT_TYPE

SST_CONVERGENCE_HELP SST_ESTIM_ACCURACY

SST_F0_ABSTOL SST_F0_RELTOL

SST_FULL_DISPLAY SST_KEEP_OPTIONS_FOR_SWEEP

SST_MAX_LINITER SST_MEMESTIM

SST_MEMORY_COMPRESS SST_MTHREAD

SST_NBTHREAD SST_NDIM_FFT

SST_NODIVERGENCE SST_NOLIMIT_LINITER

SST_NPER SST_NPT

SST_NTONE_PROCEDURE_IFUND_FOR_RESTART

SST_OSC_KEEP_PHASE_SEQUENCE

SST_OSC_PHASE_SEQUENCE SST_OVRSMP

SST_PHNOISE_SPEED SST_PLL_VCO_WITH_GLOBAL_SPECTRUM


Command OptionsSummary

Options divided into categories:

SST_PRECONDITION SST_RAMPING_FACTOR

SST_RESTART SST_SPECTRUM

SST_T0HF SST_TOT_TIME_POINTS_LIMIT

SST_TRAN_NPER SST_TSTART

SST_TSTOP SST_UIC

SST_USE_NTONE_PROCEDURE SST_VERBOSE

SSTNLCONTRIB_FILE SSTNOISE_CONTRIB_TYPE

SSTNOISE_EXCLUDE_DEVICES SSTNOISE_FILE

SSTNOISE_GLOBPART SSTNOISE_INCLUDE_DEVICES

SSTNOISE_SORT_ABS SSTNOISE_SORT_CRITER

SSTNOISE_SORT_NBMAX SSTNOISE_SORT_REL

SSTSENSRLC_FILE TUNING

Table 5-2. Time Domain Options

SST_FULL_DISPLAY SST_NPER

SST_NPT SST_RESTART

SST_TSTART SST_TSTOP

SST_TOT_TIME_POINTS_LIMIT SST_T0HF

Table 5-3. Accuracy Options

MODSST_EPS SST_ABSTOL

SST_ACCURACY SST_ESTIM_ACCURACY

SST_NDIM_FFT SST_OVRSMP

SST_PHNOISE_SPEED SST_SPECTRUM

TUNING SST_PLL_VCO_WITH_GLOBAL_SPECTRUM

Table 5-4. Convergence Options

SST_AT_TIME SST_CONVERGENCE_HELP

SST_MAX_LINITER SST_NODIVERGENCE

SST_NOLIMIT_LINITER SST_NTONE_PROCEDURE_IFUND_FOR_RESTART

Table 5-1. Eldo RF Options

Command OptionsSummary


SST_OSC_KEEP_PHASE_SEQUENCE SST_OSC_PHASE_SEQUENCE

SST_PRECONDITION SST_RAMPING_FACTOR

SST_TRAN_NPER SST_USE_NTONE_PROCEDURE

Table 5-5. Frequency Tolerance Options

SST_F0_ABSTOL SST_F0_RELTOL

Table 5-6. Noise Result Presentation Options

IMPROVED_SSTNOISE_PERF SSTNOISE_CONTRIB_TYPE

SSTNOISE_EXCLUDE_DEVICES SSTNOISE_FILE

SSTNOISE_GLOBPART SSTNOISE_INCLUDE_DEVICES

SSTNOISE_SORT_ABS SSTNOISE_SORT_CRITER

SSTNOISE_SORT_NBMAX SSTNOISE_SORT_REL

Table 5-7. Multi-threaded Simulation Options

SST_MTHREAD SST_NBTHREAD

Table 5-8. MODSST Analysis Options

MODSST_CENTRAL_FUND_OSCxx MODSST_FFT_FUND_FREQ

MODSST_FFT_NHARM MODSST_FFT_TSTART

MODSST_FULL_DISPLAY MODSST_HMAX

MODSST_FULL_DISPLAY_FORCED MODSST_USE_AVERAGE_FUND_OSC

MODSST_HMIN RF_PARTITIONING_THRESHOLD

RF_PARTITIONING_MODE

Table 5-9. Miscellaneous Analysis Options

AUTOSTOP FOUR_SOURCE_DELAY

NO_SST SST_CIRCUIT_TYPE

SST_KEEP_OPTIONS_FOR_SWEEP SST_MEMESTIM

SST_MEMORY_COMPRESS SST_UIC

SST_VERBOSE SSTNLCONTRIB_FILE

SSTSENSRLC_FILE

Table 5-4. Convergence Options


Command OptionsEldo RF Options

Eldo RF Options

Time DomainSteady-State analysis generates a frequency domain representation of signals. However, signalscan also be plotted in the time domain. Some parameters allow you to specify the way timedomain values are reconstructed.

For a single-tone simulation the SST_TSTART and SST_TSTOP options define the time windowin which the steady-state waveforms will be plotted. The SST_NPER option defines the numberof periods of the steady-state waveforms that will be plotted for a periodic signal. The SST_NPT

option is used to define the number of points per period.

• SST_TSTART

Specifies the lowest time in seconds for which the steady-state waveforms should beplotted. Default value is 0.

• SST_TSTOP

Specifies the highest time in seconds for which the steady-state waveforms should beplotted. Default value is one period of the lowest fundamental frequency.

• SST_NPER

Specifies the number of periods of the steady-state waveforms which should be plotted inthe case of periodic signals, or the number of periods of the lowest frequency harmonic forthe pseudo-periodic signals. Default value is 2 for single-tone signals, and 1 for multi-tonesignals.

NoteA maximum of two of the previous three parameters can be set simultaneously.

• SST_NPT

For single-tone signals this specifies the number of points per period. For multi-tone signalsit specifies the number of points. Default value is 64 in both cases.

For a multi-tone simulation you can reduce an excessively large number of time-points by usingthe following options; the SST_FULL_DISPLAY option is used to control the generation of theTSST waveforms, either a full or a reduced waveform can be generated. When the number ofTSST time-points generated exceeds the value of the SST_TOT_TIME_POINTS_LIMIT option,the SST_FULL_DISPLAY option will be forced to 1. The SST_TSTART and SST_TSTOP optionsdefine the time window in which the steady-state waveforms will be plotted. The SST_T0HFoption defines the period of the maximum frequency. The SST_NPER option defines the numberof periods of SST_T0HF displayed in the time window. The number of points generated duringthe period SST_T0HF is defined by SST_NPT.



• SST_FULL_DISPLAY=1|2

Specifies the duration of the plotting between time points, “1” means that between two timepoints, the duration of plotting is based on the SST_NPER option. “2” means the completewaveform will be plotted in the TSST plot time window. The time window can be adjustedwith the SST_TSTART and SST_TSTOP options. Default value is “2”, however, this canbe forced to “1” by the SST_TOT_TIME_POINTS_LIMIT option.

• SST_TOT_TIME_POINTS_LIMIT=val

Specifies the threshold at which the value of the SST_FULL_DISPLAY option is forced to“1”. The value specified corresponds to the number of plotted TSST time points given by:

Number of TSST waveforms × Number of time points

This option has no default value, therefore, if omitted the SST_FULL_DISPLAY option willnot be forced to “1”.

• SST_T0HF=val

Specifies the period of the high frequency signal fmax. Default value is 1/fmax, where fmaxis the maximum fundamental frequency defined on the .SST command. The number oftime points plotted during this time can be defined on the SST_NPT option.

• SST_RESTART=for_sst|for_pretran

Specifies whether the simulation results that were created with the .save ... sst

command will be used to restart either an SST (for_sst) or a pre-transient (for_pretran)analysis. The default is for_sst. If you have pre-transient options in the netlist andSST_RESTART is set to for_sst, the .sst file will be used to restart an SST analysis. InEldo RF v 6.7 and earlier, leaving the pre-transient options in the netlist would use the .sstfile to restart a pre-transient analysis.

AccuracyThe accuracy of steady-state results is affected by two sources of error:

• round-off error of the Newton algorithm used to solve the system of steady-stateequations

• aliasing related to the spectrum truncation (limitation of the number of harmonics)

In Eldo RF, the Newton error is controlled by the SST_ABSTOL option, this error is negligiblefor more than 99.9% of the circuits when the default value of the option is used.

The major source of inaccuracy is due to aliasing. The SST_ESTIM_ACCURACY option can beused to monitor and estimate this aliasing error.

The aliasing error can be decreased by increasing the number of harmonics, and/ or by using thefollowing options: SST_OVRSMP, SST_SPECTRUM and/ or SST_NDIM_FFT.



Another source of error exists for MODSST results called LTE (Local Truncation Error). Thiserror is similar to TRAN LTE, it is related to the integration of differential equations withnumerical methods. This error can be controlled using the MODSST_EPS option.

For help with convergence and result accuracy, please refer to Chapter 7, “ConvergenceTroubleshooting” of this manual.

• SST_OVRSMP

Defines the oversampling factor that increases the number of time points for the FFT. Thisoption is used to minimize aliasing and improve simulation accuracy. Default value is 1,meaning no oversampling. Reasonable values for sst_ovrsmp range from 1 to 5 or 10 (100is unreasonable).

• SST_SPECTRUM

Defines the spectrum truncation in the case of multi-tone analysis. Allowed values are 0 or 1(0 for a diamond truncation and 1 for a box truncation). Default value is 0.

• SST_PLL_VCO_WITH_GLOBAL_SPECTRUM

Enables the calculation of the VCO partition steady-state using the full PLL spectrumrepresentation. More harmonics are used in the analysis and therefore the simulation will bemore accurate.

• SST_NDIM_FFT

(0 or 1). Enables the simulation results to be improved in the case of aliasing problems fortwo or more tone analysis. The use of this option will slow down the simulation, but this isthe trade off for accuracy. This option is set to 0 by default.

Theoretical background: To handle N-tones, two different methods are implemented. Thefirst one is a mapping of the N-dimensions to one artificial dimension in order to use a 1-dimension FFT. The other method directly uses an N-dimension FFT. The first method isfaster because it uses less time points than the second one, but on the contrary the aliasing ismore important. Method one is used by default (sst_ndim_fft=0).

• SST_ABSTOL

This option controls the convergence criterion of the Steady-State. The default value is1×10-5. This value is well suited for most circuits, however in the case of circuits with verylow level inputs, it could be necessary to lower this value. A reasonable range for this optionis 1×10-7 to 1×10-4.



• SST_ACCURACY

This option allows you to select a different accuracy level to be used for the simulation.Each value of the option corresponds to a set of the following options: SST_SPECTRUM,SST_NDIM_FFT, SST_ABSTOL.

By default, the simulation is carried out with SST_ACCURACY set to standard. It isimportant to emphasize that the tighter the option, the higher the demand on the CPU andmemory.

• SST_ESTIM_ACCURACY=keyword

This option allows you to specify an estimation of the accuracy of the steady-state results.This information is available by setting this option and the required results are printed inASCII form in the .chi file. The printed results correspond to the estimated error (due toaliasing) on the different harmonics of the steady-state solution. This error can be displayedwith different formats (absolute magnitude (MAG), relative magnitude (DB), or percent).

The keyword structure is:

<output_type>_<number_of_contributions>

Example:

.OPTION SST_ESTIM_ACCURACY = ALL_15

In this case, for the 15 elements of the steady-state solution where the error is the maximum,Eldo RF will print the estimated error in all the different forms (PERCENT, MAG and DB).

• SST_PHNOISE_SPEED=FAST|MEDIUM|STANDARD

This can significantly reduce the phase noise simulation time for autonomous circuits(oscillators), but there is a trade-off between speed and accuracy. This command can be

Table 5-10. SST_ACCURACY

SST_ACCURACY SST_SPECTRUM SST_NDIM_FFT SST_ABSTOL

MODERATE 0 0 1e-04

STANDARD 0 0 1e-05

HIGH 1 1 1e-06

Table 5-11. SST_ESTIM_ACCURACY

output_type Available output types:PERCENT, PercentageMAG, MagnitudeDB, Magnitude in dBALL, all types

number_of_contributions Specify number of contributions as aninteger value or all:<int_val>|ALL



used for forced (non-autonomous) circuits, but only with the MEDIUM or STANDARDoptions.

FAST

Significantly reduces the phase noise simulation time. This method will provideaccurate results for approximately 80% of netlists when compared with theSTANDARD method. This is not an option for forced circuits.

MEDIUM

Reduces the phase noise simulation time by half when compared with the STANDARDmethod. This method provides accurate results for approximately 99% of netlistswhen compared with the STANDARD method.

STANDARD

Calculates the phase noise level to the highest level of accuracy. Default.

• MODSST_EPS=val

Specifies the internal accuracy of the MODSST variables (real and imaginary parts of nodevoltage harmonics). Decreasing this value will directly impact the number of computed timepoints. Default value is 1.0×10-3.

• TUNING=BACKANNOTATE|SST_BACKANNOTATE|NOSST_BACKANNOTATE

When set to SST_BACKANNOTATE the improved Eldo RF solver will be used for handlingback-annotated netlists with many parasitic elements. It can provide significant capacity andspeed improvements (up to 10 ) for all RF analyses for circuits that contain parasiticelements. The option is most efficient when the parasitics are defined in DSPF format in aseparate file. When set to NOSST_BACKANNOTATE (default) the improved Eldo RF solverwill be disabled.

NoteWhen set to BACKANNOTATE the improved analog solver will be enabled for TRAN, AC,DC and all RF analyses.

Convergence

For help with convergence and result accuracy, please refer to Chapter 7, “ConvergenceTroubleshooting” of this manual.

• SST_MAX_LINITER=VAL

Defines the maximum number of iterations for the linear solver. Default value is 20. Themore a circuit is non-linear, the more iterations required. For non-linear circuits, increasingthis option up to 100 or 200 might help.



• SST_NTONE_PROCEDURE_IFUND_FOR_RESTART=VAL

This option allows you to choose in which (M-tone) steady-state a .RESTART SST will beperformed when an N-tone steady-state is specified in the netlist (where M is less than orequal to N). (By default, a .RESTART SST command is performed in the N-tone steady-state.)

VAL specifies the number of tones of the steady-state.

For example, consider a 3-tone circuit (with the SST_USE_NTONE_PROCEDURE option) andimagine you have previously saved a 2-tone steady-state in a file named save.sst. If youspecify .RESTART SAVE.SST SST in the 3-tone circuit, the results saved in save.sst will beused to initialize the 3-tone simulation. But if you specify:

.OPTION SST_NTONE_PROCEDURE_IFUND_FOR_RESTART = 2

the results saved in save.sst are used to initialize a 2-tone simulation and, when theconvergence is reached, the 3-tone simulation is performed.

• SST_NOLIMIT_LINITER=VAL

This option is used to automatically increase the maximum number of iterations specified onthe option SST_MAX_LINITER. The maximum value that can be specified is 500.

• SST_NODIVERGENCE=KRYLOV

When the number of iterations (of the linear solver) performed by Eldo RF is equal to thevalue specified on the option SST_MAX_LINITER divergence has occurred. When thisoption is specified and divergence occurs, Eldo RF will not perform any more iterations anddivergence is ignored (Eldo RF will assume that the linear solver has converged). If thisoption is omitted and the value of SST_MAX_LINITER has been exceeded Eldo RF willperform source stepping to converge the design.

• SST_RAMPING_FACTOR

This option enables the analysis of strongly non-linear circuits by using a ramping sourceprocess. The value of the option corresponds to the increment applied to the non DCsources. (For example, let us consider SST_RAMPING_FACTOR=0.25. First Steady-Stateanalysis is performed with all RF sources at their nominal value multiplied by 0.25. Oncethis analysis is complete, a second analysis is performed with a ramping source factor set to0.5. This operation continues until the ramping source factor reaches 1.0.) By using thisoption, divergence can be avoided during the simulation, and CPU time savings can beobtained. The default value is 1.0 (means no ramping).

This option can be used in cases of N-tone autonomous circuit analysis (optionSST_USE_NTONE_PROCEDURE) and autonomous circuit analysis. The ramping factor isapplied only to all RF generators.

• SST_USE_NTONE_PROCEDURE

(0 or 1). This option controls the N-tones Steady-State analysis procedure. When activated,the simulator will perform a succession of Steady-State Analyses, each time using the resultof the previous analysis as the initial condition for the next. The first and second analyses



are one tone and at each further analysis, the number of tones is incrementally augmented,until the required number is reached. This option should save CPU time. Default value is 0.

NoteThis procedure is also available for autonomous circuit analysis and it is possible toimprove the CPU time in case of multi-tone autonomous simulation.Run the examples: vcomixerlna.cir and vcomixerlna_ntone_procedure.cir for moreinformation.

• SST_PRECONDITION=ADAPTIVE|TIME|TIME_MODERATE|TIME_ACCURATE|TIME_FOR_NOISE_ONLY|TIME_MODERATE_FOR_NOISE_ONLY|TIME_ACCURATE_FOR_NOISE_ONLY

The option SST_PRECONDITION is used to improve the convergence of the Krylov method,in some cases requiring many iterations. The greater the accuracy of the preconditioner, thegreater the reduction in the number of Krylov iterations, however, each Krylov iteration willrequire more CPU time. The potential gain in CPU time is a tradeoff between the work doneby the preconditioner and the resulting decrease in Krylov iterations. This is circuitdependent and cannot be predicted.

It is recommended that you should first use SST_PRECONDITION=TIME, because a moreaccurate preconditioner may not be required for convergence. If the simulation does notconverge, you should then try SST_PRECONDITION=TIME_MODERATE, then trySST_PRECONDITION=TIME_ACCURATE. In all cases, the accuracy of the preconditioner doesnot impact the accuracy of the simulation, it only impacts the convergence of the Krylovmethod, that is, the number of iterations and the required CPU time.

ADAPTIVE

Activates a method aimed at enhancing the efficiency of the frequency preconditionerfor strongly non-linear circuits.

TIME

Activates a time preconditioner which improves the convergence of the Krylovmethod in some cases requiring many iterations (typically those for which the optionSST_MAX_LINITER must be set). Usage up to now revealed it is particularly beneficialfor frequency divider and multiplier circuits. Restrictions are listed in Table 5-12.This option is also available for oscillators, but is recommended only for oscillator +divider circuits (although it may prove useful for other specific types of circuits).

Run the example named div2_4_time_preconditioner.cir for more information.

TIME_MODERATE

Activates a moderately accurate variant of the time preconditioner. In some raresituations, SST_PRECONDITION=TIME might induce a reduction of the Kryloviterations at each Newton step but introduce some inaccuracy that prevents the



Newton solver from converging. In such a situation, setting this option toTIME_MODERATE may prove useful.

TIME_ACCURATE

Activates a more accurate variant of the time preconditioner. In some rare situations,SST_PRECONDITION=TIME might induce a reduction of the Krylov iterations at eachNewton step but introduce some inaccuracy that prevents the Newton solver fromconverging. In such a situation, setting this option to TIME_ACCURATE may proveuseful.

TIME_FOR_NOISE_ONLY

Activates a time preconditioner for only the SSTNOISE analysis. The iterative timepreconditioner is implicitly based on a linear-time varying approximation of thequasi-periodic signal over a period. This option will enable the time preconditioner toaccelerate the computation of the noise contributions once the steady-state conditionis reached. The time preconditioner for SSTNOISE analysis can be activated byspecifying TIME_FOR_NOISE_ONLY on the option SST_PRECONDITION. Restrictionsare listed in Table 5-12.

TIME_MODERATE_FOR_NOISE_ONLY

Activates a moderately accurate variant of the time preconditioner for only theSSTNOISE analysis. The iterative time preconditioner is implicitly based on alinear-time varying approximation of the quasi-periodic signal over a period. Thisoption will enable the time preconditioner to accelerate the computation of the noisecontributions once the steady-state condition is reached. The time preconditioner forSSTNOISE analysis can be activated by specifyingTIME_MODERATE_FOR_NOISE_ONLY on the option SST_PRECONDITION. Restrictionsare listed in Table 5-12.

TIME_ACCURATE_FOR_NOISE_ONLY

Activates a more accurate variant of the time preconditioner for only the SSTNOISEanalysis. The iterative time preconditioner is implicitly based on a linear-time varyingapproximation of the quasi-periodic signal over a period. This option will enable thetime preconditioner to accelerate the computation of the noise contributions once thesteady-state condition is reached. The time preconditioner for SSTNOISE analysiscan be activated by specifying TIME_ACCURATE_FOR_NOISE_ONLY on the optionSST_PRECONDITION. Restrictions are listed in Table 5-12.

• SST_CONVERGENCE_HELP=TRANSIENT|NO_TRANSIENT|CONTINUATION|ADVANCED_NEWTON|PSEUDO_MODSST

This option activates algorithms in order to help steady-state convergence. It is particularlyadapted to high Q oscillators or circuits exhibiting convergence difficulties.

TRANSIENT

This keyword activates a transient analysis prior to the steady-state and initializes thesteady-state resolution with the transient results. When it is activated, the transientanalysis is performed for 10 periods of the fundamental frequency FUND1. Thenumber of periods can be adjusted with the following option:

.OPTION SST_TRAN_NPER=val



This algorithm already exists in Eldo RF since the v5.4 version and was used forfrequency divider analysis. It is automatically activated when frequency division isdetected or can be manually set. Use the SST_CONVERGENCE_HELP=NO_TRANSIENToption to turn this off.

NO_TRANSIENT

This keyword deactivates the transient analysis that is automatically activated whenfrequency division has been detected.

CONTINUATION

This keyword activates the continuation algorithm for oscillator steady-state analysis.

NoteRun the example named osc_sst_continuation.cir for more information.

Some restrictions about the use of these options exist:

ADVANCED_NEWTON

This keyword activates a method to enhance the convergence of Newton method forSST analysis and partly during SST OSCIL analysis (especially when a transient

Table 5-12. Convergence Option Restrictions

SST_CONVERGENCE_HELP= CONTINUATION

SST_PRECONDITION = TIME \TIME_MODERATE \TIME_ACCURATE \TIME_FOR_NOISE_ONLY \TIME_MODERATE_FOR_NOISE_ONLY \TIME_ACCURATE_FOR_NOISE_ONLY

SST, SSTAC,SSTNOISE,SSTXF Analysesfornon-autonomouscircuit

Not Available YESRestrictions: Integ() and LPF() effects inVerilog-A models, N-tone simulation

SST, SSTAC,SSTNOISE,SSTXF Analysesfor autonomouscircuit

YES YESRestrictions: as above

MODSST analysisfornon-autonomouscircuit

Not Available YESRestrictions: Can not be used with netlistscontaining R(f) and L(f) and as above

MODSST analysisfor autonomouscircuit

Not Available YESRestrictions: as above



phase is required, as for oscillator + divider circuits, or when the optionSST_CONVERGENCE_HELP=TRANSIENT is specified).

NoteThis option is irrelevant for SST OSCIL analysis whenSST_CONVERGENCE_HELP=CONTINUATION is set.

This option sometimes benefits from setting the option SST_PRECONDITION=TIME(there is a synergy between both options). This option can also be used in conjunctionwith MODSST analyses.

PSEUDO_MODSST

This keyword emulates a modulated steady-state to reach the steady-state. This optionis only available for steady-state oscillator computation. For the use of this options inpre-transient phase see “Pre-Transient Phase” on page 37.

• SST_TRAN_NPER=val

Sets the number of periods of FUND1 over which a transient analysis is performed. Defaultis 10. This transient analysis is performed only when a frequency divider is detected, or ifthe SST_CONVERGENCE_HELP=TRANSIENT option is specified. For the use of this options inpre-transient phase see “Pre-Transient Phase” on page 37.

• SST_AT_TIME=val

Is used to activate the pre-transient phase and set the transient duration. For the use of thisoptions in pre-transient phase, see “Pre-Transient Phase” on page 37.

• SST_OSC_PHASE_SEQUENCE=SEQ_1|SEQ_2|SEQ_3|SEQ_4|SEQ_5

This option is used to help steady-state convergence for highly non-linear oscillators. Formore information, see “SST_OSC_PHASE_SEQUENCE” on page 36.

• SST_OSC_KEEP_PHASE_SEQUENCE=0|1

When set to 1 the same sequence of algorithms will be used at each point of the sweepanalysis (.STEP). When set to 0 (default) the specified set of algorithms will only be usedfor the first point of the sweep. Used in conjunction with the optionSST_OSC_PHASE_SEQUENCE.

Note.RESTART or .IC commands can be used to kick the pre-transient phase and the value ofnon periodic sources during SST. This corresponds to those at the end of the pre-transientphase.

Frequency Tolerance OptionsTo improve speed without entailing accuracy convergence tests for oscillators were changed tothe following ones.



By default, a relative precision of 1.0e-6 on the oscillating frequency is required. For SSTAC,SSTNOISE and SSTXF analyses, additional precision is required and the tolerance on thecomputed oscillating frequency is set to half the perturbation frequency whenever this value issmaller than 1.0e-6 times the oscillating frequency.

In addition, you may still require extra precision on the oscillating frequency by setting itsabsolute precision and/or its relative precision with the two options specified below.

NoteWhenever both the below options are specified, the resulting smaller tolerance is used.

• SST_F0_ABSTOL=val

Sets the absolute precision of the oscillating frequency

• SST_F0_Reltol=val

Sets the relative precision of the oscillating frequency.

Noise Result PresentationSee “.SSTNOISE” on page 53 for details of the Steady-State Noise analysis command.

In order to improve the readability of the individual device noise contributions, an additionalcompact table is printed in the .chi file. Instead of presented separately as in the main tables, thedevice types are mixed, so it is easier to identify the noisiest devices. Mostly, only the deviceswith the highest contributions are of interest. Therefore, it is possible to restrict the size of both(main and compact) tables, using the three options below:

• SSTNOISE_SORT_NBMAX=val

Option used to limit the number of noisy devices whose contribution are printed. If thisoption is set then only the specified number of noisy devices with maximum contributionsare printed in the file.chi. No default value, meaning that all the devices are printed bydefault.

• SSTNOISE_SORT_REL=val

Option defining a threshold below which noisy device contributions are not printed. Thethreshold is computed as the total_noise_value multiplied by the SSTNOISE_SORT_RELvalue. Default value is 0, meaning that all the devices are printed.

• SSTNOISE_SORT_ABS=val

Option defining a threshold below which noisy device contributions are not printed. Theoption directly defines the value of the threshold. No default value, meaning that all thedevices are printed.



• IMPROVED_SSTNOISE_PERF

This option activates an improved performance algorithm for SSTNOISE analysis. It canlead to a 2× to 5× speed ratio compared to the default algorithm for most circuits. However,this algorithm requires more memory, and may also take a small number more iterations,and in a few cases might not be as robust, though the simulation results are not affected.

• SSTNOISE_FILE[=filename]

Print the SSTNOISE results (individual contributions of noisy devices) into a separate file,rather than in the normal .chi file. When filename is not specified with this optionactivated, then the results will be printed into a file named the same as the netlist but with a.sstnoise extension.

• SSTNOISE_CONTRIB_TYPE=SSB|DSB

Define whether individual contributions of SPHI and AMNOISE are printed as DSB resultsor SSB results. Default is DSB. This option only affects the individual contributions ofnoisy devices printed in the ASCII file and not the plotted results (.PLOT).

• SSTNOISE_SORT_CRITER=PHNOISE|AMNOISE|ONOISE|NAME

Choose the kind of results that are printed as individual contributions (with contributionsfrom individual sources and harmonics). When only ONOISE plots are specified then thedefault value is ONOISE. When SPHI or AMNOISE plots are requested then thedefault is PHNOISE or AMNOISE. The NAME flag is used to sort the contributionsfrom devices by device name.

• SSTNOISE_INCLUDE_DEVICES=device_type

Defines the contributions from device families that will be printed. Possible values fordevice_type are: MOS, BJT, DIOD, RES, JFET, GEN and Y. For example:

.OPTION SSTNOISE_INCLUDE_DEVICES=MOS

.OPTION SSTNOISE_INCLUDE_DEVICES=BJT

This will only print the contributions from the following device families MOS and BJT. Ifno value for device_type is specified contributions of all device families are printed.

• SSTNOISE_EXCLUDE_DEVICES=device_type

Excludes the contributions from device families that will be printed. Possible values fordevice_type are: ALL, MOS, BJT, DIOD, RES, JFET, GEN and Y. For example:

.OPTION SSTNOISE_EXCLUDE_DEVICES=RES

.OPTION SSTNOISE_EXCLUDE_DEVICES=GEN

This will print the contributions from all the device families except Resistors and Generators(individual I and V sources). Specifying keyword ALL means no individual devicecontribution will be printed.

• SSTNOISE_GLOBPART=0|1

A different algorithm is automatically used to solve the Steady-State Phase Noise of PLLcircuits with a specific setup. This is based on circuit partitioning. It can be deactivated byspecifying option SSTNOISE_GLOBPART=1. Default is 0.



Multi-Threaded Simulation OptionsSST analysis and SST based analyses can take benefit from multi-processor machines. They cansplit their internal computations on the different available processors. By default,multi-threading is activated. When activated, Eldo RF automatically identifies the number ofprocessors and tries to use the maximum of available resources. With a two processor machine,the elapsed time gain is about 25% compared to the use of a single processor. This gain canreach 45% using four processors. The following message will be displayed whenmulti-threading is used for a simulation:

Launching Multithreading: 4 processors potentially used

The following two options can be used in conjunction with this capability:

• SST_MTHREAD=0|1

Multi-threaded simulations can be activated by setting this option to 1. Setting it to 0 willdeactivate multi-threaded simulations. Default is 1.

NoteMulti-threaded simulations can be deactivated by specifying the Eldo flag-no_sst_mthread.

• SST_NBTHREAD=val

This option is used to limit the number of processors used in a multi-processor simulation.For example: if Eldo RF is run on a four processor machine, and SST_NBTHREAD is set to 2,then only two processors will be used.

MODSST Analysis OptionsMODSST analysis generates time domain and frequency-time domain representations ofsignals. At each timepoint calculated by MODSST analysis, an instantaneous spectrum isobtained. Furthermore, between two timepoints, it is possible to reconstitute the time-domainwaveform by using interpolation techniques based on Fourier components. Some parametersallow you to specify the way time domain values are reconstructed:

• MODSST_FULL_DISPLAY=0|1|2

Specifies the duration of plotting between two timepoints.

“0” means that the waveform is only plotted at the timepoint calculated by MODSSTanalysis.

“1” means that between two timepoints, the duration of plotting is based on theSST_NPER option.

“2” means that the reconstitution time-domain waveform is performed for the entireduration of simulation.



If only the TMODSST plot type is specified, then MODSST_FULL_DISPLAY is equal to 1.When running in -compat mode, MODSST_FULL_DISPLAY is forced to 1 if you set theoption to 2 in the netlist. Otherwise, it is equal to 0.

• MODSST_FULL_DISPLAY_FORCED=0|1

When running in -compat mode, MODSST_FULL_DISPLAY can be forced to 2 ifMODSST_FULL_DISPLAY_FORCED is equal to 1.

CautionForcing MODSST_FULL_DISPLAY=2 in -compat mode can produce a very large database(.wdb/.cou files).

• MODSST_CENTRAL_FUND_OSCxx=val

Specify the value of the reference frequency used to represent the results from FFTcomputations (i.e. results from the use of .OPTFOUR). In the case of modulated oscillatorsimulation, the oscillation frequency is a time-dependent variable. By default the referencefrequency value is:

o FUND_OSC_GUESS in the case of standard simulation.

o FUND_OSCxx calculated by the previous SST analysis in the case of SSTUIC.

o FUND_OSCxx specified in the restart file in the case of .RESTART SST use.

NoteThe MODSST_CENTRAL_FUND_OSC option will not affect simulation results, it will onlyaffect FOURMODSST calculations.

• MODSST_USE_AVERAGE_FUND_OSC

Provides a centered and unaliased FOURMODSST spectrum. The reference frequency forFFT computations of oscillation frequency during MODSST analysis can be automaticallycomputed as an average value of the instantaneous oscillation frequency in the FFT timewindow with this option.

• MODSST_HMAX=val

Specifies the maximal allowed time step of the MODSST analysis. If val is omitted thevalue of the option HMAX (see HMAX of the Eldo User’s Manual) will be used.

• MODSST_HMIN=val

Specifies the minimal allowed time step of the MODSST analysis. If val is omitted thevalue of the option HMIN (see HMIN of the Eldo User’s Manual) will be used.

• RF_PARTITIONING_MODE=STANDARD|FAST

This option allows true RF-analog separation. It removes any RF signal that could betransmitted to the analog part. This avoids the analog time step being controlled by residualsignals.



NoteThis mode is an approximation used to speed up simulation and is only valid if thepartitioning is setup correctly by the user.

STANDARD

Default value meaning full transmission of potential RF signal to the analog part.

FAST

Any RF signal smaller than the value defined by the RF_PARTITIONING_THRESHOLDoption on the RF/analog boundary is cancelled.

• RF_PARTITIONING_THRESHOLD

Defines the threshold value of RF signals cancelled on the RF/analog boundary. Defaultvalue is , therefore all RF signals are cancelled.

NoteAs this may impact final simulation results a warning is issued whenever a RF signal iscancelled on boundary nodes.

NoteOptions RF_PARTITIONING_MODE and RF_PARTITIONING_THRESHOLD are used whenimplementing circuit partitioning (.PART), see “.PART MODSST” on page 64 for moreinformation.

A single-tone MODSST analysis can be performed on a Spice netlist. An FFT analysis can beperformed on the single-tone MODSST analysis results during the same simulation as theMODSST analysis. The FFT analysis will obtain the two-tone SST data, this data is saved to afile with the .SAVE SST command. The saved SST data can be used to initialize a two-toneSST analysis using the .RESTART SST command. When the FFT analysis is performed, youmust specify the fundamental frequency, number of harmonics and the starting point that theFFT analysis will be performed on the FFT analysis, this is specified through the optionsMODSST_FFT_FUND_FREQ, MODSST_FFT_NHARM and MODSST_FFT_TSTART respectively. All theoptions must be specified in the same netlist with the .SAVE SST command. The FFT isinvoked using the following three options consecutively:

NoteThe options MODSST_FFT_FUND_FREQ, MODSST_FFT_NHARM and MODSST_FFT_TSTARTmust be specified in the same netlist with the .SAVE SST command. If an option isomitted the following error message will be generated:ERROR 1043: All three MODSST_FFT options must be specified. The option

MODSST_FFT_NHARM is missing:SIMULATION STOPPED!

If the .SAVE SST command is omitted the following error message will be generated:ERROR 1044: .SAVE SST must be specified with the MODSST_FFT

options:SIMULATION STOPPED!

∞



• MODSST_FFT_FUND_FREQ=val

Computes the FFT time window. Also defines the fundamental frequency for the secondtone of the saved result.

• MODSST_FFT_NHARM=val

Computes the number of sampling points. Also defines the number of harmonics for thesecond tone of the saved results.

• MODSST_FFT_TSTART=val

Specifies the starting point for the MODSST analysis when the sampling is started toprovide the data on which the FFT analysis will be performed.

Example

The following will specify an FFT analysis on the MODSST results. The extracted two-toneSST data will have two fundamental frequencies, the first is FUND_OSC_GUESS1 with 5harmonics and the second is 100MHz (specified on the option MODSST_FFT_FUND_FREQ) and 20harmonics (specified on the option modsst_fft_nharm). The FFT analysis will start at 950nson the MODSST results. The two-tone SST data will be saved to the file test_2tone.sst.

.SST OSCIL FUND_OSC_GUESS1=2.5G NHARM_OSC1=5

.MODSST 0 1u

.OPTION modsst_fft_nharm=20+ modsst_fft_fund_freq=100meg+ modsst_fft_tstart=950n

.SAVE test_2tone.sst sst

Miscellaneous• AUTOSTOP=0|1|2

This option can only be used with time domain analyses (TMODSST and TSST) simulations.This option causes Eldo RF to stop the simulation when either all of the extracted waveforminformation (.EXTRACT command) has been measured or when all of the sweepmeasurements are complete.

AUTOSTOP=0

Deactivates autostop if necessary. Default.

AUTOSTOP=1

Causes Eldo RF to stop the simulation when all extracted waveform information(.EXTRACT) has been measured.

AUTOSTOP=2

Used in multi-step simulations. Causes Eldo RF to stop when all sweepmeasurements are complete.



• FOUR_SOURCE_DELAY=val

This option can be used to deactivate modulated and periodic FOUR sources during thepre-transient phase, either fully or partially.

• NO_SST =0|1

(0 or 1). Bypass the Steady-State analysis when a Steady-State Noise or a Steady-State ACis expected. The use of this option requires a .RESTART SST in order to initialize theproblem. This option may be useful to run a Steady-State Noise analysis with moreharmonics than for the Steady-State analysis for example.

• SST_CIRCUIT_TYPE=circuit_type

Use this option to define the type of circuit to be simulated. Choose the circuit_type from alist of predefined types. Each circuit_type causes Eldo RF to automatically select theappropriate set of options for the circuit as follows:

Table 5-13. SST_CIRCUIT_TYPE Descriptions

circuit_type Keywords Options automatically selected

DIVIDER_xx (where xx isthe rank1 of the divider)

means Frequency divider by xx

SST_CONVERGENCE_HELP=TransientSST_MAX_LINITER=100 (this value can be overridden ifspecified by the user)SST_USE_NTONE_PROCEDURE=1 (this option is used formulti-tone analysis and well-suited if the frequencydivider is along FUND1)SST_PRECONDITION=Time

LNA_SS means single-tone, small non-linear conditions

Default options

LNA_LS means more severe non-linear conditions

SST_MAX_LINITER=100 (this value can be overridden ifspecified by the user)

PA SST_MAX_LINITER=100 (this value can be overridden ifspecified by the user)

MIXER_SS means N-tones, small non-linear conditions

SST_NDIM_FFT=1



• SST_KEEP_OPTIONS_FOR_SWEEP

Forces each simulation run of a sweep analysis to use the same values/options. When not seteach simulation run will use the solution computed in the previous simulation run, and in thecase of divergence the ith run starts with the set of options defined in the netlist.

• SST_MEMESTIM

This option activates the estimation of the RAM (in kBytes) required for the simulation.This estimation is done prior to the simulation. It provides two estimations: the first one isan average value and the second one is a maximum value. The exact value depends on thenumber of iterations in the linear solver iteration and this can not be predicted. Allowedvalues are 0 or 1. Default value is 0. Useful for very large circuits. This option is alsosupported with all sweeps of a sweep analysis

• SST_MEMORY_COMPRESS=LEV1

This option is used to compress the memory used in RAM by Eldo RF. When this option isspecified, a compression rate of approximately 50% is achieved, and will not effect the

MIXER_LS this set of options is well-suited if the large signal is alongFUND1

SST_CONVERGENCE_HELP=TransientSST_SPECTRUM=1SST_NDIM_FFT=1SST_USE_NTONE_PROCEDURE=1

BUFFER means single-tone strongly non-linear

SST_CONVERGENCE_HELP=TransientSST_CONVERGENCE_HELP=Advanced_NewtonSST_PRECONDITION=TimeSST_MAX_LINITER=100 (this value can be overridden ifspecified by the user)SST_USE_NTONE_PROCEDURE=1 (this option is used formulti-tone analysis and well-suited if the buffer is alongFUND1)

VCO means standard oscillator circuits

Default options (SST_OSC_PHASE_SEQUENCE=Seq_1)OSC

QUARTZ means strongly non-linear oscillators

SST_OSC_PHASE_SEQUENCE=Seq_4 (this setup presumesthe oscillator is kicked due to initial conditions)

RING

RELAX

1. The rank must be in agreement with the source frequency specification.

Table 5-13. SST_CIRCUIT_TYPE Descriptions

circuit_type Keywords Options automatically selected



accuracy of the results. Using this option may slow the simulation time. This option isparticularly useful when simulating a large design and/or a large number of linear iterations.

• SSTNLCONTRIB_FILE[=filename]

Prints the .SSTNLCONTRIB analysis results into a separate file, by default they are printedto the .chi file. When the filename is omitted from the option, the results are printed to afile with the same name as the netlist file with the extension .sstnlcontrib.

• SSTSENSRLC_FILE[=filename]

Prints the .SSTSENSRLC analysis results into a separate file, by default they are printed tothe .chi file. When the filename is omitted from the option, the results are printed to a filewith the same name as the netlist file with the extension .sstsensrlc.

• SST_UIC=0|1

This option allows you to use Steady-State analysis results as an initial condition for atransient analysis (when a .SST and a .TRAN analysis are specified in the same netlist).Allowed values are 0 or 1. When omitted, the default value is 0, when specified the defaultvalue is 1.

Note: If this option is set and a .SST and a .AC analysis are specified in the same netlistthen the Steady-State analysis results are used as an initial condition for the AC analysis (noDC analysis is performed).

• SST_VERBOSE

This option is used to control the iterative linear solver and may sometimes help the steady-state convergence. This option enables the display of information relative to steady-statestatus during the resolution (tolerance at each iteration, number of iterative solveriterations,...). Allowed values are 0 or 1. Default value is 0.


Chapter 6Additional Syntax

Special DevicesThe devices DCCUT and DCFEED correspond to capacitance and inductance respectively withan infinite value.

DC CutCxx n1 n2 DCCUT val

For .AC, .SST, .SSTAC, .SSTNOISE and .SSTXF analyses the DCCUT device corresponds toan open circuit in DC and a short circuit for all other frequencies.

For the other analyses (DC, TRAN, MODSST…) it is assumed to be a normal capacitance ofval Farads.

For more information on Capacitor syntax see Capacitor of the Eldo User’s Manual.

DC FeedLyy n3 n4 DCFEED val

For .AC, .SST, .SSTAC, .SSTNOISE and .SSTXF analyses the DCFEED device corresponds toa short circuit in DC and an open circuit for all other frequencies.

For the other analyses (DC, TRAN, MODSST…) it is assumed to be a normal inductance ofval Henrys.

For more information on Inductor syntax see Inductor of the Eldo User’s Manual.

RF to Digital and Digital to RF ConvertersIt is possible to connect a digital signal directly to a component of an RF signal and vice-versa.The component of an RF signal can be the real part, the imaginary part, the magnitude or thephase of a specified harmonic.


Additional SyntaxRF to Digital and Digital to RF Converters

Syntax

.A2D [SIM=simulator] eldo_node_name [digital_node_name]+ [MOD=MODEL_NAME] [RF_PARAM_LIST] [RF_PARAM_LIST].D2A [SIM=simulator] eldo_node_name [digital_node_name]+ [MOD=MODEL_NAME] [RF_PARAM_LIST] [RF_PARAM_LIST]

Parameters

• SIM=simulator

The parameter SIM can take the value of the digital simulator’s name, for exampleVERILOG.

• eldo_node_name

Name of the node in the analog netlist.

• digital_node_name

Name of the node in the digital description. This parameter is optional.

• MOD=model_name

Name of the model used for the convertor. If model_name is specified, there must be acorresponding .MODEL command:

.MODEL model_name A2D|ATOD parameters_list

For more information on the above parameters see .A2D and .D2A of Eldo User’sManual.

• RF_PARAM_LIST

List of available parameters as described below.

harm=(i[,j[,k]])

Parameter specifying the harmonic of the RF signal connected to the digital part.

active=MAG|PH|R|I|FREQ

Parameter specifying which component of the signal is connected to digital. Allowedkeywords are:

MAG = magnitudePH = phaseR = real partI = imaginary partFREQ = frequency

rfref=val

Parameter only required for D2A converter. It defines the complementary part of theactive parameter needed to fully define the RF signal component, that is generatedby the digital part. It defines the phase when active=PH, it defines magnitude whenactive=MAG, real part when active=R and imaginary part when active=I.

Additional SyntaxRF to Analog and Analog to RF Converters


Examples

.A2D A2D_BIT Y2.1 MOD=A2D_A_BIT harm=(1) active=R

.A2D A2D_BIT Y3.1 MOD=A2D_A_BIT harm=(1) active=I

This defines an ideal demodulator, connecting the real part and the imaginary of the firstharmonic to digital.

.D2A D2A_BIT MOD=D2A_B_BIT harm=(1) active=MAG rfref=-90

In the above example the digital world controls the magnitude of the first harmonic with a phaseequal to -90 degrees.

RF to Analog and Analog to RF ConvertersThese converters are used in the form of Y instance macromodels. This macromodel is a voltagesource connected between nodes outp and outn and its value is controlled by the definedparameters and the voltage between nodes inp and inn.

This macromodel can be used to define an RF to RF converter, an Analog to Analog converter,an RF to Analog converter as well as an Analog to RF converter depending on the specifiedvalues of input_hxx and output_hyy.

Syntax

Yxx RF_ANALOG_CONVERTER pin: inp inn outp outn+ param: [input_h_fund1=val [input_h_fund2=val [input_h_fund3=val]]]+ [input_format=val] [gain=val]+ [output_h_fund1=val [output_h_fund2=val [output_h_fund3=val]]]+ [output_format=val] [refval=val]

Parameters

• input_h_fund1, input_h_fund2, input_h_fund3

Define the harmonic (or the intermodulation product) of the input voltage v(inp, inn) used tocontrol the output voltage source. Default values are 1, 0, 0 respectively.

• input_format

Specifies the portion of the input signal that controls the output voltage source. Possiblevalues are 1, 2, 3, 4, 5 and 6 corresponding to magnitude, phase, real part, imaginary part,complex value and instantaneous frequency respectively. Default value is 5.

• gain

Coefficient that multiplies the input value to provide the output value. Default value is 1.0.

• output_h_fund1, output_h_fund2, output_h_fund3

Defines the harmonic (or the intermodulation product) of the output voltage between nodesoutp and outn that is controlled by the input. Default values are 0, 0, 0 respectively.


Additional SyntaxRF Envelope Detector

• output_format

Specifies the portion of the output signal that is controlled the input signal. Possible valuesare 1, 2, 3, 4 and 5 corresponding to magnitude, phase, real part, imaginary part andcomplex value respectively. Default value is 5.

• refval

Specifies the part of the output signal that is not controlled by the input and that is missingto generate a complex value. When output_format is 1, refval defines the phase of the outputsignal; when output_format is 2, refval defines the magnitude; when output_format is 3refval defines the imaginary part; and when output_format is 4, refval defines the real part.Default value is 1.0 when it corresponds to a magnitude, otherwise it is 0.0.

Example

YI_demod rf_analog_converter pin: rfin 0 Iout 0+ param: input_h1=1 input_format=3+ output_h1=0 output_format=3YQ_demod rf_analog_converter pin: rfin 0 Qout 0+ param: input_h1=1 input_format=4+ output_h1=0 output_format=3

YI_demod and YQ_demod correspond to I and Q demodulators.

RF Envelope DetectorThe RF envelope detector model acts as an ideal envelope detector for simulation in .MODSSTanalyses. The input signal (i.e. the voltage across inp and inn) is a full spectrum RF signal. Theoutput signal is the corresponding envelope (of the input signal) applied to the baseband (DCharmonic) coefficient of the spectrum. The output is multiplied by the gain and is delayed asdefined by the parameters Gain and Delay respectively.

Syntax

Yxx RF_ENVELOPE_DETECTOR pin: inp inn outp outn+ param: [Gain=val] [Delay=val]

Parameters

• Gain

Multiplication coefficient of the output signal i.e. the gain of the model. Default is 1.0.

• Delay

Coefficient by which the output signal is delayed (seconds). Default is 0.0.

Example

Yenv RF_Envelope_detector pin: rfin 0 rfout 0+ param: Gain = 2

Defines an RF envelope detector with a gain of two. The delay between the input and output is0.

Additional SyntaxFlexible Frequency Divider Macromodel


Flexible Frequency Divider MacromodelA flexible frequency divider model is available under the form of a parameterizable Ymacromodel. This model supports Eldo RF SST, SST PLL, SSTNOISE, SSTAC, SSTXF andMODSST analyses, but does not support Eldo AC and TRAN analyses.

This macromodel is a voltage source connected between nodes outp and outn and its value iscontrolled by the defined parameters and the voltage between nodes inp and inn. The controlpins ctrlp and ctrln can be used to control the division factor of the model. When controlpins are defined, the division factor is calculated using the following equation:

Division Factor = DIV_FACTOR + (v(ctrlp) - v(ctrln)) * Div_gain

By default, the output of the frequency divider model is a sine wave. A square wave can bedefined by specifying a positive value on the TTRANS parameter.

Syntax

Yxx RF_FREQUENCY_DIVIDER pin: inp inn outp outn [ctrlp ctrln]+ param: [Input_h_fund1=val] [Input_h_fund2=val] [Input_h_fund3=val]+ [Output_h_fund1=val] [Output_h_fund2=val] [Output_h_fund3=val]+ [Vout=val | Gain=val]+ [Div_factor=val]+ [Delay=val]+ [Out_phase=val]+ [DC_Offset=val]+ [TTRANS=val]+ [DIV_gain=val]+ [DUTY_CYCLE=val]

Parameters

• Input_h_fundx

Defines the harmonic (or the intermodulation product) of the input voltage v(inp, inn) that isused to control the output voltage source. Default value is 2, 0, 0.

• Output_h_fundx

Defines the harmonic (or the intermodulation product) output of the frequency divider. Thedefault value is automatically computed from the input frequency and the parameterDiv_factor.

• Vout

Parameter defining the output source voltage. Note that Vout and Gain are exclusiveparameters and obey the following rules:

o If neither Vout nor Gain is specified, Vout = 1.0

o If only Vout is specified, that value is used.

o If only Gain is specified, that value is used.

o If both Gain and Vout are specified, the Gain value is used.


Additional SyntaxFlexible Frequency Divider Macromodel

• Gain

Coefficient that multiplies the input value to provide the output value. Note that Vout andGain are exclusive parameters, see the Vout description.

• Div_factor

Defines the rank of division. It can be a non integer value, but it will be truncated to aninteger for an SST analysis. Default is 2.

• Delay

Specifies the delay between the input signal and the output signal. Default is 0.0.

• Out_phase

Specifies the phase shift (in degrees) of the output signal with respect to the input signal.Default is 0.0.

• DC_Offset

Specifies a DC offset at the output of the divider.

• TTRANS

Specifies the transition time (in seconds) of the output square wave. If the value specified isnot positive then the output will be a sine wave. Default is 0.0. This parameter can only beused during a MODSST analysis if the harmonic of the output signal is 0.

• DIV_gain

Specifies the gain of the control pins (ctrlp and ctrln). Default is 1.

• DUTY_CYCLE

Specifies the duty cycle of the output waveform. Default is 0.5. This parameter only has aneffect during MODSST analysis, when a squared output is specified (parameter TTRANSdefined). The value specified for this parameter must be 0.0 < DUTY_CYCLE < 1.0.

Examples

Ydiv_by_4 rf_frequency_divider pin: rfin 0 out 0+ param: input_h1=4 div_factor=4

This defines a frequency divider by 4 model. Input signal is on harmonic 4 and output signalwill be on the fundamental frequency (harmonic 1). We suppose that this model will be used inSST analysis.

Ydiv_by_100 rf_frequency_divider pin: rfin 0 out 0+ param: input_h1=1 div_factor=49.9

This defines a frequency divider by 49.9. Input signal is on harmonic 1 (fundamental frequency)and output signal will be on harmonic 0 (DC component). We suppose that this model will beused for a MODSST analysis.

Ydiv_by_100_1 rf_frequency_divider pin: rfin 0 Vout 0+ param: input_h1=1 input_h2=0 div_factor=100.1 output_h1=0

Additional SyntaxSave and Restart Capabilities


+ output_h2=1

This defines a frequency divider by 100.1. Input signal is on harmonic (1, 0), the output signalwill be on harmonic (0, 1).

Ydiv_by_100_IP rf_frequency_divider pin: rfin 0 Vout_IP 0+ param: input_h1=1 div_factor=100 out_phase=0.0Ydiv_by_100_IN rf_frequency_divider pin: rfin 0 Vout_IN 0+ param: input_h1=1 div_factor=100 out_phase=180.0Ydiv_by_100_QP rf_frequency_divider pin: rfin 0 Vout_QP 0+ param: input_h1=1 div_factor=100 out_phase=90.0Ydiv_by_100_QN rf_frequency_divider pin: rfin 0 Vout_QN 0+ param: input_h1=1 div_factor=100 out_phase=-90.0

This defines four frequency dividers that generate four shifted outputs (Vout_IP, Vout_IN,Vout_QP and Vout_QN) of a signal that is divided by 100.

Vvar var 0 four 1k ma (1) 1 -90Ydiv_by_100pm1 rf_frequency_divider pin: rfin 0 Vout1 0 var 0+ param: input_h1=1 div_factor=100 ttrans=1n

Defines a frequency divider varying between 99 and 101. The frequency division is varyingbecause the control pin voltage is varying between -1V and 1V. The output is a squarewaveform with a 1ns rise and fall time.

Save and Restart CapabilitiesThe Eldo .SAVE and .RESTART commands can handle Pre-Transient Phase, Steady-State,Steady-State Noise and Modulated Steady-State analysis results.

For further information, refer to the .RESTART command and the .SAVE commandof the Eldo User’s Manual.

Pre-Transient Phase.SAVE [FNAME] PRETRAN.RESTART [FNAME] PRETRAN

.SAVE can be used to save transient data at the end of the pre-transient phase. The data is storedin a file of name FNAME. If FNAME is not specified then a file named netlist_name.sst is created.The data in this saved file can be used to restart the pre-transient phase using the .RESTARTPRETRAN command, or to initialise a steady-state analysis using the .RESTART SSTcommand.

.RESTART PRETRAN can be used to restart the pre-transient phase from saved data. IfFNAME is not specified then a file named netlist_name.sst is assumed.


Additional SyntaxSave and Restart Capabilities

NoteIf you try to restart the pre-transient phase and the steady state analysis in the samesequence, for example:.restart data1.sst pretran

.restart data2.sst sst

then only the steady-state analysis will be restarted.

Steady-State Analysis.SAVE [FNAME] SST [TEMP=VAL] [STEP=VAL].RESTART [FNAME] SST

.SAVE can be used to save .SST simulation results in the FNAME file or to use the results in theFNAME file as an initial estimation for the .SST analysis.

The number of harmonics or the number of fundamental frequencies do not necessarily need tobe the same between the FNAME file and the .SST. For instance, single-tone SST simulationresults can be saved and used as an initial estimate for a two-tone SST analysis.

The saved results can be reused to initialize a .SST analysis.

Steady-State Noise Analysis.SAVE [FNAME] SSTNOISE [TEMP=VAL] [STEP=VAL].RESTART [FNAME] SSTNOISE

.SAVE can be used to save .SSTNOISE simulation results in the FNAME file or to use the resultsin the FNAME file as an initial estimation for the .SSTNOISE analysis.

This feature is useful when you want to avoid recomputing already calculated .SSTNOISEresults. Then for instance you wish to add plots, extend the noise frequency range or increasethe frequency resolution. The simulator will automatically detect which results can be restoredand reuse them.

NoteThis is only possible if the circuit is the same (same nodes and devices), the .SSTconfiguration is the same (fundamental frequencies, same number of harmonics and thesame input power) and the .SSTNOISE analysis has some similarities (same output nodesaround the same harmonic).

When FNAME is not supplied the simulator will use the file with the netlist name and the.sstnoise extension. Both .SAVE and .RESTART commands are allowed simultaneously in thesame netlist, then if a conflict occurs with FNAME the simulator will add the extension “_2” tothe .SAVE results.

Additional SyntaxAutomated Sweeps


Modulated Steady-State Analysis.SAVE [FNAME] SST [END|TIME=VAL [REPEAT] [ALT|SEQ]] [TEMP=VAL] [STEP=VAL].RESTART [FNAME] SST

.SAVE can be used to save .MODSST simulation results in the FNAME file or to use the results inthe FNAME file as an initial estimation for the .MODSST analysis.

The number of harmonics or the number of fundamental frequencies do not necessarily need tobe the same between the FNAME file and the .MODSST command. For instance, single-tone SSTsimulation results can be saved and used as an initial estimate for a two-tone SST analysis.

The saved results can then be reused to initialize a .MODDST analysis.

Automated Sweeps.STEP param parname valmin valmax incr [(AUTOINCR)]

A subset of the Eldo .STEP command has been extended to handle parameter sweeps withadaptive increments.

The sweep will start with the increment value incr and, with (AUTOINCR) specified, duringthe sweep the value of the increment will be automatically adapted depending on theconvergence of each sweep point. If convergence is easy, then the step is increased. Ifconvergence is difficult, then the step is decreased.


Additional SyntaxAutomated Sweeps


Chapter 7Convergence Troubleshooting

IntroductionThis chapter will help you solve some convergence problems that you may experience withEldo RF steady-state analyses.

First it will help you to identify the type of divergence problem that has occurred. The types ofdivergence problems that may occur are:

Type_1A Early divergence during the Newton iterations

Type_1B End of convergence difficult to achieve during Newton iterations

Type_2 Iterative linear solver failure

After identifying the type of divergence problem, this chapter will then propose differentactions, options or settings, that could overcome the problem. We will differentiate between thedifferent types of circuits where convergence problems may occur.

NoteAll of the options described in this chapter can be specified with the .OPTION command,see “Command Options” on page 153.

NoteIt is possible to define the type of circuit to be simulated. Choose the circuit type from alist of predefined types and specify it through an option. Each circuit_type causesEldo RF to automatically select the appropriate set of options for the circuit. This may bea first step in resolving some convergence problems before proceeding further in thischapter. See “SST_CIRCUIT_TYPE=circuit_type” on page 172.

Divergence TypesThere are three types of divergence, as follows:

Type_1AEarly divergence during the Newton iterations. This type of divergence can be detected bymonitoring the Newton Residual (through the use of the SST_VERBOSE option). This type of


Convergence TroubleshootingCircuit Types

divergence occurs when the Newton Residual dramatically increases at the beginning of thesimulation. The reason for the divergence is that the starting point is too far from the solution.

Type_1BEnd of convergence difficult to achieve during Newton iterations. This can be detected bymonitoring the Newton Residual (with the SST_VERBOSE option). This type of divergenceoccurs when the Newton Residual decreases at the beginning of the solution and then stagnates,the simulator is unable to reach the required tolerance. There may be different reasons for thisbehavior, for example:

• The number of harmonics is too small for an accurate representation of the solution.

• The problem is ill conditioned, due to small value resistors

• There are problems within the model.

Type_2Iterative linear solver failure. This may occur with very non-linear circuits, and when it occurs amessage will be issued, shown below.

Divergence due to Linear Iterative Solver failureTry to increase the SST_MAX_LINITER optional value

Circuit TypesThe type of circuits where the convergence problems can occur are:

• Forced circuits

o Single-tone

o Multi-tone

• Autonomous circuits

o High Q oscillators

o Very non-linear oscillators

The following sections in this chapter propose different actions, options or settings, that couldovercome the identified convergence problem.

Convergence TroubleshootingTroubleshooting Forced Circuits


Troubleshooting Forced CircuitsThis section describes how to overcome convergence problems with single-tone and multi-tonecircuits.

Single-Tone CircuitsThe convergence problem can occur in single-tone circuits that are:

• Circuits that do not Contain a Frequency Divider

• Circuits that Contain a Frequency Divider

Circuits that do not Contain a Frequency DividerFor this type of circuit, when convergence problems occur in Eldo RF, it will automaticallyperform a sweep (reduction) of the input power, and successively solves steady-states for thedifferent values of input power until it reaches the nominal value.

If the automatic sweep (on the input power) fails to converge or is too long:

• You can help the automatic sweep with the following option:

SST_RAMPING_FACTOR

or perform a manual customized sweep (more adapted to your case), with the use of.save and .restart.

Type_1A Convergence Solution

• Invoke a transient assistance with the following option:

SST_CONVERGENCE_HELP=TRANSIENT

• Use the corresponding options, for example:

eps, reltol, vntol, hmax, sst_tran_nper etc.

• Change the Newton algorithm with the following option:

SST_CONVERGENCE_HELP=ADVANCED_NEWTON

Type_1B Convergence Solution

• Increase the number of harmonics.

Type_2 Convergence Solution

• Increase the maximum number of iterations for the linear iterative solver, with thefollowing option:



SST_MAX_LINITER=[VAL]

• Change the preconditioner with the following option:

SST_PRECONDITION=TIME

or its more accurate variant, TIME_ACCURATE.

• Possibly oversample, as it may help this preconditioner, with the following option:

SST_OVRSMP=[VAL]

Circuits that Contain a Frequency DividerThe frequencies generated by the circuit have to be within the spectrum. This means that for adivide by N circuit, the input signal has to be specified on the harmonic N. Then the outputsignal will be on the fundamental frequency.

For this type of circuit, Eldo RF performs a transient analysis of over 10 periods of thefundamental frequency. Then it will initialize the steady-state (SST) to solve by using thetransient results of the last period.


• Tighten the transient analysis using following options:

eps, reltol, vntol, hmax etc.

• Increase the transient length with the following option:

.OPTION SST_TRAN_NPER=[VAL]


SST_CONVERGENCE_HELP=ADVANCED


• Increase the number of harmonics. As a rule of thumb the number of harmonics shouldbe at least 10 times the ratio of frequency division.


• Increase the maximum number of iterations for the linear iterative solver using thefollowing option:

SST_MAX_LINITER=[VAL]

• Change the preconditioner with the following option:





• Possibly oversample, as it may help this preconditioner, with the following option:

SST_OVRSMP=[VAL]

The recommended configuration of the options for frequency dividers is shown below:

.OPTION SST_PRECONDITION=TIME

.OPTION SST_CONVERGENCE_HELP=ADVANCED_NEWTON

Multi-Tone CircuitsPut the most difficult or the most non-linear tone along the first fundamental frequency(FUND1).

When Eldo RF has convergence problems on multi-tone circuits, it will behave in a similar wayas for single-tone circuits. It will automatically perform a sweep (reduction) of the input power,and successively solves steady-states for the different values of input power, until the nominalvalue is reached. However, for multi-tone circuits, this sweep is simultaneously applied on allinput frequencies.

If the automatic sweep (on the input powers) fails to converge or is too long:

• Perform a manual customized sweep (more specific to your case), with the use of .save

and .restart. For example, perform a single-tone simulation and save the results.Then restart the two-tone analysis from the single-tone solution and so on (by increasingthe number of tones each time). This can also be done automatically with the followingoption:

SST_USE_NTONE_PROCEDURE



SST_CONVERGENCE_HELP=ADVANCED_NEWTON


• Increase the number of harmonics.


• Increase the maximum number of iterations for the linear iterative solver with thefollowing option:

.OPTION SST_MAX_LINITER=[VAL]


Convergence TroubleshootingTroubleshooting Autonomous Circuits

Troubleshooting Autonomous CircuitsThis section describes how to overcome convergence problems with high Q oscillator and verynon-linear oscillator circuits.

Oscillators differ from forced circuits in the following way:

• They have no input power.

• The fundamental frequency is not known beforehand (with accuracy).

Oscillators also have another distinction; they have two steady-state solutions:

• The trivial one, corresponding to the DC operating point (that is easy to find but usuallynot requested).

• The correct one, corresponding to the large signal oscillations (which is much moredifficult to find).

High Q OscillatorsProviding a very accurate estimation of the oscillation frequency is mandatory for the success ofthe steady-state convergence. That is why we recommend that the simulator is given the task ofestimating the oscillation frequency (through the use of the stability analysis).

NoteThe position of the Probe is also very important. Many convergence problems occur dueto an incorrect Probe position.

• When the standard configuration fails to converge, we recommend that you use thecontinuation algorithm, which can be specified with the following option:

SST_CONVERGENCE_HELP=CONTINUATION

• If the continuation algorithm fails then try the PSEUDO_MODSST option:

SST_CONVERGENCE_HELP=PSEUDO_MODSST

Very Non-Linear OscillatorsProviding a very accurate estimation of the oscillation frequency is not as mandatory as it is fora high Q oscillator. It is recommended that you give the simulator the task of estimating theoscillation frequency.

NoteThe position of the Probe is also very important.



• When the standard configuration fails to converge, we recommend you use thecontinuation algorithm, which can be specified with the following option:

SST_CONVERGENCE_HELP=CONTINUATION

• If the continuation algorithm fails then try a combination of the following:

SST_CONVERGENCE_HELP=TRANSIENTSST_CONVERGENCE_HELP=ADVANCED_NEWTON

• If the circuit contains a frequency divider, try using the following option:



• If this fails, then try the PSEUDO_MODSST option:

SST_CONVERGENCE_HELP=PSEUDO_MODSST

• If this fails, then try the following option:

SST_OSC_PHASE_SEQUENCE=

See “SST_OSC_PHASE_SEQUENCE=SEQ_1|SEQ_2|SEQ_3|SEQ_4|SEQ_5” onpage 165 and “SST_OSC_PHASE_SEQUENCE” on page 36.


Chapter 8Working with S, Y, Z Parameters

IntroductionA set of commands in Eldo RF allows you to extract the large signal S parameters (Scatteringparameters), the Y parameters (Admittance) or the Z parameters (Impedance) in the frequencydomain for a specified circuit. The circuit can have any number of ports.

For a full example in Eldo RF, please refer to the tutorial on S parameter extraction:Tutorial #4—S-Parameter Extraction for an Amplifier.

Eldo RF enables the steady-state simulation of circuits including any number of N-port blocksdescribed by a frequency tabulation of their small-signal S (Scattering), Y (Admittance) orZ (Impedance) parameters. It does so by reading the S, Y, Z, G, H, T or A parameter data froma Touchstone® format file.

For more information on usage in the time domain, please refer to the Eldo User’sManual.

Behavioral models for describing any 2-port or 3-port network defined by its S-parameters canbe found in S-Parameters Blocks of the CommLib RF Library.

Simulation Setup for S, Y, Z ParameterExtraction

Special sources must be added at each port of the circuit to be analyzed. The number of the portand the reference impedance for S parameters must be specified in the Eldo RF control file asfollows:

Source SyntaxVyy NP NN IPORT=VAL [RPORT=VAL] [CPORT=VAL] [LPORT=VAL] [MODE=KEYWORD]Vyy NP NN IPORT=VAL ZPORT_FILE=string [CPORT=VAL] [LPORT=VAL]+ [MODE=KEYWORD]Iyy NP NN IPORT=VAL [RPORT=VAL] [CPORT=VAL] [LPORT=VAL] [MODE=KEYWORD]Iyy NP NN IPORT=VAL ZPORT_FILE=string [CPORT=VAL] [LPORT=VAL]+ [MODE=KEYWORD]


Working with S, Y, Z ParametersSimulation Setup for S, Y, Z Parameter Extraction

Parameters• yy

Name of the port.

• NP


• NN


• IPORT

This is a strictly positive number that is unique and is used as the port number: this numberis used for naming the outputs (for instance, .EXTRACT FSST yval(Sdb(1,2),fund1)).An error message will be issued if two port instances have the same value for IPORT, or ifan IPORT is missing (e.g. maximum IPORT number found in the netlist is 4, and there is noinstance with IPORT 3).

• RPORT

Value of the Reference Impedance in Ohms. Default value is 50Ω.

• CPORT

Capacitor placed in series with RPORT. Defaults to 0, in which case it behaves like a zerovoltage source (i.e. CPORT would have no effect).

• LPORT

Inductor placed in series with RPORT. Defaults to 0.

• ZPORT_FILE

Specifies the Touchstone file name that contains the port source with a complex impedance.Large signal S parameters can be extracted from a complex port impedance.

• MODE=SINGLE|COMMON|DIFFERENTIAL

Mixed-mode S parameter selection.

SINGLE specifies the port as single ended, it is dedicated to S parameter extraction. Default.

COMMON and DIFFERENTIAL specify that the port is not single ended. Such ports are splitinto two linked sources that are either common (same amplitude and same phase) ordifferential (same amplitude but opposite phases). During S parameter extraction a “non-single ended” port is equally common and differential depending on which display isrequired. During simulation (DC, AC or TRAN) this port is either common or differentialdepending on the specified mode keyword.

NotePort numbers in Vyy instances should range from 1 to the total number of ports withoutdiscontinuity. The simulation parameters FMIN, FMAX, and Number of frequency pointsfor the analysis are specified with a .AC command.

Working with S, Y, Z ParametersS, Y, Z Parameter Extraction


S, Y, Z Parameter ExtractionOnce the simulation has been setup (see “Simulation Setup for S, Y, Z Parameter Extraction” onpage 193), large signal S, Y, Z parameters can be extracted during steady-state (.SST)simulation by use of the .EXTRACT command.

Multitone and Autonomous Large-Signal S-ParameterExtraction

NoteIn addition to the method described below, you can also extract large-signal/small-signalS-parameters using steady-state AC analysis (see “.PLOT/.PRINT SSTAC” on page 78),which you may find more intuitive, faster and more compact.

It is possible to extract large signal S-parameters for any different fundamental frequency andharmonic for the input and the output ports. This feature is also supported for autonomouscircuits; for instance, the extraction of large signal S-parameters of a circuit containing anoscillator and a down-converting-mixer, where Port 1 is specified at the RF frequency at oneinput of the mixer (fund2) and Port 2 is specified at the IF frequency (fund2-fund_osc) at theoutput of the mixer.

The extraction of large signal S-parameters is conditioned by the .PLOT command.

The large signal S-parameters S(i,j) are computed as the ratio of the transmitted (or reflected)wave Bj at all the spectrum frequencies and the incident wave ai at the frequency specified atPort i.

For example:

Vrf rf 0 iport=1 rport=50 FOUR fund2 Pdbm (1) -20 0Vout_if mix_out 0 iport=2 rport=50.SST fund1=f_LO nharm1=10 fund2=f_RF nharm2=5.PLOT FSST SDB(2,1) SDB(1,1)

The .PLOT command activates the computation of the large signal S-parameters S21 and S11,where the incident wave a1 is at Port 1 at the first harmonic of the second fundamentalfrequency FUND2. The reflected wave b1, and the transmitted wave b2 are computed at all thefrequencies of the spectrum. The x-axis of the computed S-parameters corresponds to thefrequencies of the corresponding reflected or transmitted wave.

For S-Parameter Extraction.EXTRACT FSST yval(SR(i, j), fund1).EXTRACT FSST yval(SDB(i, j), fund1).EXTRACT FSST yval(SI(i, j), fund1).EXTRACT FSST yval(SM(i, j), fund1)


Working with S, Y, Z ParametersS, Y, Z Parameter Extraction

.EXTRACT FSST yval(SP(i, j), fund1)

.EXTRACT FSST yval(SGD(i, j), fund1)

For Mixed Mode S-Parameter ExtractionMixed mode S parameters can be extracted using the following syntax:

S[mn]TYPE(i,j)

TYPE can be one of the following:

R Real partM MagnitudeI Imaginary partDB Magnitude (dB)P PhaseGD Group Delay

mn specifies the mode of ports i and j respectively, can be one of the following:

cc common-commondd differential-differentialdc differential-commoncd common-differentialsc single-commonsd single-differentialcs common-singleds different-singless single-single. Default.

The default mixed mode for S parameter extraction is single-single. If no mixed extension isspecified on the output the default will change depending on how ports 1 and 2 are setup. Thedefault rule is shown in Table 8-1.

Table 8-1. Default Rule

Port 1 Port 2 Default Available quantities

Single Single SS S[ss](1,1)S[ss](1,2)S[ss](2,1)S[ss](2,2)

Single Balanced SD Sss(1,1)Ssd(1,2) Ssc(1,2)Sds(2,1) Scs(2,1)Sdd(2,2) Sdc(2,2) Scd(2,2) Scc(2,2)

Balanced Balanced DD Sdd(1,1) Sdc(1,1) Scd(1,1) Scc(1,1)Sdd(1,2) Sdc(1,2) Scd(1,2) Scc(1,2)Sdd(2,1) Sdc(2,1) Scd(2,1) Scc(2,1)Sdd(2,2) Sdc(2,2) Scd(2,2) Scc(2,2)

Working with S, Y, Z ParametersMatrix Parameter Extraction


The default can be set using the .LIN command, with syntax:

.LIN mixedmode2port=dd|dc|ds|cd|cc|cs|sd|sc|ss

Example

V1 in1 in2 iport=1 MODE=singleV2 in4 in5 iport=2 MODE=differential.PLOT AC Sdb(1,1) Sscm(1,2) Ssdp(1,2) Sddr(2,2) Scdi(2,2)

For Y-Parameter Extraction.EXTRACT FSST yval(YR(i, j), fund1).EXTRACT FSST yval(YI(i, j), fund1).EXTRACT FSST yval(YM(i, j), fund1).EXTRACT FSST yval(YDB(i, j), fund1).EXTRACT FSST yval(YP(i, j), fund1).EXTRACT FSST yval(YGD(i, j), fund1)

For Z-Parameter Extraction.EXTRACT FSST yval(ZR(i, j), fund1).EXTRACT FSST yval(ZI(i, j), fund1).EXTRACT FSST yval(ZM(i, j), fund1).EXTRACT FSST yval(ZDB(i, j), fund1).EXTRACT FSST yval(ZP(i, j), fund1).EXTRACT FSST yval(ZGD(i, j), fund1)

Where Sxx(i, j), (Yxx(i, j), Zxx(i, j)) give the influence of port j on port i.

Matrix Parameter ExtractionOnce the simulation has been setup (see “Simulation Setup for S, Y, Z Parameter Extraction” onpage 193) then G, H, T, A parameters can be extracted during an AC or FSST simulation by useof the .PRINT and .PLOT commands as described in the following subsections.

For G-Parameter Extraction.PLOT AC GR(i, j).PRINT AC GI(i, j).PRINT AC GM(i, j).PLOT AC GDB(i ,j)

Balanced Single DS Sdd(1,1) Sdc(1,1) Scd(1,1) Scc(1,1)Sds(1,2) Scs(1,2)Ssd(2,1) Ssc(2,1)Sss(2,2)

Table 8-1. Default Rule

Port 1 Port 2 Default Available quantities


Working with S, Y, Z ParametersOutput File Specification

.PRINT AC GP(i, j)

.PRINT AC GGD(i, j)

For H-Parameter Extraction.PLOT AC HR(i, j).PRINT AC HI(i, j).PRINT AC HM(i, j).PLOT AC HDB(i ,j).PRINT AC HP(i, j).PRINT AC HGD(i, j)

For T-Parameter Extraction.PLOT AC TR(i, j).PRINT AC TI(i, j).PRINT AC TM(i, j).PLOT AC TDB(i ,j).PRINT AC TP(i, j).PRINT AC TGD(i, j)

For A-Parameter Extraction.PLOT AC AR(i, j).PRINT AC AI(i, j).PRINT AC AM(i, j).PLOT AC ADB(i ,j).PRINT AC AP(i, j).PRINT AC AGD(i, j)

Where Sxx(i, j), (Yxx(i, j), Zxx(i, j)) give the influence of Port j on Port i.

Output File Specification.ffile S|Y|Z|G|H|T|A [SINGLELINE] FILENAME [HZ|KHZ|MHZ|GHZ] [RI|MA|DB]

ParametersS Specifies S (Scattering) frequency parameters tabulation.

Y Specifies Y (Admittance) frequency parameters tabulation.

Z Specifies Z (Impedance) frequency parameters tabulation.

G Specifies G (Hybrid-G) matrix parameters tabulation.

H Specifies H (Hybrid-H) matrix parameters tabulation.

T Specifies T (transfer scattering) matrix parameters tabulation.

A Specifies A (chain or ABCD) matrix parameters tabulation.



FILENAME Name of the file where the S, Y, Z, G, H, T and A parameters will be stored.

SINGLELINE This enables you to obtain the S-parameter file in single line format as shownbelow:

Freq S11 S21 S12 S22

HZ Specifies the units to be Hz. This is the default.

KHZ Specifies the units to be kHz.

MHZ Specifies the units to be MHz.

GHZ Specifies the units to be GHz.

RI Specifies Real Imaginary storage format.

MA Specifies Magnitude Angle storage format. This is the default.

DB MA with magnitude in dB.

Two-port noise parameters NFMIN_MAG, GAMMA_OPT_MAG, PHI_OPT and RNEQ areautomatically written to the specified output file when a .NOISE command is specified in thenetlist and the circuit to be analyzed is a two-port circuit.

Examples

r1 1 2 100kc1 2 0 10pfV1 1 0 iport=1 rport=100V2 2 0 iport=2 rport=20.ac dec 10 1 100meg.plot ac sdb(2,1).Ffile S sb1.par khz ri

In this example, the S parameters of an RC circuit are extracted between 1Hz and 100MHz with10 points per decade. The reference impedance is 100 for port1 and 20 for port2. Themagnitude of S21 is plotted in dB, and the extracted S parameters are stored in the file sb1.parwith the frequency in kHz. The data is stored in the form of the Real and Imaginary parts.

V1 1 0 iport=1 rport=50

R1 1 n1 1kC1 n1 0 100pR2 n1 2 1kRc1 n1 0 100k

V2 2 0 iport=2 rport=50

.ac lin 21 1meg 21meg

.noise v(n1) V1 3

.plot noise rneq gopt bopt nfmin_mag

.ffile Z Z.par kHz ma



In this example the Z parameters are being extracted between 1MHz and 21MHz with 21analysis points. The extracted Z parameters are stored in the file Z.par with the frequency inkHz. The data is stored in the form of the Magnitude Angle. As the circuit is a two-port circuitand there is a .NOISE command specified in the netlist then the two-port noise parameters arealso stored in the output file. The output file is shown below:

! Data from test# KHZ Z DB R 5.000000E+01!1.0000000000000000E+03 6.5542511585029786E+01 -5.7202544106307606E+016.4035302689753806E+01 -8.9088186330215692E+01 6.4035302689753806E+01-8.9088186330215706E+01 6.5542511585029786E+01 -5.7202544106307606E+01...

2.1000000000000000E+04 6.0025369785355387E+01 -4.3338006361865595E+003.7592014242230192E+01 -8.9956576643609139E+01 3.7592014242230199E+01-8.9956576643609139E+01 6.0025369785355402E+01 -4.3338006361865711E+00

! Noise Data: Nfmin(dB) GammaOpt PhiOpt Rneq/R01.0000000000000000E+03 5.2661113010469025E+00 9.6890047604421903E-011.7851510536508835E+02 4.8497683522933400E+01...

2.1000000000000000E+04 2.8441528902447214E+01 9.0554147314402922E-011.7956971588917614E+02 3.5225984336136335E+03

Simulating a Block Defined by its S-Parameters

For technical background and additional implementation issues, refer to TechnicalBackground of the Eldo User’s Manual.

Basic FunctionalityThe S-model implemented in Eldo is a building block that makes possible DC, AC, andTransient simulation of circuits with any number of N-ports described by their S (Scattering), Y(Admittance), or Z (Impedance) parameters, in the form of tabulated data in the frequencydomain.

In addition, Eldo RF enables the steady-state simulation of the same N-port blocks. This can beuseful to simulate a block for which only simulated or measured S-parameters are available,such as a filter or antenna. Notice that Eldo RF can extract large signal S-parameters aspreviously explained, however it can only simulate small-signal S-parameters.

For details of DC, AC, Transient simulation, refer to the Eldo User’s Manual.

The tabulated data is contained in an ASCII data file in the Touchstone® format. This format isbriefly described in “Touchstone Data Format” on page 205. When giving the instance of the



model in the Eldo netlist file, you can specify the data file either by setting an index associatedwith the file’s name, or by explicitly defining the name of the file. The optimal of the threealgorithms discussed above, Complex Pole Fitting (CPF), Digital Signal Processing (DSP), andSystem Identification (SI), is selected by the internal Eldo monitor that allows great flexibilityof simulation. You can also specify this method directly.

Detailed Functionality

For details of DC, AC, Transient simulation, refer to the Eldo User’s Manual.

• Simulation of N-ports with no restriction on N. N is determined automatically by theEldo parser according to the number of pins.

• Any number of instances of the same model, any number of different models.

• All SST and MODSST simulation types are supported.

• Input data can be specified as S, Y, Z, G, H, T or A parameters.

• Tabulated data may have linear, logarithmic, or irregular distribution in its frequencyrange. Frequency values may start from 0 Hz or any positive value. Any number ofpoints is allowed.

• The choice of the CPF, DSP, or SI algorithm can be forced through a parameter of themodel. All algorithms allow any kind of spacing. However, since internally DSPrequires linearly spaced K2+1 data points starting at zero frequency, it uses interpolationand extrapolation of the input data to satisfy this requirement.

• Simulation is possible with the Eldo default options.

• For an N-port block (N>1), port impedances can be identical, or different for each port.

• Speed-optimized C-FAS model.

Instantiating a Block Defined by S-Parameters.MODEL FBLOCK MACRO LANG=CYNAME FBLOCK PARAM:+ [M=VAL]+ [IDX_M=VAL]+ [NO_DELAY=VAL]+ [GROUPFIT=VAL]+ [SYMMETRY=VAL]+ [FORCE_PASSIVITY=VAL]+ [FORCE_REFIT=VAL]+ [EXTRAP_TO_DC=VAL]+ [POLE_REDUCTION=VAL]+ [HIGH_PRECISION=VAL]+ [MAXROW=VAL]+ [MAXCOL=VAL]



+ [IDX_F=VAL]+ STRING: FILENAME+ PIN: IP1 IN1 ... IPN INN

The first line (.MODEL) is the reference to the C-FAS model, where the entity is called FBLOCK.The other lines are the model parameters.

The keyword PARAM: precedes the list of parameters, (each one shown in brackets as they areall optional). Most of the options, except for M, IDX_M and NO_DELAY, are specific to the CPFmethod.

M is a device multiplier parameter, simulating the effect of multiple S-block elements inparallel. Default value is 1.

IDX_M is a parameter that forces a specific algorithm to be used (IDX_M=0 forces the CPFalgorithm, IDX_M=1 specifies DSP, IDX_M=2 specifies SI). The default value of IDX_M is 0(CPF).

NO_DELAY is used to allow or prevent delay extraction in the CPF or DSP methods.NO_DELAY=0 allows delay extraction, NO_DELAY=1 forbids it. The default value is 0.

GROUPFIT=1 is used in CPF to force group fitting instead of individual for every matrixcomponent. As a rule, with this option, fitting requires less effort but this might compromiseaccuracy. By default, its value is 0 that corresponds to individual fitting.

SYMMETRY=0 disables the default assumption (SYMMETRY=1) made in CPF on the fitting stagethat the original S (or Y or Z) matrix is symmetric. Matrix symmetry is a valid assumption aslong as the S-model describes a reciprocal subcircuit. We cannot simply rely on symmetry ofthe matrices in the input data. Very often, the input matrices generated by field-solvers ormeasured from reciprocal systems, are not strictly symmetric, however they should be handledas symmetric.

FORCE_PASSIVITY=val enables or disables each of the two different types of passivityenforcement available in the CPF method. These types are (1) pre-fit passivity enforcement, inwhich the original sampled data is worked with to make it “passive,” and (2) post-fitenforcement, in which poles/residues are corrected in such a way as to make the approximationstrictly passive.

FORCE_PASSIVITY=0 (default) means there is no passivity enforcement.FORCE_PASSIVITY=1 activates pre-fit passivity enforcement.FORCE_PASSIVITY=2 activates post-fit enforcement.FORCE_PASSIVITY=3 activates them both.

Pre-fit passivity enforcement is recommended for all passive devices. It removes occasionalpassivity violations from the input data (which may result from measurement errors). However,even for the passive data created by pre-fit passivity enforcement, fitting may still result in anon-passive model if this data is defined within a limited frequency range (typical case). With



two different methods of passivity enforcement, you can determine the true reason for non-passivity: poor accuracy of the input data or fitting errors. The reason for both could beincomplete frequency range, non-causality, or insufficient resolution of the input data. Forcausal, accurate, and smooth input data, fitting accuracy is quite high.

If non-reciprocal linear active devices (such as amplifiers or filters) are to be simulated, bothSYMMETRY and FORCE_PASSIVITY should be disabled.

FORCE_REFIT=1 forces fitting in CPF regardless to whether the corresponding .pls file isalready present or not. This might be needed if we want to redo fitting with different options,such as FORCE_PASSIVITY. However, you should be careful in using such an option, it shouldbe disabled after the desired fit is built. By default, FORCE_REFIT is disabled.

EXTRAP_TO_DC=1 restores a missing point at zero frequency (DC) by extrapolating the curvefrom low frequency points given in the Touchstone file. If the DC point is present in the inputdata, this option has no effect. Compared to the default case (EXTRAP_TO_DC=0) it allows, as arule, to achieve better accuracy in DC simulation when the point at zero frequency is not given.

POLE_REDUCTION=1 (default value) enables the mode of transient simulation in which some ofthe fitted poles (that are too fast, too slow or too small) are removed in order to speed-up thesolution. This mode typically gives up to 30-50% reduction in solution time when the step of thetransient solution is fixed. The decision about pole reduction is made from considering thesolution step (pole is too “fast”), or the duration of the simulation interval (pole is too “slow”).Therefore, the set of actually used poles is defined “dynamically” from considering theparameters of the .TRAN command. The generated list of poles/residues (*.pls) file remainsunaffected. Pole reduction does not considerably affect the solution accuracy. However, if theprecise simulation is needed, the option can be disabled by setting POLE_REDUCTION=0.

HIGH_PRECISION=1 increases fitting accuracy by allowing more poles than in regular mode(with default value 0). This option can be useful for verification purposes, for example if a“reference solution” is required. However, it is not recommended if the input data itself is notvery accurate. Also, since high-precision fitting produces more poles, it makes simulationslower.

MAXROW=VAL sets the limit (val/2) to the frequency points of the original dependence used infitting. By default, MAXROW=40000, that corresponds to 20,000 points.

MAXCOL=VAL sets the limit (val/2) to the maximal order of complexity for fitting in CPF. Bydefault, MAXCOL=1500, that corresponds to a order of complexity of 750. For very complicated(sharp, irregular) dependencies it is sometimes reasonable to reduce the order of complexity,especially if we have reasons not to entirely trust the input data at higher frequencies. As a rule,reducing order of complexity is a better strategy than reducing the number of points to consider(MAXROW).

The keyword STRING: is to define the name of the touchstone file, containing the input data.Path definition is allowed. Another way of defining the data file is using the parameter IDX_F.



Note that the parameter IDX_F should be defined under the keyword PARAM: together with allother parameters, not under the STRING: keyword. This parameter defines the index (integernumber) VAL associated with the S parameter file (IDX_F=VAL implies that the input parameterfile is named SBVAL.PAR).

The 2×N pins of the N-port model will be connected to nodes IP1 IN1 ... IPN INN.

A single reference node is supported. When the number of pins of the FBLOCK model is even,Eldo considers that each port has two pins. When the number of pins is odd, Eldo considers thereference pin is the same for all ports (and it is the last pin).

Any model may be instantiated as many times as required with the same or different input datafile.

Any FBLOCK instance will contribute to the global noise results of .NOISE and .SSTNOISE. Ifthe Touchstone format file contains noise parameters then they will be used to compute thenoise contribution, otherwise the simulator will use the Twiss formula.

Twiss Formula

Where:

Cy = Noise Correlation Matrix

k = Boltzmann Constant

t = Temperature

Y = Y Parameter Matrix

H = Hermitian Matrix (complex conjugate transpose)

The FBLOCK file parameter is searched with the same methodology as searching library files,see Search path priorities in the Eldo User’s Manual. This means that if the FBLOCK file is notfound in the current directory, the library where the corresponding FBLOCK instance wasfound is searched first if FBLOCK was actually read from a library. If not found, the directoriesare searched in the order specified by the option SEARCH.

Examples

.model dio D rs=4.68 bv=6.1 cjo=246p

.model Fblock macro lang=cvin 1 0 dc 5 ac 1 pulse(0 5 1n 1n 1n 5n 10n)rin 1 2 50

ytline Fblock param:+ force_passivity=1+ string: C:\s-parameterdata\lowpassfilter.s2p

C y 2kt Y YH

+( )=

Working with S, Y, Z ParametersTouchstone Data Format


+ pin: 2 0 3 0

dout 3 0 dio.ac dec 10 1 10meg.tran 10n 100n.plot ac vdb(2) vdb(3).plot tran v(1) v(2) v(3).end

In the above example, we define a block ytline. By default, the CPF method runs. Delayextraction is allowed (if feasible), fit is set to individual, the model is assumed symmetric,passivity enforcement is set for pre-fit stage; refit, extrapolation to DC, and high precision flagsare disabled, and the parameters MAXCOL, MAXROW are set to their default values, 1500 and40,000 respectively. The file name is given in conventional form, by using the keyword“string:”.

.subckt sparam_2p p1 p2 grnd

.model Fblock macro lang=cy2port FBLOCK param:+ idx_f=4+ idx_m=1+ no_delay=1+ pin: p1 grnd p2 grnd.ends sparam_2p

In this example, a block y2port refers to an S-parameter file sb4.par (since idx_f=4). Here,the S-block is described as a sub-circuit. We choose the DSP method and prevented delayextraction.

Touchstone Data FormatThe Touchstone® data format file is an ASCII text file in which data appears line by line: Nlines for each data point of N ports. The data points are stored in increasing order of frequency.

The first of these N lines consist of a frequency value and N pairs of values for S, Y, Z, G, H, Tor A parameters.

The (N-1) following lines contain N pairs of values.

Values are separated by one or more spaces or tabulations.

Touchstone data format files follow general syntax rules. The standard is available fromthe EDA Industry Working Groups website:http://www.eda.org/pub/ibis/connector/touchstone_spec11.pdf

Example of S parameters for three ports:

F SR11 SI11 SR12 SI12 SR13 SI13SR21 SI21 SR22 SI22 SR23 SI23

http://www.eda.org/pub/ibis/connector/touchstone_spec11.pdf



SR31 SI31 SR32 SI32 SR33 SI33

NoteTwo ports may also represented in single line format but will have a different parameterorder (notice that S12 and S21 are swapped), see below.Two ports on a single line: Freq S11 S21 S12 S22

Two ports on dual lines: Freq S11 S12

S21 S22

Comment lines begin with an exclamation mark (!). The first un-commented line in the filemust be a specification line. An optional specification line begins with the number symbol (#)followed by a space. Then, several optional parameters are specified in the following order:

• Frequency Unit (Hz, kHz, MHz, GHz). Default value is GHz.

• Parameter type (S, Y, Z, G, H, T, A). Default value is S.

• Data format (MA, DB, RI). Default value is MA. MA means Magnitude-angle in Voltsand degrees. DB means Magnitude in dB, and phase in degrees. RI means Real andImaginary parts.

• Reference impedance of each port (when all the ports have the same referenceimpedance, only one may be specified). Default value is all ports with the same 50 Ωreference impedance.

# [Hz|kHz|MHz|GHz] [S|Y|Z|G|H|T|A] [RI|MA|DB]+ [R Val|R1 Val1 ... Rn Valn]

• The two-port noise parameters (NFMIN, GAMMA_OPT_MAG, PHI_OPT, RNEQ) canbe used when you have specified a .NOISE command in the netlist and when the circuitto be analyzed is a two-port circuit. NFMIN is the minimal noise figure of the two-port.GAMA_OPT is the magnitude of the optimal reflection coefficient associated with theminimum noise figure. PHI_OPT is the angle of the optimal reflection coefficientassociated with the minimum noise figure. RNEQ is the equivalent noise resistance.

Example

# khz s ri r 50

Frequency values are in kHz, the data are S-parameter data, they are stored in the format Realand Imaginary part and the reference impedance is 50Ω for each Port.

Mixed Mode S-Parameter ExtractionWhen extracting mixed mode S parameters the contents of the Touchstone output data file willchange. For example the file header for a 2-port network may appear as follows:



! Data from foo! S11 = SDD11! S12 = SDS12! S13 = SDC11# HZ S RI R1 1.000000E+01 R2 1.000000E+00


Chapter 9Transmission Lines

Lossy Transmission LineLDTL (Lossy Dispersive Transmission Line) is a model implemented in Eldo to simulatetransmission lines. It is dedicated for the simulation of lossy coupled uniform lines, alsoincluding dispersive effects. This model can be used in all analysis modes (DC, AC, Transient,SST, SSTNOISE, or MODSST). To specify the line parameters to the LDTL model, fourdifferent inputs are available:

1. the first level corresponds to R, L, C and G matrices,

2. the second level uses a file at XFX output format as input to specify the line parameters,

3. the third level corresponds to the electrical parameters for a single line,

4. the fourth level corresponds to the geometrical and physical parameters for a singlestripline or up to two coupled microstrip lines.

Ways of instantiation are shown on the following pages. You can also instantiate an LDTLmodel by using a model of MODFAS type (see .MODEL of the Eldo User’s Manual) directlyin the parameter list. This model must be specified at the end of the instantiation line and cancontain any parameter. For each level, an example is provided in the directory$MGC_AMS_HOME/examples/tlines/LDTL_model.

Level 1Yxx LDTL [PIN:] P1...PN [REFin] PN+1...P2N REFout+ [PARAM: [LEVEL=1] [LENGTH=val] [SAVEFIT=val] [M=VAL]+ [R(i)=val] [L(i,j)=val] [C(i,j)=val] [G(i,j)=val] [FR1=val]]

The first level is dedicated to the simulation of an infinite number of coupled transmission lines.The Maxwell matrices (R, L, C and G) are used to describe the line system.

Parameters

• xx

Transmission line name.

• P1...PN

The N nodes at one end of the line system for a system consisting of N lines.

• REFin

Optional reference node for input signal, used to simulate differential lines.


Transmission LinesLossy Transmission Line

• PN+1...P2N

The N nodes at the other end of the line system. The line number i in the line systemconnects the nodes Pi and PN+i.

• REFout

Reference node for output signal. If REFin is not specified, then both sides of the linesystem have the same reference plane.

• LENGTH=val

Geometric length of the line system. Default value is 1 m. If LENGTH=0, Eldo uses thedefault value.

• LEVEL=1

Keyword to specify the input format. Value 1 specifies the “R,L,G,C matrices” format.Default value is 1.

• SAVEFIT=val

SAVEFIT=1 ⇒ Saves the initialization of the transmission line model (in the filecircuit_name.fit), in order to speed up the following simulations of the same netlist. Defaultvalue is 0.

• M=val

Device multiplier. Simulates the effect of multiple devices in parallel. In effect the currentvalue is multiplied by M. Default is 1. .OPTION YMFACT must be selected in order for thisoption to work.

• R(i)=val

Value of the (i,i) element per unit length of the resistance matrix: R. Default values are50 Ωm-1.

• L(i,j)=val

Value of the (i,j) element per unit length of the inductance matrix: L. Default values1×10-6Hm-1 for the self inductance and 0 for the mutual inductances.

• C(i,j)=val

Value of the (i,j) element per unit length of the capacitance matrix: C. Default values1×10-9Fm-1 for the self capacitance and 0 for the mutual capacitances.

• G(i,j)=val

Value of the (i,j) element per unit length of the conductance matrix: G. Default values are0 Sm-1.

• FR1=val

Frequency at which dispersion starts (only affects resistance). Default: no dispersion will beconsidered.

If for a line only R(1) is specified, this value is used for all lines. Specification of theTransmission line matrix parameter can be done in one of the following ways:



• Complete matrix of coefficients consisting of N×N values.

• Only the upper (or lower) triangular matrix because of the matrix symmetry.

• Only the first row (or column) of the matrix. This is normally sufficient if all lines in thesystem have the same width and spacing.

Example

Circuit name: YLDTL_level1_example.cir. Three coupled lines defined by R, L, C and Gmatrices:

Y1 LDTL 2 3 4 0 5 6 7 0+ param: LEVEL=1 length=0.677+ R(1)=15 L(1,1)=418n C(1,1)=94p G(1,1)=0.02p+ R(2)=15 L(2,2)=418n C(2,2)=94p G(2,2)=0.02p+ C(1,2)=-22p C(2,3)=-22p+ L(1,2)=125n L(2,3)=125n+ R(3)=15 L(3,3)=418n C(3,3)=94p G(3,3)=0.02p

Notice that C(1,2) and C(2,3) are both negative. This is because Eldo uses the Maxwell matrix.The capacitance matrices are based on the admittance matrix of the capacitances between theconductors. The negative values in the capacitance matrix are due to the sign convention foradmittance matrices.

Same example using a .MODEL in the instantiation:

Y1 LDTL 2 3 4 0 5 6 7 0+ param: LEVEL=1 length=0.677+ model:level1_mod

.model level1_mod MODFAS R(1)=15+ L(1,1)=418n C(1,1)=94p G(1,1)=0.02p+ R(2)=15 L(2,2)=418n C(2,2)=94p G(2,2)=0.02p+ C(1,2)=-22p C(2,3)=-22p+ L(1,2)=125n L(2,3)=125n+ R(3)=15 L(3,3)=418n C(3,3)=94p G(3,3)=0.02p

Level 2Yxx LDTL [PIN:] P1...PN [REFin] PN+1...P2N REFout+ [PARAM: [LEVEL=2] [LENGTH=val] [SAVEFIT=val]+ [XFX_IDF=val] [FP=val] [MULTIDEBYE=val]]

The second level uses a file of XFX output format as input to specify the line parameters. Thislevel is dedicated to the simulation of an infinite number of coupled transmission lines.

Parameters

The description of the global parameters (PINs, LENGTH and SAVEFIT) is as specified for theLEVEL=1 format. Level 2 specific parameters are shown below.



• LEVEL=2

Keyword to specify the input format. Value 2 specifies the XFX format. Default value is 1.

• XFX_IDF=val

Index relative to the XFX file name. If XFX_IDF=val, Eldo will search for the fileXFX_val.tlp. Eldo searches for the specified file in a specific order, see Search pathpriorities in the Eldo User’s Manual.

• FP=val

Polarization frequency to control dispersive effect on the conductance (see “TechnicalPrecision” on page 217). Default: 1.6×109.

• MULTIDEBYE=val

val=1 specifies the use of multi-pole debye model to model the dispersive effect on theconductance (recommended for modeling PCB-type dielectrics). val=0, this model is notused. (see “Technical Precision” on page 217). Default: 1.

Example

Circuit name: YLDTL_level2_example.cir. Two coupled lines defined by an XFX output file.

Y1 LDTL 1 2 3 4 0+ param: LEVEL=2 length=0.1 xfx_idf=12


Y1 LDTL 1 2 3 4 0+ param: model:level2_mod.model level2_mod MODFAS LEVEL=2 length=0.1 xfx_idf=12

The parameter xfx_idx=12 is a reference to the file XFX_12.tlp containing the lineinformation generated by XFX. Here is an example of such a file:

XFX V6.0.0.0 Report 5 Nov 15:57 1997 Setup File=ex3.xfx

Configuration Name: ANALOG1 Conductors: 2

Conductor index: 0 name: $$GND$$Conductor index: 1 name: AConductor index: 2 name: C

i j Lij Cij Ze Zo Se So Fwdx Rvsxfrom to (nh/in) (pf/in) (ohms) (ohms) (ns/ft) (ns/ft) (s/s) (v/v)--------------------------------------------------------------------- 1 1 8.631 3.742 48.02 - 2.16 - - - 1 2 6.87e-10 2.98e-10 48.02 48.02 2.16 2.16 0.000 0.000 2 2 8.631 3.742 48.02 - 2.16 - - -

: LOSS MATRICES i j Rsij Gij Rdcij Gdcijfrom to (ohm-nsec^.5) (mS-ns) (ohms) (mS) PER INCH-------------------------------------------------------------------- 1 1 0.7 0.0 0.47929 0.46827



1 2 0.00 0.0 0.00000 0.02912 2 2 0.7 0.0 0.34473 0.45333;

Level 3Yxx LDTL [PIN:] P1 [REFin] P2 REFout+ [PARAM: [LEVEL=3] LENGTH=val] [SAVEFIT=val]+ [Zc=val] [VREL=val] [TD=val] [L=val] [C=val] [R=val] [FR1=val] [M=val]]

The third level corresponds to the electrical parameters for a single line only.

Parameters


• LEVEL=3

Keyword to specify the input format. Value 3 specifies the electrical format. Default valueis 1.

• Zc=val

Characteristic impedance (Ω). If this value is not specified, it is calculated with the values ofL and C.

• VREL=val

Relative velocity. If this value is not specified, it is calculated with the values of L and C.

• TD=val

Delay for LENGTH (implies total delay calculated is LENGTH×TD). If not specified,calculated.

• L=val

Inductance per unit length. Default value: 1×10-6Hm-1.

• C=val

Capacitance per unit length. Default value is: 1×10-9Fm-1.

• R=val

Linear resistance. Default value is 50 Ωm-1.

• FR1=val

Frequency (Hz) at which dispersion starts (only affects resistance). Default: no dispersionwill be considered.

• M=val

Device multiplier. Simulates the effect of multiple devices in parallel. In effect the currentvalue is multiplied by M. Default is 1. .OPTION YMFACT must be selected in order for thisoption to work.



You can either specify directly the line parameters R, L and C, or use any combination ofelectrical parameters (L and C can be computed with electrical parameters, see “TechnicalPrecision” on page 217).

Example

Circuit name: YLDTL_level3_example.cir. One dispersive line defined by electrical parameters.

Y1 ldtl 1 2 0+ param: LEVEL=3 length=100 R=1+ ZC=50 VREL=0.66 FR1=100Meg


Y1 ldtl 1 2 0+ param: LEVEL=3 length=100 R=1+ ZC=50 model: level3_mod

.model leve3_mod MODFAS VREL=0.66 FR1=100Meg

Level 4Yxx LDTL [PIN:] P1 [REFin] P2 REFout+ [PARAM: [LEVEL=4] [LENGTH=val [SAVEFIT=val]+ [DLEV=val] [PLEV=val] [ER=val] [H=val] [W=val] [T=val]+ [RHO=val] [TAND=val] [H1=val] [FP=val] [H2=val] [S=val]+ [THICKNESS=val] [DISPERSIVE=val] [USE_ER=val] [M=val] [MULTIDEBYE=val]]

The fourth level corresponds to the geometrical parameters specification for microstrip line andstripline. Up to two coupled microstrip lines are allowed and only one single stripline can betaken into account. The structure of these transmission lines is illustrated in Figure 9-2 (coupledpair of microstrip lines) and Figure 9-3 (stripline).

Parameters


• LEVEL=4

Keyword to specify the input format. Value 4 specifies the geometrical and physicalformat. Default value is 1.

• DLEV=val

Type of line: 1 for microstripline; 2 for stripline. Default is 1.

• PLEV=val

Type of equations: 0 uses the equations from the references (1) and (2), 1 for simplifiedequations. Default is 0.



• ER=val

Dielectric relative permittivity. Default value is 9.8 (alumina).

• H=val

Dielectric thickness (m). Default value is 400×10-6m.

• W=val

Conductor width. Default value is 50×10-6 m.

• T=val

Conductor thickness. Default value is 5×10-6 m.

• RHO=val

Conductor resistivity. Default value is 17×10-9 Ωm (copper).

• TAND=val

Dielectric loss tangent. Default value is 0.

• H1=val

Conductor height, only for stripline configuration. Default value is 197.5×10-6 m.

• FP=val

Polarization frequency to control dispersive effect on the conductance (see “TechnicalPrecision” on page 217). Default: 1.6×109.

• H2=val

Height between dielectric and a possible cover plate. Only for coupled microstripconfiguration. Default: 0.0, means that no cover plate is taken into account.

• S=val

Spacing between the two conductors. Default = conductor width value (W). Only forcoupled microstrip configuration.

• THICKNESS=val

Take into account effect of finite strip thickness if val=1. Default: 0. Only for coupledmicrostrip configuration.

• DISPERSIVE=val

Take into account dispersive effect if val=1. Default: 0. Only for coupled microstripconfiguration.

• USE_ER=val

Use directly the dielectric relative permittivity (ER) to compute the characteristicimpedance (Zc) if val=1. Otherwise (if val= 0), an effective relative permittivity will becalculated and used in the Zc computation (see Technical Precision, “Level 4” on page 220for details). Default: 1.



• M=val

Device multiplier. Simulates the effect of multiple devices in parallel. In effect the currentvalue is multiplied by M. Default is 1. .OPTION YMFACT must be selected in order for thisoption to work. Only used for stripline (DLEV=2).

• MULTIDEBYE=val

val=1 specifies the use of multi-pole debye model to model the dispersive effect on theconductance (recommended for modeling PCB-type dielectrics). val=0, this model is notused. (see “Technical Precision” on page 217). Default: 1.

NoteWhen PLEV=0 and DLEV=2, it is only possible to describe an off-centered microstrip linewith the equations based on reference [1].

Examples

Circuit name: YLDTL_level4_example.cir. A microstrip line based on equations fromreference [1].

Y1 LDTL 1 2 0+ param: LEVEL=4 length=10 PLEV=0 DLEV=1+ h=400u w=50u t=5u rho=17E-09 er=9.8

Circuit name: YLDTL_level4_example2.cir. Symmetric pair of coupled microstrip lines,including finite strip thickness and dispersive effects.

Y1 LDTL 1 2 0 3 4 0+ param: LEVEL=4 length=10e-3 PLEV=1 DLEV=1+ h=635u w=88u t=2u s=90u h2=935u+ rho=1.72E-08 tand=0.01 thickness=1+ dispersive=1


Y1 LDTL 1 2 0 3 4 0+ param: LEVEL=4 length=10 PLEV=1 DLEV=1+ h=635u w=88u model:level4_mod

.model level4_mod MODFAS t=2u s=90u h2=935u+ rho=1.72E-08 tand=0.01 thickness=1+ dispersive=1

LDTL Model Error Message TreatmentA general problem which causes many errors is incorrect time delay values. These values arecalculated by multiplying the L and C matrices. Negative or null time delay values can causeerrors in the model. To avoid the simulation being stopped, here follows some advice on how toavoid errors. The error message is shown, together with advice on avoiding this type of error.



ERROR model Yxx : no time-delay in the transmission line(s), check yourinput parameters.

This means that some value(s) of the L or C matrix (or both) are bad: the diagonal of L×Cmatrix presents null value(s) (see “General Equation for Delay” on page 217). It can appearwhen some off-diagonal term(s) of the C or L matrix are too large. Normally the coupling effecton L and C decreases when the distance between the two concerned lines increase. That means:|C(1,2)| should be larger than |C(1,3)|.

ERROR model Yxx: Non physical line model (negative time delay). Check Cand/or L matrices off-diagonal terms.

This message appears only for coupled transmission line models. It means that the time-delay ofat least one line of the model is negative. So this modelization is not a physical one.

This means the diagonal of L×C matrix presents negative value(s) (see “General Equation forDelay” on page 217). It can appear when some off-diagonal term(s) of the C or L matrix are toolarge. Normally the coupling effect on L and C decreases when the distance between the twoconcerned lines increase. That means: |C(1,2)| should be larger than |C(1,3)|.

ERROR model Yxx : the diagonal of C is non-strictly-dominant : you shouldhave Sum|(C(i,j)| < |C(i,i)| (i != j)

This means some off-diagonal terms of the C matrix are too large. Therefore, the sum of all theoff-diagonal terms of one line of the matrix is not lower than the diagonal term of the line: thestrictly-dominant property is not verified. Such a property is required for the model.

WARNING model Y1 : negative diagonal value(s) : R[1][1]

This warning means that the value R[1][1] is negative. Therefore, you have to check theparameters of the model instantiation according to the LEVEL used (see Technical Precision).

Technical PrecisionHere follows some technical information about the use of the Yxx LDTL model.

General Equation for DelayThe time-delay (Td) of a single transmission line is computed as follows:

When we have a n-coupled transmission line model, the time-delay matrix is computed asfollows (L, C and Tdm are matrices):

Td Length LC×=

Tdm Length diag L C×( )×=



Tdm is a diagonal matrix. The nth element of the diagonal is the time-delay of the nth line of themodel. Therefore, these diagonal values must be positive.

Level 1To introduce skin effects in the line model (loss that is proportional to the square root offrequency), you just have to specify the FR1 parameter. Then the resistivity value will befrequency dependent:

for the ith transmission line.

Level 2Here the skin effect is introduced by the parameter Rs:

You can also introduce frequency dependent conductance by using the parameters Gs and fp(polarization frequency: FP parameter). The conductance dispersive effect can be modeled intwo ways according to the MULTIDEBYE parameter:

• One-pole debye model (MULTIDEBYE=0)

The dispersive effect is obtain according to the following equation:

• Multi-pole debye model (MULTIDEBYE=1)

By using this option, we build a complex frequency-dependent capacitance matrix:

Therefore, line conductance per unit length becomes:

where: and ; C and Gs are the user-defined matrices.

with and

R R i( ) 1 1 i+( ) fFR1-----------+

×=

R RDC RS 1 i+( ) 4πf×+=

G GDC GSi2πf 2πfp×i2πf 2πfp+------------------------------×+=

C ω( ) Cinf f jω( )Cd+=

Y jω( ) GDC jω( )Cinf jω( ) f jω( )Cd+ +=

Cinf C αGs–= Cd βGs=

α

ω22 ω0

2+

ω12 ω0

2+

-------------------

ln

4πω0

ω1------

atanω0

ω2------

atan–

-----------------------------------------------------------------= β 108( )ln

2πω0

ω1------

atanω0

ω2------

atan–

-----------------------------------------------------------------=



Finally, the function which is fitted with 15 real poles.

Note: , , , and fp the polarizationfrequency (FP parameter).

Level 3As already described, you can either specify directly the line parameters R, L and C, or use anycombination of electrical parameters. In order to discard redundant parameter sets we use thefollowing equations:

Clight = 3×108 ms-1 (the speed of light)

The skin effect is introduced by the parameter FR1:

Table 9-1. LDTL Level 3 Parameter Combinations

Input Parameters Equations

Zc, VREL,

Zc, TD,

TD, C

VREL, C

TD, L

VREL, L

any other default values for L and C

f jω( )

ω2 jω+

ω1 jω+--------------------

ln

108( )ln-------------------------------=

ω0 2π f p= ω1 104= ω2 1012

= ω 2πf=

C 1.0Zc VREL Clight××---------------------------------------------------= L Zc

VREL Clight×--------------------------------------=

C TDZc--------= L Zc TD×=

L TD2

C----------=

L 1.0

VREL2

Clight2

C××------------------------------------------------------=

C TD2

L----------=

C 1.0

VREL2

Clight2

L××-----------------------------------------------------=

R R 1 1 i+( ) fFR1-----------+

×=



Level 4Only two transmission line configurations can be described with this input format: microstripline and stripline. The following figures provides the structure of these transmission lines.Figure 9-1 and Figure 9-2 provide the structure for a single and covered pair of microstripmodels, Figure 9-3 provides the structure for a stripline.

Figure 9-1. Microstrip Line Structure

Figure 9-2. Covered Pair Microstrip Line Structure

Figure 9-3. Stripline Structure

The PLEV parameter allows the use of two sorts of equation: reference equations (PLEV=0) andsimplified equations (PLEV=1). Provided here is the reference formulation.

For a Single Microstrip Line (DLEV=1)

• From reference (1) (PLEV=0), we have the following equations:

Conductor (thickness: t)

er

w

h

Conductor pair (thickness: t)

er

wh2

h

Conductor (thickness: t)

erw

h1

h



Capacitance

Inductance

Resistance

with the characteristic impedance:

where:

With , the wave impedance; , the permittivity; and the permeability of freespace.

In reference (1), it is recommended to replace Er by Eeff in the characteristic impedancecomputation. It is done by specified the parameter USE_ER=0.

for :

and for :

For a Symmetric Pair of Coupled Microstrip Lines (DLEV=1)


Cµoεoer

Zc---------------------=

L Zc µoεoer=

R rhow t×------------=

Zcηo

2 2π er 1+--------------------------------- 1

4hw ′------ 14 8 er( )⁄+

11----------------------------- 4h

w ′------

14 8 er⁄+11

------------------------ 2 4h

w ′------

2 1 1 er⁄+2

---------------------π2++×+

ln=

w ′ w w ′∆+=

w ′∆ w1 1 er( )⁄+

2--------------------------

∆=

w∆ 1π--- 4e

t h⁄( )2 1 π⁄w t⁄ 1.1+-----------------------

2+

----------------------------------------------------------ln=

ηo εo µo

wh---- 1≤

EeffEr 1+

2---------------

Er 1–2

--------------- 1 12hw

---------+ 0.5–

0.04 1 12hw

---------– 2

++=

wh---- 1≥

EeffEr 1+

2---------------

Er 1–2

--------------- 1 12hw

---------+ 0.5–

+=



Capacitance matrix

where: and

Inductance matrix

where: and

Resistance matrix

where: and

Conductance matrix

where: and

The indices e and o indicate even and odd mode parameters respectively. The expressionof effective relative permittivity, characteristic impedance and attenuation coefficientsare given in the single microstrip line description (for more details, see reference (2)).Dispersion is taken into account (when required) in the effective permittivity andcharacteristic impedance computations (see reference (2)). Therefore, effectiveparameter matrices (R, L, C and G) values change with frequency.

For a Single Stripline (DLEV=2)


C CMˆ+ CM

ˆ–

CMˆ– C CM

ˆ+

C 1.0v p e, Z L e,×--------------------------= CM

ˆ 12--- 1.0

v p o, Z L o,⋅------------------------- 1.0

v p e, Z L e,⋅-------------------------–

=

L LMˆ

LMˆ L

L12---

Z L e,v p e,----------

Z L o,v p o,-----------+

= LMˆ 1

2---

Z L e,v p e,----------

Z L o,v p o,-----------–

=

Re Ro+2

-------------------- Re Ro–2

--------------------

Re Ro–2

-------------------- Re Ro+2

--------------------

Re10ln

10----------- αc e, Z L e,⋅ ⋅= Ro

10ln10

----------- αc o, Z L o,⋅ ⋅=

Ge Go+2

--------------------- Ge Go–2

---------------------

Ge Go–2

--------------------- Ge Go+2

---------------------

Ge10ln

10-----------

αc e,Z L e,----------⋅= Go

10ln10

-----------αc o,Z L o,-----------⋅=



Capacitance

Inductance

Resistance

with:

and:

where:

, for a centered stripline s = 0

if then

else

Dispersive effects are introduced according to the value of the geometrical parameters. So theresistivity can be frequency dependent:

You can also introduce frequency dependent conductance by using the parameters Gs and fp(polarization frequency: FP parameter). The conductance dispersive effect can be modeled intwo ways according to the MULTIDEBYE parameter:


Cµ0ε0er

Z0---------------------=

L Z0 µ0ε0er=

RDCrho

w t×------------=

Z0

η0

erC1

ε------×

----------------------=

C1

ε------

2w1 h⁄1 s h⁄– t h⁄–---------------------------------

2w1 h⁄1 s h⁄ t h⁄–+---------------------------------

2π--- 2

1 t h s–( )⁄–------------------------------- 1 1

1 t h s–( )⁄–-------------------------------+ln 1 1

1 t h s–( )⁄–-------------------------------–

1

1 t h s–( )⁄–( )2-------------------------------------- 1–ln+

2π--- 2

1 t h s+( )⁄–------------------------------- 1 1

1 t h s+( )⁄–-------------------------------+ln 1 1

1 t h s+( )⁄–-------------------------------–

1

1 t h s+( )⁄–( )2-------------------------------------- 1–ln+

+ +

+

=

s h 2 h1 t+×( )–=

wh t–----------

0.35< w10.07 h t–( )× w+

1.2------------------------------------------

=

w1 w=

R RDC 1 1 i+( )

frho---------

πµ0t2

--------------×+

×=



where fp is the polarization frequency (FP parameter).




where: and ; C is the user-defined matrix and tandthe dielectric loss tangent parameter.

with and



Reference

(1) Transmission Line Design Handbook, Brian C. Wadell, Artech House 1991.(2) Implementation of Single and Coupled Microstrip Line in APLAC, Luis Costa andMartti Valtonen, CT-33 December 1997.

G dtan C× i2π f 2πfp×i2πf 2πfp×-------------------------------×=


Y jω( ) jω( )Cinf jω( ) f jω( )Cd+=

Cinf C α dCtan–= Cd β dCtan=

α

ω22 ω0

2+

ω12 ω0

2+

-------------------

ln

4πω0

ω1------

atanω0

ω2------

atan–

-----------------------------------------------------------------= β 108( )ln

2πω0

ω1------

atanω0

ω2------

atan–

-----------------------------------------------------------------=

f jω( )

ω2 jω+

ω1 jω+--------------------

ln

108( )ln-------------------------------=

ω0 2π f p= ω1 104= ω2 1012

= ω 2πf=

Transmission LinesLossy Transmission Line: W Model


Lossy Transmission Line: W ModelThe W model is implemented in Eldo to simulate lossy coupled uniform lines includingdispersive effects. This model can be used in all analysis modes (DC, AC, Transient, SST,SSTNOISE, or MODSST). The general instantiation of a W model is shown below. Examplesare provided in the directory $MGC_AMS_HOME/examples/tlines/W_model.

RLGCfile Form

Wxx N=nb_line+ P1...PN PGNDin PN+1...P2N PGNDout+ RLGCfile=file_name L=length [FP=val]+ [MULTIDEBYE=val] [SAVEFIT=val] [COMPAT=val] [FGD=val]

Umodel Form

Wxx N=nb_line+ P1...PN PGNDin PN+1...P2N PGNDout+ Umodel=model_name L=length [SAVEFIT=val]

RLGCmodel Form

Wxx N=nb_line+ P1...PN PGNDin PN+1...P2N PGNDout+ RLGCmodel=model_name L=length [FP=val]+ [MULTIDEBYE=val] [SAVEFIT=val] [COMPAT=val] [FGD=val]

Tabular RLGCmodel Form

Wxx P1...PN PGNDin PN+1...P2N PGNDout+ N=nb_line L=length+ TABLEMODEL=table_model_name [SAVEFIT=val] [FITTABLEMODEL=val]

Parameters

• xx

W model transmission line name.

• N=nb_line

Number of lines.

• P1…PN


• PGNDin

Reference node for the P1…PN nodes of the line system.

• PN+1…P2N




• PGNDout

Reference node for the PN+1…P2N nodes of the line system.

• RLGCfile=file_name

Name of the file containing R, L, C, G, Rs and Gd matrices.

• Umodel=model_name

Name of the transmission line model. This entry allows the use of the U model (see “LossyTransmission Line: U Model” on page 236) entries in the W model.

• TABLEMODEL=table_model_name

Name of the model containing R, L, C and G tabular matrices description.

• L=length

Geometric length of the system (meter). Default value is 1.0. If L=0, Eldo uses the defaultvalue.

• FP=val

Polarization frequency to control dispersive effect on the conductance. Default: 1.6×109.

• MULTIDEBYE=val

val=1 specifies the use of multi-pole debye model to model the dispersive effect on theconductance (recommended for modeling PCB-type dielectrics). val=0, this model is notused. Default: 1.

• SAVEFIT=val

If the value is 1, this option saves the initialization of the transmission line model (in the filecircuit_name.fit), in order to speed up the following simulations of the same netlist. Defaultvalue is 0.

• COMPAT=val

If the value is 1, it specifies the model used for the dispersive effect is based onconductance, see the formula details of each W model instantiation in “RLGC File Syntax”on page 227 and “RLGC Model Syntax” on page 230. Default value is taken from the globaloption COMPAT (if .OPTION COMPAT is specified then COMPAT=1). Note that theMULTIDEBYE parameter priority is higher than COMPAT. If MULTIDEBYE is specified, thenthe value of COMPAT is zero.

• FGD=val

Cut-off frequency value. Default is zero. Can only be specified in compat mode(COMPAT=1).

• FITTABLEMODEL=val

When set to 1, it will enable a causal model (admittance and propagation) to be built fromnon-causal tabulated data. If set to 0 (default), the tabulated data will not be modified tobuild the model and the built-in models are considered to be causals.



Examples

RLGCfile Entry

Circuit name: RLGCfile_example.cir.

W1 N=2+ 1 2 0 3 4 0+ RLGCfile=2lin.rlgc L=0.97e-3

See “Example RLGC file” on page 229.

Umodel Entry

.MODEL unamel U LEVEL=3 ELEV=1 PLEV=1 DLEV=2 NL=1+ HT=1.0e-4 WD=2.0e-4 TH=5.0e-5 RHO=1.785e-8W1 N=1 1 0 2 0 Umodel=uname L=1.0e-3

RLGCmodel Entry

W1 N=2 1 2 0 4 5 0 RLGCmodel=model_rlgc L=0.97e-3

Tabular RLGCmodel Entry

W1 i1 i2 0 o1 o2 0 N=2 L=0.1 TABLEMODEL=ex1

RLGC File SyntaxThe RLGC file is a text file, which contains the values of R, L, C, G, Rs and Gd matrices perunit length. This file is order-dependent, and the order is the following:

N Number of lines.

Lo DC inductance matrix (per unit length).

Co DC capacitance matrix (per unit length).

Ro DC resistance matrix (per unit length).

Go DC conductance matrix (per unit length).

Rs Skin effect resistance matrix (per unit length):

Gd Dielectric-loss conductance matrix (per unit length). The frequencydependent conductance uses the parameters Gs and fp (polarizationfrequency: FP parameter). It can be modeled in two ways according to theMULTIDEBYE parameter:


R Ro 1 i+( ) f Rs+=

G Go

Gd

2π------- i2πf 2πfp×

i2πf 2πfp+------------------------------×+=







where: and ; C and Gs are the user-defined matrices, with:

and



• Compat dispersive model (COMPAT=1)

where fgd is a cut-off frequency; if fgd value is zero then G keeps linear dependency onthe frequency. Default is zero.

The Ro, Go, Rs and Gd matrices are optional (default value is zero). Lo and Co matrices must bedescribed in the RLGC file. Since these matrices are symmetrical, only the lower-triangularparts are specified in the RLGC file.

The diagonal terms of Lo and Co matrices must be positive non-zero; the diagonal terms of Ro,Rs, Go and Gd matrices must be non-negative. Off-diagonal terms of Co, Go and Gd are non-positive.

Comments

A comment line can be specified by an asterisk ‘*’ at the beginning of the line. This commentsout the entire line.


Y jω( ) Go jω( )Cinf jω( ) f jω( )Cd+ +=

Cinf C αGd–= Cd βGd=

α

ω22 ω0

2+

ω12 ω0

2+

-------------------

ln

4πω0

ω1------

atanω0

ω2------

atan–

-----------------------------------------------------------------= β 108( )ln

2πω0

ω1------

atanω0

ω2------

atan–

-----------------------------------------------------------------=

f jω( )

ω2 jω+

ω1 jω+--------------------

ln

108( )ln-------------------------------=

ω0 2π f p= ω1 104= ω2 1012

= ω 2πf=

G Go Gdf

1f

f gd---------

2+

------------------------------×+=



Separator

The number can be separated by any combination of the characters shown in the table below:

Example RLGC file

Filename: 2lin.rlc.

*RLGC matrices for 2 frequency-dependent lines*N (number of lines)********************2* Lo*******0.3481e-60.5458e-7 0.3481e-6* Co*******0.1593e-9-0.2578e-10 0.1651e-9* Ro*******750 50* Go*******0.2421e-3-0.4860e-4 0.2070e-3* Rs*******0.00250 0.0014

Table 9-2. RLGC Separator Characters

Character

Space

Tab

New line

,

;

(

)

[

]



* Gd*******1.2e-13-4.1e-14 1.1e-13

RLGC Model SyntaxThe RLGC model is a model, which contains the values of R, L, C, G, Rs and Gd matrices perunit length. There is no limitation on the number of coupled lines. Since the matrices aresymmetric, only the lower-triangular parts of the matrices have to be described in the RLGCmodel. Inductance and capacitance matrices (Co and Lo) have to be specified, the other matricescan be optional.

General Instantiation of the Model

.MODEL model_name W MODELTYPE=RLGC N=nb_line+ Lo=Lo_matrix_entries Co=Co_matrix_entries+ [Ro=Ro_matrix_entries] [Go=Go_matrix_entries]+ [Rs=Rs_matrix_entries] [Gd=Gd_matrix_entries]

Parameters

• N=nb_line

Number of lines.

• Lo=Lo_matrix_entries

Elements of the DC inductance matrix (per unit length).

• Co=Co_matrix_entries

Elements of the DC capacitance matrix (per unit length).

• Ro=Ro_matrix_entries

Elements of the DC resistance matrix (per unit length).

• Go=Go_matrix_entries

Elements of the DC conductance matrix (per unit length).

• Rs=Rs_matrix_entries

Elements of the skin-effect inductance matrix (per unit length):

• Gd=Gd_matrix_entries

Elements of the dielectric-loss conductance matrix (per unit length). The frequencydependent conductance uses the parameters Gs and fp (polarization frequency: FPparameter). It can be modeled in two ways according to the MULTIDEBYE parameter:


R Ro 1 i+( ) f Rs+=







where: and ; C and Gs are the user-defined matrices. with:

and



• Compat dispersive model (COMPAT=1)

where fgd is a cut-off frequency; if fgd value is zero then G keeps linear dependency onthe frequency. Default is zero.

Example RLGC Model

Circuit name: RLGCmodel_example.cir.

.MODEL model_rlgc W MODELTYPE=RLGC N=2+ Lo = 0.3481e-6+ 0.5458e-7 0.3481e-6+ Co = 0.1593e-9+ -0.2578e-10 0.1651e-9

G Go

Gd

2π------- i2πf 2πfp×

i2πf 2πfp+------------------------------×+=


Y jω( ) Go jω( )Cinf jω( ) f jω( )Cd+ +=

Cinf C αGd–= Cd βGd=

α

ω22 ω0

2+

ω12 ω0

2+

-------------------

ln

4πω0

ω1------

atanω0

ω2------

atan–

-----------------------------------------------------------------= β 108( )ln

2πω0

ω1------

atanω0

ω2------

atan–

-----------------------------------------------------------------=

f jω( )

ω2 jω+

ω1 jω+--------------------

ln

108( )ln-------------------------------=

ω0 2π f p= ω1 104= ω2 1012

= ω 2πf=

G Go Gdf

1f

f gd---------

2+

------------------------------×+=



+ Ro = 75+ 0 50+ Go = 0.2421e-3+ -0.4860e-4 0.2070e-3+ Rs = 0.0025+ 0.0014+ Gd = 1.2e-13+ -4.1e-14 1.1e-13

Tabular RLGC Model SyntaxThe Tabular model is an extension of the RLGC model, which allows to model transmissionline arbitrary frequency-dependent behavior. There is no limitation on the number of coupledlines. Inductance and capacitance tabular matrices (Co and Lo) have to be specified, the othertabular matrices are optional. Each tabular matrix is described in a .MODEL statement.


.MODEL model_name sp W MODELTYPE=TABLE N=nb_line+ LMODEL=L_freq_model CMODEL=C_freq_model+ [RMODEL=R_freq_model] [GMODEL=G_freq_model] [FITTABLEMODEL=val]

Parameters

• N=nb_line

Number of lines.

• LMODEL=L_freq_model

Name of the model containing the sampled values of the inductance matrix.

• CMODEL=C_freq_model

Name of the model containing the sampled values of the capacitance matrix.

• RMODEL=R_freq_model

Name of the model containing the sampled values of the resistance matrix. Default is zero.

• GMODEL=G_freq_model

Name of the model containing the sampled values of the conductance matrix. Default iszero.

• FITTABLEMODEL=val

When set to 1, it will enable a causal model (admittance and propagation) to be built fromnon-causal tabulated data. If set to 0 (default), the tabulated data will not be modified tobuild the model and the built-in models are considered to be causals.

Example Tablemodel

.model ex1 W MODELTYPE=TABLE N=2 LMODEL=lmod1+ CMODEL=cmod1 Rmodel=rmod1 Gmodel=gmod1



Sampled Matrix ModelThis tabular matrix model gives a frequency-varying behavior of R, L, C and G matrices.


.MODEL model_name sp N=nb_line+ SPACING=spacing_type VALTYPE=value_type+ [INFINITY=matrix_values]+ DATA=tabular_matrix_values

Parameters

• model_name

Name of the model.

• N=nb_line

Number of lines.

• SPACING=spacing_type

Data spacing format: only NONUNIFORM type is handled.

• VALTYPE=value_type

Type of matrix elements: only REAL type is handled.

• INFINITY=matrix_values

Data points at infinity.

• DATA=tabuled_matrix_values

Specified frequency value and corresponding matrix data points. As the matrices aresymmetric, only the lower-half portion is described. Syntax: DATA=(sampled_number, f1

data1 f2 data2 ...).

Example Tablemodel

As the model is a “two coupled transmission line”, the dimension of the matrices is 2.Therefore, on each line of the DATA specification, after the sample number, the first value is thefrequency, the second is the (1,1) diagonal value, the third is the (2,1) off-diagonal value, andthe last is the (2,2) diagonal value.

.model cmod1 sp N=2 SPACING=NONUNIFORM VALTYPE=REAL+ DATA=(1,( 6.602360e-11 -7.04724e-12 6.602360e-11))

.MODEL lmod1 sp N=2 SPACING=NONUNIFORM VALTYPE=REAL+ INFINITY=(4.0076e-7 4.6030e-8 4.0076e-7)+ DATA=( 20,+ (0.000000e+00 3.934460e-07 4.6030e-08 3.933460e-07)+ (3.746488e+06 4.151139e-07 4.6030e-08 4.151959e-07)+ (7.726980e+06 4.084730e-07 4.6030e-08 4.085604e-07)+ (1.196411e+07 4.054831e-07 4.6030e-08 4.055730e-07)+ (1.648352e+07 4.037715e-07 4.6030e-08 4.037628e-07)



+ (3.204884e+07 4.008228e-07 4.6030e-08 4.008166e-07)+ (5.911330e+07 3.988513e-07 4.6030e-08 3.988467e-07)+ (7.650809e+07 3.981851e-07 4.6030e-08 3.981811e-07)+ (8.650875e+07 3.978968e-07 4.6030e-08 3.978931e-07)+ (9.756098e+07 3.976313e-07 4.6030e-08 3.976278e-07)+ (1.098398e+08 3.973847e-07 4.6030e-08 3.973813e-07)+ (1.235615e+08 3.971538e-07 4.6030e-08 3.971507e-07)+ (2.962963e+08 3.958050e-07 4.6030e-08 3.958030e-07)+ (3.428571e+08 3.956319e-07 4.6030e-08 3.956300e-07)+ (4.010283e+08 3.954596e-07 4.6030e-08 3.954579e-07)+ (5.753425e+08 3.951106e-07 4.6030e-08 3.951092e-07)+ (7.145791e+08 3.949294e-07 4.6030e-08 3.949281e-07)+ (9.230769e+08 3.947392e-07 4.6030e-08 3.947380e-07)+ (1.269625e+09 3.945339e-07 4.6030e-08 3.945329e-07)+ (4.000000e+09 3.940153e-07 4.6030e-08 3.940147e-07)+ )

.MODEL rmod1 sp N=2 SPACING=NONUNIFORM VALTYPE=REAL+ DATA=( 18,+ (0.000000e+00 8.765530e-01 6.299210e-03 8.765530e-01)+ (3.746488e+06 6.028640e+00 6.299210e-03 6.028640e+00)+ (7.726980e+06 8.270684e+00 6.299210e-03 8.270684e+00)+ (1.196411e+07 1.007694e+01 6.299210e-03 1.007694e+01)+ (2.131439e+07 1.313620e+01 6.299210e-03 1.313620e+01)+ (3.803487e+07 1.727973e+01 6.299210e-03 1.727973e+01)+ (6.741573e+07 2.271433e+01 6.299210e-03 2.271433e+01)+ (7.650809e+07 2.414031e+01 6.299210e-03 2.414031e+01)+ (9.756098e+07 2.714662e+01 6.299210e-03 2.714662e+01)+ (1.098398e+08 2.845071e+01 6.299210e-03 2.845071e+01)+ (1.764706e+08 3.620734e+01 6.299210e-03 3.620734e+01)+ (1.995249e+08 3.844427e+01 6.299210e-03 3.844427e+01)+ (2.264151e+08 4.085570e+01 6.299210e-03 4.085570e+01)+ (3.428571e+08 5.012232e+01 6.299210e-03 5.012232e+01)+ (4.010283e+08 5.413628e+01 6.299210e-03 5.413628e+01)+ (7.145791e+08 7.197077e+01 6.299210e-03 7.197077e+01)+ (1.269625e+09 9.584070e+01 6.299210e-03 9.584070e+01)+ (4.000000e+09 1.690795e+02 6.299210e-03 1.690795e+02)+ )

.MODEL gmod1 sp N=2 SPACING=NONUNIFORM VALTYPE=REAL+ DATA=( 22,+ (0.000000e+00 5.977166e-11 0.000000e+00 5.977166e-11)+ (3.746488e+06 1.451137e-05 -1.821096e-06 1.451043e-05)+ (7.726980e+06 2.992905e-05 -3.755938e-06 2.992712e-05)+ (1.196411e+07 4.634076e-05 -5.815525e-06 4.633777e-05)+ (2.131439e+07 8.245729e-05 -1.036052e-05 8.245196e-05)+ (3.803487e+07 1.473209e-04 -1.848803e-05 1.473114e-04)+ (5.911330e+07 2.289642e-04 -2.873385e-05 2.289494e-04)+ (6.741573e+07 2.611221e-04 -3.276951e-05 2.611062e-04)+ (7.650809e+07 2.963396e-04 -3.718913e-05 2.963205e-04)+ (9.756098e+07 3.778840e-04 -4.742254e-05 3.778596e-04)+ (1.098398e+08 4.253437e-04 -5.339105e-05 4.254163e-04)+ (1.389961e+08 5.383752e-04 -6.756338e-05 5.383404e-04)+ (1.564859e+08 6.061286e-04 -7.606484e-05 6.060795e-04)+ (1.995249e+08 7.728220e-04 -9.698528e-05 7.727721e-04)+ (2.264151e+08 8.769759e-04 -1.100561e-04 8.769193e-04)+ (2.962963e+08 1.146647e-03 -1.440240e-04 1.147553e-03)+ (3.428571e+08 1.327992e-03 -1.666563e-04 1.327906e-03)



+ (4.757709e+08 1.842808e-03 -2.312632e-04 1.842689e-03)+ (5.753425e+08 2.228580e-03 -2.796630e-04 2.228336e-03)+ (9.230769e+08 3.575363e-03 -4.486902e-04 3.575132e-03)+ (1.959184e+09 7.588526e-03 -9.523220e-04 7.588036e-03)+ (4.000000e+09 1.549424e-02 -1.944324e-03 1.549234e-02)+ )

RLGC Model Error Message TreatmentMost of the errors you can meet with this model are the same as for the LDTL model, see“LDTL Model Error Message Treatment” on page 216.

Also, the following error message is displayed if there is a lack of value(s) in the C matrixdescription:

ERROR IN RLGC FILE 2lin.rlc : check matrix C


Transmission LinesLossy Transmission Line: U Model

Lossy Transmission Line: U ModelUxx P1...PN PGNDin PN+1...P2N PGNDout UNAME L=length [SAVEFIT=val]

The U model is implemented in Eldo to simulate lossy-coupled uniform lines. This model canbe used in all analysis modes (DC, AC, Transient, SST, SSTNOISE, or MODSST). Examplesare provided in the directory $MGC_AMS_HOME/examples/tlines/U_model.

Parameters

• xx

Transmission line name.

• P1…PN


• PGNDin

Reference node for the P1…PN nodes of the line system.

• PN+1…P2N


• PGNDout

Reference node for the PN+1…P2N nodes of the line system.

• UNAME

Name of the lossy transmission line model.

• L=length

Geometric length of the system (meter). Default value is 1.0. If L=0, Eldo uses the defaultvalue.

• SAVEFIT=val

If the value is 1, this option saves the initialization of the transmission line model (in the filecircuit_name.fit), in order to speed up the following simulations of the same netlist. Defaultvalue is 0.

Example

Circuit name: Umodel_elev1_example.cir.

U1 1 0 2 0 Umodel L=1.0e-3

Specifies a lossy transmission line U1 between nodes 1 and 2, the reference plane is the ground(node 0). The length of this transmission line is 1.0e-3 and all the parameters are specified in themodel called Umodel (.MODEL U model).



Model Syntax.MODEL UNAME U LEVEL=3 ELEV=elev_val PLEV=plev_val+ [DEV=dlev_val] [LLEV=llev_val] [Param=p_val]

Parameters

• UNAME

Name of the model.

• LEVEL=3

Selects the model of lossy transmission line.

• ELEV=elev_val

Selects the specification format:

ELEV=1 → geometrical description.

ELEV=2 → precomputed model parameters (R, L, C, and G matrices).

ELEV=3 → measured parameters.

• PLEV=plev_val

Selects the type of transmission line: planar structure (PLEV=1), coax (PLEV=2) or twinhead(PLEV=3). Only planar structure is supported.

• DLEV=dlev_val

Specifies the dielectric and ground reference configuration. Two configurations areproposed: microstrip layered dielectric (DLEV=1) and stripline (DLEV=2). Default value is 1.

• LLEV=llev_val

Reference plane inductance consideration (default is 0):

LLEV=0 → omit this inductance.

LLEV=1 → include this inductance (not supported).

• Param=p_val

Specifies parameters of the lines (depends on the specification format).

Geometric Description: ELEV=1

Restriction

Only single line can be described.

Specific Parameters

• DLEV

Type of Line;

DLEV=1 → Microstrip layered dielectric



DLEV=2 → Stripline

• NL

Number of line, default value is 1. (only single line can be described).

• HT

Conductor height, default value is 2.0e-4m.

• WD

Conductor width, default value is 3.0e-4m.

• TH

Conductor thickness, default value is 1.0e-4m.

• KD

Dielectric relative permittivity, default value is 10.0.

• RHO

Conductor resistivity. Default value is 17e-9 Ωm (copper).

Example


.MODEL Umodel U LEVEL=3 ELEV=1 PLEV=1 DLEV=2 NL=1+ HT=1.0e-4 + WD=2.0e-4 TH=5.0e-5 RHO=1.785e-8

This model describes a lossy stripline.

Precomputed Model Parameters: ELEV=2

The precomputed parameters correspond to the R, L, C and G matrices. Since these matrices aresymmetric, only the upper-triangular parts are specified.

Restriction

This description allows the specification of up to five signal conductors.

Specific Parameters

• crj

Self capacitance per unit length (Fm-1). Default value is 1.0e-9.

• cij

Mutual capacitance per unit length (Fm-1). Default value is 0.

• ljj

Self inductance per unit length (Hm-1). Default value is 1.0e-6.



• lij

Mutual inductance per unit length (Hm-1). Default value is 0.

• rjj

Resistance per unit length (Ωm-1). Default value is 0.

• grj

Self conductance per unit length (Sm-1). Default value is 0.

• gij

Mutual conductance per unit length (Sm-1). Default value is 0.

Example


.MODEL Umodel U LEVEL=3 ELEV=2 PLEV=1 r11=34.48+ r22=34.48 + r33=34.48 l11=49.76n l22=49.76n l33=49.76n+ l12=7.65n + l23=7.65n cr1=10.82p cr2=11.24p cr3=10.82p+ c12=-1.97p + c23=-1.97p gr1=0.15u gr2=0.15u gr3=0.15u

This model describes three coupled lossy transmission lines.

Measured Parameters: ELEV=3

This description corresponds to the electrical parameters.

Restriction

Only single line can be described.

Specific Parameters

• ZK

Characteristic impedance (Ω).

• VREL

Relative velocity.

• DELAY

Delay(s) for length DELEN.

• CAPL

Linear capacitance in length CLEN. Default value is 1.

• AT1

Attenuation factor in length ATLEN. Default value is 1.

• DELEN

Unit of length (m) for DELAY. Default value is 1.



• CLEN

Unit of length (m) for CAPL. Default value is 1.

• ATLEN

Unit of length for AT1. Default value is 1.

• FR1

Frequency at which dispersion starts (only affects resistance). If no value is specified, thedispersion will not be taken into account.

In order to discard redundant parameter sets, the following equations are used:

Example


.MODEL Umodel U LEVEL=3 ELEV=3 PLEV=1 ZK=50 DELAY=10n AT1=1

This model describes a single lossy transmission line with a characteristic impedance of 50 Ω.

U Model Error Message TreatmentMost of the errors you can meet with this model are the same as for the LDTL model, see“LDTL Model Error Message Treatment” on page 216.

Table 9-3. Lossy Transmission Line: U Model Parameter Combinations

Input Parameters Computation

ZK, DELAY, DELEN, CAPL and CLEN Redundant, discard CAPL and CLEN

ZK, VREL, CAPL and CLEN Redundant, discard CAPL and CLEN

ZK, DELAY and DLEN

ZK and VREL

ZK, CAPL and CLEN

CAPL, CLEN, DELAY and DELEN

CAPL, CLEN and VREL

VREL DLENDELAY CLIGHT×--------------------------------------------------=

C 1.0ZK VREL CLIGHT××------------------------------------------------------------= L ZK

VREL CLIGHT×---------------------------------------------=

C CAPLCLEN----------------= L C Z K

2×=

VREL DELENDELAY CLIGHT×--------------------------------------------------=

C CAPLCLEN----------------= L 1.0

C VREL2× CLIGH T

2×-------------------------------------------------------------=


Chapter 10Microstrip and Stripline Models

IntroductionA set of microstrip and stripline layout discontinuity structures is provided. This set is targetingRF simulations where a piece of microstrip or stripline discontinuity has to be included. Theavailable set is as follows:

Microstrip Discontinuities:

• MTEE—Microstrip T Junction

• MBEND—Microstrip Bend (Arbitrary Angle, Optimally Mitered)

• MBEND2—90-degree Microstrip Bend (Mitered)

• MBEND3—90-degree Microstrip Bend (Optimally Mitered)

• MCORN—90-degree Microstrip Bend (Unmitered)

• MSTEP—Microstrip Step in Width

• VIA2—Cylindrical Via Hole in Microstrip

Stripline Discontinuities:

• SBEND—Unmitered Stripline Bend

• STEE—Stripline T Junction

• SSTEP—Stripline Step in Width


Microstrip and Stripline ModelsMicrostrip Discontinuity Models

Microstrip Discontinuity ModelsMTEE—Microstrip T Junction

Figure 10-1. Microstrip T Junction

Symbol

Figure 10-2. Microstrip T Junction Symbol

Syntax

Yxx MTEE P1 P2 P3 P4 P5 P6 PARAM: [W1=val] [W2=val] [W3=val]+ [T=val] [Er=val] [H=val]

Parameters

Table 10-1. Microstrip T Junction Parameters

Parameter Definition Default Units

W1 Conductor width of the first arm 2.0e-3 meter

W2 Conductor width of the second arm 2.0e-3 meter

W3 Conductor width of the third arm 3.0e-3 meter

T Conductor thickness 5.0e-6 meter

W3W1

W2

1 2

3

Microstrip and Stripline ModelsMTEE—Microstrip T Junction


Model Validity Range

0.5 ≤ W1/H ≤ 2.0

0.5 ≤ W2/H ≤ 2.0

0.5 ≤ W3/H ≤ 2.0

Simulation Domains

DC, AC, TRANSIENT, and SST

References

Brian C. Wadell, “Transmission Line Design Handbook”, 1991 Artech House.

Notes

The model is based on the microstrip line symmetric T junction equations given in thementioned reference.

The model handles symmetrical T-junction only. If the specified W1 and W2 parameters are notidentical, the geometrical mean of W1 and W2 parameters is computed and used.

for non-symmetrical T-junction

Figure 10-3 illustrates the model equivalent circuit and pins connections:

Figure 10-3. Equivalent circuit Microstrip T Junction

ER Dielectric relative permittivity 4 -

H Dielectric thickness 1.6e-3 meter

Table 10-1. Microstrip T Junction Parameters


W 1 W 2 W 1 W 2⋅= =

LL

CL

P1

P2 P4

P3

P5

P6


Microstrip and Stripline ModelsMTEE—Microstrip T Junction

Example

A simple MTEE s-parameter extraction example over a range of frequencies:

.param w1 = 2.0e-3

.param w2 = 2.0e-3

.param w3 = 3.0e-3

.param t = 5.0e-6

.param Er = 4

.param h = 1.6e-3

.param frequency = 5e9

Ymtee MTEE t1a 0 t1b 0 t2 0 PARAM: W1=w1 W2=w2 W3=w3 T=t Er=Er + H=h

*** S-Parameters Extraction

V1a t1a 0 IPORT=1 RPORT=50 FOUR fund1 PdBm (1) -100 -90V1b t1b 0 IPORT=2 RPORT=50 FOUR fund1 PdBm (1) -100 -90V2 t2 0 IPORT=3 RPORT=50 FOUR fund1 PdBm (1) -100 -90

.step param frequency 1e9 7e9 100e6

.sst fund1=frequency nharm1=1

.extract fsst label=S11_Mag yval(SM(1,1),frequency)









.end

Microstrip and Stripline ModelsMBEND—Microstrip Bend (Arbitrary Angle, Optimally Mitered)


MBEND—Microstrip Bend (Arbitrary Angle, OptimallyMitered)

Figure 10-4. Microstrip Bend (Arbitrary Angle, Optimally Mitered)

Symbol

Figure 10-5. Microstrip Bend (Arbitrary Angle, Optimally Mitered) Symbol

Syntax

Yxx MBEND P1 P2 P3 P4 PARAM: [W=val] [H=val] [Er=val] [T=val]+ [RHO=val] [TAND=val] [M=val] [ANGLE=val]

Parameters

Table 10-2. Microstrip Bend (Arbitrary Angle, Optimally Mitered) Parameters


W Conductor width 2.0e-3 meter



W

1

2W

X

D

Angle




1 ≤ ER ≤ 128

0 < ANGLE < 90

0.01 ≤ W/H ≤ 100

Simulation Domains


References


Equations

The model is equivalent to a transmission line of length:


Figure 10-6. Equivalent circuit Microstrip Bend(Arbitrary Angle, Optimally Mitered)


RHO Conductor resistivity 1.7e-8 Ohm.meter

TAND Dielectric loss tangent 0.0 -

M Optimal mitre percentage 60 %

ANGLE Bend angle 60 degree

Table 10-2. Microstrip Bend (Arbitrary Angle, Optimally Mitered) Parameters


M100 X

d---------------- (%)=

d X– 2 W 1 M100---------–

=

l 2 M100 ANGLE( )sin--------------------------------------------=

P1

P2

P3

P4

TL



Example

A simple MBEND s-parameter extraction example over a range of frequencies:

.param W = 2.0e-3

.param H = 1.6e-3

.param Er = 4

.param T = 5.0e-6

.param RHO = 1.7e-8

.param TAND = 0

.param M = 60

.param angle = 60

.param fx = 5e9

Ymbend MBEND in 0 out 0 PARAM: W=W H=H Er=Er T=T RHO=RHO+ TAND=TAND M=M ANGLE=ANGLE


Vin in 0 IPORT=1 RPORT=50 FOUR fund1 PdBm (1) -100 -90Vout out 0 IPORT=2 RPORT=50 FOUR fund1 PdBm (1) -100 -90

.step param fx 1e9 7e9 100e6

.sst fund1=fx nharm1=1

.extract fsst label=S11_Mag yval(SM(1,1),fx)


.end


Microstrip and Stripline ModelsMBEND2—90-degree Microstrip Bend (Mitered)

MBEND2—90-degree Microstrip Bend (Mitered)

Figure 10-7. 90-degree Microstrip Bend (Mitered)

Symbol

Figure 10-8. 90-degree Microstrip Bend (Mitered) Symbol

Syntax

Yxx MBEND2 P1 P2 P3 P4 PARAM: [H=val] [W=val] [Er=val]

Parameters


0.2 < W/H < 6

2.36 < ER < 10.4

Simulation frequency < 12/H (Frequency in GHz, H in mm)

Table 10-3. 90-degree Microstrip Bend (Mitered) Parameters


H Substrate thickness 1.6e-3 meter


ER Dielectric constant 4 -

W

W 2

1



Simulation Domains


References

M. Kirschning, R. H. Jansen, and N. H. L. Koster. “Measurement and Computer-AidedModeling of Microstrip Discontinuities by an Improved Resonator Method,” 1983 IEEE MTT-S International Microwave Symposium Digest, May 1983, pp.495-497.

Equations

The equivalent circuit of the MBEND2 consists of 2 inductors and a capacitor, shown inFigure 10-9.

Equations used to calculate the equivalent circuit component values:

Notes

The model parameters validity ranges were tested at the corners and some typical design values.

Figure 10-9. Equivalent circuit MBEND2

Example

A simple MBEND2 s-parameter extraction example over a range of frequencies:

.param H = 1.6e-3

.param W = 2.0e-3

.param Er = 4

.param fx = 1e9

Ymbend2 MBEND2 in 0 out 0 PARAM: H=H W=W Er=Er


CH-----

WH----- 7.6 Er 3.8

WH----- 3.93 Er 0.62+( )+ +

pFm

--------=

LH----- 441.2712 1 1.062 0.177

WH-----

0.947–exp–

pF

m--------=

LL

C

P1

P2

P3

P4








.end

Microstrip and Stripline ModelsMBEND3—90-degree Microstrip Bend (Optimally Mitered)


MBEND3—90-degree Microstrip Bend (Optimally Mitered)

Figure 10-10. 90-degree Microstrip Bend (Optimally Mitered)

Symbol

Figure 10-11. 90-degree Microstrip Bend (Optimally Mitered) Symbol

Syntax

Yxx MBEND3 P1 P2 P3 P4 PARAM: [W=val] [H=val] [Er=val] [T=val]+ [RHO=val] [TAND=val]

Parameters

Table 10-4. 90-degree Microstrip Bend (Optimally Mitered) Parameters






RHO Conductor resistivity 1.7e-8 Ohm.meter

D

X

W

W

1

2




0.25 ≤ W/H ≤ 2.75

2.5 ≤ ER ≤ 25

Simulation frequency < 15/h (Frequency in GHz, H in mm)

Simulation Domains


References


Equations

The optimal miter is given by:

and is modeled as a transmission line of length:

The following figure illustrates the model equivalent circuit and pins connections:

Figure 10-12. Equivalent Circuit MBEND3

Example

A simple MBEND3 s-parameter extraction example over a range of frequencies:

.param W = 2.0e-3

.param H = 1.6e-3

.param Er = 4

TAND Dielectric loss tangent 0.0 -

Table 10-4. 90-degree Microstrip Bend (Optimally Mitered) Parameters


M 52 65e1.35

WH-----

–

+100 X

d---------------- (%)= =

L W 1.04 1.3e1.35

WH-----

–

+= m

P1

P2

P3

P4

TL



.param T = 5.0e-6

.param RHO = 1.7e-8

.param TAND = 0

.param fx = 5e9

Ymbend3 MBEND3 in 0 out 0 PARAM: W=W H=H Er=Er T=T RHO=RHO+ TAND=TAND







.end


Microstrip and Stripline ModelsMCORN—90-degree Microstrip Bend (Unmitered)

MCORN—90-degree Microstrip Bend (Unmitered)

Figure 10-13. 90-degree Microstrip Bend (Unmitered)

Symbol

Figure 10-14. 90-degree Microstrip Bend (Unmitered) Symbol

Syntax

Yxx MCORN P1 P2 P3 P4 PARAM: [W=val] [H=val] [Er=val]

Parameters

Table 10-5. 90-degree Microstrip Bend (Unmitered) Parameters




ER Dielectric constant 4 -

1

2

W

W




0.1 ≤ W/H ≤ 6

2 ≤ ER ≤ 15

Simulation Domains


References

M. Kirschning, R. H. Jansen, and N. H. L. Koster. “Measurement and Computer-AidedModeling of Microstrip Discontinuities by an Improved Resonator Method,” 1983 IEEE MTT-S International Microwave Symposium Digest, May 1983, pp. 495-497.

Equations

The equivalent circuit of a microstrip corner is a lumped network of two inductors and acapacitor, as shown in Figure 10-15.

Figure 10-15. Equivalent circuit Microstrip corner

The following equations are used to calculate the values of the model lumped components:

where H is in mm.

Example

A simple MCORN s-parameter extraction example over a range of frequencies:

.param H = 1.6e-3

.param W = 2.0e-3

LL

C

P1

P2

P3

P4

L 1 1.35 e×0.18

WH-----

1.39

–

–

0.2 nH×=

C 10.35 Er 0.25+( ) WH-----

22.6 Er 5.44+( ) W

H-----

+

0.001 H×= pF



.param Er = 4

.param fx = 5e9

Ymcorn MCORN in 0 out 0 PARAM: H=H W=W Er=Er





.extract fsst label=S11_Mag_Eldo yval(SM(1,1),fx)

.extract fsst label=S12_Mag_Eldo yval(SM(1,2),fx)

.end

Microstrip and Stripline ModelsMSTEP—Microstrip Step in Width


MSTEP—Microstrip Step in Width

Figure 10-16. Microstrip Step in Width

Symbol

Figure 10-17. Microstrip Step in Width Symbol

Syntax

Yxx MSTEP P1 P2 P3 P4 PARAM: [W1=val] [W2=val] [ER=val]+ [H=val] [F=val] [ASYMMETRICAL=val] [T=val]

Parameters

Table 10-6. Microstrip Step in Width Parameters


W1 Conductor width at port 1 2.0e-3 meter



ER Relative Dielectric constant 4 -


ASYMMETRICAL1 Selects between symmetrical andasymmetrical step structures

0 -

F Operating frequency 1e9 Hz

Symmetrical Step Asymmetrical Step

W1 W1

W2

W21

1 2

2

P3

P4

P1

P2

MSTEP



Simulation Domains


References


Equations

The equivalent circuit of a microstrip step is a lumped network of an inductor and a capacitor, asshown in Figure 10-18.

Figure 10-18. Equivalent circuit of a microstrip step in width

The model equations to calculate the lumped component values are given in the mentionedreference.

Example

A simple MSTEP s-parameter extraction example over a range of frequencies:

.param W1 = 2.0e-3

.param W2 = 0.5e-3

.param H = 1.6e-3

.param ER = 4

.param Frequency = 1e9

.param T = 5.0e-6

Ymstep MSTEP t1a 0 t1b 0 PARAM: W1=W1 W2=W2 ER=ER H=H F=FrequencyASYMMETRICAL=0 T=T


V1a t1a 0 IPORT=1 RPORT=50 FOUR fund1 PdBm (1) -100 -90V1b t1b 0 IPORT=2 RPORT=50 FOUR fund1 PdBm (1) -100 -90

.step param Frequency 1e9 5e9 100e6

.sst fund1=Frequency nharm1=1

.extract fsst label=S11_Mag yval(SM(1,1),Frequency)

1. ASYMMETRICAL only takes two values, 1 for asymmetrical, and 0 for symmetrical step.

L

C

P1

P2

P3

P4



.extract fsst label=S12_Mag yval(SM(1,2),Frequency)

.end


Microstrip and Stripline ModelsVIA2—Cylindrical Via Hole in Microstrip

VIA2—Cylindrical Via Hole in Microstrip

Figure 10-19. Cylindrical Via Hole in Microstrip

Symbol

Figure 10-20. Cylindrical Via Hole in Microstrip Symbol

Syntax

Yxx VIA2 P1 P2 PARAM: [H=val] [R=val] [COND=val] [T=val] [F=val]

H

R

T



Parameters


100um < H < 635um

0.1 < R/H < 1.5

0 < T < R

Simulation Domains


References

M. Goldfarb and R. Pucel. “Modeling Via Hole Grounds in Microstrip,” IEEE Microwave andGuided Wave Letters, Vol. 1, No. 6, June, pp.135-137.

Notes

The VIA2 is modeled as a series resistor and inductor network. The resistor and inductor valuesare based on equations given in the mentioned reference.


Table 10-7. Cylindrical Via Hole in Microstrip Parameters



R Via radius 100.0e-6 meter

COND Conductor conductivity 58.842e6 1/(Ohm.meter)


F Operating center frequency 1.0e9 Hz



Figure 10-21. Equivalent circuit Cylindrical Via Hole in Microstrip

Example

A simple VIA2 s-parameter extraction example over a range of frequencies:

.param H = 200e-6

.param R = 100e-6

.param COND = 58.824e6

.param T = 5.0e-6

.param fx = 5e9

Yvia2 VIA2 in out PARAM: H=H R=R COND=COND T=T F=fx







.end

P1

P2

L

R

Microstrip and Stripline ModelsStripline Discontinuity Models


Stripline Discontinuity ModelsSBEND—Unmitered Stripline Bend

Figure 10-22. Unmitered Stripline Bend

Symbol

Figure 10-23. Unmitered Stripline Bend Symbol

Syntax

Yxx SBEND P1 P2 P3 P4 PARAM: [W=val] [B=val] [ER=val] [T=val]+ [ANGLE=val] [F=val]

Parameters

Table 10-8. Unmitered Stripline Bend Parameters



B Ground plane spacing 280e-6 meter

T Conductor thickness 17e-6 meter

W

W

21

T1

T2Angle


Microstrip and Stripline ModelsSBEND—Unmitered Stripline Bend

Simulation Domains


References

Altschuler, H.M., and A.A. Oliner, “Discontinuities in the Center Conductor of Symmetric StripTransmission Line,” IRE Transactions on Microwave Theory and Techniques, Vol. MTT-8,May 1960, pp. 328-339 and “Addendum to ‘Discontinuities in the Center Conductor ofSymmetric Strip Transmission Line’,” Vol. MTT-10, No. 2, March 1962, p. 143.

K.C. Gupta, “Computer-Aided Design of Microwave Circuits”, 1981, ARTECH HOUSE, INC.

Arthur A. Oliner, “Equivalent Circuits for Discontinuities in balanced Strip TransmissionLine”, Microwave Theory and Techniques, IEEE Transactions on, Volume: 3 Issue: 2, Mar1955.

Notes

The equivalent circuit of an unmitered stripline bend is a lumped network of two inductors anda capacitor whose values are based on equations given in the mentioned references. Theequivalent circuit is shown in Figure 10-24.

Figure 10-24. Equivalent circuit of an unmitered stripline bend

Example

A simple SBEND s-parameter extraction example over a range of frequencies:

.param W = 0.1e-3

.param B = 280e-6

.param ER = 4.2

ER Relative dielectric constant 4.2 -

ANGLE Bend angle 60 degree

F Simulation frequency 2e9 Hz

Table 10-8. Unmitered Stripline Bend Parameters


LL

C

P1

P2

P3

P4

Microstrip and Stripline ModelsSBEND—Unmitered Stripline Bend


.param T = 17e-6

.param ANGLE = 60

.param F = 2e9

Ysbend SBEND in 0 out 0 PARAM: W=W B=B ER=ER T=T ANGLE=ANGLE F=F



.step param f 1e9 5e9 100e6

.sst fund1=f nharm1=1

.extract fsst label=S11_Mag yval(SM(1,1),f)

.extract fsst label=S12_Mag yval(SM(1,2),f)

.end


Microstrip and Stripline ModelsSTEE—Stripline T Junction

STEE—Stripline T Junction

Figure 10-25. Stripline T Junction

Symbol

Figure 10-26. Stripline T Junction Symbol

Syntax

Yxx STEE P1 P2 P3 P4 P4 P5 PARAM: [W1=val] [W2=val] [W3=val]+ [B=val] [ER=val] [T=val] [F=val]

Parameters

Table 10-9. Stripline T Junction Parameters


W1 First arm conductor width 0.1e-3 meter

W2 Second arm conductor width 0.1e-3 meter

W3 Third arm conductor width 0.2e-3 meter





W3W1

W2

1 2

3



Simulation Domains


References

Altschuler, H.M., and A.A. Oliner, “Discontinuities in the Center Conductor of Symmetric StripTransmission Line,” IRE Transactions on Microwave Theory and Techniques, Vol. MTT-8,May 1960, pp. 328-339 and “Addendum to ‘Discontinuities in the Center Conductor ofSymmetric Strip Transmission Line’,” Vol. MTT-10, No. 2, March 1962, p. 143.

K.C. Gupta, “Computer-Aided Design of Microwave Circuits”, 1981, ARTECH HOUSE, INC.

Arthur A. Oliner, “Equivalent Circuits for Discontinuities in balanced Strip TransmissionLine”, Microwave Theory and Techniques, IEEE Transactions on, Volume: 3 Issue: 2, Mar1955.

Notes

The model is based on the microstrip line symmetric T junction equations given in thementioned references.

The model handles symmetrical T-junction only. If the specified W1 and W2 parameters are notidentical, the geometrical mean of W1 and W2 parameters is computed and used.

for non-symmetrical T-junction

Example

A simple STEE s-parameter extraction example over a range of frequencies:

.param W1 = 0.1e-3

.param W2 = 0.1e-3

.param W3 = 0.2e-3

.param B = 280e-6

.param ER = 4.2

.param T = 17e-6


Ystee STEE t1a 0 t1b 0 t2 0 PARAM: W1=W1 W2=W2 W3=W3 B=B ER=ER T=TF=frequency


V1a t1a 0 IPORT=1 RPORT=50 FOUR fund1 PdBm (1) -100 -90V1b t1b 0 IPORT=2 RPORT=50 FOUR fund1 PdBm (1) -100 -90V2 t2 0 IPORT=3 RPORT=50 FOUR fund1 PdBm (1) -100 -90





W 1 W 2 W 1 W 2⋅= =





.end

Microstrip and Stripline ModelsSSTEP—Stripline Step in Width


SSTEP—Stripline Step in Width

Figure 10-27. Stripline Step in Width

Symbol

Figure 10-28. Stripline Step in Width Symbol

Syntax

Yxx SSTEP P1 P2 P3 P4 PARAM: [W1=val] [W2=val] [B=val] [T=val]+ [ER=val] [F=val]

Parameters

Simulation Domains


References


Table 10-10. Stripline Step in Width Parameters








W2 W12 1

P3

P4

P1

P2

SSTEP


Microstrip and Stripline ModelsSSTEP—Stripline Step in Width

Notes

The SSTEP is modeled as a series inductor. The inductor value is based on equations given inthe mentioned reference.


Figure 10-29. Equivalent circuit for a Stripline Step in Width

Example

A simple SSTEP s-parameter extraction example over a range of frequencies:

.param W1 = 0.10e-3

.param W2 = 0.15e-3

.param B = 280e-6

.param T = 17e-6

.param ER = 4.2


Ysstep SSTEP t1a 0 t1b 0 PARAM: W1=W1 W2=W2 B=B T=T ER=ER F=frequency


V1a t1a 0 IPORT=1 RPORT=50 FOUR fund1 PdBm (1) -100 -90V1b t1b 0 IPORT=2 RPORT=50 FOUR fund1 PdBm (1) -100 -90





.end

L

P1

P2

P3

P4


Chapter 11Behavioral Models

IntroductionCommLib RF (Verilog-AMS and VHDL-AMS) are libraries of behavioral models for RFapplications. The libraries are supplied in the mixed-signal hardware description languagesVerilog-AMS and VHDL-AMS as source code. Each model describes the essentialcharacteristic functionality of the corresponding RF block.

CommLib RF models as supplied can be used for RF system level design as well as forverification. The model sources provide a valuable lesson in behavioral modeling techniques.The sources serve as prototypes that can be customized and extended to fit the designers specialrequirements.

CommLib RF Verilog-AMS can be used in conjunction with Eldo RF and ADMS RF. It iscompatible with the Verilog-A subset supported for SST and MODSST analyses (see Chapter12, “Eldo RF and Verilog-A” for a description).

CommLib RF VHDL-AMS can be used in conjunction with ADMS RF. It is compatible withthe VHDL-AMS subset supported for SST and MODSST analysis (see Chapter 13, “Eldo RFand Questa ADMS” for a description of VHDL-AMS usage with ADMS RF).

The models available in CommLib RF include the following:

• Low Noise Amplifiers

• Power Amplifiers

• Voltage Controlled Oscillators

• Mixers

• Filters

• Functions

For more information on CommLib RF, please refer to the CommLib RF VHDL-AMSLibrary and CommLib RF Verilog-AMS Library documentation.


Behavioral ModelsUsing CommLib RF Verilog-AMS with ADMS RF

Using CommLib RF Verilog-AMS with ADMS RFCommLib RF Verilog-AMS is provided as source code for use with ADMS RF. This allowsyou to modify any model(s) or re-run the compilation for any reason. A script, quickcompile, isprovided to assist you with the generation of the library and compilation of the CommLib RFVerilog-AMS models.

Test ExamplesEldo netlists (.cir extension) which exercise one or more test examples for each part model canbe found in the same directory as the models (i.e.$MGC_AMS_HOME/libraries/commlib_rf/admsRF/verilogams.

Model Instantiation

From an Eldo Netlist

In the netlist you have to add a line defining the model you are going to use as well as thelogical library name where the model exists. Then you can instantiate the model as shown in thefollowing example:

Using CommLib RF Verilog-AMS with Eldo RFVerilog-A

CommLib RF Verilog-AMS is provided in the following way for use with Eldo RF Verilog-A:

• source code which allows users to modify any model(s) or re-run the compilation forany reason. A script is provided to help you compile the CommLib RF Verilog-AMSmodels.

With the source code, you can run the quickcompile script to perform the compilation.

Test ExamplesEldo netlists (.cir extension) which exercise one or more test examples for each part model canbe found in the same directory as the models (i.e.$MGC_AMS_HOME/libraries/commlib_rf/eldoRF/verilogams.

* LNA_1_1_rf Test

.model lna_1_1_rf macro lang=verilog lib=<user_defined_library>

Y_lna lna_1_1_rf in out+ param : Gain=10 IIP3=-5 IIP2=-3 P1dB=-15 NF=5

Behavioral ModelsUsing CommLib RF Verilog-AMS with Eldo RF Verilog-A


Model Instantiation

From an Eldo Netlist

• source codeSource files can be used inside Eldo netlists directly using the .verilog command wherethe source file will be compiled every run and a .ai file is created as in the followingexample.

• compiled library of all the modelsThe required compiled model can be instantiated inside Eldo netlist using the.use_veriloga command which takes the complied file name (.ai file) as an argument asin the following example.

The compiled file (.ai file) is generated when compiling the source file using a vlac commandas follows:

vlac <file_name.va>

* LNA_1_1_rf Test

.verilog <source_file_name> -vlac

.model lna_1_1_rf macro lang=veriloga


* LNA_1_1_rf Test

.use_veriloga <compiled_file_name>



Behavioral ModelsUsing CommLib RF Verilog-AMS with Eldo RF Verilog-A


Chapter 12Eldo RF and Verilog-A

Eldo RF and the Verilog-A CompilerVerilog-A is a language that can be used to help design analog systems with high-levelbehavioral descriptions, as well as structural descriptions of systems and components.

The default Verilog-A implementation in Eldo is the same as that used by Verilog-AMS inQuesta ADMS. Compilation is done in the same way for both products. Prior to the 2009.1release, a separate Eldo Verilog-A compiler (vlac) translated Verilog-A source code into ananalog intermediate file (.ai file) for use by Eldo/Eldo RF for simulation. This flow is stillavailable, with the -vlac command line flag of Eldo, see Running Eldo from the CommandLine in the Eldo User’s Manual.

If a netlist contains a .verilog statement for compilation, the pre-2009.1 vlac compiler will beused (in this case the -vlac option is optional).

Including Verilog-A Modules in an Eldo RFNetlist

For further information on Verilog-A usage in Eldo RF, please refer to the Verilog-A Usage inEldo Simulator in the Eldo Verilog-A User’s Manual. The below is a summary.

Verilog-A modules can be used in an Eldo simulation by any of the following methods:

1. Including a Verilog-A source file that contains the modules in an Eldo netlist with a.verilog statement. See .VERILOG.

2. Including a Verilog-A source file in an Eldo netlist with the .HDL command. See .HDL.

3. Referencing a compiled Verilog-A file that contains the modules in an Eldo netlist witha .use_veriloga statement. The compiled Verilog-A file is the output of the Verilog-Acompiler in a previous compilation, either within Eldo or standalone (pre-2009.1 -vlacflow). The Verilog-A compiler is not invoked for the .use_veriloga statement. See.USE_VERILOGA.

The .model statement can also be used in conjunction with the .hdl, .verilog, and .use_verilogacommands to reference a compiled Verilog-A module in an Eldo netlist. The compiled Verilog-A module is the output of the Verilog-A compiler in a previous compilation, either within Eldoor Questa ADMS. The Verilog-A compiler is not invoked for the .model statement. See.MODEL.


Eldo RF and Verilog-AIncluding Verilog-A Modules in an Eldo RF Netlist

Compiling Verilog-A Source Files

Verilog-A Compiler OptionsThe Verilog-A(MS) (valog) compiler options can be set inside the netlist using option VAOPTS.All the .verilog and .hdl commands will then use the options specified when the Verilog-Acompiler is called. If several .option VAOPTS statements are specified only the last one willbe used.

VA_INCLUDE_PATH VariableThe VA_INCLUDE_PATH environment variable is used by the Verilog-A compiler to find thestandard include files, disciplines.vams, and constants.vams specified on the compiler directive`include. The compiler will search in the directory specified on `include; if the file is not foundit will search the path specified by VA_INCLUDE_PATH. This is a runtime variable (hiddenfrom the environment).

.VERILOGTo incorporate Verilog-A modules (contained in a Verilog-A source file) in an Eldo netlist,specify the source file name in a .verilog statement in the netlist. The .verilog statement is in theform:

.verilog [.verilog-options] <file_name>

where file_name (which may include a relative or absolute pathname) is the name of theVerilog-A source file to be compiled.

.verilog-options depend on the simulator and the flow, and can be:

• for Eldo standalone with the default flowQuesta ADMS valog options can be specified, see the valog command in the QuestaADMS User’s Manual

• for Eldo standalone with the pre-2009.1 (-vlac) flowvlac compiler options -ai and -o can be specified

• for Questa ADMSQuesta ADMS valog options can be specified, see the valog command in the QuestaADMS User’s Manual (specify -oldvams to switch to the pre-2009.1 flow)

Search paths can be specified for Verilog-A source files with the Eldo command line argument-hdlpath. Specify the directory where the source Verilog-A files are located with this argument.For example:

eldo test.cir -hdlpath /lib/veriloga



.HDLThis is an alternative to the .verilog command. To incorporate Verilog-A modules (contained ina Verilog-A source file) in an Eldo netlist, specify the source file name in a .hdl statement in thenetlist. The .hdl statement is in the form:

.HDL FILE=filename [MODULE=module_name] [ALIAS=alias_name]

where filename (which may include a relative or absolute pathname) is the name of theVerilog-A source file to be compiled.

• FILE=filename

Verilog-A file.

• MODULE=module_name

Module name. Optional. If specified, only that module is loaded from the specifiedVerilog-A file.

• ALIAS=alias_name

Specifies an alias instead of the module name defined in the Verilog-A file. Optional. Thiscould be useful if you want to load modules of the same name from different source files.

The .hdl command supports default .va Verilog-A filename extensions. When only a prefix isspecified, .va is assumed as the extension, for example:

.hdl file=myfile Eldo looks for file myfile.va and not myfile

.hdl file=myfile.vams Eldo looks for file myfile.vams

Instantiating Verilog-A ModulesAny compiled Verilog-A modules can now be instantiated and used in an Eldo simulation. Theinstantiation command differs depending on the compilation command used:

• Y instantiation for .verilog or .hdl compilation

• X instantiation for .hdl compilation

Verilog-A modules can also be referenced using .model statements. Referencing the module bya .model statement is not mandatory when the module has been compiled using .hdl or .verilog.For modules compiled outside of these commands you must use .use_veriloga (see“.USE_VERILOGA” on page 278) or a .model (see “.MODEL” on page 279) statements toreference the module.

Modules compiled using the .verilog or .hdl commands can be instantiated by having thefollowing statement in the netlist:



Y<inst_name> module_name+ [PORT:] node_name node_name+ [GENERIC: param=val param=val] | [PARAM: param=val param=val]

where:

• <inst_name> is the name of the instance.

• module_name is the name of the Verilog-A module in the Verilog-A source file.

• node_name corresponds to the terminals or ports of the Verilog-A module.

• param=val overrides the values of parameters param in the module. Parameters arescalars values, the declaration of arrays of parameters is not allowed. The valuesspecified can be of type real or integer. The syntax checking done by Eldo will producean error if necessary.

The instance can then be used in an Eldo simulation, just like any other instances.

Modules compiled using the .hdl command can use the Y instance syntax above or the Eldosubcircuit instantiation syntax:

X<inst_name>+ node_name node_name+ module_name+ [PARAM:] param=val param=val

Referencing Verilog-A Compiled Files

.USE_VERILOGAThis usage of this command differs between the default flow and the pre-2009.1 -vlac flow.

In the default flow, the .use_veriloga command with no additional arguments is supported.

.use_verilog[a]

All the modules of the working library will be loaded. With this feature you can compilemodules outside of Eldo, with the Questa ADMS valog command, and specify a .use_verilogastatement to load all compiled modules. In this case the .model statement is not required.

In the pre-2009.1 -vlac flow, to incorporate Verilog-A modules (contained in a Verilog-Acompiled file and obtained using the vlac command) in an Eldo netlist without recompiling thesource file, specify the .ai file name in a .use_veriloga statement in the netlist. The.use_veriloga statement is in the form:

.use_verilog[a] <file_name>

where <file_name> is the name of the .ai file.



If <file_name> does not use an absolute path, it is searched according to the following rules:

1. -ai <directory> argument set on the Eldo command line

2. -outpath <directory> argument set on the Eldo command line

NoteThe Verilog-A compiler is not invoked for the .use_veriloga statement.

As in the Instantiating Verilog-A Modules, the y<instance_name> ... statement is required forthe Verilog-A modules in the .ai file to be available for simulation.

.MODELTo incorporate Verilog-A modules (contained in a Verilog-A compiled description) in an Eldonetlist use the Eldo .model statement:

.model <module_name> macro lang=veriloga

where:

• module_name is the name of the Verilog-A module in the Verilog-A source file.

NoteLIB=logical_lib_name is not supported in Eldo Verilog-A. Any name specified will beignored. This parameter is required in Questa ADMS to specify the logical library nameto map with the physical library name in order to find the Verilog-A module.

Verilog-A instances can use the same syntax as Eldo built-in devices for models. Syntax:

.model model_name model_type <param=val> param=val

• model_name is the user-defined model name reference. The Verilog-A device instancesmust use this to refer to the .model definition.

• model_type is the Verilog-A module name or module alias (for .HDL only)

• param=val overrides the values of parameters param in the module. param is amodule parameter name.

Every Verilog-A module can have one or more associated model cards. model_type must notconflict with any Eldo built-in model types. Eldo gives priority to its built-in models. Whenmodel_type does not match any of the built-in model types then Eldo attempts to load theVerilog-A modules.

To use this feature the Verilog-A module name (or its alias) must not conflict with the followingbuilt-in devices keywords:



NMOS, PMOS, D, NPN, PNP, NJF, PJF, CORE, R, C, CAP, RES, IND, L, DTOA, ATOD,D2A, A2D, BIDIR, U, W, S, MACRO, NEUR, OPA, TFT, NSOI, PSOI, S3RN13, S3RP13,S3DN13, S3DP13, SOIN1V4, SOIN2V, LPNP, LNPN, MWRN, MWRP, CDN, CDP, RN, RP,PROM, ASGA, MODFAS, DUMMY, FASMOD, NSW, VSWITCH, PSW, PWL, LOGIC,SP, Y.

ExampleThe following code (mixer.vla) shows how an ideal mixer can be implemented and coded inVerilog-A.

// mixer.vla`include <discipline.h>module mixer(rf,lo,out) ; inout rf,lo,out; electrical rf,lo,out,int; parameter real rin_rf=50; parameter real rin_lo=50; parameter real rout=50; parameter real conv_gain=1.0; analog begin I(rf) <+ V(rf)/rin_rf; I(lo) <+ V(lo)/rin_lo; I(out,int) <+ V(out,int)/rout; V(int) <+ conv_gain * V(rf) * V(lo); endendmodule

An example Eldo RF netlist (vlog.cir), which contains a reference to the Verilog-A module isgiven below:

*** Vlog.cir.verilog mixer.vla.model mixer macro lang=veriloga

Vlo lo 0 iport=1 rport=50 four fund1 MA (1) 1.0 -90Vrf rf 0 iport=2 rport=50 four fund2 MA (1) 1.0 0.0

Ymix mixer rf lo out param : conv_gain=4.0Rout out 0 50

.sst fund1=900Meg fund2=910Meg nharm1=3 nharm2=3

.plot fsst VDB(out)

.end

Eldo RF and Verilog-ARestrictions Related to RF Analysis


Restrictions Related to RF AnalysisVerilog-A modules can be used in any Eldo RF analysis (SST, SSTNOISE, SSTAC, SSTXF,MODSST). The Verilog-A language subset supported for RF analysis is described in the tablebelow:

Table 12-1. Verilog-A Restrictions Related to RF Analysis

Status RF Analysis Comments

Analog Operators

ddt() Supported All* -

idt() Partiallysupported

All* idt(x): supported (DC component is 0)idt(x,a): not supportedidt(x,a,assert): not supportedidt(x,a,assert,abstol): not supported

delay() Partiallysupported

All* delay(expr, time_delay): supporteddelay(expr, tdelay max): not supported

transition() Not Supported - -

slew() Not Supported - Will be supported for MODSST in afurther release

laplace_zp() Supported All* -

laplace_zd() Supported All* -

laplace_np() Supported All* -

laplace_nd() Supported All* -

zi_zp() Supported All* -

zi_zd() Supported All* -

zi_np() Supported All* -

zi_nd() Supported All* -

Analog Events

cross() Supported .SST, .MODSST -

timer() Not supported - Will be supported for MODSST in afurther release

Simulation Environment

$realtime() Not supported - Will be supported for MODSST in afurther release

$temperature() Supported All* -

$strobe() Supported All* -


Eldo RF and Verilog-ARF Modeling Examples

*All means: .SST, .SSTAC, .SSTXF, .SSTNOISE and .MODSST.

RF Modeling ExamplesThe Modeling Examples are a set of behavioral VHDL-AMS, Verilog-AMS and Verilog-Amodels, with graded levels of complexity, for communications and multimedia applications.The models are organized in categories or sub-libraries. The categories covered by the modelingexamples are: A/D, D/A, Amplifiers/Comparators, PLL, Delta-Sigma, Filters, Control,Functions, DC-DC, Modulators/Demodulators, Digital, Sources, SerDes, and RF.

The modeling examples cover subsystem models as well as systems basic design units.

For further information on the Verilog-A models, see the Modeling Examples forVerilog-A.

$display() Supported All* -

Analysis Dependent

Analysis() Supported All* -

ac_stim() Not supported - Will be supported for SSTAC in afurther release

white_noise() Supported .SSTNOISE -

flicker_noise() Supported .SSTNOISE -

Noise_table() Supported .SSTNOISE -

Miscellaneous

discontinuity() Supported .MODSST -

bound_step() Supported .MODSST -

last_crossing() Not supported - Will be supported for MODSST in afurther release

final_step() Not supported - Will be supported for MODSST in afurther release

initial_step() Not supported - Will be supported for MODSST in afurther release

Table 12-1. Verilog-A Restrictions Related to RF Analysis

Status RF Analysis Comments


Chapter 13Eldo RF and Questa ADMS

IntroductionNoteA Questa ADMS RF license is required to use the RF algorithms in Questa ADMS.

Many digital communication systems include tightly integrated RF, analog mixed-signal andDSP functions. RF carriers severely slow classical mixed-signal transient simulation. ADMSRF specifically targets this challenge by combining the mixed-signal and RF capabilities ofQuesta ADMS and Eldo RF.

Such systems integrate an RF front-end together with baseband complex digital signalprocessing. Verifying such systems such as direct conversion receivers or automatic gaincontrol loops mandates simulation solutions where the transistor-level RF part can be simulatedsimultaneously with the baseband part, and doing so in a practical CPU time.

ADMS RF supports descriptions using any mix of SPICE (Eldo), Verilog-A, Verilog, VHDLand VHDL-AMS. It uses advanced mixed time-frequency algorithms which computes a time-varying spectrum. The spacing of time points is chosen to follow the slow-varying basebandinformation, rather than the fast-varying RF carriers. These algorithms have the same accuracyas regular but slow circuit-level transient simulation.

This results in huge speedup ratios over regular transient simulation, without compromisingaccuracy. Two or three orders of magnitude are usual figures with the typical baseband-to-carrier frequency ratios in wireless networks (WLAN) applications for example.

ADMS RF is able to handle RF, analog and digital functions in the same simulation, as well asmixing modulated, analog, digital signals, and mixing different abstraction levels.Parameterized built-in converters are automatically inserted between different types ofdescriptions. ADMS RF combines the capability of Questa ADMS with the steady-state basedanalyses (.SST, .SSTNOISE, .SSTAC, .SSTXF) and the modulated steady-state (.MODSST)analysis of Eldo RF.

The simulator commands are completely compatible with the current simulators (Questa ADMSand Eldo RF). The Questa ADMS graphical user interface is available for interactivelycontrolling the simulation session. The modulated steady-state analysis (MODSST) has to berequested in the simulator command file as well as some specific commands related to thisanalysis (such as plot, extract commands).


Eldo RF and Questa ADMSIntroduction

Details about using Questa ADMS can be found in the Questa ADMS User’s Manual. Foran introduction to ADMS RF, using an example for which the tool brings real benefits interms of performance and usability, please refer to the ADMS RF Tutorial—AGC Loop.

Restrictions when using Eldo RF with Questa ADMS• Only the CMD-file (Spice syntax) will handle plotting, probing or listing waves during

simulation. The Questa ADMS Tcl add wave/log/list commands, and the Net windowmenus have not yet been extended to allow this.

NoteThe vacom VHDL-AMS compiler will issue a warning message when the value of aquantity is used in the condition of an if...use statement. For ADMS RF, the message maybe safely ignored. In particular, do not use the 'above attribute in ADMS RF as themessage suggests. Use of 'above will yield incorrect results without further warning inADMS RF.

Circuit Partitioning

MODSST Circuit PartitioningEldo RF and ADMS RF have the capability to use simultaneously the transient algorithm andthe MODSST (MODulated Steady-StaTe) algorithm. Transient and MODSST algorithms areused selectively for specified subcircuit instances. In order to use this feature, you must specifywhich instances have to be simulated using the MODSST algorithm. The .PART commandinstructs Eldo to use the MODSST algorithm in place of the regular transient algorithm, for acertain selection of instances.

For further details, see “.PART MODSST” on page 64.

Steady-State Circuit PartitioningEldo RF and ADMS RF have the capability to partition a netlist into individual partitions,compute the steady-state for each partition, then perform the steady-state analysis for thecomplete netlist to find the steady-state solution. You should list all devices that have the samefunctionality and frequency spectrum in the same SST partition, this will help convergence andreduce the simulation time.

For further details, see “.RFBLOCK” on page 67.

Eldo RF and Questa ADMSUsing Eldo RF from the Questa ADMS GUI


RF Modeling ExamplesThe Modeling Examples are a set of behavioral VHDL-AMS, Verilog-AMS and Verilog-Amodels, with graded levels of complexity, for communications and multimedia applications.The models are organized in categories or sub-libraries. The categories covered by the modelingexamples are: A/D, D/A, Amplifiers/Comparators, PLL, Delta-Sigma, Filters, Control,Functions, DC-DC, Modulators/Demodulators, Digital, Sources, SerDes, and RF.

The modeling examples cover subsystem models as well as systems basic design units.

For further information, see the Modeling Examples for VHDL-AMS andModeling Examples for Verilog-A.

Using Eldo RF from the Questa ADMS GUIAfter invoking the Questa ADMS GUI with the vasim command (without any design unitspecified), or Simulate > Start Simulation if Questa ADMS is already running, the LoadDesign dialog will be displayed, see Load Design Dialog in the Questa ADMS User’s Manual.

• Select New in the Design tab to open the Eldo Commands dialog. This allows you tospecify Eldo and Eldo RF equivalent commands.

Eldo Commands DialogIn order to generate the command file for RF simulation, the MODSST simulation analysis isavailable in the Eldo Commands dialog. When editing the CMD-File window, this MODSSTanalysis can be specified as shown in Figure 13-1.



Figure 13-1. Selecting MODSST Analysis in the Questa ADMS GUI

Simulation time The Modulated Steady-State analysis duration in seconds.Related Eldo RF parameter: MODSST_TSTOP.

Print period The printing or plotting increment for the printer output (inseconds). Related Eldo RF parameter: MODSST_TPRINT.

A .MODSST analysis must be specified in conjunction with a .SST command. Selecting theModSST radio button in the Eldo Commands dialog opens a new dialog (Figure 13-2) forspecifying the different harmonic data.



Figure 13-2. Modulated Steady State Dialog in the Questa ADMS GUI

• Click Add or Edit to open another dialog (Figure 13-3), which allows the values for anyharmonic to be added or changed.

Figure 13-3. Fundamental Frequency Specification Dialog in theQuesta ADMS GUI

Index Specifies the index of the fundamental frequencies andcorresponding harmonics. The number should range from 1 tothe total number of fundamental frequencies. Related Eldo RFparameter: xx.

Fund Freq value Specifies the value of the fundamental frequency. RelatedEldo RF parameter: FUND.


Eldo RF and Questa ADMSVHDL-AMS RF Subset

Number of Harmonics Specifies the maximum number of harmonics correspondingto the fundamental frequency. Related Eldo RF parameter:NHARM.

• After setting the required values, click OK to return to the Modulated Steady Statedialog, then click Close in that dialog to return to the Eldo Commands dialog.

• Click Save to save the command file, and return to the Load Design dialog.

• Click Load to load the design with this command file.

VHDL-AMS RF SubsetThe following table characterizes the language supported by ADMS RF by comparing it to theVHDL-AMS language subset supported in Questa ADMS.

Table 13-1. VHDL-AMS RF Subset Support

Description Supported?

Design entities and configurations

Entity declarations Supported as VHDL-AMS subset

Architecture bodies Supported as VHDL-AMS subset

Configuration declarations Supported as VHDL-AMS subset

Subprograms and packages

Subprogram declarations Supported as VHDL-AMS subset

Subprogram bodies Supported as VHDL-AMS subset

Subprogram overloading Supported as VHDL-AMS subset

Resolution functions Supported as VHDL-AMS subset

Package declarations Supported as VHDL-AMS subset

Package bodies Supported as VHDL-AMS subset

Conformance rules Supported as VHDL-AMS subset

Types and natures

Scalar types Supported as VHDL-AMS subset

Composite types Not implemented for pure RF quantities

Access types Not implemented

File types Supported as VHDL-AMS subset

Natures Not implemented for RF analysis, record is notimplemented (only complex types for QuestaADMS)



Declarations

Type declarations Supported as VHDL-AMS subset

Subtype declarations Supported as VHDL-AMS subset

Objects Supported as VHDL-AMS subset (aliases aresupported only on the following objects: file,variable, constants, quantity, signal, terminal)

'ABOVE Supported

'DELAYED Supported

'STABLE Supported

'QUIET Supported

'TRANSACTION Supported

'DOT Supported

'INTEG Supported

'ZOH Not supported in RF simulation

'LTF Supported

'ZTF Supported

'REFERENCE Supported

'CONTRIBUTION Not implemented

'RAMP Supported in MODSST analysis. Can only beapplied to a baseband signal or quantity

'SLEW Supported in MODSST analysis. Can only beapplied to a baseband signal or quantity

Attribute declarations Supported as VHDL-AMS subset

Component declarations Supported as VHDL-AMS subset

Group template declarations Not implemented

Group declarations Not implemented

Nature declaration Supported as VHDL-AMS subset

Specifications

Attribute specification Supported as VHDL-AMS subset

Configuration specification Supported as VHDL-AMS subset

Disconnection specification Not implemented





Step limit specification Not implemented

Names

Simple names Supported as VHDL-AMS subset

Selected names Supported as VHDL-AMS subset

Indexed names Supported as VHDL-AMS subset

Slice names Not supported in RF simulation

Attribute names Supported

'LEFT Supported

'DELAYED Supported

Expressions

Rules for expressions

Operators All are supported except ABS

Operands Supported as VHDL-AMS subset

Static expressions Supported as VHDL-AMS subset

Universal expressions Supported as VHDL-AMS subset

Linear Forms Supported as VHDL-AMS subset

Sequential statements

Wait statement Supported as VHDL-AMS subset

Assertion statement Supported as VHDL-AMS subset

Report statement Supported as VHDL-AMS subset

Signal assignment statement Supported as VHDL-AMS subset

Variable assignment statement Supported as VHDL-AMS subset

Procedure call statement Supported as VHDL-AMS subset

If statement Supported as VHDL-AMS subset

Case statement Supported as VHDL-AMS subset

Loop statement Supported as VHDL-AMS subset

Next statement Supported as VHDL-AMS subset

Exit statement Supported as VHDL-AMS subset

Return statement Supported only in a function

Null statement Supported as VHDL-AMS subset





Notes

• All digital statements are supported only in Questa ADMS, but are not considered in RFsimulation. Digital statement are considered differently, according to the analysis, asfollows:

o For any RF simulation, the digital part is used for computing the quiescent point(VHDL-AMS equivalent to DC), anywhere it is in the design.

o For the MODSST analysis, the digital part is computed normally when part of thebaseband.

o If an RF analysis is requested after a transient, the digital part is computed normallyuntil the last transient point.

For the latter two points, the last digital computed value (either DC or last transientpoint) is used as a parameter in the RF part if requested by the description.

• Complex are not supported in RF.

• Records are not supported in RF.

Break statement Supported as VHDL-AMS subset




Chapter 14Eldo RF Tutorials

IntroductionA summary of the circuits used in this chapter, together with a brief description of the Eldo RFsubject areas dealt with are listed in the table below. These circuits are available in thedirectory: $MGC_AMS_HOME/examples/rfic/

Table 14-1. Tutorials Summary

TutorialNo.

Circuit Name Eldo Description

1 ampli.cir Single-tone Steady-State analysis and Two-port NoiseExtraction

2 1dB_PAE.cir Power Efficiency and 1dB Compression Point Extraction

3 IM3.cir Multi-tone analysis: IM3 and IP3 Extraction

4 Sparameter_1tone.cir S-parameter Extraction

51 Gilbert_Cell.Noise.cir Mixer Steady-State and Noise Analysis

6 Gilbert_Cell.Stac.cir Mixer Steady-State AC Analysis

7 Gilbert_Cell.Stxf.cir Mixer Steady-State TF Analysis

8 div2_4.cir Frequency Divider by 4 Steady-State Analysis

9 osc_sst.cir Single-tone Autonomous Steady-State Analysis

10 vco_step.cir Voltage Controlled Oscillator

11 vco_phnoise.cir Phase Noise Extraction

12 dig_mod_source.cir Digitally Modulated Sources

13 ampli_ACPR.cir ACPR Computations

14 ampli_va.cir Verilog-A Usage

15 ampli_NPR_sst.cir NPR Computation with Steady-State Analysis

16 ampli_NPR_modsst.cir NPR Computation with Modulated Steady-StateAnalysis

17 mqam_evm.cir EVM and BER Computations

18 Contours.cir Load Pull Contours


Eldo RF TutorialsTutorial #1—Single-Tone Steady-State Analysis and Two-Port Noise Extraction for an Amplifier

Tutorial #1—Single-Tone Steady-State Analysisand Two-Port Noise Extraction for an Amplifier

This simple circuit simulation gives a general introduction to the syntax of Eldo RF byperforming a single-tone steady-state analysis and two-port noise extraction on an amplifier.

Figure 14-1. Amplifier Circuit

Eldo RF Features Used

• Input source specified in terms of Power

• Steady-State analysis definition (.SST)

• Plot output results (.PLOT TSST, .PLOT FSST)

• Total Harmonic Distortion

Complete Netlist

.include ampli.cktx1 in load vdd AMPLI_CKTVdc vdd 0 3.3Vout load 0 rport=50 iport=2Vin in 0 RPORT=50 iport=1 FOUR fund1 PdBm (1) Pin -90.param Pin=-20* Steady-state analysis definition and plots

19 lnamixer_im3.cir SST simulation using the .RFBLOCK command

20 lnamix.cir SST simulation using the .SSTNLCONTRIB command

21 vcomixerlna_S_parameter.cir Multitone Large Signal S Parameters Extraction

1. You must run Gilbert_Cell.cir to obtain the expected Gilbert_Cell.sst that is used in the .restart inGilbert_Cell.Noise.cir.

Table 14-1. Tutorials Summary

TutorialNo.

Circuit Name Eldo Description

RLVin

RPORTLOAD

VDD

IN

Pin



.sst fund1=900MegaHerz nharm1=10* Plots.plot tsst v(load).plot fsst vdb(load)* Total harmonic distortion extraction.extract fsst label=THD disto(vdb(load), fund1, 10Meg, 10G)* Noise Analysis for Two-port extractions.sstnoise v(load) lin 10 750Meg 1050Meg.snf input=Vin output=Vout.extract sstnoise label=SNF@750MegHz yval(snf, 750Meg).extract sstnoise label=BOPT@750MegHz yval(bopt, 750Meg).extract sstnoise label=GOPT@750MegHz yval(gopt, 750Meg).extract sstnoise label=GAMMA_OPT@750MegHz+ yval(gamma_opt_mag, 750Meg).extract sstnoise label=PHI_OPT@750MegHz+ yval(phi_opt, 750Meg).plot sstnoise snf nfmin rneq.plot sstnoise gopt bopt gamma_opt_mag phi_opt* Noise Circles Plot into a Smith Chart.plot sstnoise nc (smith).end

Netlist Explanation

.include ampli.cktx1 in load vdd AMPLI_CKTVdc vdd 0 3.3Vout load 0 rport=50 iport=2Vin in 0 RPORT=50 iport=1 FOUR fund1 PdBm (1) Pin -90.param Pin=-20

The above lines define the circuit description and the input source. There are two points tounderline about the source description:

• The source is of the Fourier type:

FOUR fund1 PdBm (1) Pin -90

To specify that the source is multi-tone, with the fundamental frequencies defined afterthe keyword FOUR (only one tone in this case, fund1).

• The source, described here as a voltage source with an internal resistor RPORT, isspecified directly in terms of Power, in dBm format. This is the one that is used mostoften in RF design for specifying the input supply. The power value at fund1 is Pin(set to -20 dBm through the .param command), with an angle of -90 (to generate asinusoidal waveform, as a cosinusoidal waveform is the default). fund1 value isdefined in the .sst command (see below).

iport=1 means that port number 1 is being used here. In this case, it is meaningless as there isonly one port, however different numbers are used mainly for s-parameter extraction.

The definition of the source as a Power allows easy definition in units used in RF design, andavoids a more complex definition in Voltage. The equivalence for the input source with voltageunits would have been:



RPORTIN vn1 IN z0VIN vn1 0 four fund1 MA (1) vamp -90.param vamp=sqrt(8*z0*1m*exp(log(10)*Pin/10)).param z0=50* Steady-State analysis definition.sst fund1=900MegaHz nharm1=10

The steady-state analysis is activated through the command .sst.

The fundamental frequency fund1 is set to 900 MHz. We want to compute a steady-stateanalysis with 10 harmonics, so nharm1 parameter is set to 10. This implies that the steady-statefrequency spectrum will have components at 900, 1800, ... 9000 MHz, plus DC.

* Plots.plot tsst v(load).plot fsst vdb(load)

The steady-state analysis results are plotted in time and frequency domain.

The syntax for the plots are similar to the one for time domain and frequency domain, exceptthat the keywords tran and ac are respectively replaced with tsst and fsst.

* Total harmonic distortion extraction.extract fsst label=THD disto(vdb(load), fund1, 10Meg, 10G)

The total harmonic distortion (THD) is the ratio between the power at fund1 and the sum ofthe powers from the other harmonics.

It can be computed with EZwave, or with a .extract command using the disto function.The fundamental frequency has to be specified with the frequency window [fmin, fmax] onwhich the THD is measured.

* Noise Analysis for Two-port extractions.sstnoise v(load) lin 10 750Meg 1050Meg.snf input=Vin output=Vout.extract sstnoise label=SNF@750MegHz yval(snf, 750Meg).extract sstnoise label=BOPT@750MegHz yval(bopt, 750Meg).extract sstnoise label=GOPT@750MegHz yval(gopt, 750Meg).extract sstnoise label=GAMMA_OPT@750MegHz+ yval(gamma_opt_mag, 750Meg).extract sstnoise label=PHI_OPT@750MegHz+ yval(phi_opt, 750Meg).plot sstnoise snf nfmin rneq.plot sstnoise gopt bopt gamma_opt_mag phi_opt* Noise Circles Plot into a Smith Chart.plot sstnoise nc (smith)

These commands allow you to extract and plot two port noise parameters (NFMIN, GOPT,BOPT, RNEQ, GAMMA_OPT_MAG and PHI_OPT) and noise figure (SNF). Additionally,noise circles can be plotted in a Smith Chart.



See “Two-Port Noise Parameters” on page 108 for further information.

Simulation Results

The first waveform (Figure 14-2) represents the frequency spectrum with 10 harmonics(specified with nharm1=10 in the .sst command). The second curve (Figure 14-3) is the timedomain waveform recreated on two cycles from the steady-state point. The third and fourthwaveforms (Figure 14-4) represent the Two-port parameter values. Finally, a Smith chart(Figure 14-5) is displayed which shows the noise circles.

The simulation performed in less than 1 second, and the THD value is -62dB, that corresponds to0.07%.

Figure 14-2. Single-tone Steady-State Analysis and Two-port NoiseExtraction Simulation Results 1

Eldo RF TutorialsTutorial #2—Power Efficiency and 1dB Compression Point Extraction for an Amplifier


Tutorial #2—Power Efficiency and 1dBCompression Point Extraction for an Amplifier

The same amplifier is used in this example as in Tutorial #1—Single-Tone Steady-StateAnalysis and Two-Port Noise Extraction for an Amplifier. Therefore the original netlist isidentical, but commands are added in order to compute the power efficiency, power addedefficiency and 1dB compression point.



• Parametric Steady-State analysis

• Power efficiency and power added efficiency computation

• 1 dB compression point extraction

Complete Netlist

.include ampli.cktRL LOAD 0 50Vdc VDD 0 3.3XCKT IN LOAD VDD AMPLI_CKTVin IN 0 RPORT=50 iport=1 FOUR fund1 PdBm (1) Pin -90* Power sweep.param Pin=-20.step param Pin -20 10 0.5* Steady-State analysis definition.sst fund1=900MegaHz nharm1=10.option sst_max_liniter=100* Plots.plot tsst v(load).plot fsst vdb(load)* Power efficiency: Pout at fund1 / Pdc.extract fsst label=PE YVAL(Pm(RL), fund1)/YVAL(Pm(Vdc), 0).extract MAX(meas(PE))* Power added efficiency: (Pout-Pin) at fund1 / Pdc.extract fsst label=PAE (YVAL(Pm(RL), fund1) -+ YVAL(Pm(Vin), fund1))/YVAL(Pm(Vdc), 0).extract MAX(meas(PAE))

RLVin

RPORTLOAD

VDD

IN

Pin



* Gain.extract fsst label=Gain YVAL(PdBm(RL), fund1) -+ YVAL(PdBm(Vin), fund1)* 1dB compression point.extract fsst label=POUTdBm YVAL(PdBm(RL), fund1).extract fsst label=PINdBm Yval(PdBm(Vin), fund1).extract sweep label=IP1dB+ yval(meas(PINdBm), xcompress(meas(POUTdBm), 1.0)).extract sweep label=OP1dB compress(meas(POUTdBm), 1.0).end

Netlist Explanation

.include ampli.cktRL LOAD 0 50Vdc VDD 0 3.3XCKT IN LOAD VDD AMPLI_CKTVin IN 0 RPORT=50 iport=1 FOUR fund1 PdBm (1) Pin -90* Power sweep.param Pin=-20.step param Pin -20 10 0.5* Steady-State analysis definition.sst fund1=900MegaHz nharm1=10.option sst_max_liniter=100* Plots.plot tsst v(load).plot fsst vdb(load)

For the definition of the input source, the steady-state analysis and the plots, see Tutorial #1—Single-Tone Steady-State Analysis and Two-Port Noise Extraction for an Amplifier.

The input power is swept from -20 dBm to 10 dBm with an increment of 0.5 in order to performa parametric steady-state analysis. It is mandatory for measuring the power efficiency and the1dB compression point, which are obtained from the characteristics of the output powerfunction of the input power Pin.

Different options related to the steady-state analysis can be defined. The circuit simulated ishighly non-linear for higher values of the input power, so the maximum number of iterations ofthe linear solver set by default to 20 might not be enough. This parameter can be modifiedthrough the option: sst_max_liniter.

* Power efficiency: Pout at fund1 / Pdc.extract fsst label=PE YVAL(Pm(RL), fund1)/YVAL(Pm(Vdc), 0).extract max(meas(PE))

The power efficiency is the ratio between the output power Pout and the DC power Pdc. It iscomputed with a .extract fsst command, that is similar to a .extract ac except that thekeyword ac is replaced with fsst. This extraction is performed for each Pin value that is sweptduring the parametric analysis. The yval function returns the value of the power in magnitudePm at the specified frequency (fund1 for the power in the load RL and 0 for the DC power).



The maximum value of the power efficiency on all the parametric analyses is computed with anextract using the max function. As PE is a label, we cannot use it directly, only with the measurefunction meas.

* Power added efficiency: (Pout-Pin) at fund1 / Pdc.extract fsst label=PAE (YVAL(Pm(RL), fund1) -+ YVAL(Pm(Vin), fund1))/YVAL(Pm(Vdc), 0).extract MAX(meas(PAE))

The power added efficiency and its maximum on all the parametric runs is extracted exactly inthe same way as the power efficiency, except that its expression is slightly different: PAE =(Pout - Pin) / Pdc

* Gain.extract fsst label=Gain YVAL(PdBm(RL), fund1) -+ YVAL(PdBm(Vin), fund1)

The gain is the ratio between the output power and the input power when in Magnitude units, orthe difference between the values if we are in dB or dBm.

* 1dB compression point.extract fsst label=POUTdBm YVAL(PdBm(RL), fund1).extract fsst label=PINdBm Yval(PdBm(Vin), fund1).extract sweep label=IP1dB yval(meas(PINdBm),+ xcompress(meas(POUTdBm), 1.0)).extract sweep label=OP1dB compress(meas(POUTdBm), 1.0)

The 1dB compression point is the point where an increase in the input amplitude results in anoutput that is 1dB lower than that the ideal projection if the amplifier had no non-linearity. It isobtained from the characteristic Pout=f(Pin) at fund1 for a sweep on Pin, given here by thePoutdBm extracted waveform. It characterizes the non-linearity of the circuit.

There are two functions especially defined to measure this 1dB compression point, these are,xcompress for the x-axis value (Pin) and compress for the y-axis value (Pout). The syntax isgiven above with a .extract sweep applied on these functions.

NoteThe IP1dB computation using only the xcompress function (.extract sweeplabel=IP1dB xcompress(meas(POUTdBm), 1.0)) would return the value of Pin(parameter swept during the simulation) that differs a small amount from the actualpower delivered by the input supply Vin, except at the adaptation. Therefore, we extractthe IP1dB at the value of Vin (meas(PINdBm)) corresponding to the Pin value returnedby the xcompress function.

To view the following figure:

• Open the 1dB_PAE.ext file inside EZwave.

• Expand the EXT file shown in the Waveform List window.



• Then drag GAIN, PAE, PE, and POUTDBM individually into the Wave window.

The results plotted in Figure 14-7 have been generated from the .extract commands with theinput power Pin in the x-axis.

The other results from the .extract commands are given below:

MAX(MEAS(PE)) = 4.0424E-01 (or 40.4%)MAX(MEAS(PAE)) = 3.9544E-01 (or 39.5%)MAX(MEAS(GAIN)) = 2.8128E+01 (or 28.1dB)

for the power efficiency and power added efficiency maxima, that can be measured on theEZwave waveforms.

IP1DB = -1.1621E+01 (in dBm)OP1DB = 1.4277E+01 (in dBm)

for the coordinates of the 1dB compression point.

The simulation performed in less than 1 minute.



Simulation Results

Figure 14-7. Power Efficiency and 1 dB Compression Point ExtractionSimulation Results


Eldo RF TutorialsTutorial #3—Multi-Tone Analysis, and IM3 and IP3 Extraction for an Amplifier

Tutorial #3—Multi-Tone Analysis, and IM3 andIP3 Extraction for an Amplifier

The same amplifier as Tutorial #1—Single-Tone Steady-State Analysis and Two-Port NoiseExtraction for an Amplifier and Tutorial #2—Power Efficiency and 1dB Compression PointExtraction for an Amplifier is used here. The netlist is similar, but two tones are introduced inthe input to allow computation of IM3 and IP3.



• Multi-Tone Steady-State analysis

• Third order intermodulation (IM3) computation

• Third order intercept point (IP3) extraction

Complete Netlist

.include ampli.cktRL LOAD 0 50Vdc VDD 0 3.3XCKT IN LOAD VDD AMPLI_CKTVin IN 0 RPORT=50 iport=1 FOUR fund1 fund2 PdBm (1, 0)+ pin -90 (0, 1) pin -90.param pin=-20.param fund_1=900Meg fund_2=901Meg* Steady-State analysis definition.sst fund1=fund_1 nharm1=5 fund2=fund_2 nharm2=5.option sst_spectrum=1* Plots.plot fsst vdb(in) vdb(load).plot fsst PdBm(vin).plot fsst PdBm(RL).plot tsst v(LOAD)* functions for direct IIP3-OIP3 extracts.extract fsst label=IIP3_ref+ iipx(PdBm(Vin), PdBm(RL), fund_1, 2*fund1-fund2).extract fsst label=IIP3_ref+ oipx(PdBm(RL), fund1, 2*fund1-fund2)

RLVin

RPORTLOAD

VDD

IN

Pin



* definition of the third intermodulation.extract fsst label=IM3+ (yval(PdBm(RL), fund1) - yval(PdBm(RL), 2*fund1-fund2))* definition of the third order intercept point.extract fsst label=PINdBm YVAL(PdBm(Vin), fund_1).extract fsst label=IIP3 0.5*meas(IM3) + meas(PindBm).extract fsst label=OIP3+ 0.5*meas(IM3) + yval(PdBm(RL), fund1).end

Netlist Explanation

Vin IN 0 RPORT=50 iport=1 FOUR fund1 fund2 PdBm (1, 0)+ pin -90 (0, 1) pin -90.param pin=-20.param fund_1=900Meg fund_2=901Meg* Steady-State analysis definition.sst fund1=fund_1 nharm1=5 fund2=fund_2 nharm2=5.option sst_spectrum=1* Plots.plot fsst vdb(in) vdb(load).plot fsst PdBm(vin).plot fsst PdBm(RL).plot tsst v(LOAD)

The input source Vin is described here as a two tone source, with two fundamental frequenciesfund1 and fund2 (see Tutorial #1—Single-Tone Steady-State Analysis and Two-Port NoiseExtraction for an Amplifier for more details). The syntax (1,0) or (0,1) defines the index of theharmonics according to fund1 and fund2. Here both fund1 and fund2 are definedindependently, with an input power in dBm of pin and an angle of -90.

fund1 and fund2 values are defined in the .sst command description, the two-tonesteady-state analysis will be performed with five harmonics on both fund1 and fund2

(nharm1=5 and nharm2=5).

The option sst_spectrum defines the spectrum truncation. When set to one, a box truncation isused, that will take sums and differences of the harmonics. The default is the diamondtruncation that will eliminate the higher order combinations.

* definition of the third intermodulation.extract fsst label=IM3+ (yval(PdBm(RL), fund_1) - yval(PdBm(RL), 2*fund1-fund2))

As there are two fundamentals at the input, these two signals can intermodulate (linearcombination of fund1 and fund2: a*fund1 + b*fund2) and produce harmonics that fall in aundesired channel, these intermodulated signals being created by the non-linearity of theamplifier. Especially when the two RF fundamentals are close to each other and also close to thedesired channel that we do not want to affect, the intermodulation distortion of the third order(IM3) has to be characterized. There are two IM3 terms at the harmonics 2*fund1-fund2 and2*fund2-fund1.



The IM3 is defined as the difference between the power in the load PdBm(RL) at fund1 and thepower in the load at 2*fund1-fund2. The extract function of Eldo on the fsst analysis allowsthe computation of the IM3.

* definition of the third order intercept point.extract fsst label=PINdBm YVAL(PdBm(Vin), fund1).extract fsst label=IIP3 0.5*meas(IM3) + meas(PindBm).extract fsst label=OIP3+ 0.5*meas(IM3) + yval(PdBm(RL), fund1)

The third intercept point IP3 is a measure to characterize the performance of non-linear systemswhen mixing signals. It can be measured on a graph by tracing the tangents on the waveformsPout=f(Pin) at fund1 and 2*fund2-fund1, and then taking the interception.

The value is then extracted from the IM3. The coordinates of the IP3 point are given by theformula:

IIP3=IM3/2 + Pin for the x-axis value (Power in Vin)OIP3=IM3/2 + Pout (at fund1) for the y-axis value (Power in the output)

NoteIn the extract syntax, an IM3 which is the label of a previous extract can only be accessedby using the meas function.

Both coordinates of IP3 (IIP3 and OIP3) can also be extracted through the use of the built-infunctions IIPxx() and OIPxx():

* functions for direct IIP3-OIP3 extracts.extract fsst label=IIP3_ref+ iipx(PdBm(Vin), PdBm(RL), fund1, 2*fund1-fund2).extract fsst label=IIP3_ref+ oipx(PdBm(RL), fund1, 2*fund1-fund2)

IMn and IPn can be obtained in the same way. The simulation takes less than 1 second.

Figure 14-9 shows the power on the load RL in dBm. This is a greatly enlarged view of the areaaround fund1 and fund2. The other harmonics generated by the combination of fund1 andfund2 harmonics, and especially 2*fund1-fund2 can then be clearly seen.

The other results from the extract commands are given below:

IM3 = 5.1971E+01 (in dBm)

For the third intermodulation:

IIP3 = -1.9446E+00 (in dBm)OIP3 = 2.6099E+01 (in dBm)

for the coordinates of the third order interception point.



Simulation Results

To view the simulation results shown in Figure 14-9, you must zoom in so that the minimumand maximum values on the X-axis are 894MHZ and 906MHz respectively. To do this, selectthe X-axis using the right mouse button, the X Axis pop-up menu will be displayed. SelectZoom Range X in the pop-up menu, the X - Axis Min/Max dialog box will be displayed. Enterthe minimum and maximum values in the Min and Max fields respectively, and click OK.

Figure 14-9. Multi-tone Analysis—IM3 and IP3 Extraction Simulation Results


Eldo RF TutorialsTutorial #4—S-Parameter Extraction for an Amplifier

Tutorial #4—S-Parameter Extraction for anAmplifier

The same amplifier is used here as in Tutorial #1—Single-Tone Steady-State Analysis andTwo-Port Noise Extraction for an Amplifier, Tutorial #2—Power Efficiency and 1dBCompression Point Extraction for an Amplifier and Tutorial #3—Multi-Tone Analysis, andIM3 and IP3 Extraction for an Amplifier. To compute the S-parameters of the amplifier, twosources on each side of the circuit are added to determine all entries of the 2×2 S-parametermatrix.



• S-parameter extraction

• Input and output impedance extraction

• Available Gain circle and Power Gain circle

• Stability circles

Complete Netlist

.include ampli.cktVdc VPOS 0 3.3XCKT IN LOAD VPOS AMPLI_CKTVin IN 0 RPORT=50 iport=1 FOUR fund1 PdBm (1) p1 -90Vout LOAD 0 RPORT=Rout iport=2 FOUR fund1 PdBm (1) p2 -90.TEMP 50.0.OP.param Rout=50.sst fund1=fund_1 nharm1=10.param fund_1=900Meg.param p1=-10.param p2=-20* s-params on port1.extract fsst label=STEADY_STATE_S11 yval(sdb(1,1), fund1).extract fsst label=STEADY_STATE_S21 yval(sdb(2,1), fund1)* s-params on port2.extract fsst label=STEADY_STATE_S12 yval(sdb(1,2), fund1)

PORT1

RPORT RPORT

PORT2P1 P2

VDD

LOADIN

Vin Vout



.extract fsst label=STEADY_STATE_S22 yval(sdb(2,2), fund1)* Input impedance extraction.defwave Zin=50.0*1.0 + s(1,1) / 1.0 - s(1,1).extract fsst label=STEADY_STATE_Zin_r yval(wr(Zin), fund1).extract fsst label=STEADY_STATE_Zin_i yval(wi(Zin), fund1)* Output impedance extraction.defwave Zout=50.0*1.0 + s(2,2) / 1.0 - s(2,2).extract fsst label=STEADY_STATE_Zout_r yval(wr(Zout),fund1).extract fsst label=STEADY_STATE_Zout_i yval(wi(Zout),fund1)* Available Gain circle and Power Gain circle plots.plot fsst GAC(smith).plot fsst GPC(smith)* Stability circle plots.plot fsst LSC(smith).plot fsst SSC(smith).end

Netlist Explanation

The two sources Vin and Vout are defined with the parameters; fund1, p1 and p2 respectively.With port1 (iport=1) active, s11 and s21 are computed. With port2 (iport=2) active (port1 isthen passive, reduction to RPORT only), s12 and s22 are computed.

.include ampli.cktVdc VPOS 0 3.3XCKT IN LOAD VPOS AMPLI_CKTVin IN 0 RPORT=50 iport=1 FOUR fund1 PdBm (1) p1 -90Vout LOAD 0 RPORT=Rout iport=2 FOUR fund1 PdBm (1) p2 -90.param Rout=50.sst fund1=fund_1 nharm1=10.param fund_1=900Meg.param p1=-10.param p2=-20

The values of the s-params can be extracted directly with the .extract command. Two sets ofiterations are performed, one with Vin as a source and Vout as RPORT to get sdb(1,1) andsdb(2,1), another with Vout as a source and Vin as RPORT to get sdb(1,2) and sdb(2,2). The s-params values are for a given value of the input power Pin, they are of course different fromthose obtained with a small signal ac analysis around the DC operating point. The onescomputed here are done for a point at the steady-state, taking into account the non-linearities ofthe amplifier.

* s-params on port1.extract fsst label=STEADY_STATE_S11 yval(sdb(1,1), fund1).extract fsst label=STEADY_STATE_S21 yval(sdb(2,1), fund1)* s-params on port2.extract fsst label=STEADY_STATE_S12 yval(sdb(1,2), fund1).extract fsst label=STEADY_STATE_S22 yval(sdb(2,2), fund1)

From the s-param values the z-params Z11 and Z22 can be extracted. wr(Z11) gives the realpart of the wave Z11 defined through the .defwave command, wi(Z11) gives the imaginarypart.

* Input impedance extraction



.defwave Zin=50.0*1.0 + s(1,1) / 1.0 - s(1,1)

.extract fsst label=STEADY_STATE_Zin_r yval(wr(Zin), fund1)

.extract fsst label=STEADY_STATE_Zin_i yval(wi(Zin), fund1)* Output impedance extraction.defwave Zout=50.0*1.0 + s(2,2) / 1.0 - s(2,2).extract fsst label=STEADY_STATE_Zout_r yval(wr(Zout),fund1).extract fsst label=STEADY_STATE_Zout_i yval(wi(Zout),fund1)

From the s-param values we can also compute the following .plot commands:

.plot fsst GAC(smith)

.plot fsst GPC(smith)* Stability circle plots.plot fsst LSC(smith).plot fsst SSC(smith)

These commands plot a smith chart with the following circles: Available Gain circle, PowerGain circle and Stability circles (LSC and SSC). See “Two-Port Constant Gain Circles” onpage 111 and “Two-Port Stability Circles” on page 112 for further information.

Simulation Results

The s-params values extracted for P1=-10dBm and P2=-20dBm are given below:

STEADY_STATE_S11 = -6.5152E-01STEADY_STATE_S21 = 2.0171E+01STEADY_STATE_S12 = -2.9719E+01STEADY_STATE_S22 = -6.6369E+00

The input and output impedances extracted for P1=-10dBm and P2=-20dBm are given below:

STEADY_STATE_ZIN_R = 5.5405E+00STEADY_STATE_ZIN_I = -6.9783E+01STEADY_STATE_ZOUT_R = 3.3902E+01STEADY_STATE_ZOUT_I = -4.0239E+01

NoteThe above results are located at the end of the Sparameter_1tone.chi file.

The Smith charts obtained are shown below:



Figure 14-11. Two Port Constant Gain Circle—GAC



Figure 14-12. Two Port Constant Gain Circle—GPC



Figure 14-13. Two Port Stability Circle—LSC



Figure 14-14. Two Port Stability Circle—SSC

Eldo RF TutorialsTutorial #5—Mixer Steady-State and Noise Analysis for a Gilbert Cell


Tutorial #5—Mixer Steady-State and NoiseAnalysis for a Gilbert Cell

A steady-state noise analysis simulates the effects of frequency conversion and frequencycorrelation of device noise due to large-signal, time-varying circuit operating point.

Figure 14-15. Gilbert Cell Circuit


• Multi-tone Steady-State analysis

• Multi-tone Steady-State noise analysis

• Large-signal conversion gain

• Restart from a previous sst simulation

• Noise figure

Complete Netlist

VCS N$225 0 2V* Input LO signalVIN1 IN1PP IN1NN FOUR fund1 MA (1) 50m -90.0* input RF power in dBm.param pwr_RF=-35* Input RF signalVBB1 BB1PP BB1NN iport=1 rport=50 FOUR fund2+ PdBm (1) pwr_RF -90.0V_Dcinit___VCC VCC 0 DC 5RIN1P IN1PP IN1PPP 25.0CIN1P IN1PPP IN1P 1uRIN1N IN1NN IN1NNN 25.0CIN1N IN1NNN IN1N 1u* this resistance is replaced by VBB1's RPORT*RBB1P BB1PP BB1PPP 25.0CBB1P BB1PPP BB1P 1u* this resistance is replaced by VBB1's RPORT*RBB1N BB1NN BB1NNN 25.0

FOUR

FOUR

RF

LO

IF



CBB1N BB1NNN BB1N 1uROUT1N OUT1 0 10MegCOUT1N OUT1N OUT1NN 1uLOUT1N OUT1NN OUT1 1mROUT1P OUT2 0 10MegCOUT1P OUT1P OUT1PP 1uLOUT1P OUT1PP OUT2 1mRReso nnn30 nnn31 800CReso nnn30 nnn31 100pLReso nnn30 nnn31 2.533u* Steady-state and steady-state noise analyses definition.restart Gilbert_Cell.sst sst.sst fund1=910meg fund2=900meg nharm1=5 nharm2=4.sstnoise v(out1p, out1n) lin 25 1Meg 60Meg*Single Side Band Noise figure (RF)*.snf input=(VBB1) output=(ROUT1N, ROUT1P)*+ input_sideband=(-1, 0)*Double Side Band Noise figure (RF & IM).snf input=(VBB1)+ output=(ROUT1n, ROUT1P)+ input_sideband=( (-1, 0) (1, 0) )*IEEE Double Side Band Noise figure*.snf input=(VBB1)*+ output=(ROUT1N, ROUT1P)*+ input_sideband= (-1, 0)*+ input_temp=290*Default Noise figure (all sidebands)*.snf input=(VBB1) output=(ROUT1N, ROUT1P)* Plots.plot fsst vdb(out1,out2).plot sstnoise onoise.plot sstnoise snf...

Netlist Explanation

VCS N$225 0 2V* Input LO signalVIN1 IN1PP IN1NN FOUR fund1 MA (1) 50m -90.0* input RF power in dBm.param pwr_RF=-35* Input RF signalVBB1 BB1PP BB1NN iport=1 rport=50 FOUR fund2+ PdBm (1) pwr_RF -90.0V_Dcinit___VCC VCC 0 DC 5

The LO and RF inputs are defined to be of type FOUR with fundamental frequencies fund1 andfund2, specified as voltage with magnitude MA of 50mV (large-signal for LO) and power PdBm,in dBm, of -35dBm (large-signal for RF), the angles being -90.

* Steady-state and steady-state noise analyses definition.restart Gilbert_Cell.sst sst.sst fund1=910meg fund2=900meg nharm1=5 nharm2=4.sstnoise v(out1p, out1n) lin 25 1Meg 60Meg*Single Side Band Noise figure (RF)*.snf input=(VBB1) output=(ROUT1N, ROUT1P)*+ input_sideband=(-1, 0)



*Double Side Band Noise figure (RF & IM).snf input=(VBB1)+ output=(ROUT1n, ROUT1P)+ input_sideband=( (-1, 0) (1, 0) )*Default Noise figure (all sidebands)*.snf input=(VBB1) output=(ROUT1N, ROUT1P)* Plots.plot fsst vdb(out1,out2).plot sstnoise onoise.plot sstnoise snf

A two-tone steady-state analysis is performed, which can be restarted from a previous run withthe .restart sst command, thus saving simulation time.

The .sstnoise analysis is run for a frequency range 1MHz - 60MHz, with an output voltagenoise computed on the differential output between nodes out1p and out1n. This frequencyrange corresponds to an area in the neighborhood of the IF frequency

.

The Noise Figure .snf is defined as:

The Single Sideband Noise Figure is defined as:

RF frequency corresponds to the (-1, 0) sideband.

The Double Sideband Noise Figure is defined as:

The IEEE SSB Noise Figure is defined as:

F IF F LO F RF Fund1 Fund2 10 MHz=–=–=( )

Total Noise at the Output Contribution from the Load (ouput) Noise–Contribution from the Source (input)Noise at Specifed Sidebands

------------------------------------------------------------------------------------------------------------------------------------------------------------------------

SN FSSBTotal Noise Load Noise–

Source Noise at RF Frequency-------------------------------------------------------------------------=

fp IF Noise Frequency=

RF Noise Frequency F LO IF Noise Frequency 1 0,–( )⇒( )–=

SN F DSBTotal Noise Load Noise–

Source Noise at RF Frequency Source Noise at IM Frequency+---------------------------------------------------------------------------------------------------------------------------------------------------------=

fp IF Noise Frequency=

RF Noise Frequency F LO IF Noise Frequency 1 0,–( )⇒–=

IM Noise Frequency F LO IF Noise Frequency 1 0,( )⇒+=



The Source Noise contributes to the Total Noise only from the side bands defined ininput_sideband.

For the Default Noise Figure, all sidebands are used to compute the Source Noise Contribution,which gives an output roughly equivalent to the Double Sideband results. The default value ofthe input_sideband parameter is (*,*).

Access to the output noise onoise and the spot noise figure snf is given with the command.plot sstnoise.

Simulation Results

The output voltage spectrum with the harmonics from the two fundamentals LO and RFcombination, as well as the output noise spectrum and the noise figure are shown below.

The simulation takes 1 minute to be performed.

Figure 14-16. Mixer Steady-State and Noise Analysis Simulation Results 1

SN F IEEETotal Noise Load Noise–

Source Noise at RF Frequency at 290K----------------------------------------------------------------------------------------------=



Figure 14-17. Mixer Steady-State and Noise Analysis Simulation Results 2


Eldo RF TutorialsTutorial #6—Mixer Steady-State AC Analysis for a Gilbert Cell

Tutorial #6—Mixer Steady-State AC Analysis fora Gilbert Cell

The Gilbert cell circuit is a mixer. A steady-state two-tone analysis can be performed to obtainthe output spectrum. With Eldo RF, you have the option to compute an AC analysis around thesteady-state by performing a steady-state analysis for the large-signal LO and a steady-state ACanalysis for the RF signal. The advantage of such a method is a faster simulation, the drawbackis that the RF signal is considered as a small-signal.

Figure 14-18. Gilbert Cell Circuit


• Steady-State AC analysis

• Small-signal conversion gain of a mixer

Complete Netlist

VCS N$225 0 2V* Input LO signalVIN1 IN1PP IN1NN FOUR fund1 MA (1) 50m -90.0* Input RF signal defined as an AC sourceVBB1 BB1PP BB1NN AC 1 -90.0V_Dcinit___VCC VCC 0 DC 5RIN1P IN1PP IN1PPP 25.0CIN1P IN1PPP IN1P 1uRIN1N IN1NN IN1NNN 25.0CIN1N IN1NNN IN1N 1uRBB1P BB1PP BB1PPP 25.0CBB1P BB1PPP BB1P 1uRBB1N BB1NN BB1NNN 25.0CBB1N BB1NNN BB1N 1u

RF IF

LO

AC

FOUR



ROUT1N OUT1 0 10MegCOUT1N OUT1N OUT1NN 1uLOUT1N OUT1NN OUT1 1mROUT1P OUT2 0 10MegCOUT1P OUT1P OUT1PP 1uLOUT1P OUT1PP OUT2 1mRReso nnn30 nnn31 800CReso nnn30 nnn31 100pLReso nnn30 nnn31 2.533u* Steady-State and Steady-State AC analysis definition.sst fund1=910meg nharm1=5.sstac lin 50 880meg 909.9meg* Conversion gain.defwave ConvGain_Val=vm(out2, out1).h(-1)* Plots.plot fsst vdb(out2, out1).plot sstac vdb(out2, out1)* Extract.extract sstac label=ConversionGain+ yval(wdb(ConvGain_Val), 10meg)...

Netlist Explanation

VCS N$225 0 2VVIN1 IN1PP IN1NN FOUR fund1 MA (1) 50m -90.0VBB1 BB1PP BB1NN AC 1 -90.0V_Dcinit___VCC VCC 0 DC 5

The LO signal is defined to be type FOUR with a fundamental frequency fund1, specified as avoltage with a magnitude MA of 50mV and an angle of -90.

The RF signal is defined as a small signal AC source.

* Steady-State and Steady-State AC analysis definition.sst fund1=910meg nharm1=5.sstac lin 50 880meg 909.9meg

Here a steady-state analysis (.sst) and a steady-state AC analysis (.sstac) are defined. An ACanalysis is computed at the steady-state around the 5 harmonics of fundamentalfund1=910MHz, for an input frequency fp varying linearly from 880MHz to 909.9MHz.

* Conversion gain.defwave ConvGain_Val=vm(out2, out1).h(-1)

The ConvGain_Val wave is defined as the harmonic at -fund1 on the output with the specificsyntax .h(-1). It is used to extract the conversion gain value at fp-fund1.

* Plots.plot fsst vdb(out2, out1).plot sstac vdb(out2, out1)

The voltage difference between out2 and out1 is plotted for the single-tone steady-stateanalysis (.plot fsst) and for the AC analysis done around the steady-state analysis (.plotsstac). The .plot fsst command will plot the voltage difference between out2 and out1 of



the whole spectrum, and .plot sstac will plot the voltage difference for the spectrum partdefined in the .sstac command.

* Extract.extract sstac label=ConversionGain+ yval(wdb(ConvGain_Val), 10meg)

The .extract sstac command is used to extract the value of the conversion gain at 10 MHz.

Simulation Results

The two plots are shown below.

Figure 14-19. Mixer Steady-State AC Analysis Simulation Results

Eldo RF TutorialsTutorial #7—Mixer Steady-State TF Analysis for a Gilbert Cell


Tutorial #7—Mixer Steady-State TF Analysis for aGilbert Cell

SSTXF is a small signal analysis that can be performed on top of a large-signal Steady-State(.SST analysis). SSTXF computes the transfer functions from any source at any frequency to asingle output at a single frequency. This analysis models frequency conversion effects and isused to compute useful quantities such as conversion efficiency, image and sideband rejectionand power supply rejection.


• Steady-State TF analysis

• Small-Signal rejection and conversion gain of a mixer

Complete Netlist

VCS N$225 0 2V* Input LO signalVLO IN1PP IN1NN FOUR fund1 MA (1) 50m -90.0* Input RF portVRF BB1PP BB1NN 0V_Dcinit___VCC VCC 0 DC 5RIN1P IN1PP IN1PPP 25.0CIN1P IN1PPP IN1P 1uRIN1N IN1NN IN1NNN 25.0CIN1N IN1NNN IN1N 1uRBB1P BB1PP BB1PPP 25.0CBB1P BB1PPP BB1P 1uRBB1N BB1NN BB1NNN 25.0CBB1N BB1NNN BB1N 1uROUT1N OUT1 0 10MegCOUT1N OUT1N OUT1NN 1uLOUT1N OUT1NN OUT1 1mROUT1P OUT2 0 10MegCOUT1P OUT1P OUT1PP 1uLOUT1P OUT1PP OUT2 1mRReso nnn30 nnn31 800CReso nnn30 nnn31 100pLReso nnn30 nnn31 2.533u* Steady-state and steady-state XF analyses definition.sst fund1=910meg nharm1=5.sstxf v(out2, out1) lin 50 0.1meg 20meg* Conversion gain.defwave ConvGain_Val=xf(VRF).h(-1)* Image rejection.defwave ImagRej_Val=xf(VRF).h(1)* Power supply Rejection.defwave PowRej_Val=xf(VCS).h(0)* Plots.plot fsst vdb(out2, out1).plot sstxf xfdb(VRF).plot sstxf xfdb(VCS)* Extract.extract sstxf label=Conversion_Gain



+ yval(wdb(ConvGain_Val), fund1-10meg).extract sstxf label=Image_Rejection+ yval(wdb(ImagRej_Val), fund1+10meg).extract sstxf label=Powsup_Rejection+ yval(wdb(PowRej_Val), 10meg)...

Netlist Explanation

VCS N$225 0 2VVLO IN1PP IN1NN FOUR fund1 MA (1) 50m -90.0VRF BB1PP BB1NN 0V_Dcinit___VCC VCC 0 DC 5

The large signal LO is defined as a FOUR with a fundamental frequency fund1, specified as avoltage with a magnitude MA of 50mV and an angle of -90. The RF port is considered as a smallsignal.

* Steady-state and steady-state XF analyses definition.sst fund1=910meg nharm1=5

Here a steady-state analysis (.sst) is defined with a fundamental frequency fund1=910MHzand 5 harmonics.

.sstxf v(out2, out1) lin 50 0.1meg 20meg

Here a steady-state TF analysis (.sstxf) is defined. It tells the simulator to compute a smallsignal analysis on top of the large-signal steady state. This is used to calculate all the transferfunctions between all the sources in the circuit at all the frequencies (around each harmonics ofthe steady-state) and one output at a single frequency. The output as well as the outputfrequency (let’s call it fp) are specified on the .sstxf command line. The specified output isthe voltage between nodes out2 and out1 and the analysis is performed for different values offp varying linearly from 0.1MHz to 20MHz.

* Conversion gain.defwave ConvGain_Val=xf(VRF).h(-1)

The conversion gain is the transfer function between the RF port at the RF frequency fund1-fp

and the output at frequency fp. It is displayed as xfdb(VRF).h(-1).

* Image rejection.defwave ImagRej_Val=xf(VRF).h(1)

The image rejection is the transfer function between the RF port at the image frequencyfund1+fp and the output at the frequency fp. It is displayed as XFDB(VRF).h(1).

* Power supply Rejection.defwave PowRej_Val=xf(VCS).h(0)

Finally the power supply rejection is the transfer function between the supply source at fp andthe output at the same frequency and is displayed as xf(VCS).h(0).



* Plots.plot fsst vdb(out2, out1).plot sstxf xfdb(VRF).plot sstxf xfdb(VCS)

The first statement plots the voltage difference between out2 and out1 for the single-tonesteady-state analysis. The second statement plots the results of the computed transfer functionbetween source VRF and output port (out2, out1), displayed in dB. The third statement plotsthe results of the computed transfer function between source VCS and output port (out2,out1), displayed in dB.

* Extract.extract sstxf label=Conversion_Gain+ yval(wdb(ConvGain_Val), fund1-10meg).extract sstxf label=Image_Rejection+ yval(wdb(ImagRej_Val), fund1+10meg).extract sstxf label=Powsup_Rejection+ yval(wdb(PowRej_Val), 10meg)...

The .extract sstxf command is used to extract the values of the conversion gain, imagerejection and power supply rejection at fund1-fp, fund1+fp and fp respectively.

Simulation Results

The results from the extractions of conversion gain, image rejection and power supply rejectionare shown below:

CONVERSION_GAIN= 5.6001E+00IMAGE_REJECTION= 5.5397E+00POWSUP_REJECTION= -3.1622E+01

The result for image rejection is very similar to the result for conversion gain because this celldoes not contain any built-in image rejection circuitry, it is a plain mixer.

The plot for Conversion Gain (fund1-fp), is shown in Figure 14-20.



Figure 14-20. Mixer Steady-State TF Analysis Simulation Results 1

The plot for Image Rejection (fund1+fp) is shown in Figure 14-21.




The plot for Power Supply Rejection (fp) is shown in Figure 14-22.

Eldo RF TutorialsTutorial #8—Steady-State Analysis for a Frequency Divider


Tutorial #8—Steady-State Analysis for aFrequency Divider

This circuit describes how frequency dividers are handled with Eldo RF. There are no newdefinitions or commands used in this tutorial.

Figure 14-23. Frequency Divider Circuit

Complete Netlist

* Input signal frequency is at harmonic 4 of* the output frequencyvck ck mc four fund1 ma (4) 1.65 -90* Steady-state analysis with fundamental set at 12.5Meg.param ff1=’50Meg/4’.sst fund1=ff1 nharm1=40* Simple bsim3v3 models.model n nmos level=53.model p pmos level=53* Power suppliesvmc mc 0 1.65Vvdd vdd 0 3.3Vvss vss 0 0V* Global nets.global vss vdd* Circuit description.subckt inverter out inm1 out in vss vss n w=4u l=0.35um2 out in vdd vdd p w=6u l=0.35uco out vss 10ff.ends inverter

F0

F0/4

F0/2÷2

÷2



.subckt hff out in clock nclockm1 nin nclock nout vss n w=7u l=0.35um2 nin clock nout vdd p w=9u l=0.35um3 nout clock fb vss n w=7u l=0.35um4 nout nclock fb vdd p w=9u l=0.35uxa out nout inverterxb fb out inverterxc nin in inverterc1 nin vss 10fc2 nout vss 15fc3 fb vss 12f.ends hff

.subckt ff out nout in clockx1 int in clock nclock hffx2 out int nclock clock hffx3 nout out inverterca int vss 10fcb out vss 12fx4 nclock clock inverter.ends ff

xc1 b0 nb0 nb0 ck ffxc2 b1 nb1 nb1 b0 ff

* Plots.plot tsst V(CK).plot tsst V(B0).plot tsst V(B1).plot fsst VDB(B0).plot fsst VDB(B1).end

Netlist Explanation

* Input signal frequencyvck ck mc four fund1 ma (4) 1.65 -90

An input Fourier source is defined. It has a magnitude MA of 1.65V, with a fundamentalfrequency of 4*fund1 (the factor 4 is specified with (4)), which corresponds to harmonic four ofthe output frequency. An angle of -90 is specified to generate a sine equivalent waveform(cosine is taken by default).

* Steady-state analysis with fundamental set at 12.5Meg.param ff1=’50Meg/4’.sst fund1=ff1 nharm1=40

The steady-state analysis is performed with a fundamental frequency fund1 at ff1=12.5 MHz,which is one quarter of the input frequency (divided by four).

As frequency dividers are very non-linear circuits, the simulation is performed with manyharmonics (nharm1=40) to better represent the highly non-linear signals.



* Plots.plot tsst V(CK).plot tsst V(B0).plot tsst V(B1).plot fsst VDB(B0).plot fsst VDB(B1)

The frequency spectrum (.plot fsst) of the two divider outputs are plotted, (B0 for divide bytwo and B1 for divide by four), and the input source and dividing outputs equivalent timedomain waveforms are regenerated (.plot tsst).

Simulation Results

The simulation results shown in Figure 14-24 have been zoomed in. To zoom into a specificrange select on the Y-axis, select the Y-axis of the plot VDB(B0) with the right mouse button,and select Zoom Range in the pop up dialog box. Enter a minimum value of -60 and amaximum value of 10 in the Min and Max fields respectively. Repeat this for plot VDB(B1),specifying a Min and Max range of -40 and 10 respectively.

Figure 14-24. Steady-State Analysis Simulation Results 1



Figure 14-25. Steady-State Analysis Simulation Results 2

Eldo RF TutorialsTutorial #9—Steady-State Analysis of Autonomous Circuit for an Oscillator


Tutorial #9—Steady-State Analysis ofAutonomous Circuit for an Oscillator

A steady-state analysis of a VCO is performed in order to extract the oscillating frequencygenerated by this circuit.



• Steady-State stability analysis

• Steady-State oscillator analysis

• Probe added in the circuit

Complete Netlist

* Device models.include "bsim3v3.inc".model d d is=1e-15 cj=2.3p vj=0.5* Power supplyvdd vdd 0 1.5* VCO control voltage [0.5V - 2.5V].param vctrl=1.2vcontrol vcontrol 0 ’vctrl’* CMOS cross connected pairm1 vp vn 0 0 n w=250u l=0.5um2 vn vp 0 0 n w=250u l=0.5u* Diodes for capacitance tuningd1 vp vcontrol dd2 vn vcontrol d* Parametrized quality factor.param qfact=6* Self resistances deduced from the Q factor (Q=w.L/R).param rl=’2*3.14159*1.8*3.2/qfact’

VcontrolVprobe



* Inductors (spiral)l1 vp xvp 3.2nHr1 xvp com ’rl’c1 vp 0 0.1pccom1 com 0 0.1pl2 vn xvn 3.2nHr2 xvn com ’rl’c2 vn 0 0.1pccom2 com 0 0.1p

* Bias stagem3 vdd b com vdd p w=200u l=0.5um4 vdd b b vdd p w=200u l=0.5uibias b 0 3m

* PROBE device added.sstprobe vp 0* Operating point analysis (useful to get gm(m1/m2)* and diode capacitances).op* Steady-state analysis for autonomous circuit.sst oscil nharm_osc1=10

* Plots.plot fsst vdb(vp, vn).plot tsst v(vp, vn) v(vp) v(vn).plot fsst i(vdd).end

Netlist Explanation

* Device models.include "bsim3v3.inc".model d d is=1e-15 cj=2.3p vj=0.5* Power supplyvdd vdd 0 1.5* VCO control voltage [0.5V - 2.5V].param vctrl=1.2vcontrol vcontrol 0 ’vctrl’* CMOS cross connected pairm1 vp vn 0 0 n w=250u l=0.5um2 vn vp 0 0 n w=250u l=0.5u* Diodes for capacitance tuningd1 vp vcontrol dd2 vn vcontrol d* Parametrized quality factor.param qfact=6* Self resistances deduced from the Q factor (Q=w.L/R).param rl=’2*3.14159*1.8*3.2/qfact’* Inductors (spiral)l1 vp xvp 3.2nHr1 xvp com ’rl’c1 vp 0 0.1pccom1 com 0 0.1pl2 vn xvn 3.2nHr2 xvn com ’rl’c2 vn 0 0.1pccom2 com 0 0.1p



* Bias stagem3 vdd b com vdd p w=200u l=0.5um4 vdd b b vdd p w=200u l=0.5uibias b 0 3m* PROBE device added.sstprobe vp 0

The circuit is a VCO with a input voltage source vcontrol.

A specific component, probe, is inserted to achieve the oscillator analysis. This probe does notaffect the other types of analyses that can be specified in the netlist, it is only active duringsteady-state analysis for determining the oscillating frequency. A probe has to be added inparallel with the resonating part of the circuit, and should be unique.

Operating point analysis (useful to get gm(m1/m2) and diode capacitances.op* Steady-state analysis for autonomous circuit.sst oscil nharm_osc1=10* Plots.plot fsst vdb(vp, vn).plot tsst v(vp, vn) v(vp) v(vn).plot fsst i(vdd)

The oscillator analysis is defined by the .sst oscil command and 10 harmonics will becomputed in this case.

When no estimated oscillation fundamental frequency (corresponding to the parameterfund_osc_guess1) is specified as is the case here, a Local Stability analysis is performedprior to the oscillation analysis. The estimate of the oscillation frequency found by the stabilityanalysis is then used for computing the oscillator analysis (the fund_osc_guess1 parameterwill take this estimated value).

NoteFor circuits with more than 500 nodes, it is mandatory to define the fund_osc_guess1

parameter in the .sst oscil command, because the Local Stability analysis is unable tohandle circuits with so many nodes.

The .sst oscil analysis returns the value of the oscillator frequency.

Simulation Results

The simulation first returns an estimate of the oscillator frequency (1.71 GHz), and afterrunning the steady-state analysis gives the final computed value: 1.82 GHz. This value can alsobe measured on the EZwave frequency spectrum and the time domain waveforms.



Figure 14-27. Steady-State Analysis of Autonomous Circuit Simulation Results 1



Figure 14-28. Steady-State Analysis of Autonomous Circuit Simulation Results 2


Eldo RF TutorialsTutorial #10—Sweeping the Oscillator Frequency

Tutorial #10—Sweeping the Oscillator FrequencyThis is an identical circuit to Tutorial #9—Steady-State Analysis of Autonomous Circuit for anOscillator. Multiple runs are done by sweeping the control voltage which sets the oscillatorfrequency.



• Parametric Steady-State oscillator analysis

• Extraction of the fundamental oscillation frequency

Complete Netlist

* Device models.include "bsim3v3.inc".model d d is=1e-15 cj=2.3p vj=0.5* Power supplyvdd vdd 0 1.5* VCO control voltage [0.5V - 2.5V].param vctrl=1.2vcontrol vcontrol 0 'vctrl'* CMOS cross connected pairm1 vp vn 0 0 n w=250u l=0.5um2 vn vp 0 0 n w=250u l=0.5u* Diodes for capacitance tuningd1 vp vcontrol dd2 vn vcontrol d* Parametrized quality factor.param qfact=6* Self resistances deduced from the Q factor (Q=w.L/R).param rl='2*3.14159*1.8*3.2/qfact'* Inductors (spiral)l1 vp xvp 3.2nHr1 xvp com 'rl'c1 vp 0 0.1pccom1 com 0 0.1pl2 vn xvn 3.2nHr2 xvn com 'rl'c2 vn 0 0.1pccom2 com 0 0.1p

Vcontrol

Vprobe



* Bias stagem3 vdd b com vdd p w=200u l=0.5um4 vdd b b vdd p w=200u l=0.5uibias b 0 3m* Operating point analysis (useful to get gm(m1/m2) and* diode capacitances).op* Steady-state analysis for autonomous circuit.sst oscil fund_osc_guess1=2giga nharm_osc1=10* PROBE device added.sstprobe vp 0* Plots.plot fsst vdb(vp, vn).plot tsst v(vp, vn) v(vp) v(vn)* Sweep the control voltage and extract oscillation frequency.step param vctrl 0.5 2.5 0.1* Extraction of oscillation frequency.extract fsst label=fosc fund_osc.end

Netlist Explanation

...* Operating point analysis (useful to get gm(m1/m2) and* diode capacitances).op* Steady-state analysis for autonomous circuit.sst oscil fund_osc_guess1=2giga nharm_osc1=10* PROBE device added.sstprobe vp 0* Plots.plot fsst vdb(vp, vn).plot tsst v(vp, vn) v(vp) v(vn)* Sweep the control voltage and extract oscillation frequency.step param vctrl 0.5 2.5 0.1* Extraction of oscillation frequency.extract fsst label=fosc fund_osc

The netlist is very similar to that used in Tutorial #9—Steady-State Analysis of AutonomousCircuit for an Oscillator except that we sweep the input source value, which will induce a sweepof the oscillator frequency.

NoteIf the oscillator analysis does not converge, you can try again and move the probe whichmay have not been defined to be in the right place.* Steady-state analysis for autonomous circuit.sst oscil fund_osc_guess1=2giga nharm_osc1=10

In this tutorial, we perform an oscillator analysis by explicitly specifying the fund_osc

parameter with an estimated oscillation frequency of 2 GHz. No Local Stability analysis will beperformed, the first analysis will start from the fund_osc frequency specified. Each point in theparametric analysis will use the result from a previous run as an initial guess.



* Plots.plot fsst vdb(vp, vn).plot tsst v(vp, vn) v(vp) v(vn)* Extraction of oscillation frequency.extract fsst label=fosc fund_osc

The fundamental oscillation frequency is extracted for each run, through the .extract

command with the keyword fund_osc.

Simulation Results

Figure 14-30 shows the oscillation frequency as a function of the control voltage. To view as isshown below, open the vco_step.swd file in EZwave. In the Waveform List panel of theEZwave window, navigate to the extracted results by selecting vco_step > EXT and doubleclick the waveform FOSC.

Figure 14-30. Sweeping the Oscillator Frequency Simulation Results

Eldo RF TutorialsTutorial #11—Phase Noise Extraction for an Oscillator


Tutorial #11—Phase Noise Extraction for anOscillator

The identical circuit to Tutorial #9—Steady-State Analysis of Autonomous Circuit for anOscillator and Tutorial #10—Sweeping the Oscillator Frequency is used here. A steady-statenoise analysis is performed to compute the phase noise of the VCO.




• Steady-State noise analysis

• Phase noise extraction

NoteThis is an example of .SSTNOISE after .SST. Run example vco_phnoise_modsst.cir foran example of .SSTNOISE after .MODSST.

Complete Netlist

* Device models.include "bsim3v3.inc".model d d is=1e-15 cj=2.3p vj=0.5* Power supplyvdd vdd 0 1.5* VCO control voltage [0.5V - 2.5V].param vctrl=1.2vcontrol vcontrol 0 ’vctrl’* CMOS cross connected pairm1 vp vn 0 0 n w=250u l=0.5um2 vn vp 0 0 n w=250u l=0.5u* Diodes for capacitance tuningd1 vp vcontrol dd2 vn vcontrol d* Parametrized quality factor.param qfact=6* Self resistances deduced from the Q factor (Q=w.L/R).param rl='2*3.14159*1.8*3.2/qfact'* Inductors (spiral)l1 vp xvp 3.2nH

Vcontrol

Vprobe



r1 xvp com 'rl'c1 vp 0 0.1pccom1 com 0 0.1pl2 vn xvn 3.2nHr2 xvn com 'rl'c2 vn 0 0.1pccom2 com 0 0.1p* Bias stagem3 vdd b com vdd p w=200u l=0.5um4 vdd b b vdd p w=200u l=0.5uibias b 0 3m* Operating point analysis (useful to get gm(m1/m2) and*+ diode capacitances).op* Steady-state analysis for autonomous circuit.sst oscil nharm_osc1=10* Steady-state noise analysis.sstnoise v(vp, vn) harm (1) dec 10 100 100000k* PROBE device added.sstprobe vp 0* Plots.plot fsst vdb(vp, vn).plot tsst v(vp, vn) v(vp) v(vn).plot tsst i(vdd).plot sstnoise db(sphi).end

Netlist Explanation

* Operating point analysis (useful to get gm(m1/m2) and*+ diode capacitances).op* Steady-state analysis for autonomous circuit.sst oscil nharm_osc1=10* Steady-state noise analysis.sstnoise v(vp, vn) harm(1) dec 10 100 100000k* PROBE device added.sstprobe vp 0

A steady-state noise analysis is performed following the oscillator analysis. Frequencies arerelative to the carrier (harm(1)). -1/f noise is important below 100k offset from the carrier, andtranslates to a -30dB/dec slope, instead of -20dB/dec. Phase noise at 600kHz offset isapproximately -115dBc/Hz.

To extract the phase noise with as much accuracy as possible, we compute the oscillatoranalysis with 10 harmonics.

The phase noise can be plotted directly with the syntax db(sphi) and has the unit dBc/Hz.

* Plots.plot fsst vdb(vp, vn).plot tsst v(vp, vn) v(vp) v(vn).plot tsst i(vdd).plot sstnoise db(sphi)



Simulation Results

The output shows the 1/f noise effect for frequencies lower than 50 kHz (-30dB per decadeslope) and white noise effect (-20dB per decade) for upper frequencies.

Figure 14-32. Phase Noise Extraction Simulation Results


Eldo RF TutorialsTutorial #12—Digitally Modulated Sources

Tutorial #12—Digitally Modulated SourcesThis example demonstrates the usage of the different digitally modulated sources. It shows howto:

• Set up a modulated steady-state analysis to obtain I and Q plots, and IQ trajectorydiagrams

• Generate a constellation diagram from I and Q plots

There is no circuit associated with this example, the sources are simply driving a 50 Ω loadresistor.

Eldo RF Feature Used

• Modulated Steady-State analysis

Complete Netlist

* Digitally modulated sources *

*The fundamental (carrier) frequency.param valfund1 = 900meg

* A GFSK sourceVgfsk gfsk 0 RPORT=50 GFSK fdev=1meg beta=.5+ four fund1 MA (1) 2.0 -90+ pattern tsymb=Ts PRBS12 INIT_PRBS="011100101100"Rgfsk gfsk 0 50

* A GMSK sourceVgmsk gmsk 0 RPORT=50 GMSK+ four fund1 MA (1) 2.0 -90+ pattern tsymb=Ts RANDOMRgmsk gmsk 0 50

* A 8-PSK sourceVmpsk mpsk 0 RPORT=50 MPSK m=8 lpf=root_raised_cosine+ four fund1 MA (1) 2.0 -90+ pattern tsymb=Ts RANDOMRmpsk mpsk 0 50

* A 16-QAM sourceVqam mqam 0 RPORT=50 MQAM m=16 lpf=gaussian beta=0.5+ save_iqfile=""mqam_gaus.dat""+ four fund1 MA (1) 2.0 -90+ pattern tsymb=Ts PRBS7Rqam mqam 0 50

* An OQPSK sourceVoqpsk oqpsk 0 RPORT=50 OQPSK lpf=raised_cosine beta=0.3+ four valfund1 MA (1) 2.0 -90.0+ pattern tsymb=Ts PRBS25 FEEDBACK=(1, 5, 8, 18)Roqpsk oqpsk 0 50

* A PI/4 QPSK source



Vpi4qpsk pi4qpsk 0 RPORT=50 PI4QPSK lpf=raised_cosine beta=0.5+ four valfund1 MA (1) 2.0 -90.0+ pattern tsymb=Ts PRBS12 INIT_PRBS="101010010111"Rpi4qpsk pi4qpsk 0 50

* A 4-FSK sourceVmfsk mfsk 0 RPORT=50 MFSK M=4 fdev=1meg lpf=gaussian beta=0.5+ four valfund1 MA (1) 2.0 -90.0+ pattern tsymb=Ts PRBS9Rmfsk mfsk 0 50

* A I-Q modulatorViqmod iqmod 0 RPORT=50 IQMOD+ four valfund1 MA (1) 2.0 -90.0+ IQFILE=""mqam_nofilt.dat""Riqmod iqmod 0 50

* A (32 QAM I) Base Band sourceVbbandI bbandI 0 RPORT=50 MQAM m=32+ four fund1 ma (0) 1 0+ pattern tsymb=Ts prbs19RbbandI bbandI 0 50

* A (32 QAM Q) Base Band sourceVbbandQ bbandQ 0 RPORT=50 MQAM m=32+ four fund1 ma (0) 1 -90+ pattern tsymb=Ts prbs19RbbandQ bbandQ 0 50

* An EDGE sourceVedge edge 0 rport=50 EDGE+ four valfund1 MA (1) 2.0 -90.0+ pattern tsymb=Ts RANDOMRedge edge 0 50

* An HPSK sourceVhpsk hpsk 0 50 rport=50 HPSK+ spread_factor_ctrl=4 spread_code_ctrl=walsh2+ spread_factor_data=16 nb_ch_data=5 scrambling=WCDMA_UPLINK+ lpf=root_raised_cosine beta=0.65+ four valfund1 MA (1) 1.0 -90.0+ pattern tsymb=10u PRBS15Rhpsk hpsk 0 50

* Modulation parameters.param Ts=1u.param nbs=512.param modsst_tstop=Ts*nbs

* Steady-State parameters.sst fund1=valfund1 nharm1=2

* Modulated Steady-State parameters.modsst 0 modsst_tstop

* Plot definitions.plot fmodsst vi(gfsk).h(1) vr(gfsk).h(1).plot fmodsst vi(gmsk).h(1) (versus) vr(gmsk).h(1)



.plot fmodsst vi(mpsk).h(1) (versus) vr(mpsk).h(1)

.plot fmodsst vi(oqpsk).h(1) (versus) vr(oqpsk).h(1)

.plot fmodsst vi(pi4qpsk).h(1) vr(pi4qpsk).h(1)

.plot fmodsst vi(pi4qpsk).h(1) (versus) vr(pi4qpsk).h(1)

.plot fmodsst vi(mqam).h(1) (versus) vr(mqam).h(1)

.plot fmodsst vi(iqmod).h(1) (versus) vr(iqmod).h(1)

.plot fmodsst vi(mfsk).h(1) vr(mfsk).h(1)

.plot fmodsst vi(edge).h(1) (versus) vr(edge).h(1)

.plot fmodsst vi(hpsk).h(1) (versus) vr(hpsk).h(1)

.plot fmodsst vr(bbandI).h(0)

.plot fmodsst vr(bbandQ).h(0)

.end

Netlist Explanation

The carrier frequency is defined as 900 MHz and is used to set the fundamental frequency forthe .SST analysis:

.param valfund1 = 900meg

A number different sources are instantiated in the netlist: GFSK, GMSK, MPSK, MQAM,OQPSK, PI4QPSK and so on (see netlist). The instantiations are similar and we will describethe implementation of the GFSK source by way of example (refer also to “Digitally ModulatedSources” on page 127):

* A GFSK sourceVgfsk gfsk 0 RPORT=50 GFSK fdev=1meg beta=.5+ four fund1 MA (1) 2.0 -90+ pattern tsymb=Ts PRBS12 INIT_PRBS="011100101100"Rgfsk gfsk 0 50

The GFSK source supports optional parameters (not shown in the example, because the defaultvalues are used) which specify the Gaussian filter properties. These parameters are the BETA(normalized bandwidth, default value 0.5), and the I (IASC) and Q (QASC) channels amplitudescales (default values 1.0).

The carrier used for this instance is a Fourier source, four, at fund1 (the carrier frequency,900 MHz), with an amplitude of 2.0 Volts, and an initial angle of -90 degrees (to obtain a sineshape instead of a cosine shape).

The modulation pattern for the source is introduced by the pattern keyword. In this example,the symbol period is given (tsymb = Ts = 1µs), and a random sequence of symbols is chosen(using the RANDOM keyword).

It is also possible to specify a delay for the application of the pattern, using the optional delayparameter. The default value of the delay is 0.0, which is used in our case. A non-zero valueallows a shift of the pattern. This can be useful if several sources must be instantiated and madeactive at different times. An explicit list of symbols can be used instead, see “Modulation Signal(PATTERN)” on page 148. Using a random sequence means that a pseudo-random sequence ofsymbols will be internally generated by Eldo RF, and used as the digital modulation pattern.



After the sources instantiations, there follows the modulation parameters and the steady-stateand modulated steady-state specifications:

* Modulation parameters.param Ts=1u.param nbs=512.param modsst_tstop=Ts*nbs

* Steady-state parameters.sst fund1=valfund1 nharm1=2

* Modulated steady-state parameters.modsst 0 modsst_tstop

The steady-state specification is a regular .SST directive, specifying the fundamentalfrequency, vafund1 (which equals 900 MHz) and the number of harmonics (2).

The modulated steady-state analysis is specified with the .MODSST directive, which is verysimilar to a transient (.TRAN) directive. Only the TPRINT (0) and TSTOP (modsst_tstop)times are given. The analysis is defined to last 512 cycles, that is, 512 times the duration of onesymbol, or 512×1µs = 512µs. Using the .param statements simplifies the reading of the netlistand any future modifications.

The plot definition statements, see below, use a specific syntax to retrieve the time-varyingspectrum components of the output. The notation .h(1) means “harmonic number 1” which isthe carrier in this case. The different components of the spectrum components may be accessedusing this notation, using the node name as the base name (vr(gfsk) for example).

* Plot definitions.plot fmodsst vi(gfsk).h(1) vr(gfsk).h(1).plot fmodsst vi(gmsk).h(1) (versus) vr(gmsk).h(1).plot fmodsst vi(mpsk).h(1) (versus) vr(mpsk).h(1).plot fmodsst vi(oqpsk).h(1) (versus) vr(oqpsk).h(1).plot fmodsst vi(pi4qpsk).h(1) vr(pi4qpsk).h(1).plot fmodsst vi(pi4qpsk).h(1) (versus) vr(pi4qpsk).h(1).plot fmodsst vi(mqam).h(1) (versus) vr(mqam).h(1).plot fmodsst vi(iqmod).h(1) (versus) vr(iqmod).h(1).plot fmodsst vi(mfsk).h(1) vr(mfsk).h(1).plot fmodsst vi(edge).h(1) (versus) vr(edge).h(1).plot fmodsst vi(hpsk).h(1) (versus) vr(hpsk).h(1).plot fmodsst vr(bbandI).h(0).plot fmodsst vr(bbandQ).h(0)

During a modulated steady-state analysis, each signal can be represented as a time-varyingspectrum, this spectrum being a truncated Fourier series (the SST representation). In otherwords, the amplitude (and phase) of each Fourier coefficient is varying over time, due to themodulation.

The Fourier coefficient corresponding to the fundamental (carrier) may be denoted as a complexvalue:



The I and Q components correspond to the real and imaginary parts respectively of the Fouriercoefficient, that is, VR().h(1) and VI().h(1) respectively. The plot definitions generate plotsshowing the evolution, over time, of these real and imaginary parts of the first Fouriercoefficient in the spectrum. This is actually a way to visualize the I and Q channels.

Simulation Results

Trajectory Diagrams

A trajectory diagram is a plot of the imaginary component (on the y-axis) against the realcomponent (on the x-axis). For an example, see Figure 14-34.

Trajectory diagrams can be obtained in a number of ways:

1. By specify the plot in the netlist using the (versus) option, for example for the PI/4QPSKsource:

.plot fmodsst vi(pi4qpsk).h(1) (versus) vr(pi4qpsk).h(1)

2. By right-clicking a wave and selecting Plot as... > complex plane, see Figure 14-33.

Figure 14-33. Plotting in the Complex Plane

3. By using the Waveform Calculator. Using the PI/4QPSK source as an example:

a. Open the results file, dig_mod_source.swd, in EZwave.

b. Go to the Wave window which contains the waves for the real and imaginary partsfor the PI/4QPSK source, that is VR(PI4QPSK).H(1) and VI(PI4QPSK).H(1).

c. Select Edit > Options. The EZwave Display Preferences window is displayed.

d. Click on the Transformations branch and make sure that the Complex Planetransformation is selected in the Apply By Default column, then click OK.

e. Open the Waveform Calculator by clicking the Waveform Calculator button in thetoolbar, selecting Tools > Waveform Calculator, or typing CTRL+K.

I j Q⋅+



f. Select the Complex panel of the Waveform Calculator, then add the real andimaginary parts to the complex function:

complex(wf(“<dig_mod_source/FMODSST>VR(PI4QPSK).H(1)”),wf(“<dig_mod_source/FMODSST>VI(PI4QPSK).H(1)”))

g. Click Eval. This creates a waveform called wf0.

h. Click Plot. The wf0 waveform, which is the trajectory diagram, appears in a newWave window.

Figure 14-34. IQ Trajectory Diagram for a PI/4QPSK Source

Constellation Diagrams

A constellation diagram can be obtained by sampling the trajectory diagram in the middle ofeach symbol. For example, you can generate the constellation diagram for the PI/4QPSK sourceas follows:

1. Select the PI/4QPSK trajectory diagram and open the Waveform Calculator

2. Select the RF panel then click cd (Constellation Diagram). The Constellation Diagramwindow is displayed.

3. Click the Add Selected Waveforms icon.



4. Enter the following values in the Parameter Setup frame:

o Delay: 0.5e-6 (corresponding to the middle of the first symbol)

o Symbol Period: 1.0e-6

5. Click OK. The Constellation Diagram window is closed.

6. Return to the Waveform Calculator and note that a new waveform has been created,named Constellation_wf0.

7. Click Plot to plot the last created waveform. The constellation diagram is generated, seeFigure 14-35.

Figure 14-35. Constellation Diagram for a PI/4QPSK Source

Eldo RF TutorialsTutorial #13—ACPR Computations for an Amplifier


Tutorial #13—ACPR Computations for anAmplifier

This example shows how to compute the ACPR (Adjacent Channel Power Ratio) for a simpleamplifier, using the modulated Steady-State analysis. The ACPR is typically used tocharacterize the spectral regrowth effect out of amplifiers or more generally systems driven bydigitally modulated signals. The modulated Steady-State analysis with the support of built-indigitally modulated sources is ideally suited to simulate such effects.


• Modulated Steady-State analysis

• Adjacent Channel Power Ratio

Complete Netlist

* ACPR - amplifier.include ampli.cktx1 in load vdd AMPLI_CKTVdc vdd 0 3.3RL load 0 50Vin in 0 RPORT=50 iport=1 PI4QPSK lpf=raised_cosine+ beta=0.35 four fund1 PdBm (1) Pin -90 pattern DELAY=0+ TSYMB=Ts RANDOM.param Pin=-5.param Ts= 1u.param nbs=257.param modsst_tstop=Ts*nbs.sst fund1=900MegaHerz nharm1=10.modsst 0u modsst_tstop.plot fmodsst vr(load).h(1) vi(load).h(1).plot fmodsst vr(in).h(1) vi(in).h(1)* Setup FFT parameters.param t_tstart=Ts.param t_tstop=modsst_tstop.param nbpts=nbs*4.optfour tstart=t_tstart tstop=t_tstop nbpt=nbpts+ normalized=1 interpolate=0.plot fourmodsst fourdb(v(load).h(1)).plot fourmodsst fourdb(v(in).h(1)).option sst_max_liniter=100* For ACPR.param offset2=1.2Meg.param B1=1.2Meg.param B2=30k.extract fourmodsst label=out_main+ PIB(fourm(v(load).h(1)), -B1/2,B1/2).extract fourmodsst label=out_lower2+ PIB(fourm(v(load).h(1)), -offset2-B2/2, -offset2+B2/2).extract fourmodsst label=out_acpr_lower_2_dBc+ 10.*log10(meas(out_lower2) / meas(out_main)).extract fourmodsst label=in_main+ PIB(fourm(v(in).h(1)), -B1/2,B1/2).extract fourmodsst label=in_lower2



+ PIB(fourm(v(in).h(1)), -offset2-B2/2, -offset2+B2/2).extract fourmodsst label=in_acpr_lower_2_dBc+ 10.*log10(meas(in_lower2) / meas(in_main)).END

Netlist Explanation

The carrier frequency is defined as 900Meg. This frequency will be used to set the fundamentalfrequency for the .SST analysis.

Vin in 0 RPORT=50 iport=1 PI4QPSK lpf=raised_cosine+ beta=0.35 four fund1 PdBm (1) Pin -90 pattern DELAY=0+ TSYMB=Ts RANDOM

A PI/4QPSK source is used to drive an amplifier. The PI/4QPSK source supports a number ofparameters. The lpf=raised_cosine parameter indicates a Raised Cosine filter for thebaseband low-pass filter. Other possible filters are a Gaussian filter or a Square Root RaisedCosine Filter. In the case of our filter, the beta parameter indicates the roll-off factor for thefilter.

The carrier used for this instance is a FOURier source operating at fund1 (the carrier frequency,900MHz), with an amplitude of -5dBm, and an initial angle of -90 degrees (to obtain a ‘sine’shape instead of a ‘cosine’ shape). The amplitude of -5dbm is chosen to force the amplifier intoits non-linear region, so that the ACPR measurement is meaningful (or at least easilyvisualized).

The modulation pattern for the source is introduced with the pattern keyword. In this example,the symbol period is given (tsymb = Ts = 1µs), and a random sequence of symbols is chosen(using the RANDOM keyword). An explicit list of symbols can also be used instead (see “DigitallyModulated Sources” on page 127). Using a random sequence means that a pseudo-randomsequence of symbols will be internally generated by Eldo RF, and used as the digitalmodulation pattern.

* Modulation parameters.param Ts=1u.param nbs=257.param modsst_tstop=Ts*nbs* Steady-state parameters.sst fund1=900Meg nharm1=10* Modulated steady-state parameters.modsst 0 modsst_tstop

The Steady-State specification (.SST) is a regular .SST directive, specifying the fundamentalfrequency (900Meg) and the number of harmonics (10).

The Modulated Steady-State analysis is specified with the .MODSST directive. It is very similarto a transient (.TRAN) directive. Only the start and stop times are given. The analysis is definedto last 257 cycles, i.e. 257 times the duration of one symbol, i.e. 257×1µs = 257µs. Usage of.param statements simplifies both reading and further modifications of the netlist.

* Setup FFT parameters



.param t_tstart=Ts

.param t_tstop=modsst_tstop

.param nbpts=nbs*4

.optfour tstart=t_tstart tstop=t_tstop nbpt=nbpts+ normalized=1 interpolate=0

To compute the ACPR, we need to do an FFT of the complex time-varying spectrum of thesignals, to obtain the modulation spectrum, and then compute the “channel-power” to “adjacent-channel-power” ratio.

The FFT is performed using the regular .optfour command of Eldo. The parameters for the.optfour command include the start and stop times, and the number of points to use (theseparameters define the position of the sampling points). The positions of the tstart and tstop

parameters for the FFT are chosen so that they coincide with the symbol boundaries.

Choosing four points per symbol for the FFT is arbitrary. The number of points will affect thefrequency resolution of the FFT.

.plot fourmodsst fourdb(v(load).h(1))

.plot fourmodsst fourdb(v(in).h(1))

The actual FFT plot commands are shown above. These waveforms will be used to compute theACPR. They are specified using the .plot fourmodsst commands. Both the input and theoutput spectrum are plotted in this example. Remember that V(load).h(1) for exampledesignates the complex time-varying Fourier coefficient number 1 for the load signal.

* For ACPR.param offset2=1.2Meg.param B1=1.2Meg.param B2=30k.extract fourmodsst label=out_main+ PIB(fourm(v(load).h(1)), -B1/2,B1/2).extract fourmodsst label=out_lower2+ PIB(fourm(v(load).h(1)), -offset2-B2/2, -offset2+B2/2)

To compute the ACPR, the PIB() function (PIB stands for Power In Band) is used. Thisfunction returns the power in a given frequency band, by summing the Fourier coefficients inthis band. The PIB() function is used to compute the power in the channel (in this case, we havedefined the channel as being 1.2 MHz wide, and the adjacent channel of interest as being a30kHz-wide band, located 1.2 MHz away from the main channel center). The PIB(<spectrum>,<fmin>, <fmax>) function takes absolute frequency values (<fmin> and <fmax>) as arguments.

.extract fourmodsst label=out_acpr_lower_2_dBc+ 10.*log10(meas(out_lower2) / meas(out_main))

The ACPR is simply the ratio of the adjacent-to-main power values, expressed in dBc. Theexample computes the ACPR at the input and at the output. The non linearity of the amplifierdegrades the ACPR by about 28 dB (the numerical values are shown in the .chi file).



The plots for the ACPR, where the “shoulders” effect, due to the spectral regrowth is clearlyvisible at the output, are in Figure 14-36. To reproduce the results simply drag and drop onewaveform onto the other.

Simulation Results

Figure 14-36. ACPR Computations Simulation Results

Eldo RF TutorialsTutorial #14—Verilog-A Usage


Tutorial #14—Verilog-A UsageThis example shows how to use a Verilog-A model in a simulation with Eldo RF. The setupcorresponds to a two-tone test of an amplifier. The amplifier model has a non-linear transferfunction, and thus generates harmonics and intermodulation products. Compression andintermodulation effects are studied. This tutorial also illustrates the #ifdef facility of Eldo.

Complete Netlist

* Verilog-A example

* Source code of the Verilog-A modules:.include_veriloga model.vla

* Mapping of the 'ampli' module:.model ampli macro lang=veriloga

* Two-tone test:Vrf in 0 rport=50 iport=1 four fund1 fund2 pdbm (1,0)+ prf -90 (0,1) prf -90

* Verilog-A instance* (the 'ampli' source code is defined in model.vla):Yamp ampli in out params: gain=10 sat=1

* Load resistor:RL out 0 50

.param prf = -15#ifdef im3* Sweep input power to obtain an IP3 plot:.step param prf -40 10 incr 2#endif

* Definition of a two-tone SST analysis:.param f1=1.9G f2=1.901G.sst fund1=f1 nharm1=7 fund2=f2 nharm2=7

* Extract main output and third-order intermodulation:.extract fsst label=pout yval(pdbm(RL),f1).extract fsst label=p3 yval(pdbm(RL),2*f1-f2)

#ifndef im3* Output plots:.plot tsst v(in) v(out).plot fsst vdb(out)#endif

.option noascii

.end



Netlist Explanation

The first instruction in the netlist (.include_veriloga model.vla) includes the Verilog-A sourcecode in the netlist. Eldo will automatically compile the modules in the model.vla file, and createa compiled (binary) version of each module. This is completely user-transparent.

The model.vla file contains the source code of the Verilog-A model. Here is the content of themodel.vla file:

`include "discipline.h"module ampli(in, out); inout in, out; electrical in, out; electrical int;

parameter real cin = 0.0; parameter real rin = 50; parameter real rout = 50.0; parameter real cout = 0.0; parameter real gain = 10; parameter real sat = 1;

analog begin I(in) <+ V(in) / rin; I(in) <+ cin * ddt(V(in)); I(out) <+ cout * ddt(V(out)); I(out, int) <+ V(out, int) / rout; V(int) <+ sat * tanh(V(in)*gain); endendmodule

The model uses an hyperbolic tangent function (tanh()) to introduce a non-linearity in thetransfer function. The gain and the output saturation values can be controlled through the gainand sat parameters respectively. The amplifier model is setup to present 50 Ohms input andoutput impedances.

Using a Verilog-A instance in the netlist is quite simple. First, a .model statement must beincluded, to declare the ampli entity as a Verilog-A model. Second, a generic macro-modelinstantiation is used, with the standard Y statement of Eldo. The parameters which are declaredin the Verilog-A module can be modified on an instance-by-instance basis. In this case, we passvalues for the gain and sat parameters.

.model ampli macro lang=verilogaYamp ampli in out params: gain=10 sat=1

Back to the ampli_va.cir netlist, the input setup corresponds to a two-tone test. We use 1.9GHzand 1.901GHz for the fundamental frequencies. The VRF power source delivers prf dBm(under matching conditions) on each of these frequencies:

Vrf in 0 rport=50 iport=1 four fund1 fund2 pdbm (1,0)+ prf -90 (0,1) prf -90



In the ampli_va.cir netlist, you will notice #ifdef statements. This allows to use the same .cir fileto run different simulations. If the im3 symbol is defined, the simulation is a sweep analysis onthe prf parameter (the input power). If the im3 symbol is not defined, the simulation is a simpletwo-tone test at a single input power value (-15dBm). A symbol can be defined either with a#define statement in the same netlist, or at run-time, using the -define command-line switch ofEldo. This is the method we will use here.

Please refer to the previous tutorials for detailed explanations concerning the setup of two-tonesimulation, and extraction of intermodulation products.

Simulation Results

Running eldo ampli_va.cir will produce the following results:

Figure 14-37. Verilog-A Usage Simulation Results 1




On this plot, we clearly see the intermodulation products, and the saturation effect on the output(the top and bottom of the output sinusoid (V(OUT)) are affected by the tanh()-like transferfunction, thus the saturation effect.

Running eldo -define im3 ampli_va.cir will “define” the im3 symbol and thus trigger thesweep analysis for an IM3 plot. The results are shown below. The compression of the mainoutput and the third-order intermodulation product are clearly visible. The nice thing with atanh() function is the “smoothness” of the compression, and the realistic behavior of the outputs(whereas polynomial models tend to exhibit a fancy behavior when used at high input levels).




This kind of model can be used in lieu of real transistor-level descriptions, and still capture theiroverall behavior. The accuracy of the modeling is up to you, knowing that simple models willrun faster than complicated ones.

The example directory also contain an example showing the combination of this same amplifierwith a mixer, also in Verilog-A.


Eldo RF TutorialsTutorial #15—NPR Computation with Steady-State Analysis for an Amplifier

Tutorial #15—NPR Computation with Steady-State Analysis for an Amplifier

This example shows how to calculate the NPR (Noise Power Ratio) for a simple amplifier,using the steady state analysis.The NPR is used to calculate the effect of signal of the adjacentchannels (noise) on the desired channel.

The signal in the desired channel is generated as intermodulation products of the signals in theadjacent channels and is caused by the circuit non-linearities.

Complete Netlist

ampli.cir* amplifier

.include ampli.cktx1 in load vdd AMPLI_CKT

Vdc vdd 0 3.3RL load 0 50

VNPR_N inn 0 four fund1 fund2 MA+ (1, -30) vali random+ (1, -29) vali random+ (1, -28) vali random+ (1, -27) vali random+ (1, -26) vali random+ (1, -25) vali random+ (1, -24) vali random+ (1, -23) vali random+ (1, -22) vali random+ (1, -21) vali random+ (1, -20) vali random+ (1, -19) vali random+ (1, -18) vali random+ (1, -17) vali random+ (1, -16) vali random+ (1, -15) vali random+ (1, -14) vali random+ (1, -13) vali random+ (1, -12) vali random+ (1, -11) vali random+ (1, -10) vali random+ (1, -9) vali random+ (1, -8) vali random+ (1, -7) vali random+ (1, -6) vali random+ (1, -5) vali random

VNPR_P inn inp four fund1 fund2 MA+ (1, 30) vali random+ (1, 29) vali random+ (1, 28) vali random+ (1, 27) vali random+ (1, 26) vali random



+ (1, 25) vali random+ (1, 24) vali random+ (1, 23) vali random+ (1, 22) vali random+ (1, 21) vali random+ (1, 20) vali random+ (1, 19) vali random+ (1, 18) vali random+ (1, 17) vali random+ (1, 16) vali random+ (1, 15) vali random+ (1, 14) vali random+ (1, 13) vali random+ (1, 12) vali random+ (1, 11) vali random+ (1, 10) vali random+ (1, 9) vali random+ (1, 8) vali random+ (1, 7) vali random+ (1, 6) vali random+ (1, 5) vali random

.param vali=0.01 fund1=900megRIN inp in 50.sst fund1=fund1 nharm1=1 fund2=0.1meg nharm2=350.option sst_spectrum=1 sst_max_liniter=150.plot fsst vdb(load) vdb(inp)*For ACPR.param BW_MAIN=0.8Meg.param BW_UPPER=2.5Meg.extract fsst label=out_main+ PIB(vm(load), fund1-BW_MAIN/2, fund1+BW_MAIN/2).extract fsst label=out_upper+ PIB(vm(load), fund1+0.5Meg, fund1+0.5Meg+BW_UPPER).extract fsst label=out_npr+ 10.0*log10(meas(out_upper)/ meas(out_main)).step param vali 0.01 0.08 0.01

.END

Netlist Explanation

VNPR_N inn 0 four fund1 fund2 MA...VNPR_P inn inp four fund1 fund2 MA...

These source voltages represent the noise model chosen to calculate the NPR. The noise signalsare modeled as a sum of sinusoidal signals, spaced by 0.1 Meg, with random phase between[0,360]. It is represented in two bands of frequencies:

• VNPR_N represents 25 frequencies in the band (900Meg-3Meg, 900Meg-0.5Meg).

• VNPR_P represents 25 frequencies in the band (900Meg+0.5Meg, 900Meg+3Meg).

.sst fund1=fund1 nharm1=1 fund2=0.1meg nharm2=350



The steady state specification (.SST) directive, specifying the fundamental frequencies (900Meg and 0.1 Meg) and the number of harmonics for fundamental 2 as 350. This High number ofharmonics is chosen to obtain a lot of combinations between the noise frequencies.

.option sst_spectrum=1 sst_max_liniter=150

Different options related to the steady state analysis can be defined. The circuit simulated ishighly non-linear for higher values of the input power, so the maximum number of iterations ofthe linear solver set by default to 20 might not be enough, this parameter can be modifiedthrough the option: sst_max_liniter.

Simulation Results

.plot fsst vdb(load) vdb(inp)

The steady state analysis results and the input source are plotted in the frequency domain asshown below:

Plot 1

.extract fsst label=out_main+ PIB(vm(load), fund1-BW_MAIN/2, fund1+BW_MAIN/2).extract fsst label=out_upper+ PIB(vm(load), fund1+0.5Meg, fund1+0.5Meg+BW_UPPER)

The PIB() function (PIB stands for Power In Band) returns the power in a given frequencyband, by summing the Fourier coefficients in this band. The PIB() function is used to computethe noise power in the desired channel and in the adjacent channel.

.extract fsst label=out_npr+ 10.0*log10(meas(out_upper)/ meas(out_main))

This command extracts the NPR at the output of the Amplifier

.step param vali 0.01 0.08 0.01

To observe the effect of the amplitude of the input signal on the NPR of the output, we cansweep the vali parameter.



Figure 14-40. NPR Computation with Steady-State Simulation Results 1

Plot 2

The plot below shows the NPR versus the input amplitude:



Figure 14-41. NPR Computation with Steady-State Simulation Results 2

Eldo RF TutorialsTutorial #16—NPR Computation with Modulated Steady-State Analysis for an Amplifier


Tutorial #16—NPR Computation with ModulatedSteady-State Analysis for an Amplifier

This example shows how to calculate the NPR (Noise Power Ratio) for a simple amplifier,using the modulated steady state analysis. The NPR is used to calculate the effect of signal ofthe adjacent channels (noise) on the desired channel.

The signal in the desired channel is generated as intermodulation products of the signals in theadjacent channels and is caused by the circuit non-linearities.

Complete Netlist

ampli.cir* amplifier

.include ampli.cktx1 in load vdd AMPLI_CKT

Vdc vdd 0 3.3RL load 0 50

VNPR_N inn 0 four 0.1meg MA+ (8970) vali random+ (8971) vali random+ (8972) vali random+ (8973) vali random+ (8974) vali random+ (8975) vali random+ (8976) vali random+ (8977) vali random+ (8978) vali random+ (8979) vali random+ (8980) vali random+ (8981) vali random+ (8982) vali random+ (8983) vali random+ (8984) vali random+ (8985) vali random+ (8986) vali random+ (8987) vali random+ (8988) vali random+ (8989) vali random+ (8990) vali random+ (8991) vali random+ (8992) vali random+ (8993) vali random+ (8994) vali random+ (8995) vali random

VNPR_P inn inp four 0.1meg MA+ (9005) vali random+ (9006) vali random+ (9007) vali random+ (9008) vali random+ (9009) vali random



+ (9010) vali random+ (9011) vali random+ (9012) vali random+ (9013) vali random+ (9014) vali random+ (9015) vali random+ (9016) vali random+ (9017) vali random+ (9018) vali random+ (9019) vali random+ (9020) vali random+ (9021) vali random+ (9022) vali random+ (9023) vali random+ (9024) vali random+ (9025) vali random+ (9026) vali random+ (9027) vali random+ (9028) vali random+ (9029) vali random+ (9030) vali random

.param vali=0.01RIN inp in 50.sst fund1=900meg nharm1=1.modsst 0 30u.optfour tstart=20u tstop=30u nbpt=1024 normalized=1+ interpolate=0.plot fourmodsst fourdb(v(inp).h(1)).plot fourmodsst fourdb(v(load).h(1))

* For ACPR.param BW_MAIN=0.8Meg.param BW_UPPER=2.5Meg.extract fourmodsst label=out_main+ PIB(fourm(v(load).h(1)), -BW_MAIN/2, BW_MAIN/2).extract fourmodsst label=out_upper PIB(fourm(v(load).h(1)),+ 0.5Meg, 0.5Meg+BW_UPPER).extract fourmodsst label=out_npr+ 10.0*log10(meas(out_upper) /meas(out_main)).step param vali 0.01 0.08 0.01

.END

Netlist Explanation

VNPR_N inn 0 four 0.1meg MA...VNPR_P inn inp four 0.1meg MA...

As in the previous tutorial, the noise signals in the adjacent channels is modeled as a sum ofsinusoidal signals with random phases. However, in this case the signals are considered asmodulation signals, which allows to run a single tone simulation.

.sst fund1=900meg nharm1=1



The Steady-State specification (.SST) is a regular .SST directive, specifying the fundamentalfrequency (900 Meg) and the number of Harmonics (1).

.modsst 0 30u

.optfour tstart=20u tstop=30u nbpt=1024 normalized=1+ interpolate=0

The Modulated Steady-State Analysis is specified with the .MODSST directive.

To compute the NPR, we need to do an FFT of the complex time-varying spectrum of thesignals, to obtain the modulation spectrum, and then compute the “noise power ratio”. The FFTis performed using the regular .optfour command of Eldo. The parameters for the .optfour

command include the start and stop times, and the number of points to use.

Simulation Results

.plot fourmodsst fourdb(v(inp).h(1))

.plot fourmodsst fourdb(v(load).h(1))

Eldo RF will generate the spectrum around the first harmonic for nodes inp and load, and plotthe results in dB (fourdb) as shown below:

Plot 1

.extract fourmodsst label=out_main+ PIB(fourm(v(load).h(1)), -BW_MAIN/2, BW_MAIN/2).extract fourmodsst label=out_upper PIB(fourm(v(load).h(1)),+ 0.5Meg, 0.5Meg+BW_UPPER)

The PIB() function (PIB stands for Power In Band) returns the power in a given frequencyband, by summing the Fourier coefficients in this band. The PIB() function is used to computethe noise power in the desired channel and in the adjacent channel.

.extract fourmodsst label=out_npr+ 10.0*log10(meas(out_upper) /meas(out_main))

This command extracts The NPR at the output of the Amplifier

.step param vali 0.01 0.08 0.01

To observe the effect of the amplitude of the input signal on the NPR of the output, we cansweep the vali parameter.



Figure 14-42. NPR Computation with Modulated Steady-State Simulation Results 1



Plot 2

The plot below shows the NPR versus the input amplitude:

Figure 14-43. NPR Computation with Modulated Steady-State Simulation Results 2


Eldo RF TutorialsTutorial #17—EVM and BER Computations

Tutorial #17—EVM and BER ComputationsThis example shows how to compute the EVM and BER statistics for a MQAM modulatedsignal using the modulated steady state analysis. EVM and BER are useful figures of meritsproviding a diagnostic on the overall system quality. They characterize the perturbationbetween a real signal and a reference or ideal one.

Let us explain the assumptions and the algorithms used for the estimation of EVM and BER.

EVM can be computed from the constellation diagram of a modulated signal and theconstellation diagram of a reference signal. If IMEAS, QMEAS and IREF, QREF, stand respectivelyfor the I and Q components of a modulated signal and its reference signal, then we provide threequantities:

These quantities are expressed in terms of their RMS values with Magnitude error and EVMbeing expressed in percentage and normalized to the reference signal.

Bit Error Rate is estimated only for MPSK and MQAM modulated signals. This estimation iscomputed like EVM from two constellation diagrams. We assume that the perturbation betweenthe real and the ideal diagrams follows a Gaussian law with a standard deviation deduced fromthe EVM calculations.

For MPSK, we suppose that only the phase error influences the detection quality (consequentlyσ is the Phase Error). The BER is then defined as the probability for the instantaneous phaseerror to be greater than the decision angle (π/M).

For MQAM, we define the I and Q errors as the RMS values of the differences in I and Qbetween the real and ideal diagrams. We consider that the detection process only depends onthese two measurements. We define two Gaussian laws for the I and Q errors (σI and σQ). TheBER is then defined as the probability for the instantaneous I error to be greater than the Idecision distance (half the Inphase distance between 2 adjacent ideal positions) or for theinstantaneous Q error to be greater than the Q decision distance (half the Quadrature distancebetween 2 adjacent ideal positions). The calculation is slightly different for the particular caseswhere M=8 or M=32.

In this example, we focus on the influence of Gaussian filtering on a simple 16QAM signal.

Phase error arctanQMEAS

I MEAS----------------- arctan

QREF

I REF-------------–=

Magnitude error I MEAS2

QMEAS2

+ I REF2

QREF2

+–=

EVM I MEAS I REF–( )2QMEAS QREF–( )2

+=



Complete Netlist

mqam_evm.cir* Tutorial example EVM, BER for QAM

* No filter versionc1 out 0 1pfR1 out 0 0.5iout 0 out MQAM m=16 lpf=no_fil four fund1+ MA (1) 1.0 -90+ pattern delay=10n tsymb=50n PRBS9

* filtered versionc2 outf 0 1pfR2 outf 0 0.5ioutf 0 outf MQAM m=16 beta=0.5 lpf=gaussian+ four fund1 MA (1) 1.0 -90+ pattern delay=10n tsymb=50n PRBS9

.sst fund1=1gigaherz nharm1=1

.modsst 0 15000n

.option hmax=2n

.plot fmodsst vr(out).h(1) vi(out).h(1)

.plot fmodsst vr(outf).h(1) vi(outf).h(1)

.end

Netlist Explanation

We instantiate two independent circuits. In the first one, we specify an unfiltered 16QAMmodulated source with a pseudo-random bits sequence.

* No filter versionc1 out 0 1pfR1 out 0 0.5iout 0 out MQAM m=16 lpf=no_fil four fund1+ MA (1) 1.0 -90+ pattern delay=10n tsymb=50n PRBS9

In the second one, we specify the same modulated source with Gaussian filtering (BT=0.5).Note the bits sequence is identical in both cases, defined by PRBS9.

* filtered versionc2 outf 0 1pfR2 outf 0 0.5ioutf 0 outf MQAM m=16 beta=0.5 lpf=gaussian+ four fund1 (1) 1.0 -90+ pattern delay=10n tsymb=50n PRBS9

The Steady-State specification (.SST) is a regular .SST directive, specifying the fundamentalfrequency (1 GHz, equal to the carrier frequency) and the number of harmonics (1).

.sst fund1=1gigaherz nharm1=1



The Modulated Steady-State analysis is specified with a .modsst directive. The maximumallowed timestep is set to 2ns, using the .option hmax=2n directive, in order to obtain a goodaccuracy.

.modsst 0 15000n

.option hmax=2n

The plot definition statements use a specific syntax to retrieve the time-varying spectrumcomponents of the output. The notation .h(1) means “harmonic number 1” which is the carrierin this case. The I (vr) and Q (vi) components of the ideal and filtered modulated signal aredisplayed as a function of the simulation time.

.plot fmodsst vr(out).h(1) vi(out).h(1)

.plot fmodsst vr(outf).h(1) vi(outf).h(1)

Simulation Results

Open the mqam_evm.swd file using EZwave. The real and imaginary waveforms will be plottedautomatically for V(OUT).H(1) and V(OUTF).H(1).

To compute the EVM and BER results we need to create the constellation diagrams forV(OUT).H(1) and V(OUTF).H(1). To do this follow the steps below:

• Before the constellation diagrams can be generated a complex plane transformationmust be applied to the real and imaginary parts for both sets of plots. This is done usingthe Waveform Calculator. Open the Waveform Calculator by clicking the WaveformCalculator button in the toolbar or selecting Tools > Waveform Calculator.

• Select the complex function from the Complex panel. Add the real and imaginarywaveforms to the function by using the Add Selected Waveforms button in theWaveform Calculator toolbar. To add the real and imaginary waveforms to the functionselect the real waveform, now select the Add Selected Waveforms button (shown inFigure 14-44). Now place the curser after the comma at end of the function, select theimaginary waveform and click the Add Selected Waveforms button. Now select the Evalbutton to evaluate the function and plot the result by selecting the Plot button. This isshown below for V(OUT).H(1):



Figure 14-44. EVM and BER Computations Complex Function

• Repeat the above for the waveform V(OUTF).H(1), the two plots wf0 and wf1 are shownin Figure 14-45.

NoteThe Complex Plane transformation must be selected in the Transformations tab in theEZwave Options window.



Figure 14-45. EVM and BER Computations Complex Plane Plots

• To create the constellation diagrams start by selecting the RF panel in the WaveformCalculator. Click on CD, the Constellation Diagram window will be displayed. Add thewf0 waveform to the Source Waveform field and enter the following: Delay = 35n,Symbol Period = 50n, as shown below:



Figure 14-46. EVM and BER Computations Constellation Diagram Setup

• Then click OK.

• Click Plot to generate the constellation diagram. Repeat this for the wf1 waveform. Theconstellation diagrams for wf0 and wf1 are shown below:

Figure 14-47. EVM and BER Computations Constellation Diagram Plots



To compute the EVM and BER results follow the steps below:

• In the RF panel of the Waveform Calculator click Evmber. The Error VectorMagnitude and Bit Error Rate window will be displayed.

• In the Source Waveforms field, add the constellation diagrams for wf0 and wf1. wf0should be specified as the reference waveform.

• In the Parameter Setup field select the Use Bit Error Rate MQAM Param checkbox andspecify the M parameter as 16.

• Figure 14-48 shows the Error Vector Magnitude and Bit Error Rate window with thewaveforms and M parameter specified. Click OK to calculate the results.

Figure 14-48. EVM and BER Computations EVM and BER Setup

• The results are displayed in the Rslts tab under “Expression Results”. The results shouldbe as follows:Phase Error = 2.50 degreesMag Error = 7.17%EVM = 8.33%BER MQAM = 2.89e-9

Eldo RF TutorialsTutorial #18—Load Pull Contours


Tutorial #18—Load Pull ContoursThe same amplifier is used in this example as in Tutorial #1. The netlist is identical, butcommands are added in order to compute Load Pull contours.

Load Pull contour is a set of points representing all the loads dissipating a given amount ofpower. The Load Pull contour is used to determine which load will dissipate the most power orto help match a load to a Power Amplifier for a maximum power transfer.

Load pull contours are obtained by putting a source at the output of the analyzed block and bysweeping the magnitude and the phase of this source in order to reach all the possiblefunctioning conditions. Once this has been performed the contours are the collection of pointscorresponding to specified situations, generally the same dissipated power or same inputreflection coefficient.


• SST analysis definition (.sst)

• Multiple parameter sweeps (source magnitude and phase)

• Contour extractions

• Plot output results (.plot contour)

Complete Netlist

*Contours.cir.include ampli.cktVdc VPOS 0 3.3XCKT IN LOAD VPOS AMPLI_CKT

Vin IN 0 RPORT=50 iport=1 FOUR fund1 PdBm (1) p1 -90Vout LOAD 0 RPORT=Rout iport=2 FOUR fund1 MA (1) mag2 ph_out

.TEMP 50.0

.OP

.param Rout=50

.param Z0 = 50

.param ph_out=-90

.sst fund1=900Meg nharm1=10

.param p1=-10

.param mag2=0.05

.defwave Zin=-V(in) / I(Vin)

.defwave Zout=V(load) / I(Vout)



.step param mag2 0.05 1.4 0.05

.step param ph_out 30 360 30



.extract fsst label=Zout_r yval(wr(Zout), fund1)

.extract fsst label=Zout_i yval(wi(Zout), fund1)

.extract fsst label=gamma_in yval(wm(gamma_in), fund1)

.extract fsst label=pm_out yval(pm(Vout), fund1)

.extract catvect sweep label=G80r xycond(meas(Zout_r),+ ==meas(gamma_in) == 0.80).extract catvect sweep label=G80i xycond(meas(Zout_i), meas(gamma_in)+ == 0.80)

.extract catvect sweep label=G85r xycond(meas(Zout_r),+ meas(gamma_in) == 0.85).extract catvect sweep label=G85i xycond(meas(Zout_i),+ meas(gamma_in) == 0.85)






** Display of Gamma_in (input reflexion coefficient) contours.plot contour meas(G80r) meas(G80i) (scattered).plot contour meas(G85r) meas(G85i) (scattered).plot contour meas(G90r) meas(G90i) (scattered).plot contour meas(G95r) meas(G95i) (scattered).plot contour meas(G100r) meas(G100i) (scattered).plot contour meas(G105r) meas(G105i) (scattered).plot contour meas(G110r) meas(G110i) (scattered)

.extract catvect sweep label=P90r xycond(meas(Zout_r),+ meas(pm_out) == 0.009).extract catvect sweep label=P90i xycond(meas(Zout_i),+ meas(pm_out) == 0.009)

.extract catvect sweep label=P95r xycond(meas(Zout_r),+ meas(pm_out) == 0.0095).extract catvect sweep label=P95i xycond(meas(Zout_i),



+ meas(pm_out) == 0.0095)





.extract catvect sweep label=P120r xycond(meas(Zout_r),+ meas(pm_out) == 0.0120).extract catvect sweep label=P120i xycond(meas(Zout_i),+ meas(pm_out) == 0.0120)** Display of Pm_out (output power) contours.plot contour meas(P90r) meas(P90i) (scattered).plot contour meas(P95r) meas(P95i) (scattered).plot contour meas(P100r) meas(P100i) (scattered).plot contour meas(P105r) meas(P105i) (scattered).plot contour meas(P110r) meas(P110i) (scattered).plot contour meas(P115r) meas(P115i) (scattered).plot contour meas(P120r) meas(P120i) (scattered)

Netlist Explanation

Vout LOAD 0 RPORT=Rout iport=2 FOUR fund1 MA (1) mag2 ph_out.step param mag2 0.05 1.4 0.05.step param ph_out 30 360 30

The above lines define the output source with a sweep of the magnitude and phase.

Note that the output source must be passive and must not affect the design functionality. Thiscan be verified in EZwave. The following graph contains all Zout_R waveforms: the real part ofZout value must be always positive or equal to 0.



Figure 14-49. Zout_R values Pull Load Contours Simulation Results

Defwave and extract commands allow to define the contour. Defwave and defmac commandsdefine the input impedance Zin, the output impedance Zout and the input reflection coefficientgamma_in.

.defwave Zin=-V(in) / I(Vin)

.defwave Zout=V(load) / I(Vout)



Extract commands allow to extract Zout and gamma_in at the fundamental frequency:

.extract fsst label=Zout_r yval(wr(Zout), fund1)

.extract fsst label=Zout_i yval(wi(Zout), fund1)

.extract fsst label=gamma_in yval(wm(gamma_in, fund1)

.extract fsst label=pm_out yval(pm(Vout),fund1)

Example of extraction of a contour corresponding to an input reflection coefficient equal to0.80:



.extract catvect sweep label=G80r xycond(meas(Zout_r),+ meas(gamma_in) == 0.80).extract catvect sweep label=G80i xycond(meas(Zout_i),+ meas(gamma_in) == 0.80).plot contour meas(G80r) meas(G80i) (scattered)

This contour will print all points in a scatter plot corresponding to Zout providing thatgamma_in is equal to 0.80.

Simulation Results

The following steps must be performed to obtain the load pull contour waves:

1. Load the Contours.wdb file into EZwave

2. Each wave has a real and imaginary part, plot the imaginary part against the real part. Todo this drag the plots for the input reflection coefficient G80R and G80I into the samewave window. Select the plot G80R with the right mouse button, and select Set as X axisin the pop up window. The G80I(G80R) will be displayed in a new wave window.Repeat this for G85I, G90I, G95I, G100I, G105I, and G110I.

3. In order to analyze the simulation results, all contours should be on the same graph.Drag and drop the plots of real vs imaginary (e.g. G80I(G80R)) into the same wavewindow, shown in Figure 14-50.



Figure 14-50. Load Pull Contours Simulation Results

4. Repeat steps one and two for output power plots P90I, P95I, P100I, P105I, P110I, P115Iand P120I.

5. Drag the plots created in step 4 into a new graph of the same wave window. The graphsfor the “output power” and the “input reflection coefficient”, shown Figure 14-51.



Figure 14-51. Final Waveforms—Load Pull Contours Simulation Results

Summary

The purpose of Load Pull contours is to determine which load will dissipate the most power. Forthis circuit, this point can be obtained from the second graph: the load pull contour for theoutput power.

The load corresponding to Re(Zout)=50 and Im(Zout)=40 dissipate the most power (center ofall contours).


Eldo RF TutorialsTutorial #19—SST Simulation using the .RFBLOCK Command

Tutorial #19—SST Simulation using the.RFBLOCK Command

The purpose of this tutorial is to show how the .RFBLOCK command can be used to reduce themulti-tone SST simulation time for large non-linear circuits.

The tutorial files are available in the directory: $MGC_AMS_HOME/examples/rfic/RFBLOCK/


• Steady-State simulation using the .RFBLOCK command


• Plot output results (.PLOT TSST and .PLOT FSST)

• Probe added in the circuit

Circuit Description

The schematic diagram of the circuit is shown in Figure 14-52. The circuit consists of a LowNoise Amplifier (LNA), a mixer, a Voltage Controlled Oscillator (VCO), a divider (divide bytwo), and a buffer. The VCO provides a 3.888GHz signal to the divider, therefore the frequencyof the LO signal at the input of the mixer is 1.944GHz. The 1.950GHz signal at the input of theLNA is down converted to 6MHz by the mixer.

Figure 14-52. Schematic diagram of the circuit



Analysis Description

The efficiency of the .RFBLOCK command is a function of the circuit as well as the SSTsimulation parameters. This example will show how changes in the input power will effect thesimulation time and the efficiency of the .RFBLOCK command to reduce the simulation time.

Four SST simulations are performed. Two SST simulations are performed with an input powerof -60dBm, the .RFBLOCK command is omitted from the netlist in the first simulation and usedin the second simulation. The goal is to compute the gain and the third-order intermodulationproduct of the RX part (LNA and mixer). This is a three-tone SST simulation. The first tone isthe frequency of the signal provided by the VCO connected to the divider. The second and thethird tones are the frequencies of the two signals provided by the RF port at the input of theLNA. Two SST simulations are performed with an input power of -20dBm, the .RFBLOCKcommand is omitted from the netlist in the first simulation and used in the second simulation.The goal is to study the effect of the input power on the gain of both the LNA and the mixer.There is only a single-tone at the input of the LNA.

IM3 Simulation

The circuit contains a VCO, therefore, the circuit is an autonomous circuit and must besimulated using the .SST OSCIL command. This command should be associated with the.SSTPROBE source.

The full netlist is shown in the file rxlo.spi. Extracts of the netlist used to simulate the IM3 areshown below:

*parameters.PARAM prf=-60.PARAM frf1=1950MEG frf2='frf1+1MEG'

*input sourceVRF RFP RFN RPORT=50 IPORT=1 FOUR FUND2 FUND3 PDBM (1,0) prf -90 (0,1)+ prf -90

*analysis.SST OSCIL NHARM_OSC1=20 FUND2=frf1 NHARM2=5 FUND3=frf2 NHARM3=5.SSTPROBE XVCO.LCP XVCO.LCN FUND_OSC1 RANK=2

To plot the TSST waveforms the SST_FULL_DISPLAY and SST_STOP options are used todefine the time window. The SST_TSTOP option is set to 170n which is one period of the 6MHzoutput signal.

*options.OPTION SST_VERBOSE=1 SST_FULL_DISPLAY=1 SST_TSTOP=170n

*extractions.EXTRACT FSST LABEL=FOSC FUND_OSC.EXTRACT FSST LABEL=GLNA YVAL(VDB(RFINP,RFINN),frf1)+ -YVAL(VDB(RFP,RFN),frf1).EXTRACT FSST LABEL=GMIX YVAL(VDB(OUTP,OUTN),frf1-FUND_OSC)+ -YVAL(VDB(RFINP,RFINN),frf1).EXTRACT FSST LABEL=VOUT_1 YVAL(VDB(OUTP,OUTN),frf1-FUND_OSC)



.EXTRACT FSST LABEL=VOUT_3 YVAL(VDB(OUTP,OUTN),(2*frf1-frf2)-FUND_OSC)

.EXTRACT FSST LABEL=IM3 MEAS(VOUT_1)-MEAS(VOUT_3)

.EXTRACT FSST LABEL=IIP3_DBM YVAL(PDBM(VRF),frf1) + MEAS(IM3)/2

*plots.PLOT TSST V(RFP,RFN) V(RFINP,RFINN) V(INP_DIV,INN_DIV).PLOT TSST V(LOP,LON) V(OUTP,OUTN) V(OUTP_DIV,OUTN_DIV).PLOT FSST V(RFP,RFN) V(RFINP,RFINN) V(INP_DIV,INN_DIV).PLOT FSST V(LOP,LON) V(OUTP,OUTN) V(OUTP_DIV,OUTN_DIV)

IM3 Simulation without the .RFBLOCK Command

When the circuit is simulated without the .RFBLOCK command, the options below are requiredto achieve convergence:

.OPTION SST_AT_TIME=200n

.OPTION SST_NDIM_FFT=1

.OPTION SST_CONVERGENCE_HELP=Advanced_Newton

The SST_AT_TIME option is used to set the transient duration, the SST_NDIM_FFT option isused to increase accuracy and reduce the aliasing problems, and the SST_CONVERGENCE_HELPoption is used to activate the Advanced Newton algorithm to help Steady-State convergence.

.OPTION SST_AT_TIME=200n

.OPTION SST_NDIM_FFT=1

.OPTION SST_CONVERGENCE_HELP=Advanced_Newton

To run the example perform the following:

eldo lnamixer_im3.cir

IM3 Simulation with the .RFBLOCK Command

The circuit is broken down into three blocks, this will help convergence and reduce thesimulation time. The first block consists of the VCO (where the .SSTPROBE is defined). Thesecond block consists of the most non-linear part of the circuit (divider associated to the buffer).The third block consists of all other parts of the circuit (i.e. LNA and the mixer). The commandsused are shown below:

.RFBLOCK NAME=VCO INST=(XVCO)

.RFBLOCK NAME=DIV_BUF INST=(XDIV,XBUF)

For the circuit simulated with the .RFBLOCK command specific options are required. TheSST_TRAN_NPER option sets the number of periods of FUND1 over which a transient analysis isperformed. The options EPS and RELTOL are used to set the internal simulator accuracy for thetransient analysis phase:

.OPTION SST_TRAN_NPER=15

.OPTION EPS=1u RELTOL=1u

To run the example perform the following:



eldo lnamixer_im3_rfblock.cir

Compression Simulation

The only differences between IM3 simulation (three-tone) and compression simulation(two-tone) is that the input port provides a single-tonexwavee signal thus FUND3 is notrequired. The full netlist is shown in the file rxlo_comp.spi. The following has been extractedfrom the netlist:

*input sourceVRF RFP RFN RPORT=50 IPORT=1 FOUR FUND2 PDBM (1) prf -90*analysis.SST OSCIL NHARM_OSC1=20 FUND2=1950MEG NHARM2=5

Two simulations are performed, one simulation contains the .RFBLOCK command the otherdoes not, as described in the sections IM3 Simulation with the .RFBLOCK Command and IM3Simulation without the .RFBLOCK Command respectively. To run the examples perform thefollowing:

eldo lnamixer_comp_rfblock.cireldo lnamixer_comp.cir

Simulation Results

Some of the simulation results for the IM3 simulation using .RFBLOCK command are shown inFigure 14-53 and Figure 14-54. Figure 14-53 shows the signal at the output of the mixer in thetime-domain (TSST). Figure 14-54 shows a fraction of the spectrum of the signal at the outputof the mixer (FSST).

Figure 14-53. Simulation Results — TSST



Figure 14-54. Simulation Results — FSST

The following shows the characteristics extracted from the simulation results:

EXTRACT for SST results FSST TEMPERATURE = 27.000 Celsius *FOSC = 1.944G *GLNA = 25.499 *GMIX = 10.556 *VOUT_1 = -34.841 *VOUT_3 = -108.689 *IM3 = 73.848 *IIP3_DBM = -23.277

Table 14-2 shows the simulation times for the four SST simulations. In this example, the.RFBLOCK command is efficient only for high input power simulation when the circuit is highlynon-linear.

NoteThe .RFBLOCK command associated to SST analysis is currently not supported for PLLsimulation.

Conclusion

The multi-tone SST simulation using the .RFBLOCK command may be a good way to reducesimulation time. However, the gain depends strongly on both circuit and simulation parameters.In this example, the .RFBLOCK command was very efficient when the circuit was under nonlinear condition (i.e. large input power).

Table 14-2. Simulation Times for Each Netlist

Without .RFBLOCK With .RFBLOCK

Pin=-60dBm, three-tone 6 mins 13 mins

Pin=-20dBm, two-tone 9 mins 2 mins

Eldo RF TutorialsTutorial #20—SST Simulation using the .SSTNLCONTRIB Command


Tutorial #20—SST Simulation using the.SSTNLCONTRIB Command

The purpose of this tutorial is to show how the Steady-State Non-linear Contributors analysis(.SSTNLCONTRIB), can be used to identify the non-linear devices that contribute non-linearity toeither a specified voltage output or a specified current through a voltage source.

Eldo RF Feature Used

• Steady-State simulation using the .SSTNLCONTRIB command

Complete Netlist

.GLOBAL

.PARAM LLOAD=1.9n

.PARAM CLOAD=2.7p

.PARAM tr=200p

.PARAM LMIX=0

.PARAM vlo=500m

.PARAM WQUAD=300u

.PARAM WDRIVER=300u

.PARAM VDD=2.5

.PARAM flo=1944MEG

.PARAM frf1=1950MEG

.PARAM frf2='frf1+1MEG'

.PARAM prf=-60

.LIB ./RFBLOCK/cmos.lib

.SUBCKT MIXER GNDI LON LOP OUTN OUTP RFN RFP VDDIC0 NET059 GNDI 1nI0 VDDI NET059 DC 500uR3 VDDI LON 10KR4 LON GNDI 20KR16 LOP GNDI 20KR2 VDDI OUTN 350R1 VDDI OUTP 350R15 VDDI LOP 10KL1 NET02 DBIAS LMIXL0 NET2 DBIAS LMIXXMSOU NET059 NET059 GNDI GNDI N_MOSFET Lm=0.5u Wf=25u N=2XMBIAS DBIAS NET059 GNDI GNDI N_MOSFET Lm=0.5u Wf=25u N=16 XMQD4 OUTN LON DDR2 DDR2 N_MOSFET Lm=0.24u Wf='WQUAD/20' N=20XMQD2 OUTN LOP DDR1 DDR1 N_MOSFET Lm=0.24u Wf='WQUAD/20' N=20XMQD1 OUTP LON DDR1 DDR1 N_MOSFET Lm=0.24u Wf='WQUAD/20' N=20XMDRN DDR2 RFN NET02 NET02 N_MOSFET Lm=0.24u Wf='WDRIVER/20' N=20XMQD3 OUTP LOP DDR2 DDR2 N_MOSFET Lm=0.24u Wf='WQUAD/20' N=20XMDRP DDR1 RFP NET2 NET2 N_MOSFET Lm=0.24u Wf='WDRIVER/20' N=20

.ENDS

.SUBCKT IND3 1 2C1 2 NET010 100fC0 1 NET012 100fR2 NET010 0 1KR1 NET012 0 1KR0 NET1 2 'LLOAD*1e9'



L0 1 NET1 LLOAD.ENDS

.SUBCKT IND2 1 2C1 2 NET010 100fC0 1 NET012 100fR2 NET010 0 1KR1 NET012 0 1KR0 NET1 2 1L0 1 NET1 1n

.ENDS

.SUBCKT LNA GNDI INN INP OUTN OUTP VDDIVMIDP OUTP NET033 DC 0V0 OUTN NET034 DC 0 I5 NET034 GNDI DC 1mI4 VDDI NET031 DC 150uI6 NET033 GNDI DC 1mXMBUFN VDDI NET058 NET034 GNDI N_MOSFET Lm=0.24u Wf=10u N=20 XMSOU NET031 NET031 GNDI GNDI N_MOSFET Lm=0.5u Wf=25u N=2XMBUFP VDDI NET036 NET033 GNDI N_MOSFET Lm=0.24u Wf=10u N=20XMDRP NET43 INP NET46 GNDI N_MOSFET Lm=0.24u Wf=10u N=20XMDRN NET35 INN NET29 GNDI N_MOSFET Lm=0.24u Wf=10u N=20XMCASP NET036 VDDI NET43 GNDI N_MOSFET Lm=0.24u Wf=10u N=20XMCASN NET058 VDDI NET35 GNDI N_MOSFET Lm=0.24u Wf=10u N=20 R1 INN NET031 50KR0 INP NET031 50KC0 NET031 GNDI 1nC3 VDDI GNDI 100pC1 VDDI NET036 CLOADC2 VDDI NET058 CLOADXI3 NET058 VDDI IND3XI2 NET036 VDDI IND3XI0 GNDI NET29 IND2XI1 GNDI NET46 IND2

.ENDS

XMIX 0 LON LOP OUTN OUTP RFINN RFINP NET029 MIXERROUT OUTN OUTP 10KC0 NET028 NET026 300fC1 RFP RFN 1.4pL2 RFN NET026 8nL0 RFP NET028 8nVRF RFP RFN RPORT=50 IPORT=1 FOUR FUND2 FUND3 PDBM (1,0) prf -90 (0,1) prf+ -90XLNA 0 NET028 NET026 RFINN RFINP NET013 LNAV1 LOP LON PULSE ( vlo -vlo 1n tr tr '0.5/flo-tr' '1/flo' )VDDMIX NET029 0 DC VDDVDDLNA NET013 0 DC VDD.OPTION NOASCII AEX NUMDGT=3 ENGNOT SST_NDIM_FFT=1.OP.DC.SST FUND1=flo NHARM1=20 FUND2=frf1 NHARM2=5 FUND3=frf2 NHARM3=5.EXTRACT FSST LABEL=GLNA YVAL(VDB(RFINP,RFINN),frf1)-+ YVAL(VDB(RFP,RFN),frf1).EXTRACT FSST LABEL=GMIX YVAL(VDB(OUTP,OUTN),frf1-flo)+ - YVAL(VDB(RFINN, RFINP),frf1).EXTRACT FSST LABEL=S11_dB YVAL(SDB(1,1),frf1)



.EXTRACT FSST LABEL=IIP3LNA_DBM YVAL(PDBM(VRF),frf1)+ + (YVAL(VDB(RFINP,RFINN),frf1)-YVAL(VDB(RFINP,RFINN),2*frf1-frf2))/2.EXTRACT FSST LABEL=IIP3_DBM YVAL(PDBM(VRF),frf1)+ + (YVAL(VDB(OUTP,OUTN),frf1-flo)-YVAL(VDB(OUTP,OUTN),(2*frf1-frf2)-+ flo))/2

.SSTNLCONTRIB V(OUTP,OUTN) HARM(1,2,-1) SORT_NBMAX=4

.OPTION SSTNLCONTRIB_FILE=lnamix.sstnl

.ALTER

.PARAM LLOAD=1.7n

.PARAM CLOAD=2.0p

Circuit and Analysis Description

The schematic of the whole circuit is shown in Figure 14-55. The circuit consists of a LowNoise Amplifier (LNA) (shown in Figure 14-56), and a mixer (shown in Figure 14-57). Thegoal is to compute the gain and the third-order intermodulation product of the circuit. This is athree-tone SST simulation. The first tone is the frequency of the signal provided by the LOsource to the mixer (FLO=1944MHz). The second and the third tones are the frequencies of thetwo signals provided by the RF port at the input of the LNA (FRF1=1950MHz,FRF2=1951MHz).

Figure 14-55. Schematic of the circuit



Figure 14-56. Schematic of the LNA

Figure 14-57. Schematic of the Mixer

In this example, the devices that are responsible for the third-order intermodulation product(2×FRF1-FRF2) at the output of the circuit are listed in a specific output file using the optionSSTNLCONTRIB_FILE. For the purpose of this example the results are written to the filelnamix.sstnl:

.OPTION SSTNLCONTRIB_FILE=lnamix.sstnl

The node and the frequency where the contribution analysis is performed is specified on the.SSTNLCONTRIB command. For the purpose of this example only four devices are listed:

.SSTNLCONTRIB V(OUTP,OUTN) HARM(1,2,-1) SORT_NBMAX=4



Two simulations are performed, in the first simulation the gain of the LNA is set high and in thesecond simulation it is set low. The purpose of this is to identify the effect that the gain of theLNA has on the non-linearity of the output of the whole circuit. To run the example perform thefollowing:

eldo lnamixer.cir

Simulation Results

The following has been extracted from the results file lnamix.sstnl. It shows the characteristicsextracted from the netlist for the two LNA’s that were used to quantify the effect of the gain ofthe LNA on the non-linearity at the output of the whole circuit:

EXTRACT for SST results FSST TEMPERATURE = 27.000 Celsius ALTER index 0 *GLNA = 25.854 *GMIX = 10.524 *S11_DB = -13.406 *IIP3LNA_DBM = -13.938 *IIP3_DBM = -24.457

EXTRACT for SST results FSST TEMPERATURE = 27.000 Celsius ALTER index 1 *GLNA = 16.348 *GMIX = 10.508 *S11_DB = -14.225 *IIP3LNA_DBM = -5.163 *IIP3_DBM = -14.571

The following shows the non-linear contributors. The first part (ALTER index 0) is dedicated tothe analysis results using a high gain LNA. The second part (ALTER index 1) is dedicated tothe analysis results using a low gain LNA. For the circuit using the low gain LNA, the mixerdoes not contribute a significant amount of non-linearity to the output when compared with theLNA.

************ SSTNLCONTRIB RESULTS ************

TEMPERATURE = 27.000 Celsius ALTER index 0

**************************************************************


.SSTNLCONTRIB command # 1, Output: V(OUTP, OUTN) HARM(1,2,-1)

**************************************************************

subckt instance XMIX : 110.48%



subckt instance XMQD2 : 24.41% device instance MAIN : 24.87% device instance DJDB_AREA : -0.36% device instance DJDB_PERIM : -0.11%




subckt instance XMDRP : 5.16% device instance MAIN : 5.02% device instance DJDB_AREA : 0.10% device instance DJDB_PERIM : 0.03%

subckt instance XLNA : -10.48%

subckt instance XMDRP : 40.64% device instance MAIN : 40.45% device instance DJDB_AREA : 0.15% device instance DJDB_PERIM : 0.04%

subckt instance XMDRN : 40.64% device instance MAIN : 40.45% device instance DJDB_AREA : 0.15% device instance DJDB_PERIM : 0.04%

subckt instance XMBUFN : -24.34% device instance MAIN : -24.41% device instance DJSB_AREA : 0.06% device instance DJSB_PERIM : 0.02%

subckt instance XMBUFP : -24.34% device instance MAIN : -24.41% device instance DJSB_AREA : 0.06% device instance DJSB_PERIM : 0.02%

subckt instance XMCASP : -21.54% device instance MAIN : -23.13% device instance DJDB_AREA : 1.12% device instance DJDB_PERIM : 0.35% device instance DJSB_AREA : 0.10% device instance DJSB_PERIM : 0.03%

************ SSTNLCONTRIB RESULTS ************

TEMPERATURE = 27.000 Celsius ALTER index 1



**************************************************************


.SSTNLCONTRIB command # 1, Output: V(OUTP, OUTN) HARM(1,2,-1)

**************************************************************

subckt instance XLNA : 202.84%

subckt instance XMDRN : 48.90% device instance MAIN : 49.15% device instance DJDB_AREA : -0.19% device instance DJDB_PERIM : -0.06%

subckt instance XMDRP : 48.90% device instance MAIN : 49.15% device instance DJDB_AREA : -0.19% device instance DJDB_PERIM : -0.06%

subckt instance XMCASN : 36.45% device instance MAIN : 36.56% device instance DJSB_AREA : -0.19% device instance DJDB_AREA : 0.11% device instance DJSB_PERIM : -0.06% device instance DJDB_PERIM : 0.03%

subckt instance XMCASP : 36.45% device instance MAIN : 36.56% device instance DJSB_AREA : -0.19% device instance DJDB_AREA : 0.11% device instance DJSB_PERIM : -0.06% device instance DJDB_PERIM : 0.03%

subckt instance XMBUFN : 16.06% device instance MAIN : 16.19% device instance DJSB_AREA : -0.10% device instance DJSB_PERIM : -0.03%

subckt instance XMIX : -102.84%

subckt instance XMQD2 : -18.51% device instance MAIN : -18.82% device instance DJDB_AREA : 0.24% device instance DJDB_PERIM : 0.07%






subckt instance XMDRN : -15.08% device instance MAIN : -15.03% device instance DJDB_AREA : -0.04% device instance DJDB_PERIM : -0.01%

Conclusion

The .SSTNLCONTRIB analysis can be used to detect the possible origins of unwanted spuriousspectral components. With these results, you can try to reduce them by modifying parts of thedesign that have been identified by the analysis, for example, the subckt instance XMIX.

Eldo RF TutorialsTutorial #21—Multitone Large Signal S Parameters Extraction


Tutorial #21—Multitone Large Signal SParameters Extraction

The same amplifier is used in this tutorial as in Tutorial #14—Verilog-A Usage. The netlist issimilar but additional commands are included to compute multitone S parameter extraction.

It is possible to extract large signal S parameters for any fundamental frequency and harmonicfor the input and the output ports. The extraction of large signal S parameters of a circuitcontaining an oscillator and a down-converting-mixer is illustrated in this tutorial.

In this example, Port 1 is specified at the fundamental frequency fund2 (that is, the RFfrequency) at one input of the Low Noise Amplifier (this is described in the Verilog-A tutorial)and Port 2 is specified at the IF frequency (fund2-fund_osc) at the output of the mixer.

Figure 14-58 show the synoptic of the circuit.

Figure 14-58. Synoptic of Oscillator and Down-Converting Mixer

Complete Netlist

The complete netlist is in the file vcomixerlna_S_parameter.cir in$MGC_AMS_HOME/examples/rfic.



Netlist Explanation

Explanations of the Verilog-A model and circuit have already been covered in Tutorial 14, see“Tutorial #14—Verilog-A Usage” on page 357, therefore only the multitone S parametersextraction is explained here. The commands are at the end of the netlist.

* S parameters plots.plot fsst Sdb(1,1) Sdb(2,1) Sdb(1,2) Sdb(2,2)

The .plot command activates the computation of the large signal S parameters Sxx and Sxy,where the incident wave a1 is at port 1 at the first harmonic of the second fundamentalfrequency fund2. The reflected wave b1, and the transmitted wave b2 are computed at allfrequencies of the spectrum. The x-axis of the computed S parameters corresponds to thefrequencies of the corresponding reflected or transmitted wave.

* S parameters extraction.extract fsst yval(S(2,1),fund_osc-fund2).extract fsst yval(S(1,1),fund2)

The first .extract command extracts the S21 (transmission coefficient) value in db at the IFfrequency. The second .extract command extracts the S11 (reflection coefficient) value at theRF frequency.

* input capacitor sweep.param Rinput=30.param cinput=30p.step param cinput dec 10 0.1p 100p

The .step command allows you to see the effects of a changing circuit parameter on the Sparameter.

Simulation Results

Figure 14-59 shows the S21 db value versus frequency, Figure 14-60 shows the S11 parameteron a Smith chart and Figure 14-61 shows the S21 parameter on a polar chart. The high value ofS21 is 25.3 dB at IF frequency (equal to Flo - Frf).

To view the output as shown in Figure 14-59, open the vcomixerlna_S_parameter.wdb file inEZwave. In the Waveform List panel of the EZwave window, navigate to the directory FSSTand drag and drop the plot S(2,1)_1.



Figure 14-59. S21 Value in dB Versus Frequency

To obtain the results shown in Figure 14-60, drag and drop YVAL(S(1,1), FUND2) in thededicated extract folder. Right-Click on the curve and choose the Smith Chart transformation.

Figure 14-60. S11 Plot on Smith Chart

To obtain the results shown in Figure 14-61, drag and drop YVAL(S(2,1), FUND_OSC-FUND2) in the dedicated extract folder. Right-Click on the curve and choose the Polar Charttransformation.



Figure 14-61. S11 Plot on Polar Chart

Conclusion

Eldo RF enables the computation of large signal S parameters with a circuit containing anoscillator and a down-converting mixer. With the appropriate extract command andtransformation in EZwave, it is very easy to see the variation of the S parameter at anyfrequency versus a parameter of a circuit.


Chapter 15Eldo RF Tutorial—Mixer Simulations

ObjectiveThe objective of this three-part extended tutorial is to describe the simulation setup that isneeded to analyze the behavior of an RF mixer using Eldo RF.

• Part I describes the basic setup required to perform a functional simulation.

• Part II describes the simulation of third-order intermodulation (IM3) and intercept points(IP3).

• Part III describes mixer noise and noise figure (NF) simulations.

Part I shows how to ‘import’ an existing netlist (circuit description) into Eldo RF. This is a verycommon way for an open point tool such as Eldo RF to be used.

Also, the note shows how to setup parametric analyses in order to verify the performance over arange of operating conditions.

The design aspects are not covered by this note. The mixer is assumed to be available as aregular SPICE subcircuit.

This extended tutorial guides the reader through a step-by-step procedure which shows how tosetup an Eldo RF simulation and extract the relevant information from the results.

This note can be considered as a complement to the standard product tutorials (these tutorialscan be found in the documentation), and also the netlists described in this extended tutorial mayserve as ‘templates’ for your own simulations, as they cover very common simulations.

Ready-to-run netlists are associated with the extended tutorial, to illustrate the simulations.

Part I

ArchitectureThe mixer is one of the basic building blocks in RF transceivers. In the receive path, its basicfunction is to ‘downconvert’ the input signal from a high frequency (RF) to a lower outputfrequency (also known as the intermediate frequency IF), prior to demodulation. The inputsignal has already gone through the antenna, a duplexer filter, maybe another filter, and a lownoise amplifier before it is presented to the mixer. The lower output frequency is obtained bymultiplying the input signal by a periodic signal at a frequency LO, usually generated by a


Eldo RF Tutorial—Mixer SimulationsPart I

frequency synthesizer. The output frequency is equal to the difference between the inputfrequency (‘RF’) and the ‘LO’ frequency. Depending on the application, this may be the finaloutput frequency, or there may be additional downconversion stages to further reduce thefrequency.

We will assume that the mixer description is contained in a regular SPICE subcircuit. We willalso assume that all ports are differential ports. For example the RF signal is applied betweenpins RF_P and RF_N. The power supplies are labeled VSS and VDD. Of course, there may beadditional pins, such as bias voltages for example.

Mixer circuit.subckt MIXER RF_P RF_N LO_P LO_N OUT_P OUT_N VSS VDD*circuit netlist…*….ENDS

The simulation netlist, mixer.cir, will instantiate the mixer using the following syntax:


X1 RF_P RF_N LO_P LO_N OUT_P OUT_N VSS VDD MIXER

We will continue to build our netlist incrementally from this point. Each newly added elementwill be shown in bold characters (as the X1 instantiation statement above).

We will now connect a load (50 Ohms) and the power supplies. We simply need a resistor andgrounded DC voltage sources. We chose 3V as the power supply value. It is easy to adapt toanother power supply value of course.


LO

RF OUT




RLOAD OUT_P OUT_N 50VSS VSS 0 DC 0VVDD VDD 0 DC 3V

The first simulation we want to run with our mixer is a simple functional simulation, where wewill verify that the downconversion operation actually occurs inside the mixer. To setup such asimulation, we need to drive the RF input and the LO input with the appropriate signals, and tosetup the simulation parameters.

The Input SignalsWhat kind of signals do we want? For the RF input, we need a sinusoidal signal, with a givenfrequency and a given amplitude, or equivalently power. Same thing for the LO input.

Let us start with the LO input. Assume we want a 900MHz signal with a 50mV amplitude. Also,assume we want a 50 Ohm resistance for this input voltage source. Translating thesespecifications into a netlist statement for the simulator, we obtain:

VLO LO_P LO_N rport=50 four 900Meg ma (1) 50m –90

How does this read? This statement instantiates a special kind of voltage source, dedicated toRF operation. The name of the source is VLO. It is connected at the LO port, between nets LO_P

and LO_N. Note that this is a ‘floating’ voltage source (i.e. none of the pins is grounded). Next,we give the resistance value (50 Ohms) with the rport keyword (rport=50). The fourkeyword indicates that the source is a special ‘Fourier’-type source. Fourier sources are sourceswhose value is the sum of fundamental frequencies and their harmonics, as in a Fourier series. AFourier source can represent rather complex signals, the description of which is embedded in asingle statement. In this case, this is pretty simple, as we want a single frequency (900 MHz),and we specify this with ‘900Meg’. Then we specify the amplitude of the signal by giving itsamplitude (in Volts) and its initial phase (in degrees). The ma keyword is a format selector, andit stands for magnitude. The surprising (1) indicates that we are providing the amplitude(50mV) and phase (-90 deg) for the harmonic 1 of the signal. The –90 factor is here tocompensate for the fact that, by default, the signals in the Fourier source are co-sinusoids,instead of the more usual sinusoids.

Similarly, we can describe the RF input:

VRF RF_P RF_N rport=50 four 930Meg ma (1) 5m –90

This is not very different from the LO signal. This time we connect the source at the RF port ofcourse, and we specify also a single-tone signal at 930MHz, with an amplitude of 5mV.

Here is how our netlist looks like now:

Mixer circuit.subckt MIXER RF_P RF_N LO_P LO_N OUT_P OUT_N VSS VDD*circuit netlist…



*….ENDS





Steady-State Analysis—A bit of TheoryNow, we need to specify the analysis we want to run with this circuit. As a first analysis, we justwant to look at the output spectrum, to check that the intermediate frequency (IF=930MHz-900MHz=30MHz in this case) is generated by our mixer.

We will do so by specifying the parameters of a steady-state analysis. The steady-state analysismay be considered as an additional analysis of the simulator, very much like transient, dc, smallsignal, noise etc.

In the case of a steady-state analysis, the time-domain solution (i.e. the input voltages, the nodevoltages and the device currents) is assumed to be a Fourier series. Each voltage waveform iswritten as a Fourier series, that is a sum of complex functions such as:

where is a frequency and are, respectively, an amplitude and a phase. Of course theFourier series is truncated, which means that the summation is carried over a finite number offrequencies (otherwise the simulator would have an infinite number of unknowns to solve for).We will detail the truncation mechanisms a little bit later.

Intermodulation ProductsThe frequencies are chosen to include the fundamental frequencies, a finite number ofharmonics of these fundamental frequencies, and a limited number of said intermodulationproducts, i.e. frequencies which are linear integer combination of these fundamentalfrequencies. These harmonics and intermodulation frequencies appear in the spectrum of theinternal and output voltages because of circuit non-linearities. If the circuit is absolutely linear,no harmonics and spurious products would appear. When doing a SPICE ‘.AC’ analysis, i.e. asmall-signal analysis, the basic assumptions are that the circuit can be linearized around the DCbias point, and that all signals are sufficiently ’small’ to be represented as sinusoids at a singlefrequency. The SPICE .AC analysis cannot be used to analyze harmonics nor intermodulationproducts.

V t( ) ai expj 2π f i t ϕ i+⋅ ⋅( )

⋅i

∑=

f i ai ϕ i,



In our mixer example, we have two (natural) fundamental frequencies, which are, respectively,the RF and the LO frequencies. This means that the voltage spectra for the circuit will containenergy at the fundamental frequencies (900MHz and 930MHz), at their harmonics (1800MHz,2700MHz, 3600MHz… and also 1860MHz, 2790MHz, etc.), and also at frequencies that arelinear integer combinations of the fundamentals (930MHz-900MHz=30MHz,930MHz+900MHz=1830MHz, 2*930MHz-900MHz=960MHz, etc.)

If we want to write this a little bit more formally, in the case of two fundamental frequencies,this is the way it looks like (when the two fundamental frequencies are not harmonically related,as is often the case, such signals are called ‘quasi-periodic’ signals):

where indexes i and j cover a finite range, defined by what is called the ‘truncation’ model. Thetruncation model is the way to limit the number of index combinations, that is the number ofindividual harmonics and intermodulation products which are considered.

Truncation ModelsOne way to limit the number of frequencies is to impose the simultaneous conditions:

, where N and M are integers.

If we choose N=M=5 for example, it means we include in the spectrum all frequencies such as:

, with , and , and

This is called the ‘box’ truncation model, because if we plot the frequencies in a two-dimensional diagram, it looks roughly like a rectangle (square in this case). Note that onlypositive frequencies are retained.

Another common way to limit the number of frequencies is to impose the conditions:

, where N and M are integers.

If we choose M=4 and N=5 for example, it means we include in the spectrum all frequenciessuch as:

, with , , and , and

This is called the ‘diamond’ truncation model, because if we plot the frequencies in a two-dimensional diagram, it looks roughly like a diamond-shaped area (we’ll need a little bit ofimagination though). Note that only positive frequencies are retained in the simulator.

The ‘diamond’ truncation model is the one used by Eldo RF by default. To use the ‘box’truncation model, we need to enter an option directive in the netlist:

V t( ) ai j, expj 2π f i j, t ϕ i j,+⋅ ⋅( )

⋅i j,∑=

f ij i f i⋅ j f 2⋅+=( )

i M j N≤,≤

f ij i f 1⋅ j f 2⋅+= i 5≤ j 5≤ f ij 0≥

i M j N i j+ max M N,( )≤,≤,≤

f ij i f 1⋅ j f 2⋅+= i j+ 5≤ i 4≤ j 5≤ f ij 0≥



.option sst_spectrum=1

The ‘box truncation’ model with M=N=5 and The ‘diamond truncation’ model with M=4 and N=5

The figures above depict symbolically the frequency selection (truncation) models.

Steady State Analysis ParametersBack to our mixer, let us specify the steady state analysis we want to run:

.SST fund1=900Meg nharm1=5 fund2=930Meg nharm2=5

How does this read? The .SST keyword stands of course for Steady State, this is the analysisspecification. The fund1/nharm1 and, respectively, fund2/nharm2 keywords are used tospecify the frequencies of the independent tones we want to use in the circuit, and the respectivenumber of harmonics we want to consider. In our case, we have two tones. The first tone is at900MHz, this is the LO input frequency. The second tone is at 930MHz, and this is the RFfrequency.

In the case of multi-tone simulations (a mixer simulation is a two-tone simulation – LO andRF), it is recommended to use the first fundamental for the strongest signal, here the LO, and touse the other fundamental for the weaker signals (which in this case is the RF). This is notmandatory but it often helps convergence.

Why did we choose to specify 5 harmonics? This is more or less a rule of thumb. The properselection of the number of harmonics to consider is based on accuracy and performanceconsiderations. The more harmonics in a signal, the more accurate its representation can be,particularly if the signal has a significant high-frequency spectral content. Also, increasing thenumber of harmonics for each tone increases the number of intermodulation products includedin the output spectrum. As more harmonics are requested, the simulator will need more memoryand more time to determine all the Fourier coefficients.

This is even worse in the case of a two-tones analysis, as the number of frequencies (and thusthe number of unknowns) grows roughly as k.N2, as can be easily seen from the truncationmodel figures above.

i i

j j



Selecting 5 harmonics is usually a good initial choice. It may be necessary to increase thisnumber if the simulator has difficulty to converge, or if the circuit is highly non-linear.However, this must be done carefully to avoid an excessive growth of the memory usage.

The number of harmonics does not need to be the same for all tones. In many ‘real’ mixers , theoperation with respect to the LO signal is much more non-linear than the operation with respectto the RF. So it may be desirable to use more harmonics along the LO direction (say 6 or 7), andless along the RF (say 3 or 4).

Here is our new netlist now:







In the above netlist, the numerical values of the frequencies are repeated both in the sourcespecification (VRF and VLO) and in the analysis statement (.SST). This is not so important, butthis is obviously an error-prone methodology.

Fortunately, we can use symbols rather than explicit numerical values in the sourcespecifications. The fund1 and fund2 symbols automatically designate the frequenciesspecified in the .SST analysis, and they can be used in the source specifications.

NoteThe .SST statement can appear after the source specifications.

Here is our modified netlist, including this modification:



RLOAD OUT_P OUT_N 50



VSS VSS 0 DC 0VVDD VDD 0 DC 3V

VRF RF_P RF_M rport=50 four fund2 ma (1) 5m –90

VLO LO_P LO_M rport=50 four fund1 ma (1) 50m –90


This way, the explicit frequency values 930MHz and 900MHz appear once only.

We are almost done with the setup of the simulation. The only thing we need to add arestatements which specify what simulation outputs we want to look at.

Exactly like for other analysis types, such as transient or DC, we will use .PLOT statements togenerate the graphical outputs we want to analyze.

The only thing we want to look at is the output spectrum:

.PLOT FSST VDB(OUT_P,OUT_N)

The familiar .PLOT statement will allow us to graphically analyze the output spectrum. TheFSST keyword stands for Frequency-Steady-State, it is similar to the TRAN, DC and ACkeywords used for other regular analysis. We specify VDB() to plot the differential outputvoltage in dB.

Here is our final netlist:




VRF RF_P RF_N rport=50 four fund2 ma (1) 5m –90

VLO LO_P LO_N rport=50 four fund1 ma (1) 50m –90


.PLOT FSST VDB(OUT_P,OUT_N)

Deriving a ‘grounded’ version (where all ports are referenced to ground) is straightforward:

Mixer circuit.subckt MIXER RF_P RF_N LO_P LO_N OUT_P OUT_N VSS VDD*circuit netlist…*…



.ENDS

X1 RF 0 LO 0 OUT 0 VSS VDD MIXER

RLOAD OUT 0 50VSS VSS 0 DC 0VVDD VDD 0 DC 3V

VRF RF 0 rport=50 four fund1 ma (1) 5m –90

VLO LO 0 rport=50 four fund2 ma (1) 50m –90


.PLOT FSST VDB(OUT)

The file mixer.sst.cir contains a similar netlist.

Running this simulation will generate the kind of plot shown below.

Also interesting is the intermodulation table which is generated in the ASCII output file,mixer.sst.chi. This table lists all the intermodulation products which are computed by thesimulator. In this case, the default diamond truncation model was used. The table shows theactual frequencies present in the signals, and the linear combinations that produce these



frequencies. For example, the second line in the table, which is shown in bold, corresponds tothe IF frequency at 30MHz.

******* FREQUENCY & INTERMODULATION TABLE *******

FUND1: 900000000.0Hz NHARM1: 5 FUND2: 930000000.0Hz NHARM2: 5

FREQ FUND1 FUND2 0.0 : 0 0

30000000.0 : -1 1 60000000.0 : -2 2 840000000.0 : 3 -2 870000000.0 : 2 -1 900000000.0 : 1 0 930000000.0 : 0 1 960000000.0 : -1 2 990000000.0 : -2 3 1770000000.0 : 3 -1 1800000000.0 : 2 0 1830000000.0 : 1 1 1860000000.0 : 0 2 1890000000.0 : -1 3 2670000000.0 : 4 -1 2700000000.0 : 3 0 2730000000.0 : 2 1 2760000000.0 : 1 2 2790000000.0 : 0 3 2820000000.0 : -1 4 3600000000.0 : 4 0 3630000000.0 : 3 1 3660000000.0 : 2 2 3690000000.0 : 1 3 3720000000.0 : 0 4 4500000000.0 : 5 0 4530000000.0 : 4 1 4560000000.0 : 3 2 4590000000.0 : 2 3 4620000000.0 : 1 4 4650000000.0 : 0 5

Conclusion of Part IThis concludes the first part of this extended tutorial. Parts II and III describe simulations ofthird-order intermodulation (IM3 and IP3) and mixer noise respectively.

Eldo RF Tutorial—Mixer SimulationsPart II


Part II

Simulation of Third-Order IntermodulationOne of the problems which gives headaches to designers of RF amplifiers and mixers is thethird-order intermodulation. As explained in Part I of this extended tutorial, intermodulationhappens as soon as the circuit exhibits some degree of non linearity.

Intermodulation can affect amplifiers as well as mixers. For amplifiers, the basic phenomenon isthat two signals with frequencies, say f1 and f2, will generate output frequencies at (2.f1-f2) and(2.f2-f1). If frequencies f1 and f2 are close to each other, the third-order intermodulationproducts at 2.f1-f2 and 2.f2-f1 will be close to f1 and f2 too, and may well fall right within thedesired channel and corrupt it (‘corruption’ here has to be understood as a degradation of thesignal-to-noise ratio). These particular products are troublesome because they appear in thechannel and cannot be filtered out. Fith-order products (at 3.f2-2.f1 and 3.f1-2.f2) can also be aproblem, but they are usually much lower than the third-order ones.

Intermodulation products involving higher harmonics (3.f1-f2 for example), even if undesirable,can usually be filtered by subsequent filtering stages in a receive or transmit path of an RFsystem. The two close signals at f1 and f2 could be the signal of interest and an adjacent channel‘interferer’.

For mixers, the basic phenomenon is that two signals within the RF frequency band withfrequencies, say f1 and f2, will generate output frequencies at fLO – (2.f1-f2) and fLO – (2.f2-f1). If f1 and f2 are close to each other, fLO – 2.f1-f2 and fLO – 2.f2-f1 can fall within the IFfrequency band, degrading the signal-to-noise ratio of the communication system. So it is nearlyalways a design goal to minimize the levels of these spurious intermodulation products, whichmost of the time implies controlling the non-linearity of the component or block.

Third-order intermodulation is such a common problem that standard (in the world of RFdesign) ways of characterizing this effect exist and are widely accepted. Figures such as IM3,IIP3 and OIP3 are part of the specifications of many RF elements. This extended tutorial willexplain how Eldo RF can be put at work to extract these specifications in the case of a mixer.

Three-Tone Steady-State AnalysisAs we are working with a mixer, we will need a three-tone analysis. The first two tones will beused for the within-band interferers. The third tone will be used for the LO signal.



The .SST command we need to specify these fundamental frequencies is very similar to theone we developed in Part I. We simply added a third tone.

We chose to assign 930MHz and 931MHz to model the interferers, and to keep the LO at900MHz for this analysis.

.SST fund1=900Meg nharm1=4 fund2=930Meg nharm2=4+ fund3=931Meg nharm3=4

Input SignalsIn Part I, we learnt how the signals are represented for steady-state analysis. In particular, weexplained that signals (and inputs) are expressed as sums of sinusoidal signals.

To model our two interferers, we want to create a signal such as:

(1)

In Eldo RF, each FOUR voltage source can carry such a sum of elementary signals. In fact, wecan use a FOUR source to inject any linear combination of the fundamental frequencies wespecify in the .SST analysis. Most of the time, this feature is only partially used, to create suchsimple sums as in (1) above, or to deliberately inject a signal presenting harmonic distortion, butthis is rather handy.

Here is the way we will model our RF source:

VRF RF 0 rport=50 four fund2 fund3+ ma (1,0) 5m –90 (0,1) 5m –90

The fund2 and fund3 keywords indicate that the actual value of the voltage source will bespecified by a sum of linear combinations of the fund2 and fund3 frequencies from the .SST

command. Thus the (1,0) notation actually designates 1.fund2+0.fund3, that is fund2 infact. Similarly, (0,1) designates fund3. After each (i,j) pair, we give the associatedamplitude and angle. In our case, the two interferers have the same amplitude (5mV), and the

RF

LO

OUT

VRF a1 2π fund2t⋅( )sin⋅ a2 2π fund3t⋅( )sin⋅+=



same angle (-90 degrees). The source still has a 50 Ohms impedance, specified with the rport

keyword.

The LO signal is similar to the LO signal used in Part I, except we have assigned the LOfrequency as being the first tone (fund1):


Here is our new netlist:






.SST fund1=900Meg nharm1=4 fund2=930Meg nharm2=4 fund3=931Meg nharm3=4

.PLOT FSST VDB(OUT)

Before we continue, we will introduce parameters for the fundamental frequencies. As we maywant to make them vary or do some computations such as 2.fund2-fund3, etc. it will be saferand easier. We simply introduce .param statements for the frequencies and use the symbolsrather than literal values in the .SST command:




.param f1=930Meg

.param f2=931Meg

.param flo=900Meg





.SST fund1=flo nharm1=4 fund2=f1 nharm2=4 fund3=f2 nharm3=4

.PLOT FSST VDB(OUT)

The file mixer.im3.cir contains a similar netlist.

Running this type of simulation produces an output spectrum like the one shown below. Wenotice that many intermodulation products are generated. Those appearing at high frequencieswill be filtered by subsequent stages in a real system.

However, those appearing in the vicinity of the IF, at 30MHz in our case, may be a problem.Thus we want to measure how small these spurious intermodulation products are. The IM3measure is shown on the plot below, which is simply a zoomed version of the previous one,around IF frequency. We clearly see the third-order intermodulation products, and we canmeasure the distance relative to the IF level as being roughly 70dB.



This ‘rough’ measurement technique is not fully satisfactory. It would be much nicer to have anexact ‘extraction’ of the wanted result computed by the simulator, rather than rely on a tediousmanual operation.

Automatic Measurement of the IM3We can achieve that with the .extract command of Eldo RF (this command actually belongsto Eldo, and has been extended to support the FSST analysis).

Before we write the full extraction command, let us detail the .extract command a little bit.The .extract command has the following general form:

.extract <analysis> label=<username> <expression>

The <analysis> keyword indicates from which results the quantity must be extracted. In ourcase we want to extract from an FSST waveform. The label keyword allows us to provide aname for the quantity we want to extract. This name will be used to refer the quantity in otherexpressions if needed. In our case IM3 seems adequate. The <expression> is a math.expression where waveform values, operators, numbers and functions can be used.



One of the most useful functions we can use in the <expression> above, is the yval()function. It is used to retrieve a value from a waveform. The yval(<wave>, <x>) functiontakes two arguments. It returns the value of the wavefom <wave> when the abcissa is <x>. Inour case, assuming we want to retrieve the value of VDB(OUT) at the IF, the waveform namewould be VDB(OUT), and the x value would be 30MHz, so we would use something likeyval(VDB(OUT), 30Meg).

So, using this information, we may write the following extraction command to compute the IM3value:

.extract FSST label=IM3+ yval(vdb(out), f1-flo) – yval(vdb(out), 2*f1-f2-flo)

Although it’s a little bit hard to read, this is actually pretty straightforward. We simply computethe difference between the output level at the IF and the intermodulation product at2*f1-f2-flo.

We use the symbols we have defined as symbols (f1, f2 and flo) in the expression, whichmakes the statement ‘portable’ if we want to modify one of the frequencies.

Eldo RF will compute this expression, and log the result to the output ASCII file, mixer.im3.chi.

Automatic Measurement of the Third-Order InterceptPoint (IP3)

The IM3 itself is certainly useful as a measure of linearity. However, the third-order interceptpoint (IP3) is at least as useful and is present in the specifications of RF blocks. At low inputsignal levels, the power of the main output exceeds that of third-order intermodulation products.However, as the input power level increases, the third-order products grows much more rapidlythan the main output. The IP3 is defined as the point at which the extrapolated main output andthe extrapolated third-order product reach the same level.

The following diagram illustrates the typical behavior of third-order intermodulation products:



It is relatively easy to show that, typically, the amplitude of third-order products grows as thecube of the input signal amplitude (for low input levels, where behavior can be linearized).Interested readers are refered to virtually any RF textbook1,2 for a demonstration of this fact.Plotted on a log scale, this means that the slopes of the main output and the third-order productare 1 and 3 respectively.

With a little bit of geometry, it is easy to compute an estimate of the IP3 point from a single IM3measurement realized at a (low) power input level P0.

The main output can be written as:

Y1 = O1 + PIN – P0

Input power (dBm)

Outputpower(dBm)

IIP3

OIP3

IP3

Mainoutput

Third-orderoutput

Outputpower(dBm)

IIP3

OIP3

IP3

Mainoutput

Third-orderoutput

P0

O1

O3Inputpower(dBm)

IM3



Where PIN is the input power level.

The third-order output is:

Y3 = O3 + 3(PIN – P0)

Putting Y1=Y3 and solving for PIN yields:

O1 + PIN – P0 = O3 + 3(PIN-P0)

And thus:

PINIP3 = IIP3 = (O1 – O3)/2 +P0 = IM3/2 + P0

PINIP3 or IIP3 is the ‘input-referred’ third-order intercept point.

Computing the output-referred intercept point is equally straightforward:

POUTIP3 = OIP3 = O1 + IIP3 – P0 = O1 + IM3/2

These computations are only valid if the chosen P0 level is sufficiently ‘small’ of course.

Let us see how we can translate this into a simulation. As a first step we will modify our netlistso that we can easily manipulate power values instead of amplitudes:




.param f1=930Meg

.param f2=931Meg

.param flo=900Meg

.param p0=-40

VRF RF 0 rport=50 four fund2 fund3+ pdbm (1,0) p0 –90 (0,1) p0 –90



.PLOT FSST PDBM(RLOAD)



We have picked a value of –40dBm for the input power, and we made this value a parametrizedone thanks to a .param statement. We have changed the format for the RF input fromma-gnitude to pdbm. Also we now plot the output power rather than the amplitude in dB, usingthe pdbm(RLOAD) which reads ‘the power dissipated in resistor RLOAD, expressed in dBm’.

Now let us add the .extract commands which will compute the IM3, IIP3 and OIP3 values:




.param f1=930Meg

.param f2=931Meg

.param flo=900Meg

.param p0=-40





.extract FSST label=O1 yval(vdb(out), f1-flo)

.extract FSST label=O3 yval(vdb(out), 2*f1-f2-flo)

.extract FSST label=IM3 meas(O1) – meas(O3)

.extract FSST label=IIP3 meas(IM3)/2 + P0

.extract FSST label=OIP3 meas(O1) + meas(IM3)/2

It is very easy to follow the computations in the .extract commands above. They are nothingbut a direct translation from the theory. We have split the computations into simple ones, whichis usually much more readable than using a single three-lines-long formula.

Notice that when reusing an extracted result in a .extract expression, we need to use themeas() function. This is a requirement from the equation parser in Eldo. So we have to write:


rather than:

.extract FSST label=OIP3 O1 + IM3/2



which would be more intuitive.

The results are printed in the output ASCII file, toward the end of the file.

The extraction mechanism is flexible, and this is rather handy when we need to do some kind of‘customized’ calculus. For example, we could compute the IM3 as the difference between themain output level and the average level of the two third-order products at 2.f1-f2 and 2.f2-f1. Until now, we had picked 2.f1-f2 for our estimation, but this was arbitrary. The twoproducts usually have levels which are close, but this is not always true, so using the averagevalue or the max() of the two can make sense.

Here is the netlist which uses the average value:




.param f1=930Meg

.param f2=931Meg

.param flo=900Meg

.param p0=-20






.extract FSST label=O3a yval(vdb(out), 2*f1-f2-flo)

.extract FSST label=O3b yval(vdb(out), 2*f2-f1-flo)

.extract FSST label=IM3 meas(O1) –(meas(O3a) +meas(O3b))/2



The file mixer.im3.auto.cir contains a similar netlist.



Parametric (Sweep) AnalysisUntil now, we have extracted ‘nominal’ performance value. For example we did not sayanything about the temperature. By default, the temperature is set to 27 degrees Celsius by thesimulator, so all our IM3 and IP3 extractions were done for T=27C.

What happens if the temperature is 0C or 100C? We need to verify the behavior over a range oftemperature values.

To do so we will simply make the temperature a design parameter, and use the regular .stepcommand to perform the analysis over a user-specified range of parameter values.

Making the temperature a design parameter is easy:

.param xtemper=27

.temp xtemper

Sweeping the temperature is also easy:

.step param xtemper list 0 20 40 60 80 100

We can keep the same .extract commands as before. The simulator will now produce twooutput files. The .wdb files will contains all the runs (6 in this case). The .wdb file will containmore interesting data. Each quantity extracted through a .extract command is plotted versusthe .step parameter, i.e. the temperature in this case. So we get IM3 (temperature) directly forexample. The .wdb file is generated by Eldo and can be read by EZwave.

Here is the netlist:




.param f1=930Meg

.param f2=931Meg

.param flo=900Meg

.param p0=-20








.extract FSST label=O3a yval(vdb(out), 2*f1-f2-flo)

.extract FSST label=O3b yval(vdb(out), 2*f2-f1-flo)

.extract FSST label=IM3 meas(O1) –(meas(O3a) +meas(O3b))/2



.param xtemper=27

.temp xtemper

.step param xtemper list 0 20 40 60 80 100

This technique allows us to parameterize any design variable and to verify the behavior of thecircuit when the variable covers a specified range, usually imposed by the circuit specifications.

To complete our verification, the next thing we will probably want to do is to extract themaximum IM3 value when, for example, the temperature varies . This is probably the value wewould put in a datasheet.

We can achieve that by inspection of the IM3 waveform in the .wdb file, but again it would bepreferable to automate the procedure.

To automatically extract this maximum value, we will use a specific form of the .extractcommand, in which we use the sweep keyword for the analysis:

.extract sweep label=DS_IM3 max(meas(IM3))

The max() function used in the command above returns the maximum value of the argument foreach run (i.e. each temperature, corresponds a certain IM3 value, and we want to extract themaximum value over the whole set of runs).

The extracted value is written at the very end of the ASCII output file (.chi).

Conclusion of Part IIThis concludes Part II. We have seen how to extract usual specifications such as third-orderintermodulation products (IM3), third-order intercept points (IP3). We have seen how tocustomize the extraction if required. And we have also learnt how to run parametric analyses.The next part will focus on mixer noise analysis.

Eldo RF Tutorial—Mixer SimulationsPart III


Part III

Simulation of the Mixer Noise—IntroductionMixers tend to be ‘noisy’ devices. This is one of the main reasons why low noise amplifiers areusually needed in the first stages of a receiver. Knowing the noise generated by the mixer allowsthe computation of interesting figures such as the noise figure or noise factor, and also toestimate the sensitivity of the receiver. Also, one often used figure of merit is the so-called‘spurious-free dynamic range’ or SFDR value.

In this part we will examine how to extract these figures from various simulations. First it isimportant to understand what ‘noise’ the simulator computes. Apart from capacitors andinductors (the ideal ones), devices tend to generate ‘noise’. Different physical mechanisms areinvolved, and it is not the intent of this tutorial to review the theory. Briefly, the noise generatedby the devices is usually modeled as an equivalent noise spectral density, which is a frequencydependant function. Thermal noise, shot noise, flicker noise are all noise sources due todifferent physical mechanism, but they can be modeled as noise spectral densities, allowing tocompute how much noise (as an rms square voltage) the device generates in a given bandwidth.

With a SPICE-type simulator, noise analysis is performed in conjunction with the small signalanalysis. The circuit is linearized around its DC operating point, and the contribution of all noisesources is summed at a given output. All node voltages and currents are assumed to be smallsinusoidal signals, all at the same frequency.

Although valuable for the analysis of linear circuits, this is clearly not an appropriate modelingof what happens inside an RF mixer. Eldo RF provides a steady-state noise analysis which isbased on the steady-state behavior of the circuit, rather than its DC operating point. The circuitis ‘linearized’ around the large-signal time-varying operating point, and the noise contributions(the amplitudes of the noise signals) are assumed to be small enough that this linearization isvalid. The device noise models and equations are then used to compute the correlation matrix ofthe noise sources at user-specified frequencies. The simulator actually computes the frequenciesthat get translated to these output frequencies due to the non-linear behavior of the circuit. Thefundamental difference with SPICE-type noise analysis is that the phenomenon of frequencytranslation is fully handled.

Simulation of the Mixer Output NoiseThe steady-state noise analysis is associated with a specific keyword, SSTNOISE. Setting upthe analysis requires specifying the output where we want to investigate the noise spectrum, andthe frequency range of interest.

For example, we could set:

.SSTNOISE V(OUT) LIN 100 1Meg 100Meg



This specifies a steady-state noise analysis. Node OUT will be the summation point. For adifferential output, we could have used V(OUT_P, OUT_N) instead. The output frequencies wewant to look at are in the vicinity of the IF (30 MHz). Here, we will compute 100 frequencypoints, from 1MHz to 100MHz.

NoteThe syntax is very similar to the classical .NOISE and .AC statements of SPICE.

For a noise analysis, we don’t need the three-tones setup from part II. A simple two-tones isenough (actually, we could even connect the RF input to zero without changing much theoutput).




.param f1=930Meg

.param f2=931Meg

.param flo=900Meg

.param rf_amp=1mV

VRF RF 0 rport=50 four fund2 ma (1) rf_amp –90


.SST fund1=flo nharm1=4 fund2=f1 nharm2=4


.PLOT SSTNOISE ONOISE DB(ONOISE)

With the .PLOT command shown above, we will plot the total output noise (in V per sqrt(Hz))and the same quantity in dB.

Sorted Contributions of the DevicesIt is often interesting to know which device or which harmonic is responsible for the outputnoise, or to what extent. Eldo RF provides output tables which contain such information. Eachdevice is listed, sorted by its contribution, and for each device we may know which harmonic isresponsible for which amount of noise. Also the contribution per harmonic is given. This data,although voluminous, is highly useful when it comes to size devices for example.



Below are some excerpts from the .chi file when a .SSTNOISE analysis was run:

MU NOISE PSD (SQ V/Hz) : 9.4059E-20 ( 6.5% of ONOISE) MOST IMPORTANT HARMONIC CONTRIBUTIONS

TOTAL (0) (-1) (1) (2) (-2)RD 0.0000E+00 0.0000E+00 0.0000E+00 0.0000E+00 0.0000E+000.0000E+00RS 0.0000E+00 0.0000E+00 0.0000E+00 0.0000E+00 0.0000E+000.0000E+00ID 9.4018E-20 9.4018E-20 4.1457E-27 3.7292E-27 5.0999E-33 4.9556E-34FN 4.1453E-23 4.1453E-23 3.8552E-28 5.4882E-29 5.8080E-36 1.3003E-35TOTAL 9.4059E-20 9.4059E-20 4.5312E-27 3.7841E-27 5.1057E-335.0857E-34

MB1 NOISE PSD (SQ V/Hz) : 4.3197E-22 MOST IMPORTANT HARMONIC CONTRIBUTIONS

TOTAL (0) (-1) (1) (-2) (2)RD 0.0000E+00 0.0000E+00 0.0000E+00 0.0000E+00 0.0000E+000.0000E+00RS 0.0000E+00 0.0000E+00 0.0000E+00 0.0000E+00 0.0000E+000.0000E+00ID 4.3195E-22 4.3195E-22 6.7367E-28 1.8456E-29 3.1321E-34 7.8026E-37FN 2.6638E-26 2.6638E-26 1.9804E-31 5.7032E-34 2.7476E-38 2.4005E-41TOTAL 4.3197E-22 4.3197E-22 6.7387E-28 1.8456E-29 3.1324E-347.8029E-37

**** TOTAL OUTPUT NOISE VOLTAGE (SQ V/Hz) Total : 1.44898E-18

CONTRIBUTION FROM THE HARMONICS: (-10) : 1.90775E-50 ( -9) : 4.42883E-51 ( -8) : 1.62908E-50 ( -7) : 1.04240E-51 ( -6) : 2.18357E-50 ( -5) : 3.94955E-50 ( -4) : 9.63689E-44 ( -3) : 3.08649E-38 ( -2) : 3.43621E-31 ( -1) : 6.18331E-25 ( 0) : 1.44898E-18 (100.0% of ONOISE) ( 1) : 8.44023E-26 ( 2) : 1.91524E-32 ( 3) : 3.68917E-39 ( 4) : 5.70062E-44 ( 5) : 1.24117E-49 ( 6) : 1.07857E-50 ( 7) : 9.53155E-51 ( 8) : 1.52791E-50 ( 9) : 1.69096E-50



( 10) : 7.38521E-51

Noise Figure and Noise FactorA more interesting figure is the Noise Figure (Nf), which is a measure of the degradation of thesignal-to-noise ratio from the input to the output. This can be easily computed by the simulatortoo. A specific directive, .SNF allows this. The definition of the noise figure is a ratio:

Nf = (total_noise – noise_from_the_load) / (noise_from_the_input)

The input noise must be computed from the input resistance(s) (the port resistance of VRF in ourcase), and the noise due to the load must be computed from the load resistor (RLOAD).

.SNF INPUT=(VRF) OUTPUT=(RLOAD) INPUT_SIDEBAND=(-1,0)

This command will compute the Noise Figure using the information supplied about whichresistor is the input (source) resistor or port, and which resistor is the output (load) resistor.

The INPUT_SIDEBAND parameter requires some explanations. The computation of the noisefigure for a mixer has different definitions. The Single Side Band (SSB) and the Double SideBand (DSB) noise figures coexist in the literature. To accommodate these definitions, Eldo RFlets you define which ‘sidebands’ are to be considered in the computation of the denominator inthe Nf expression above (the ‘noise_from_the_input’).

Let us review the theory of operation first. The non-linear steady-state noise analysis computesthe noise spectrum at so-called ‘output noise frequencies’ Fout (the frequencies we list on the.SSTNOISE command line). In the case of a mixer, we are usually interested in the noisespectrum around the IF frequency.

The noise at these output noise frequencies results from the ‘folding’ of the noise at manydifferent frequencies, i.e.

In the case of a mixer, the fi frequencies are the harmonics of the LO. A mixer causes frequency‘translation’, and the noise spectrum are translated, with different transfer functions, fromfi+Fout to Fout. The figure below depicts the ‘paths’ followed by the noise spectrum around thefirst two harmonics of f0.

f i Fout+ i f 0 Fout+⋅=

Fout 1.f0+Fout 2.f0+Fout



However when computing the noise figure for a mixer, not all sidebands are considered as noisesources. In particular, the SSB and DSB definitions differ in the way the denominator, in the Nfexpression above, is handled.

For both SSB and DSB, the only frequencies considered as noise sources when computing the‘noise_from_the_source’ term in the denominator are the input and image sidebands. Higherorder harmonics are not considered as contributing to the input source noise.

For a standard SSB calculation, only the input sideband is included in the frequencies that createthe contribution of the input source to the noise. For a DSB calculation, not only the inputsideband is included, but also the image sideband (the symetrical with respect to f0) is included.Usually the DSB figure is thus about 3dB lower than the SSB figure (though it does not have tobe exact, because the noise spectrum and the transfer functions for the input and the image donot have to be strictly identical).

For an SSB figure, the treatment of the sidebands for the input source in the denominator isshown above.

For a DSB figure, the treatment is different, it includes the image sideband as well, as shownabove.

The INPUT_SIDEBAND keyword allows you to list which sidebands must be considered in thecalculation, and thus lets you achieve either SSB or DSB figures.

The designation of the sidebands corresponds to the indexes (k1,k2) in k1.fund1+k2.fund2+fout.

The LO frequency is the ‘fund1’ in our case. As we have a two-tone setup we need to designatethe frequencies with two integer indexes, one for fund1 and one for fund2.

Fout

Fout



So the upper sideband (the image) would be designated by (1,0). The upper sideband indeed isat 1.fund1+0.fund2+fout.

However for the lower sideband (the input), this is less easy. The upper sideband is atfund1-fout, which is not in the form ‘k1.fund1+k2.fund2+ fout’. So we use the harmonic –1 ofthe LO to designate this sideband (-1.fund1+0.fund2+fout). This is a negative frequency, but thisis equivalent.

So, for an SSB figure, we would use the following statement (including only the inputsideband):

.SNF INPUT=(VRF) OUTPUT=(RLOAD) INPUT_SIDEBAND=(-1,0)

But for a DSB figure, we would use the following statement (including both input and imagesidebands):

.SNF INPUT=(VRF) OUTPUT=(RLOAD) INPUT_SIDEBAND=((-1,0),(1,0))

If you want to use your own definition, you can use any list you like, those above are simply thelists needed to obtain the conventional definitions (SSB or DSB).

Whatever the definition of SNF that we choose, plotting this SNF quantity uses a .plotcommand:

.PLOT SSTNOISE SNF




.param f1=930Meg

.param f2=931Meg

Fout 1.fund1+0.fund2+Fout-1.fund1+0.fund2+Fout

imageinput



.param flo=900Meg

.param rf_amp=1mV

VRF RF 0 rport=50 four fund2 ma (1) rf_amp –90


.SST fund1=flo nharm1=4 fund2=f1 nharm2=4


.PLOT SSTNOISE ONOISE DB(ONOISE)

.SNF INPUT=(VRF) OUTPUT=(RLOAD) INPUT_SIDEBAND=(-1,0) ! SSB

.PLOT SSTNOISE SNF

The netlist above generates a unitless quantity, and is called the Noise Factor in manytextbooks. The Noise Figure is defined by these same textbooks as the Noise Factor expressedin dB:

Nfig = 10.log(Nfact)

To obtain this Noise Figure, we could use the following code:

.defwave nfig=db(snf)

.plot sstnoise w(nfig)

The .defwave command defines a new waveform, which can be printed (.print) or plotted(.plot). It must be referred to with the w() function, as shown above. The db() functionreturns decibels.

NoteThe definitions of Noise Figure and Noise Factor are sometimes swapped by someauthors1.

Computing the SensitivityFrom the Noise Factor, we can compute the sensitivity of the mixer. The sensitivity is theminimum input signal level which can be detected with an acceptable output signal-to-noiseratio.

The standard formula1 for a matched input is:

S = -174dBm + Nfactor + 10.log(BW) + SNRmin(dB)

Integrating this equation in our netlist so that we obtain the sensitivity is straightforward (thevalues for the bandwith (100K) and the minimum SNR (12dB) are just examples – they arecompletely application-dependant of course)

.param BW = 100K



.param SNR_min = 12

.defwave nfactor=db(snf)/2

.extract sstnoise label=sensitivity+ -174+yval(w(nfactor), f1)+10*log10(BW)+SNR_min

The value is given in dBm.

Computing the Spurious Free Dynamic RangeAnother interesting figure we can extract from our mixer simulation is an estimate of thespurious-free dynamic range. As various definitions are used, we’ll start by making clear whatwe mean by SFDR, and then we will illustrate this with a graphic, which will make it clearerstill.

When the input signal is varied from a very low level to higher and higher levels, the behaviorof the mixer changes radically. At first, the output level is ‘buried’ into the noise, i.e. it is belowthe output noise floor. For a given input level, the output level starts to raise from the noise floorlevel. This is the minimum input level (L1) the component can handle. If the mixer is welldesigned, there will be a certain range of input levels where the output level will growproportionally, without distortion.

From this L1 level, as the input level continues to increase, several phenomenon will take place.Spurious output frequencies will start to appear, and (usually later) the gain will start to drop(i.e. compression will occur). One interesting input level (L2) corresponds to the moment whenthe level of the third-order intermodulation products reaches the level of the noise floor. Thedefinition of the third-order SFDR we will use is the difference between these two input levels:SFDR = L2 - L1

Obviously the dynamic range provided by this computation is rather theoretical, and we need todecrease it by a minimum SNR value to obtain a meaningful value. This way the SFDR depictsthe input range which provides a usable signal (minimum SNR), yet undistorted by third-orderproducts.

The graphic below depicts the graphical construction for the theoretical SFDR.



The derivation of the SFDR follows:

The main output can be written as:

Y1 = O1 + PIN – P0

Where PIN is the input power level.

The third-order output is:

Y3 = O3 + 3(PIN – P0)

L1 corresponds to:

NFLOOR = O1 + L1 – P0

Whereas L2 corresponds to:

NFLOOR = O3 + 3(L2 – P0)

Thus we have:

L2 – L1 = (NFLOOR – O3)/3 + P0 – NFLOOR + O1 – P0L2 – L1 = SFDR = (NFLOOR – O3)/3 – NFLOOR + O1

First we need to obtain an estimate of the noise floor. To do so, we will integrate the outputnoise. The Eldo integ() function allows integration of a waveform. We will need to convertthe output noise, onoise, into a power spectral density (PSD=onoise*onoise/RLOAD), and tointegrate it.

This is achieved with the following code (we put the result in dBm, thus the 10.log() factor):

.defwave PSD=onoise*onoise/50

Noise Floor

Outputpower(dBm)

L2

Third-orderoutput

SFDR

L1



.extract sstnoise label=NFLOOR_DBM+ 10 * log10(1e3 * integ(w(PSD), 1Meg, 100Meg))

Then computing the SFDR is straightforward – this is a direct translation from the theoreticalequation we just derived:

.extract sstnoise label=SFDR+ (meas(NFLOOR_DBM)-meas(O3))/3 –meas(NFLOOR_DBM) +meas(O1)

Combining all this into a netlist, we obtain:




.param f1=930Meg

.param f2=931Meg

.param flo=900Meg

.param p0=-20




.extract FSST label=O1 yval(pdbm(RLOAD), f1-flo)

.extract FSST label=O3 yval(pdbm(RLOAD), 2*f1-f2-flo)


.defwave PSD=onoise*onoise/50

.extract sstnoise label=NFLOOR_DBM+ 10 * log10(1e3 * integ(w(PSD), 1Meg, 100Meg))

.extract sstnoise label=SFDR+ (meas(NFLOOR_DBM)-meas(O3))/3 –meas(NFLOOR_DBM)+ meas(O1)

Note that we changed the way O1 and O3 are extracted, to have everything expressed in dBm.

For a more realistic simulation, we should have included the bandpass filter centered on the IFfrequency which usually follows the mixer.

This will combine the results of the intermodulation simulation (.SST) with those from the noiseanalysis (.SSTNOISE) and provide a result with a single simulation. However, it may be more



effective to perform the noise analysis with less tones, to speed-up the computations. In thiscase we would first run a noise analysis with the RF input connected to ground (this will notchange the output noise much anyway) to extract the noise floor value, and then to use the resultwithin the three-tones IM3 analysis. The drawback is that it requires manual re-entry of thenoise floor value into the netlist of the second simulation. The advantage is a considerably fastersimulation, particularly if the number of noise frequencies is large.

This type of simulation illustrates the flexibility of the simulator to extract miscellaneousfigures of merit. Once the concept of .extract is fully assimilated, it is very easy to performany kind of complex extraction.

Conclusion of Part IIIThis concludes Part III. We have seen how to extract usual specifications such Noise Figure,Noise Factor, Sensitivity and Spurious-Free Dynamic Range.

References1. RF Microelectronics. Behzad Razavi. Prentice Hall. 1998.

2. The design of CMOS radio-frequency integrated circuits. Thomas H. Lee. Cambridge.1998.


Chapter 16Eldo RF Tutorial—Port Impedance and

Admittance Modeling

ObjectiveThis tutorial was written for those who need a Thévenin or Norton equivalent circuit (i.e.resistor-capacitance, resistance-inductance, conductance-capacitance or conductance-inductance) of a port impedance or admittance, respectively. This may happen if a blockdesigner does not want to include the netlist of a given block in a simulation, but would still liketo get realistic models of the load and/or source. Using a lumped element equivalent of a portimpedance or admittance can also be useful to analyze stability issues (tracking a nodeimpedance variation over frequency). Yet another example would be to maximize an impedancein a tank circuit, in order to maximize voltage swing.

The document is organized in 8 short sections, each section showing sample netlist code.

Sections 1 and 2 deal with port impedance modeling using small-signal (AC) analysis, andlarge-signal analysis (SST), respectively. Sections 3 and 4 describe the equivalent case ofadmittance modeling. In section 5, previous results are applied to the case of a low noiseamplifier (LNA). Particularly, it is shown how large-signal Z or Y parameters relate to small-signal parameters at low input levels.

Finally, sections 6 and 7 show how to use S-parameter files, to take into account the dependenceon the frequency.

1—Converting Port Impedance intoThévenin Equivalent Circuit (AC Analysis)

To obtain an equivalent lumped circuit representation of the impedance, we use the Zparameters analysis of Eldo. The real part of the impedance is simply ZR(1,1) (the real part ofZ(1,1)). Depending on the sign of ZI(1,1), we compute an equivalent series capacitance or seriesinductance, at the frequency of interest.

* Converting port impedance into Thévenin equivalent circuit* (ac-domain)

* ammeter allows zero voltage sources* aex creates a separate data file,* <simu_name>.aex containing the .extract results* engnot print data in engineering format* noascii do not plot the curves (in ascii format) into .chi-file


Eldo RF Tutorial—Port Impedance and Admittance Modeling2—Converting Port Impedance into Thévenin Equivalent Circuit (SST Analysis)

.option ammeter aex engnot noascii

* define the frequency of interest.param frequ=1G

* voltage source and the example circuitV1 1 0 iport=1 rport=50Rload 1 2 50Cload 2 0 25pLload 2 0 1n

* analysis.ac dec 10 100meg 10G

* extracted lumped resistance value at the frequency of interest.extract ac label=”R_serial” yval(zr(1,1),frequ)* plot the resistance over the defined frequency range.plot ac zr(1,1)

* extracted lumped reactance value at the frequency of interest.defwave cap_ind=+ eval(zi(1,1)>0?zi(1,1)/(2*3.14159*freq):1/(zi(1,1)*2*3.14159*freq)).extract ac+ label=” C_serial if<0 or L_serial if>0 “ yval(w(cap_ind),frequ)

* plot the reactance value over the defined frequency range.plot ac zi(1,1).plot ac wr(cap_ind).end

2—Converting Port Impedance intoThévenin Equivalent Circuit (SST Analysis)

Z parameters (as well as S and Y parameters) are also available in large-signal analysis (with theSST analysis of Eldo RF). The setup is similar to the small-signal analysis.

* Converting port impedance into Thévenin equivalent circuit* (sst-domain)

* ammeter allows zero voltage sources* aex creates a separate data file, <simu_name>.aex* engnot print data in engineering format* noascii do not plot the curves (in ascii format) into .chi-file.option ammeter aex engnot noascii


* voltage source and the example circuitV1 1 0 iport=1 rport=50 four frequ PdBm (1) 0 -90Rload 1 2 50Cload 2 0 25pLload 2 0 1n

* analysis

Eldo RF Tutorial—Port Impedance and Admittance Modeling3—Converting Port Admittance into Norton Equivalent Circuit (ac-domain)


.sst fund1=frequ nharm1=5

* extracted lumped resistance value at the frequency of interest.extract fsst label="R_serial" yval(zr(1,1),frequ)* extracted lumped reactance value at the frequency of interest.extract fsst label="zi" yval(zi(1,1),frequ).extract fsst label="C_serial if<0 or L_serial if>0"+ eval(meas(zi)>0?meas(zi)/(2*3.14159*frequ):1/+ (meas(zi)*2*3.14159*frequ))

.end

3—Converting Port Admittance intoNorton Equivalent Circuit (ac-domain)

In this section, we use the same ideas to obtain a Norton equivalent (i.e. an admittance) ratherthan a Thévenin equivalent.

* Converting port admittance into Norton equivalent circuit* (ac-domain)



* voltage source and the example circuitV1 1 0 iport=1 rport=50Rload 1 0 50Cload 1 0 25pLload 1 0 1n

* analysis.ac dec 10 100meg 10G

* extracted lumped resistance and conductance value at the frequency ofinterest.defwave res=1/yr(1,1).defwave cond=yr(1,1).extract ac label=”R_parallel” yval(w(res),frequ).extract ac label=”G_parallel” yval(w(cond),frequ)

* plot the resistance and conductance over the defined frequency range.plot ac wr(res).plot ac wr(cond)

* extracted lumped admittance value at the frequency of interest.defwave cap_ind=+ eval(yi(1,1)>0?yi(1,1)/(2*3.14159*freq):1/(yi(1,1)*2*3.14159*freq)).extract ac+ label=”C_parallel if>0 or L_parallel if<0” yval(w(cap_ind),frequ)


Eldo RF Tutorial—Port Impedance and Admittance Modeling4—Converting Port Admittance into Norton Equivalent Circuit (SST Analysis)

* plot the admittance value over the defined frequency range.plot ac yi(1,1).plot ac wr(cap_ind)

.end

4—Converting Port Admittance intoNorton Equivalent Circuit (SST Analysis)

* Converting port admittance into Norton equivalent circuit* (sst-domain)



* voltage source and the example circuitV1 1 0 iport=1 rport=50 four 1G PdBm (1) 0 -90Rload 1 2 50Cload 2 0 25pLload 2 0 1n

* analysis.sst fund1=1G nharm1=5

* extracted lumped resistance and conductance values at the frequency ofinterest.defwave res=1/yr(1,1).extract fsst label="R_parallel" yval(w(res),frequ).extract fsst label="G_parallel" yval(yr(1,1),frequ)* extracted reactance value at the frequency of interest*.extract fsst label="yi" yval(yi(1,1),frequ).extract fsst label="C_parallel if>0 or L_parallel if<0"+ eval(meas(yi)>0?meas(yi)/(2*3.14159*frequ):1/+ (meas(yi)*2*3.14159*frequ))

.end

5—Input Impedance and Admittance of a LNAA simple LNA (Figure 16-1) was analyzed as an application of the techniques developed insections 1 to 4. We also show the convergence between small-signal and large-signal models,when the input level is small enough.

Eldo RF Tutorial—Port Impedance and Admittance Modeling5—Input Impedance and Admittance of a LNA


Figure 16-1. Schematic of LNA

The frequency of analysis is 2GHz.

The input-referred 1dB compression point of the LNA is -10dBm.

We have extracted small-signal impedance and admittance values, and also the large signalvalues. At -50dBm input level, the LNA is operating almost linearly, and thus the large-signal(SST) model gives nearly the same result as the small signal (AC) model. At -10dBm, the LNAhas entered compression, and the large-signal waveforms are distorted, thus the large-signalmodel gives different results.

* Thévenin equivalent of the input impedance of the lna in ac-domain "R_SERIAL" = 43.2459 "C_SERIAL IF<0 OR L_SERIAL IF>0" = -132.7260P

* Thévenin equivalent of the input impedance of the lna in sst-domain, *Pin = -50dBm "R_SERIAL" = 43.2526 "ZI" = -598.8904M "C_SERIAL IF<0 OR L_SERIAL IF>0" = -132.8750P

* Thévenin equivalent of the input impedance of the lna in


Eldo RF Tutorial—Port Impedance and Admittance Modeling6—How to Save S-Parameters into a File?

* sst-domain, Pin = -10dBm "R_SERIAL" = 47.0004 "ZI" = -3.0056 "C_SERIAL IF<0 OR L_SERIAL IF>0" = -26.4764P

------------------------------------------------------

* Norton equivalent of the input admittance of the lna in ac-domain "R_PARALLEL" = 43.2542 "G_PARALLEL" = 23.1191M "C_PARALLEL IF>0 OR L_PARALLEL IF<0" = 25.5066F

* Norton equivalent of the input admittance of the lna in ac-domain,* Pin = -50dBm "R_PARALLEL" = 43.2609 "G_PARALLEL" = 23.1155M "YI" = 320.0654U "C_PARALLEL IF>0 OR L_PARALLEL IF<0" = 25.4700F

* Norton equivalent of the input admittance of the lna in ac-domain,* Pin = -10dBm "R_PARALLEL" = 47.1926 "G_PARALLEL" = 21.1898M "YI" = 1.3551M "C_PARALLEL IF>0 OR L_PARALLEL IF<0" = 107.8318F

6—How to Save S-Parameters into a File?Eldo can also export the S parameters of a block (typically a one-port or two-port network) to afile, and later reuse this file for another simulation (see section 8). The file is saved inTouchstone format. The S parameter file is essentially a table that lists the S parameter values(either in real/imaginary or in db/phase etc. format) as a function of the frequency.

* How to save s-parameters into a file?

* ammeter allows zero voltage sources.option ammeter

* voltage source and the example circuitV1 1 0 iport=1 rport=50Rload 1 2 50Cload 2 0 25p

* analysis.ac dec 10 100Meg 1G

* stores the S parameters into the sb1.par using the MHz as the unit* for frequency and magnitude-phase as the format for the S parameters.Ffile S sb1.par mhz ma

* plot s11.plot ac sdb(1,1)

.end

Eldo RF Tutorial—Port Impedance and Admittance Modeling7—How to use a Saved S-Parameter File as Part of the Circuit?


7—How to use a Saved S-Parameter File as Partof the Circuit?

Eldo and Eldo RF can ‘read’ a small-signal S parameter file, in Touchstone format. The filemay have been saved by Eldo itself (see section 6), or generated by a measurement equipment.Thus it is very easy to ‘replace’ a given block with its S (or Y or Z) parameter description, andto simulate it. Transient, DC, AC and SST analyses are supported. Of course the ‘replaced’block has to be reasonably linear for the results to be accurate.

* How to use a saved s-parameter file as a part of the circuit?

* ammeter allows zero voltage sources.option ammeter

* voltage source and the example circuitV1 1 0 iport=1 rport=50

* analysis.ac dec 10 100Meg 1G

* includes saved s-parameters as a part of the circuit* sb1.par must be located in the simulation directory.model fblock macro lang=cyblock fblock 1 0 param: idx_f=1* number one is equivalent to number one in the sb1.par

* note: another way topoint to the file is to use:* yblock fblock 1 0 param: STRING: sb1.par

* plot s11 (waveform is identical to the one in section 7).plot ac sdb(1,1)

.end

Further InformationFor more information, please read “Working with S, Y, Z Parameters” on page 193, and theEldo RF Tutorials chapter.


Eldo RF Tutorial—Port Impedance and Admittance ModelingFurther Information


Chapter 17ADMS RF Tutorial—AGC Loop

IntroductionThe objective of this tutorial is to provide an introduction to Questa ADMS RF (ADMS RF),using an example for which the tool brings real benefits in terms of performance and usability.

In this tutorial ADMS RF will be used to efficiently simulate an automatic gain control (AGC)loop. The system is driven with a digitally modulated signal (GMSK). The different blocks inthis loop are described in SPICE (transistor-level), in Verilog-A (behavioral), and in VHDL.The gain control is performed digitally. The automatic gain control system employs aquadrature detector with low frequency filters. Its purpose is to maintain the amplitude of anincoming RF signal relatively constant, so that the requirements on the analog/digital convertersdriving the digital signal processor (DSP) can be relaxed. The circuit is shown in Figure 17-1.

Figure 17-1. Synoptic of the AGC Loop

The AGC loop consists of the following elements:

1. Differential Logarithmic RF Mixers

Two independent Gilbert mixers are used to ‘demodulate’ an incoming GMSK signalinto I and Q information. These mixers are described at the transistor level (using a 0.25RF CMOS process from UMC). About 130 transistors are used. The device model isBSIM3v3.2.

2. Analog Processing


ADMS RF Tutorial—AGC LoopIntroduction

Various analog processing blocks, such as low-pass-filters, adders etc. are used to buildan image of the amplitude of the incoming signal. These blocks are described inVerilog-A, as behavioral models.

3. Digital Control Processing

This block is described in VHDL. It actually includes an A/D converter together withthe gain computation. This operation is performed digitally.

Differential Logarithmic MixersThe RF mixers are classical Gilbert mixers, built in CMOS. They use current outputs. Theconversion gain is programmable with a four-bit control word.

The output current mirrors are controlled by the four-bit word, and their gain ratio can takevalues of the form (1+x)/(1-x), where:

This relationship approximates an exponential (ex), when x is ‘small’.

To validate the transistor-level design of the mixers, large signal steady–state simulations (SST)are performed with Eldo RF. Then the conversion gain (RF to IF) is plotted as a function of thecontrol word.

Practically, the mixers have a programmable conversion gain, ranging (approximately) fromG0-20dB to G0+20dB, where G0 is the nominal conversion gain. However the variation of thegain with the control word is really linear on a +/-10dB range.

Analog ProcessingOnce the incoming RF signal has been ‘demodulated’ by the differential mixers, the systemcomputes a signal which will be used to control the gain of the mixers. This signal isproportional to the modulus of the I/Q information. The I and Q components (the outputs fromthe mixers) are independently low-pass filtered, squared and summed. This is done to capturethe instantaneous input signal power without requiring to be phase coherent, since cos2θ(e) +sin2θ(e) = 1.

With a GMSK input signal, this signal should be constant (ideally, i.e. when ignoring all non-idealities in the processing chain and in the channel as well), because GMSK is a ‘constant-envelope’ modulation scheme. This would not necessarily be the case with other modulationformats.

x b0 20⋅ b1 21⋅ b2 22⋅ b3 23⋅+ + +=

ADMS RF Tutorial—AGC LoopNetlist Explanation


Digital ProcessingThe magnitude signal (shown as MAGF on the synoptic diagram Figure 17-1) is then passed tothe digital gain control system. It is first converted from A-to-D, and it is then compared to aninternal reference (the internal current gain ‘level’).

The digital control block operates in a very simple way. If the magnitude signal is too low, thenthe level is increased by one unit. If the magnitude signal is too high, then the level is decreasedby one unit. The comparisons occur at the rhythm of a digital clock, the period of which isseveral microseconds. During the clock period, the gain is not changed, it is simply heldconstant. The input level is compared to an internal reference level (corresponding to the code‘7’, i.e. 0111 in this case).

All of this processing is described in VHDL. The input signal is conveniently described as ‘real’type signal in VHDL. This makes the A-to-D conversion implicit.

The VDHL coding-style is totally independent of the simulator and of the algorithm. AnyVHDL model can be used with ADMS RF. The same holds for Verilog of course.

The output signal is a four-bit bus that drives the logarithmic mixers. Note that ADMS RFautomatically inserts the adequate signal converters (electrical-to-real for the input signal andbit-to-electrical for the outputs).

This automatic converter insertion makes it very easy to setup this kind of simulation, althoughthe algorithms are sophisticated.

Various types of connections are allowed, between VHDL, Verilog and SPICE/Verilog-Asignals, just like Questa ADMS itself allows.

Netlist ExplanationInvoking ADMS RF is the same as invoking the ‘regular’ Questa ADMS. In the file modsst.cir.The modulated steady-state algorithm is invoked in the normal way, refer to Eldo RF andQuesta ADMS for more information:

.param Tsymbol=3.7u

.sst fund1=1G nharm1=5

.modsst 0 ‘100*Tsymbol’

The modulated steady-state analysis algorithm is a mixed time-frequency algorithm. During themodulated steady–state analysis, (analog) signals are represented as truncated Fourier series,with time-varying coefficients. At each time point, the simulator solves for these coefficients.However the spacing between the time points picked by the simulator depends on the rate ofchange of the spectrum (i.e. the Baseband signal). It does not depend on the rate of change ofthe RF carrier(s), as occurs in a transient analysis.


ADMS RF Tutorial—AGC LoopSimulation with ADMS RF

Using a simplified notation, and assuming there is a single fundamental (carrier) at frequency‘fund1’, the signals are represented as:

S(t) = vr(out).h(0) + vr(out).h(1).cos(w.t) - vi(out).h(1).sin(w.t) + vr(out).h(2).cos(2.w.t) - vi(out).h(2).sin(2.w.t) + vr(out).h(3).cos(3.w.t) - vi(out).h(3).sin(3.w.t) + …

with

The display of the results is a little bit more complicated, compared to plain transientsimulation. The DC component can be displayed with the command:

.plot fmodsst vr(node).h(0)

The time-varying spectrum around any given harmonic (i) can be displayed by using:

.plot fmodsst vr(node).h(i) vi(node).h(i)

The line above will display the I(t) and Q(t) information of a digitally modulated signal. Formore information on the .plot command please refer to “Steady-State Analysis Results” onpage 76.

In modulated steady-state analysis, the coefficients of the Fourier expansion are all complexquantities. This is why they are accessed through the vr(), vi(), vm(), vp() or vdb()functions.

Apart from the .modsst command and the .plot commands, the rest of the netlist is a regularQuesta ADMS (Eldo/SPICE) netlist.

In this tutorial, the top-level is a SPICE netlist, with SPICE, Verilog-A, and VHDL instances.

Simulation with ADMS RFCreate a working directory for this example, and navigate to it by entering the following UNIXshell commands:

mkdir <directory_name>cd <directory_name>

Using the command below, copy all of the files from the AGC example directory in to yourworking directory.

cp $MGC_AMS_HOME/examples/rfic/admsrf/AGC/* .

Before you can compile a design you must create a working library using the command vlib,from the UNIX command line type:

vlib admsrf

w 2π fund1×=

ADMS RF Tutorial—AGC LoopSimulation


This command creates a design library called admsrf.

You will then need to map this library as the work library, using the vmap command:

vmap work admsrf

This modifies a file called modelsim.ini which stores information such as the locations oflibraries, by associating a physical library name to a logical library name.

You also need to map a logical library name, admsrf, to the physical library, admsrf. This isbecause the .cir file, modsst.cir (see later), contains a reference to that logical library.

vmap admsrf admsrf

The next step is to compile all of the files; the vacom command is used to compile the VHDLfiles, the valog command is used to compile the Verilog-A files.

vacom gaincontrol.vhdvalog -oldvams agc1.va

SimulationThere are two methods of simulation:

1. Interactive Mode

2. Batch Mode

Interactive Mode• From the UNIX command line type:

vasim &

The Questa ADMS Main window and Load Design dialog are displayed.

• In this example the top design is Eldo, therefore select the Eldo radio button (locatednear the top of the dialog), see Figure 17-2.



Figure 17-2. Load Design Dialog

• Select the modsst.cir file listed in the Eldo Design pane and click Load.

Alternatively, you can specify the modsst.cir file with the -cmd option of the vasimcommand. This invokes the simulator and loads the design in GUI mode without theneed for the Load Design dialog:

vasim -cmd modsst.cir



There is no requirement to create a command file because the file modsst.cir containsthe simulation parameters.

In either case (using the Load Design dialog or not), you will see a series of progressmessages in the Transcript window, ending with “Load done”.

• From the Main window menu, select View > All windows to view all of the GUIwindows.

The Structure window shows the hierarchy of the circuit, with colors indicating thedescription level; Red for SPICE, Green for VHDL-AMS, and Orange for Verilog-AMS. See Figure 17-3.

Figure 17-3. Questa ADMS User Interface

• Select the following from the Objects window and then right-click and select Wave >Selected Net:

o rf_p, rf_n, out_p_i, out_n_i, b0, b1, b2, b3, out_p_q, out_n_q, mag, magf and att.

Note the corresponding “add wave” command in the Transcript window.

• From the Main window menu, select Simulate > Run > Run-All


ADMS RF Tutorial—AGC LoopSimulation Results

This invokes the simulation. Simulation messages are displayed in the Transcriptwindow and the progress of the simulation is shown on the status bar. The simulationtakes approximately 5 minutes.

Batch ModeBatch mode simulation allows you to run the whole simulation from the command line.

To invoke the simulation in batch mode, type the following command.

vasim -c -cmd modsst.cir -do modsst.do

• vasim specifies that you want to invoke the simulator.

• -c specifies the simulation is to be run from the command line.

• -cmd modsst.cir specifies the name of the commmand file, which contains thesimulation parameters.

• -do modsst.do informs Questa ADMS that all of the simulation commands are in thefile modsst.do.

Simulation ResultsThe results are displayed graphically in the Wave (EZwave) window, see Figure 17-4. Use thescroll bar to view all of the waves, this is located at the right hand side of the window. In thiswindow you can edit the way in which you view the results. Note that the EZwave User’s andReference Manual can be accessed from the Help menu.



Figure 17-4. Wave Window

Multiple waves can be plotted on the same graph within EZwave, to do this select the wavename on the right hand side, then drag and drop the wave onto the destination wave. For theexample, do the following:

• Drag V(magf) onto V(mag)

• Drag VI(rf_p,rf_n).H(1) onto VR(rf_p,rf_n).H(1)

• Drag VR(out_p_i,out_n_i).H(0) onto VR(out_p_q,out_n_q).H(0)

To overlap a number of waves, hold down the Ctrl key and select V(b1), V(b2), and V(b3), thenrelease the Ctrl key and drag the selected onto V(b0).

Figure 17-5 shows the groups of overlapped waves.

To validate the AGC loop, the input signal (a 1GHz GMSK signal with a random bit–streampattern) is artificially attenuated or amplified by V(att), see the fourth graph in Figure 17-5.



When the amplitude of the input signal changes, the change is tracked by the analog processingblock. The digital control block then computes a new gain to compensate the amplitude change.The compensation is never exact, as the gain can take only discrete values (the digital controlblock in VHDL outputs a four bits control word).

The second and third graphs in Figure 17-5 show the input and output I/Q information.

In the fifth graph, the digital signals (b0, b1, b2 anf b3) are shown as digital (VHDL std_logic)signals

The input amplitude is changed twice, at 110us and 260us. After each change, the outputamplitude first tends to follow the input, and after a short transient response during which thecontrol block adjusts the digital gain, it returns back to its nominal level, or close to this level,which is the desired mode of operation for the AGC loop.

Figure 17-5. Overlapped Input and Output Waveforms

The glitches visible in the output waveforms occur because the CMOS switches in the currentmirrors of the mixers are turned on or off abruptly. The bit-to-electrical converters are set up so

ADMS RF Tutorial—AGC LoopConclusion


that the transitions on the control signals (B0:B3) occur in 10ns. This causes glitches because ofthe overlap capacitances.

This example displays the accurate behavioral–modeling solution of ADMS RF; the critical RFmixers are modeled using a 0.25 RF CMOS process, so that every glitch is captured.

ConclusionThis tutorial demonstrated the use of ADMS RF for the simulation of a digital AGC loop. Thiskind of system is an example of tightly integrated RF and DSP functions. ADMS RF provides afast, efficient and elegant solution for this kind of problem.

For this example, a speedup ratio of 400 is observed compared to regular (transient) simulationwith Questa ADMS.

NoteYou can run the transient.cir file to compare the transient simulation to the MODSSTsimulation, but be aware that this simulation will take more than 30 hours!


ADMS RF Tutorial—AGC LoopConclusion


Chapter 18ADMS RF Tutorial—ZigBee Chain

IntroductionThe objective of this tutorial is to provide an example for the use of Questa ADMS RF (ADMSRF). The example shows that the tool brings important benefits in terms of performance andusability.

In this ADMS RF tutorial, we show how to model and simulate the transmission path in aZigBee transceiver. The ZigBee wireless technology and the underlying IEEE 802.15.4standard are used to achieve low-cost and low-power communication for applications withrelatively low data rates.

The blocks in this loop are described in SPICE (transistor-level), in Verilog-A, and in VHDL-AMS.

OverviewThe chain contains models for modulation, power amplification, transmission, demodulationand computation of the Binary Error Rate (BER).

Figure 18-1. The ZigBee Chain

The complete synoptic is shown in Figure 18-2:


ADMS RF Tutorial—ZigBee ChainZigBee Modulation

Figure 18-2. Synoptic of the Complete ZigBee Chain

ZigBee Modulation

Principles of the 802.15.4 PHY Level 2.4 GHz BandThe physical level (PHY) of the 802.15.4 standard specifies two bands of operation. The first is2.4 GHz and the second is 868-915 MHz. In this example, we focus on the 2.4 GHz band.

For the 2.4 GHz PHY, the standard specifies how the data coding, spreading and modulationhave to be performed. Starting from the raw baseband bit stream, bits are examined by groups offour bits. Each four-bit sequence is mapped to one symbol out of 16 possible symbols.

Each symbol is then mapped to a 32-chip sequence. These sequences are pseudo-randomsequences, and they are nearly orthogonal. The chip stream itself is OQPSK modulated withhalf-sine pulse-shaping.

In terms of data rate, the standard specifies this as 250 kbit/s. Thus the symbol rate is four timesless, that is, 62.5 ksymbol/s. And the chip rate is 32 times higher, that is, 2 Mchips/s, seeFigure 18-3.



Figure 18-3. Mapping, Spreading and Modulation

Description of ZigBee Modulation

Digital SectionIn this design, two std_logic_src.vhd VHDL models are used to generate bit sequencesclk_main and reset. clk_main is a 4µs periodic clock and reset is a start synchronization.

The pseudo_random_gen.vhd VHDL model generates random bit sequences. A random bit isgenerated every 4µs (clock period).

In the serial_parallel_1xn.vhd VHDL module, rnd_bit and clk_main are used to generate arandom integer range 0 to 15. This integer, symbol_code, is computed every 16µs andsymbol_ready is a signal which passed to “1” when a new symbol_code is ready.

With symbol_code input, the VHDL model, send_chip_seq.vhd, converts each symbol to asequence of 32 bits using the next correspondence table given in the section 6.5 of the IEEE802.15.4 specification. The period chip is 0.5µs, therefore the chip rate is 2Mchips/s.

Table 18-1. Symbol-to-Chip Mapping

DataSymbol(decimal)

Data Symbol(binary)(b0, b1, b2, b3)

Chip Values(c0, c1, ... c30, c31)

0 0000 1 1 0 1 1 0 0 1 1 1 0 0 0 0 1 1 0 1 0 1 0 0 1 0 0 0 1 0 1 1 1 0

1 1000 1 1 1 0 1 1 0 1 1 0 0 1 1 1 0 0 0 0 1 1 0 1 0 1 0 0 1 0 0 0 1 0

2 0100 0 0 1 0 1 1 1 0 1 1 0 1 1 0 0 1 1 1 0 0 0 0 1 1 0 1 0 1 0 0 1 0

3 1100 0 0 1 0 0 0 1 0 1 1 1 0 1 1 0 1 1 0 0 1 1 1 0 0 0 0 1 1 0 1 0 1

4 0010 0 1 0 1 0 0 1 0 0 0 1 0 1 1 1 0 1 1 0 1 1 0 0 1 1 1 0 0 0 0 1 1



Then, chips with even indices are written on the I_chip signal and chips with odd indices onQ_chip for OQPSK encoding.

Figure 18-4. O-QPSK Chip Offsets (Tc=period_chip=0.5µs)

Analog SectionI_chip and Q_chip signals are coded in the oqpsk_encoding.vhd VHDL model with an O_QPSKencoding with pulse shaping. Chips equal to 1 are coded with a half-positive sinusoid periodand chips equal to 0 with a half-negative sinusoid period.

5 1010 0 0 1 1 0 1 0 1 0 0 1 0 0 0 1 0 1 1 1 0 1 1 0 1 1 0 0 1 1 1 0 0

6 0110 1 1 0 0 0 0 1 1 0 1 0 1 0 0 1 0 0 0 1 0 1 1 1 0 1 1 0 1 1 0 0 1

7 1110 1 0 0 1 1 1 0 0 0 0 1 1 0 1 0 1 0 0 1 0 0 0 1 0 1 1 1 0 1 1 0 1

8 0001 1 0 0 0 1 1 0 0 1 0 0 1 0 1 1 0 0 0 0 0 0 1 1 1 0 1 1 1 1 0 1 1

9 1001 1 0 1 1 1 0 0 0 1 1 0 0 1 0 0 1 0 1 1 0 0 0 0 0 0 1 1 1 0 1 1 1

10 0101 0 1 1 1 1 0 1 1 1 0 0 0 1 1 0 0 1 0 0 1 0 1 1 0 0 0 0 0 0 1 1 1

11 1101 0 1 1 1 0 1 1 1 1 0 1 1 1 0 0 0 1 1 0 0 1 0 0 1 0 1 1 0 0 0 0 0

12 0011 0 0 0 0 0 1 1 1 0 1 1 1 1 0 1 1 1 0 0 0 1 1 0 0 1 0 0 1 0 1 1 0

13 1011 0 1 1 0 0 0 0 0 0 1 1 1 0 1 1 1 1 0 1 1 1 0 0 0 1 1 0 0 1 0 0 1

14 0111 1 0 0 1 0 1 1 0 0 0 0 0 0 1 1 1 0 1 1 1 1 0 1 1 1 0 0 0 1 1 0 0

15 1111 1 1 0 0 1 0 0 1 0 1 1 0 0 0 0 0 0 1 1 1 0 1 1 1 1 0 1 1 1 0 0 0

Table 18-1. Symbol-to-Chip Mapping

DataSymbol(decimal)

Data Symbol(binary)(b0, b1, b2, b3)

Chip Values(c0, c1, ... c30, c31)

ADMS RF Tutorial—ZigBee ChainPower Amplification


Figure 18-5. Sample Baseband Chip Sequences with Pulse Shaping

Next, I_out and Q_out signals are filtered with two low-pass filters (flt_butter_low.vhd) tosmooth the effect of the digital encoding. The filters are low-pass filters of second order with1MHz cut off frequency and have an unitary gain.

Then, I_out_F and Q_out_F are passed to an up-converting quadrature mixer (mixer.va). Thein-phase I_carrier and quadrature Q_carrier components of the 2.4 GHz carrier are generatedfrom ideal sources. The I_mod and Q_mod are combined with the Verilog-A module add.vabefore being passed to the power amplification.

Figure 18-6. IQ Modulation

Power AmplificationThe PA is a transistor-level implementation using a 0.25 µm RF process. It has a simpledifferential structure.

When used in its linear region, the voltage gain is about 7dB, and the delivered power is 0 dBm.Up to 100mV input level, the PA operates linearly, and the output phase is independent of thepower. Above this input level, compression occurs, and the phase is shifted.


ADMS RF Tutorial—ZigBee ChainTransmission

TransmissionTwo models handle the transmission simulation: a model adding phase noise model(phase_noise.vhd) and a multipath model (multipath.vhd).

Phase Noise AdditionThe RF-to-Analog converter macromodel (see “RF to Analog and Analog to RF Converters” onpage 177) allows the translation of any component from the frequency domain (such as realpart, imaginary part, magnitude, phase and complex value for a given harmonic) to anotherfrequency domain component. Two RF-to-Analog converters are used to separate magnitude(V) and phase (φ) of the PA output v(Pa_outp, Pa_outn).

The phase is modified by adding phase noise ∆ϕ

φ is replaced by ϕ + ∆ϕ

As a consequence, a new signal V_noise is computed by adding noise. The real and imaginaryparts are computed such that:

Real(V_noise) = V × cos(ϕ + ∆ϕ)

Imaginary(V_noise) = V × sin(ϕ + ∆ϕ)

Two RF-to-Analog converters and a linear voltage controlled source, E, are then used torecompose a single signal v(Pa_outp_n, Pa_outn_n).

Channel and Multipath ModelizationThe VHDL-AMS module multipath.vhd simulates multipath and echo effect propagation. Inthis model, four different attenuations are handled.

Figure 18-7. Multipath Effects

The output signal is the sum of the four delayed and attenuated signals:

V out( ) Ai ω0t φi+( )cosi

∑=

ADMS RF Tutorial—ZigBee ChainZigBee Demodulation


ZigBee DemodulationThe demodulation is done in four steps:

1. The signal is OQPSK demodulated,

2. The chips are stocked in 32 bit size buffers,

3. The signal is decoded into symbols, and then

4. The symbols are converted to serial form and written to the output Dout

OQPSK DemodulationIn the first step, the signal is down-converted. The phase is extracted with an RF-to-Analogconverter configured as follows:

input_h1=1input_format=2gain=1output_h1=0output_format=3

This is explained as follows. The phase of the first harmonic of the input signal will betranslated to the real part of the harmonic 0 of the output signal (refer to “RF to Analog andAnalog to RF Converters” on page 177 for more information). This translation is necessary inVHDL-AMS descriptions. Only baseband signals are computed in the VHDL-AMS model.

In the next step, the instantaneous phase phi_inst is sampled with a 1µs clock period, resultingin phi_inst_s. The latter is thus compared to the reference phase phi_ref. At this moment, thedelay shifts of the I and Q channels are not visible. That is why the signal is demodulated as a 4-PSK signal.

Table 18-2. Decision Regions

Chips BufferizationThe resulting chips are stocked in the buffer table (size: 32 bits) dout_vect_2_4G before beingcompared to the sequence reference. The sequences references are shown in Table 18-1.

1 π to -π/2 chip=“00”

2 -π/2 to 0 chip =“01”

3 0 to π/2 chip=“11”

4 π/2 to π chip=“10”


ADMS RF Tutorial—ZigBee ChainBER Measurement

Chip to SymbolIf the values contained in the buffer table dout_vect_2_4G equal a reference sequence, acorresponding 4 bits symbol is written to out_buf_2_4G which is a std_logic_vector.

If dout_vect is different from the references sequences, it is probably due to an error oftransmission or a synchronization problem. In this case, a function (count_diff_2_4G) is calledto determine the closest possible symbol. The result of the function is written to out_buf_2_4G.

NoteIn verbose mode, the function count_diff_2_4G displays a message with the chosensymbol.

Symbol to Serial BitIn this step, the values contained in out_buf_2_4G are converted to a serial form and written tothe output Dout.

If Least parameter is equal to 1, the MSB is written to the first position otherwise the LSB iswritten first.

BER MeasurementTo evaluate the quality of the transmission, a BER measurement is required. The modelcompares the emitted chips with the demodulated chips.

The module ber.vhd computes two measurements:

1. An instantaneous BER measurement

2. A total BER measurement

The instantaneous measurement allows the visibility of the quality of the transmission at anytime. This measurement is computed as follows:

where:

Xn = 1 if there is an error chip, otherwise Xn = 0

µ is a constant which is proportional to the time response of the instantaneous BER. If µincreases, the time response decreases.

The total BER measurement is computed as:

bern bern 1– µ X n bern 1––( )+=

ADMS RF Tutorial—ZigBee ChainDetailed Netlist Explanation


Detailed Netlist ExplanationThe netlist is a regular Questa ADMS (Eldo) netlist except for the parts concerning:

1. The presence of two RF-to-Analog converters

2. Simulation commands for sst and modsst analysis

The outputs of the module in phase_noise.vhd are two real quantities: the real part and theimaginary part of the noisy signal.

yre rf_analog_converter pin: PA_OUTP_RE PA_OUTN_RE PA_OUT_RE 0+ param:+ input_h1=0+ input_format=3+ gain=1+ output_h1=1+ output_format=3

yim rf_analog_converter pin: PA_OUTP_IM PA_OUTN_IM PA_OUT_IM 0+ param:+ input_h1=0+ input_format=3+ gain=1+ output_h1=1+ output_format=4Enoise PA_OUTP_N PA_OUTN_N value=v(PA_OUT_RE)+v(PA_OUT_IM)

This part of the design above allows a new modulation of the noisy signal.

The two RF-to-Analog converters realize the up-converting around harmonic 1 (carrierfrequencies) and the linear voltage controlled source, E, is used to recompose a complex signal.

v(pa_outp_n, pa_outn_n) is a complex noisy signal around harmonic 1.

Invoking ADMS RF is as simple as invoking the regular Questa ADMS simulator. In the netlist,zigbee_tutorial.cir, the modulated steady-state algorithm is invoked as usual:

.SST FUND1=2.4G NHARM1=4

.MODSST 0 300u

The modulated steady-state analysis algorithm is a mixed time-frequency algorithm. During themodulated steady-state analysis, (analog) signals are represented as truncated Fourier series,with time-varying coefficients. At each time point, the simulator solves the Kirchoff laws forthese coefficients. However, the spacing between the time points picked by the simulatordepends on the rate of change of the spectrum. It does not depend on the rate of change of theRF carrier(s), as occurs in a transient analysis.

ber_total(%)number_of_false_chipstotal_number_of_chips-------------------------------------------------------- 100×=


ADMS RF Tutorial—ZigBee ChainDetailed Netlist Explanation

The plots lines are:

.plot tmodsst v(I_OUT_F) v(Q_OUT_F) v(I_MOD) v(Q_MOD)

.plot fmodsst v(PA_INP,PA_INN).h(1) v(PA_OUTP,PA_OUTN).h(1)+ v(PA_OUTP_N,PA_OUTN_N).h(1) v(IN_MOD_P,IN_MOD_N).h(1)

.plot tmodsst is a recomposition of the transient waveform, equivalent as a .plot tran.

.plot fmodsst displays complex quantities. If no harmonic number is given, all harmonicswill be displayed. In this tutorial, only useful harmonics for the demonstration are plotted.

In modulated steady-state analysis, the coefficients of the Fourier expansion are all complexquantities. This is why they are accessed through the vr(), vi(), vm(), vp() or vdb() functionswith .plot fmodsst commands. In this example, the .plot fmodsst commands display thecomplex form of the signal. Complex quantities are visible through EZwave.

Apart from the .modsst command and the .plot commands, the rest of the netlist is a regularQuesta ADMS (Eldo/SPICE) netlist.

In this tutorial, the top-level is a SPICE netlist, with SPICE, Verilog-A, and VHDL-AMSinstances.

ADMS RF Tutorial—ZigBee ChainRunning the Simulation


Example Files

SPICE Netlist

ZigBee Modulation Files

Transmission Files

ZigBee Demodulation File

Computation of BER

Running the SimulationCreate a working directory for this example, and navigate to it by entering the following UNIXshell commands:

zigbee_tutorial.cir Top SPICE

std_logic_src.vhd VHDL

pseudo_random_gen.vhd VHDL

serial_parallel_1xn.vhd VHDL

send_chip_seq.vhd VHDL

oqpsk_encoding.vhd VHDL-AMS

flt_butter_low.vhd VHDL-AMS

mixer.va Verilog-A

add.va Verilog-A

mgc_quantities.vhd VHDL-AMS

mgc_commlibpkg.vhd VHDL-AMS

mgc_DCDC_pkg.vhd VHDL-AMS

mgc_filterpkg.vhd VHDL-AMS

phase_noise.vhd VHDL-AMS

multipath.vhd VHDL-AMS

zigbee_demod.vhd VHDL-AMS

ber.vhd VHDL



mkdir <directory_name>cd <directory_name>

Use the command below to copy all of the files from the ZigBee example directory to yourworking directory.

cp $MGC_AMS_HOME/examples/rfic/admsrf/zigbee/* .

Before you can compile a design you must create a working library using a valib, command,then compile all of the files using vacom commands, then run the simulation with a vasim

command. These commands are supplied in a script file, named runme.csh. To run this scriptfrom the UNIX command line, type:

source runme.csh

The first line of the script cleans delete files from previous runs if they exist. The second line is:

valib zigbee

As with “regular” Questa ADMS this creates:

• a design library called zigbee

• a file called modelsim.ini

The next lines of the script compile all of the files; the vacom commands compile the VHDLfiles, and the valog commands compile the Verilog-A files.

Then the script runs the simulation with the command line:

vasim -cmd zigbee_tutorial.cir

A series of progress messages is displayed in the Questa ADMS Main window, ending withLoad done.

To view all of the windows, from the Questa ADMS Main window select View > All windows.This displays seven new windows, including the EZwave viewer.

The Structure window shows the hierarchy of the circuit, with colors indicating the descriptionlevel (Red indicates SPICE, Green indicates VHDL, and Orange indicates Verilog-A).

Figure 18-8 shows a typical instance of the user interface when the design has just been loaded.



Figure 18-8. Questa ADMS User Interface

Prior to the simulation, the EZwave window is empty.

Drag-and-drop the following zigbee_tutorial signals from the Objects window into the EZwavewindow:

clk_main, reset, rnd_bit, symbol_ready, symbol_code, chip, i_chip, q_chip, ber andber_total

and the following boundaries/terminals:

i_out, q_out, i_out_f and q_out_f

and the following ydemod signal:

out_int

This must be done before running the simulation. The contents of the Objects window aredetermined by the selection in the Structure window.


ADMS RF Tutorial—ZigBee ChainSimulation Results

In the Questa ADMS Main window, select Simulate > Run > Run-All to start the simulation.Simulation messages are displayed in the Questa ADMS Transcript window and the progress ofthe simulation (as a percentage) is shown on the status bar.

NoteThe simulation takes approximately 7 minutes when run on a 2.8 GHz Linux host.

Simulation ResultsTo see the complete list of signals available, click on the Toggle Waveform List Visibility iconon the toolbar of the EZwave window,

Drag the following signals from the TRAN zigbee_tutorial folder to the wave window:

clk_main, rnd_bit, reset, symbol_code and symbol_ready

The resulting plot shows the random bit sequence rnd_bit, the generated symbol symbol_code,and the signal symbol_ready, see Figure 18-9.



Figure 18-9. Simulation Results — First Set of Signals

The results are displayed graphically in the EZwave window. Use the scroll bar to view all ofthe waves. The scroll bar is located at the right-hand-side of the window, and from here you canchange the way the results are displayed. Help on using EZwave is available online by clickingHelp on the far-right of the menu bar.

To see other signals in another wave window, a simple method is to drag a signal outside of thewave window (you can do this provided the existing wave window is not maximised); a newwave window containing the signal is created. For example, drag the following signals into anew window to get the result shown in Figure 18-10. You will have to zoom in to see thewaveforms exactly as shown in the diagram:

• symbol_code, chip, i_chip, q_chip, i_out and q_out



Figure 18-10. More Simulation Results

The chip signals show the chips associated with the symbol (symbol_code). The signals i_outand q_out are the results of the OQPSK modulation

In the wave shown in Figure 18-11, the signals V(i_out_f) and V(q_out_f) , from the TMODSSTzigbee_tutorial folder, have been plotted to analyze the filter effects:



Figure 18-11. Filtering Effects

Constellation PlotsThe Constellation Diagram function of EZwave makes it easy to generate the constellationdiagrams corresponding to RF signals. A function in the Waveform Calculator creates thediagram with just one mouse click. The parameters that must be provided are the symbol periodand the offset delay.

The waveform calculator then automates the creation of a constellation diagram by performingthe signal sampling and graph creation; the sampled signal is shown in scattered display modein the IQ space.

We will use this function to generate the constellation diagrams corresponding to the signal atthe input and the output of the PA, and at the output of the adding phase noise model andmultipath model.

1. Select Tools > Waveform Calculator in the EZwave window. The WaveformCalculator window appears.



2. If not already selected, select RF from the toolbar dropdown list inside the WaveformCalculator window. The buttons in the waveform calculator change.

3. Select cd (Constellation Diagram). The Constellation Diagram dialog box, seeFigure 18-12, appears:

Figure 18-12. Constellation Diagram Dialog Box

4. Select the waveform V(:zigbee_tutorial:pa_inp,pa_inn).H(1) from the SourceWaveform drop-down list. If there are no source waveforms listed then return to theEZwave window, select at least one of the waveforms from the tree and repeat.

5. Enter 13.36u in the Delay field

6. Enter 0.5u in the Symbol Period field

7. Click Apply. A constellation plot appears in an EZwave window.

8. Select each of the following waveforms in turn and click Apply for each waveform:

V(:zigbee_tutorial:pa_outp,pa_outn).H(1)

V(:zigbee_tutorial:pa_outp_n,pa_outn_n).H(1)

V(:zigbee_tutorial:in_mod_p,in_mod_n).H(1)

Corresponding constellation plots appear in the EZwave window.

9. Combine the individual plots into a single plot by draging-and-droping the constellationplot keys into the same plot to obtain a plot similar to that shown in Figure 18-13.



Figure 18-13. Constellation Diagram

The inner constellation is the usual OQPSK constellation at the PA input. The outerconstellation corresponds to the PA output. Amplitudes at the output are larger because of thePA gain, and the pattern is due to the phase shift occurring in the PA. The rotation of theconstellation corresponds to the phase shift of the PA. Output phase noise constellation andmultipath constellation shows the disturb signal. The level of noise is voluntarily high to createtransmission error.

BER MeasurementThese simulations show detection of errors in ZigBee demodulation (see message in the QuestaADMS Transcript window) and BER measurement. Figure 18-14 shows the demodulationsresults and BER measurement. Here we can see the signals ydemod:out_int and symbol_code.The demodulation is done correctly although there are many errors of transmission. The bersignal shows where the quality of transmission is bad and ber_total shows the BERmeasurement as a percentage.

The transmission is successful with a total Binary Error Rate of 11 percent, which is very high.


ADMS RF Tutorial—ZigBee ChainConclusion

Figure 18-14. BER Measurement

ConclusionThis tutorial has shown how to use ADMS RF for high-level analysis of a complete chain forthe 802.15.4 standard (ZigBee). Digital encoding, spreading and OQPSK modulation wereperformed digitally and simulated with VHDL models. Analog filtering used VHDL-AMSmodels, and the RF section was modeled with VerilogA and SPICE transistor level models.

Transmission, digital decoding and BER measurement were also simulated with VHDL-AMSmodels. And constellation diagrams were easily generated.

Everything was performed using a single environment, ADMS RF, using only standard designlanguages.


Glossary

ACPRAdjacent Channel Power Ratio. Adjacent Channel Power characterizes the degree of spectrumregrowth for non-linear circuits (e.g. Power Amplifier).

AGCAutomatic Gain Control.

Autonomous CircuitCircuit subject to free oscillation. Oscillation frequency is one of the unknowns to be determinedby steady-state analysis, contrary to non-autonomous circuits, where all frequencies aredetermined by excitation.

CCDFCumulative Complementary Density Function.

CCKComplementary Code Keying.

CDMACode Division Multiple Access.

DFTDiscrete Fourier Transform.

DSPDigital Signal Processing.

EVMError Vector Magnitude.

FDMAFrequency Division Multiple Access.

Fundamental FrequenciesFrequencies that are incommensurate (not harmonically related), or frequencies that are veryhigh harmonics of a common frequency. For example, 100 MHz and 101 Mhz are harmonics of1 Mhz, but can be considered as fundamental frequencies.

GFSKGaussian Frequency Shift Keying.

Glossary


GMSKGaussian Minimum Shift Keying.

Harmonic Balance (HB)A frequency domain algorithm to compute the steady-state of circuits under periodic or quasi-periodic excitation.

I & QIn-phase and Quadrature-phase.

IQMODIQ-Modulator.

IMxIntermodulation compression point.

IPxIntercept point.

Large Signal Steady-State (SS)The state reached by a circuit submitted to periodic or quasi-periodic large signal excitationwhen all transients have died out.

LINCLInear amplification with Non-linear Components.

LNALow Noise Amplifier.

LOLocal Oscillator.

LTELocal Truncation Error.

MODSSModulated Steady-State.

MODSST AnalysisModulated Steady-State analysis. This analysis is dedicated to handle modulated signals. Itallows combination of Steady-State analysis with respect to the high frequency part of the signal(the carrier) and Transient analysis with respect to the modulation.

MPSKM-ary Phase Shift Keying.

MQAMM-ary Quadrature Amplitude Modulator.

Glossary


Multi-Tone SignalSignal composed of several fundamental frequencies and their harmonics. Multi-tone signals arealso called pseudo-periodic.

NLNon-Linear.

NPRNoise Power Ratio. It measures the spurious free dynamic range of a multi-channel amplifier,such as a base station.

OQPSKOffset Quaternary Phase Shift Keying.

PAPower Amplifier. An important block in a transmitter flow design.

Parametric Steady-State AnalysisSeries of Steady-State analyses where each one is performed with a different value of a circuitparameter. Simulation results can be output as a function of the swept parameter.

PFDPhase-Frequency Detector

PI4QPSKπ/4 Quaternary Phase Shift Keying.

PIBPower In Band.

PLLPhase Lock Loop.

Pseudo-Periodic SignalMulti-tone signal.

SC CircuitsSwitched Capacitor circuits.

SFDRSpurious Free Dynamic Range

Single ToneSignal composed of a fundamental frequency and its harmonics. Single tone signals are periodic.

SSTSteady-State.

Glossary


Steady-State (SS)Large Signal Steady-State.

Steady-State Analysis (SST Analysis)A new kind of analysis in circuit simulation that allows the user to find the steady-state of acircuit submitted to large signal periodic or quasi-periodic excitation.

Steady-State AC Analysis (SSTAC)SSTAC is much faster than multi-tone SST. SSTAC is particularly useful to compute the small-signal conversion gain/loss.

Steady-State Noise Analysis (SSTNOISE)This will compute the output noise spectrum, the noise contribution of any noisy device, as wellas the Noise figure.

Steady-State Transfer Function Analysis (SSTXF)SSTXF computes transfer functions from any source at any frequency in the circuit to a singleoutput at a singe frequency.

TDMATime Division Multiple Access.

VCOVoltage Controlled Oscillator.

VGAVoltage Gain Amplifier.

WPANWireless Personal Area Network.

481

A B F GDC E H I J K L M N O P Q R S T U V XW Y Z


— Symbols —.AGE command, 73.EXTRACT command, 101

FMODSST, 101FOURMODSST, 101FSST, 101FUND_OSC, 102SSTAC, 101SSTJITTER, 101SSTNOISE, 101SSTSTABIL, 101SSTXF, 101TMODSST, 101TSST, 101

.FOUR command, 115

.MODSST command, 62Analysis options, 168

.OP RF command, 74

.OPTFOUR command, 98

.OPTION command, 153

.PLOT commandCONTOUR, 99FMODSST, 92FOURMODSST, 98FSST, 76MODSST, 92SSTAC, 78SSTJITTER, 88SSTNOISE, 81SSTSTABIL, 89SSTXF, 80TMODSST, 97TSST, 76

.PRINT commandFMODSST, 92FOURMODSST, 98FSST, 76MODSST, 92SSTAC, 78SSTJITTER, 88

SSTNOISE, 81SSTSTABIL, 89SSTXF, 80TMODSST, 97TSST, 76

.RESTART command, 181

.SAVE command, 181

.SNF command, 57

.SST command, 28

.SST STABIL command, 43

.SSTAC command, 44

.SSTNLCONTRIB command, 47

.SSTNOISE command, 53

.SSTPROBE command, 119

.SSTXF command, 46

— Numerics —2-port

see Two-port8PSK

see EDGE

— A —Accuracy options, 157Adjacent Channel Power Ratio (ACPR), 103,

477Admittance parameters, 193ADMS RF

AGC Loop Tutorial, 445 to 455ZigBee Chain Tutorial, 457 to 476

AGC LoopTutorial, 445 to 455

Age Analysis, 73Amplifier Tutorials

1dB Compression Point, 301ACPR Computations, 353IM3 and IP3 Extraction, 306Multi-tone Analysis, 306NPR Computation with Modulated Steady-

State Analysis, 367

Index



NPR Computation with Steady-StateAnalysis, 362

Power Efficiency, 301Single-Tone Steady-State Analysis, 294S-Parameter Extraction, 310Two-Port Noise Extraction, 294

Analog to RF Converters, 177Analysis

see Steady-State AnalysisAnalysis of a Fractional (non-integer rank)

PLL, 42automated sweeps, 183Automatic Gain Control (AGC), 477Autonomous circuit, 477

— B —Bandwidth Power, 103Baseband sources, 128Behavioral Models, 271 to 273BFACTOR, 108BOPT, 109

— C —CCDF function, 92Circuit Partitioning (.PART MODSST), 64Circuit Partitioning (.RFBLOCK), 67Code Division Multiple Access (CDMA), 477Command Syntax

.EXTRACT command, 101FSST, 101FUND_OSC, 102SSTAC, 101SSTNOISE, 101SSTXF, 101TSST, 101

.FOUR command, 115

.MODSST command, 62

.OPTFOUR command, 98

.OPTION command, 153

.PLOT commandCONTOUR, 99FMODSST, 92FOURMODSST, 98FSST, 76MODSST, 92SSTAC, 78

SSTJITTER, 88SSTNOISE, 81SSTSTABIL, 89SSTXF, 80TMODSST, 97TSST, 76

.PRINT commandFMODSST, 92FOURMODSST, 98FSST, 76MODSST, 92SSTAC, 78SSTJITTER, 88SSTNOISE, 81SSTSTABIL, 89SSTXF, 80TSST, 76

.RESTART command, 181

.SAVE command, 181

.SNF command, 57

.SST command, 28

.SST STABIL command, 43

.SSTAC command, 44

.SSTNOISE command, 53

.SSTPROBE command, 115

.SSTXF command, 46Additional syntax, 175 to 183Analysis, 27 to 74Display, 75 to 114Options, 153 to 174Sources, 115 to 152

CommLib RF, 271Complementary Code Keying (CCK), 477Constant Gain Circles

see Two-port Constant Gain CirclesConstant Gain Circles. See Two-port Constant

Gain CirclesContours

Load Pull, 99Convergence

Troubleshooting, 185 to 191Convergence options, 160Cumulative Complementary Density Function

(CCDF), 477

483Eldo RF User’s Manual, AMS 2009.2


— D —DC operating point

calculation for RF, 31, 74local stability analysis, 89

DCCUT (Capacitor), 175DCFEED (Inductor), 175Device Description

Lossy Transmission Line, 209U Model, 236W Model, 225

Digital Signal Processing (DSP), 477Digital to RF Converters, 175Digitally Modulated Sources, 127 to 148

Baseband, 128EDGE (8PSK), 139GFSK, 131GMSK, 130HPSK, 142IQMOD, 137MFSK, 135MPSK, 134MQAM, 136OFDM, 140OQPSK, 132PI4QPSK, 133Tutorial, 346

Discrete Fourier Transform (DFT), 477Display Commands

Frequency domain, 76Time domain, 76

Divergencetypes of, 185

— E —EDGE, 139Edge Specific filter, 151Error Vector Magnitude (EVM), 477

— F —Filtering Information

Edge Specific filter, 151Gaussian filter, 150Raised Cosine filter, 150Square Root Raised Cosine filter, 151

Flexible Frequency Divider Macromodel, 179FOUR, 116

Frequency Divider TutorialSteady-State Analysis, 331

Frequency Division Multiple Access (FDMA),477

Frequency domainDisplay Commands, 76

Fundamental frequencies, 477

— G —GA, 105GAC, 111Gain Parameters

see Two-port Gain ParametersGAM, 106GAMMA_OPT_MAG, 110GASM, 106GAUM, 107Gaussian filter, 150Gaussian Frequency Shift Keying (GFSK), 477Gaussian Minimum Shift Keying (GMSK),

478GFSK, 131Gilbert Cell Tutorials

Mixer Steady-State AC Analysis, 322Mixer Steady-State and Noise Analysis,

317Glossary, 477 to 480GMSK, 130GOPT, 109GP, 106GPC, 112

— H —Harmonic Balance (HB), 478HPSK, 142

— I —I & Q, 478Impedance parameters, 193IMx, 478Intercept point (IP), 478Invoking Eldo RF, 23IQMOD, 137IQ-Modulator (IQMOD), 478

— K —KFACTOR, 107



— L —Large Signal Steady-State (SS), 478Linear amplification with non-linear

components (LINC), 478LO, 478Load Pull contours, 99Load Stability Circle (LSC), 112Local Oscillator (LO), 478Local Truncation Error (LTE), 478Lossy Transmission Line Model, 209

LDTL, 209Level 1, 209Level 2, 211Level 3, 213Level 4, 214U Model, 236W Model, 225

Low Noise Amplifier (LNA), 478

— M —M-ary Phase Shift Keying (MPSK), 478M-ary Quadrature Amplitude Modulator

(MQAM), 478MFSK, 135Microstrip Models, 241 to ??, 241 to 270

90-degree Microstrip Bend, 248, 251, 254Cylindrical Via Hole in Microstrip, 260MBEND, 245MBEND2, 248MBEND3, 251MCORN, 254Microstrip Bend, 245Microstrip Step in Width (MSTEP), 257Microstrip T Junction, 242MTEE, 242Stripline Step in Width (SSTEP), 269Stripline T Junction (STEE), 266Unmitered Stripline Bend (SBEND), 263VIA2, 260

Mixer SimulationsTutorial, 403 to 435

ModelsBehavioral, 271 to 273Lossy Transmission Line, 209

Level 1, 209

Level 2, 211Level 3, 213Level 4, 214U Model, 236W Model, 225

Microstrip, 241 to ??, 241 to 270MODSST Analysis, 478Modulated Steady-State (MODSS), 478Modulation Signal

Filtering, 150PATTERN, 148

Monte Carlo analysis, 61MPSK, 134MQAM, 136MUFACTOR, 108Multi-threaded simulation options, 168Multitone Large Signal S Parameters

ExtractionTutorial, 399

Multi-tone signal, 479

— N —NC, 111NFMIN, 109Noise Circles

see Two-port Noise CirclesNoise Parameters

see Two -port Noise ParametersNoise Power Ratio (NPR), 479Noise result presentation, 166Non-Linear (NL), 479

— O —OFDM, 140Offset Quaternary Phase Shift Keying

(OQPSK), 479Optimization capability, 22Options, 153 to 174

Accuracy, 157MODSST_EPS, 160SST_ABSTOL, 158SST_ACCURACY, 159SST_ESTIM_ACCURACY, 159SST_NDIM_FFT, 158SST_OVRSMP, 158SST_PHNOISE_SPEED, 159



SST_PLL_VCO_WITH_GLOBAL_SPECTRUM, 158

SST_SPECTRUM, 158TUNING, 160

Convergence, 160SST_AT_TIME, 165SST_CONVERGENCE_HELP, 163SST_MAX_LINITER, 160SST_NODIVERGENCE, 161SST_NOLIMIT_LINITER, 161SST_NTONE_PROCEDURE_IFUND

_FOR_RESTART, 161SST_OSC_KEEP_PHASE_SEQUEN

CE, 165SST_OSC_PHASE_SEQUENCE, 165SST_PRECONDITION, 162SST_RAMPING_FACTOR, 161SST_TRAN_NPER, 165SST_USE_NTONE_PROCEDURE,

161Frequency tolerance, 165

SST_F0_ABSTOL, 166SST_F0_RELTOL, 166

Miscellaneous, 171AUTOSTOP, 171FOUR_SOURCE_DELAY, 172NO_SST, 172SST_CIRCUIT_TYPE, 172SST_KEEP_OPTIONS_FOR_SWEEP

, 173SST_MEMESTIM, 173SST_MEMORY_COMPRESS, 173SST_UIC, 174SST_VERBOSE, 174SSTNLCONTRIB_FILE, 174SSTSENSRLC_FILE, 174

MODSST analysis, 168MODSST_CENTRAL_FUND_OSCx

x, 169MODSST_FFT_FUND_FREQ, 171MODSST_FFT_NHARM, 171MODSST_FFT_TSTART, 171MODSST_FULL_DISPLAY, 168MODSST_FULL_DISPLAY_FORCE

D, 169

MODSST_HMAX, 169MODSST_HMIN, 169MODSST_USE_AVERAGE_FUND_

OSC, 169RF_PARTITIONING_MODE, 169RF_PARTITIONING_THRESHOLD,

170Multi-threaded simulations, 168

SST_MTHREAD, 168SST_NBTHREAD, 168

Noise result presentation, 166IMPROVED_SSTNOISE_PERF, 167SSTNOISE_CONTRIB_TYPE, 167SSTNOISE_EXCLUDE_DEVICES,

167SSTNOISE_FILE, 167SSTNOISE_GLOBPART, 167SSTNOISE_INCLUDE_DEVICES,

167SSTNOISE_SORT_ABS, 166SSTNOISE_SORT_CRITER, 167SSTNOISE_SORT_NBMAX, 166SSTNOISE_SORT_REL, 166

Time domain, 156SST_FULL_DISPLAY, 157SST_NPER, 156SST_NPT, 156SST_RESTART, 157SST_T0HF, 157SST_TOT_TIME_POINTS_LIMIT,

157SST_TSTART, 156SST_TSTOP, 156

OQPSK, 132Oscillator Tutorials

Phase Noise Extraction, 343Steady-State Analysis of Autonomous

Circuit, 335Sweeping the Oscillator Frequency, 340

— P —Parameter Extraction, 78, 193 to 207Parametric Steady-State Analysis, 479PATTERN Modulation Signal, 148PFD, 479Phase Lock Loop (PLL), 479



PHI_OPT, 110Pi/4 Quaternary Phase Shift Keying

(PI4QPSK), 479PI4QPSK, 133Port Impedance and Admittance Modeling

Tutorial, 437 to 443Power Amplifier (PA), 479Power in a Bandwidth (PIB), 103Power In Band (PIB), 479Pseudo-periodic signal, 479

— Q —Quality factor

estimation for autonomous circuits, 89Questa ADMS, 23, 283

— R —Raised Cosine filter, 150Restart Capabilities, 181RF Envelope Detector, 178RF to Analog Converters, 177RF to Digital Converters, 175RF_ANALOG_CONVERTER, 177RF_ENVELOPE_DETECTOR, 178RF_FREQUENCY_DIVIDER, 179RNEQ, 109

— S —S, Y, Z Parameter Extraction, 78, 193 to 207

Source Syntax, 195Save and Restart Capabilities, 181SC (Switched Capacitor ) circuits, 479Scattering parameters, 193Signal Modulation, 148Single tone, 479S-Model

FBLOCK, 202Syntax, 201

Source Stability Circle (SSC), 112Sources, 115

Baseband, 128Digitally Modulated Sources, 127EDGE, 139GFSK, 131GMSK, 130HPSK, 142

IQMOD, 137MFSK, 135Modulation Signal, 148

Filtering, 150Edge Specific, 151Gaussian, 150Raised Cosine, 150Square Root Raised Cosine, 151

MPSK, 134MQAM, 136Multi-tone Source, 116OFDM, 140OQPSK, 132Phase Noise Source, 121PI4QPSK, 133Probe Source, 119VCO, 125

Spurious Free Dynamic Range (SFDR), 479Square Root Raised Cosine filter, 151Stability Circles. See Two-port Stability

CirclesSteady-State (SST), 479Steady-State Analysis (SST Analysis), 480Steady-State Analysis Results

AC, 78Noise, 81TF, 80

Steady-State Analysis TypesAC Analysis (SSTAC), 19, 44, 480Analysis for Autonomous Circuits, 17, 33Analysis for Non-autonomous Circuits, 17,

29Local Stability Analysis, 19, 43Modulated Steady-State, 62

Algorithm, 21Autonomous Circuits, 20Non-autonomous Circuits, 20

Modulated Steady-State AnalysisFrequency domain, 98Time domain, 97Time-Frequency domain, 92

Monte Carlo, 61Noise Analysis (SSTNOISE), 19, 53, 480

Spot Noise Figure (SNF), 57Non-linear Contribution Analysis, 47



Parametric, 18PLL Analysis (SST_PLL), 39RLC Sensitivity Analysis

(SSTSENSRLC), 50Transfer Function Analysis (SSTXF), 19,

46, 480Steady-State, Large signal (SS), 480Stripline Models

see Microstrip Modelssweeps

automated, 183

— T —TGP, 107Time Division Multiple Access (TDMA), 480Time domain

Display Commands, 76Time domain options, 156Touchstone Data Format, 205Troubleshooting

Autonomous circuits, 190Convergence, 185 to 191Multi-tone circuits, 189Oscillators

High Q, 190Very non-linear, 190

Single-tone circuits, 187Tutorials, 293 to 402

ADMS RF AGC Loop, 445 to 455ADMS RF ZigBee Chain, 457 to 476Amplifier-ACPR Computations, 353Digitally Modulated Sources, 346Eldo RF Mixer Simulations, 403 to 435Eldo RF Port Impedance and Admittance

Modeling, 437 to 443EVM and BER computations, 372Frequency Divider, 325Gilbert Cell-Mixer Steady-State AC

Analysis, 322Gilbert Cell-Mixer Steady-State and Noise

Analysis, 317IM3 and IP3 Extraction, 306Load Pull contours, 379Multitone Large Signal S Parameters

Extraction, 399

NPR Computation with Modulated Steady-State Analysis, 367

NPR Computation with Steady-StateAnalysis, 362

Oscillator-Phase Noise Extraction, 343Oscillator-Steady-State Analysis of

Autonomous Circuit, 335Oscillator-Sweeping the Oscillator

Frequency, 340Power Efficiency and 1 dB Compression

Point Extraction, 301Single-Tone Steady-State, 294S-Parameter Extraction, 310Verilog-A Usage, 357

Two-port Constant Gain CirclesGAC, 111GPC, 112

Two-port Gain ExtractGA (Available Power Gain), 105GAM (Maximum Available Power Gain),

106GASM (Maximum Available Stable Gain),

106GAUM (Maximum Unilateral Transducer

Power Gain), 107GP (Power Gain), 106TGP (Transducer Power Gain), 107

Two-port Noise CirclesNC, 111

Two-port Noise ParametersBOPT, 109GAMMA_OPT_MAG, 110GOPT, 109NFMIN, 109PHI_OPT, 110RNEQ, 109YOPT, 109

Two-port Stability CirclesLoad Stability Circle (LSC), 112Source Stability Circle (SSC), 112

Two-port Stability FactorsBFACTOR, 108KFACTOR, 107MUFACTOR, 108MUFACTOR_L, 108



MUFACTOR_S, 108

— V —Verilog-A, 22

Compiler, 275Restrictions, 281Tutorial, 357

Voltage Controlled Oscillator (VCO), 480Voltage Gain Amplifier (VGA), 480

— W —Wireless Personal Area Network (WPAN), 480

— Y —YOPT, 109

— Z —ZigBee Chain

Tutorial, 457 to 476

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13. EXPORT. Software is subject to regulation by local laws and United States government agencies, which prohibit export ordiversion of certain products, information about the products, and direct products of the products to certain countries and certainpersons. Customer agrees that it will not export Software or a direct product of Software in any manner without first obtainingall necessary approval from appropriate local and United States government agencies.

14. U.S. GOVERNMENT LICENSE RIGHTS. Software was developed entirely at private expense. All Software is commercialcomputer software within the meaning of the applicable acquisition regulations. Accordingly, pursuant to US FAR 48 CFR12.212 and DFAR 48 CFR 227.7202, use, duplication and disclosure of the Software by or for the U.S. Government or a U.S.Government subcontractor is subject solely to the terms and conditions set forth in this Agreement, except for provisions whichare contrary to applicable mandatory federal laws.

15. THIRD PARTY BENEFICIARY. Mentor Graphics Corporation, Mentor Graphics (Ireland) Limited, Microsoft Corporationand other licensors may be third party beneficiaries of this Agreement with the right to enforce the obligations set forth herein.

16. REVIEW OF LICENSE USAGE. Customer will monitor the access to and use of Software. With prior written notice andduring Customer’s normal business hours, Mentor Graphics may engage an internationally recognized accounting firm toreview Customer’s software monitoring system and records deemed relevant by the internationally recognized accounting firmto confirm Customer’s compliance with the terms of this Agreement or U.S. or other local export laws. Such review may includeFLEXlm or FLEXnet (or successor product) report log files that Customer shall capture and provide at Mentor Graphics’request. Customer shall make records available in electronic format and shall fully cooperate with data gathering to support thelicense review. Mentor Graphics shall bear the expense of any such review unless a material non-compliance is revealed. MentorGraphics shall treat as confidential information all information gained as a result of any request or review and shall only use ordisclose such information as required by law or to enforce its rights under this Agreement. The provisions of this section shallsurvive the termination of this Agreement.

17. CONTROLLING LAW, JURISDICTION AND DISPUTE RESOLUTION. The owners of the Mentor Graphics intellectualproperty rights licensed under this Agreement are located in Ireland and the United States. To promote consistency around theworld, disputes shall be resolved as follows: This Agreement shall be governed by and construed under the laws of the State ofOregon, USA, if Customer is located in North or South America, and the laws of Ireland if Customer is located outside of Northor South America. All disputes arising out of or in relation to this Agreement shall be submitted to the exclusive jurisdiction ofPortland, Oregon when the laws of Oregon apply, or Dublin, Ireland when the laws of Ireland apply. Notwithstanding theforegoing, all disputes in Asia (except for Japan) arising out of or in relation to this Agreement shall be resolved by arbitration inSingapore before a single arbitrator to be appointed by the Chairman of the Singapore International Arbitration Centre (“SIAC”)to be conducted in the English language, in accordance with the Arbitration Rules of the SIAC in effect at the time of thedispute, which rules are deemed to be incorporated by reference in this section. This section shall not restrict Mentor Graphics’right to bring an action against Customer in the jurisdiction where Customer’s place of business is located. The United NationsConvention on Contracts for the International Sale of Goods does not apply to this Agreement.

18. SEVERABILITY. If any provision of this Agreement is held by a court of competent jurisdiction to be void, invalid,unenforceable or illegal, such provision shall be severed from this Agreement and the remaining provisions will remain in fullforce and effect.

19. MISCELLANEOUS. This Agreement contains the parties’ entire understanding relating to its subject matter and supersedes allprior or contemporaneous agreements, including but not limited to any purchase order terms and conditions. Some Softwaremay contain code distributed under a third party license agreement that may provide additional rights to Customer. Please seethe applicable Software documentation for details. This Agreement may only be modified in writing by authorizedrepresentatives of the parties. All notices required or authorized under this Agreement must be in writing and shall be sent to theperson who signs this Agreement, at the address specified below. Waiver of terms or excuse of breach must be in writing andshall not constitute subsequent consent, waiver or excuse.

Rev. 090402, Part No. 239301

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