implementation design of the converter-based galvanic isolation for low voltage dc distribution

The 2014 International Power Electronics Conference

Implementation Design of the Converter-Based

Galvanic Isolation for Low Voltage DC Distribution

A. Mattsson, v. Vaisanen, P. Nuutinen, T. Kaipia, A. Lana, P. Peltoniemi, P. Silventoinen, J. Partanen. LUT Energy

Lappeenranta University of Technology Lappeenranta, Finland

Abstract-In this paper the implementation of the galvanic

isolation in the low voltage direct current (LVDC) network

is discussed. The galvanic isolation can be implemented

either with a 50 Hz transformer located at the customer AC output of the customer-end inverter (CEI) or with an

isolated DC-DC converter at the input DC network side of

the CEI. However, the former solution results in a

significant increase of the volume and mass of the system and has a high amount of no-load losses. Therefore, an

isolated DC-DC converter is studied in this paper. Based on

calorimetric measurements conducted for the passive 50 Hz

transformer, the requirements for the DC-DC converter are

set. Selection of the converter topology is carried out and the

selected topology is studied by simulations and calculations.

The results are compared with the 50 Hz transformer.

Keywords- Converter, DC-DC, DC distribution.

I. INTRODUCTION

The L VDC system is an emerging power distribution technology that has been under intense research and development globally. L VDC power distribution has been proposed for versatile use-cases from datacenters [1] to public utility grid distribution [2], [3], [4], [5]. The LVDC distribution research network [6] studied in this paper utilizes voltage levels of ±750 VDC Customers are connected to either between +750 VDC and 0 VDC or between 0 VDC and -750 VDC The customer-end 50 Hz AC voltage of 230/400 V ACRMS is supplied by the CEI. The power demand in the end-user installations, and thus the load of the CEI varies on a wide range [7]. This distinguishes the distribution system application of the converter systems from the common industry applications.

Even though the design and implementation of the CEI plays a significant role in the optimization of the L VDC system, one of the main contributors to increased efficiency and power density is the galvanic isolation between the DC network and the customer-end AC network. In this paper the galvanic isolation is addressed and a selection for the configuration of the isolation is carried out by utilizing simulated and calculated comparisons.

Two methods for the galvanic isolation are discussed. The simpler of the two methods is the utilization of a passive transformer on the AC output side of the CEI. The main drawbacks of this method are its poor power density, limited performance under unbalanced load and

978-1-4799-2705-0/14/$31.00 ©2014 IEEE 587

distorted currents and a significant increase in the volume and mass which restrict its use. The second method is the utilization of power electronics and a high frequency transformer on the DC input side of the CEI in the form of an isolated DC-DC converter. The advantage of this method is its high power density and a possibility for increased efficiency which result in a reduction of the volume and mass of the system. However, as a drawback the complexity of the system increases which may increase the cost. The DC-DC converter also introduces a possibility to regulate the input DC-voltage of the CEI.

II. GALVANIC ISOLATION

The system under study utilizes an earth isolated (IT) bipolar DC network with a positive +750 VDC pole, a negative -750 VDC pole and a middle pole which is represented as 0 VDC However, as the network is floating the middle pole can have a potential difference of 750 VDC referenced to earth and therefore 0 VDC should be thought as 0 VDC only when referenced to the DC poles. In contrast the customer-end AC network is a functionally earthed (TN) network in which the neutral conductor is connected to earth due to protection and safety requirements and therefore, the two networks have to be galvanically isolated from each other. The scheme of the L VDC distribution network is depicted in Fig. 1.

+750VDC

- -- -

OVDC

Fig. 1. Scheme of the L VDC distribution network. The rectifier substation is connected to the 20 kY distribution network by a doubletier transformer with Ddy5 connection. The DC network is an earth isolated (IT) network and the CEls have 50 Hz isolation transformers at the AC output to allow the customer-end functionally earthed (TN) network earthing which is required due to protection and safety requirements.

The galvanic isolation can be implemented utilizing two different methods. Diagrams for the different isolated configurations are depicted in Fig. 2. In the six-pack inverter bridge configurations depicted in Fig. 2a and 2b, the minimum required DC-voltage level is 565 VDC In


the single phase converter-inverter configuration depicted in Fig. 2c, a minimwn of 325 VDC is required as the converter outputs would be floating and therefore, the maximwn achievable voltage difference between two phases is 2 UDe,OU!'

750 VDC Inverter CustomerAC network

DC network f----t---t-_____ =-.--T7,\,O;- L1 L2

1--_-.=<::.='-->o....><.fL-- ��N

(.) 750VDC

DC network Isolated DC-DC converter

f----t---t-.. .--,--ILl 1-----H .. I--r4-L2 f-tI-.r++--L3

750VDC

Isolated DC network DC-DC

PEN

«) Fig. 2. Different configurations for the customer-end galvanic isolation. Three-phase 1: 1 ratio 230/400 Y ACRMS 50 Hz transformer at the AC output of a three-phase inverter (a), with an isolated DC-DC converter at the DC input of a three-phase inverter (b) and with a DCDC converter and a single-phase inverter configuration feeding single phases of a three-phase AC network (c).

The L VDC research site discussed in [6] has been implemented by utilizing the configuration in Fig. 2a. Even though the selected method is the simplest of the three, the 50 Hz isolation transformer adds a significant amount of volume and mass to the CEI resulting in a significant decrease in its power density. Also, a high percentage of the total losses of the CEI are generated by the transformer as depicted by calorimetric measurement results in Fig. 3 .

� '" '" .Q :v " 6:.

800 --- catotal --- IGBT bridge

600 --- Transformer --- LC-fitter

400

200

o L=====�====�==�� 0 3 4 5 6 7 8

OJtput power IkVVJ Fig. 3. Total losses and losses per component of the CEI measured with a calorimeter. The nominal power of the CEI is ?nom = 16 kY A and the switching frequency islsw = 16 kHz.

To achieve a higher power density and a higher power to weight ratio the configurations in Fig. 2b and Fig. 2c that utilize an isolated DC-DC converter are more feasible. Furthermore, the introduction of wide bandgap (WBG) switching devices utilizing Silicon Carbide (SiC)

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and Galliwn Nitride (GaN) has enabled the realization of high efficiency power converter systems [8], [9], [10], [11]. However, the new technology is still relatively expensive and the possibility of increased efficiency needs to be compared against the increase in the cost of the converter.

Utilization of DC-DC converters in direct customerend DC distribution for datacenters and households have previously been studied for instance in [1], [12], [13]. Implementation of the galvanic isolation in the L VDC network with an isolated DC-DC converter has also been proposed in [14]. In this paper a comparison of the characteristics of an isolated DC-DC converter and a 50 Hz isolation transformer is carried out. A selection for the converter topology is presented and the topology is analyzed by simulations and calculations.

A. Volume and mass The isolation transformer of the system in [6] is a

16 kVA, 400 VACRMS, 1:1 ratio unit with Dyn 11 vector group. It weighs 93 kg and measures 42.5 cm x 37.0 cm x 18.5 cm. The power to weight ratio and power density of the transformer are thus very poor. An isolated 19.2 kW DC-DC converter with a power density of 10 W/cm3 applying WBG devices was studied in [13]. The converter in [13] is designed for a voltage level of 384 VDC. However, the rating is achieved by connecting several 48 VDC modules in series and in parallel. Therefore, if the nwnber of series connected modules is doubled and the number of paralleled modules is halved the converter could be configured for a 750 VDC system while sustaining the power density and nominal power ratings. The power density of the converter in [13] is 18 times higher in comparison to the 16 kVA isolation transformer. Therefore, if the transformer is replaced with a converter solution it would be possible to significantly reduce the volume and mass of the CEI.

B. Efficiency As depicted in Fig. 3, the transformer contributes to a

high percentage of the total losses of the CEI. However, Fig. 3 also shows that the losses of the IGBT bridge are even higher in comparison and therefore, a hard switched topology cannot be used in the isolated converter as a reduction of the losses is required. The measured transformer losses in combination with the transformer cost set the requirements for the efficiency and cost of the isolated converter.

III. DESIGN OF THE CONVERTER

A selection for the configuration in Fig. 2c was made due to the properties of the load. Therefore, the system will consist of a minimum of three converters, one per each of the phases. The converter is connected between the DC network and the CEI and therefore, the input voltage of the converter is �n = 750 VDC. The converter was selected to be operated similarly to the isolation


transformer and therefore, an output voltage value of UOtlt = Uin = 750 VDC was selected.

A. Selection of the nominal power One of the main concerns in the converter design is the

selection of the nominal power Pnom. The range of possible values of Pnom can be narrowed down based on the properties of the load [7]. The typical load type for the studied L VDC system is a detached house utilizing electrical heating. Therefore, the peak load is in the range of Pmax = 10-16 kVA which represents the required sum of the nominal power of all the converters that are supplying the inverter. As the configuration depicted in Fig. 2c was selected the maximum nominal power of one converter becomes Pnom,max = 5300 W when P max = 16 k V A. However, due to the highly varied load a case in which each of the phases utilizes a minimwn of two parallel converter-inverter modules was included in the study. Therefore, the maximum nominal power value of one converter becomes Pnom,max = 2700 W. For illustrative purposes, a range of Pnom = 200--6000 W in which the difference between the consecutive values of Pnom is Pnom,step = 200 W while Pnom = 200-1000 W and Pnom,step = 1000 W while Pnom = 1000-6000 W was selected. The value of Pnom = 6000 W defines the case in which only a single converter per phase is operated and therefore, no converters are connected in parallel.

B. Selection of the converter topology Different types of isolated converter topologies have

previously been studied in [15]. Even though the primary effort in [15] was put on fuel cell applications utilizing a low input voltage and current-fed converter topologies, a comparison of voltage-fed topologies was also carried out. Some of the studied topologies are also suitable for high input voltage and low current applications like the L VDC system which utilizes an input voltage level of UDe = 750 VDC in combination of low load currents.

Especially the phase-shifted full-bridge with a voltage doubler secondary side rectifier (PSFBVD) [15], [16] has favorable characteristics if utilized in the L VDC system. The PSFBVD is a soft-switching topology that utilizes a full-bridge on the primary side and a voltage doubler consisting of two semiconductors and two capacitors on the secondary side. The secondary side can also be implemented with diodes but then the possibility of bidirectional power flow is lost. The primary side lagging-leg switches are operated in phase-shift relative to the leading-leg switches which in combination with the magnetizing and leakage inductance of the isolation transformer is utilized to enable zero voltage switching (ZVS) for the primary side switches. The leakage inductance in combination with the secondary side voltage doubler capacitors form an LC-resonance circuit which allows the secondary and also the primary side to switch under zero current switching (ZCS). A more thorough explanation of the working principle and schematic of the topology can be found in [15], [16]. The voltage doubler also behaves as an LC-filter and therefore

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a bulky output inductor is not required. As a result, the voltage stress on the rectifier semiconductors is clamped to the output voltage and RC snubbers that are typically required to suppress the voltage ringing can also be eliminated. The PSFBVD topology enables ZVS operation of the switches over a wide load range [16] which is required in the L VDC system. The main drawback of the PSFBVD topology is its poor output voltage regulation as a function of the duty cycle [16]. However, as the L VDC system utilizes an inverter which regulates the output voltage of the system to a value of 230/400 V ACRMS and 50 Hz, a regulated DC-voltage is not a requirement. The lack of a sophisticated control of the DC-DC converter would also considerably simplify the system. The converter could be operated at a fixed duty cycle and the output voltage of the converter and therefore, the input voltage of the inverter would vary as a function of the network voltage similar to the configuration in Fig. 2a. If a duty cycle value near Deff = 0.5 is selected the DC-voltage is still well regulated as a function of the load [16].

C. Design criteria The configuration of the converter highly depends on

the selected design criteria. The converter can for example be optimized for maximwn efficiency, power density, minimization of the production cost, or total lifetime cost. Because the converters are to be utilized in a distribution network, the most significant criteria is the total lifetime cost of the converter which can be defined as

T min f (Cinv (t) + Closs (t) + Cmterr (t) + C main (t»dt

o T

'" min L [Cinv (t) + Closs (t) + Cmterr (t) + C main (t)] t=i

(1)

in which Cinv is the investment cost, Closs the loss cost, Cnterr the interruption cost, and Cmain the maintenance cost [17]. A utilization period of tu = lOa was selected for the study as this roughly estimates the expected lifetime of the converter. For the purpose of this paper only Cov and Closs are included in the calculations as Coterr and Cmain represent the cost due to failure and replacement of the component and downtime of the system. The value of Cinv is selected to include the cost of the main power semiconductors, gate drivers, DC capacitors, transformers and heat sink. The converter is assumed to be operated at a fixed duty cycle and therefore, a sophisticated control system is not a requirement. The cost of the losses Closs is calculated as present value of an annuity by

1- (1 + pt', Closs = P,o, C e _-'--....!...L_ p (2)

in which p is the interest rate and Ce the market price of electricity [17]. A value of p = 5% and Ce = 5.0 cent/kWh was used in the calculations.


The losses are calculated by using a measured average load profile for detached households [7]. The annual energy consumption was selected to be E = 20 MWh as this represents an average value for a detached household with direct electrical heating. The load profile of the selected customer type is illustrated in Fig. 4.

400 � � 300 ::l

1ij > "0 200 � Q) .'" iii 100 Qi a::

o �� o 1000 2000 3000 4000 5000 6000 7000 8000 Tirre index value [h]

Fig. 4. Measured average load profile for the selected customer type which represents a percentage value of the average hourly power of the customer [7].

The parallel converters are operated in a way in which a minimum of three converters, one per phase, are in operation at all times and when the nominal power of the single converter is exceeded the required number of parallel converters are turned on to supply the increased load as defined by

NmodlLleson ;:: Fioad , NmodlLleson E[1,2,3 ... m] . (3) , Pnom '

By only operating the required number of converters Nmodules,on to supply the load, the no-load losses of the system can be minimized. The total number of the required converter modules for the customer is calculated by

�n� [ ] NmodlLles ;:: -- , NmodlLles E 1,2,3 ... m Pnom (4)

and is used to determine the investment cost of the system.

D. Transformer parameters The effect of Pnom and Isw on the transformer

parameters was studied to determine if dividing the nominal power of the converter among several smaller converters would be feasible and how much the power density of the system could be increased by increasing the value of Isw. The transformers were designed using ETD and PM cores [18]. The magnetic material of the cores is N87 and N97 ferrite [19], [20]. For the winding, copper foil and litz was selected. Cov of the transformer was calculated using retail prices for 250 pcs quantities for the cores [21] and the price of copper per ton [22] for the winding. The losses of the transformer were also included in the design process and the cost of the losses was calculated by (2) and (3) and by utilizing the load curve in Fig. 4. Because the design process includes the

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effect of the costs Cinv and Closs the resulting parameters for the transformer do not necessarily result in the maximum efficiency of the system if the increased efficiency is shadowed by an increase in investment cost.

Fig. 5 shows the effect of �om and Isw on the volume of the transformer core when the design goal is set for minimum lifetime cost and copper foil is used for the winding. The foil height is defined as 80% of the height of the winding window and the minimum thickness of the copper is 0.025 rum. The thickness of the insulation is the same in each case.

120

ME 100 .£

� 80 ,,0

60

40

Fig. 5. Core volume of a single transformer as a function of Pnom and fsw when the design goal is set for minimum lifetime cost. The number of transformers is defined by (4)

Fig. 5 shows that decreasing the value of Pnom does not have an effect on the core selection when Pnom = 200--4000 W while Isw = 140-200 kHz. Similar behavior is observed when Pnom = 200-2000 W while Isw = 40--140 kHz. The AC flux density Bac of the transformer is defined by

(5)

in which k is a waveform coefficient and Ac the core cross-sectional area [23]. As can be seen from (5) the value of Upri, which is constant, limits the selection of the core. However, from (5) it can also be seen that by increasing the value oflsw or Npri results in the reduction of Bac and the selection of a smaller core model becomes possible if the higher number of winding turns can be implemented or the design is not limited by the reduction in thermal performance of the smaller core. Because the dependency between Isw and Bac is linear but the dependency between Isw and core loss density is exponential a significant increase in Isw does not necessarily provide further reduction of the volume of the core as the core loss density to Bac to Isw ratio begins to saturate.

Fig. 6 illustrates the number of winding turns Npri as a function of Pnom and Isw when the design goal is set for minimum lifetime cost and copper foil is used for the winding. The number of secondary turns is Npr/2 in each of the cases. The foil height is defined as 80% of the


height of the winding window and the mmnnum thickness of the copper is 0.025 mm. The thickness of the insulation is the same in each case.

50

_ 40 "-.,

30

20

Fig. 6. Winding turns Npri of a single transformer as a function of Pnom and Isw when the design goal is set for minimum lifetime cost and copper foil is used for the winding.

By comparing Fig. 5 and Fig. 6 it can be seen that the design is limited by the maximum number of winding turns Npri which results in the selection of an unnecessarily large core if the value of Pnom is decreased. Because the thickness of the copper foil cannot be decreased indefinitely the thickness and cross-sectional area of the foil limits further increasing of Npri' Therefore, calculations were carried out for the same input parameters using litz wire for the winding. The litz wire calculation methodology is presented in [24]. Utilization of litz wire makes it possible to reduce the cross-sectional area of the winding significantly compared to the foil and therefore, a higher value of Npri can be implemented when the available window area is the same. However, as a result the same value of Npri can also be fitted to a smaller core and therefore, a smaller core can be selected if any other parameter is not limiting the design.

Fig. 7 illustrates the volume of the transformer core as a function of Pnom and/sw when the design goal is set for minimum lifetime cost and litz wire is used for the winding. Fig. 8 depicts the number of winding turns Npri as a function of Pnom and/sw when the design goal is set for minimum lifetime cost and litz wire is used for the winding. The number of secondary turns is Npr/2 in each of the cases. By comparing Figs 5-8 the effect of the possibility to reduce the winding cross-sectional area by using litz wire and therefore, increase the value of Npri is obvious. Npri is increased significantly when Pnom < 4 kW compared to the foil case and therefore, the value of Bac decreases. As a result the value of Ac and therefore, the core volume can be decreased.

Fig. 9 illustrates the total core volume of the transformers when the number of transformers is calculated by (4). Fig. 9 shows that the minimum volume of the transformers is achieved when Pnom = 800 W. When Pnom is reduced further the total volume begins to increase exponentially as the number of transformers begins to increase rapidly as a function of Pnom and the

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volume of the single core cannot be decreased further as the smallest ETD core is already selected in several cases.

120

100

Mi 80 . 60 6 ,,0

40

20

4000 5000

6000 200 150

Fig. 7. Core volume of a single transformer as a function of Pnom and fsw when the design goal is set for minimum lifetime cost and litz wire is used for the winding.

350

300 250

c 200 .,�

150

100

50

Fig. 8. Winding turns Npri of a single transformer as a function of Pnom andlsw when the design goal is set for minimum lifetime cost and litz wire is used for the winding.

600

500

;; .� 300 ,,0

200

200

Fig. 9. Total core volume Vcore.lol of the transformers as a function of Pnom andlsw when the design goal is set for minimum lifetime cost and litz wire is used for the winding. The number of transformers is defined by (4).

Fig. 10 illustrates the lifetime cost Ctot of the transformers as a function of Pnom and/sw when litz wire is used as the winding and the cost of the cores, winding


and losses are combined. In the loss calculation (2) and (3) are used, and (4) is used to calculate the required number of transformers for the investment cost.

300

250

� 200 � � 150

u� 100

Pnom!W]

Fig. 10. Total lifetime cost Clmns of the transformers as a function of Pnom andlsw when the design goal is set for minimum lifetime cost and litz wire is used for the winding. The number of transformers is as defined by (4).

Fig. 10 shows that the minimum value for Ctrans is reached when Pnom = 600-3000 W while Isw = 70-200 kHz. Based on the transformer cost Ctrans we can observe that the selection of the value of Pnom could be narrowed down to values of Pnom = 800-3000 W. A preference toward values of Isw = 70-200 kHz is also observed. Results for the foil are not discussed further as the litz wire resulted in reduced values of Ctrans when Pnom < 4000 W. The differences were in the favor of the foil in some cases when Pnom � 4000 W but the differences were negligible. Because the selection cannot be made based purely on the transformer cost, the semiconductor, DC capacitance, gate driver and heat sink cost was also evaluated.

E. Semiconductor selection Because the studied nominal power range is so wide,

semiconductors with two different current ratings were selected [25], [26]. The selection of the semiconductors for each combination of Pnom and Isw was carried out comparing the safe operating area of the switching device against the pulse length and peak current of the switch which is defined as

1 OJJoutT peak = n

2 (6)

in which n is the transformer turn ratio, lout the output current of the converter and OJr the resonance frequency of the converter [15]. Because the converter is supplying an inverter generating a 50 Hz sine wave, the value of lout is modulated as a function of the output current of the inverter and needs to be considered when selecting the semiconductor switches. The converter is operated near the maximum duty ratio and therefore Deff= 0.43. Due to the voltage doubler configuration the current Ipeak is doubled at the secondary side switches and therefore, becomes 2Ipeak when compared to the value of Ipeak on the

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primary side. The primary side has four switches and the secondary side two switches. The selection of the semiconductors is illustrated in Table I.

TABLE! SELECTION OF THE SEMICONDUCTOR SWITCHES

Secondary / Primary side

S

P

Transistor

[23]

[22]

[23]

[22]

Number of semiconductors per

switch

3

2

2

Pnom [kW]

>4.3

2.2-4.3

0.3-2. ! < 0.3

>4.3

0.6-4.3

<0.6

Due to ZVS/ZCS switching and the utilization of silicon carbide semiconductors the switching loss is assumed negligible and only the conduction losses are included in the loss calculation. The RMS current of a single switch is calculated by

(7)

in which Llk is the leakage inductance of the transformer and Cr the resonance capacitance [15]. For the secondary side semiconductors IRMs is calculated with n = 1 and for the primary side with n = NseJNpri' The value of IRMs is divided between parallel semiconductors as defined by Table I and the value of lout between parallel converters as defined by (3). Fig. 11 depicts the total lifetime cost of the main semiconductors as a function of Pnom andlsw.

"

2000 1800 1600

� 1400 E 1200

o'f, 1000 800 600

Pnom [WI 6000 200 fsw [kHz[

Fig. 11. Total lifetime cost Cscm of the main semiconductors as a function of Pnom and Isw. The number of semiconductors is as defined by (4) and Tab[e 1. The values of Pnom = 1000-6000 W have been calculated with a step of Pnom.slep = 200 W to emphasize the points in which the semiconductor configuration is changed.

From Fig. 11 the points in which the number of parallel semiconductors or the number of parallel converters is increased as well as the points in which the semiconductor is changed to the more expensive


alternative can clearly be observed as an increase in the value of the cost Csem• Based solely on the semiconductor cost Csem an obvious preference toward a single Pnom> 5.2 kW converter per phase is observed. If the results in Fig. 11 are compared against the transformer cost Ctrans in Fig. lO it can be seen that Csem is clearly the more dominating of the two. Further, the dominating component in the value of Csem is the investment cost of the semiconductor switches. From Fig. 11 it is also obvious that this type of configuration of the converter is not a feasible solution if Pnom < 2.7 kW.

F. Total cost of the converter In order to determine the most feasible configuration

the cost of the heat sinks, DC-capacitance and gate drivers were also evaluated. For the gate drivers a similar configuration as in [27] was selected and prices for the main components in 250 pcs quantities was used in the calculation making the estimated cost 90 € for the required six drivers. The required thermal resistance of the heat sink for a 5WC rise above 5WC ambient was calculated from the semiconductor losses at P max and a function for the relation between the price per KlW and Pnom was formulated using the price data from [28].

Essentially the total DC-capacitance is the same in each of the cases and therefore, it can be said that the total cost is very similar. In the multiple converter cases the capacitance is merely divided into smaller parts among the parallel converters. The output side capacitance of the converter is considered part of the inverter as the inverter includes a DC-capacitance also in the 50 Hz isolation transformer configuration and is therefore not included in the calculations. A similar value as in [6] was selected for the network side DCcapacitance of the converter, making the cost of the capacitor 4 €/kW if a similar 250 pcs price [29] is used as in the previous cases. Fig. 12 depicts the total cost of the converter as a function of Pnom and/sw.

5000

4000 g � 3000

6 ,,0 2000

1000

Fig. 12. Total lifetime cost Ccmw of the converters as a function of ?nom and Isw. Ccmw includes the cost of the transformers, main semiconductors, heat sink, DC-capacitance, gate drivers and losses.

By comparing Fig. 11 and Fig. 12 it can be seen that the dominating component in the total lifetime cost Cconv of the converter is the cost Csem of the main semiconductor switches. The total cost of the converter was then compared to the 50 Hz isolation transformer. The loss cost of the 50 Hz transformer was calculated by

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(2) using the data in Fig. 3 and Fig. 4. Retail price of 900€ of the same transformer was used as the investment cost.

Fig. 13 illustrates the difference in lifetime cost between the passive 50 Hz transformer and the converter system as a function of Pnom and/sw of the converters. The nominal power of the 50 Hz transformer is 16 kW in each case.

Pnom[W]

Fig. 13. Difference in total lifetime cost CdifT between the converters and the 16 kW 50 Hz isolation transformer as a function of Pnom and fsw of a single converter.

Fig. 11 shows that despite the high cost of the converter, a clear advantage in losses still make it a feasible option if a single Pnom = 6000 W converter per phase is used. The main contributor to the difference in this case are the no-load losses, which are decreased significantly if the converter is used. However, in the parallel converter cases the advantage is shadowed by the high cost of the increasing number of sem iconductors and the difference changes in the favor of the 50 Hz transformer. The effect of the paralleling should also be investigated on the inverter side to determine its effect on the inverter cost and this will be discussed in further publications. The transformer gained a clear advantage of the paralleling and therefore, if the semiconductor cost could be decreased the parallel converter system could prove to be a feasible option. The parallel converters should also be studied as a partly series connected system which would enable the use of semiconductors with lower voltage ratings and decrease the cost further.

IV. CONCLUSION

Galvanic isolation in the L VDC network is discussed and a comparison between two isolation methods is carried out. The passive isolation method utilizing a 50 Hz transformer is shown to have poor power density and high losses. The alternative method utilizing an isolated converter is discussed in more depth and a selection for a suitable converter topology is carried out. Due to the properties of the load a case in which the nominal power could be divided between several parallel converters was also studied. A comparison of the converter parameters as a function of the switching frequency and nominal power was presented. In the studied application a preference toward a single converter per phase was observed due to a significant increase in the semiconductor cost if several converters are


paralleled. However, with the parallel system a further reduction of the losses is made possible. The conduction losses of the semiconductors and the transformer winding decrease as the load current is shared among parallel converters. Further, the no load losses could be decreased if the parallel converters are turned off during low loads. However, this requires the transformer designed for the lower nominal power to have lower core losses. The total lifetime cost of the two isolation methods was compared for a ten year utilization period. It was concluded that despite the relatively high cost of the converter system a significant decrease in the losses made the converter a feasible alternative over the 50 Hz transformer. However it is to be noted that the estimated investment cost of th� converter system does not reflect the actual production cost if the converters were to be manufactured and should be considered for illustrative purposes only.

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implementation design of the converter-based galvanic isolation for low voltage dc distribution

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