integrated 200–240-ghz fmcw radar transceiver module

3808 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 61, NO. 10, OCTOBER 2013

Integrated 200–240-GHz FMCWRadar Transceiver ModuleTomas Bryllert, Member, IEEE, Vladimir Drakinskiy,

Ken B. Cooper, Member, IEEE, and Jan Stake, Senior Member, IEEE

Abstract—We present a 220-GHz homodyne transceiver moduleintended for frequency modulated continuous wave radar appli-cations. The RF transceiver circuits are fabricated on 3- m-thickGaAs membranes, and consist of a Schottky diode based trans-mitter frequency doubler that simultaneously operates as asub-harmonic down-converting mixer. Two circuits are usedin a balanced configuration to improve the noise performance.The output power is 3 dBm over a 40-GHz bandwidth (BW)centered at 220 GHz, and the receiver function is characterized bya typical mixer conversion loss of 16 dB. We present radar imagesat 4-m target distance with up to 60-dB dynamic range using a30- s chirp time, and near-BW-limited range resolution. Themodule is intended for applications in high-resolution real-time3-D radar imaging, and the unit is therefore designed so that itcan be assembled into 1-D or 2-D arrays.

Index Terms—Frequencymodulated continuous wave (FMCW),imaging radar, millimeter wave, radar, Schottky diode, semicon-ductor membrane, transceiver.

I. INTRODUCTION

R ADAR technology has widespread use in today’s society.Atmospheric radars are used to predict the weather and to

monitor climate change. Aviation radars help airplanes to nav-igate and land in bad weather conditions and NASA’s spaceprobe Curiosity landed safely on the surface of Mars a fewmonths ago, with the help of an altitude radar. The implementa-tions of radars can look very different, but all systems are sub-jected to the following two fundamental limitations.1) The law of diffraction, which states that the cross-rangeresolution of an optical element, at best, can be

Manuscript received July 30, 2013; accepted August 05, 2013. Date of pub-lication September 09, 2013; date of current version October 02, 2013. Thiswork was supported by the Swedish Civil Contingencies Agency, MSB, by theU.S. Department of Homeland Security, Science and Technology Directorate,and by Security Link, a Strategic Research Area for security and crisis manage-ment, funded by the Swedish Government. A portion of this work was carriedout at the Jet Propulsion Laboratory, California Institute of Technology, undera contract with the National Aeronautics and Space Administration (NASA).T. Bryllert, V. Drakinskiy, and J. Stake are with the Terahertz and Mil-

limetre Wave Laboratory, Department of Microtechnology and Nanoscience,Chalmers University of Technology, SE-41296, Göteborg, Sweden. (e-mail:[email protected]).K. B. Cooper is with the Jet Propulsion Laboratory (JPL), California Institute

of Technology, Pasadena, CA 91109 USA.Color versions of one or more of the figures in this paper are available online

at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/TMTT.2013.2279359

where

is the cross-range resolution;

is the wavelength;

is the distance to the target;

is the diameter of the antenna/lens.

2) The relationship between the bandwidth (BW) of a signaland its shortest possible duration in time (related by Fouriertransform). This relation sets the range resolution of anyradar signal independent of the signal modulation format

where

is the range resolution;

is the speed of light;

is the signal BW.

Looking at these two equations it is clear that for a fixed distanceand for a fixed size of the antenna aperture, the resolution willdepend on: 1) the wavelength of the radiation: shorter wave-length better cross-range resolution and 2) the BW of thesignal: larger BW better range resolution. These two pointsare the fundamental reasons to develop millimeter-wave- andsubmillimeter-wave imaging radar [1].For many applications, it is desirable to acquire 3-D radar

images at a high frame rate (video rate). There are two differentapproaches, each with many variations, to accomplish this. Oneway is to use a single transceiver and employ rapid scanning, ei-ther of the beam or of the transceiver itself, to image the target.The other is to use an array of transceivers to acquire the im-ages–much in the sameway as with a digital camera. In practice,a combination of these two alternatives may be the best optionbecause the technology to build a full focal plane array of radartransceivers at submillimeter-wave frequencies is not available,while a single transceiver cannot provide the necessary framerate.The available output power from solid-state sources at fre-

quencies over 100 GHz falls off sharply with frequency [2], andmany of the devices are not suited for pulsed operation. There-fore, pulse compression techniques, like frequency modulatedcontinuous wave (FMCW) radar, are popular modes of opera-tion, especially at submillimeter-wave frequencies [3], [4]. Themodule presented in this paper is specifically designed to op-erate in pulse compression mode.The device technology that we use for the transceiver cir-

cuits is based on Schottky diodes on thin (3- m) semiconductor

0018-9480 © 2013 IEEE

BRYLLERT et al.: INTEGRATED 200–240-GHz FMCW RADAR TRANSCEIVER MODULE 3809

membranes. This technology has been demonstrated at frequen-cies up to 1 THz, both for power generation and for down-con-version [5], [6]. In addition to a high-performance device tech-nology, the membrane circuit technology enables accurate as-sembly and integration. Assembly is critically important at sub-millimeter-wave frequencies. Traditional techniques, like wirebonding or flip-chip soldering, introduce parasitics (inductanceand electrical length from the bond wire/solder pads) and it ischallenging to maintain assembly tolerances on the microm-eter level for a high and consistent performance. The substratethickness (typically 50 m) and the high dielectric constantof semiconductor substrates ( for Si, GaAs, InP) alsomakes it difficult to fabricate broadband transitions from chipto rectangular waveguide.The semiconductor membrane technology addresses the inte-

gration issues by utilizing chip-integrated transitions from cir-cuit to rectangular waveguide; the thin membrane substrate al-lows for broadband transitions by minimizing the amount ofdielectric material. DC and ground connections are fabricatedvia integrated on-chip beam leads, which means that a low-in-ductance ground connection can be placed close to ( 40 m)the semiconductor devices; this is useful for broadband circuitdesign. The importance of a solid assembly technique is evengreater when a larger number of circuits are to be integrated ina single housing, as in a transceiver array.Integrated radar- and transceiver circuits at frequencies up

to 220 GHz have been developed using other device technolo-gies. In particular, monolithic microwave integrated circuit(MMIC) solutions using SiGe or III–V semiconductors havebeen presented. SiGe offers a high integration level, and in[7] a complete 140-GHz four-channel transceiver chip, waspresented with a sensor double-sideband noise figure of 18.5 dBand a transmit power of 1 dBm. Current state-of-the-art SiGeHBT technologies have a cutoff frequency of 300 GHz,which means that designs at 220 GHz are challenging. Still,in [8], a 220-GHz receiver with a chip-level single-sideband(SSB) noise figure of 18 dB is demonstrated. In [9]–[11], thebuilding blocks for 220-GHz radar transceivers, fabricated inGaAs mHEMT and InP HEMT technologies, are presented. Achip-level noise figure of 5 dB is demonstrated, as well as highoutput power ( 20 dBm), using power-combining techniques.However, radar demonstrations using III–V technology at fre-quencies greater than 200 GHz still rely on hybrid integrationof more than one chip, as well as integration with separateantennas (e.g., lenses or waveguide antennas). The work de-scribed here introduces a general radar transceiver architecture,which, in contrast to transistor-based MMIC designs, can bereadily extended to frequencies up to 1 THz using existingGaAs Schottky diode technology.

II. TRANSCEIVER MODULE ARCHITECTURE

The transceiver is packaged in a gold-plated brass waveguideblockwith dimensions 21mm 21mm 33mm.A single hornantenna is used both to transmit and receive the radar signal, anda printed circuit board (PCB) is integrated in the module for IFand dc connections. The RF circuits, consisting of two identicalchips assembled in a balanced configuration, are placed in the

Fig. 1. (a) CAD drawing of the transceiver module showing the different parts.(b) Zoom-in of the two RF circuits assembled in the block. (c) Top view ofone of the block halves, showing the waveguide layout. 1: input 90 hybrid,2: additional 45 phase shifter, 3: chip channel, 4: hole for the IF transition, 5:output 90 hybrid.

Fig. 2. Block diagram of the radar transceiver.

-plane split of the waveguide block. A computer-aided design(CAD) drawing of the module is shown in Fig. 1.The input signal to the unit is an FMCW signal centered at110 GHz. The average input power over the frequency sweep

should be 13 dBm. This signal is frequency multiplied by afactor of 2, and is then transmitted out via the horn antenna.The radar return signal is received using the same antenna, andis down-converted to baseband using the same devices that wereused for frequency multiplication. The circuits operate both asa frequency doubler and as a sub-harmonic mixer. The blockdiagram of the module is shown in Fig. 2.The input hybrid (135 ) and the output hybrid (270 ) are im-

plemented using waveguide branch-line couplers. A branch-linecoupler has a 90 phase shift so the 135 phase shift at the inputside is obtained by using an additional single-stub phase shifter[12]. The output coupler has an extra 180 phase shift origi-nating from symmetry of the output coupling from the circuits[see Fig. 1(b)–(c)]. The isolated ports of the waveguide couplersare extended out to the sides of the block to enable measure-ments of input return loss and output isolation. The 180 trans-former at the IF side operates at baseband ( 1 200 MHz) and


Fig. 3. Common mode rejection ratio (CMRR) in decibels, as a function ofphase balance and amplitude balance of the input signal reaching the two trans-ceiver circuits.

is implemented using a center-tap transformer. The center-tap(CT in Fig. 2) is used to apply dc bias to the circuits.In principle, a single (unbalanced) chip could be used for fre-

quency multiplication and down-conversion, but this approachdoes not provide any noise immunity to the local oscillator. AnyAMmodulation, or even amplified thermal noise, will get mixedto the IF band, and this will appear as a dramatically increasednoise figure of the transceiver. To some degree, amplitude mod-ulation that is repeatable from chirp to chirp can be compen-sated for in signal processing, but random amplitude variationsand amplified thermal noise will always be present. In the bal-anced design, the desired radar return signal appears as a dif-ferential signal at the IF port, while the input noise appears incommon mode. In this way, the noise can be rejected in the IFtransformer. The magnitude of the noise rejection depends onthe phase and amplitude balance between the circuits and in theinput hybrid. Fig. 3 shows the theoretical common mode rejec-tion as a function of phase and amplitude balance. There arenoise mechanisms that will not be rejected by the balanced con-figuration. The most important mechanism is the excess noisegenerated in the Schottky diodes when they are driven by theinput signal. As we shall see in Section IV, this noise is present,but it does not seriously limit the radar performance.

III. CIRCUITS AND WAVEGUIDE COUPLERS

A. RF Circuits

The two RF chips that are used in the transceiver moduleare identical, and are based on the balanced frequency doublerdesign first presented by Erickson et al. [13]. Fig. 4 shows one-half of the transceiver module with the two circuits assembled,as well as a schematic of the circuit layout.A frequency doubler using this topology can operate either

in varacator mode, where the nonlinear junction capacitance ofthe Schottky diodes is used to generate harmonics, or in a resis-tive mode, where the nonlinear current–voltage characteristic isused. The mode of operation is determined by the bias point ofthe multiplier and in practice it is often a mix of varactor andresistive modes. In order for the circuits to operate as a sub-har-monic mixer and multiplier simultaneously, the mode of opera-tion should be partly resistive. The circuits can then be viewed asa switching mixer with the conductance waveform having twicethe frequency of the fundamental input signal; hence, it operates

Fig. 4. (a) Photograph of one-half of the transceiver module showing the twochips assembled [compare with Fig. 1(b)–(c)]. (b) Schematic drawing of a cir-cuit showing the matching sections of the circuit. 1: Beam-lead ground connec-tion. 2: Schottky diodes. 3: matching sections. 4: Transmission line (suspendedstripline). 5: Waveguide probe. 6: Mechanical support beam lead. 7: Low-passfilter. 8: Beam-lead used as transition to the IF PCB. (c) Schematic drawing ofthe diode configuration.

as a sub-harmonic mixer. The total conversion loss, which is thesum of the multiplier conversion loss and the mixer conversionloss, is optimized in the design. The dc-bias voltage is used toset the mode of operation, and can be used to tune the perfor-mance of the circuits, as well as to optimize the performance fordifferent input power levels. A bias voltage of 0.5 V was usedin the radar experiments.The GaAs semiconductor membrane technology that is used

for the circuits was pioneered by the Jet Propulsion Laboratory(JPL) [14] and is also described in [15]. The circuits used in thecurrent work were fabricated in the Nanofabrication Laboratory,Chalmers University of Technology.A single RF circuit was assembled in a test block, which

comprised the same waveguide matching structure as the radarmodule, but without the hybrids and the IF PCB. This modulewas characterized with respect to output power, and the result isshown in Fig. 5 for a span of 180–240 GHz. The single circuit isdesigned to operate over a full waveguide band (180–280 GHz),but the measurements were made over this smaller BW becauseof limitations in the available test hardware.

B. Waveguide Branch-Line Couplers

The waveguide branch-line couplers were designed withmanufacturability in mind. The number of coupling slots (four)was kept at a minimum, and the width of the slots were de-signed to be as wide as possible (width/depth 1/3), whilestill maintaining an acceptable BW, amplitude balance, andphase balance. Standard milling tools are readily available withan aspect ratio of 3:1. On the same note, the 45 phase shifterthat is used together with the input coupler to form the inputhybrid consists of a single waveguide stub only. Fig. 6 showsthe simulated results for the input hybrid. The output hybridis identical to the input hybrid, but scaled in frequency andwith the extra 45 phase shifter removed. Due to the reliablesimulation tools for passive waveguide structures, the hybrids


Fig. 5. Simulated results and measured performance of a single (unbalanced)radar chip operating as a 2 frequency doubler (transmitter). The chip is onlymeasured up to 240 GHz due to lack of available input power.

Fig. 6. Simulated results from the input 135 waveguide hybrid. The powerand phase balance is better than 0.6 dB and 1.5 , respectively.

were not manufactured as breakout parts, and hence, were notmeasured individually.

IV. RADAR MODULE MEASUREMENTS

A photograph of two manufactured modules can be seen inFig. 7. Both the IF and the dc connectors are surface mountMMCX, and are soldered directly onto the IF PCB. The IFtransitions from the RF circuits to the IF PCB consist of two300- m-diameter PCB nails that are soldered to the PCB andthen connected to the IF beam leads of the RF circuits with silverepoxy.

A. Fixed Frequency Characterization

The transmit function was tested with respect to output powerand conversion loss of the 2 multiplier, and the result is shownin Fig. 8. The output power is 3–5 dBm over the frequencyband 200–240 GHz, with a conversion loss better than 10 dB.The down-conversion function of the transceiver is measuredby feeding a test signal back into the transceiver output port.This signal is offset by 50 MHz from . The test signalis fed through a 10-dB attenuator to minimize reflections and

Fig. 7. Photograph of two radar transceivers. The right module has the lid re-moved to display the IF PCB.

Fig. 8. Performance of the transmit and receive function for the radar trans-ceiver at 20-mW input power.

the influence of the transmitted signal; the power of the inputsignal, as well as the test signal, is monitored using directionalcouplers. A typical mixer conversion loss of 16 dB is measuredacross the 200–240-GHz band (Fig. 8). The ripple in measuredcurve is probably due to the measurement setup.The receiver noise figure was characterized at the point of

lowest conversion loss (200-GHz output frequency). At this fre-quency, the absolute noise power was measured as a function ofIF frequency using an IF amplifier with a gain of dBand a noise figure of dB. The IF power was recordedwith a spectrum analyzer at 1-MHz BW (Fig. 9). The receiverSSB noise figure is calculated from

(1)

(2)

(3)


Fig. 9. IF noise spectrum at an input (output) frequency of 100 GHz (200GHz).The displayed noise spectral density is adjusted for the 50-dB IF gain and the1-MHz BW used in the measurement. A measurement with the input powerswitched off is included for reference. The increase in noise power beyond300 MHz is due to the limited BW of the IF transformer.

using• dB;• dB;• ;• dBm;• K;• MHz.The SSB noise figure is calculated to be 17 dB atGHz.

B. Chirped Frequency Noise Characterization

To illustrate the difference in noise performance between atransceiver using two circuits in a balanced setup and a unitusing a single circuit, we have measured IF noise spectra fromboth configurations. An FMCW chirped signal with the fol-lowing characteristics was used:• BW (input/output) 0.6/1.2 GHz;• center frequency (input/output) 110/220 GHz;• chirp time 30 s.The output port was terminated in a matched load. Fig. 10

shows the results from the measurements. The balanced config-uration suppress the IF noise by up to 25 dB. The reduction in IFnoise translates directly to an improved sensitivity of the radarsystem. In the balanced configuration, the IF noise level is in-creased by 5–7 dB compared to when the FMCW chirp is turnedoff; this excess noise consists of uncorrelated noise generatedfrom the diodes in the two RF circuits, as well as any noise thatis not suppressed due to the finite amplitude and phase balancebetween the circuits.

C. Radar Characterization

To demonstrate the feasibility of the mixer/multiplier trans-ceiver for security imaging applications, radar measurementshave been performed in the test setup described in [3]. The setuphas a 40-cm-diameter main reflector antenna that focuses theradar beam at a distance of 4 m. The available chirp BW waslimited by the backend hardware and was between 5.4–10.8GHz in the different experiments. The chirp was generated using

Fig. 10. Comparison of IF noise power using balanced and unbalanced trans-ceiver modules.

Fig. 11. Radar spectra at different strengths of the return signal. The excessnoise at high return power is from the phase noise that is mixed down to the IFfrequency band. The chirp time was 30 s and the chirp BW was 10.8 GHz.

a direct digital synthesizer (DDS) at 1.6–3.4 GHz. This chirpwas then up-converted to 36.4–38.2 GHz and frequency multi-plied ( 3) to 109.2–114.6 GHz. This signal was then amplifiedand fed into the transceiver unit. The input power to the trans-ceiver unit was on average 13 dBm, but with variations of dBover the BW. The chirped signal carries some unwanted, but re-peatable amplitude and phase modulation originating mainly inthe RF hardware (mixers, multipliers, and amplifiers). The influ-ence of this modulation is minimized by carrying out a calibra-tion measurement on a mirror target and applying the resultingcalibration data to the subsequent measurements; the procedureis described in [16].In Fig. 11, a mirror target placed at 4-m distance and tilted

off-normal to the incident radar beam, was used to produce radarreturn signals with different strengths. A signal-to-noise (S/N)ratio of 60 dB was measured at 10.8-GHz chirp BW and with30- s chirp time. The 60-dB S/N ratio corresponds to the max-imum dynamic range achieved with the current backend elec-tronics and chirp time. The noise floor increases by up to 20 dBat strong return signals due to the noise carried by the trans-mitted signal, which is mixed down to the IF band. In a 3-D


Fig. 12. Demonstration of radar range resolution. The target consists of a sheetof cardboard in front of a mirror. The separation distance is varied from 3 to7 cm.

imaging application, where multiple layers should be resolvedin range, this means that the power returned from a highly re-flective layer, rather than the noise figure of the receiver, can setthe noise floor of the radar data.The range resolution was verified by placing a sheet of semi-

transparent cardboard in front of a mirror target. The separationbetween the cardboard and the mirror was adjusted to 3–7 cm.The chirp BW in the experiment was 10.8 GHz, which gives atheoretical limit of the range resolution ( ) of

cm (4)

In Fig. 12, the results of the measurements are shown. At aseparation of 3 cm, the IF return from the 5-mm-thick cardboardand the mirror are distinguishable with a contrast of 3 dB. Thecardboard was slightly tilted to reduce the multiple reflectionsbetween the cardboard and the mirror which gave some ambi-guity in the separation distance. AHanning windowwas appliedto the time-domain data before the spectra were calculated.

D. Radar Imaging

3-D radar images were acquired by raster scanning of thewhole radar setup, including the reflector antenna. This imagingtechnique is used to verify the operation of the transceiver unit,and is not suitable for fast scanning. For near real-time scan-ning, fast secondary mirrors can be used, keeping the rest of theradar stationary [16].We use a mock-up mannequin as the target for the radar

imaging and scan a 60 60 cm area of the scene using40 36 pixels. A single chirp per pixel is recorded, using achirp time of 30 s and a chirp BW of 5.4 GHz. A short chirptime is important to increase the frame rate in radar imagingapplications. The minimum data collection time for the imagepresented here ( ), not taking the mechanical scanning intoaccount, is equal to the number of pixels multiplied by the chirptime s ms. This ideal data collec-tion time corresponds to a frame rate of 23 Hz, but in practice,

Fig. 13. Mannequin spectrum data. (inset) Mannequin range-gated total powerplot in log-scale. The rectangle in the inset is used for the S/N approximation.The black dot is the data point for the mannequin spectrum displayed in thefigure.

without a dense array of transceivers, only a much slower framerate is possible because of the need for mechanical scanning.In Fig. 13, we present a typical IF spectrum from a single

pixel on the torso of the mannequin. An estimate of the S/N ratiofor the pixel is calculated by taking the difference between thepower of the peak frequency bin and the power of the medianfrequency bin; the median power is shown with a dotted line inFig. 13. This S/N metric is then averaged for all the pixels on thetorso of the mannequin (black rectangle in the inset of Fig. 13).The average value is

dB (5)

A high S/N ratio is important in imaging application where itis desirable to see through lossy or reflective layers, like thickclothing or packaging material. The results here show that themixer/multiplier transceiver has good enough performance topenetrate fairly strongly attenuating layers ( dB one-wayattenuation).The inset of Fig. 13 shows a grayscale image of the total

power returned from the target. The data is range gated, keepingdata from a 0.3-m depth of field centered at the range of the man-nequin. The range-gating removes the clutter from the wall andthe objects behind the mannequin.It has been shown that a very powerful way to use submil-

limeter-wave radar data for imaging is to rely exclusively onthe range information. Additional algorithms can be used to se-lect the range data that should be presented, depending on thetype of targets that are anticipated. In this paper, we take themost basic algorithm to demonstrate that 3-D radar imaging isindeed feasible with the integrated modules that we have pre-sented. In Fig. 14, we plot the range for the strongest peak inthe IF spectrum, which clearly resolves the shape of the man-nequin. Due to technical limitations in the backend electronics,a relatively low RF BW of 5.4 GHz was used. The theoreticalrange resolution using this BW is 3 cm, which makes the sur-face of the mannequin look rough. The modules themselves can


Fig. 14. 3-D plot of range data using the mannequin as a target. The range ofthe maximum amplitude peak is used.

Fig. 15. Photograph of 4 transceiver units assembled into a 2 2 matrix. Theantenna separation is 21 mm.

handle up to 40 GHz of RF BW however, which would give atheoretical range resolution of 4 mm.For video-rate radar imaging, it is an advantage to use

multiple transceivers. With multiple transceivers the beam candwell longer on each pixel, which improves the S/N ratio in theradar signal. Multiple transceivers will also relax the demandson the high-speed scanning system; it is very challenging toscan a single beam over a target at video rate. Fig. 15 shows aphotograph of four transceiver modules assembled into a 2 2array (only two horn antennas are assembled). A small array,up to eight elements, is likely feasible using separate blocks, asshown in Fig. 15; for a larger number of transceivers, the circuitpackaging should be reduced in size with antennas integratedin the block.

V. CONCLUSION

In conclusion, we have developed an integrated FMCW radartransceiver for 200–240 GHz. 3-D radar imaging was demon-strated, and a S/N ratio of up to 60 dB was measured at 30- schirp time. Two RF circuits are used in a balanced architecturethat uses the same devices for the transmit and the receive func-tion; this means that a single antenna can be used for themodule.The circuit technology used for the modules is Chalmers mem-brane monolithic integrated circuits, where the whole RF circuitis fabricated on 3- m-thick GaAs membranes. The active de-vices are Schottky diodes that are operational all the way up to1 THz. The modules can be assembled into transceiver arrays,which is important in order to achieve video-rate radar imaging.

ACKNOWLEDGMENT

The authors would like to thank C.-M. Kihlman for the man-ufacturing of the waveguide blocks.

REFERENCES

[1] R. Appleby and H. B. Wallace, “Standoff detection of weapons andcontraband in the 100 GHz to 1 THz region,” IEEE Trans. AntennasPropag., vol. 55, no. 11, pp. 2944–2956, Nov. 2007.

[2] G. Chattopadhyay, “Technology, capabilities, and performance of lowpower terahertz sources,” IEEE Trans. Terahertz Sci. Technol., vol. 1,no. 1, pp. 33–53, Sep. 2011.

[3] K. B. Cooper, R. J. Dengler, N. Llombart, T. Bryllert, G. Chattopad-hyay, E. Schlecht, J. Gill, C. Lee, A. Skalare, I. Mehdi, and P. H. Siegel,“Penetrating 3-D imaging at 4 and 25 meter range using a submil-limeter-wave radar,” IEEE Trans. Microw. Theory Techn., vol. 56, no.12, pp. 2771–2778, Dec. 2008.

[4] D. A. Robertson, P. N. Marsh, D. R. Bolton, R. J. C. Middleton, R. I.Hunter, P. J. Speirs, D. G. Macfarlane, S. L. Cassidy, and G. M. Smith,“340 GHz 3-D radar imaging test bed with 10 Hz frame rate,” Proc.SPIE, vol. 8362 , pp. –10, 2012, Art. ID 836206.

[5] A. Maestrini, J. S. Ward, J. J. Gill, C. Lee, B. Thomas, R. H. Lin, G.Chattopadhyay, and I. Mehdi, “A frequency-multiplied Source withmore than 1 mW of power across the 840–900 GHz band,” IEEE Trans.Microw. Theory Techn., vol. 58, no. 7, pp. 1925–1932, Jul. 2010.

[6] T. W. Crowe, W. L. Bishop, D. W. Porterfield, J. L. Hesler, and R.M. Weikle, II, “Opening the terahertz window with integrated diodecircuits,” IEEE J. Solid-State Circuits, vol. 40, no. 10, pp. 2104–2110,Oct. 2005.

[7] M. Jahn, R. Feger, C. Pfeffer, T. F. Meister, and A. Stelzer, “A SiGe-based 140-GHz four-channel radar sensor with digital beamformingcapability,” in IEEEMTT-S Int. Microw. Symp. Dig., Jun. 17–22, 2012.

[8] E. Öjefors, B. Heinemann, and U. R. Pfeiffer, “Subharmonic 220- and320-GHz SiGeHBT receiver front-ends,” IEEE Trans. Microw. TheoryTechn., vol. 60, no. 5, pp. 1397–1404, May 2012.

[9] M. Abbasi, S. E. Gunnarsson, N. Wadefalk, R. Kozhuharov, J. Svedin,S. Cherednichenko, I. Angelov, I. Kallfass, A. Leuther, and H. Zi-rath, “Single-chip 220-GHz active heterodyne receiver and transmitterMMICswith on-chip integrated antenna,” IEEE Trans.Microw. TheoryTechn., vol. 59, no. 2, pp. 466–478, Feb. 2011.

[10] A. Tessmann, A. Leuther, H. Massler, M. Kuri, M. Riessle, M. Zink,R. Sommer, A. Wahlen, and H. Essen, “Metamorphic HEMT amplifiercircuits for use in a high resolution 210GHz radar,” in IEEECompoundSemicond. Integr. Circuit Symp., Oct. 14–17, 2007, pp. 1–4.

[11] V. Radisic, K. M. K. H. Leong, S. Sarkozy, X. Mei, W. Yoshida, P.-H.Liu, W. R. Deal, and R. Lai, “220-GHz solid-state power amplifiermodules,” IEEE J. Solid-State Circuits, vol. 47, no. 10, pp. 2291–2297,Oct. 2012.

[12] P. J. Sobis, J. Stake, and A. Emrich, “A 170 GHz 45 hybrid for submil-limeter wave sideband separating subharmonic mixers,” IEEEMicrow.Wireless Compon. Lett., vol. 18, no. 10, Oct. 2008.

[13] N. R. Erickson, B. J. Rizzi, and T. W. Crowe, “A high power doublerfor 174 GHz using a planar diode array,” in 4th Int. Space TerahertzTechnol. Symp., 1993, pp. 287–296.


[14] S. Martin, B. Nakamura, A. Fung, P. Smith, J. Bruston, A.Maestrini, F.Maiwald, P. Siegel, E. Schlecht, and I. Mehdi, “Fabrication of 200 GHzto 2700 GHz multiplier devices using GaAs and metal membranes,” inIEEE MTT-S Int. Microw. Symp. Dig., May 20–24, 2001, vol. 3, pp.1641–1644.

[15] H. Zhao, V. Drakinskiy, P. Sobis, J. Hanning, T. Bryllert, A. Tang, andJ. Stake, “Development of a 557 GHz GaAs monolithic membrane-diode mixer,” in Proc. 24th Int. Indium Phosphide Relat. Mater. Conf.,Aug. 2012, pp. 102–105.

[16] K. B. Cooper, R. J. Dengler, N. Llombart, B. Thomas, G. Chattopad-hyay, and P. H. Siegel, “THz imaging radar for standoff personnelscreening,” IEEE Trans. Terahertz Sci. Technol., vol. 1, no. 1, pp.169–182, Sep. 2011.

Tomas Bryllert (A’09–M’13) was born in Växjö,Sweden, in 1974. He received the M.S. degree inphysics and Ph.D. degree in semiconductor physicsfrom Lund University, Lund, Sweden, in 2000 and2005, respectively.In 2006, he joined the Microwave Electronics

Laboratory, Chalmers University of Technology,Göteborg, Sweden, where his main research interestwas device and circuit technology for terahertzfrequency multipliers. From 2007 to 2009, he waswith the Submillimeter Wave Advanced Technology

(SWAT) Group, Jet Propulsion Laboratory, California Institute of Technology,Pasadena, CA, USA, where he was involved with terahertz imaging and radarsystems. He is currently with the Terahertz and Millimetre Wave Labora-tory, Department of Microtechnology and Nanoscience (MC2), ChalmersUniversity of Technology. He is also cofounder and Chief Executive Officer(CEO) of Wasa Millimeter Wave AB, a company that develops and fabricatesmillimeter-wave products.

Vladimir Drakinskiy was born in Kurganinsk,Russia, in 1977. He received the Diploma degreein physics and informatics (with honors) fromthe Armavir State Pedagogical Institute, Armavir,Russia, in 2000.From 2000 to 2003, he was a Post-Graduate

Student and Junior Research Assistant with thePhysics Department, Moscow State PedagogicalUniveristy, Moscow, Russia. Since 2003, he hasbeen with the Department of Microtechnologyand Nanoscience (MC2), Chalmers University of

Technology, Göteborg, Sweden. From 2003 to 2005, he was responsible formixer chips fabrication for the Herschel Space Observatory. Since 2008, he

has been a Research Engineer with the Department Microtechnology andNanoscience, Chalmers University of Technology. He is currently responsiblefor the terahertz Schottky diodes process line at MC2, Chalmers Universityof Technology. His research interests include microfabrication and nanofab-rication techniques, detectors for sub-millimeter and terahertz ranges, andsuperconducting thin films.

Ken B. Cooper (M’06) received the A.B. degree inphysics (summa cum laude) from Harvard College,Harvard University, Cambridge, MA, USA, in 1997,and the Ph.D. degree in physics from the CaliforniaInstitute of Technology, Pasadena, CA, USA, in2003.Following postdoctoral research in supercon-

ducting microwave devices, he joined the JetPropulsion Laboratory, California Institute ofTechnology, Pasadena, CA, USA, as a Member ofthe Technical Staff in 2006. His current research

interests include submillimeter-wave radar, spectroscopy, and device physics.

Jan Stake (S’95–M’00–SM’06) was born in Udde-valla, Sweden, in 1971. He received the M.Sc. de-gree in electrical engineering and Ph.D. degree in mi-crowave electronics from the Chalmers University ofTechnology, Göteborg, Sweden, in 1994 and 1999,respectively.In 1997, he was a Research Assistant with the Uni-

versity of Virginia, Charlottesville, VA, USA. From1999 to 2001, he was a Research Fellowwith theMil-limetre Wave Group, Rutherford Appleton Labora-tory, Didcot, U.K. He then joined Saab Combitech

Systems AB, as a Senior RF/microwave Engineer, until 2003. From 2000 to2006, he held different academic positions with the Chalmers University ofTechnology. From 2003 to 2006, was also Head of the Nanofabrication Labora-tory, Department of Microtechnology and Nanoscience (MC2). During Summer2007, he was a Visiting Professor with the SubmillimeterWave Advanced Tech-nology (SWAT) Group, Jet Propulsion Laboratory (JPL), California Instituteof Technology, Pasadena, CA, USA. He is currently a Professor and Head ofthe Terahertz and Millimetre Wave Laboratory, MC2. He is also cofounder ofWasa Millimeter Wave AB, Göteborg, Sweden. His research involves sourcesand detectors for terahertz frequencies, high-frequency semiconductor devices,graphene electronics, and terahertz measurement techniques and applications.Prof. Stake serves as a topical editor for the IEEE TRANSACTIONS ON

TERAHERTZ SCIENCE AND TECHNOLOGY.

integrated 200–240-ghz fmcw radar transceiver module

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