partial-resonant buck–boost and flyback dc–dc converters

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 8, AUGUST 2014 4357

Partial-Resonant Buck–Boost and FlybackDC–DC Converters

Hamidreza Keyhani, Student Member, IEEE, and Hamid A. Toliyat, Fellow, IEEE

Abstract—This paper introduces innovative nonisolated and iso-lated soft-switched dc–dc topologies with the step-up/down ability.The nonisolated topology is constructed by adding a small ac ca-pacitor in parallel with the main inductor of the conventional buck–boost converter and replacing its semiconductor devices with thereverse-blocking switches. By using a novel control scheme, truezero-voltage switching is realized at both turn-on and turn-off of thepower switches irrespective of the input voltage, output voltage, orload value. The isolated form of the converter is created by substi-tuting the main inductor with an air-gapped high-frequency trans-former, similar to the flyback converter. In this case, two smaller accapacitors are placed on both sides of the transformer to realize softswitching as well as passive clamping; no extra clamping circuitis required. The basic operation of the proposed partial-resonantconverters includes four modes and is described in detail. The com-prehensive analysis of the topologies is carried out as well. Variousexperimental results in different operating conditions are providedto verify the performance of the proposed power converters.

Index Terms—Air-gapped transformer, dc–dc converter, reso-nant power conversion, soft switching.

I. INTRODUCTION

PULSE-WIDTH modulated (PWM) dc–dc power convertersare well developed and are widely used. These converters

are simple and relatively low cost. However, PWM topologieshave the disadvantages of considerable switching losses, highswitching stresses, and significant electromagnetic interferencedue to the hard-switching operation [1]. Soft-switching topolo-gies incorporating zero-voltage and zero-current switching havebeen introduced to increase the conversion efficiency, switch-ing frequency, and power density of converters [1], [2]. Severalsoft-switching dc–dc topologies have been proposed in the lit-erature; they can be mainly categorized as quasi-resonant, mul-tiresonant, resonant-transition, active-clamp, phase-controlled,and resonant-load converters [3]–[26]. Quasi-resonant convert-ers are obtained by replacing the power switch in conventionalPWM converters with a switch network containing resonant el-ements [3]–[7]. Although the output voltage in these converterscan be regulated by varying the switching frequency, the switchcould suffer from excessive voltage or current stresses. Multires-onant converters are the extension of quasi-resonant converters

Manuscript received October 4, 2013; revised December 15, 2013; acceptedJanuary 13, 2014. Date of current version March 26, 2014. Recommended forpublication by Associate Editor G. Moschopoulos.

The authors are with the Department of Electrical and Computer Engineer-ing, Texas A&M University, College Station, TX 77843–3128 USA (e-mail:[email protected]; [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2014.2301094

providing soft switching for all the semiconductor devices inthe circuit [7]–[9]. Resonant-transition dc–dc converters shapethe switching waveforms of the conventional PWM converterswithout significantly increasing the voltage and current stressesof the power devices with the expense of adding an auxiliaryresonant circuit [10], [11]. This auxiliary circuit is composedof a switch and resonant components. Active-clamp topolo-gies employ a large capacitor and auxiliary switch to make aresonant circuit with the transformer leakage inductance in iso-lated converters, which leads to zero-voltage switching (ZVS)in addition to nondissipative voltage clamping [12]–[14]. In aphase-controlled topology, a full-bridge network is loaded byan effective inductive load to achieve ZVS for the primary-side semiconductor devices [15], [16]. In this way, the switch-ing frequency is fixed and the output voltage is controlled viaphase control. In resonant-load dc–dc converters, a square wave-form is applied to a resonant tank network connected to a loadthrough a rectifier [17]–[26]. By moving the switching fre-quency closer to or further from the resonant frequency of thetank network, the load voltage can be controlled. According tothe formation of the resonant network, resonant-load convert-ers can also be subdivided as series resonant [17], [18], paral-lel resonant [19], [20], series-parallel LCC resonant [21]–[23],and LLC topologies [24]–[26]. Care should be taken in theconsideration of various soft-switching dc–dc converters; sev-eral introduced resonant converters have noticeable drawbacksbesides their soft-switching privileges. The drawbacks includepoor performance over a wide range of input voltages and loadresistances, poor efficiency at the light load due to current cir-culation in the tank elements, increased conduction losses, andhigh device voltage stress [1].

The soft-switched partial-resonant dc–dc converters in theisolated and nonisolated configurations are introduced in thispaper based on the authors’ previous work on the resonant ac-link converters [27]–[29]. The nonisolated topology is the newreformation of the conventional buck–boost converter with theadvantage of soft switching. An ac capacitor and two reverse-blocking (RB) switches are added to achieve this benefit. Com-pared to the quasi-resonant converters, the true ZVS happensin the proposed topology at both turn-on and turn-off of all thesemiconductor devices regardless of the load level and volt-age values. Galvanic isolation can be realized by employingan air-gapped high-frequency transformer similar to the flybackconverter without any snubber circuitry.

II. PROPOSED RESONANT DC–DC CONVERTERS

The basic partial-resonant buck–boost dc–dc converter isproposed in Fig. 1(a). Similar to the conventional buck–boost

0885-8993 © 2014 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications standards/publications/rights/index.html for more information.

4358 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 8, AUGUST 2014

Fig. 1. Proposed partial-resonant dc–dc converters. (a) Nonisolated buck–boost topology, and (b) isolated flyback topology.

converter, inductor L is responsible for transferring power fromthe input to the output. This inductor is charged from the inputand then discharged to the output cycle-by-cycle. Small ac ca-pacitor C is placed in parallel with this inductor. The main roleof capacitor C is to produce partial resonances with the induc-tor to realize ZVS for the power devices, as will be shown later.As the figure depicts, the converter needs two (RB switches. AnRB-switch can be realized by a conventional reverse-conductingswitch (IGBT or MOSFET) in series with a diode. However, thenewly available individual RB-switches can be employed withthe advantage of lower total on-state voltage [28].

Fig. 1(b) shows the isolated topology. In this converter, themagnetizing inductance of the high-frequency transformer isused for transferring power. In order to reach an appropriateinductance value, the transformer may need to have an air gap.Two smaller capacitors, C1 and C2 , provide partial resonance.These capacitors are placed on both sides of the transformerto also provide paths for the currents of the primary and sec-ondary leakage inductances and subsequently, avoid voltagespikes when the input and output switches are turned OFF [27].As a result, no extra snubber circuit is required.

III. PRINCIPLE OF OPERATION

The operation of the proposed soft-switched dc–dc converterin both nonisolated and isolated configurations is composedof four modes in each switching cycle. The converter’s maininductance charges through the input in mode 1 and dischargesto the output in mode 3. Modes 2 and 4 are for partial resonanceof the main inductance with its parallel capacitance to achieveZVS at both turn-on and turn-off of the power switches. Atypical switching cycle of the isolated partial-resonant topologyis shown in Fig. 2, and its corresponding operating modes areshown in Fig. 3. The state-plane diagram is also depicted in

Fig. 2. Waveforms of the proposed isolated converter. (a) Magnetizing in-ductance voltage, (b) magnetizing inductance current, (c) input current, and(d) output current.

Fig. 4. The detailed operating modes of the isolated converterare explained in detail as follows by ignoring the transformer’sleakage inductances and winding resistances. Their effects willbe shown later.

A. Mode 1 [Inductance Charges From the Input, Fig. 3(a)]

Switch S1 is turned ON to connect the input dc voltage acrossthe magnetizing inductance of the transformer, LM . As a result,LM charges in the positive direction. This mode is allowed to rununtil the average of the input current meets the input referencecurrent, I∗i . Subsequently, switch S1 is turned OFF.

Note that due to the existence of capacitors C1 and C2 , themagnetizing inductance voltage, vL (t), decreases slowly whenswitch S1 is turned OFF. As a result, the voltage of switch S1goes up slowly in its turn-off transition. Therefore, the turn-offof switch S1 occurs at almost zero voltage. Switch S2 has asimilar ZVS in its turn off. As the ZVS behavior at the turn-offof the power switches is caused by the existence of C1 and C2 ,it happens at any input voltage, output voltage, or load value.

KEYHANI AND TOLIYAT: PARTIAL-RESONANT BUCK–BOOST AND FLYBACK DC–DC CONVERTERS 4359

Fig. 3. Operating modes of the proposed isolated partial-resonant converter.(a) Mode 1, (b) mode 2, (c) mode 3, (d) mode 4.

B. Mode 2 [Partial Resonance, Fig. 3(b)]

LM starts to partially resonate with its total parallel capaci-tance, Ct = C1 + n2C2 (n is the transformer’s turns ratio), andtherefore, vL (t) starts to drop. This partial resonance is per-mitted to run until vL (t) becomes equal to the output reflectedvoltage (−Vout /n), which then allows the converter to go tomode 3 with soft transition. During this mode, the magnetizinginductance current, iL (t), reaches its positive peak value, ILP+ ,and can be expressed as follows:

ILP+ =

√I2E 1 +

C1 + n2C2

LMV 2

in (1)

Fig. 4. State-plane diagram of the proposed converter.

where IE 1 is the magnetizing inductance current at the end ofmode 1, and Vin is the input dc voltage.

C. Mode 3 [Inductance Discharges to Output, Fig. 3(c)]

The load-side switch, S2 , is turned ON to discharge the mag-netizing inductance energy to the output. This mode continuesuntil iL (t) reaches a small value of IE 3 , which will be definedlater. After that, this mode ends and switch S2 turns OFF. Inorder to leave a certain amount of current in LM at the end ofmode 3 in the digital control implementation, iL (t) is constantlymeasured. Once iL (t) reaches the desired value, switch S2 isturned OFF and the converter goes to mode 4.

The magnetizing inductance voltage is equal to the outputreflected voltage at the beginning of mode 3 when switch S2is turned ON. As a result, the turn-on transition of switch S2occurs at zero voltage. The zero-voltage turn-on also happensfor switch S1 similarly. This ZVS at the turn-on of the powerswitches does not depend on the load level and voltage values.

D. Mode 4 [Partial Resonance, Fig. 3(d)]

LM and Ct resonate together again. This considerable partialresonance is maintained until vL (t) is equal to the input voltagein the down-going direction, which then permits the converterto go to mode 1 with soft transition. Therefore, vL (t) shouldgo higher than the input voltage during this mode as shown inFig. 2. In the case that Vout /n is more than the input voltage,this process happens naturally, and IE 3 can be selected as zero(the peak value of the magnetizing inductance voltage, VLP , willbe equal to Vout /n). However, when Vout /n is smaller than theinput voltage, a certain amount of current should be left in themagnetizing inductance at the end of mode 3 to force vL (t) topeak higher than the input voltage as follows:

IE 3 =

√√√√C1 + n2C2

LM

(V 2

LP −(

Vout

n

)2)

(2)

where VLP is the predetermined peak value of the magnetizinginductance voltage and can be selected to be 10% to 15% higherthan the input voltage in practice.


In addition, iL (t) reaches its negative peak value, ILP−, dur-ing mode 4, which can be given by

ILP− = −

√C1 + n2C2

LMVLP . (3)

The proposed converters’ control scheme only requires themeasurement of the main inductance voltage and current asthey include the information of the input and output voltagesand currents. In case of the isolated topology where the mag-netizing inductance current cannot be measured directly, thetransformer’s primary and secondary currents should be mea-sured and properly subtracted (by including the turns ratio) toreach the magnetizing inductance current. Note that by knowingthe exact value of the main inductance in the isolated and noniso-lated topologies, the inductance current can be easily estimated,and only the inductance voltage measurement is required.

In the digital control implementation of the proposed convert-ers, the switches may be turned ON with a delay from the exactdesirable instants due to the discrete sampling. This delay maycreate an unwanted hard-switching operation. In order to avoidthis problem, the switches can be turned ON sooner while theyhave negative voltages. The switches do not conduct at this timebecause they are RB switches and their voltages are negative.Once they become forward biased, they start conducting. By us-ing this switching scheme, ZVS happens truly for both switches.Switch S1 can be turned on when vL (t) reaches its positive peakvalue, and switch S2 can be turned ON as soon as the switch S1is turned OFF. As Fig. 2 shows, once vL (t) reaches its positivepeak value, iL (t) passes zero in the positive direction. So, thezero crossing of iL (t) can be used as a sign to appropriately turnon switch S1 .

The proposed topologies can transfer power in the reverse di-rection in addition to the forward direction by replacing the RB-switches by bidirectional switches. The bidirectional switchescan be made by using two back-to-back reverse-conductingswitches in series or two RB-switches in antiparallel. In thereverse power flow, the magnetizing inductance is charged firstfrom the output in the negative direction. After a resonant mode,the inductance is discharged to the input by properly turning onthe converter’s input-side switches. Although the proposed con-verters are capable of bidirectional operation, more switchesare required to take advantage of this feature, which partiallyundermines any benefit it may provide.

IV. ANALYSIS OF THE CONVERTER

The starting point of the analysis of the proposed isolatedpartial-resonant topology is the specified peak voltage of themagnetizing inductance. Knowing that vL (t) peaks at VLP duringmode 4, iL (t) at the end of mode 4 can be given by

IE 4 =

√C1 + n2C2

LM(V 2

LP − V 2in). (4)

The input reference current is equal to the average of the iL (t)in mode 1 as follows:

I∗i =(IE 4 + IE 1) T1

2T(5)

where T1 is the time length of mode 1, and T is the switchingperiod. Applying the inductor principal equation to the trans-former’s magnetizing inductance in mode 1 gives

IE 1 =VinT1 + LM I4

LM. (6)

By substituting (6) into (5) and then solving for T1 , the timelength of mode 1 can be given by

T1 = −LM IE 4

Vin+

√(LM IE 4

Vin

)2

+2LM TI∗i

Vin. (7)

A similar analysis gives the duration of mode 3 as follows:

T3 = −nLM IE 3

RIout+

√(nLM IE 3

RIout

)2

+2n2LM T

R(8)

where Iout is the output dc current and can be found accordingto the input–output power balance from the following:

Iout =

√VinI∗i

R. (9)

The time lengths of modes 2 and 4 can be expressed by

T2 =√

LM (C1 + n2C2)

[sin−1

⎛⎝ Vin

ILP+

√C1 + n2C2

LM

⎞⎠

+ sin−1

⎛⎝ RLIo

nILP+

√C1 + n2C2

LM

⎞⎠

](10)

T4 =√

LM (C1 + n2C2)

×[2π − sin−1

(RLIo

nVLP

)− sin−1

(Vin

VLP

)]. (11)

Finally, the sum of the time intervals of all the modes of theconverter is equal to the switching period as follows:

T1 + T2 + T3 + T4 = T. (12)

Substituting (7), (8), (10), and (11) into (12) results in a set ofimplicit equations that should be solved simultaneously to findthe switching period and other operating parameters.

Fig. 5 depicts the graph of the output voltage of the pro-posed isolated converter versus the input reference current forthe different values of the input dc voltage. According to thefigure, the converter’s output voltage is a function of the inputreference current, and as the input reference current increases,the output dc voltage of the converter rises nonlinearly. The fig-ure clearly illustrates that the converter works in the step-downmode of operation in the low input reference currents and grad-ually switches to the step-up mode of operation by increasingthe input reference current. In order to have a fixed output volt-age, a closed-loop controller can be employed by measuring theoutput voltage of the converter and adjusting the input reference


Fig. 5. Output current versus input reference current for the various values ofthe input voltage and R = 120 Ω.

current according to the desired output voltage. As the maininductance in the proposed topologies is rather small (the in-ductance current is quasi-discontinuous), the dynamic responsecan be fast by selecting a proper bandwidth for the voltage loop.

The graph of the switching frequency, the positive peak valueof the magnetizing inductance current, and the absolute value ofthe negative peak of the magnetizing inductance current versusL′

M for the various values of Cs are shown in Fig. 6. L′M and

Cs are the magnetizing inductance and total parallel capacitanceseen from the secondary side of the transformer (L′

M = n2LM

and Cs = C2+C1/n2). According to the figure, as the totalparallel capacitance reduces, the magnitudes of the magnetizingcurrent peak values decrease; this can be verified by consider-ing (1) and (3). According to (10) and (11), the reduction of thetotal parallel capacitance also lessens the duration of the reso-nant modes where there is no transfer of power; consequently,iE 1 in (1) reduces as well. It is apparent that the switching fre-quency increases by lessening the resonant intervals due to thereduction of the total parallel capacitance. As Fig. 6 displays,the switching frequency and the peak values of the magnetizinginductance current also depend on the magnetizing inductancevalue; the reduction of the magnetizing inductance increases themagnitudes of the magnetizing current peak values as well asthe switching frequency. Note that the transformer’s turns ratiowas selected as 0.92 in this analysis, which will be discussedlater.

The graph of the switching frequency and the peak valuesof the magnetizing inductance current versus L′

M and Cs for agiven operating point (see Fig. 6) can be used to design the pro-posed converter effectively. The magnitudes of the magnetizingcurrent positive and negative peak values specify the conductionpower loss of the switches and the transformer power loss; there-fore, lowering these peak values yields smaller power losses onthe switches and transformer. In addition, it is desirable to in-crease the switching frequency as much as possible to shrinkthe size of passive components. In order for the converters torun properly according to their operating principle, the maininductance voltage and current should be measured frequently

Fig. 6. Switching frequency and the magnitudes of the magnetizing currentpeak values versus the total parallel capacitance for various values of the mag-netizing inductance, Vin = 300 V, R = 120 Ω, n = 0.92, and Pout = 750 W.

in each switching cycle (at least 40–50 times). Therefore, theswitching frequency should be much smaller than the samplingrate of the digital controller. As shown later in the experimentalresults section, the sample time was 1.1 μs by employing a de-cent TI’s microcontroller (TMS320F28335). This sample timeincludes the ADC conversion time and the required processingtime. Therefore, the magnetizing inductance and the total par-allel capacitance were selected as 225 μH and 100 nF to setthe switching frequency at about 22 kHz at the given operatingpoint. The switching frequency can be moved higher providedthat a faster digital controller with a smaller sample time is em-ployed. The proposed control scheme can also be implementedby using an FPGA or integrated analog design, which mainlyeliminates this problem.

Although the proposed partial-resonant isolated dc–dc con-verter can work with the transformer’s turns ratio of 1 in bothstep-up and step-down operations, the turns ratio can be selectedoptimally for the optimal operation. Fig. 7 shows the magne-tizing inductance rms current, IL,rms , versus the transformer’sturns ratio for different input voltages at the given operating


Fig. 7. Magnetizing inductance rms current versus the transformer’s turnsratio for various input voltage values, R = 120 Ω, L′

M = 225 μH, Cs =100 nF, VLP = 1.1Vin , and Pout = 750 W.

point. As the graph shows, the magnetizing inductance rms cur-rent reaches its lowest value when n is almost equal to 0.92 forthe input voltage of 300 V (same as the output voltage). This oc-curs because as the turns ratio decreases from unity, the mode-3magnetizing inductance voltage, −Vout /n, decreases. If Vout /nreaches the required peak value of the magnetizing inductancevoltage, there is no need to leave any current in the magnetizinginductance at the end of mode 3 as stated before and VLP =Vout /n. As a result, the mode-4 duration reaches its minimumvalue [note the second term in (11)], and the converter operatesmore efficiently. Decreasing n lower than the optimized valuewill cause an increase in VLP , which lengthens the mode-4 du-ration [note to the third term in (11)]. As Fig. 7 reveals, theoptimum turns ratio depends on the input voltage; in the case ofa different input voltage, the optimum turns ratio can be foundby multiplying 0.92 by the output–input voltage ratio.

Fig. 8 shows the switching frequency and the magnitudes ofthe magnetizing current peak values versus the output power fordifferent values of the input voltage. According to the figure,the positive peak value of the magnetizing inductance currentis completely dependent on the output power, and as the out-put power reduces, ILP+ decreases almost linearly. This occursbecause as the output power drops, the input reference cur-rent should decrease accordingly. Therefore, the time-lengthsof modes 1 and 3 decrease, which results in the reduction ofIE 1 in (1) and consequently ILP+ . In addition, the reductionin duration of the odd-numbered modes results in the increaseof the switching frequency. Note that according to (3), ILP−does not depend on the output power, but it may depend onthe input voltage (VLP should be higher than the input voltage).The switching frequency and the magnitudes of the magnetizingcurrent peak values at a very small amount of the output powercan be given by

f |Po u t≈0 =1

2π√

LM (C1 + n2C2)(13)

Fig. 8. Switching frequency and the magnitudes of the magnetizing currentpeak values versus the output power for various input voltage values, R =120 Ω, L′

M = 225 μH, Cs = 100 nF, and n = 0.92.

ILP+ |Po u t≈0 = − ILP−|Po u t≈0 =

√C1 + n2C2

LMVLP . (14)

According to Fig. 8, the change of the input dc voltage at agiven output power affects the operating parameters of the pro-posed converter moderately. Although the increment of the inputvoltage at any given power increases the switching frequency,the change of the peak values of the magnetizing inductancecurrent depends on the output power and input voltage values atthe operating point of the converter.

V. EXPERIMENTAL RESULTS

To test the performance of the proposed topology, a pro-totype of the proposed isolated partial-resonant converter wasbuilt and tested as shown in Fig. 9. TMS320F28335 microcon-troller was employed to control the prototype converter. Thesample time of the digital control was minimized to reach bet-ter control performance, which resulted in the sample time of1.1 μs. An air-gapped Ferrite transformer was designed with the


Fig. 9. Developed prototype of the proposed isolated topolgy.

Fig. 10. Experimental results with 300 V input at 750 W. Top: magnetizinginductance voltage (300 V/div), and bottom: magnetizing inductance current(20 A/div) versus time (12 μS/div).

secondary-side magnetizing inductance of 225 μH and turnsratio of 0.92. Since the transformer has a considerable air gaplength with a high-frequency ac current, the air gap fringingfield can cause huge eddy current losses in adjacent coil con-ductors [30]. To avoid this problem, windings were not placedin the immediate vicinity of the gap and multistranded wireswere employed. The total leakage inductance and winding re-sistance of the transformer were measured as 2 μH and 95 mΩ,respectively. The parallel capacitors C1 and C2 were chosen as47 nF to have Cs of about 103 nF at the primary-side of thetransformer.

In the first test, the prototype converter was run with the in-put voltage of 300 V to supply 750 W to a 120 Ω load, whichyielded the output voltage of 300 V. Fig. 10 shows the mag-netizing inductance current and the transformer primary-sidevoltage, which is almost equal to the magnetizing inductancevoltage. As the figure shows, the transformer’s leakage induc-tances cause some ringing at the beginning of modes 1 and 3,which are damped by the parallel capacitors and dissipated in thewinding resistances of the transformer. The resulting switchingfrequency and the positive and negative peak values of the mag-netizing inductance current were 22.4 kHz, 18.9 A, and −6.4 A,respectively, at this test condition. These values are very closeto the analytical results shown in Fig. 8. The small differencesare mainly due to the nonlinearity and power loss of the com-ponents. The measured efficiency was 96.2%. Figs. 11 and 12

Fig. 11. Experimental results with 300 V input at 750 W. Top: voltage ofswitch S1 (400 V/div), and bottom: current of switch S1 (20 A/div) versus time(5 μS/div).

Fig. 12. Experimental results with 300 V input at 750 W. Top: voltage ofswitch S2 (400 V/div), and bottom: current of switch S2 (20 A/div) versus time(5 μS/div).

depict the voltage and current waveforms of switches S1 and S2 .As the figure shows, the true ZVS happens at both turn-on andturn-off of the power switches as expected. The magnetizinginductance voltage and current of the prototype converter withthe output power of 375 W (50% nominal load) and 75 W (10%nominal load) are displayed in Figs. 13 and 14. As the figuresshow, reducing the output power mainly decreases the durationof the odd-numbered modes where power is transferred. Theswitching frequency for the output power of 375 and 75 Wwas measured as 25 and 30.1 kHz, respectively, which is inagreement with the analysis results shown in Fig. 8. The softtransitions between the modes clearly verify the soft-switchingoperation of the converter in these cases.

In order to verify the converter operation in step-up and step-down modes of operation, two more tests with ±100 V changein the input voltage were run. Fig. 15 shows the magnetizinginductance voltage and current with the input voltage of 200 V,and Fig. 16 depicts these parameters with the input voltageof 400 V. The output load and power at these test conditionswere 120 Ω and 750 W, same as the first test. The change ofthe input voltage can be revealed by observing the magnetiz-ing inductance voltage during mode 1, which is decreased in





Fig. 15 and increased in Fig. 16. As a result, the mode-1 dura-tion is increased with the 200 V input and decreased with the300 V input to draw the same amount of power from the input.Therefore, consistent with the analytical results, the switchingfrequency decreased to 18.7 kHz in the step-up operating caseand increased to 24.6 kHz in the step-down test condition. Note


Fig. 17. Efficiency of the prototype converter versus output power for differentvalues of the input voltage.

that the magnetizing inductance current at the end of mode 3 iszero with the input voltage of 200 and 300 V (see Figs. 10 and15) while it is nonzero with the 400 V input (see Fig. 16), ac-cording to (2). As a result, VLP and |ILP−| are higher in Fig. 16.The measured efficiency of the designed partial-resonant fly-back converter versus the output power for different values ofthe input voltage is shown in Fig. 17. The efficiency of theconventional flyback converter was measured 1.9% lower thanthe proposed converter at nominal operating condition. This de-crease was 3.5% at light-load condition. Note that the switchingfrequency of the prototype converter was limited by the digitalcontroller sampling rate. In case a faster controller is used, theswitching frequency can be selected higher, and as a result, thepartial-resonant converter can show more advantage in terms ofefficiency.

VI. CONCLUSION

A new generation of soft-switched buck–boost dc–dc convert-ers was introduced in this paper. The partial-resonant networkis composed of a small inductor in parallel with a small ac ca-pacitor. The main role of the inductor is to transfer power bycharging from the input and discharging to the output or viceversa. The parallel capacitor creates partial resonances to realizezero-voltage turn-on and turn-off for the converter’s switches.


The inductor can be simply replaced by an air-gapped high-frequency transformer to achieve galvanic isolation. Detailedanalysis of the proposed topology was given to reveal its be-havior at various operating points. According to this analysis,the converter’s main components can be properly selected to fitthe desired working conditions. Multiple experimental resultswere presented to demonstrate the effective performance of theproposed topology in different operating conditions.

REFERENCES

[1] M. K. Kazimierczuk and D. Czarkowski, Resonant Power Converters, 2nded. Hoboken, NJ, USA: Wiley, 2011.

[2] R. Erickson and D. Maksimovic, Fundamentals of Power Electronics, 2nded. New York, NY, USA: Kluwer, 2004.

[3] I. Aksoy, H. Bodur, and A. F. Bakan, “A new ZVT-ZCT-PWM DC–DCconverter,” IEEE Trans. Power Electron., vol. 25, no. 8, pp. 2093–2105,Aug. 2010.

[4] K. H. Liu and F. C. Lee, “Zero-voltage switching technique in DC/DCconverters,” IEEE Trans. Power Electron., vol. 5, no. 3, pp. 293–304, Jul.1990.

[5] J. Dudrik and N. D. Trip, “Soft-switching PS-PWM DC–DC converter forfull-load range applications,” IEEE Trans. Ind. Electron., vol. 57, no. 8,pp. 2807–2814, Aug. 2010.

[6] H. Rongjun and S. K. Mazumder, “A soft-switching scheme for an iso-lated DC/DC converter with pulsating DC output for a three-phase high-frequency-link PWM converter,” IEEE Trans. Power Electron., vol. 24,no. 10, pp. 2276–2288, Oct. 2009.

[7] F. C. Lee, “High-Frequency quasi-resonant and multi-resonant convertertechnologies,” in Proc. 14th Annu. Conf. Ind. Electron. Soc., Oct. 1988,vol. 3, pp. 509–521.

[8] W. A. Tabisz and F. C. Lee, “Zero-voltage-switching multi-resonanttechnique—A novel approach to improve performance of high frequencyquasi-resonant converters,” IEEE Trans. Power Electron., vol. 4, no. 4,pp. 450–458, Oct. 1989.

[9] R. Laouamer, M. Brunello, J. Ferrieux, O. Normand, and N. Buchheit, “Amulti-resonant converter for non-contact charging with electromagneticcoupling,” in Proc. Int. Conf. Ind. Electron., Control Instrum., Nov. 9–14,1997, vol. 2, pp. 792–797.

[10] G. Hua, C. S. Leu, and F. C. Lee, “Novel zero-voltage-transition PWMconverters,” in Proc. IEEE Power Electron. Spec. Conf., 1992, pp. 55–61.

[11] W. J. Gu and K. Harada, “A novel self-excited forward DC–DC converterwith zero-voltage-switched resonant transitions using a saturable core,”IEEE Trans. Power Electron., vol. 10, no. 2, pp. 131–141, Mar. 1995.

[12] R. Watson, F. C. Lee, and G. Hua, “Utilization of an active-clamp cir-cuit to achieve soft switching in flyback converters,” IEEE Trans. PowerElectron., vol. 11, no. 1, pp. 162–169, Jan. 1996.

[13] L. Jong-Jae, J. Kwon, K. Eung-Ho, and B. Kwon, “Dual series-resonantactive-clamp converter,” IEEE Trans. Ind. Electron., vol. 55, no. 2,pp. 699–710, Feb. 2008.

[14] M. Chen and J. Sun, “Reduced-order averaged modeling of active-clampconverters,” IEEE Trans. Power Electron., vol. 21, no. 2, pp. 487–494,Mar. 2006.

[15] J. A. Sabate, V. Vlatkovic, R. B. Ridley, and F. C. Lee, “High-voltage,high-power, ZVS, full-bridge PWM converter employing an active snub-ber,” in Proc. Appl. Power Electron. Conf. Expo., Mar. 1991, pp. 158–163.

[16] C. Zhao, S. D. Round, and J.W. Kolar, “Full-order averaging modellingof zero-voltage-switching phase-shift bidirectional DC–DC converters,”IET Power Electron., vol. 3, no. 3, pp. 400–410, May 2010.

[17] R. King and T. Stuart, “A normalized model for the half bridge seriesresonant converter,” IEEE Trans. Aerosp. Electron. Syst., vol. AES-17,no. 2, pp. 180–193, Mar. 1981.

[18] V. Vorperian and S. Cuk, “A complete DC analysis of the series resonantconverter,” in Proc. IEEE Power Electron. Spec. Conf., Jun. 1982, pp. 85–100.

[19] S. D. Johnson, A. F. Witulski, and R. W. Erickson, “A comparison ofresonant topologies in high-voltage DC applications,” IEEE Trans. Aerosp.Electron. Syst., vol. 24, no. 3, pp. 263–274, Jul. 1988.

[20] S. D. Johnson and R. W. Erickson, “Steady-state analysis and design of theparallel resonant converter,” IEEE Trans. Power Electron., vol. 3, no. 1,pp. 93–104, Jan. 1988.

[21] H. Chen, E. Sng, and K. Tseng, “Generalized optimal trajectory controlfor closed loop control of series-parallel resonant converter,” IEEE Trans.Power Electron., vol. 21, no. 5, pp. 1347–1355, Sep. 2006.

[22] M. K. Kazimierczuk, N. Thirunarayan, and S. Wang, “Analysis of series-parallel resonant converter,” IEEE Trans. Aerosp. Electron. Syst., vol. 29,no. 1, pp. 88–98, Jan. 1993.

[23] D. Fu, F. C. Lee, Y. Qiu, and F. Wang, “A novel high-power-densitythree-level LCC resonant converter with constant-power-factor-controlfor charging applications,” IEEE Trans. Power Electron., vol. 23, no. 5,pp. 2411–2420, Sep. 2008.

[24] Y. Gu, Z. Lu, L. Hang, Z. Qian, and G. Huang, “Three-level LLC seriesresonant dc/dc converter,” IEEE Trans. Power Electron., vol. 20, no. 4,pp. 781–789, Jul. 2005.

[25] K. Jin and X. Ruan, “Hybrid full-bridge three-level LLC resonantconverter—A novel dc–dc converter suitable for fuel cell power system,”IEEE Trans. Ind. Electron., vol. 53, no. 5, pp. 1492–1503, Oct. 2006.

[26] X. Fang, H. Hu, F. Chen, U. Somani, E. Auadisian, J. Shen, and I. Batarseh,“Efficiency-oriented optimal design of the LLC resonant converter basedon peak gain placement,” IEEE Trans. Power Electron., vol. 28, no. 5,pp. 2285–2296, May 2013.

[27] H. Keyhani, H. A. Toliyat, M. Todorovic, R. Lai, and R. Datta, “An iso-lated resonant AC-Link Three-Phase AC–AC converter using a single HFtransformer,” IEEE Trans. Ind. Electron., 2014.

[28] H. Keyhani and H. A. Toliyat, “A new generation of buck–boost resonantAC-link DC–DC converters,” in Proc. Appl. Power Electron. Conf. Expo.,Mar. 2013, pp. 1383–1390.

[29] H. Keyhani and H. A. Toliyat, “A ZVS single-inductor multi-input multi-output DC–DC converter with the step up/down capability,” in Proc.Energy Convers. Congr. Expo., Sep. 2013, pp. 5546–5552.

[30] W. A. Roshen, “Fringing field formulas and winding loss due to an airgap,” IEEE Trans. Magn., vol. 43, no. 8, pp. 3387–3394, Aug. 2007.

Hamidreza Keyhani (S’11) received the B.S. degreefrom the University of Shiraz, Iran, in 2001, the M.S.degree from Sharif University of Technology, Tehran,Iran, in 2004, and the Ph.D. degree from Texas A&MUniversity, College Station, TX, USA, in 2014, all inelectrical engineering.

From 2002 to 2010, he worked in industry anddeveloped several power electronic circuits and sys-tems, including switching dc power supplies in vari-ous voltage and power levels, power converters, un-interruptible power supplies, and automatic voltage

stabilizers. During 2011 and 2013, he worked toward the Ph.D. degree on soft-switched high-frequency-link power converters and published six journal andeight conference papers. His research interests and experience include mod-eling, design and control of power converters, switching power supplies, andrenewable energy systems.

Hamid A. Toliyat (S’87–M’91–SM’96–F’08) joinedthe Department of Electrical and Computer Engineer-ing, Texas A&M University, College Station, TX,USA, in 1994, where he is currently Raytheon En-dowed Professor.

Dr. Toliyat received the prestigious Cyrill VeinottAward in Energy Conversion from the IEEE PowerEngineering Society in 2004, Patent and InnovationAward from Texas A&M University System in 2007,TEES Faculty Fellow Award in 2006, DistinguishedTeaching Award in 2003, E.D. Brockett Professor-

ship Award in 2002, Eugene Webb Faculty Fellow Award in 2000, and TexasA&M Select Young Investigator Award in 1999. He also received the SpaceAct Award from NASA in 1999, and the Schlumberger Foundation TechnicalAwards in 2001 and 2000. He is the recipient of the 2008 Industrial ElectronicsSociety Electric Machines Committee Second Best Paper Award, the recipientof the IEEE Power Engineering Society Prize Paper Awards in 1996 and 2006,and the 2006 IEEE Industry Applications Society Transactions Third Prize Pa-per Award. He has supervised more than 80 graduate students, postdocs andresearch engineers, published more than 400 technical papers, presented morethan 80 invited lectures all over the world, and has 13 issued and pending U.S.patents. He is the Author of DSP-Based Electromechanical Motion Control,CRC Press, 2003, the Coeditor of Handbook of Electric Motors—2nd Edition,Marcel Dekker, 2004, and the Coauthor of Electric Machines–Modeling, Con-dition Monitoring, and Fault Diagnosis, CRC Press, Florida, 2013.

partial-resonant buck–boost and flyback dc–dc converters

Documents