multifrequency waveguide orthomode transducer

2604 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 8, AUGUST 2005

Multifrequency Waveguide Orthomode TransducerShashi Bhushan Sharma, Vijay Kumar Singh, and Soumyabrata Chakrabarty

Abstract—This paper presents the design and development ofa multifrequency probe-coupled orthomode transducer (OMT)using a circular waveguide as the primary waveguide and arectangular waveguide as the secondary waveguide. Design is pre-sented for a common OMT operating at 6.6, 10.65, 18, and 21 GHzusing four cascaded circular waveguide sections with differentcross-sectional dimensions. An innovative design technique is usedto minimize the inter-port coupling and to maximize the power inthe dominant mode to get the required radiation performance atall the frequency bands using a common radiating aperture. Thesimulated and measured parameters of the OMT and the horn fedby this OMT have been presented.

Index Terms—Cascaded waveguide, corrugated horn, multifre-quency orthomode transducer (OMT), probe coupling.

I. INTRODUCTION

M ICROWAVE radiometers operate in the receive mode atwidely separated frequency bands, which are sensitive

to geophysical parameters. A single aperture antenna operatingat all the frequency bands is preferred for radiometers since in-dependent antennas for each frequency band will require largersatellite space and weight. For scanning microwave radiome-ters, an offset parabolic reflector antenna is generally used with acorrugated horn, which is designed to operate optimally at mul-tiple frequency bands [1]–[3]. For desired radiation patterns ofthe multifrequency horn, an orthomode transducer (OMT) mustprovide dominant mode purity at each frequency band at theinput of the horn. Higher order mode generation and couplingbetween different frequency ports are major problems for thedesign of a multifrequency OMT. Though OMTs have been inuse for a long time, the literature elaborating the analysis and de-sign is limited. Design of dual-band OMTs and the techniquesfor bandwidth enhancement are reported in the literature [4],[5]. To the best of the authors’ knowledge, the design of theOMT for multifrequency operation for more than two frequencybands is not available in the literature. For a common OMT op-erating at four frequencies, the different waveguide sections cor-responding to different frequencies have to be cascaded. As a re-sult of cascading of waveguide sections with different cross-sec-tional dimensions, the lower frequency waveguide sections be-come oversized for higher frequencies and supports higher ordermodes generated due to discontinuities in the form of junc-tions (step or taper) between successive sections [6], [7] andpower-sensing probes or slots on the walls of the primary wave-guide [8], [9].

Manuscript received October 21, 2004; revised January 18, 2005.The authors are with the Microwave Sensors Antenna Division, Antenna

Systems Group, Space Applications Centre, Indian Space ResearchOrganization, Ahmedabad 380 015, India (e-mail: [email protected]).

Digital Object Identifier 10.1109/TMTT.2005.852754

In this paper, a novel configuration for the OMT is presented,which gives optimum performance in terms of mode purity, re-turn loss, inter-port isolation, and radiation characteristics formultifrequency band operation. This technique has been em-ployed to develop a single OMT to operate at four frequencybands at 6.6, 10.65, 18, and 21 GHz. Numerical data on simu-lated modal amplitudes and comparison of simulated and mea-sured return loss, isolation between ports, the radiation patternof corrugated horn [3] fed by the multifrequency OMT havebeen presented.

II. DESIGN AND ANALYSIS

The design goals for this multifrequency OMT was to achieve23 dB or better isolation between two orthogonal ports,17 dB or better return loss, and minimum insertion loss at the

frequency bands 6.6 GHz 125 MHz, 10.65 GHz 150 MHz,18 GHz 200 MHz, 21 GHz 200 MHz. Additionally, higherfrequencies should be decoupled with the lower frequencyports by 18 dB or better for the same polarization. In OMT,coaxial probes [8] or slots [9] are used for power coupling.Maximum power is coupled when a short termination is usedat a distance of from the location of the probe, and thedepth of probe is chosen so as to match the impedance seenby the probe to the characteristic impedance of the coaxialline [8]. This configuration is not suitable in the case whena common aperture corrugated horn is to be operated for anumber of frequency bands. For multifrequency OMT, differentwaveguide cross sections, which correspond to the dominantmode propagation at that frequency band, have to be cascadedsuch that the cross-sectional dimensions at higher frequencybands are at cutoff for the lower frequency bands. The wave-guide sections for lower frequency bands become oversized athigher frequencies and support higher order modes, which areexcited because of structural discontinuities. Apart from this,the higher frequency dominant mode signal gets coupled tothe lower frequency power-sensing ports, thereby increasingthe insertion loss of the device. In the current case, the higherorder modes are generated because of: 1) transition in the formof step or tapered discontinuity between two waveguides ofdifferent cross sections and 2) a probe that senses power at alower frequency band acts as a radial discontinuity for higherfrequency signal. Since the design goal is to ensure dominantmode purity at each frequency band, the higher order modeshave to be suppressed and all the frequency ports have to bedecoupled. The dominant mode purity at each frequency bandof the OMT will ensure the desired radiation patterns of thecorrugated horn antenna. Hence, it is worthwhile to estimate themodal amplitude of different higher order modes generated be-cause of the discontinuities, as discussed above. Finite-elementmethod (FEM)-based electromagnetic (EM) software [Ansoft’s

0018-9480/$20.00 © 2005 IEEE

SHARMA et al.: MULTIFREQUENCY WAVEGUIDE OMT 2605

Fig. 1 Multifrequency waveguide sections joined with step junctions.

High Frequency Structure Simulator (HFSS)] has been used toestimate power of the different higher order modes and the EMmodeling of the transitions and to arrive at an optimum design.The design steps for the development of a multifrequency modetransducer are explained below.

A. Modal Analysis of Step Junctions

Fig. 1 shows four straight circular waveguide sectionsand joined together to form stepped waveguide tran-

sitions. When a pure mode is incident in sectionand corresponding to 21, 18, 10.65, and 6.6 GHz, re-

spectively, it is of interest to evaluate the modal power in theoutput waveguide section , which is oversized for 21, 18, and10.65 GHz and supports the higher order modes generated dueto step junctions between waveguide pairs , and ,and .

For the three-dimensional (3-D) model of the step junctionsused in HFSS, the diameters of sections andare chosen as 9.4, 11, 19, and 32.54 mm for the propagation ofthe dominant mode at 21, 18, 10.65, and 6.6 GHz, respec-tively. The lengths of the individual sections have been selectedas 34, 53, 54, and 78 mm, respectively. The higher order propa-gating modes supported at the waveguide section that feeds acorrugated horn are

and at 18 GHz. Along with thesemodes, additional propagating higher order modes at 21 GHzare and . At 10.65 GHz, and

are the higher order propagating modes in section .The step junction discontinuities generally couple power in thehigher order and modes. From themodal analysis results, it is found that the dominant mode pu-rity is not achievable in section and almost half of the powergets coupled to higher order modes ( and ) at 18 and21 GHz. Additionally, step discontinuity also causes reflectionof the input power.

A corrugated horn fed by the mode transducer of Fig. 1yields poor radiation performance. The cross-pol level at 18and 21 GHz degrades to 13.5 dB, as compared to the caseof pure mode, giving a cross-pol level of the order of

27.5 dB. In order to minimize reflected power and the powercoupled to higher order modes at higher frequencies, the stepjunctions have to be replaced by gradual tapered junctions.Section II-B deals with design and modal analysis of differentwaveguide sections joined by tapered sections.

B. Waveguide Sections Cascaded With Tapered Sections

Fig. 2 shows four circular waveguide sections andjoined together by a tapered section between two successive

Fig. 2 Multifrequency waveguide sections joined with tapered transition.

waveguide sections. The dimensions of the straight waveguidesections are same as mentioned above for Fig. 1.

The taper angle and length of the tapered sections betweenwaveguide sections and

have to be optimized in order to minimize the power inthe higher order modes and maximize the power in the desireddominant mode. A 3-D model of the structure has beeninputted to Ansoft’s HFSS with the initial taper angle and lengthof the taper between two successive waveguide sections, andoptimization was carried out to minimize power in higher ordermodes. The optimum flare angles for the geometry are between3 –6 .

The modal power was computed at the output waveguide sec-tion having the largest cross-sectional dimension consid-ering a unity power incident at the input waveguide sections ateach frequency. It is found that, for optimum flare angles, thepower in the desired mode is of the order of 0.044 dBat 18 and 21 GHz and better at 10.65 and 6.6 GHz for the opti-mized transitions. The power coupling in the higher order modesis negligible and the reflected power is less than 21.6 dB at allthe frequencies. The radiation pattern of a corrugated horn fedwith the geometry of Fig. 2 exhibits symmetrical patterns andcross pol better than 27 dB at all the four frequency bands.Thus, the desired radiation performance of a corrugated horncan be achieved at each frequency band if the horn is fed bywaveguide sections joined with tapered sections, ensuringmode purity.

C. Effect of Coaxial Probes

For exciting the mode in the circular waveguide section,a coaxial probe [8] is used. Since a common aperture OMT is tobe used for all the frequency bands to excite the horn antenna,the waveguide sections for individual frequency bands cannotbe short terminated for maximum power coupling. As seen inFig. 2, the waveguide section for the 6.6-GHz frequency bandis terminated by waveguide section at 10.65 GHz througha tapered transition, which is at cutoff for 6.6 GHz. Similarly,an 18-GHz section is at cutoff for a 10.65- and 21-GHzsection is at cutoff for 18 GHz. In this configuration, thelocation of the probe from the cutoff region, which is in the formof a tapered transition, can be optimized for a particular depth ofthe probe for maximum power coupling to or from the primarycircular waveguide.

Modal analysis using Ansoft’s HFSS was carried out at21 GHz in the presence of coaxial probes in 6.6-, 10.65-, and18-GHz waveguide sections to compute the power coupledto the higher order modes in the output section . Probe or


TABLE IPOWER (IN DECIBELS) IN DOMINANT AND HIGHER ORDER MODES

IN SECTION D FOR TAPERED SECTIONS

post discontinuities generally couple power in the higher orderand modes.

The analysis results are shown in Table I for all frequencies.The Table I shows that the power is coupled to higher ordermodes and also to the lower frequency ports, which effectivelyreduces the power in the desired mode. The return lossalso deteriorates to the order of 10 dB at 18 and 21 GHz dueto the presence of coaxial probes at 6.6 and 10.65 GHz.

In the presence of probes, the cross-polar performance ofa corrugated horn at 18 and 21 GHz deteriorates to around

11 dB, as compared to 27 dB for the configuration of Fig. 2where no probes are considered at lower frequency bands.

It has been found through simulation and also through an-tenna pattern measurement that the cross-polar performance at18 and 21 GHz gets improved if the depth of the probe at 6.6and 10.65 GHz is reduced from its optimum value for maximumpower coupling and impedance matching. Radiation pattern per-formance is further improved if the higher frequency and lowerfrequency ports of same polarization are decoupled. However,with the reduction of the depth of the probes, the impedancematching gets deteriorated at 6.6- and 10.65-GHz ports, thoughthere is an improvement of cross-polar performance at 18 and21 GHz. Thus, the design challenge for this type of multifre-quency mode transducer is to ensure mode purity in the outputsection at all the frequency bands and at the same timeto achieve optimum power coupling, impedance matching, andport-to-port isolation. The overall design of the multifrequencyOMT is presented in Section II-D.

D. Design of Multifrequency OMT

The current design of the mode transducer is based oncoupling from primary cascaded circular waveguide sections tooutput rectangular waveguides WR-137 for 6.6 GHz, WR-75for 10.65 GHz, and WR-42 for 18 and 21 GHz. The schematicof the mode transducer (a circular-to-rectangular waveguideend launcher) at 6.6 GHz is shown in Fig. 3.

The orthogonal ports at the same frequency band have beenseparated by an axial distance of and the angular spacingof 90 to achieve the desired isolation between the two ports.As described in Section II-C, the reduction of the depth of theprobes in the lower frequency waveguide sections from theirresonant depths (quarter-wavelength) improves the cross-polarperformance at higher frequency bands due to reduced power inthe higher order modes.

Fig. 3 Schematic of OMT at a single frequency band.

The probe depths were reduced from resonant depths from11.6 to 7.5 mm in the 6.6-GHz section, from 7 to 4.2 mm inthe 10.65-GHz section, and from 3.6 to 2.5 mm in the 18-GHzsection. The undesirable effect of the reduction in the depthsof the probes is that the real part of the impedance seen by thecoaxial probe is reduced with a reactive impedance, which re-sults in the deterioration of return loss at that frequency. Forexample, the simulated return loss with reduced depth probe isonly 4.5 dB, as compared to a full-depth probe, where it isbetter than 17 dB at 6.6 GHz. The real part of the impedanceseen by the coaxial probe of reduced depth is transformed to therectangular waveguide impedance by multisection ridge wave-guide sections [10], [11] by properly optimizing the heights andwidths of the ridge sections. The reactance due to the reduceddepth probe was cancelled by using a stub pin in the coaxialsection, shorting the inner and outer conductor of the coaxialsection (like a single stub), as shown in the Fig. 3. The shortingpins at 18 and 21 GHz were not required in the coaxial sectionsof the mode transducer. The location of the steps of the ridgesin the rectangular waveguide with respect to the coaxial sectionhave been found to significantly affect inter-port isolation. Forexample, a displacement of 0.25 mm of the step from its op-timum position of 0.5 mm from the onset of the coaxial sectionreduced the isolation of the 18-GHz signal with a 6.6-GHz portfrom 41 to 10.8 dB. Step locations were optimized for bestisolation between lower and higher frequencies.

Modal power distribution and coupling of power to otherports have been computed in the presence of optimized modetransducers consisting of optimized step transformers and lowerdepth probes giving a best return loss at 6.6 and 10.65 GHz.The simulated results for the optimized mode transducers arepresented in Table II, which shows that the maximum power isconfined in the dominant mode at all the frequencies. Thereturn loss at 18 and 21 GHz with optimized mode transducersalso improved to the order of 15 dB, as compared to 10 dBfor the case of full depth coaxial probes present at lower fre-quencies, as described in Section II-C. The return loss at 6.6and 10.65 GHz was optimized for better than 17 dB. Table IIshows improved port-to-port isolation and reduced coupling tohigher order modes than shown in Table I.

At 18 GHz, the simulated radiation patterns of a horn fedby the OMT of optimized step transformers and reduced depthprobes in the lower frequency sections are given in Fig. 4. Sim-ulated results show 9 dB better cross-polar performance of the


TABLE IIPOWER (IN DECIBELS) IN DOMINANT AND HIGHER ORDER MODES

IN SECTIOND IN THE PRESENCE OF MODE TRANSDUCERS

Fig. 4 Patterns of corrugated horn at 18 GHz with optimized mode transducersat 6.6 and 10.65 GHz.

Fig. 5 Return loss for 6.6-GHz circular-to-rectangular waveguide modetransducer for both orthogonal ports.

horn fed with the OMT of reduced depth probes than the fulldepth probes in the lower frequency sections. This improvementis due to the higher isolation with lower frequency ports and lesscoupling to higher order modes.

The simulated and measured return loss and isolation be-tween orthogonal ports is presented in Fig. 5 at 6.6 GHz.

The measured isolation between orthogonal ports at 6.6 GHzis better than 36 dB at the specified bandwidth of 250 MHz.At 10.65 GHz. 15-dB return-loss bandwidth of 300 MHzis achieved by using circular-to-ridged rectangular wave-guide mode transducer. Measured decoupling of 18 dB wasachieved for the 10.65-GHz signal with the 6.6-GHz coaxialprobe. An isolation of better than 29 dB was achieved be-tween orthogonal ports over the band.

The measured results for an 18-GHz OMT are shown inFig. 6. The measured isolation between orthogonal ports at

Fig. 6 Return loss and isolation for 18-GHz OMT.

Fig. 7 Photograph and drawing of the eight-port OMT with orthogonal ports.

18 GHz is of the order of 25 dB over the band. The measuredisolation of the 18-GHz signal is better than 30 and 20 dBwith 6.6- and 10.65-GHz ports, respectively, for same polariza-tion.

At 21 GHz, with the probe depth of 3.1 mm, 15-dB re-turn-loss bandwidth obtained was 360 MHz. The measuredisolation between orthogonal ports is of the order of 25 dBover the band. The isolation of the 21-GHz signal with 6.6and 10.65 GHz was better than 20 dB over the band. Themeasured isolation of the 21-GHz signal with the 18-GHz portwas only from 7 to 10 dB over the band, which could not beimproved due to the comparable size of the OMT at 21 GHz tothat of 18 GHz. The poor isolation adds to increased insertionloss at 21 GHz. The measured insertion loss of the OMT is 0.5,0.7, 1.1, and 1.6 dB at 6.6, 10.65, 18, and 21 GHz, respectively.The photograph and drawing of the developed eight-port OMTwith orthogonal ports at all the frequency bands is shown inFig. 7.

A common aperture conical corrugated horn [3] yieldinggood pattern symmetry and cross-polar performance whenexcited with a pure dominant mode was tested with thecurrent OMT at all four frequency bands. The horn [3] wasdesigned to feed an offset parabolic reflector of focal length toa diameter ratio of 1.8, requiring an edge illumination angleof 13.65 . The measured co-polar and cross-polar radia-tion patterns of the horn fed with this OMT are presented inFigs. 8–11 for one polarization. Similar patterns are achievedfor orthogonal polarization. The measured patterns at 21 GHzshowed slight asymmetry due to a larger ratio of higher ordermodes to dominant mode power, as compared to 18 GHz. A


Fig. 8 Radiation patterns of the horn at 6.6 GHz.

Fig. 9 Radiation patterns of the horn at 10.65 GHz.

Fig. 10 Radiation patterns of the horn at 18 GHz.

Fig. 11 Radiation patterns of the horn at 21 GHz.

larger ratio of higher order mode to dominant mode power isdue to the poor isolation of 21 GHz with the 18-GHz port.

III. CONCLUSION

A novel design of four frequency-band OMT to feed acommon corrugated horn has been presented. In the multi-frequency environment, the methods of controlling power inthe higher order modes and improving isolation of higherfrequencies with lower frequency ports has been described.Modal analysis has been performed to estimate the effects ofsymmetrical step, taper, and asymmetrical probe discontinuitiesin the main waveguide, particularly at higher frequencies. Anoptimum configuration of a multifrequency OMT yieldinga desired isolation of orthogonal ports, isolation of higherfrequencies with lower frequency ports of the same polariza-tion, and maximum power in the dominant mode hasbeen obtained. Optimum radiation performance of the hornexcited with the presented OMT has been achieved for all fourfrequency bands for both polarizations. It was not possible tofabricate the OMT device as a single piece. This was fabricatedin a number of pieces and assembled to make the eight-portdevice. The slight deviation of the measured data from simu-lated data may be attributed to fabrication tolerances and minorassembly and alignment errors. The modal-analysis-baseddesign approach presented in this paper may be applied to thedesign of multifrequency OMTs at other frequency bands.

ACKNOWLEDGMENT

The authors thank Dr. K. N. Shankara, Space ApplicationsCentre (SAC), Ahmedabad, India, for his support. The authorsalso acknowledge and are thankful for the help provided by theEngineers of Microwave Sensors Antenna Division, AntennaSystems Group, SAC.

REFERENCES

[1] E. G. Njoku, J. M. Stacey, and F. T. Barath, “The seasat scanningmulti-channel microwave radiometer (SMMR): Instrument descriptionand performance,” IEEE J. Ocean. Eng., vol. OE-5, no. 2, pp. 100–115,Apr. 1980.

[2] S. B. Sharma, “Antenna system for the multi-frequency scanning mi-crowave radiometer: MSMR,” IEEE Antennas Propag. Mag., vol. 42,no. 3, pp. 21–29, Jun. 2000.

[3] S. B. Sharma and V. K. Singh, “Design of common aperture hybrid modecorrugated horn for multifrequency scanning microwave radiometer,”IETE Tech. Rev., vol. 16, no. 1, pp. 47–52, Jan.–Feb. 1999.

[4] J. Uher, J. Boernemann, and U. Rosenberg, Waveguide componentsfor antenna feed systems: Theory and CAD. Norwood, MA: ArtechHouse, 1993.

[5] S. J. Skinner and G. L. James, “Wide band orthomode transducers,”IEEE Trans. Microw. Theory Tech., vol. 39, no. 2, pp. 294–300, Feb.1991.

[6] W. J. English, “The circular waveguide step discontinuity mode trans-ducer,” IEEE Trans. Microw. Theory Tech., vol. MTT-21, no. 10, pp.633–636, Oct. 1973.

[7] K. Tomiyasu, “Conversion of TE by a large conical junction,” IEEETrans. Microw. Theory Tech., vol. MTT-17, no. 5, pp. 277–279, May1969.

[8] W. W. S. Lee and E. K. N. Yung, “The input impedance of a co-axialline fed probe in a cylindrical waveguide,” IEEE Trans. Microw. TheoryTech., vol. 42, no. 8, pp. 1468–1473, Aug. 1994.

[9] S. B. Sharma, S. B. Chakrabarty, and V. K. Singh, “Moment methodanalysis of a slot coupled circular waveguide orthomode transducer,”Microwave Opt. Technol. Lett., vol. 34, no. 4, pp. 285–289, Aug. 30,2002.

[10] T.-S. Chen, “Calculation of parameters of ridge waveguides,” IRE Trans.Microw. Theory Tech., vol. MTT-5, no. 1, pp. 12–17, Jan. 1957.

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Shashi Bhushan Sharma was born in Morad-abad, India, in 1947. He received the B.E. degreeelectronics and communication and M.E. degreein microwave engineering from the Universityof Roorkee, Roorkee, India, in 1970 and 1972,respectively, and the Ph.D. degree in microwaveengineering from Gujarat University, Ahmedabad,India, in 1987.

He possesses over 32 years of academic and di-versified research and development experience in thedesign and development of antenna systems for satel-

lite communication and remote sensing. He is currently the Group Director ofthe Antenna Systems Group (ASG), Space Applications Centre (SAC), IndianSpace Research Orgainzation (ISRO), Ahmedabad, India. He has authored orcoauthored over 100 publications.

Dr. Sharma was the recipient of the 1992 Dr. Vikram Sarabhai ResearchAward in the field of electronics, telematics, and automation for his outstandingcontributions to the development of various types of antenna systems for ground,airborne, and spaceborne systems.

Vijay Kumar Singh was born in Bahraich Dis-trict, Uttar Pradesh, India, on August 21, 1967.He received the B.Tech. degree in electronics andtelecommunication engineering from the JhuggilalKamlapati Institute of Applied Physics and Tech-nology, Allahabad University, Allahabad, India, in1990, the M.Tech. degree in electronics engineering(microwaves) from the Institute of Technology,Banaras Hindu University, Varanasi, India, in 1992,and is currently working toward the Ph.D. degree atGujarat University, Ahmedabad, India.

Since, 1993, he has been with the Antenna Systems Group (ASG), Space Ap-plications Centre (SAC), Indian Space Research Orgainzation (ISRO), Ahmed-abad, India. Working as Project Manager for Radar Imaging Satellite (RISAT),as well as Oceansat-II missions, he is currently involved in the design and devel-opment of spaceborne active phased-array antennas and scanning scatterometerantennas, respectively. His area of interest is multimode couplers, transducers,wide-band multifrequency feeds, reflectors and dual-polarized microstrip an-tennas for satellite remote sensing application.

Soumyabrata Chakrabarty was born on January3, 1966, in the Karimganj District, Assam, India. Hereceived the B.E. degree in electronics and telecom-munication engineering (with honors) from GauhatiUniversity, Guwahati Assam, India, in 1988, theM.E. degree in electronics and telecommunicationengineering from Jadavpur University, Calcutta,India, in 1992, and the Ph.D. degree in engineeringfrom the Indian Institute of Technology, Kharagpur,India, in 1996.

He is currently with the Antenna Systems Group(ASG), Space Applications Centre (SAC), Indian Space Research Orgainzation(ISRO), Ahmedabad, India, as Senior Scientist/Engineer and Deputy DivisionalHead of the Microwave Sensors Antenna Division, where he has been involvedin the development of antennas related to microwave remote sensing. His areaof interest is computational electromagnetics, microwaves, and millimeter-waveantennas.

multifrequency waveguide orthomode transducer

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