lte physical layer v2

LTE Physical Layer- 3PCA-L1 Certification email: @yahoo.com Page 7

1. Downlink Physical Layer

1.1 OFDMA Principles OFDMA (Orthogonal Frequency division Multiple Access) is a multicarrier scheme. Multicarrier schemes subdivide bandwidth into parallel subchannels, ideally each non-frequency-selective (spectrally-flat gain), overlapping but orthogonal. This avoids need of guard-bands, makeing OFDM highly spectrally efficient, as subchannels can be perfectly separated at the receiver. This makes receiver less complex, attractive for high-rate mobiles. Robustness has to be built up against time-variant channels by employing channel coding. LTE downlink combines OFDM with channel coding and Hybrid Automatic Repeat reQuest (HARQ). OFDM is ideal for broadcast/DL applications for low receiver complexity. OFDM has efficient implementation by means of the FFT. It uses Cyclic Prefix to avoid ISI, enabling block-wise processing. Orthogonal subcarriers avoid spectrum wastage in intersubcarrier guard-bands. Parameters flexibility allows balance the tolerance of Doppler and delay spread.

Key OFDMA points (a) Orthogonal subcarriers with very small inter-subcarrier guard-bands. (b) It makes use of a CP to avoid ISI, enabling block-wise processing. (c) Efficient implementation by means of the FFT. (d) Achieves high transmission rates of broadband transmission, with low receiver complexity. (e) Balanced tolerance of Doppler and delay spread depending on the deployment scenario. (f) It can be extended to a multiple-access scheme, OFDMA, in a straightforward manner. (g) Suited for broadcast or downlink applications because of low receiver complexity while

requiring a high transmitter complexity (expensive PA).

First OFDM patent filed at Bell Labs in 1966, initially only as analog. In 1971, Discrete Fourier Transform (DFT) was proposed. Later in 1980, application of the Winograd Fourier Transform (WFT) or Fast Fourier Transform (FFT) was employed. OFDM then became modulation of choice for ADSL and wireless systems. OFDM tended to focus broadcast systems such as - Digital Video Broadcasting (DVB) and Digital Audio Broadcasting (DAB), and WLANs. Main thing to control in OFDM was PAPR and thats why in low power WLAN it was good. First cellular mobile based on OFDM was proposed in 1985 by IEEE to LTE downlink. Other benefits of OFDM was to operate in different bandwidth according to spectrum availability.

1.1.1 OFDM - Orthogonal Multiplexing Principle Challenge is always in having a symbol period Ts < channel delay spread Td. This generates Intersymbol Interference (ISI), needing complex equalization procedure. Equalization complexity usually is in proportion to square of (channel impulse response length). Data symbols are first serial-to-parallel converted for modulation on M parallel subcarriers.This increases symbol duration by a factor of approx M, > channel delay spread.


Fig 2.1.1.1 OFDM Signal Processing This operation makes time-varying channel impulse response substantially constant during each modulated OFDM symbol. Resulting long symbol duration is virtually unaffected by ISI compared to the short symbol duration. A Serial to Parallel (S/P) converter collects serial data symbols into a data block Sk = [Sk [0] , Sk [1] , . . . , Sk [M 1]]T of dimension M, where the subscript k is the index of an OFDM symbol (spanning the M sub-carriers). The M parallel data streams are first independently modulated resulting in the complex vector Xk = [Xk [0] , Xk [1] , . . . , Xk [M 1]]T . In principle it is possible to use different modulations (e.g. QPSK or 16QAM) on each sub-carrier, the channel gain may differ between sub-carriers, and thus some sub-carriers can carry higher data-rates than others. The vector of data symbols Xk then passes through an Inverse FFT (IFFT) resulting in a set of N complex time-domain samples xk = [xk[0], . . . , xk[N 1]]T . In a practical OFDM system, the number of processed subcarriers is greater than the number of modulated sub-carriers (i.e. N M), with the unmodulated sub-carriers being padded with zeros.

Fig 2.1.1.2 OFDMA tramsmission and reception

A guard period is created at the beginning of each OFDM symbol, to eliminate the remaining impact of ISI. A Cyclic Prefix (CP) is added at the beginning of each symbol xk. The CP is generated by duplicating the last G samples of the IFFT output and appending them at the beginning of xk. This yields the time domain OFDM symbol [xk[N G], . . . , xk[N 1], xk[0], . . . , xk[N 1]]T . CP length G should be longer than the longest channel impulse response to be supported. The CP converts the linear (i.e. aperiodic) convolution of the channel into a circular (i.e. periodic) one which is suitable for DFT processing. The IFFT output is then Parallel-to-Serial (P/S) converted for transmission through frequency-selective channel. Here is an example of OFDM LTE signal. At the receiver, the reverse operations are performed to demodulate the OFDM signal, CP are removed and ISI-free block of samples is passed to the DFT. If number of subcarriers N is designed to be a power of 2, a highly efficient FFT implementation may be used to transform the signal back to the frequency domain. Among the N parallel streams output from the FFT, the modulated subset of M subcarriers are selected and further processed by the receiver. Let x(t) be the signal symbol transmitted at time instant t . The received signal in a multipath environment is then given by r(t) = x(t) * h(t) + z(t), where h(t) is the continuous-time impulse response of the channel, represents the convolution operation and z(t) is the additive noise. Assuming that x(t) is band-limited to [1/2Ts ,1/2Ts], the continuous-time signal x(t) can be sampled at sampling rate Ts such that the Nyquist criterion is satisfied. Due to multipath, several replicas of the transmitted signals arrive at the receiver at different delays. The received discrete-time OFDM symbol k including CP, under the assumption that the channel impulse response has a length smaller than or equal to G, Receiver has to process equalization to recover xk[n] signals. CP of OFDM changes the linear convolution into a circular one. The circular convolution is very efficiently transformed by an FFT into a multiplicative operation in frequency domain. Hence, the transmitted signal over a frequency-selective (multipath) channel is converted into a transmission over N parallel flat-fading channels in


the frequency domain: Rk[m] = Xk[m] H[m] + Zk[m]. As a result the equalization is much simpler than for single-carrier systems and consists of just one complex multiplication per subcarrier.

1.1.2 Peak-to-Average Power Ratio and Sensitivity Major drawbacks of OFDM is that it has a high Peak-to-Average Power Ratio (PAPR). The amplitude variations of OFDM signal can be very high, however PAs of RF transmitters are linear only within a limited dynamic range. Hence, OFDM signal is likely to suffer from non-linear distortion caused by clipping, giving out-of-band spurious emissions and in-band corruption of the signal. To avoid such distortion, PAs should have large power back-offs, leading to inefficient amplification. Let x[n] be the signal after IFFT. PAPR of an OFDM symbol is defined as the square of the peak

amplitude divided by the mean power, i.e. PAPR = Max,n{|x[n]|2} / E{|x[n]|

2}

It is observed that a high PAPR does not occur very often. However, when it does occur, degradation due to PA non-linearities may be expected. PAPR Reduction Techniques Many techniques are studied for reducing the PAPR, but not specified for downlink. An overview of possibilities is provided below.

1. Clipping and filtering . Signal may be clipped, but it causes spectral leakage into adjacent channels, resulting in reduced spectral efficiency, in-band noise, degrading BER. To avoid this problem, oversample the original signal by padding with zeros and processing it using a longer IFFT. Oversampled signal is clipped and then filtered to reduce the out-of-band radiation. This may be is used in LTE.

2. Selected mapping. Whichever phase vector gives Least PAPR, that is used. To recover phase information, separate control signalling is used to tell which phase vector was used. It is not used.

3. Coding techniques. Use code words with lowest PAPR. Complementary codes have good properties to combine both PAPR and forward error correction. It is not used.

Sensitivity to Carrier Frequency Offset and Time-Varying Channels OFDM orthogonality relies that transmitter and receiver operate with exactly same frequency reference, else perfect orthogonality of subcarriers is lost, causing subcarrier leakage (Inter-Carrier Interference (ICI). UE local oscillator frequency drifts are usually greater than in the eNodeB and are typically due to temperature and voltage variation and phase noise. This difference between the reference frequencies is referred as Carrier Frequency Offset (CFO). The CFO can be larger than subcarrier spacing - divided into integer part and fractional part. Frequency error fo = (T+e)df. Where, df is subcarrier spacing,, T is an integer and 0.5


where d is the timing offset in samples corresponding to a duration equal to To.

This phase shift can be recovered as part of the channel estimation operation, with cyclic prefix but not zero-padding.In the general case of a channel with delay spread, for a given CP length the maximum tolerated timing offset without degrading the OFDM reception is reduced by an amount equal to the length of the channel impulse response: To TCP Td. For greater timing errors, ISI and ICI occur. Timing synchronization becomes more critical in long-delay spread channels. Initial timing is achieved by the cell-search and synchronization procedures. Thereafter, for continuous tracking of timing-offset, either CP correlation or Reference Signals (RSs) is used. If an OFDM system, CP is sufficiently designed of lengthG samples such that Channel impulse Response L


further overhead, sub-carrier spacing is kept at 7.5kHz and an extended CP of approx 33 s is used.

Fig 2.1.5.1 -FDD Frame Structure

Fig 2.1.5.2 TDD Frame Structure

Note that, with normal CP, the CP for the first symbol in each 0.5 ms slot is slightly longer than the next six symbols, to accommodate an integer (7) number of symbols in each slot, with assumed FFT block-lengths of 2048. For 20 MHz, FFT order of 2048 is assumed for efficient implementation. However, in practice the implementer is free to use other Discrete Fourier Transform sizes. These parameterizations are designed to be compatible with a sampling frequency of 30.72 MHz, which is 8*3.84Mhz(UMTS sampling rate), for backward compatibility. Thus, the basic unit of time in LTE, is defined as Ts = 1/30.72 s. Lower sampling frequencies (and proportionally lower FFT orders) are always possible to reduce RF and baseband processing complexity for narrower BW: Example, for 5 MHz, FFT order and sampling frequency could be 512 and fs = 7.68 MHz respectively, while only 300 subcarriers are actually modulated with data. For simple implementation, direct current (d.c.) subcarrier is left unused, to avoid d.c. offset errors.

1.1.6 Transmission Resource Structure LTE downlink, consist of user-plane and control-plane data from higher protocol stack layers multiplexed with physical layer signalling. A DL resources possess dimensions of time(slot),


frequency(multiple of 180khz) and space(layer). Layer is defined by multiple antenna transmission and reception. The largest unit of time is the 10 ms radio frame, further subdivided into ten 1 ms subframes, each of which is split into two 0.5 ms slots. Each slot has seven OFDM symbols in normal CP (six if extended CP). In frequency domain, resources are grouped in units of 12 subcarriers (15*12kHz=180 kHz), such that one unit of 12 subcarriers for a duration of one slot is termed a Resource Block (RB).

Fig 2.1.6.1 FDD Downlink Frame sample

Smallest unit of resource is the Resource Element (RE) - one subcarrier for a duration of one OFDM symbol. A RB comprised of 84 REs in normal CP (72 RE in extended CP). Within certain RBs, some REs are reserved for synchronization signals (PSS/SSS), reference signals (RS), control signalling and critical broadcast system information (CFICH,PHICH,PDCCH). Remaining REs are used for data transmission(PDSCH), and are usually allocated in pairs (in time domain) of RBs.

Fig 2.1.6.2 TDD Downlink Frame sample

Two types of frame structure are defined:

1. Frame Structure Type 1(Frequency Division Duplex,FDD) assumes all subframes are available for DL, in paired radio spectrum, or standalone downlink carrier.

2. Frame Structure Type 2(Time Division Duplexing , TDD) in unpaired spectrum, basic structure of RBs and REs remains same, but only a subset of subframes are available for


downlink; remaining subframes are used for uplink or for special subframes which allow switching between DL & UL. In the centre of the special subframes a guard period is provided which allows UL timing to be advanced.

Signal Structure Physical layer translate data into reliable signal for transmission between eNodeB and UE. Each block of data is first protected against transmission errors, first with a Cyclic Redundancy Check (CRC), and then with coding; The initial scrambling stage is applied to all DL channels and helps interference rejection. Scrambling sequence uses order- 31 Gold code, which are not cyclic shifts of each other. Scrambling sequence generator is re-initialized every subframe (except PBCH), based on cell-id, subframe number (within a radio frame), UE identity and codeword id. Scrambling sequence generator is similar to pseudo-random sequence used for Reference Signals, only difference is the method of initialization. A fast-forward of 1600 places is applied at initialization to ensure low cross-correlation between sequences used in adjacent cells. Following scrambling, data bits from each channel are mapped to modulation symbols depending on modulation scheme, then mapped to layers, precoded, mapped to RE, and finally translated into a complex-valued OFDM signal by IFFT. To communicate with eNodeB cells, UE must first identify the DL from one of these cells and synchronize with it. This is achieved by means of special synchronization signals embedded into the OFDM structure by cell search and synchronization. Then UE estimates DL radio channel to perform demodulation of received DL signal, based on pilot signals (reference signals) inserted into DL signal. The channel designs are explained next.


1.2 Synchronization and Cell Search Cell Search executes synchronization for time and frequency parameters, necessary to demodulate DL and to transmit UL with correct timing and acquires some critical system parameters. Three major synchronization requirements:

1. symbol timing determines correct symbol start position and sets the FFT window position; 2. carrier frequency synchronization to reduce frequency errors by oscillator & Doppler shift; 3. sampling clock synchronization.

1.2.1. Synchronization Sequences and Cell Search in LTE Two relevant cell search procedures exist in LTE:

1. Initial synchronization,

UE detects a cell and decodes all information required to register. This is required, for example, when UE is switched on, or it has lost connection to the serving cell.

2. New cell identification,

When UE is already connected and is detecting a new neighbour cell. UE reports new cell measurements to Serving cell for handover. Procedure is repeated periodically until either Scell quality becomes satisfactory again, or UE moves to another cell.

Fig 2.2.1.1 FDD and TDD Synchronization Signalling

Synchronization procedure detects specially designed Primary Synchronization Signal (PSS) and Secondary Synchronization Signal (SSS). This enables time and frequency synchronization, provides the UE with physical cell identity (PCI) and CP, and informs UE whether cell uses FDD or TDD. Here is figure explaining relative location of PSS and SSS in frame structure of FDD and TDD respectively.

Fig 2.2.1.2 Cell Synchronization Process

In initial synchronization, UE proceeds to decode PBCH for critical system information (SI). For new cell identification, UE does not need to decode PBCH; it makes quality-level measurements (RSRP/RSRQ) and reports to the serving cell.


The sync signals are transmitted periodically, twice per 10 ms radio frame.

In FDD cell, PSS is always located in the last symbol of the 0th and 10

th slots of each frame,

thus enabling UE to acquire the slot boundary timing independently of the CP. SSS is located in the symbol immediately preceding PSS, for coherent detection of SSS relative to PSS.

In TDD cell, PSS is located in 2nd

symbol of the 2nd

and 12th slots, while the SSS is located 3

symbols earlier and falls in previous slot. Position of SSS changes depending on CP length of the cell. At this stage, CP length is unknown and SSS is blindly detected by checking for SSS at expected positions.

PSS in a given cell is same in every subframe, SSS may change & thus UE knows the position of the 10 ms radio frame boundary. PSS and SSS are transmitted in the central six Resource Blocks (RBs), irrespective of the system BW (6 to 110 RBs), without knowing BW. The PSS and SSS are each comprised of a sequence of length 62 symbols, mapped to the central 62 subcarriers around d.c. subcarrier which is left unused. Five REs at each extremity of each sync sequence are not used. Thus a UE can detect the PSS and SSS with size-64 FFT and a lower sampling rate if all 72 subcarriers were used. In case of MIMO at eNodeB, PSS and SSS are always transmitted from same antenna port in a subframe, while between different subframes they may be transmitted from different antenna ports for diversity. PSS and SSS sequence indicate one of 504 unique PCI, grouped into 168 groups of three identities. The three identities in a group are assigned to cells under same eNodeB. Three PSS sequences are used to indicate the cell identity within the group, and 168 SSS sequences are used to indicate the identity of the group. PSS uses ZadoffChu sequences

1.2.2. ZadoffChu Sequences ZadoffChu (ZC) sequences (Generalized Chirp-Like (GCL) sequences) are non-binary unit-amplitude sequences, which satisfy a Constant Amplitude Zero Autocorrelation (CAZAC) property. The ZC sequence of odd-length NZC is given by

aq(n) = exp [j2q (n(n + 1)/2 + ln)/ NZC ]

where q {1, . . . , NZC 1} is the ZC sequence root index, n = 0, 1, . . . , NZC 1, l N is any integer (In LTE l = 0). ZC sequences have the following important properties. Property 1. A ZC sequence has constant amplitude which limits PAPR and generates bounded and time-flat interference to other users, and its NZC-point DFT also has constant amplitude. Property 2. ZC sequences of any length have ideal cyclic autocorrelation (correlation with circularly shifted version of itself is a delta function). ZC periodic autocorrelation is exactly zero for 0 and it is non-zero for = 0, whereas PN periodic autocorrelation shows significant peaks, some above 0.1, at non-zero lags. CAZAC sequence allows multiple orthogonal sequences to be generated from the same ZC sequence. Indeed, if the periodic autocorrelation of a ZC sequence provides a single peak at the zero lag, the periodic correlation of the same sequence against its cyclic shifted replica provides a peak at lag NCS, where NCS is the number of samples of the cyclic shift. This creates a Zero-Correlation Zone (ZCZ) between the two sequences. As a result, as long as the ZCZ is dimensioned to cope with the largest possible expected time misalignment between them, the two sequences are orthogonal for all transmissions within this time misalignment. Property 3. The absolute value of the cyclic cross-correlation function between any two ZC sequences is constant and equal to 1/NZC, if |q1 q2| (where q1 and q2 are the sequence indices) is relatively prime with respect to NZC . Selecting NZC as a prime number results in NZC 1 Zaddoff-Chu sequences which have the optimal cyclic cross-correlation between any pair. Cyclic extension or truncation preserves both the constant amplitude property and the zero cyclic autocorrelation property for different cyclic shifts. The DFT of a ZC sequence xu(n) is a weighted cyclicly-shifted ZC sequence Xw(k) such that w = 1/u mod NZC. This means that a ZC sequence can be generated directly in the frequency domain without the need for a DFT operation.


1.2.3. Primary Synchronization Signal (PSS) Sequences There are 0 to 503 (total 504) PCI are available to be assigned. Every cell will have a PCI where

PCI=3* NID,1 + NID,2.

The NID,1 is the PCI group ID - , NID,1 can have values 0 to 167 and

NID,2 is the PCI local (may be sector) ID - NID,2 can have values 0 to 2.

How does the PCI optimization affect my Network? Well, this is the main parameter, by which the

PSS, SSS and reference signals will be generated. Even your scrambling code with which every DL

and UL signal will be scrambled, will depend on this. So, every generated signals uniqueness

depends on this parameter. Lets understand how some of the signals are generated based on PCI.

If PSS, SSS, RS and other generated signals are not unique, then my every operation will be affected

and it may reflect as latency in Synchronization detection, Interference and lower SINR values for

signals, which will end up in low CQI.

1.2.4. PSS Generation The PSS represented by d(u,n) is generated from a frequency-domain Zadoff-Chu sequence

according to

61,...,32,31

30,...,1,0)(

63

)2)(1(

63

)1(

ne

nend

nnuj

nunj

u

where the Zadoff-Chu root sequence index u is given by following table.

(2)IDN

Root index u

0 25

1 29

2 34

The mapping of PSS to resource elements depends on the frame structure, FDD or TDD. The

sequence d(u,n) is mapped to the resource elements according to

231

61,...,0 ,

RBsc

DLRB

,

NNnk

nnda lk


For FDD, the PSS is mapped to the last OFDM symbol in slots 0 and 10. For TDD, the PSS is apped

to the third OFDM symbol in subframes 1 and 6. Resource elements (k,l) in the OFDM symbols used

for transmission of the primary synchronization signal where

66,...63,62,1,...,4,5

231

RBsc

DLRB

n

NNnk

are reserved and not used for transmission of the primary synchronization signal. N=0,1,61 are used with above formulae.

PSS is constructed from a freq-domain ZC sequence of length 63, with middle element punctured to avoid transmitting on d.c. subcarrier. This set of roots for ZC sequences was chosen for its good periodic autocorrelation and cross-correlation properties. These sequences have a low-frequency offset sensitivity (maximum undesired autocorrelation peak /desired correlation peak) at a certain frequency offset, giving best robustness. Also the ZC sequences are robust against frequency drifts. Thus, PSS can be easily detected during the initial synchronization with a frequency offset up to 7.5 kHz. The selected root combination satisfies time-domain root-symmetry, sequences 29 and 34 are complex conjugates of each other and can be detected with a single correlator. UE must detect PSS without any prior knowledge of the channel, so noncoherent correlation is required for PSS timing detection.

1.2.5. Secondary Synchronization Signal (SSS) Sequences SSS maximum length M-sequences, can be created by cycling through every possible state of a shift register of length n, resulting in M-sequence of length 2n 1. Two length-31 BPSK secondary sync codes (SSC1(even (di)) and SSC2(odd d(i)) are interleaved to construct SSS sequence in frequency-domain. Two codes are two different cyclic shifts of a single length-31 M-sequence. Cyclic shift indices are derived from a function of PCI group. Two codes are alternated between the first and second SSS in each radio frame.

5 subframein )(

0 subframein )()12(

5 subframein )(

0 subframein )()2(

)(11

)(0

)(11

)(1

0)(

1

0)(

0

10

01

1

0

nzncns

nzncnsnd

ncns

ncnsnd

mm

mm

m

m

with

30,30

2)1(,2)1(

31mod131

31mod

(1)ID

(1)ID(1)

ID

01

0

NqqqN

qqqNm

mmm

mm

Thus UE determines the 10 ms radio frame timing from a single observation of a SSS. SSC2 is scrambled by a sequence that depends on the index of SSC1. Sequence is then scrambled by a code that depends on the PSS. Scrambling code is mapped to the PCI within the group corresponding to the target eNodeB. The resource mapping is done as per the following:


2 typestructure framefor 11 and 1 slotsin 1

1 typestructure framefor 10 and 0 slotsin 2

231

61,...,0 ,

DLsymb

DLsymb

RBsc

DLRB

,

N

Nl

NNnk

nnda lk

SSS sequences are spectrally flat. PSS, the SSS can be detected with a frequency offset up to 7.5 kHz. Channel is known based on the PSS sequence first and then SSS detection is done. However, in the case of synchronized neighbouring eNodeBs, coherent detector performance can be degraded. If an interfering eNodeB employs the same PSS, phase difference between them can have adverse impact on estimation of the channel coefficients. If BW of the channel is less than the six RB for SSS, impact may be bad, hence minimum 6 RB legth is chisen. M-sequence and WalshHadamard matrices are similar and index remapping is done. This reduces complexity of SSS

detector, as complexity = N log2 N with N=32, complexity= 32 log2 32 = 160.

1.2.6. Cell Search Performance A new cell detection delay for UE and report it to S-eNB should be less than acceptable threshold. eNodeB uses the reports to prepare intra- or inter-frequency handover. A multicell environment with three cells with different transmitted powers with synchronized and unsynchronized eNodeBs should be analysed. For the propagation channel, various multipath fading, at least two receive antenna, UE speeds, (5 km/h, 300 km/h) should be considered. Cell search performance is measured as 90-percentile(maximum time required to detect a target cell 90%of the time) identification delay. After detection of PSS-SSS, RSRP is measured. For initial synchronization case the time taken to decode PBCH is adapted, and not just of reporting of measurements on RS.For inter-frequency handover, performance can be derived from the intra-frequency performance timing.

Coherent Versus Non-Coherent Detection A coherent detector uses knowledge of the channel, while a non-coherent detector uses an optimization metric of average channel statistics. In PSS, non-coherent (No channel estimation available) detection is used, while for SSS, coherent (channel estimation) or non-coherent techniques can be used.

1.2.7. Reference Signals and Channel Estimation In any communication system signal x transmitted by A passes through a radio channel H (exhibit multipath fading, causing ISI) and suffers additive noise before being received by B. To remove ISI, equalization, detection algorithms and knowledge of Channel Impulse Response (CIR) is used. OFDMA is quite robust against ISI by CP which allows very good equalization at receiver. Coherent detection uses amplitude and phase information exchanged between eNodeBs and UEs. This comes at a price of overhead of channel estimation by exploiting known signals which do not carry any data, sacrificing spectral efficiency. Known reference signals are inserted into the transmitted signal structure. Reference signals(known) are multiplexed with data symbols (unknown at receiver) in either frequency, time or code domains. Time multiplexing, known preamble-based training transmission also is another technique. Orthogonal RS multiplexing is the most common technique. OFDM transmission is a two-dimensional lattice in time and frequency, which helps multiplexing of RSs mapped to specific REs according to a specific pattern. Since RS are sent only on particular OFDM REs (particular symbols and subcarriers), channel estimates for non-RS REs have to be computed via interpolation.

1.2.8. Design of Reference Signals in LTE In DL, three different types of RS are provided:


1. Cell-specific RSs ( common RSs). 2. UE-specific RSs, in the data for specific UEs. 3. MBSFN-specific RSs, for MBSFN.

1.2.9. Cell-Specific Reference Signals (CRS) In an OFDM-based system, an equidistant arrangement of RS in the lattice structure diamond shape achieves the Minimum Mean Squared Error (MMSE) estimate. In time domain, RS spacing in is governed by maximum Doppler spread (highest speed-540km/h(150m/s)) to be supported. Doppler shift is fd = (fc v/c) where fc is the carrier frequency, v is UE speed in m/s, and c= 3 * 10

8 m/s. Considering fc = 2 GHz (2*10

9Hz) and v = 500 km/h, fd

=(2*109*150/3*10

8) =1000 Hz. According to Nyquists sampling theorem, minimum sampling

frequency to reconstruct the channel is Tc = 1/(2fd)= 0.5 ms. This implies that two RS/slot are needed to estimate channel correctly.

Fig 2.2.8.1 Cell RS for 1, 2 and 4 antenna

In frequency domain, there is one RS every six subcarriers on each symbols including RS symbol, but staggered so that within each RB there is one RS every 3 subcarriers. This spacing is goverened by expected coherence BW of channel, governed by channel delay spread. The 90% and 50% coherence BW are given respectively by Bc,90% = 1/50d=20kHz and Bc,50% = 1/5d=200kHz where d is the r.m.s delay spread=1000ns. In LTE spacing between two RS in frequency, is 45 kHz (3 symbols), enough to resolve expected frequency domain variations of the channel. RS patterns are designed to work with MIMO antennas defined for multiple antenna ports at eNodeB. An antenna port may be either a single physical antenna, or a combination of multiple physical antenna elements. The transmitted RS in a given antenna port defines the antenna port from the point of view of the UE, and enables UE to derive channel estimate for that antenna port.


Fig 2.2.8.2 Antenna ports and Physical antennas

Up to eight cell-specific antenna ports may be used by eNodeB, requiring UE to derive up to eight separate channel estimates. For each antenna port, a different RS pattern is designed, to minimize intra-cell interference between multiple transmit antenna ports. Rp is used for RS Tx on antenna port p. Also, when a RE is used for RS on one antenna port, corresponding RE on other antenna ports is set to zero to limit interference. Mark that, density of RS for third and fourth antenna ports is half of the first two, to reduce overhead. In cells with a high prevalence of high-speed users, use of four antenna ports is unlikely, RSs with lower density can provide sufficient channel estimation accuracy.

Fig 2.2.8.3 Antenna port example of port 0 and port 5

All the RSs (cell-specific, UE-specific or MBSFN specific) are QPSK modulated to ensure low PAPR. The signal can be written as r(l,ns,m) = 1/2[1-2c(2m)] + j1/2[1-2c(2m+1)] where m is RS index, ns = slot number and l =symbol number within slot, c(i) is length-31 Gold sequence, with different initialization values depending on type of RSs. RS sequence carries unambiguously one of the 504 different cell identities, Ncell ID. For the cell-specific RSs, a cell-specific frequency shift (Ncell ID mod 6) is also applied. This shift avoids collisions between common RS from up to six adjacent cells. Transmission power of RS is boosted, up to max 6 dB relative to surrounding data symbols, designed to improve channel estimation. If adjacent cells also transmit high-power RS on same REs, interference will prevent the gain.

1.2.10. UE-Specific Reference Signals(URS) UE-specific RS may be used in addition to CRSs, embedded only in a specific UEs scheduled RBs, using a distinct antenna port. UE is expected to use them to derive the channel estimate for demodulating data in PDSCH RBs.


Fig 2.2.9 Port 5, UE specific Reference Signals

A typical usage of URSs is beam-forming of data transmissions to specific UEs. Rather than using physical antennas of other CRS antenna ports, eNodeB may use a correlated array of antenna elements to generate a narrow beam in the direction of a particular UE. Beam will experience different channel response requiring URSs to demodulate the beamformed data coherently. The URS structure is chosen not to collide with CRSs, and hence URSs does not affect CRSs. URSs have a similar pattern as CRSs allowing re-use similar channel estimation algorithms. Density is half of CRS, minimizing the overhead.

1.2.11. RS-Aided Channel Modelling and Estimation Channel estimation problem is related to physical propagation, number of transmit and receive antennas, BW, frequency, cell configuration and relative speed.

1. frequencies and BW determine the scattering. 2. Cell deployment governs multipath, delay spread and spatial correlation. 3. Relative speed sets time-variations.

Propagation conditions characterize the channel in three dimension (frequency, time and spatial) domains. Each MIMO multipath channel component can experience different scattering conditions across the three domains. LTE specifications do not mandate any specific channel estimation technique, and there is therefore complete freedom in implementation provided that the performance requirements are met and the complexity is affordable.

1.2.12. Frequency Domain Channel Estimation The natural approach to estimate the whole CTF is to interpolate its estimate between the reference symbol positions. As a second straightforward approach, the CTF estimate over all subcarriers can be obtained by IFFT interpolation. More elaborate linear estimators derived from both deterministic and statistical viewpoints are proposed -Least Squares (LS), Regularized LS, Minimum Mean-Squared Error (MMSE) and Mismatched MMSE. It is seen that IFFT and linear interpolation methods yield lowest performance. The regularized LS and the mismatched MMSE perform exactly equally. Optimal MMSE estimator outperforms any other estimator. MMSE-based channel estimation suffers the least band-edge degradation, while all the other methods presented are highly adversely affected.

1.2.13. Time-Domain Channel Estimation Time-Domain (TD) channel estimation requires sufficient memory for buffering soft values of data over several symbols while the channel estimation is carried out. However, correlation between consecutive symbols decreases as UE speed increases. TD correlation is inversely proportional to the UE speed sets a limit on the possibilities for TD filtering in high-mobility conditions.

Finite and Infinite Length MMSE-(TD-MMSE) The statistical TD filter which is optimal in terms of Mean Squared Error (MSE) can be approximated in the form of a finite impulse response filter. It can be observed that, unlike Frequency-Domain (FD) MMSE filtering, the size of the matrix to be inverted for a finite-length TD-MMSE estimator is independent of the channel length L but dependent on the chosen FIR order M. Similarly to the FD counterpart, the TD-MMSE estimator requires knowledge of the PDP, the UE speed and the noise variance.


Normalized Least-Mean-Square(NLMS) An adaptive estimation approach can be considered which does not require knowledge of second-order statistics of both channel and noise. A feasible solution is the Normalized Least-Mean-Square (NLMS) estimator. It can be observed that the TD-NLMS estimator requires much lower complexity compared to TD-MMSE as no matrix inversion is required, as well as not requiring any a priori statistical knowledge. Other adaptative approaches could also be considered such as Recursive Least Squares (RLS) and Kalman-based filtering.

1.2.14. Spatial Domain Channel Estimation(SD-MMSE) LTE UE is designed for MIMO. Consequently, whenever the channel is correlated in the spatial domain, the correlation can be exploited to provide a further means for enhancing the channel estimate. If it is desired to exploit spatial correlation, a natural approach is again offered by Spatial Domain (SD) MMSE filtering.

lte physical layer v2

Documents

ofdm patent

makeing ofdm

benefits of ofdm

modulated ofdm symbol

ofdm signal processing

low receiver complexity

channel coding

orthogonal subcarriers