in0 thesis 初稿 - national chiao tung university ·...
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國 立 交 通 大 學 電機與控制工程研究所
碩 士 論 文
晶片內部 5-Gbs 低功率脈波訊號傳輸介面
5-Gbs Low Power On-Chip Pulse Signaling Interface
研 究 生方盈霖
指導教授蘇朝琴 教授
中 華 民 國 九 十 六 年 九 月
2
晶片內部 5-Gbs 低功率脈波訊號傳輸介面
5-Gbs Low Power On-Chip Pulse Signaling Interface
研 究 生方盈霖 Student Ying-Lin Fang
指導教授蘇朝琴 教授 Advisor Chau Chin Su
國 立 交 通 大 學
電機與控制工程研究所
碩士論文
A Thesis
Submitted to Department of Electrical and Control Engineering
College of Electrical Engineering and Computer Science
National Chiao Tung University
in partial Fulfillment of the Requirements
for the Degree of
Master
in
Electrical and Control Engineering
September 2007
Hsinchu Taiwan Republic of China
中 華 民 國 九 十 六 年 九 月
i
晶片內部 5-Gbs 低功率脈波訊號傳輸介面
研究生 方盈霖 指導教授 蘇朝琴 教授
國立交通大學電機與控制工程研究所
摘 要
本論文提出一個在晶片內部的脈波傳輸介面來達到 SOC 晶片內的長距離低
功率消耗的傳輸接收電路研究內容包括脈波傳輸端電路脈波接收端電路與晶
片內部的差動傳輸線傳輸端部分實現電容偶合的方式產生脈波訊號經由長距
離的差動傳輸線在接收端同樣利用電容偶合的方式將脈波訊號偶合回接收端電
路設計上首先經由利用較高的傳輸端端電組來將傳送出的脈波訊號振幅提
升並配合提出的傳輸端 de-emphasis 電路架構可將脈波的尾端消除以減少訊
號 ISI 效應接收端則是自偏壓電路將脈波訊號載回接收端共模準位經過放大
及電路將脈波訊號放大後由栓鎖閘將訊號轉回 NRZ 訊號整個傳輸電路在台積電
RF013μm 製程下傳輸 5Gbps 的亂數資料功率消耗在傳輸端 34mW接收端
32mW 總功率消耗 64mW傳輸距離 5mm
關鍵字 脈波傳輸電容偶合晶片內部傳輸高速傳輸電路
ii
5-Gbs Low Power On-Chip Pulse Signaling Interface
Student Ying Lin Fang Advisor Chau Chin Su
Department of Electrical and Control Engineering National Chiao Tung University
Abstract
In this thesis we propose an on-chip pulse signaling communication It can be
used for long distance and low power interconnection on SOC The pulse signaling
communication consists of a transmitter an on-chip transmission-line and a receiver
By increase the termination resistance at the near end we can increase the amplitude
of the transmitted pulse signal And then a de-emphasis circuit is employed to reduce
the ISI effect both in the transmitter and in the receiver A TSMC 013um RF process
was utilized in our design In the simulation result 5Gbps signal transmission can be
achieved through a 5mm-length differential interconnect The power consumption at
Tx and Rx are 32mW and 34mW respectively and the total power consumption is
66mW
Keyword AC coupled pulse signaling capacitive coupling on-chip
communication driver receiver de-emphasis equalization
iii
誌 謝
首先要感謝我的指導教授 蘇朝琴老師感謝老師將近五年時間內從大二基
礎電子學到研究所論文的栽培您費心指導我在實驗室內的所有學習到做研究應
有的精神與態度將陪我進入碩士後的人生旅途
感謝實驗室內煜輝鴻文鴻愷(丸子)仁乾孟昌盈杰等博士班學長
們的指導忠傑緯達(阿達)瑛佑冠羽(KU)柏成(CGU)俊銘(阿銘)順
閔宗諭匡良楙軒智琦冠宇(小冠)等歷屆碩班學長傳授所學汝敏慶
霖(教主)永祥(祥哥)英豪(小馬)議賢村鑫威翔(小潘潘)存遠勇志
(阿伯)季慧雅婷挺毅俊彥繁祥(孔哥)碩廷子俞等同學和學弟妹在
碩班日子的相處和學習依萍俊秀雅雯上容佳琪嘉琳(大姐)志憲(totoro)
等 918 之友在生活上的照顧謝謝你們陪我度過碩班的日子
此外也要由衷感謝蔡中庸老師給予了我兩年碩士生涯的所有助教機會這些
助教收入雖然不至優渥但也等同於一筆獎助學金足夠讓我在研究之餘不至於
太過擔心交大生活上所需的開銷蔡媽妳對教學的熱忱與不放棄任何學生的愛心
在這段期間深深撼動著我也讓我更懂得回饋與感恩
平日研究外的生活兩位家教小妹瑜和宜安妳們給了我很多八九年級生
新穎的想法西洋棋社的 zayinacmonkyy浩浩阿杰mediclightningfox
給了很多棋藝上的切磋電控 16 號家族的鎮源(ican)普暄虹君維祐永遠
忘不了你們給的畢業禮物和家族的溫情室友們宇軒(鯉魚)旻廷依寰(Hubert)
將近五年的朝夕相處感謝你們的包容和帶給我的快樂
最後要感謝老爸老媽爺爺奶奶老哥老妹及女友上容在這段期間內
無盡的鼓勵與支持你們對我付出的關心是永遠報答不完的僅將自己這段時間
的努力與收穫獻給我愛的你們
盈霖 20070904
List of Contents
iv
List of Contents Abstract helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
Acknowledgements helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
List of Contents helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip iv
List of Tables helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vi
List of Figureshelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vii
Chapter 1 Introduction 1 11 Motivationhelliphelliphelliphelliphelliphelliphelliphelliphellip1
12 Link Components 3
13 Thesis Organizationhelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip3
Chapter 2 Background Study5 21 High Speed Serial Link Interface helliphelliphelliphelliphelliphelliphelliphelliphellip5
211 PECL Interfacehelliphelliphelliphelliphellip7
212 CML Interfacehelliphelliphellip7
213 LVDS Interfacehelliphelliphellip8
22 AC Coupled Communicationhelliphelliphellip 10
221 AC Coupled Communicationhelliphelliphellip10
222 Typical pulse transmitterreceiver11
Chapter 3 On-Chip Pulse Signaling14 31 Pulse Signaling Schemehelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip15
32 On-Chip Pulse Signaling Analysishelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip 16
33 On-Chip Differential Transmission Linehelliphelliphelliphelliphelliphelliphelliphelliphellip 18
331 Mode of Transmission Linehelliphellip helliphelliphelliphelliphelliphelliphelliphelliphellip 18
332 Geometry Analysis of Transmission Linehelliphelliphelliphelliphelliphellip20
Chapter 4 5Gbps Pulse Transmitter Receiver Circuit Design23 41 Pulse Transmitter Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip23
411 Voltage Mode Transmitter with De-emphasishelliphelliphelliphellip23
412 The Proposed De-emphasis Circuithelliphelliphelliphelliphelliphellip25
42 Pulse Receiver Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip28
421 Receiver End Terminationhelliphelliphelliphellip 29
List of Contents
v
422 Self-Bias and Equalization Circuithelliphelliphelliphellip 31
423 Inductive Peaking Amplifierhelliphelliphelliphellip 32
424 Non-clock Latch with Hysteresishelliphelliphellip35
Chapter 5 Experiment39 51 Simulation Setup39
52 Measurement Considerations41
53 Experiment45
Chapter 6 Conclusion47
Bibliography49
List of Tables
vi
List of Tables Table 21 Electrical characteristics of the CMLhelliphelliphelliphellip8
Table 22 Electrical characteristics of LVDShellip9
Table 31 Parasitic RLC of the on-chip transmission linehellip20
Table 51 Specification Tablehellip43
Table 52 Comparison of high speed data communicationhellip45
List of Figures
vii
List of Figures Fig 11 On-chip long distance interconnection2
Fig 12 Basic link components the transmitter the channel and the receiver3
Fig 21 A generalized model of a serial link6
Fig 22 High speed interfacehelliphelliphellip6
Fig 23 (a) output structure of PECL hellip 7
Fig 23 (b) input structure of PECLhellip 7
Fig 24 (a) output structure of CML hellip 8
Fig 24 (b) input structure of CMLhellip 8
Fig 25 (a) output structure of LVDS helliphellip 9
Fig 25 (b) input structure of LVDS hellip 9
Fig 26 (a) AC coupled communication helliphellip 10
Fig 26 (b) pulse waveform in channelhellip 10
Fig 27 Voltage mode transmitter 11
Fig 28 (a) pulse receiver 12
Fig 28 (b) equivalent circuit with enable on 12
Fig 29 Low swing pulse receiver 13 Fig 31 Pulse signaling scheme 15
Fig 32 (a) On-chip pulse signaling model17
Fig 32 (b) Transient response of the transmitterhelliphelliphellip 17
Fig 33 On-chip transmission line as a distributed RLC transmission line 19
Fig 34 (a) Micro-strip structure and the parasitic effect 20
Fig 34 (b) Cross section of the on-chip transmission line 20
Fig 35 Transmission line dimension vs characteristic impedancehellip 22
Fig 36 Transmission line dimension vs parasitic RtotalCtotalhelliphellip 22
Fig 41 Output pulse signal at transmitter with different termination resistance24
Fig 42 Voltage mode transmitter (a) without de-emphasis 25
Fig 42 (b) with de-emphasis 25
Fig 43 Coupled pulse signal with de-emphasis circuit 26
Fig 44 Pulse transmitter26
Fig 45 Pulse Transmitter design flowcharthellip 27
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
2
晶片內部 5-Gbs 低功率脈波訊號傳輸介面
5-Gbs Low Power On-Chip Pulse Signaling Interface
研 究 生方盈霖 Student Ying-Lin Fang
指導教授蘇朝琴 教授 Advisor Chau Chin Su
國 立 交 通 大 學
電機與控制工程研究所
碩士論文
A Thesis
Submitted to Department of Electrical and Control Engineering
College of Electrical Engineering and Computer Science
National Chiao Tung University
in partial Fulfillment of the Requirements
for the Degree of
Master
in
Electrical and Control Engineering
September 2007
Hsinchu Taiwan Republic of China
中 華 民 國 九 十 六 年 九 月
i
晶片內部 5-Gbs 低功率脈波訊號傳輸介面
研究生 方盈霖 指導教授 蘇朝琴 教授
國立交通大學電機與控制工程研究所
摘 要
本論文提出一個在晶片內部的脈波傳輸介面來達到 SOC 晶片內的長距離低
功率消耗的傳輸接收電路研究內容包括脈波傳輸端電路脈波接收端電路與晶
片內部的差動傳輸線傳輸端部分實現電容偶合的方式產生脈波訊號經由長距
離的差動傳輸線在接收端同樣利用電容偶合的方式將脈波訊號偶合回接收端電
路設計上首先經由利用較高的傳輸端端電組來將傳送出的脈波訊號振幅提
升並配合提出的傳輸端 de-emphasis 電路架構可將脈波的尾端消除以減少訊
號 ISI 效應接收端則是自偏壓電路將脈波訊號載回接收端共模準位經過放大
及電路將脈波訊號放大後由栓鎖閘將訊號轉回 NRZ 訊號整個傳輸電路在台積電
RF013μm 製程下傳輸 5Gbps 的亂數資料功率消耗在傳輸端 34mW接收端
32mW 總功率消耗 64mW傳輸距離 5mm
關鍵字 脈波傳輸電容偶合晶片內部傳輸高速傳輸電路
ii
5-Gbs Low Power On-Chip Pulse Signaling Interface
Student Ying Lin Fang Advisor Chau Chin Su
Department of Electrical and Control Engineering National Chiao Tung University
Abstract
In this thesis we propose an on-chip pulse signaling communication It can be
used for long distance and low power interconnection on SOC The pulse signaling
communication consists of a transmitter an on-chip transmission-line and a receiver
By increase the termination resistance at the near end we can increase the amplitude
of the transmitted pulse signal And then a de-emphasis circuit is employed to reduce
the ISI effect both in the transmitter and in the receiver A TSMC 013um RF process
was utilized in our design In the simulation result 5Gbps signal transmission can be
achieved through a 5mm-length differential interconnect The power consumption at
Tx and Rx are 32mW and 34mW respectively and the total power consumption is
66mW
Keyword AC coupled pulse signaling capacitive coupling on-chip
communication driver receiver de-emphasis equalization
iii
誌 謝
首先要感謝我的指導教授 蘇朝琴老師感謝老師將近五年時間內從大二基
礎電子學到研究所論文的栽培您費心指導我在實驗室內的所有學習到做研究應
有的精神與態度將陪我進入碩士後的人生旅途
感謝實驗室內煜輝鴻文鴻愷(丸子)仁乾孟昌盈杰等博士班學長
們的指導忠傑緯達(阿達)瑛佑冠羽(KU)柏成(CGU)俊銘(阿銘)順
閔宗諭匡良楙軒智琦冠宇(小冠)等歷屆碩班學長傳授所學汝敏慶
霖(教主)永祥(祥哥)英豪(小馬)議賢村鑫威翔(小潘潘)存遠勇志
(阿伯)季慧雅婷挺毅俊彥繁祥(孔哥)碩廷子俞等同學和學弟妹在
碩班日子的相處和學習依萍俊秀雅雯上容佳琪嘉琳(大姐)志憲(totoro)
等 918 之友在生活上的照顧謝謝你們陪我度過碩班的日子
此外也要由衷感謝蔡中庸老師給予了我兩年碩士生涯的所有助教機會這些
助教收入雖然不至優渥但也等同於一筆獎助學金足夠讓我在研究之餘不至於
太過擔心交大生活上所需的開銷蔡媽妳對教學的熱忱與不放棄任何學生的愛心
在這段期間深深撼動著我也讓我更懂得回饋與感恩
平日研究外的生活兩位家教小妹瑜和宜安妳們給了我很多八九年級生
新穎的想法西洋棋社的 zayinacmonkyy浩浩阿杰mediclightningfox
給了很多棋藝上的切磋電控 16 號家族的鎮源(ican)普暄虹君維祐永遠
忘不了你們給的畢業禮物和家族的溫情室友們宇軒(鯉魚)旻廷依寰(Hubert)
將近五年的朝夕相處感謝你們的包容和帶給我的快樂
最後要感謝老爸老媽爺爺奶奶老哥老妹及女友上容在這段期間內
無盡的鼓勵與支持你們對我付出的關心是永遠報答不完的僅將自己這段時間
的努力與收穫獻給我愛的你們
盈霖 20070904
List of Contents
iv
List of Contents Abstract helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
Acknowledgements helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
List of Contents helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip iv
List of Tables helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vi
List of Figureshelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vii
Chapter 1 Introduction 1 11 Motivationhelliphelliphelliphelliphelliphelliphelliphelliphellip1
12 Link Components 3
13 Thesis Organizationhelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip3
Chapter 2 Background Study5 21 High Speed Serial Link Interface helliphelliphelliphelliphelliphelliphelliphelliphellip5
211 PECL Interfacehelliphelliphelliphelliphellip7
212 CML Interfacehelliphelliphellip7
213 LVDS Interfacehelliphelliphellip8
22 AC Coupled Communicationhelliphelliphellip 10
221 AC Coupled Communicationhelliphelliphellip10
222 Typical pulse transmitterreceiver11
Chapter 3 On-Chip Pulse Signaling14 31 Pulse Signaling Schemehelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip15
32 On-Chip Pulse Signaling Analysishelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip 16
33 On-Chip Differential Transmission Linehelliphelliphelliphelliphelliphelliphelliphelliphellip 18
331 Mode of Transmission Linehelliphellip helliphelliphelliphelliphelliphelliphelliphelliphellip 18
332 Geometry Analysis of Transmission Linehelliphelliphelliphelliphelliphellip20
Chapter 4 5Gbps Pulse Transmitter Receiver Circuit Design23 41 Pulse Transmitter Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip23
411 Voltage Mode Transmitter with De-emphasishelliphelliphelliphellip23
412 The Proposed De-emphasis Circuithelliphelliphelliphelliphelliphellip25
42 Pulse Receiver Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip28
421 Receiver End Terminationhelliphelliphelliphellip 29
List of Contents
v
422 Self-Bias and Equalization Circuithelliphelliphelliphellip 31
423 Inductive Peaking Amplifierhelliphelliphelliphellip 32
424 Non-clock Latch with Hysteresishelliphelliphellip35
Chapter 5 Experiment39 51 Simulation Setup39
52 Measurement Considerations41
53 Experiment45
Chapter 6 Conclusion47
Bibliography49
List of Tables
vi
List of Tables Table 21 Electrical characteristics of the CMLhelliphelliphelliphellip8
Table 22 Electrical characteristics of LVDShellip9
Table 31 Parasitic RLC of the on-chip transmission linehellip20
Table 51 Specification Tablehellip43
Table 52 Comparison of high speed data communicationhellip45
List of Figures
vii
List of Figures Fig 11 On-chip long distance interconnection2
Fig 12 Basic link components the transmitter the channel and the receiver3
Fig 21 A generalized model of a serial link6
Fig 22 High speed interfacehelliphelliphellip6
Fig 23 (a) output structure of PECL hellip 7
Fig 23 (b) input structure of PECLhellip 7
Fig 24 (a) output structure of CML hellip 8
Fig 24 (b) input structure of CMLhellip 8
Fig 25 (a) output structure of LVDS helliphellip 9
Fig 25 (b) input structure of LVDS hellip 9
Fig 26 (a) AC coupled communication helliphellip 10
Fig 26 (b) pulse waveform in channelhellip 10
Fig 27 Voltage mode transmitter 11
Fig 28 (a) pulse receiver 12
Fig 28 (b) equivalent circuit with enable on 12
Fig 29 Low swing pulse receiver 13 Fig 31 Pulse signaling scheme 15
Fig 32 (a) On-chip pulse signaling model17
Fig 32 (b) Transient response of the transmitterhelliphelliphellip 17
Fig 33 On-chip transmission line as a distributed RLC transmission line 19
Fig 34 (a) Micro-strip structure and the parasitic effect 20
Fig 34 (b) Cross section of the on-chip transmission line 20
Fig 35 Transmission line dimension vs characteristic impedancehellip 22
Fig 36 Transmission line dimension vs parasitic RtotalCtotalhelliphellip 22
Fig 41 Output pulse signal at transmitter with different termination resistance24
Fig 42 Voltage mode transmitter (a) without de-emphasis 25
Fig 42 (b) with de-emphasis 25
Fig 43 Coupled pulse signal with de-emphasis circuit 26
Fig 44 Pulse transmitter26
Fig 45 Pulse Transmitter design flowcharthellip 27
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
i
晶片內部 5-Gbs 低功率脈波訊號傳輸介面
研究生 方盈霖 指導教授 蘇朝琴 教授
國立交通大學電機與控制工程研究所
摘 要
本論文提出一個在晶片內部的脈波傳輸介面來達到 SOC 晶片內的長距離低
功率消耗的傳輸接收電路研究內容包括脈波傳輸端電路脈波接收端電路與晶
片內部的差動傳輸線傳輸端部分實現電容偶合的方式產生脈波訊號經由長距
離的差動傳輸線在接收端同樣利用電容偶合的方式將脈波訊號偶合回接收端電
路設計上首先經由利用較高的傳輸端端電組來將傳送出的脈波訊號振幅提
升並配合提出的傳輸端 de-emphasis 電路架構可將脈波的尾端消除以減少訊
號 ISI 效應接收端則是自偏壓電路將脈波訊號載回接收端共模準位經過放大
及電路將脈波訊號放大後由栓鎖閘將訊號轉回 NRZ 訊號整個傳輸電路在台積電
RF013μm 製程下傳輸 5Gbps 的亂數資料功率消耗在傳輸端 34mW接收端
32mW 總功率消耗 64mW傳輸距離 5mm
關鍵字 脈波傳輸電容偶合晶片內部傳輸高速傳輸電路
ii
5-Gbs Low Power On-Chip Pulse Signaling Interface
Student Ying Lin Fang Advisor Chau Chin Su
Department of Electrical and Control Engineering National Chiao Tung University
Abstract
In this thesis we propose an on-chip pulse signaling communication It can be
used for long distance and low power interconnection on SOC The pulse signaling
communication consists of a transmitter an on-chip transmission-line and a receiver
By increase the termination resistance at the near end we can increase the amplitude
of the transmitted pulse signal And then a de-emphasis circuit is employed to reduce
the ISI effect both in the transmitter and in the receiver A TSMC 013um RF process
was utilized in our design In the simulation result 5Gbps signal transmission can be
achieved through a 5mm-length differential interconnect The power consumption at
Tx and Rx are 32mW and 34mW respectively and the total power consumption is
66mW
Keyword AC coupled pulse signaling capacitive coupling on-chip
communication driver receiver de-emphasis equalization
iii
誌 謝
首先要感謝我的指導教授 蘇朝琴老師感謝老師將近五年時間內從大二基
礎電子學到研究所論文的栽培您費心指導我在實驗室內的所有學習到做研究應
有的精神與態度將陪我進入碩士後的人生旅途
感謝實驗室內煜輝鴻文鴻愷(丸子)仁乾孟昌盈杰等博士班學長
們的指導忠傑緯達(阿達)瑛佑冠羽(KU)柏成(CGU)俊銘(阿銘)順
閔宗諭匡良楙軒智琦冠宇(小冠)等歷屆碩班學長傳授所學汝敏慶
霖(教主)永祥(祥哥)英豪(小馬)議賢村鑫威翔(小潘潘)存遠勇志
(阿伯)季慧雅婷挺毅俊彥繁祥(孔哥)碩廷子俞等同學和學弟妹在
碩班日子的相處和學習依萍俊秀雅雯上容佳琪嘉琳(大姐)志憲(totoro)
等 918 之友在生活上的照顧謝謝你們陪我度過碩班的日子
此外也要由衷感謝蔡中庸老師給予了我兩年碩士生涯的所有助教機會這些
助教收入雖然不至優渥但也等同於一筆獎助學金足夠讓我在研究之餘不至於
太過擔心交大生活上所需的開銷蔡媽妳對教學的熱忱與不放棄任何學生的愛心
在這段期間深深撼動著我也讓我更懂得回饋與感恩
平日研究外的生活兩位家教小妹瑜和宜安妳們給了我很多八九年級生
新穎的想法西洋棋社的 zayinacmonkyy浩浩阿杰mediclightningfox
給了很多棋藝上的切磋電控 16 號家族的鎮源(ican)普暄虹君維祐永遠
忘不了你們給的畢業禮物和家族的溫情室友們宇軒(鯉魚)旻廷依寰(Hubert)
將近五年的朝夕相處感謝你們的包容和帶給我的快樂
最後要感謝老爸老媽爺爺奶奶老哥老妹及女友上容在這段期間內
無盡的鼓勵與支持你們對我付出的關心是永遠報答不完的僅將自己這段時間
的努力與收穫獻給我愛的你們
盈霖 20070904
List of Contents
iv
List of Contents Abstract helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
Acknowledgements helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
List of Contents helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip iv
List of Tables helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vi
List of Figureshelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vii
Chapter 1 Introduction 1 11 Motivationhelliphelliphelliphelliphelliphelliphelliphelliphellip1
12 Link Components 3
13 Thesis Organizationhelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip3
Chapter 2 Background Study5 21 High Speed Serial Link Interface helliphelliphelliphelliphelliphelliphelliphelliphellip5
211 PECL Interfacehelliphelliphelliphelliphellip7
212 CML Interfacehelliphelliphellip7
213 LVDS Interfacehelliphelliphellip8
22 AC Coupled Communicationhelliphelliphellip 10
221 AC Coupled Communicationhelliphelliphellip10
222 Typical pulse transmitterreceiver11
Chapter 3 On-Chip Pulse Signaling14 31 Pulse Signaling Schemehelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip15
32 On-Chip Pulse Signaling Analysishelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip 16
33 On-Chip Differential Transmission Linehelliphelliphelliphelliphelliphelliphelliphelliphellip 18
331 Mode of Transmission Linehelliphellip helliphelliphelliphelliphelliphelliphelliphelliphellip 18
332 Geometry Analysis of Transmission Linehelliphelliphelliphelliphelliphellip20
Chapter 4 5Gbps Pulse Transmitter Receiver Circuit Design23 41 Pulse Transmitter Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip23
411 Voltage Mode Transmitter with De-emphasishelliphelliphelliphellip23
412 The Proposed De-emphasis Circuithelliphelliphelliphelliphelliphellip25
42 Pulse Receiver Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip28
421 Receiver End Terminationhelliphelliphelliphellip 29
List of Contents
v
422 Self-Bias and Equalization Circuithelliphelliphelliphellip 31
423 Inductive Peaking Amplifierhelliphelliphelliphellip 32
424 Non-clock Latch with Hysteresishelliphelliphellip35
Chapter 5 Experiment39 51 Simulation Setup39
52 Measurement Considerations41
53 Experiment45
Chapter 6 Conclusion47
Bibliography49
List of Tables
vi
List of Tables Table 21 Electrical characteristics of the CMLhelliphelliphelliphellip8
Table 22 Electrical characteristics of LVDShellip9
Table 31 Parasitic RLC of the on-chip transmission linehellip20
Table 51 Specification Tablehellip43
Table 52 Comparison of high speed data communicationhellip45
List of Figures
vii
List of Figures Fig 11 On-chip long distance interconnection2
Fig 12 Basic link components the transmitter the channel and the receiver3
Fig 21 A generalized model of a serial link6
Fig 22 High speed interfacehelliphelliphellip6
Fig 23 (a) output structure of PECL hellip 7
Fig 23 (b) input structure of PECLhellip 7
Fig 24 (a) output structure of CML hellip 8
Fig 24 (b) input structure of CMLhellip 8
Fig 25 (a) output structure of LVDS helliphellip 9
Fig 25 (b) input structure of LVDS hellip 9
Fig 26 (a) AC coupled communication helliphellip 10
Fig 26 (b) pulse waveform in channelhellip 10
Fig 27 Voltage mode transmitter 11
Fig 28 (a) pulse receiver 12
Fig 28 (b) equivalent circuit with enable on 12
Fig 29 Low swing pulse receiver 13 Fig 31 Pulse signaling scheme 15
Fig 32 (a) On-chip pulse signaling model17
Fig 32 (b) Transient response of the transmitterhelliphelliphellip 17
Fig 33 On-chip transmission line as a distributed RLC transmission line 19
Fig 34 (a) Micro-strip structure and the parasitic effect 20
Fig 34 (b) Cross section of the on-chip transmission line 20
Fig 35 Transmission line dimension vs characteristic impedancehellip 22
Fig 36 Transmission line dimension vs parasitic RtotalCtotalhelliphellip 22
Fig 41 Output pulse signal at transmitter with different termination resistance24
Fig 42 Voltage mode transmitter (a) without de-emphasis 25
Fig 42 (b) with de-emphasis 25
Fig 43 Coupled pulse signal with de-emphasis circuit 26
Fig 44 Pulse transmitter26
Fig 45 Pulse Transmitter design flowcharthellip 27
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
ii
5-Gbs Low Power On-Chip Pulse Signaling Interface
Student Ying Lin Fang Advisor Chau Chin Su
Department of Electrical and Control Engineering National Chiao Tung University
Abstract
In this thesis we propose an on-chip pulse signaling communication It can be
used for long distance and low power interconnection on SOC The pulse signaling
communication consists of a transmitter an on-chip transmission-line and a receiver
By increase the termination resistance at the near end we can increase the amplitude
of the transmitted pulse signal And then a de-emphasis circuit is employed to reduce
the ISI effect both in the transmitter and in the receiver A TSMC 013um RF process
was utilized in our design In the simulation result 5Gbps signal transmission can be
achieved through a 5mm-length differential interconnect The power consumption at
Tx and Rx are 32mW and 34mW respectively and the total power consumption is
66mW
Keyword AC coupled pulse signaling capacitive coupling on-chip
communication driver receiver de-emphasis equalization
iii
誌 謝
首先要感謝我的指導教授 蘇朝琴老師感謝老師將近五年時間內從大二基
礎電子學到研究所論文的栽培您費心指導我在實驗室內的所有學習到做研究應
有的精神與態度將陪我進入碩士後的人生旅途
感謝實驗室內煜輝鴻文鴻愷(丸子)仁乾孟昌盈杰等博士班學長
們的指導忠傑緯達(阿達)瑛佑冠羽(KU)柏成(CGU)俊銘(阿銘)順
閔宗諭匡良楙軒智琦冠宇(小冠)等歷屆碩班學長傳授所學汝敏慶
霖(教主)永祥(祥哥)英豪(小馬)議賢村鑫威翔(小潘潘)存遠勇志
(阿伯)季慧雅婷挺毅俊彥繁祥(孔哥)碩廷子俞等同學和學弟妹在
碩班日子的相處和學習依萍俊秀雅雯上容佳琪嘉琳(大姐)志憲(totoro)
等 918 之友在生活上的照顧謝謝你們陪我度過碩班的日子
此外也要由衷感謝蔡中庸老師給予了我兩年碩士生涯的所有助教機會這些
助教收入雖然不至優渥但也等同於一筆獎助學金足夠讓我在研究之餘不至於
太過擔心交大生活上所需的開銷蔡媽妳對教學的熱忱與不放棄任何學生的愛心
在這段期間深深撼動著我也讓我更懂得回饋與感恩
平日研究外的生活兩位家教小妹瑜和宜安妳們給了我很多八九年級生
新穎的想法西洋棋社的 zayinacmonkyy浩浩阿杰mediclightningfox
給了很多棋藝上的切磋電控 16 號家族的鎮源(ican)普暄虹君維祐永遠
忘不了你們給的畢業禮物和家族的溫情室友們宇軒(鯉魚)旻廷依寰(Hubert)
將近五年的朝夕相處感謝你們的包容和帶給我的快樂
最後要感謝老爸老媽爺爺奶奶老哥老妹及女友上容在這段期間內
無盡的鼓勵與支持你們對我付出的關心是永遠報答不完的僅將自己這段時間
的努力與收穫獻給我愛的你們
盈霖 20070904
List of Contents
iv
List of Contents Abstract helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
Acknowledgements helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
List of Contents helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip iv
List of Tables helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vi
List of Figureshelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vii
Chapter 1 Introduction 1 11 Motivationhelliphelliphelliphelliphelliphelliphelliphelliphellip1
12 Link Components 3
13 Thesis Organizationhelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip3
Chapter 2 Background Study5 21 High Speed Serial Link Interface helliphelliphelliphelliphelliphelliphelliphelliphellip5
211 PECL Interfacehelliphelliphelliphelliphellip7
212 CML Interfacehelliphelliphellip7
213 LVDS Interfacehelliphelliphellip8
22 AC Coupled Communicationhelliphelliphellip 10
221 AC Coupled Communicationhelliphelliphellip10
222 Typical pulse transmitterreceiver11
Chapter 3 On-Chip Pulse Signaling14 31 Pulse Signaling Schemehelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip15
32 On-Chip Pulse Signaling Analysishelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip 16
33 On-Chip Differential Transmission Linehelliphelliphelliphelliphelliphelliphelliphelliphellip 18
331 Mode of Transmission Linehelliphellip helliphelliphelliphelliphelliphelliphelliphelliphellip 18
332 Geometry Analysis of Transmission Linehelliphelliphelliphelliphelliphellip20
Chapter 4 5Gbps Pulse Transmitter Receiver Circuit Design23 41 Pulse Transmitter Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip23
411 Voltage Mode Transmitter with De-emphasishelliphelliphelliphellip23
412 The Proposed De-emphasis Circuithelliphelliphelliphelliphelliphellip25
42 Pulse Receiver Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip28
421 Receiver End Terminationhelliphelliphelliphellip 29
List of Contents
v
422 Self-Bias and Equalization Circuithelliphelliphelliphellip 31
423 Inductive Peaking Amplifierhelliphelliphelliphellip 32
424 Non-clock Latch with Hysteresishelliphelliphellip35
Chapter 5 Experiment39 51 Simulation Setup39
52 Measurement Considerations41
53 Experiment45
Chapter 6 Conclusion47
Bibliography49
List of Tables
vi
List of Tables Table 21 Electrical characteristics of the CMLhelliphelliphelliphellip8
Table 22 Electrical characteristics of LVDShellip9
Table 31 Parasitic RLC of the on-chip transmission linehellip20
Table 51 Specification Tablehellip43
Table 52 Comparison of high speed data communicationhellip45
List of Figures
vii
List of Figures Fig 11 On-chip long distance interconnection2
Fig 12 Basic link components the transmitter the channel and the receiver3
Fig 21 A generalized model of a serial link6
Fig 22 High speed interfacehelliphelliphellip6
Fig 23 (a) output structure of PECL hellip 7
Fig 23 (b) input structure of PECLhellip 7
Fig 24 (a) output structure of CML hellip 8
Fig 24 (b) input structure of CMLhellip 8
Fig 25 (a) output structure of LVDS helliphellip 9
Fig 25 (b) input structure of LVDS hellip 9
Fig 26 (a) AC coupled communication helliphellip 10
Fig 26 (b) pulse waveform in channelhellip 10
Fig 27 Voltage mode transmitter 11
Fig 28 (a) pulse receiver 12
Fig 28 (b) equivalent circuit with enable on 12
Fig 29 Low swing pulse receiver 13 Fig 31 Pulse signaling scheme 15
Fig 32 (a) On-chip pulse signaling model17
Fig 32 (b) Transient response of the transmitterhelliphelliphellip 17
Fig 33 On-chip transmission line as a distributed RLC transmission line 19
Fig 34 (a) Micro-strip structure and the parasitic effect 20
Fig 34 (b) Cross section of the on-chip transmission line 20
Fig 35 Transmission line dimension vs characteristic impedancehellip 22
Fig 36 Transmission line dimension vs parasitic RtotalCtotalhelliphellip 22
Fig 41 Output pulse signal at transmitter with different termination resistance24
Fig 42 Voltage mode transmitter (a) without de-emphasis 25
Fig 42 (b) with de-emphasis 25
Fig 43 Coupled pulse signal with de-emphasis circuit 26
Fig 44 Pulse transmitter26
Fig 45 Pulse Transmitter design flowcharthellip 27
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
iii
誌 謝
首先要感謝我的指導教授 蘇朝琴老師感謝老師將近五年時間內從大二基
礎電子學到研究所論文的栽培您費心指導我在實驗室內的所有學習到做研究應
有的精神與態度將陪我進入碩士後的人生旅途
感謝實驗室內煜輝鴻文鴻愷(丸子)仁乾孟昌盈杰等博士班學長
們的指導忠傑緯達(阿達)瑛佑冠羽(KU)柏成(CGU)俊銘(阿銘)順
閔宗諭匡良楙軒智琦冠宇(小冠)等歷屆碩班學長傳授所學汝敏慶
霖(教主)永祥(祥哥)英豪(小馬)議賢村鑫威翔(小潘潘)存遠勇志
(阿伯)季慧雅婷挺毅俊彥繁祥(孔哥)碩廷子俞等同學和學弟妹在
碩班日子的相處和學習依萍俊秀雅雯上容佳琪嘉琳(大姐)志憲(totoro)
等 918 之友在生活上的照顧謝謝你們陪我度過碩班的日子
此外也要由衷感謝蔡中庸老師給予了我兩年碩士生涯的所有助教機會這些
助教收入雖然不至優渥但也等同於一筆獎助學金足夠讓我在研究之餘不至於
太過擔心交大生活上所需的開銷蔡媽妳對教學的熱忱與不放棄任何學生的愛心
在這段期間深深撼動著我也讓我更懂得回饋與感恩
平日研究外的生活兩位家教小妹瑜和宜安妳們給了我很多八九年級生
新穎的想法西洋棋社的 zayinacmonkyy浩浩阿杰mediclightningfox
給了很多棋藝上的切磋電控 16 號家族的鎮源(ican)普暄虹君維祐永遠
忘不了你們給的畢業禮物和家族的溫情室友們宇軒(鯉魚)旻廷依寰(Hubert)
將近五年的朝夕相處感謝你們的包容和帶給我的快樂
最後要感謝老爸老媽爺爺奶奶老哥老妹及女友上容在這段期間內
無盡的鼓勵與支持你們對我付出的關心是永遠報答不完的僅將自己這段時間
的努力與收穫獻給我愛的你們
盈霖 20070904
List of Contents
iv
List of Contents Abstract helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
Acknowledgements helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
List of Contents helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip iv
List of Tables helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vi
List of Figureshelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vii
Chapter 1 Introduction 1 11 Motivationhelliphelliphelliphelliphelliphelliphelliphelliphellip1
12 Link Components 3
13 Thesis Organizationhelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip3
Chapter 2 Background Study5 21 High Speed Serial Link Interface helliphelliphelliphelliphelliphelliphelliphelliphellip5
211 PECL Interfacehelliphelliphelliphelliphellip7
212 CML Interfacehelliphelliphellip7
213 LVDS Interfacehelliphelliphellip8
22 AC Coupled Communicationhelliphelliphellip 10
221 AC Coupled Communicationhelliphelliphellip10
222 Typical pulse transmitterreceiver11
Chapter 3 On-Chip Pulse Signaling14 31 Pulse Signaling Schemehelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip15
32 On-Chip Pulse Signaling Analysishelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip 16
33 On-Chip Differential Transmission Linehelliphelliphelliphelliphelliphelliphelliphelliphellip 18
331 Mode of Transmission Linehelliphellip helliphelliphelliphelliphelliphelliphelliphelliphellip 18
332 Geometry Analysis of Transmission Linehelliphelliphelliphelliphelliphellip20
Chapter 4 5Gbps Pulse Transmitter Receiver Circuit Design23 41 Pulse Transmitter Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip23
411 Voltage Mode Transmitter with De-emphasishelliphelliphelliphellip23
412 The Proposed De-emphasis Circuithelliphelliphelliphelliphelliphellip25
42 Pulse Receiver Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip28
421 Receiver End Terminationhelliphelliphelliphellip 29
List of Contents
v
422 Self-Bias and Equalization Circuithelliphelliphelliphellip 31
423 Inductive Peaking Amplifierhelliphelliphelliphellip 32
424 Non-clock Latch with Hysteresishelliphelliphellip35
Chapter 5 Experiment39 51 Simulation Setup39
52 Measurement Considerations41
53 Experiment45
Chapter 6 Conclusion47
Bibliography49
List of Tables
vi
List of Tables Table 21 Electrical characteristics of the CMLhelliphelliphelliphellip8
Table 22 Electrical characteristics of LVDShellip9
Table 31 Parasitic RLC of the on-chip transmission linehellip20
Table 51 Specification Tablehellip43
Table 52 Comparison of high speed data communicationhellip45
List of Figures
vii
List of Figures Fig 11 On-chip long distance interconnection2
Fig 12 Basic link components the transmitter the channel and the receiver3
Fig 21 A generalized model of a serial link6
Fig 22 High speed interfacehelliphelliphellip6
Fig 23 (a) output structure of PECL hellip 7
Fig 23 (b) input structure of PECLhellip 7
Fig 24 (a) output structure of CML hellip 8
Fig 24 (b) input structure of CMLhellip 8
Fig 25 (a) output structure of LVDS helliphellip 9
Fig 25 (b) input structure of LVDS hellip 9
Fig 26 (a) AC coupled communication helliphellip 10
Fig 26 (b) pulse waveform in channelhellip 10
Fig 27 Voltage mode transmitter 11
Fig 28 (a) pulse receiver 12
Fig 28 (b) equivalent circuit with enable on 12
Fig 29 Low swing pulse receiver 13 Fig 31 Pulse signaling scheme 15
Fig 32 (a) On-chip pulse signaling model17
Fig 32 (b) Transient response of the transmitterhelliphelliphellip 17
Fig 33 On-chip transmission line as a distributed RLC transmission line 19
Fig 34 (a) Micro-strip structure and the parasitic effect 20
Fig 34 (b) Cross section of the on-chip transmission line 20
Fig 35 Transmission line dimension vs characteristic impedancehellip 22
Fig 36 Transmission line dimension vs parasitic RtotalCtotalhelliphellip 22
Fig 41 Output pulse signal at transmitter with different termination resistance24
Fig 42 Voltage mode transmitter (a) without de-emphasis 25
Fig 42 (b) with de-emphasis 25
Fig 43 Coupled pulse signal with de-emphasis circuit 26
Fig 44 Pulse transmitter26
Fig 45 Pulse Transmitter design flowcharthellip 27
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
List of Contents
iv
List of Contents Abstract helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
Acknowledgements helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellipii
List of Contents helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip iv
List of Tables helliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vi
List of Figureshelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip vii
Chapter 1 Introduction 1 11 Motivationhelliphelliphelliphelliphelliphelliphelliphelliphellip1
12 Link Components 3
13 Thesis Organizationhelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip3
Chapter 2 Background Study5 21 High Speed Serial Link Interface helliphelliphelliphelliphelliphelliphelliphelliphellip5
211 PECL Interfacehelliphelliphelliphelliphellip7
212 CML Interfacehelliphelliphellip7
213 LVDS Interfacehelliphelliphellip8
22 AC Coupled Communicationhelliphelliphellip 10
221 AC Coupled Communicationhelliphelliphellip10
222 Typical pulse transmitterreceiver11
Chapter 3 On-Chip Pulse Signaling14 31 Pulse Signaling Schemehelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip15
32 On-Chip Pulse Signaling Analysishelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip 16
33 On-Chip Differential Transmission Linehelliphelliphelliphelliphelliphelliphelliphelliphellip 18
331 Mode of Transmission Linehelliphellip helliphelliphelliphelliphelliphelliphelliphelliphellip 18
332 Geometry Analysis of Transmission Linehelliphelliphelliphelliphelliphellip20
Chapter 4 5Gbps Pulse Transmitter Receiver Circuit Design23 41 Pulse Transmitter Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip23
411 Voltage Mode Transmitter with De-emphasishelliphelliphelliphellip23
412 The Proposed De-emphasis Circuithelliphelliphelliphelliphelliphellip25
42 Pulse Receiver Circuithelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphelliphellip28
421 Receiver End Terminationhelliphelliphelliphellip 29
List of Contents
v
422 Self-Bias and Equalization Circuithelliphelliphelliphellip 31
423 Inductive Peaking Amplifierhelliphelliphelliphellip 32
424 Non-clock Latch with Hysteresishelliphelliphellip35
Chapter 5 Experiment39 51 Simulation Setup39
52 Measurement Considerations41
53 Experiment45
Chapter 6 Conclusion47
Bibliography49
List of Tables
vi
List of Tables Table 21 Electrical characteristics of the CMLhelliphelliphelliphellip8
Table 22 Electrical characteristics of LVDShellip9
Table 31 Parasitic RLC of the on-chip transmission linehellip20
Table 51 Specification Tablehellip43
Table 52 Comparison of high speed data communicationhellip45
List of Figures
vii
List of Figures Fig 11 On-chip long distance interconnection2
Fig 12 Basic link components the transmitter the channel and the receiver3
Fig 21 A generalized model of a serial link6
Fig 22 High speed interfacehelliphelliphellip6
Fig 23 (a) output structure of PECL hellip 7
Fig 23 (b) input structure of PECLhellip 7
Fig 24 (a) output structure of CML hellip 8
Fig 24 (b) input structure of CMLhellip 8
Fig 25 (a) output structure of LVDS helliphellip 9
Fig 25 (b) input structure of LVDS hellip 9
Fig 26 (a) AC coupled communication helliphellip 10
Fig 26 (b) pulse waveform in channelhellip 10
Fig 27 Voltage mode transmitter 11
Fig 28 (a) pulse receiver 12
Fig 28 (b) equivalent circuit with enable on 12
Fig 29 Low swing pulse receiver 13 Fig 31 Pulse signaling scheme 15
Fig 32 (a) On-chip pulse signaling model17
Fig 32 (b) Transient response of the transmitterhelliphelliphellip 17
Fig 33 On-chip transmission line as a distributed RLC transmission line 19
Fig 34 (a) Micro-strip structure and the parasitic effect 20
Fig 34 (b) Cross section of the on-chip transmission line 20
Fig 35 Transmission line dimension vs characteristic impedancehellip 22
Fig 36 Transmission line dimension vs parasitic RtotalCtotalhelliphellip 22
Fig 41 Output pulse signal at transmitter with different termination resistance24
Fig 42 Voltage mode transmitter (a) without de-emphasis 25
Fig 42 (b) with de-emphasis 25
Fig 43 Coupled pulse signal with de-emphasis circuit 26
Fig 44 Pulse transmitter26
Fig 45 Pulse Transmitter design flowcharthellip 27
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
List of Contents
v
422 Self-Bias and Equalization Circuithelliphelliphelliphellip 31
423 Inductive Peaking Amplifierhelliphelliphelliphellip 32
424 Non-clock Latch with Hysteresishelliphelliphellip35
Chapter 5 Experiment39 51 Simulation Setup39
52 Measurement Considerations41
53 Experiment45
Chapter 6 Conclusion47
Bibliography49
List of Tables
vi
List of Tables Table 21 Electrical characteristics of the CMLhelliphelliphelliphellip8
Table 22 Electrical characteristics of LVDShellip9
Table 31 Parasitic RLC of the on-chip transmission linehellip20
Table 51 Specification Tablehellip43
Table 52 Comparison of high speed data communicationhellip45
List of Figures
vii
List of Figures Fig 11 On-chip long distance interconnection2
Fig 12 Basic link components the transmitter the channel and the receiver3
Fig 21 A generalized model of a serial link6
Fig 22 High speed interfacehelliphelliphellip6
Fig 23 (a) output structure of PECL hellip 7
Fig 23 (b) input structure of PECLhellip 7
Fig 24 (a) output structure of CML hellip 8
Fig 24 (b) input structure of CMLhellip 8
Fig 25 (a) output structure of LVDS helliphellip 9
Fig 25 (b) input structure of LVDS hellip 9
Fig 26 (a) AC coupled communication helliphellip 10
Fig 26 (b) pulse waveform in channelhellip 10
Fig 27 Voltage mode transmitter 11
Fig 28 (a) pulse receiver 12
Fig 28 (b) equivalent circuit with enable on 12
Fig 29 Low swing pulse receiver 13 Fig 31 Pulse signaling scheme 15
Fig 32 (a) On-chip pulse signaling model17
Fig 32 (b) Transient response of the transmitterhelliphelliphellip 17
Fig 33 On-chip transmission line as a distributed RLC transmission line 19
Fig 34 (a) Micro-strip structure and the parasitic effect 20
Fig 34 (b) Cross section of the on-chip transmission line 20
Fig 35 Transmission line dimension vs characteristic impedancehellip 22
Fig 36 Transmission line dimension vs parasitic RtotalCtotalhelliphellip 22
Fig 41 Output pulse signal at transmitter with different termination resistance24
Fig 42 Voltage mode transmitter (a) without de-emphasis 25
Fig 42 (b) with de-emphasis 25
Fig 43 Coupled pulse signal with de-emphasis circuit 26
Fig 44 Pulse transmitter26
Fig 45 Pulse Transmitter design flowcharthellip 27
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
List of Tables
vi
List of Tables Table 21 Electrical characteristics of the CMLhelliphelliphelliphellip8
Table 22 Electrical characteristics of LVDShellip9
Table 31 Parasitic RLC of the on-chip transmission linehellip20
Table 51 Specification Tablehellip43
Table 52 Comparison of high speed data communicationhellip45
List of Figures
vii
List of Figures Fig 11 On-chip long distance interconnection2
Fig 12 Basic link components the transmitter the channel and the receiver3
Fig 21 A generalized model of a serial link6
Fig 22 High speed interfacehelliphelliphellip6
Fig 23 (a) output structure of PECL hellip 7
Fig 23 (b) input structure of PECLhellip 7
Fig 24 (a) output structure of CML hellip 8
Fig 24 (b) input structure of CMLhellip 8
Fig 25 (a) output structure of LVDS helliphellip 9
Fig 25 (b) input structure of LVDS hellip 9
Fig 26 (a) AC coupled communication helliphellip 10
Fig 26 (b) pulse waveform in channelhellip 10
Fig 27 Voltage mode transmitter 11
Fig 28 (a) pulse receiver 12
Fig 28 (b) equivalent circuit with enable on 12
Fig 29 Low swing pulse receiver 13 Fig 31 Pulse signaling scheme 15
Fig 32 (a) On-chip pulse signaling model17
Fig 32 (b) Transient response of the transmitterhelliphelliphellip 17
Fig 33 On-chip transmission line as a distributed RLC transmission line 19
Fig 34 (a) Micro-strip structure and the parasitic effect 20
Fig 34 (b) Cross section of the on-chip transmission line 20
Fig 35 Transmission line dimension vs characteristic impedancehellip 22
Fig 36 Transmission line dimension vs parasitic RtotalCtotalhelliphellip 22
Fig 41 Output pulse signal at transmitter with different termination resistance24
Fig 42 Voltage mode transmitter (a) without de-emphasis 25
Fig 42 (b) with de-emphasis 25
Fig 43 Coupled pulse signal with de-emphasis circuit 26
Fig 44 Pulse transmitter26
Fig 45 Pulse Transmitter design flowcharthellip 27
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
List of Figures
vii
List of Figures Fig 11 On-chip long distance interconnection2
Fig 12 Basic link components the transmitter the channel and the receiver3
Fig 21 A generalized model of a serial link6
Fig 22 High speed interfacehelliphelliphellip6
Fig 23 (a) output structure of PECL hellip 7
Fig 23 (b) input structure of PECLhellip 7
Fig 24 (a) output structure of CML hellip 8
Fig 24 (b) input structure of CMLhellip 8
Fig 25 (a) output structure of LVDS helliphellip 9
Fig 25 (b) input structure of LVDS hellip 9
Fig 26 (a) AC coupled communication helliphellip 10
Fig 26 (b) pulse waveform in channelhellip 10
Fig 27 Voltage mode transmitter 11
Fig 28 (a) pulse receiver 12
Fig 28 (b) equivalent circuit with enable on 12
Fig 29 Low swing pulse receiver 13 Fig 31 Pulse signaling scheme 15
Fig 32 (a) On-chip pulse signaling model17
Fig 32 (b) Transient response of the transmitterhelliphelliphellip 17
Fig 33 On-chip transmission line as a distributed RLC transmission line 19
Fig 34 (a) Micro-strip structure and the parasitic effect 20
Fig 34 (b) Cross section of the on-chip transmission line 20
Fig 35 Transmission line dimension vs characteristic impedancehellip 22
Fig 36 Transmission line dimension vs parasitic RtotalCtotalhelliphellip 22
Fig 41 Output pulse signal at transmitter with different termination resistance24
Fig 42 Voltage mode transmitter (a) without de-emphasis 25
Fig 42 (b) with de-emphasis 25
Fig 43 Coupled pulse signal with de-emphasis circuit 26
Fig 44 Pulse transmitter26
Fig 45 Pulse Transmitter design flowcharthellip 27
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
List of Figures
viii
Fig 46 Simulation of coupled pulse signal with de-emphasishelliphellip 28
Fig 47 Receiver end circuit blockhelliphelliphellip 29
Fig 48 Receiver end termination scheme helliphellip 30
Fig 49 Impedance looking from the transmission line to the receiver 31
Fig 410 Receiver end Self-bias and equalization circuithelliphelliphellip 32
Fig 411 Pulse waveform after equalization circuit 32
Fig 412 Receiver end Inductive peaking amplifierhelliphelliphellip 33
Fig 413 (a) Inductive peaking cell 34
Fig 413 (b) Pulse waveform after inductive peaking amplify 34
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier 34
Fig 415 Receiver end Non-clock latch with hysteresis 35
Fig 416 Non-clock latch with hysteresis circuit 36
Fig 417 Simulation of non-clock latch with hysteresis circuit 37
Fig 418 Jitter from the receiver 37
Fig 51 System Architecture 40
Fig 52 Common source circuit connects to pad for output signal measurement 41
Fig 53 System simulation results 42
Fig 54 Corner case simulation results 43
Fig 55 Layout 44
Fig 56 Measurement instruments 46
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 1 Introduction
1
Chapter 1 Introduction
11 Motivation
With the COMS technology grows in recent years there has been a great interest
in SOC design It results in large chip size and high power consumption In
conventional chip design the overall system efficiency depends on the performance
of individual module However with the distance between modules increases the
module-to-module data communication bandwidth becomes an important issue of
SOC design Because the long distance communication not only decays the signal
amplitude but also requires high power consumption to transmit the signal The long
distance on-chip transmission line has a large parasitic resistance and the resistance
has frequency dependence due to the skin effect The large parasitic resistance and
parasitic capacitance make the signal decay greatly Furthermore the power
consumption of an electrical signal in SOC is governed by two components The first
component is due to leakage current and DC path between VDD and GND known as
the static power and the second one is due to switching transient current and short
circuit transient current known as the dynamic power [1] In conventional high speed
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 1 Introduction
2
link design current mode logic (CML) Positive emitter coupled logic (PECL) low
voltage differential signaling (LVDS) are mostly used They all require a current
source to drive the communication data and the current source will increase the power
consumption especially when high speed data is transmitted
In this thesis we will explore the pulse signaling to improve the power efficiency
of long distance on-chip interconnection The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal and
that treat the DC component as redundant The pulse signaling transmits data by using
AC coupled method and that consumes only the dynamic power Therefore the pulse
signaling would reduce the power consumption of the on-chip data communication
Fig 11 On-chip long distance interconnection
Fig 11 shows the high speed data communication in the SOC design The
distance from driver end to receiver end is in the range of 1000um to 5000um The
long distance transmission line goes through the space between modules for saving
chip area In this way a long distance transmission of small area overhead is required
Besides there are also two targets to design this pulse signaling The first is the
structure must be simple and easy for implementation The second is the circuit
should operate at high speed and consume low power for SOC application These
design methodology and much more circuit details will be discussed in the following
chapters
TX
RX
Module 1
Module 2
Module 3
Module 4
SOC
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 1 Introduction
3
12 Link Components
A typical high speed link is composed of three components a transmitter a
transmission line and a receiver Fig 12 illustrates these components At near end
the transmitter converts the digital data into an analog signal stream and sends that to
the transmission line During the signal propagates to the receiver the transmission
line attenuates the signal and introduces noise In order to achieve high data rate and
reduce the distortion a transmission line with low attenuation at high frequency is
required At far end the signal comes into the receiver The receiver must be able to
resolve small input and then amplify the incoming signal After that the receiver
converts the signal back into binary data
Fig 12 Basic link components the transmitter the channel and the receiver
In this thesis we use a pulse form signal stream which known as return-to-zero
(RZ) signaling We also use a differential structure of transmission line to reduce the
common mode disturbance At the far end a receiver with a built-in latch is able to
convert the RZ pulse back into the non-return-to-zero (NRZ) signal Then the
converted NRZ data can be used for further receiver end usage
13 Thesis Organization
This thesis describes a voltage mode pulse transmitter a pulse receiver and an
on-chip transmission line In chapter 2 we study the background of the high-speed
serial link interface and the AC coupled communication Some typical structures are
also introduced in this chapter In chapter 3 the on-chip pulse signaling is present An
Transmitter Receiver
Timing Recovery
Data out Data in
Transmission Line
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 1 Introduction
4
analysis of on chip pulse signaling and an on-chip differential transmission are
discussed Chapter 4 shows the design of the pulse transmitter and the pulse receiver
The experimental results and conclusions are addressed in the last two chapters The
results are also compared with some prior works
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 2 Background Study
5
Chapter 2 Background Study
System on chip (SOC) is the trend of todayrsquos IC design and the high speed
interface [2] [3] between chip modules is an important design issue In the electrical
industry positive-referenced emitter-coupled logic (PECL) current mode logic
(CML) and low voltage differential signaling (LVDS) are the structures commonly
used in high speed link interface design However a current source is required to
produce the IR drop that makes the signal swing The current source not only
consumes the dynamic power but also the static power In order to save more power
AC coupled method is introduced because it only consumes the dynamic power by
coupling pulse mode data Besides with the growing of chip size on-chip
interconnect is also getting more attention as on-chip interconnect is becoming a
speed power and reliability bottleneck for the system In this chapter we will briefly
overview the background of high-speed serial link interface and AC coupled
communication Of course some typical structures are also introduced in this chapter
21 High Speed Link Interface
High speed links for digital systems are widely used in recent years In the
computer systems the links usually appear in the processor to memory interfaces and
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 2 Background Study
6
in the multiprocessor interconnections In the computer network the links are used in
the long-haul interconnect between far apart systems and in the backplane
interconnect within a switch or a router Fig 21 is a generalized structure of a high
speed serial link [4] It consists of a transmitter a transmission line and a receiver
The transmitter side serializes the parallel data and delivers the synchronized data into
the transmission line Timing information is embedded in this serial data which is
sent over a single interconnect The receiver side receives the serial data and recovers
its timing Then the serial data returns to the parallel one
Fig 21 A generalized model of a serial link
As the demand for high-speed data transmission grows the interface between
high speed integrated circuits becomes critical in achieving high performance low
power and good noise immunity Fig22 shows the typical structure of a high speed
interface The interface usually appears in the chip-to-chip high speed data
communication [5] Three commonly used interfaces are PECL CML and LVDS
The signal swing provided by the interface depends on the IR drop on the termination
resistance and that decides the power consumption The signal swing of most designs
is in the range of 300mV to 500mV
Fig 22 High speed interface
Serial to Parallel
Receiver N
N
Phase Lock Loop
Clock Recovery
Transmitter Parallel to
Serial
Transmission Line
IR~300-500mV
Iout
D
D
ZL
RT-T RT-R
ZL
I
Chip1
Channel
Chip2
Gnd
Vdd
Gnd
Vdd
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 2 Background Study
7
211 PECL Interface PECL [6] is developed by Motorola and originates from ECL but uses a positive
power supply The PECL is suitable for high-speed serial and parallel data links for its
relatively small swing Fig 23 (a) shows the PECL output structure It consists of a
differential pair that drives source followers The output source follower increases the
switching speeds by always having a DC current flowing through it The termination
of PECL output is typically 50Ω that results in a DC current and makes Vop and Von to
be (Vdd-Vov) The PECL has low output impedance to provide good driving capability
Fig23 (b) shows the input structure of PECL It consists of a differential current
switch with high input impedance The common mode voltage is around (Vdd-IR) to
provide enough operating headroom Besides decoupling between the power rails is
required to reduce the power supply noise
Fig 23 (a) output structure of PECL (b) input structure of PECL
212 CML Interface CML [6] [7] is a simple structure for high speed interface Fig 24 (a) shows the
output structure of the typical CML which consists of a differential pair with 50Ω
drain resistors The signal swing is supplied by switching the current in a
common-source differential pair Assuming the current source is 8mA typically with
the CML output connected a 50Ω pull-up to Vdd and then the single-ended CML
output voltage swing is from Vdd to (Vddminus04V) That results in the CML having a
differential output swing of 800mVpp and a common mode voltage of (Vddndash04V) Fig
Von Vop D D
(a) (b)
Vop
Von
D
RI
D
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 2 Background Study
8
24 (b) shows the CML input structure which has 50Ω input impedance for easy
termination The input transistors are source followers that drive a differential-pair
amplifier
Fig 24 (a) output structure of CML (b) input structure of CML
In conclusion the signal swing provided by the CML output is small and that
results in low power consumption Besides the termination minimizes the back
reflection thus reducing high-frequency interference Table 21 [6] summarizes the
electrical characteristics of a typical CML
Table 21 Electrical characteristics of the CML [6]
Parameter min typ max Unit
Differential Output Voltage 640 800 1000 mVpp
Output Common Mode Voltage Vdd-02 V
Single-Ended Input Voltage Range Vddndash06 Vdd+02 V
Differential Input Voltage Swing 400 1200 mVpp
213 LVDS Interface LVDS [6] [8] has several advantages that make it widely used in the telecom and
network technologies The low voltage swing leads to low power consumption and
makes it attractive in most high-speed interface Fig 25 (a) shows the output
structure of LVDS The differential output impedance is typically 100Ω A current
VonVop
50Ω50Ω
Vdd
VonVop
50Ω 50Ω
Vip
Vin
Vdd
(a) (b)
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 2 Background Study
9
steering transmitter provides Ib (at most 4mA) current flowing through a 100Ω
termination resistor The signal voltage level is low which allows low supply voltage
such as 25V The input voltage range of the LVDS can be form 0V to 24V and the
differential output voltage can be 400mV These allow the input common mode
voltage in the range from 02V to 22V Fig 25 (b) shows the LVDS input structure
It has on-chip 100Ω differential impedance between Vip and Vin A level-shifter sets
the common-mode voltage to a constant value at the input And a Schmitt trigger with
hysteresis range relative to the input threshold provides a wide common-mode range
The signal is then send into the following differential amplify for receiver usage
Table 22 [6] summarizes the LVDS input and output electrical characteristic
Fig 25 (a) output structure of LVDS (b) input structure of LVDS
Table 22 Electrical characteristics of LVDS [6]
Parameter Symbol Cond min typ max UnitOutput high voltage VOH 1475 V Output low voltage VOL 0925 V Differential Output Voltage |Vod| 250 400 mV Differential Output Variation |V od| 25 mV Output Offset Voltage Vos 1125 1275 mV Output Offset Variation |V os| 25 mV Differential Output Impedance 80 120 Ω
Output Current Short together 12 mA
Output Current Short to Gnd 40 mA
Input Voltage Range Vi 0 24 V
Vo
Vip Vin
(a) (b)
Vocm Von Vop
Ib
Ib
Vin
Vip Vin
Vip
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 2 Background Study
10
Differential Input Voltage |Vid| 100 mV
Input Common-Mode Current Input Vos=12V 350 uA
Hysteresis Threshold 70 mV Differential Input Impedance Rin 85 100 115 Ω
22 AC Coupled Communication
221 AC coupled communication The AC coupled interface [9] [10] [11] uses pulse signal to transmit data It has
the advantage of less power consumption The AC component actually carries all the
information of a digital signal and thus an AC coupled voltage mode driver can be
used to improve the power consumption because only the dynamic power is presented
Coupling capacitor acts like a high-pass filter that passes the AC component of the
data to the data bus
(a)
(b)
Fig 26 (a) AC coupled communication [12] (b) pulse waveform in transmission line
GND
VDD300mV
GND
VDD 1ns
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 2 Background Study
11
Fig 26 (a) shows the AC coupled communication [12] The transmitter (in chip
1) and the receiver (in chip 4) are connected to the data bus through an on-chip
coupling capacitor Cc at pad Both end of the data bus are terminated by the
impedance matching resistors Z0 with termination voltage Vterm Fig 26 (b) indicates
the waveform of the pulse communication The input full swing data sequence is
coupled to pulse form by the capacitor at the transmitter output The amplitude of the
pulse signal is roughly 300mV at the data rate of 1Gbps The receiver obtains the
pulse mode data through the coupling capacitor and then amplifies the decayed signal
to full swing
222 Typical pulse transmitter receiver Pulse signaling [15] [16] typically uses a voltage-mode transmitter [17] as shown
in Fig 28 The transmitter output is connect to a coupling capacitor and then to the
transmission line Return impedance matching at the transmitter output is provided by
the termination resistor The voltage mode transmitter provides a full swing as well as
high edge rate output to drive the transmission line Without DC current consumption
the voltage mode transmitter consumes less power than a current mode transmitter
Fig 27 voltage mode transmitter
Fig 29 (a) shows the pulse receiver proposed by Jongsun Kim in 2005 [12] The
enable function is design for saving power if the circuit is in the off mode The
equivalent circuit with enable on is shown in Fig 29 (b) The cross-couple structure
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 2 Background Study
12
not only senses the pulse signal but also provides the latch function to transform the
pulse signal into NRZ data
(a)
(b)
Fig 28 (a) pulse receiver (b) equivalent circuit with enable on
Fig 210 shows another low swing pulse receiver proposed by Lei Luo in 2006
[14] The diode connected feedback M1~M4 limits the swing at the out of inverter
Meanwhile M5 also M6 provide a weak but constant feedback to stabilize the bias
voltage making it less sensitive to the input pulses Source couple logic M7~M9
further amplifies the pulse and cross-coupled M11 amp M12 serves as a clock free latch
to recover NRZ data In addition a clamping device M10 limits the swing of long 1rsquos
or 0rsquos and enable latch operation for short pulse which improves the latch bandwidth
The structure is able to receive pulse of the swing as small as 120mVpp
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 2 Background Study
13
Fig 29 low swing pulse receiver
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 3 On-Chip Pulse Signaling
14
Chapter 3 On-Chip Pulse Signaling
This chapter introduces the on-chip pulse signaling It includes the pulse
signaling mathematic model and the geometry of the on-chip differential transmission
line On-chip transmission line [18] [19] has a large parasitic resistance The
resistance is frequency dependent due to the skin effect The large parasitic resistance
and parasitic capacitance make the signal decay greatly [20] Thus a wide line is
required for long distance and high-speed signal transmission However the wider
line and space between lines require more layout area That will increase the costs of
the SOC That is why a low cost and large bandwidth transmission line is desired
Moreover the characteristic impedance of transmission line is also discussed in this
chapter Theoretically impedance matching among the transmitter end the receiver
end and the transmission line is required That can prevent the signal from reflecting
in the transmission line In this thesis a single end termination is implemented The
single termination method can guarantee the signal reflects at most once The
designed termination impedance is 75Ω for less attenuation The width space and
length of the transmission line are 23um 15um and 5000um respectively The
transmission is implemented with Metal 6 and Metal 5 of TSMC RF013um
technology
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 3 On-Chip Pulse Signaling
15
31 Pulse Signaling Scheme
Fig 31 shows the on-chip pulse signaling scheme The transmitter and the
receiver are coupled to the on-chip transmission line through the coupling capacitor
Cctx and Ccrx Transmitter and receiver ends of the data bus are terminated by the
impedance Ztx and Zrx respectively with the terminated voltage Vterm Furthermore the
transmission line acts like the distributed RLC that decays the transmitted data The
coupling capacitor makes it a high-pass filter that transmits the transient part of the
input data The DC component is blocked The pulse signaling method is base on the
fact that the AC component actually carries all the information of a digital signal The
DC component is treated as redundant In this way the pulse data acts as return to
zero signaling (RZ) In contrast to non-return to zero (NRZ) signaling pulse signaling
has been used to reduce the power consumption by only dissipating the dynamic
power at transient time
Fig 31 Pulse signaling scheme
In Fig 31 full swing data is fed into the transmitter input The transmitted data
in the transmission line is a pulse coupled by the coupling capacitor at the transmitter
output After long distance of transmission the received pulse is coupled to the
On-Chip Transmission Line Cascaded Driver Hysteresis Receivers
Full Swing
Coupling Pulse Coupling
Pulse Full
Swing
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 3 On-Chip Pulse Signaling
16
hysteresis receiver by the coupling capacitor as well The receiver end biases the pulse
to the DC common mode voltage of the hysteresis receiver After that the hysteresis
receiver transfers the RZ pulse data to NRZ full swing data
32 On-chip pulse signaling analysis
Fig 32 (a) defines the on-chip pulse signaling model and Fig 32 (b) is the
corresponding transient response of the transmitter Let Zt be the characteristic
impedance of the transmission line Ztx and Zrx the termination resistances at the
transmitter end and receiver end Cctx and Crtx the coupling capacitors Cd and Cg the
parasitic capacitance at transmitter output and receiver input respectively The transfer
function from transmitter output (A) to near end transmission line (B) is
tx tctx tx t
AB
tx t tx td ctx ctx
1 +(Z Z )sC (Z Z )H (s)= 1 1 1+ +(Z Z ) +(Z Z )
sC sC sC
times
tx t ctx d
tx t ctx d ctx d
(Z Z )C C s=(Z Z )C C s+(C +C )
(31)
Equation (31) shows the high-pass characteristic and that HAB(s) is dominated by
tx t ctx d(Z Z )C C in high frequency In other words large coupling capacitor and
termination resistance are better for transferring the pulse signal Moreover the
transfer function from the near end (B) to the far end C) is
crx grx
ch crx gBC
crx gch ch rx
ch crx g
C +C1( Z )sC sC C
H (s)= C +C1R + sL +( Z )sC sC C
Because Ccrx and Cg are typically in the order of femto-farad (Ccrx+Cg)sCcrxCg ltlt1
and the transfer function becomes
chrx
BC
ch ch
chrx
1( )1sC +ZH (s) 1R + sL +( )1sC +
Z
asymp (32)
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 3 On-Chip Pulse Signaling
17
Equation (32) shows the channel bandwidth is roughly 1RchCch The parasitic
resistance and capacitance of the transmission line limits the channel bandwidth
Furthermore the transfer function from the far end (C) to the receiver input (D) is
gCD
crx g
1sC
H (s)= 1 1+sC sC
crx
crx g
C=C +C
2 ch ch
ch ch ch chrx rx
1= L RC L s +(R C + )s+( +1)Z Z
(33)
Equation (33) shows that the receiver pulse is proportional to the value of coupling
capacitor (Ccrx )
Fig 32 (a) On-chip pulse signaling model (b) Transient response of the transmitter
Node A
Node B
Vp
Vterm
Vdd
tr td Pulse Width≒ tr + td
(a)
A
B C
D
Z
(b)
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 3 On-Chip Pulse Signaling
18
The step response is important in analyzing the pulse signaling A step input
voltage at node A results in a transient on node C Transforming a square wave into a
short triangular pulse wave on the transmission line The amplitude (Vp) becomes
p eff cV = R I Atx t c
dV= (Z Z )Cdt
(34)
and the voltage value at node B is
eff eff
t-R C
B p termV (t)=V e +V t-t
p term=V e +V (35)
Where eff tx t eff ctx d eff ctx d ctx dR = (Z Z ) 1C = 1C +1C C = C C (C +C ) and eff efft = R C
Equation (34) shows that the amplitude of the pulse is proportional to three parts the
equivalent value of the termination resistance parallel to the characteristic resistance
coupling capacitance and the slew rate at the transmitter output As illustrated in Fig
32 (b) the transmitter needs to provide a large amplitude value of pulse signal for the
decay in the transmission line However after the transition time the transmitted
waveform becomes steady and the coupling pulse decays according to the RC time
constant The pulse width roughly equals to the rise time during the pulse transition
plus the RC decay time If the amplitude is too large it creates the pulse tail and that
leads to the ISI effect This ISI effect not only limits the communication speed but
also increases the jitter at the receiver end In other words large coupling capacitance
and large termination resistance are good for transferring the pulse signal but that also
create the ISI issue In this thesis we bring up an equalization method to reduce the
ISI effect The details will be discussed in the chapter 4
33 On-Chip Differential Transmission Line
331 Mode of Transmission Line As the wire cross-section dimension becomes smaller and smaller due to the
technology scaling down and the wire length increases due to growing in chip size
not only the capacitance but also the resistance of the wire becomes significant
Besides the wire inductance becomes important as well for the faster circuit operating
speed and relatively lower resistance from the new technology The current
distribution inside the conductor changes as the frequency increases This change is
called the skin effect and that results in variations of resistance and inductance value
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 3 On-Chip Pulse Signaling
19
Fig 33 shows the on-chip transmission line When ωL is comparable to R then the
wire should be modeled as a distributed RLC network The frequency dependent
behavior of the transmission line decays the pulse signal in the channel and makes the
amplitude too small to receive at the far end
Fig 33 On-chip transmission line as a distributed RLC transmission line
Fig 34 (a) shows the proposed on-chip differential transmission line which is
fabricated by TSMC 013microm RF technology A co-planar transmission line is placed
in Metal 6 Metal 5 below is reserved for ground shielding A micro-strip structure is
used in GSGSG placing lsquoSrsquo for signal and lsquoGrsquo for ground The ground path is not only
for signal return but also for the ground shielding The transmission line model is
analyzed and extracted to build the distributed RLC parameters by PTM [21] Fig 34
(b) illustrates the cross section of the differential transmission line The total length (l)
of the line is 5mm The line width (w) is 23microm and line-to-line space (s) is 15microm
On-Chip Transmission Line
Characteristic Impedance Z0
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 3 On-Chip Pulse Signaling
20
Fig 34 (a) Micro-strip structure and the parasitic effect (b) Cross section of the
on-chip transmission line
332 Geometry Analysis of Transmission Line
Table 31 shows the data obtained from PTM The dimension is obtained form
the TSMC 013RF technology document The thickness (t) of Metal 6 is 037microm the
height (h) from Metal 5 to Metal 6 is 045microm and the dielectric constant (k) is 39
The total parasitic RLC value divided by the total length of the transmission line
obtains the parasitic RLC in unit length The distributed parasitic parameters are
Rul=2585Ωmm Lul=174nHmm and Cul=30670 fFmm
Table 31 Parasitic RLC of the on-chip transmission line
Dimension RLC ( 5mm ) RLC ( mm ) w = 23microm s = 15microm l = 5000microm t = 037microm h = 045microm k = 39
R = 129259Ω L = 8728nH M12 = 6876 nH M13 = 6183 nH M14 = 5779nH K12 = 0787 K13 = 0708 K14 = 0662
Rul =258518 Ωmm Lul =17456 nHmm Cg =251088 fFmm Cc =27807 fFmm Cul =306702 fFmm
k h w s
Gnd
l
S G S G G t
Ccouple Ccouple Ccoupl Ccouple
S G SG G
Gnd
Cground Cground
Metal 6
Metal 5
(a)
(b)
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 3 On-Chip Pulse Signaling
21
Cc = 125544 fF Cg = 139035 fF Ct = 153351 fF
According to the data mentioned above the characteristic impeadance (Z0) of the
transmission line is
0ul ul
ul
R j LZj C
ωω+
=
ul
ul
LC
cong (36)
The characteristic impedance of the designed differential transmission is 75Ω Fig 35
shows the relationship between the dimension of the transmission line versus its
characteristic impedance Fig 36 also illustrates the relationship between the
dimension of transmission line versus its parasitic RtotalCtotal which is equivalent to
the bandwidth of the line These two figures tell that smaller width as well as spacing
results in larger characteristic impedance value From (31) we know that it is good
for coupling pulse because the high pass characteristic However small width as well
as spacing results in large parasitic RtotalCtotal which makes the signal decay greatly
On the contrary wider width and spacing of the transmission line results in better
frequency performance of the transmission line But it also requires more layout area
which increases the cost of SOC The design of w=32microm and s=15microm meets the
characteristic impedance of 75Ω And there are three main reasons for this value
Firstly 75Ω of characteristic impedance makes the parasitic RtotalCtotal lt 2E-10
(ΩF) which decay the signal roughly 18dB Secondly 75Ω is close to 77Ω which
theoretically causes minimum attenuation Thirdly the values of the width and
spacing reduce the layout area as well as the costs
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 3 On-Chip Pulse Signaling
22
Fig 35 Transmission line dimension vs characteristic impedance
Fig36 Transmission line dimension vs parasitic RtotalCtotal
40
50
60
70
80
90
100
110
120
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Z 0 (Ω
)05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Characteristic Impedance (Z0)
1E-10
15E-10
2E-10
25E-10
3E-10
35E-10
4E-10
45E-10
5E-10
55E-10
6E-10
50E-07 10E-06 15E-06 20E-06 25E-06 30E-06 35E-06 40E-06
T-Line Spacing (m)
Rto
tal
Cto
tal (Ω
F) 05um
1um
2um
6um
4um
w=23um
05 1 15 2 25 3 35 4 T-Line Spacing (s) (um)
T-Line Dimension (sw) vs Rtotal Ctotal (ΩF)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
23
Chapter 4 5Gpbs Pulse TransmitterReceiver Circuit Design
This chapter introduces the pulse transmitter and pulse receiver operating at
5Gbps At the near end we increase the termination resistance for better high-pass
characteristics The equalization circuit built in the transmitter also cuts the pulse tail
and depresses the ISI effect as discussed in chapter 32 At the far end three functions
are required biasing the received pulse to the common mode voltage of the receiver
amplifying the received pulse amplitude and transferring the RZ pulse to NRZ data
These functions are implemented by a self-bias and equalization circuit an inductive
peaking amplifier and a non-clock latch The following sections will discuss these
circuits and their design methodology
41 Pulse Transmitter Circuit
411 Voltage Mode Transmitter In high-speed serial link design the near end typically uses a current mode driver
as discussed in Chapter 2 but the pulse signaling must use a voltage mode driver as
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
24
shown in Fig 42 (a) The transmitter output is connect to a coupling capacitor (Cctx)
and then to the on-chip transmission line Return impedance matching at the
transmitter output is provided by the termination resistances (Ztx) There are two
advantages to use a voltage mode transmitter rather than a current mode one First the
voltage mode transmitter is much easier to drive the transmission line of high input
impedance Secondly the transmission line loss as well as the high frequency
pass-band characteristic of the transmission line requires a full swing and high edge
rate transmitter output The voltage mode transmitter is quite suit for that Besides the
voltage mode transmitter consumes dynamic power and has no DC current component
on the transmission line That results in less power consumption than a current mode
transmitter Although dynamic supply current will introduce the synchronous
switching noise (SSN) we can add slew rate control circuit or add decoupling
capacitor between power rails to reduce the effect
Fig 41 Output pulse signal at transmitter with different termination resistance
As discussed in Chapter 3 the high pass characteristic of the transmitter passes
the transient part of the input data and blocks the DC signal component Equation (31)
and (34) also implies that we can use a large termination resistor to generate the
better high-pass characteristics Fig 41 shows different values of the termination
resistors and the corresponding pulse signals The larger value of the termination
Ztx -- 1000Ω hellip 150Ω - 75Ω
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
25
resistance is the larger amplitude of the pulse signal will be delivered The trade off is
that the large termination resistance will introduce a pulse tail within a bit time This
tail not only limits the maximum transmission speed but also increases the ISI effect
In section 412 we introduce an equalization method to solve this problem
412 The Proposed De-emphasis Circuit Fig 42 (b) shows a de-emphasis structure for the equalization The structure
consists of a buffer and a coupling de-emphasis capacitor (Ccde) connected to the
differential node of the transmitter output The main idea is to use the structure to
generate a complementary pulse and add the complementary pulse to the original
signal In this way the output pulse signal at the near end can be describes as
y(n)= x(n)+kx(n) (41)
Fig 42 Voltage mode transmitter (a) without de-emphasis (b) with de-emphasis
Cctx
Ninv
Ztx
Cctx Ccde
Ninv
Ztx
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
26
Where x(n) is a full swing data at the last stage of the transmitter x(n)rsquo is the
complement of x(n) and k is a weighting factor which depends on the delay of the
buffer and the value of the de-emphasis capacitor Fig 43 illustrates the circuit
behavior of the transmitter Take the positive terminal for example The Vop is a
positive full swing data at the transmitter output and the x is the coupling pulse The
Von is negative as contrast to Vop The buffer at negative terminal delays the full swing
data And then the de-emphasis capacitor adds the complement pulse xrsquo to the
positive terminal to remove the pulse tail The de-emphasis circuit can reduce the ISI
effect and increase the maximum transmission speed
Fig 43 Coupled pulse signal with de-emphasis circuit
413 Pulse Transmitter Design Flow and simulation results
Fig 44 Pulse transmitter
Td
Data
Coupling Pulse
Vop
Von
Pulse x
xrsquo
y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
27
Fig 44 shows the pulse transmitter circuit which consists of a series of cascaded
inverters as the driver (last stage of size Ninv) a de-emphasis buffer and the coupling
capacitors The driver uses n cascaded inverters and each one of them is larger than
the preceding stage by a width factor f Then the relationship between the output
loading capacitor (CL) and the driver input capacitance (Cgi) can model to be
n L L
gigi
C Cf = n = log logfCC
rArr (42)
Let the equivalent pull-up resistance (Rpu) and the equivalent pull-down resistance
(Rpd) of the inverter are equal The time constant (τ ) of the first stage is ( Rpu+
Rpd)Cgi2 and total delay (td) becomes fτ (log(CLCgi)logf) Take dtddf=0 can
obtain the best choice of f (usually take f=easymp2~3) And then the loading of the driver
including de-emphasis circuit becomes
CL=fnCgi=Cctx+Cgbufferr (43)
Let the gate capacitance of the buffer is x factor larger than Cgi (Cgbufferr= xCgi) then
the coupling capacitor can be calculated
Cctx=fnCgi-Cgbufferr=(fn-x) Cgi (44)
Fig 45 Pulse Transmitter design flowchart
Slew Rate (or Power Budget ) Decide Ninv
Rising Time of Input Data Decide Td
Equation (44) Decide Cctx
Equation (31) Decide Ztx
Decide Ccde For De-emphasis (Cut Pulse Tail)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
28
Besides from equation (31) the proper termination resistance (Ztx) can be
decided Fig 45 shows the design flowchart of the transmitter The buffer delay (Td)
depends on the process and we can obtain the best result by fine tuning the de-couple
capacitor (Ccde) Fig 46 is the simulation of the pulse signal at the transmitter output
with de-emphasis circuit included The diamond eye shows the de-emphasis circuit
cuts the pulse tail and reduces the ISI effect in one data period (200ps) The voltage
swing at the near end is 400mVpp with using a coupling capacitor of 280fF and a
de-emphasis capacitor of 140fF
Fig 46 Simulation of coupled pulse signal with de-emphasis
42 Pulse Receiver Circuit
A pulse receiver is used to transfer RZ pulse into NRZ data Since the coupling
capacitor blocks the DC signal the receiver end needs to bias amplify and then
convert the RZ pulse signal into NRZ data Fig 47 illustrates the receiver end circuit
blocks Because the incoming signal at far end is a short pulse with amplitude smaller
then 50mV a self-bias and equalization circuit is used to bias the pulse to the receiver
common-mode voltage A differential pre-amplify is then connected to amplify the
small pulse After that a non-clock latch with hysteresis has the ability to filter out the
incoming noise from imperfect termination and common-mode disturbances such as
--- Without de-emphasis With de-emphasis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
29
ground bounce The recovered NRZ data can then fed to a traditional clock and data
recovery circuit to generate receiver end clock and re-sample the NRZ data
Fig 47 Receiver end circuit block
421 Receiver End Termination
To minimize reflections either or both side of the transmission line should be
impedance matched In this thesis a receiver end termination is implemented
Terminating at the receiver side reduces reflections and allows the transmitter side to
have high-pass behavior as a transmitted pulse encounters the high impedance
transmitter There is reflection noise at the far end transmission line due to the
forward reflection at the un-terminated near end transmission line This forward
reflection noise is absorbed by the far end termination resistor and thus no further
backward reflection noise shown at the near end termination line Fig 48 illustrates
the receiver end termination The Ceq is the equivalent capacitance of the receiver end
coupling capacitor (Ccrx) in serial with the receiver end gate capacitance (Cg) In
general a node capacitance of a digital circuit is roughly 20fF The Ccrx in our design
is 240fF Thus the equivalent capacitance Ceq=CcrxCg(Ccrx+Cg) asymp20fF In other
words the input impedance at receiver end becomes
Driver End 200mV Pulse
Non-Clock Latch With Hysteresis
Self-Bias amp Equalization
CircuitA
Full Swing NRZ Data
Receiver End lt50mV Pulse
5mm
Pre-amplifier
gt200mV Pulse
A
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
30
rxeq rx rx
rx rxeq eq rx eq
rxeq
1RsC R R1Z = R = = =1sC 1+ sRC 1+2 fR C jR +sC
π (45)
Where the Rrx is the receiver end termination resistance which is equal to the
characteristic resistance of the transmission line (75Ω) and the magnitude is
rxrx rx2 2
rx eq
RZ = R1 +(2 fR C )π
asymp (46)
Equation (46) tells that Zrx is equal to Rrx in low frequency range According to
TSMC 013RF technology the maximum rise time (Tr) of a single inverter is roughly
40ps over 12V power rails This means that the edge rate of a pulse signal is fasymp03
Tr =9GHz The termination resistance in our design matches well over the frequency
range including at the data rate (5Gbps) as well as at the signal edge rate (9GHz)
Fig 48 Receiver end termination scheme
Fig 49 shows the simulation results of the impedance matching at the receiver
end Though at high frequencies the parasitic gate capacitance and the coupling
capacitor reduce Zrx the impedance still matches well across a wide frequency range
A
B C
D
ZZrx
Ceq
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
31
Fig 49 Impedance looking from the transmission line to receiver
422 Self-Bias and Equalization Circuit
Fig 410 shows the self-bias and equalization circuit The circuit not only
automatically generates the receiver end common-mode voltage but also reduces the
low frequency component of the incoming pulse signal As illustrated in the dash line
area an inverter with input connected to output generates the Vdd2 common mode
voltage The common mode voltage is then connected to the receiver differential ends
through two transmission gates The size of the self-bias inverter in our design is the
same as the pre-amplifier in the next stage for the matching consideration Besides
the transmission line has a low-pass response which is due to the skin effect The
low-pass response results in a long tail on the pulse signal If there is no equalization
the tail will cause the ISI effect and reduce the timing margin at the receiver end In
Fig 410 the dotted line area indicates the equalization circuit which consists of an
inverter and a de-emphasis capacitor The circuit de-emphasizes the low frequency
components by generating a small complementary pulse and adding the
complementary pulse to the original signal The output pulse signal of the self-bias
and equalization circuit can be express in
y(n)= x(n)+kx(n)
Impedance Matching (Zrx)
Impedance
Frequency (log) (Hz)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
32
Fig 410 Receiver end Self-bias and equalization circuit
Where x(n) is the pulse signal at the output of the self-bias and equalization circuit
x(n)rsquo is the complement of x(n) and k is the weighting factor which depends on the
delay of the inverter and the size of the de-emphasis capacitor Fig 411 shows the
behavior of the circuit with using a coupling capacitor of 240fF and a de-emphasis
capacitor of 5fF
Fig 411 Pulse waveform after equalization circuit
423 Inductive Peaking Amplifier
An inductive peaking amplifier is composed of parallel inverters as the gain
stage with its output connected to an inductive peaking cell as shown in Fig 412 The
main idea of the circuit is to use the inverter gain for signal amplification and the
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
x
xrsquo
y
x
xrsquo
y
Td
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
33
inductive peaking for improving the high-frequency performance Fig 413 (a) shows
the inductive peaking cell Each inductive peaking cell consists of a small size
inverter and a resistor The inverter is configured as diode connected by a resistor
which is implemented with a transmission gate as illustrated in Fig 412 The diode
connected inductive peaking cell lowers the output resistance of the inverter and so
does the gain As a result the 3dB frequency increases as the output resistance
decreases At low frequency the output resistance is roughly 1gm while it roughly
equals to the resistance of a transmission gate at high frequency It is intended to
design the resistance of a transmission gate to be larger than 1gm so it increases gain
and extends bandwidth at high frequency
Fig 412 Receiver end Inductive peaking amplifier
In other words the attached inductive peaking cell at the output of the inverter
sacrifices gain but obtains bandwidth to enhance the high-frequency performance
Moreover as illustrated in Fig 413 (b) this circuit not only amplifies the high
frequency component but also depresses the low frequency pulse tail A lightly
transition overshoot due to the inductive peaking results in good performance in the
high frequency part The pulse signal is amplified larger than 200mV and fed into the
hysteresis latch
Self-bias And Equalization Circuit
Non-Clock Latch With Hysteresis
Pre-amplify With Inductive Peaking
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
34
Fig 413 (a) Inductive peaking cell (b) Pulse waveform after inductive peaking amplify
Although the inductive peaking cell doesnrsquot have any actual inductor inside the
parasitic capacitance of inverter (Cgdp Cgdn) combining the resistor generates a high
frequency zero that improves the high frequency performance The frequency
response is like adding an inductor This broadband technique is so called the
inductive peaking Fig 414 shows the gain-bandwidth plot of the inductive peaking
amplifier The 3dB bandwidth extends from 388GHz to 113GHz after the inductive
peaking cell is used The receiver sensitivity depends on the equalization as well as
the peaking ability of the amplify stage
Fig 414 Gain-Bandwidth plot of inductive peaking amplifier
No Inductive Peaking f3db=388GHz
Inductive Peaking f3db=113GHz
i OC gdp
C gdnR
200ps
lt50mVPulse
gt200mV Pulse
(a) (b)
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
35
424 Non-clock latch with hysteresis The non-clock latch transforms the RZ pulse into NRZ data So that the
recovered NRZ data can then be fed to a traditional clock and data recovery circuit to
generate the receiver end clock to re-sample the NRZ data Fig 415 illustrates the
non-clock latch structure established from four inverters and Fig 416 shows its
schematic Via and Vib are the differential inputs and Voa and Vob are their
corresponding differential outputs Two small size inverters are connected back to
back at the differential output to generate the hysteresis range
Fig 415 Receiver end Non-clock latch with hysteresis
As demonstrated in Fig 416 and Fig 417 when the input Via goes from logic
high (H) to the trigger point (Vtrp-) the corresponding output Voa will goes from logic
low (L) to the threshold point (Vth) If we let p =β p ox pminC (WL)μ n =β
n oxCμ nmin(WL) (where micro is the mobility of transistor Cox is the oxide capacitance
and (WL) is the aspect ratio of the transistor) Then the total current flows through
the node Voa becomes
px py nx nyI + I = I + I
- 2 2ddp dd TRP tp p dd tp
V1 1X [(V -V )-V ] + Y [(V - ) -V ]2 2 2
β β=
- 2 2ddn TRP tn n tn
V1 1= X (V -V ) + Y ( -V )2 2 2
β β (47)
Self-bias And Equalization Circuit
Pre-amplify Stage
Non-Clock Latch With Hysteresis
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
36
Where X is size of the input inverter and Y is the size of the back to back inverter pair
With the following replacements
1 pB = X β 2 nB = X β 3 dd tpB = (V -V ) 4 tnB =V
2 2dd dd5 p dd tp n tn
V VB = (Y [(V - )-V ] +Y ( -V ) )2 2
β β
2 21 1 2A = (B - B ) 2 l 2 2 4 1 3 1 2 2 4 1 3A = (B + B )(B B - B B )-(B - B )(B B + B B )
2 2
3 1 3 2 4 5A = (B B ) -(B B ) - B
(47) becomes
- 2 -
1 TRP 2 TRP 3A (V ) + A (V )+ A = 0
The trigger point can be solved
[ ] TRP-V = roots( A1 A2 A3 ) (48)
Fig 416 Non-clock latch with hysteresis circuit
Take TSMC 013RF technology for example for X = 3 Y = 1 ddV = 12(V)
2p = 15166e - 3 (AV )β 2
n = 18198e - 3 (AV )β tpV = 403817e - 3 (V) and
tnV = 403596e - 3(V) we can obtain the trigger point ( TRP-V ) of 05578V The proper
design of X and Y values makes the hysteresis range to filter out the incoming noise
and the interference
Voa
Vob
Via
Vib
HrarrVTRP-
LrarrVth
X
Y
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
37
Fig 417 Simulation of non-clock latch with hysteresis circuit
Fig 418 Jitter from the receiver
Data
Ideal Pulse
Over Peaking
Vp
Vp
Vp
Vp Jitter
Trigger PointOf Latch
VTRP+VTRP
-
Voa
Via
VOL
VOH
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 4 5Gbps Pulse TransmitterReceiver Circuit Design
38
Inductive peaking amplifier may over peak the pulse signal for some corner
cases especially in the SS corner This is because the resistance of the peaking cell
will change and results in different peak ability Fig 418 illustrates the circuit
behavior of the pulse receiver This over peaking behavior will contribute some jitter
at the receiver end However the hysteresis range of the latch can solve this problem
As long as the peaking amplitude is smaller than the hysteresis range the latch can
ignore the over peaking
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 5 Experiment
39
Chapter 5 Experiment
One prototype chip was designed in TSMC 013RF technology to verify the
research ideas presented in previous chapters The prototype chip contains a voltage
mode transmitter a 5mm long differential transmission line and a pulse receiver
Besides in order to drive the large parasitic capacitance on the output pad open drain
output drivers are used for the receiver differential output In this way the 5Gbps
high-speed data can be measured on the oscilloscope The simulated peak-to-peak
jitter at the pulse receiver output is 437ps The die size of the chip is 884times644microm2
with the power consumption of 34mW at the transmitter end and 32mW at the
receiver end
51 Simulation Setup
A test chip is implemented in TSMC 013microm RF technology The building block
of the system is shown in Fig 51 We apply a sequence of random data to the input
of the voltage mode pulse transmitter The data are transported in pulse mode through
the coupling capacitor (Cctx) to the on-chip transmission line As illustrated in Fig
51 the differential structure of the transmission line is built in Metal 6 and we use
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 5 Experiment
40
Metal 5 as the shielding ground In addition the differential structure of the
transmission line also has the shielding grounds in the arrangement of lsquoGSGSGrsquo
between signal paths The differential scheme has higher immunity to reject noise and
the shielding ground reduces the environmental interference as well as the crosstalk
between signal lines At far end the pulse data is coupled to the hysteresis receiver by
the coupling capacitors (Ccrx) The receiver then transforms RZ pulses signal to NRZ
data
Fig 51 System Architecture
In order to drive a large parasitic capacitance of an output pad open drain output
drivers are used for receiver differential output as shown in Fig52 The package of
the chip is modeled as a pad (C=1pF) connected to a pin (C=1pF) on the printed
circuit board through a pounding wire (L=2nH) Thus the open drain circuit will
introduce a current (Ics) from the outside-chip power supply The current goes through
the impedance matching resistor (50Ω) and drives the large parasitic capacitance on
O
OCcr Rx
Zrx=75Ω Vterm
D
D
Tx
C ct
Ztx Vterm
Cctx=280fF Cde=140fF
Ccrx=240fF Cde=5fF
Ztx=1000Ω
5mm on-chip differential line with shielding ground
5Gpbs eye
G S G S G
Cross-section of T-Line
M5(G) M5(G)
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 5 Experiment
41
the package One oscilloscope with 100nF bypass capacitor connected to the pin is
used to probe the signal at the chip output Therefore we can observe the eye diagram
on the oscilloscope
Fig 52 Common source circuit connects to pad for output signal measurement
52 Experiment and Results
Fig 53 shows the system simulation results The transmitter end sends the
differential pulse signal with an amplitude of 400mVpp After a 5mm long on-chip
transmission line the pulse amplitude decays to 50mVpp at the receiver front end The
pre-amplifier stage amplifies the pulse amplitude to 400mVpp and fed it to the
hysteresis latch The latch then turns the RZ pulse signal into the full swing NRZ data
After the open drain circuit as discussed in section 51 the oscilloscope can measure
the received data with a swing of 240mVpp Fig 54 shows the eye diagrams at the
receiver output in different process corners The eye diagram of the receiver shows
the peak-to-peak jitter is 437ps (0218UI) in the TT case The worse case takes place
in SS corner and the jitter is 827ps (0413UI)
Vpp= Ics x 50 PAD
Buffer Receiver
50Ω
Oscilloscope
100nF
PIN2nH
1pF
div Ldt
=
Inside Chip
Common Source (Open Drain) Connect To PAD
Ics
1pF 50Ω
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 5 Experiment
42
Fig 53 System simulation results
The proposed on-chip pulse signaling is implemented by National Chip
Implement Center (CIC) in TSMC 013RF technology The core area is
5847micromtimes4115microm including a transmitter of 961micromtimes578microm a receiver of
801micromtimes521microm and a 5mm long on-chip transmission line The total area is
884micromtimes644microm as shown in Fig 55 Table 51 lists the chip summary The power
consumption of the transmitter is 34mW and it is 32mW for the receiver at the data
rate of 5Gbps
400mVpp
~240mV
Full Swing
Rx after Pre-Amp
400mVpp
50mVpp
Input Data
Tx Pulse
Rx Pulse
400mVpp
~240mV
Full Swing
Latch
Oscilloscope
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 5 Experiment
43
Fig 54 Corner simulation results
Table 51 Specification Table
Item Specification (unit) Process TSMC 013microm RF Supply Voltage 12V Data Rate 5Gbs channel (at 25GHz) BER lt10-12 Coupling Caps Tx (280+140)fF Rx (240+5)fF Link 5mm and 75Ω on chip micro-strip line Jitter of receiver data (pk-to-pk) 437ps (0218UI) Transmitter End Layout Area 57microm x96microm Receiver End Layout Area 52microm x78microm Core Layout Area 884microm x644microm
Pulse Driver 3402mW Power dissipation Pulse Receiver 3213mW
Total 6615mW
TT 437ps (0218UI)
FF 296ps (0148UI)
FS 525ps (0262UI)
SF 497ps (0248UI)
SS 827ps (0413UI)
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 5 Experiment
44
Fig 55 Layout
Table 52 lists the comparison of the pulse signaling and other on-chip
communications Our work uses TSMC 013RF technology to implement an on-chip
5Gbps pulse signaling The total communication distance is 5mm with a differential
transmission line of width in 23microm and line-to-line spacing in 15microm The line
consumes small area overhead as compares to other on-chip differential lines [22]
[23] Compared to the twisted differential wire method [24] our work uses a wider
width as well as the wider line-to-line space But the twisted differential wire method
has only 40mVpp voltage swing at the far end However our work has full swing data
at receiver output and that can be used for further receiver end usage The power
consumption of the transmitter in our work is 068pJbit and the total power is
132pJbit Our work is the lowest in power consumption
5mm On-Chip Differential Transmission Line
Tx
Rx
gndtx
vip
vddtx
vddrx
vpon vpop
vbrx
gndrx
gndcs
884um
vin vbtx
Decouple Cap
644um
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 5 Experiment
45
Table 52 Comparison of high speed data communication
Communication Chip-to-chip On-chip On-chipYear 2005 [12] 2006 [14] 2005 [22] 2006 [23] 2006 [24] This work
Method Pulse Signaling
Pulse Signaling
Differential T-Line
Differential T-Line
Twisted Differential
Wires
Pulse Signaling
Technology 01microm 018microm 018microm 018microm 013microm 013um Data rate 1Gbps 3Gbps 4Gbps 5Gbps 3Gbps 5Gbps
T-Line (microm) - - w=40 s=42
w=40 s=40
w=04 s=04
w=23 s=15
T-Line length 10cm FR4 15cm FR4 2mm 3mm 10mm 5mm Tx Power
(mW) 29 5 317mA 35 - 34
Rx Power (mW) 27 10 24mA 3 - 32
Total power (mW) 56 15 - 65 6 66
Tx Powerbit (pJbit) 29 167 - 07 05 068
Total Powerbit (pJbit) 56 5 - 13 2 132
53 Measurement Considerations
The whole measurement environment is shown in Fig 56 We use a N4901B
Serial BERT to generate 5Gbps random data and send that into the chip The data pass
through the pulse transmitter on chip transmission line and then into the pulse
receiver The output data of the chip are sent back to the N4901B Serial BERT for the
bit error rate (BER) measurement At the same time the output data are also sent to
an Agilent 86100B for the eye diagram measurement We expect the output data rate
is 5Gbps with 240mV swing and the peak-to-peak jitter of 021UI Besides three HP
E3610A DC Power Supplies are used One is for the output open drain circuit and the
other two are for the transmitter and the receiver power rails The power consumption
is also measured by a Ktythley 2400 Source Meter
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Chapter 5 Experiment
46
Fig 56 Measurement instruments
Agilent 86100B Wide-Bandwidth
Oscilloscope
HP E3610A DC Power Supply
(Tx Rx CS)
N4901B Serial BERT 5 Gbps
Input [vipvin]
Output [vpopvpon]
BER Measurement
Jitter Measurement
Supply 12V0V
Ktythley 2400 Source Meter
Power Measurement
Power Measurement
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Bibliography
47
Chapter 6 Conclusion
In this thesis we have proposed a 5Gbps on-chip pulse signaling interface
Different from previous researches of pulse signaling or other on-chip communication
our near end architecture uses a high termination resistor combing the de-emphasis
scheme to reduce ISI effect as well as to increase the maximum data rate At far end
the self-bias circuit the pre-amplifier and the non-clock latch compose the receiver
circuit The self-bias circuit generates the common mode voltage for the receiver such
that the pulse mode data can be received The amplifier stage and the non-clock latch
increase the amplitude of the pulse signal and then transfer the RZ pulse signal into
NRZ data The latch also has an input hysteresis range that can filter out the incoming
noise from imperfect termination and common-mode disturbances such as the ground
bounce The receiver circuit is design in a simple scheme and easy for
implementation
In chapter 3 we have analyzed the pulse signaling as well as the on-chip channel
model We have also designed a 5mm on-chip differential transmission line in our
chip The characteristic impedance of the line is 75Ω to minimize the attenuation
Furthermore the geometry of the line is chosen in the width of 23microm and the spacing
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Bibliography
48
of 15microm Our on-chip transmission line has small area overhead as compares to other
works
The simulation results show that the receiver has a peak-to-peak jitter of 407ps
from 12V supply The power consumption of the transmitter is 068pJbit and total
power is 132pJbit This on-chip pulse signaling is fabricated in TSMC 013microm RF
technology The total chip occupies 884micromtimes644microm of area including a transmitter of
57micromtimes96microm a receiver of 52micromtimes78microm and an on-chip transmission line The chip
will be sent back and measured in October 2007
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Bibliography
49
Bibliography [1] Hamid Hatamkhani Chin-Kong Ken Yang ldquoPower Analysis for High-Speed
IO Transmittersrdquo IEEE Symposium On VLSI Circuit Digest of Technical Papers pp142-145 Jan 2004
[2] Ramin Farjad-Rad Chih-Kong Ken Yang Mark A Horowitz and Thomas H
Lee ldquoA 04- m CMOS 10-Gbs 4-PAM Pre-Emphasis Serial Link Transmitterrdquo IEEE J Solid-State Circuits VOL 34 NO 5 MAY 1999
[3] Bashirullah R Wentai Liu Cavin R III Edwards D ldquoA 16 Gbs adaptive
bandwidth on-chip bus based on hybrid currentvoltage mode signalingrdquo IEEE J Solid-State Circuits Vol 41 Issue 2 pp 461-473 Feb 2006
[4] Chih-Kong Ken Yang ldquoDESIGN OF HIGH-SPEED SERIAL LINKS IN
CMOSrdquo PhD Dissertation Stanford University California Dec 1998 [5] Kun-Yung Chang ldquoDesign Of A CMOS Asymmetric Serial Linkrdquo PhD
Dissertation Stanford University California Aug 1999 [6] ldquoIntroduction to LVDS PECL and CMLrdquo MAXIM High-FrequencyFiber
Communications Group Application Note HFAN-10 (Rev 0 900) Some parts of this application note first appeared in Electronic Engineering Times on July 3 2000 Issue 1120
[7] Ching-Te Chiu Jen-Ming Wu Shuo-Hung Hsu Min-Sheng Kao Chih-Hsien
Jen Yarsun Hsu ldquoA 10 Gbs Wide-Band Current-Mode Logic IO Interface for High-Speed Interconnect in 018μm CMOS Technologyrdquo IEEE International SOC Conference pp 257-260 25-28 Sept 2005
[8] Mingdeng Chen Silva-Martinez J Nix M Robinson MErdquo Low-voltage
low-power LVDS driversrdquo IEEE J Solid-State Circuits Vol 40 Issue 2 pp 472-479 Feb 2005
[9] Gabara TJ Fischer WC ldquoCapacitive coupling and quantized feedback
applied to conventional CMOS technologyrdquo IEEE J Solid-State Circuits Vol 32 No 3 pp 419-427 March 1997
[10] Kuhn SA Kleiner MB Thewes R Weber W ldquoVertical signal
transmission in three-dimensional integrated circuits by capacitive couplingrdquo IEEE International Symposium on Circuits and Systems Vol 1 pp 37 ndash 40 May 1995
[11] Mick S Wilson J Franzon P ldquo4 Gbps High-Density AC Coupled
Interconnectionrdquo Custom Integrated Circuits Conference pp 133 ndash 140 May 2002
[12] J Kim I Verbauwhede and M-C F Chang ldquoA 56-mW 1-Gbspair pulsed
signaling transceiver for a fully AC coupled busrdquo IEEE J Solid-State Circuits vol 40 no 6 pp 1331ndash1340 Jun 2005
[13] Mick S Luo L Wilson J Franzon P ldquoBuried bump and AC coupled
interconnection technology rdquoIEEE TRANSACTIONS ON ADVANCED PACKAGING VOL 27 No 1 Feb 2004
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006
Bibliography
50
[14] L Luo J MWilson S E Mick J Xu L Zhang and P D Franzon ldquoA 3 Gbs
AC coupled chip-to-chip communication using a low swing pulse receiverrdquo IEEE J Solid-State Circuits Vol 41 No 1 pp 287-296 Jan 2006
[15] Min Chen and Yu CaordquoAnalysis of pulse signaling for low-power on-chip
global bus designrdquo Proceedings of the 7th International Symposium on Quality Electronic Design Mar 2006
[16] Jongsun Kim Jung-Hwan Choi Chang-Hyun Kim Chang AF Verbauwhede
I ldquoA low power capacitive coupled bus interface based on pulsed signalingrdquo Custom Integrated Circuits Conference pp 35 ndash 38 Oct 2004
[17] Gijung Ahn Deog-Kyoon Jeong Gyudong Kim ldquoA 2-Gbaud 07-V swing
voltage-mode driver and on-chip terminator for high-speed NRZ data transmissionrdquo IEEE J Solid-State Circuits Vol 35 Issue 6 pp 915-918 June 2000
[18] Gomi S Nakamura K Ito H Sugita H Okada K Masu K ldquoHigh speed
and low power on-chip micro network circuit with differential transmission linerdquo International Symposium on System-on-Chip pp 173-176 Nov 2004
[19] Chang RT Talwalkar N Yue CP Wong SS ldquoNear speed-of-light
signaling over on-chip electrical interconnectsrdquo IEEE J Solid-State Circuits Vol 38 Issue 5 pp 834-838 May 2003
[20] So Young Kim ldquoModeling And Screening On-Chip Interconnection
Inductancerdquo PhD Dissertation Stanford University California July 2004 [21] httpwww-deviceeecsberkeleyedu~ptm [22] Ito H Sugita H Okada K Masu K ldquo4 Gbps On-Chip Interconnection
using Differential Transmission Linerdquo Asian Solid-State Circuits Conference pp 417-420 Nov 2005
[23] Takahiro Ishii Hiroyuki Ito Makoto Kimura Kenichi Okada and Kazuya
Masu ldquoA 65-mW 5-Gbps On-Chip Differential Transmission Line Interconnect with a Low-Latency Asymmetric Tx in a 180 nm CMOS Technologyrdquo IEEE A- Solid-State Circuits Conference pp131-134 Jul 2006
[24] Danieumll Schinkel Eisse Mensink Eric A M Klumperink Ed (A J M) van
Tuijl and Bram Nauta ldquoA 3-Gbsch Transceiver for 10-mm Uninterrupted RC-Limited Global On-Chip Interconnectsrdquo IEEE J Solid-State Circuits vol 41 no 1 pp 297ndash306 Jan 2006