IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 11, NOVEMBER 2009 2783

Broadband Micro-Coaxial Wilkinson DividersNegar Ehsan, Student Member, IEEE, Kenneth Vanhille, Member, IEEE, Sébastien Rondineau, Member, IEEE,

Evan D. Cullens, Student Member, IEEE, and Zoya B. Popovic, Fellow, IEEE

Abstract—This paper presents several micro-coaxial broadband2 : 1 Wilkinson power dividers operating from 2 to 22 GHz, a 11 : 1bandwidth. Circuits are fabricated on silicon with PolyStrata tech-nology, and are implemented with 650 m 400 m air-supportedmicro-coaxial lines. The measured isolation between the outputports is greater than 11 dB and the return loss at each port is morethan 13 dB over the entire bandwidth. The footprints of these di-viders can be miniaturized due to the high isolation between ad-jacent coaxial lines and their tight bend radius. For higher powerhandling, larger lines with a cross section of 1050 m 850 mare also demonstrated. The effect of mismatch at the output portsis investigated in order to find the power loss in the resistors.

Index Terms—Coaxial components, coaxial transmission lines,power dividers.

I. INTRODUCTION

T HE Wilkinson power divider was first introduced in 1960as a distributed -way circuit with lumped resistors

[1]. Various distributed and lumped extensions of this in-phasehigh-isolation divider have been researched to date [2]. For asingle-section two-way divider the bandwidth is around 3 : 1 fora voltage standing-wave ratio (VSWR) of 2 : 1. Several methodshave been developed to increase the bandwidth of such dividers.In 1968, Cohn introduced a multisection divider, containingpairs of equal-length transmission lines in series and shuntresistors between the pairs [3]. This divider can theoreticallyachieve 10 : 1 bandwidth with with an impedance rangeof 90–50 and resistor values of 100–600 . High isolationcan be maintained between output ports over a broad bandwidth.Other broadband Wilkinson-type dividers are designed mostlyin microstrip [4]. Algorithms to find the correct number of sec-tions, characteristic impedances, and lumped resistor values aregiven for ideal components [5], [6]. The greatest experimentallydemonstrated bandwidth with the Wilkinson divider to the bestof the authors’ knowledge is 4 : 1 (3–12 GHz) using a strip-lineconfiguration with four sections [5].

Manuscript received May 15, 2009; revised August 07, 2009. First publishedOctober 13, 2009; current version published November 11, 2009. This work wasfunded by the Defense Advanced Research Projects Agency (DARPA) under theDMT Program, U.S. Army Contract W15P7T-07-C-P437, and is approved forPublic Release, Distribution Unlimited; DARPA case #13204, 3/16/09.

N. Ehsan, E. D. Cullens, and Z. B. Popovic are with the Department of Elec-trical and Computer Engineering, University of Colorado at Boulder, Boulder,CO 80309 USA (e-mail: negar.ehsan@colorado.edu; evan.cullens@colorado.edu; zoya@colorado.edu).

K. Vanhille is with Nuvotronics LLC, Blacksburg, VA 24060 USA (e-mail:kvanhille@nuvotronics.com).

S. Rondineau is with TSM Antennas, Santa Maria RS 97105-030, Brazil.Color versions of one or more of the figures in this paper are available online

at http://ieeexplore.ieee.org.Digital Object Identifier 10.1109/TMTT.2009.2032345

Fig. 1. (a) Rendering of the miniaturized 2–22-GHz Wilkinson divider imple-mented in the five-layer fabrication process. (b) Cross section of the five-layermicro-coaxial line. The height of layers 1, 3, and 5 is 100 �m, and layers 2 and 4are 50-�m tall. The inner conductor is supported by 100-�m wide, 18-�m-highdielectric straps that are embedded in the bottom of layer 3. � � ��� �m,� � ��� �m, and � � ���� �m are the dimensions for a 50- line.

The goal of this paper is to demonstrate broadband Wilkinsondividers implemented in wafer-scale PolyStrata technology [7],which provides both low loss and small footprints simultane-ously; the latter is illustrated in Fig. 1(a). The micro-coaxiallines are air filled and fabricated using sequential metal de-position and standard photolithography. A large number ofmillimeter-wave narrowband components such as resonators,hybrids, antennas, and narrowband dividers have already beendemonstrated with this process [8]–[11].

The PolyStrata micro-coaxial lines have loss as low as0.1 dB/cm (0.08 dB/wavelength) at 38 GHz [12], and veryhigh isolation of 60 dB at the -band for neighboringlines sharing a common ground wall [13]. The characteristicimpedance of the lines is constant over a broad range offrequencies since the TEM mode is dominant up to around450 GHz, depending on the line geometry [14]. The low dis-persion, accompanied by low loss and high isolation, makes thePolyStrata process uniquely suitable for ultra-broadband minia-turized components, such as the 11 : 1 bandwidth Wilkinsondivider/combiner presented here.

Since a Wilkinson divider requires resistors, in this studywe demonstrate for the first time specially designed 3-D sur-face mount pads for hybrid integration of 0402 and 0303 stan-

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2784 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 11, NOVEMBER 2009

Fig. 2. Circuit schematic of the broadband Wilkinson divider with indicated characteristic impedances, lengths, and resistor values. This model consists of fivepairs of transmission lines with five resistors between each pair. The input transmission line is composed of four sections.

dard packaged resistors. This hybrid assembly method can beextended to other lumped passive and active components. Thedividers utilize the available discrete values of surface-mountbroadband resistors, and follow the constraints of the PolyStratafabrication rules, which determine the achievable characteristicimpedances and the overall size of the circuit.

Contributions of this work are as follows.• Section II details the design procedure and describes the

fabrication process and its limitations. In particular, thecircuit optimization process, as well as the full-wave elec-tromagnetic (EM) simulation including hybrid resistors, ispresented. In addition, the miniaturization in terms of foot-print reduction for a given coaxial line size is discussed.

• Section III presents the measured results on two versionsof a smaller cross-section design, including a discussion oncalibration using PolyStrata standards.

• The final two sections discuss unbalanced loads, powerdissipation in the resistors, and possible extensions of thiswork to other types of dividers.

II. FABRICATION AND DESIGN PROCEDURE

A. Fabrication

The PolyStrata process involves sequential deposition ofcopper layers and photoresist on a silicon wafer. Copper layerthickness ranges from 10 to 100 m, with gap-to-height andwidth-to-height aspect ratios of 1 : 1.2 and 1 : 1.5, respec-tively. Fig. 1(b) shows the cross section of a five-layer 50-micro-coaxial line in which layers 1, 3, and 5 are 100- m thickand layers 2 and 4 are 50- m thick. The inner conductor issupported by 100- m-long dielectric straps with periodicityof 700 m. After the desired layers have been deposited, thephotoresist filling all space unoccupied by copper and dielectricstraps is rinsed away (“released”) through 200 m 200 mrelease holes on layers 1, 2, 4, and 5 with 700- m periodicity.Characteristic impedances available for a micro-coaxial linewith this layer configuration range from 8 to 54 [15].

B. Design Procedure

As mentioned in Section I, a general broadband Wilkinsondivider consists of pairs of transmission lines with shunt re-sistors between them [3]. The number of sections, characteristicimpedances, and resistor values determine the bandwidth. SinceCohn’s basic design of a 50- Wilkinson requires impedances

(a) (b)

Fig. 3. Full-wave simulation model of two passive sockets designed for a shunt(a) 0402 and (b) 0303 chip resistor package. The lumped resistor is connectedbetween the two pads; for (b), the two micro-coaxial lines share an outer con-ductor wall.

Fig. 4. FEM and circuit simulation comparison. �� � and �� � are�3.02 and �3.70 dB at 12 GHz, respectively. Conductor losses are not takeninto account in the circuit simulator.

greater than those available with the layer configuration shownin Fig. 1(b), we chose to transform the input impedance of 50to a lower impedance at the point of the division. Port 1 is,therefore, composed of several sections of line with impedancesvarying from 50 to approximately 30 . A constraint of 54was placed on the maximum allowable impedance for all trans-mission line sections in Fig. 2, and Ansoft Designer [16] was

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EHSAN et al.: BROADBAND MICRO-COAXIAL WILKINSON DIVIDERS 2785

Fig. 5. (a) Photograph of the fabricated straight Wilkinson divider for both 0402 and 0303 sockets. Resistors are mounted on the divider with 0303 passive sockets.(b) Photograph of an 0303 passive socket with no resistor. (c) Photograph of the fabricated miniaturized Wilkinson divider.

Fig. 6. (a) �� � and �� �. (b) Return loss and isolation for the straight divider. (c) Measured phase and amplitude imbalance for straight divider; note that thescale covers 1 in phase and 0.1 dB in amplitude.

used to optimize the divider with - input and outputports. Fig. 2 shows the circuit model in which the characteristicimpedances and the lengths of the transmission lines are chosenfrom the results of the optimization process, and the resistorsin between them are chosen based on the availability of resistorvalues in 0402 and 0303 packages. Due to the impedance con-straint, six transmission line sections were required to achievethe desired return loss and isolation over the desired bandwidth.

The circuit from Fig. 2 was modeled in Ansoft’s HighFrequency Structure Simulator (HFSS) [17] for the five-layerprocess and in a straight geometry [unlike the reduced-foot-print version shown in Fig. 1(a)]. The cross sections of eachtransmission line section are calculated using the method from[14]. In order to place surface mount resistors in shunt betweentransmission line sections, 3-D assembly pads, referred to as“passive sockets,” are designed for both 0303 and 0402 standardpackages, as shown in Fig. 3. The design of the passive socketsdepends on the cross-sectional geometry of the transmissionlines at the location where the socket is placed. If the field dis-tribution is concentrated in the upper gap (layer 4), designingthe sockets becomes more difficult due to field leakage fromthe open area of the outer conductor. Since the fields were notvery concentrated in the upper gap layer, there was not muchdifficulty in this design. This issue will be discussed in moredetail for a divider with input and output ports of 12.5 inSection V.

In the finite-element method (FEM) simulation, the resistorsare modeled as purely resistive impedance sheets. Fig. 4 showsthe comparison between circuit simulations and the FEM resultswith the 0303 resistor packages. For the circuit model, idealtransmission lines and resistors are used; however, in the HFSS

model, the conductor losses, as well as parasitics due to the “pas-sive sockets” and resistors are taken into account, resulting inincreased insertion loss. This device has a VSWR better than2 : 1 from 2 to 22 GHz, and a better than 20-dB return loss andisolation from 4 to 18 GHz.

C. Miniaturization

In order to reduce the footprint of the straight Wilkinson di-vider, we miniaturized it by bending the transmission lines asshown in Fig. 1(a). The length of the miniaturized divider is18 mm, significantly less than the 48-mm-long straight divider.This is facilitated by the high isolation between adjacent micro-coaxial lines [18] and judicious EM design of the tight bends.The difficulties encountered during the design of the miniatur-ized Wilkinson divider include maintaining the characteristicimpedances of the transmission lines through the curves andcareful design of the passive sockets.

III. PROTOTYPE PERFORMANCE

Fig. 5 contains photographs of a fabricated straight Wilkinsondivider, a passive socket for 0303 package surface mount com-ponent, and a miniaturized Wilkinson divider. The dividers weremeasured with an Agilent E8364B PNA four-port network an-alyzer, Cascade Microtech 250- m-pitch coplanar waveguide(CPW) microwave probes, and a Cascade Summit 9000 probestation. Two calibration methods were performed; the first wasa three-port short-open-load-thru (SOLT) implemented in theCPW on an alumina substrate. This calibration method removesthe effect of the cables and probes up to the probe tips. Thestraight Wilkinson divider was measured with this calibration

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2786 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 11, NOVEMBER 2009

Fig. 7. Simulated and measured �-parameter results of the miniaturizedWilkinson divider with 0402 resistors. Measured �� � at 12 GHz is�3.71 dB.

method. Fig. 6(a) and (b) shows the comparison between -pa-rameter simulation and measurement results of this divider. Onepossible reason for the reduced performance in measuredand above 16 GHz is imperfection in the manual mountingand positioning of the resistors in the sockets.

A broadband Wilkinson divider ideally has no amplitude orphase imbalance due to perfect symmetry [3]. In the actual cir-cuit, however, due to fabrication imperfection, the symmetry isbroken and the measured amplitude and phase imbalance arecalculated from

dB (1)

and

deg (2)

Fig. 6(c) shows the measured amplitude and phase imbalanceof the straight Wilkinson divider. The phase imbalance is betterthan 1 and the amplitude imbalance is better than 0.1 dB.

The second calibration method is performed via an on-waferPolyStrata thru-reflect-line (TRL) calibration standard with twoline standards to cover the full bandwidth. The TRL calibrationmethod removes the effect of the cables, probes, and probe toPolyStrata transition. The miniaturized Wilkinson divider wascalibrated through this method; specifically, the divider wasmeasured with the PNA using the three-port TRL calibrationand three 250- m-pitch CPW microwave probes. Fig. 7 showsthe comparison between simulated and measured results ofthe miniaturized Wilkinson divider. Measured amplitude andphase imbalance, from (1) and (2), are 0.1 dB and 1 ,respectively.

IV. POWER CONSIDERATIONS

A. Effect of Mismatched Loads

In an ideal Wilkinson power divider, the current in the resis-tors is zero. However, this is not the case when there are slight

TABLE IRMS CURRENT THROUGH RESISTORS FOR 1-W INPUT

POWER AND VARIOUS LOAD MISMATCHES

Fig. 8. Transient response simulation from � � � ns to � � ��� ns of current ineach resistor. Ports 2 and 3 are connected to 40- and 60-� loads, respectively.The analysis was performed for a 2-ns duration with 0.1-ps steps.

mismatches at the output ports, or in the worst case, when a de-vice or line at an output port fails as short or open. It is impor-tant to know how much current can flow through the resistorsfor specific mismatches in order to select a resistor with ap-propriate power handling characteristics. Both the output portmismatches and the current flow in the resistor are investigatedfor the ideal broadband divider shown schematically in Fig. 2.Table I gives the rms current in each resistor for different outputport impedances ranging from 40 to 60 with 1 W of inputpower at 10 GHz. The maximum current in the resistors for 60-and 40- loads at the two ports is 4.71 mA in . Fig. 8shows the simulated transient response of the current flow ineach resistor for this case. The transient analysis was performedwith a circuit simulator for 2-ns duration in 0.1-ps steps. Notethat is the first resistor into which the current flows since thisresistor is closest to the mismatched output ports. The selected0402 (USMRG2040AN) and 0303 (USMRG3000AN) resistorscan dissipate 3.45 and 3.89 W, respectively, providing a signifi-cant safety margin.

B. RF Power Handling

The power-handling capability of the dividers in this paper islimited primarily by the transmission lines’ thermal and elec-trical breakdown limits, as previously discussed in detail byEhsan et al. [15]. In that study, it was concluded that the elec-trical breakdown limit of 120 W for the five-layer 50- line issignificantly above the thermal limits. micro-coaxial lines withlarger cross sections allow greater power-handling capability,increased design flexibility, and lower insertion loss. Fig. 9(a)

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EHSAN et al.: BROADBAND MICRO-COAXIAL WILKINSON DIVIDERS 2787

Fig. 9. (a) Cross section of the 11-layer micro-coaxial line; the height of layers 2, 10, and 11 is 50 �m; all other layers are 100-�m high. � � ���� �m,� � ��� �m, and � � ��� �m are the dimensions for a 50-� line with this cross section. (b) Simulated and measured �� � and �� � for the 11-layerWilkinson divider. (c) Return loss, isolation, and match at the output ports.

Fig. 10. Circuit schematic of the broadband Wilkinson divider designed for the11-layer process. The values of characteristic impedances, resistors, and lengthof each section are given in Table II.

TABLE II11-LAYER 50-� WILKINSON PARAMETERS

shows the cross section of such a line, with 11 layers. The avail-able characteristic impedances for micro-coaxial lines with the11-layer configuration range from 6 to 140 . A broadbandminiaturized micro-coaxial Wilkinson divider was designed andfabricated with this larger cross-section line. Fig. 10 shows thecircuit schematic of this divider, which consists of four paralleltransmission lines and four resistors in the divider section, andthree transmission lines sections in the input line. Table II showstheir values. Although fewer transmission line sections are usedin this design, the resulting bandwidth is still 2–22 GHz. Thisis due to the greater range of characteristic impedances avail-able with the 11-layer process compared to the five-layer line.This divider was modeled in HFSS and then fabricated in thePolyStrata process. Fig. 9(b) and (c) shows the simulated andmeasured results of this divider. Due to a process error that hassubsequently been identified and corrected, the top layer, i.e.,11, was not completely in intimate contact with the rest of thedevice. This seam was filled using conductive epoxy. Despitethis problem, the transmission coefficient in the 11-layerdivider is approximately 0.4 dB better than the five-layer di-viders due to the larger cross section of the transmission lines.

TABLE III11-LAYER 50-32-� WILKINSON PARAMETERS

Fig. 11. Electric field distribution in 12.5-� lines in 11-layer configuration forinner conductor height of (left) 700 �m and (right) 600 �m.

V. DISCUSSION

The implemented divider networks were all designed for50- ports. The intended application for this component isa miniaturized power combining broadband amplifier. Sinceactive devices have low input and output impedances, it isadvantageous to have the Wilkinson divider also performpartial impedance matching to the active device. Therefore,a broadband Wilkinson divider was designed in the 11-layerPolyStrata configuration with an input port impedance of 50and two 32- output ports. The circuit schematic is the sameas that of the 11-layer divider in Section IV-B and shown inFig. 10. The characteristic impedances and lumped resistorvalues for this type of design are given in Table III.

Another divider with low input and output impedances wasdesigned for 12.5- ports in the 11-layer PolyStrata process.The characteristic impedances selected for this divider are basedon an optimization process and range from 12.5 to 22 , andthe resistor values are 25, 50, 100, and 100 . The input portconsists of one section that is 12.5 , unlike the other dividersin this paper. Since the characteristic impedances necessary forthis divider are relatively low, the inner conductor height shownin Fig. 9(a) is increased to 700 m (layers 3–9). This can cause

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2788 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 57, NO. 11, NOVEMBER 2009

Fig. 12. Rendering of the miniaturized 12–12-� Wilkinson divider imple-mented for the 11-layer fabrication process. Release holes and straps are notshown in this figure. The outer conductor is rendered with 50% transparency toshow the inside of the divider.

problems in the design of the passive sockets since the electricfield distribution is highly concentrated in the top (layer 10)and bottom (layer 2) gaps between the inner and outer con-ductors. In order to reduce the exposure of the electric fieldto the open areas of the passive sockets, the inner conductoris designed to be on layers 3–8 (i.e., vertically offset from thecenter), leaving a gap of 150 m on the top between the innerand outer conductors. This causes the electric field distribu-tion to be concentrated in layer 2, and less concentrated be-tween layers 9 and 10, making the structure suitable for imple-menting the sockets. Fig. 11 shows the electric field distributionfor the two 12.5- lines implemented in the 11-layer configura-tion. This divider exhibits 2–22-GHz bandwidth, and its overalllength after miniaturization is 14.5 mm. Fig. 12 shows a 3-Drendering of the 12.5–2.5- divider without release holes anddielectric straps.

VI. CONCLUSION

In summary, we demonstrated 11 : 1 bandwidth Wilkinson di-viders implemented in micro-coaxial lines. Isolation betweenthe output ports is better than 11 dB for 2–22 GHz, and the re-turn loss is better than 13 dB over that range. Insertion loss variesfrom less than 0.2 dB at 4 GHz to 0.7 dB at 18 GHz due to theincrease from the skin-effect loss in the copper. Several coaxialgeometries for different power-handling capabilities were con-sidered. The power loss through the resistors was determinedfor 20% mismatch in the magnitude of the load impedanceswith at most 3.2-mW power dissipation in a 200- resistor for1 W of input power. The PolyStrata process enables hybrid in-tegration of standard surface mount components, which can beextended to active devices for a PolyStrata power combined am-plifier. Alternatively, it is possible to detach the coaxial copperstructure from the silicon substrate and integrate it with circuitsmade in different technologies.

ACKNOWLEDGMENT

The authors would like to thank the Nuvotronics Microfabri-cation Team, Blacksburg, VA, and Prof. D. Filipovic, Universityof Colorado at Boulder.

REFERENCES

[1] E. Wilkinson, “An � -way hybrid power divider,” IRE Trans. Microw.Theory Tech., vol. MTT-8, no. 1, pp. 116–118, Jan. 1960.

[2] J. R. Blodgett, “Power combiner/splitter” U.S. Patent 5430418,Jul. 4, 1995. [Online]. Available: http://www.google.com/patents?hl=en&lr=&vid=USPAT5430418&id=T4EeAAAAEBAJ&oi=fnd

[3] S. B. Cohn, “A class of broadband three-port TEM-mode hybrids,”IEEE Trans. Microw. Theory Tech., vol. MTT-19, no. 2, pp. 110–116,Feb. 1968.

[4] G. S. Makineni and W. T. Joines, “Comparison of the broadband per-formance of two-way power dividers and combiners,” Microw. Opt.Technol. Lett., vol. 17, no. 1, pp. 29–37, 1998.

[5] R. B. Ekinge, “A new method of synthesizing matched broad-bandTEM-mode three-ports,” IEEE Trans. Microw. Theory Tech., vol.MTT-19, no. 1, pp. 81–88, Jan. 1971.

[6] H. Oraizi and A. Sharifi, “Design and optimization of broadband asym-metrical multisection wilkinson power divider,” IEEE Trans. Microw.Theory Tech., vol. 54, no. 5, pp. 2220–2231, May 2006.

[7] Z. Popovic, S. Rondineau, D. S. Filipovic, D. Sherrer, C. Nichols,J. Rollin, and K. Vanhille, “An enabling new 3-D architecture formicrowave components and systems,” Microw. J., vol. 51, no. 2, pp.66–87, Feb. 2008.

[8] K. J. Vanhille, D. L. Fontaine, C. Nichols, D. S. Filipovic, and Z.Popovic, “Quasi-planar high-� millimeter-wave resonators,” IEEETrans. Microw. Theory Tech., vol. 54, no. 6, pp. 2439–2446, Jun. 2006.

[9] K. Vanhille, D. S. Filipovic, C. Nichols, D. Fontaine, W. Wilkins, E.Daniel, and Z. Popovic, “Balanced low-loss ��-band �-coaxial hy-brids,” in IEEE MTT-S Int. Microw. Symp. Dig., 2007, pp. 1157–1160.

[10] M. Lukic and D. Filipovic, “Surface-micromachined dual ��-bandcavity backed patch antenna,” IEEE Trans. Antennas Propag., vol. 55,no. 7, pp. 2107–2110, Jul. 2007.

[11] Y. Saito, D. Fontain, J. M. Rollin, and D. Filipovic, “Monolithic micro-coaxial power dividers,” Electron. Lett., vol. 45, no. 9, pp. 469–470,2009.

[12] D. Filipovic, M. Lukic, Y. Lee, and D. Fontaine, “Monolithic rectan-gular coaxial lines and resonators with embedded dielectric support,”IEEE Microw. Wireless Compon. Lett., vol. 18, no. 11, pp. 740–742,Nov. 2008.

[13] K. Vanhille, “Design and characterization of microfabricated three-dimensional millimeter-wave components,” Ph.D. dissertation, Dept.Elect. Eng., Univ. Colorado at Boulder, Boulder, CO, 2007.

[14] M. Lukic, S. Rondineau, Z. Popovic, and D. Filipovic, “Modeling of re-alistic rectangular�-coaxial lines,” IEEE Trans. Microw. Theory Tech.,vol. 54, no. 5, pp. 2068–2076, May 2006.

[15] N. Ehsan, E. Cullen, K. Vanhille, D. Frey, S. Rondineau, R. Actis,S. Jessup, R. Lender, Jr., A. Immorlica, D. Nair, D. Filipovic, and Z.Popovic, “Micro-coaxial lines for active hybrid-monolithic circuits,”presented at the IEEE MTT-S Int. Microw. Symp., 2009.

[16] Ansoft Designer. Ansoft Corporation, Pittsburgh, PA, 2006. [Online].Available: http: //www.ansoft.com/products/hf/ansoft_designer/

[17] High Frequency Structure Simulator (HFSS). Ansoft Corporation,Pittsburgh, PA, 2007. [Online]. Available: http://ansoft.com/hfss/

[18] Y. Saito and D. Filipovic, “Analysis and design of monolithic rect-angular coaxial lines for minimum coupling,” IEEE Trans. Microw.Theory Tech., vol. 55, no. 12, pp. 2521–2530, Dec. 2007.

Negar Ehsan (S’05) was born in Tehran, Iran. Shereceived the B.S. and M.S. degrees in electrical andcomputer engineering (with a minor in applied math-ematics) from the University of Colorado at Boulder,in 2006, and is currently working toward the Ph.D.degree in electrical engineering at the University ofColorado at Boulder.

Her research interests include the of designingpassive and active broadband microwave circuits,advanced full-wave EM modeling techniques,miniaturization of microwave circuits, and mil-

limeter-wave antenna arrays.Ms. Ehsan was the recipient of the 2006 Distinguished Senior Award pre-

sented by the Department of Electrical Engineering, University of Colorado atBoulder.

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EHSAN et al.: BROADBAND MICRO-COAXIAL WILKINSON DIVIDERS 2789

Kenneth Vanhille (S’00–M’07) received the B.S.degree in electrical engineering from Utah StateUniversity, Logan, in 2002, and the M.S. and Ph.D.degrees in electrical engineering from the Universityof Colorado at Boulder, in 2005 and 2007, respec-tively.

He is currently a Senior Engineer and ProgramManager with Nuvotronics LLC, Blacksburg, VA.His technical interests include high-frequency pack-aging techniques, millimeter-wave components andsystems, and antenna design.

Sébastien Rondineau (M’04) received the Ph.D.degree from the University of Rennes 1, Rennes,France, in 2002.

From 2002 to 2006, he was a Research Fellow withthe University of Colorado at Boulder. From 2006to 2008, he was a Research Assistant Professor withthe Microwave and Active Antenna Laboratory, Uni-versity of Colorado at Boulder, and is currently Di-rector of the Research and Development Department,TSM Antennas, Santa Maria, Brazil. His research in-terests include the method of analytical regulariza-

tion in computational electromagnetics, mode matching, conformal mapping,microscale interconnects, Butler matrices, propagation and scattering of waves,homogeneous and inhomogeneous dielectric lenses, discrete lens arrays and an-tennas, dispersion controlled 2-D Rotman lenses, nonlinear polymers applied torectification-modulation for lasers, terahertz and microwave signals, and SigInt/ComInt in electronic warfare.

Evan D. Cullens (S’04) received the B.S. degree inelectrical engineering from Kansas State University(KSU), Manhattan, in 2006, the M.S.E.E. degreefrom the University of Colorado at Boulder, in 2008,and is currently working toward the Ph.D. degree atthe University of Colorado at Boulder.

While with KSU, he was a McNair Scholar andthe Supervisor of the nuclear reactor. He also per-formed research for the KSU Wireless Hardware De-sign Group on a UHF proximity microtransceiver forinteroperable Mars communications. His current re-

search involves hybrid integration techniques of active and passive componentsinto micro-coaxial lines.

Zoya B. Popovic (F’02) received the Dipl.Ing. degreefrom the University of Belgrade, Serbia, Yugoslavia,in 1985, and the Ph.D. degree from the California In-stitute of Technology, Pasadena, in 1990.

Since 1990, she has been with the Universityof Colorado at Boulder, where she is currently theHudson Moore Jr. Chaired Professor of Electricaland Computer Engineering. In 2001, she was aVisiting Professor with the Technical Universityof Munich. Her research interests include high-ef-ficiency, low-noise and broadband microwave

and millimeter-wave circuits, quasi-optical millimeter-wave techniques forimaging, smart and multibeam antenna arrays, intelligent RF front ends, RFoptics, and wireless powering for batteryless sensors.

Dr. Popovic was the recipient of the 1993 and 2006 Microwave Prizes pre-sented by the IEEE Microwave Theory and Techniques Society (IEEE MTT-S)for best journal paper. She was the recipient of the 1996 URSI Issac Koga GoldMedal, and the Humboldt Research Award for Senior U.S. Scientists from theGerman Alexander von Humboldt Stiftung. She was also the recipient of the2001 HP/ASEE Terman Medal for combined teaching and research excellence.

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