with high selectivity by adding an inductive...

> REPLACE THIS LINE WITH YOUR PAPER IDENTIFICATION NUMBER (DOUBLE-CLICK HERE TO EDIT) <

1

Abstract— A Frequency Selective Surfaces (FSS) is essentially a

spatial filter. Compared to a connectorized traditional filter with

fixed ports, an FSS can separate signals at different frequencies by

both transmission and reflection. In this paper, we propose an

approach to design FSSs with high selectivity. This is achieved by

adding an inductive surface on the other side of a resonant surface.

The proposed FSS will have both a passband and a stopband.

Such a response has two advantages. One is that by choosing the

frequencies properly, the stopband can improve the selectivity of

the passband significantly, and vice versa. Another advantage is

that the proposed FSS can operate effectively as a diplexer by

separating signals at the passband and stopband.

The center frequency and bandwidth of the passband and

stopband can be independently designed. The proposed FSS has a

small element size, low profile and insensitive response to

surrounding dielectric materials. The proposed design method has

many potential applications. An FSS has been fabricated and

tested. Experimental results have validated the proposed theory

and design method.

Index Terms: Frequency selective surface (FSS), periodic

structures, artificial magnetic conductor (AMC), filter, diplexer

I. INTRODUCTION

requency selective surface (FSSs) are usually constructed by

patches, slots or arbitrary geometrical structures within a

metallic screen supported by dielectric materials.

Characteristics such as the array element type and shape, the

presence or absence of dielectric slabs, the dielectric substrate

parameters and inter-element spacing will determine the

response of the FSS, such as its transfer function, bandwidth,

and its dependence on the angular wave (incident and

polarization angles). Early researches with respect to FSSs

using an infinite planar array [1], a two-layer dipole array [2] or

metallic mesh structures [3] focus on the theoretical calculation

of scattering or reflection properties when plane

electromagnetic waves are incident on the structures. Currently,

FSSs have been the subject of intense investigations for a wide

range of applications for decades, such as bandpass filters [4-7]

and high impedance electromagnetic surfaces [8].

An artificial magnetic conductor (AMC) or high impedance

surface (HIS) consists of an FSS placed above a perfect electric

conductor (PEC) ground plane, with a dielectric material in

between [9]. It displays a 0o reflection coefficient phase at a

given frequency [8]. An AMC can be applied to control the

propagation of electromagnetic waves in certain patterns. These

structures can improve the performance of an antenna [8]. It

exhibits selectivity in suppressing surface waves to improve the

gain of antennas. A U-slot patch antenna integrated to a

modified Jerusalem cross FSS was proposed in [10] to improve

the antenna gain, bandwidth and return loss at 2.45 and 5.8 GHz

for Bluetooth and WLAN applications. A split-ring shaped slot

Manuscript received 27 January 2019. The authors are with the Department

of Electrical Engineering and Electronics, The University of Liverpool, UK. (e-

mail:[email protected]).

antenna was integrated to an FSS to enhance the bandwidth and

gain of the antenna in [11].

In general, good selectivity characteristics of FSSs refer to

the rapid decline of the sideband of the transmission

coefficients. Many designs have been reported on the

optimization of selectivity of FSSs [12-17]. Such designs

include the use of substrate integrated waveguide cavities [12],

two layers of metallic unit cells [13], a periodic array of

gridded-triple square loop [14], an identical tripole resonators

[15] and loop-shaped resonators [16]. The selectivity of FSSs

can be enhanced by cascading substrate integrated waveguide

cavities [12]. The coupling between capacitive patches in

different layers [13] in the FSS can induce a good selection

feature. A quad-band FSS [14] can be obtained by cascading

three metallic layers to achieve high selectivity. The aperture-

coupled resonator structure [15] consisting of identical tripole

resonators also exhibits high selectivity.

The target of this study is to propose a low-profile FSS design

that shows high selectivity. This is achieved by adding an

inductive layer on the other side of a resonant layer. By doing

so, both a passband and a stopband can be generated. The two

bands can improve the selectivity of each other. Another

advantage is that the performance of the FSS is very stable

when it is attached to other dielectric material.

An equivalent circuit is derived to analyzed the proposed

design. With the equivalent circuit, it was demonstrated that the

center frequency and the bandwidth of the passband and the

stopband can be synthesized independently. This will provide

great flexibility for the design. The FSS also have a small

element size and a low profile.

The proposed FSS design has many potential applications.

The passband and stopband can be designed so that the FSS can

operate as a diplexer to separate signals at these two bands. This

is because, for an FSS, both the reflected and the transmitted

signals can be utilized. While in a traditional filter, usually only

the transmitted signal is utilizable. The insensitivity to

surrounding dielectric materials suggests that FSS can be

potentially used as part of an AMC to improve the gain of RFID

tag antennas. Such applications will be exploited as future work.

In this paper, Section II discusses the equivalent circuit of an

FSS. Section III describes the procedure to design the proposed

FSS. The experimental setup and measurement results to verify

the proposed design method are provided in Section IV.

Conclusions and future work are finally described in Section V.

II. EQUIVALENT CIRCUIT OF AN FSS

This section discusses the equivalent circuit of an FSS. In

general, at least for mechanical and miniaturization reasons,

most FSSs are supported by dielectric slabs. The dielectric

materials have strong effect on the bandwidth, resonant

Frequency Selective Surface with High Selectivity by

Adding an Inductive Layer Muaad Hussein, Zhenghua Tang, Yi Huang and Jiafeng Zhou

F


2

frequency and element size. FSSs can be classified to two types

based on the equivalent circuit: capacitive surfaces and

inductive surfaces. It is well known that the equivalent circuit

for a patch is a shunt capacitor (Cp), and for a mesh is a shunt

inductor (Li) [3].

For a mesh-type FSS layer, when it is attached to an infinitely

wide dielectric material with dielectric constant εr, effect of the

dielectric material on the FSS can be explained by using the

equivalent circuits of the FSS and the dielectric slab. The

approximate value of the intrinsic inductance can be calculated

using the formula [18]

𝐿 = 𝜇𝑜𝜇𝑒 (𝑃 2𝜋)⁄ ln csc (𝜋𝑤 2𝑃⁄ ) (1)

where P is the strip length and w is the strip width of the meshes.

𝜇𝑜 and 𝜇𝑒 denote the permeability of free space and the

effective relative permeability of the structure, respectively.

(a)

(b)

Fig. 1. Equivalent circuit of an FSS (a) without the inductive layer (b) with

the inductive layer and also an extra layer of dielectric layer.

The dielectric slab can be represented by a short piece of

transmission lines h1 as shown in Fig. 1. The characteristic

impedance of the transmission line is 𝑍𝑑 = 𝑍0 √𝜀𝑟⁄ , where Z0 =

377 Ω is the free space impedance, and 𝜀𝑟 is the effective

relative permittivity of the structure.

For a capacitive patch-type FSS, when it is attached an

infinitely wide dielectric slab, the equivalent circuit can be

obtained using a procedure similar to the one for an inductive

FSS. The approximate value of the corresponding equivalent

capacitance Cp can be calculated by [18]:

𝐶𝑝 = 𝜀0𝜀𝑒 (2𝑃 𝜋)⁄ ln csc (𝜋𝑔 2𝑃⁄ ) (2)

The capacitance is determined by the patch length P, the gap

g between adjacent patches and the effective dielectric constant

ɛe of the structure.

From (1), it can be noticed that the inductance value of the

meshes of the FSS is mainly determined by its strip width. An

increase in strip width of the FSS will result in the inductance

to be reduced. Similarly, a decrease in the width of the meshes

will lead to an increase of the inductance. The changes of the

inductive value of the FSS will significantly affecting its

characteristic impedance. This will influence the S-parameters

of the FSS when electromagnetic waves are incident on the FSS

It can also be observed from above that, when an FSS is

attached to a dielectric material, the change in the bandwidth

and resonant frequency can be attributed to two reasons. Firstly

the dielectric material directly changes the effective dielectric

constant of the structure and thereby the intrinsic capacitance.

Secondly, it is because the equivalent circuit for the FSS will be

changed by adding a series transmission line with a

characteristic impedance of Zd, as shown in Fig. 1(a).

(a) (b)

Fig. 2. Structures of (a) the resonant surface and (b) the inductive surface.

-40

-20

0

-40

-20

0

S p

aram

eter

s (d

B)

S21

S11

(a)

(b)

8.06.04.02.0

S11

S21

S p

aram

eter

s (d

B)

Frequency(GHz)

0.0

Fig. 3. The calculated S-parameters of (a) the original FSS (resonant surface)

without the inductive layer and (B) the proposed FSS with the inductive layer.

III. FSS CIRCUIT DESIGN

The choice of elements is of great importance when

designing an FSS. Some structures are inherently narrowband

or wideband. Different FSS types can be chosen based on the

application requirements. These requirements usually include a

level of dependence on the polarization and incidence angle of

incoming waves and bandwidth. In general, FSS structures with

a small element size, compared to the wavelength, will be in

insensitive to incident angles [19]. In this paper, a new approach


3

is proposed to implement FSS structures with miniaturized

elements and high selectivity.

The proposed FSS element is realized by using a resonant

surface (the top side) and an inductive surface (the bottom side),

as shown in Fig. 2. They are separated by a dielectric substrate

with a thickness of d. The equivalent circuit of the proposed

element is shown in Fig. 1(b). As shown in Fig. 2(a), the top

layer of the proposed FSS is a resonant surface consisting of an

inductive mesh and four capacitive patches as introduced in [19,

20]. The main physical features of the resonant surface in Fig.

2(a) can be described by the circuit model of Fig. 1. The

corresponding equivalent component values for the resonant

surface can be approximated based on the circuit theory in [21].

In the equivalent circuit, C2 represents the capacitance between

adjacent patches in one element, L1 is the inductance of the outer

square ring and C1 is the mutual capacitance between adjacent

elements.

-28

-21

-14

-7

0

-60

-40

-20

0

S11(d

B) L

2=2.5 nH

L2=10 nH

L2=30 nH

L2=50 nH(a)

(b)

L2=2.5 nH

L2=10 nH

L2=30 nH

L2=50 nH

S21(d

B)

Frequency(GHz)0 2 4 6 8

Fig. 4. The calculated (a) S11 and (b) S21 of the proposed FSS with the inductive

layer of different inductances L2.

The values of lumped-element components in Fig. 1 are

approximately C1=0.25, C2=0.27 pF, L1= 2.4 nH and L2 = 2.5

nH. The calculated reflection and transmission coefficients of

the equivalent circuit are shown in Fig. 3. It can be observed

that the resonant surface has a passband at 5.7 GHz and

stopband at 4.5 GHz. By adding the inductive layer, the

inductance L2 is introduced into the equivalent circuit as shown

in Fig. 1(b). The passband frequency is shifted to 3.46 GHz and

the resonant frequency of the stopband remains unchanged as

shown in Fig. 3. The corresponding relative bandwidth of the

10-dB stopband at 4.5 GHz is widened from 18.5% to 31.2%

while the passband bandwidth is significantly narrowed with

the effect the inductance L2.

In order to observe the influence of the inductance L2 on S-

parameters more clearly, the S-parameters for the equivalent

circuit of the proposed FSS is calculated and illustrated as in

Fig. 4 with different values of L2. When the inductance L2 is

increased from 2.5 nH to 50 nH, the resonant frequency is

shifted form 3.46 GHz to 1.2 GHz. The relative bandwidth is

widened from 0 to 60.4%. When the passband and the stopband

frequencies are close to each other, the FSS will achieve good

selectivity.

The resonant surface, with the equivalent circuit shown in

Fig. 1(a), has a bandpass response. This is determined by the

equivalent circuit of L1 in parallel with C2. The impedance of

the parallel circuit at frequency ω can be calculated by:

𝑍𝑝 = 𝑗𝜔𝐿1/(1 − (𝜔

𝜔0𝑃)

2

) (3)

where

𝜔0𝑝 = 1/√𝐿1𝐶2 (4)

is the resonant angular frequency of the parallel resonator. By

connecting this LC parallel circuit with C1 in series, it can easily

calculate that the circuit will have both a stopband and a

passband. It has a passband because at around the resonant

frequency 𝜔0𝑝 , the parallel LC circuit has an impedance of

infinity. The total impedance of C1 in series with the LC circuit

(L1//C2) is still close to infinity. Therefore the passband is

maintained. At frequencies lower than the passband, the

impedance of the parallel LC circuit, as can be calculated from

(3), is inductive. This effective inductance in series with C1 will

form a series resonator, which will realize a stopband.

The proposed FSS can potentially be used as an AMC to

improve the performance of antennas. This will be exploited in

the future work. For such applications, the FSS will usually be

attached to other material. For example, it can be used as RFID

tags for IoT applications. If the FSS is attached to a dielectric

material with a thickness of h2, represented by Zex as shown in

Fig. 1(b), the response of the FSS will be affected. This extra

layer can be considered as a transmission line in a similar way

to Zd. The equivalent circuit model of the extra layer is

composed of a series inductor (Lt) and a shunt capacitor (Ct),

which can be obtained using the Telegrapher’s model for TEM

transmission lines [22].

In the proposed design, an inductive mesh layer, as shown in

Fig. 2(b), is added on the other side of the substrate of Zd. Since

the stopband of the resonant FSS is mainly determined by L1,

C1 and C2 as shown in Fig. 1(a), the extra dielectric layer Zex

and the inductive layer L2 have little effect on the stopband

frequency. It is because the effective impedance of the three

components L1, C1 and C2 is still close to zero at the stopband

frequency, with or without L2 and Zex.

This added inductive layer can improve the performance of

the FSS on three aspects. Firstly, the added parallel inductance

will significantly reduce the passband frequency of the FSS.

This will effectively reduce the element size of the FSS

element, making it insensitive to incident angles. In the original

circuit as shown in Fig. 1(a), there is a passband because the

circuit of L1, C1 and C2 has a high input impedance of infinity

at the resonant frequency ω0p as defined in (4). By adding the

inductive layer, with the influence of L2, the input impedance is

no longer high. As indicated in (3), the parallel circuit of L1//C2

is inductive at frequencies lower than frequency ω0p. This

effective impedance combined with L2 and C1 will resonate at a

lower frequency, which will produce a passband. The passband

frequency is much lower than that of the original FSS.

Effectively, the element size is reduced very significantly.

Secondly, from the analysis above, it is obvious the stopband

and the passband of the proposed FSS can be independently

synthesized. The stopband is mainly determined of L1, C1 and

C2. The passband can be tuned by changing the value of L2. The

corresponding bandwidths can also be independently designed


4

by choosing the values of the four components properly. If the

passband and the stopband are close to each other, the

selectivity of the passband (or the stopband) can be significantly

improved. Since both the transmitted wave and the reflected

wave can be fed to the next stage, the FSS can potentially be

used as a diplexer.

Thirdly, the inductance of the inductive layer will have a

much stronger effect on the input impedance of the FSS than

the extra layer of Zex as shown in Fig. 1(b). Thus, the extra

transmission line has very little effect on the passband

frequency. Furthermore, the response of mesh layer itself, being

inductive surface, is not significantly influenced by surrounding

dielectric materials. As a result, the overall performance of the

FSS will be very insensitive to nearby dielectric materials. This

feature is very useful for antenna applications when the FSS is

attached to other materials.

To demonstrate the proposed design method, a prototype

FSS was designed on a 1.6 mm thick FR4 substrate with a

dielectric constant of 4.3. The geometry parameters after

optimization are shown in Table I.

TABLE I

GEOMETRY PARAMETERS OF THE PROPOSED RESONANT

SURFACE (unit:mm)

Parameter Dx Dy h Wl Ws

Value 6 6 1.6 0.2 0.2

Parameter Wi Wp εr s b

Value 0.2 2.3 4.3 0.1 0.2

-40

-30

-20

-10

0

0 2 4 6 8 10

-40

-30

-20

-10

0

Frequency(GHz)

S p

aram

eter

s (d

B)

S21

S11(a)

S p

aram

eter

s (d

B)

S21

S11

(b)

Fig. 5. The simulated S-parameters of (a) the original FSS without the inductive

layer and (b) the proposed FSS with the inductive layer.

The S-parameters for the structure in the absence and presence

of the inductive surface are simulated by CST. The results are

illustrated in Fig. 5 (a) and (b), respectively. The passband

frequency at 5.8 GHz of the original resonant surface is shifted

own to 3.5 GHz when the inductive layer is added on the other

side of the resonant surface. The corresponding bandwidth of

the passband is changed as well, from 23.5% to 29.4%.

The proposed FSS with 3×3 array elements is shown in Fig.

6. To demonstrate the insensitivity of the proposed FSS to

surrounding materials, the structure is simulated in four cases:

the inductive layer is not attached to any dielectric material,

attached to a 0.25 mm thick slab of Roger 3035 (RO3035) with

ɛr = 3.5, Roger 3006 (RO3006) with ɛr = 6.15 and Roger 3210

(RO3210) slab with ɛr = 10.2. The simulated transmission

coefficients over the frequency range from 0 to 10 GHz for the

proposed FSS in the four cases mentioned above are shown in

Fig. 7. When the dielectric constant of the additional dielectric

materials is increased from 1.0 (no extra dielectric material) to

10.2 (RO3210), the four curves as illustrated in Fig. 7 are almost

completely overlapping. This is because the change of the

dielectric constant has little effect on the response of the

proposed FSS. In order to observe the response more clearly,

the transmission coefficient at the passband and the stopband

are enlarged and embedded into Fig. 7. The stopband frequency

is shifted from 4.5 GHz to 4.47 GHz when the dielectric

constant of the attached material is increased from 1 to 10.2.

The resonant frequency shift is less than 0.7%. The passband

frequency changes no more than this.

Fig. 6. Proposed FSS with 3×3 array elements.

0 1 2 3 4 5

-40

-30

-20

-10

0

4.4 4.5 4.6

-34

-32

-30

3.3 3.4 3.5 3.6

-3.2

-2.8

-2.4

-2.0

S21(d

B)

Frequency(GHz)

Not attached

Slab with r=3.5

Slab with r=6.15

Slab with r=10.2

Fig. 7. Simulated transmission coefficients of the proposed structure when

placed on 0.25 mm thick Roger slabs with dielectric constants ɛr = 3.5, 6.15

and 10.2, respectively. The stopband and passband details are enlarged and embedded in the figure.


5

To further observe the effect of the inductive layer on the

response of the FSS, the S-parameters with inductive strip

widths of Wi = 0.03, 0.1, 0.2 and 1.0 mm are simulated and

shown in Fig. 8. As the strip width increases, the relative

bandwidth of the passband decreases significantly and the

passband edges become steeper. This indicates that the

selectivity of the FSS can be enhanced by choosing the value of

L2. If any higher inductance value is desired, a thinner strip

width or a meander line structure can be used.

-12

-8

-4

0

0 2 4 6 8

-60

-40

-20

0

S11(d

B)

Wi=0.03 mm

Wi=0.1 mm

Wi=0.2 mm

Wi=1.0 mm

(a)

Wi=0.03 mm

Wi=0.1 mm

Wi=0.2 mm

Wi=1.0 mm

Frequency(GHz)

S21(d

B)

(b)

Fig. 8. Simulated transmission coefficients of the proposed FSS with

different inductive strip widths of Wi=0.03 mm, 0.1 mm, 0.2 mm and 1.0 mm.

(a)

(b)

Fig. 9. A photograph of the fabricated prototype of the proposed FSS (a) the

resonant surface and (b) the inductive surface.

IV. EXPERIMENTAL RESULTS

An FSS consisting of 30 × 30 elements is fabricated on a 180

mm × 180 mm FR-4 substrate as shown in Fig. 9. Dimensions

of the structure are summarized in Table I. The resonant surface

(top side) is shown in Fig. 9(a), while the inductive surface

(bottom side) is illustrated in Fig. 9(b). A vector network

analyzer and two horn antennas are used for the measurement.

The transmission coefficient between the two horn antennas

was measured without the FSS, and then measured with the FSS

between the antennas. Then, the measured transmission with

the FSS is normalized with respect to the measured data without

the FSS.

The prototype FSS is tested when it is attached to a 0.25 mm

thick slab of Roger 3035 substrate (εr = 3.5), Roger 3006 (εr =

6.15) and Roger 3210 substrate (εr = 10.2). The experimental

setup in an anechoic chamber is shown in Fig. 10. To compare

the measured and simulated S21 more clearly, the curves of the

measured S11 is shifted down by 0.279 GHz and shown in Fig.

11. It can be seen that the resonant frequency of the proposed

FSS is very stable when the FSS is placed on different dielectric

materials. The structure exhibits a passband at 3.5 GHz and a

stopband at 4.5 GHz as designed.

Fig. 10. Experimental setup in an anechoic chamber.

-30

-20

-10

0

3.52.5 4.54.03.0

S2

1(d

B)

Frequency(GHz)

Not attached, simulated

Not attached, measured

Slab with r=3.5, simulated

Slab with r=3.5, measured





2.0

Fig. 11. Simulated and measured transmission coefficients of the proposed

FSS when placed on a 0.25 mm thick dielectric slab with various dielectric

constants (εr=3.5, 6.15, 10.2).


6

The FSS was also tested under various polarization angles,

and the performance was also very stable (not shown here). The

response of the FSS is the same for vertical and horizontal

polarizations due to the symmetrical nature of the proposed

element.

V. CONCLUSION AND FUTURE WORK

In this paper, a study on the effect of an additional inductive

surface on selectivity of frequency selective surfaces has been

carried out. It has been demonstrated that, by combining a

resonant surface and an inductive surface, the FSS can generate

a passband and a stopband. The stopband performance is

mainly determined by the resonant surface. The inductive layer

can significantly lower the passband frequency, which

effectively reduces the size of the FSS element. The passband

response can be tuned by changing parameters of the inductive

layer only. Thus the passband and the stopband can be

independently synthesized.

The selectivity of the FSS can be improved by controlling the

passband and stopband frequency and bandwidth. Theoretical

and experimental results show that the performance of the FSS

has been significantly improved by adding the inductive layer.

The proposed FSS can be useful for many potential

applications, which will be exploited in future work. For

example, the proposed FSS can be used as a diplexer to separate

signals at different frequency bands, or as a artificial magnetic

conductor for antenna applications, to enable them achieving

consistent performance when attached to various materials with

arbitrary thicknesses.

REFERENCES

[1] R. Ott, R. Kouyoumjian, and L. Peters, “Scattering by a Two‐

Dimensional Periodic Array of Narrow Plate,” Radio Science, vol. 2, no. 11, pp. 1347-1359, Nov. 1967.

[2] B. Munk, and R. Luebbers, “Reflection properties of two-layer

dipole arrays,” IEEE Transactions on Antennas and Propagation, vol. 22, no. 6, pp. 766-773, Nov. 1974.

[3] R. Ulrich, “Far-Infrared Properties of Metallic Mesh and Its

Complementary Structure,” Infrared Physics, vol. 7, no. 1, pp. 37-&, 1967.

[4] C. Winnewisser, F. Lewen, and H. Helm, “Transmission characteristics of dichroic filters measured by THz time-domain

spectroscopy,” Applied Physics a-Materials Science & Processing,

vol. 66, no. 6, pp. 593-598, Jun. 1998. [5] S. Govindaswamy, J. East, F. Terry, E. Topsakal, J. L. Volakis, and

G. I. Haddad, “Frequency-selective surface based bandpass filters in

the near-infrared region,” Microwave and Optical Technology Letters, vol. 41, no. 4, pp. 266-269, May 20, 2004.

[6] F. Bayatpur, and K. Sarabandi, “Single-layer high-order

miniaturized-element frequency-selective surfaces,” IEEE Transactions on Microwave Theory and Techniques, vol. 56, no. 4,

pp. 774-781, Apr. 2008.

[7] B. Schoenlinner, A. Abbaspour-Tamijani, L. C. Kempel, and G. M. Rebeiz, “Switchable low-loss RF MEMS Ka-band frequency-

selective surface,” IEEE Transactions on Microwave Theory and

Techniques, vol. 52, no. 11, pp. 2474-2481, Nov. 2004. [8] D. Sievenpiper, L. J. Zhang, R. F. J. Broas, N. G. Alexopolous, and

E. Yablonovitch, “High-impedance electromagnetic surfaces with a

forbidden frequency band,” IEEE Transactions on Microwave Theory and Techniques, vol. 47, no. 11, pp. 2059-2074, Nov. 1999.

[9] D. J. Kern, D. H. Werner, A. Monorchio, L. Lanuzza, and M. J.

Wilhelm, “The design synthesis of multiband artificial magnetic conductors using high impedance frequency selective surfaces,”

IEEE Transactions on Antennas and Propagation, vol. 53, no. 1, pp.

8-17, Jan. 2005.

[10] H.-Y. Chen, and Y. Tao, “Performance improvement of a U-slot

patch antenna using a dual-band frequency selective surface with

modified Jerusalem cross elements,” IEEE Transactions on antennas and Propagation, vol. 59, no. 9, pp. 3482-3486, 2011.

[11] B. Mandal, A. Chatterjee, and S. K. Parui, "A wearable button

antenna with FSS superstrate for WLAN health care applications." RF and Wireless Technologies for Biomedical and Healthcare

Applications (IMWS-Bio), IEEE MTT-S International Microwave

Workshop Series on pp. 1-3. 2014. [12] G.-Q. Luo, W. Hong, H.- J. Tang, and K. Wu, “High Performance

Frequency Selective Surface Using Cascading Substrate Integrated

Waveguide Cavities, ” IEEE Microwave and Wireless Components Letters, vol. 16, no 12, Dec. 2006.

[13] F. Deng, X.-Q. Yi, and W.-J. Wu, “Design and Performance of a

Double-Layer Miniaturized-Element Frequency Selective Surface, ” IEEE Antennas and Wireless Propagation Letters, vol. 12, 721.

2013.

[14] M.-b. Yan, J.-F. Wang, H. Ma, S.-B. Qu, J.-Q. Zhang, C.-L.Xu, L.

Zheng, and A.-X Zhang, “A Quad-Band Frequency Selective

Surface With Highly Selective Characteristics, ” IEEE Antennas and Wireless Propagation Letters, vol. 26, no. 8, Aug. 2016.

[15] D.-S. Wang, B.-J. Chen, and C.-H Chan, “High-Selectivity

Bandpass Frequency-Selective Surface in Terahertz Band,” IEEE Transactions on Terahertz Science and Technology, vol. 6, no. 2,

Mar. 2016.

[16] M. Kartal, J. J. Golezani, and B. Doken, “A Triple Band Frequency Selective Surface Design for GSM Systems by Utilizing a Novel

Synthetic Resonator,” IEEE Transactions on Antennas and

Propagation, vol. 65. no.5, May 2017. [17] Y.-Q. Pang , Y.-F. Li, J.-Q. Zhang, Z. Xu, and S.-B. Qu, “Design of

Frequency Selective Surface Based on Spoof Surface Plasmon

Polariton Modes,” IEEE Antennas and Wireless Propagation Letters, vol. 17, no 6, June 2018.

[18] M. Al-Joumayly, and Nader Behdad, “A New Technique for Design

of Low-Profile, Second-Order, Bandpass Frequency Selective Surfaces, ” IEEE Transactions on Antennas and Propagation, vol.

57, no. 2, Feb. 2009.

[19] M. N. Hussein, J. Zhou, Y. Huang, M. Kod, and A. P. Sohrab, “A

Miniaturized Low-Profile Multilayer Frequency-Selective Surface

Insensitive to Surrounding Dielectric Materials,” IEEE

Transactions on Microwave Theory and Techniques, 2017. [20] M. Hussein, J. Zhou, Y. Huang, A. Sohrab, and M. Kod, "Frequency

selective surface with simple configuration stepped-impedance

elements." 10th European Conference on Antennas and Propagation (EuCAP) pp. 1-4, 2016.

[21] N. Marcuvitz, Waveguide handbook: IET, 1951.

[22] W. R. Eisenstadt, and Y. Eo, “S-Parameter-Based IC Interconnect Transmission Line Characterization,” IEEE Transactions on

Components, Hybrids, and Manufacturing Technology, vol. 15, no.

4, Aug. 1992.

with high selectivity by adding an inductive...

Documents