substrate integrated waveguide filters

Because of the inherent structural flexibility in coupling design and topologi-cal arrangement, substrate integrated waveguide (SIW) filter topologies enjoy better out-of-band frequency selectivity and/or in-band phase response with the allocation of finite transmission zeros (FTZs). In the first article in this series, basic design rules and fundamental electrical characteristics have been

presented that indicate the superior performances of SIW structures and their filter ap-plications. Advanced design techniques and innovative structure features have recently been reported in a large number of publications. They include cross couplings realized by physical and nonphysical paths and SIW filters with dual-mode or multimode techniques. Miniaturization-enabled techniques including low-temperature cofired ceramic (LTCC) technology have been developed and applied to the development of SIW filters to reduce the size for low-gigahertz applications using nontransverse electromagnetic (non-TEM) modes. Wideband SIW filters, multiband SIW filters, and reconfigurable SIW filters have also been reported by various research groups. This article reviews these advanced and innovative SIW filter technologies, and related examples are presented and discussed.

Xiao-Ping Chen and Ke Wu

Date of publication: 8 September 2014Digital Object Identifier 10.1109/MMM.2014.2332886

Xiao-Ping Chen and Ke Wu ([email protected]) are with the Poly-Grames Research Center, Department of Electrical Engineering, Ecole Polytechnique (University of Montreal), Center for Radiofrequency

Electronics Research of Quebec, Montreal, Quebec H3T 1J4, Canada.

Substrate Integrated Waveguide Filters

September/October 2014 1211527-3342/142014IEEE

image licensed by ingram publishing

122 September/October 2014

SIW Filters with FTZsFilters with FTZs located on the imaginary axis or sym-metrically on the real axis or in all four quadrants of the complex frequency plane are known to have bet-ter out-of-band frequency selectivity and/or in-band phase response. The FTZs on the imaginary axis of the complex frequency plane will lead to high selectivity performance and excellent stopband characteristics, while FTZs on the real axis or in all the four quadrants are used for achieving a linear phase response in the passband. There are two methods that have been used to produce FTZs. The first method makes use of cross couplings with nonphysical couplings by higher-order modes or physical coupling structures to produce mul-tiple paths for signal flow. If two different signal paths

yield the same magnitude but opposite phase, they would cancel each other out, and an FTZ on the imagi-nary axis is then produced. However, positive and nega-tive couplings must be simultaneously realized for the opposite phases. On the other hand, if two signal paths generate the same magnitude and phase, an FTZ on the real axis is then produced. In this case, all of the cou-plings can have the same sign.

The second method involves FTZs on the imagi-nary axis that can be extracted to realize bandstop resonators. In an extracted-pole filter, it is not neces-sary to simultaneously realize both types of coupling because the FTZ on the imaginary axis is produced by the bandstop resonator that can also produce a trans-mission pole in the passband. In addition, every FTZ can be tuned independently by changing the resonant frequency of the bandstop resonator.

SIW technology, a part of the family of substrate integrated circuits, is very suitable for the realization of the above-mentioned filter topologies due to its inherent flexibility for the implementation of multiple paths and phase correlations and controls for FTZs. A fourth-order, linear-phase filter with two FTZs that are symmetrically located along the real axis was pro-posed and realized by a single-layer printed circuit board (PCB) process based on an SIW platform in [1]. In this case, all of the direct coupling and cross cou-pling between the first SIW cavity and fourth SIW cav-ity were realized by a section of SIW evanescent mode for positive coupling. The variation of in-band group time delay of the filter was smaller than 0.5% over 50% of the passband around the center frequency. Multilay-ered topologies provide more freedom in the design of coupling paths and control mechanisms.

To obtain FTZs on both the imaginary axis and the real axis using only magnetic coupling, [2] presented a sixth-order Ka-band self-equalized pseudoellipti-cal SIW filter, the geometric configuration, structural topology, and frequency responses of which are all shown in Figure 1. Two bandstop SIW cavity reso-nators, 1 and 6, are responsible for the generation of two FTZs on the imaginary axis, while the cross cou-pling between SIW cavity resonators 2 and 5 are for two FTZs that are symmetrically located on the real axis. The filter was synthesized and designed using an extracted-pole technique [3] and was fabricated on a low-cost single-layer PCB substrate.

On the single-layer SIW platform, where only mag-netic iris coupling can be realized, it is a rather chal-lenging task to simultaneously design positive and negative coupling networks for the generation of FTZs on the imaginary axis. A novel structure using a bal-anced microstrip line with a pair of metalized via-holes placed between transverse electric (TE)101-mode-based SIW cavity resonators 1 and 4 to invert the phase of sig-nal was presented for the first time in [4]. A mixed cou-pling, including both positive and negative couplings,

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Figure 1. The (a) geometric configuration and (b) structural topology. (c) and (d) The frequency responses of a sixth-order self-equalized pseudoelliptical SIW filter [2].

September/October 2014 123

which cancel each other out, is produced. The structure can be optimized to ensure that the negative coupling is stronger than its positive counterpart, and a small amount of the negative coupling can be canceled out by tuning the width of the magnetic postwall iris. This structure is well suited for SIW implementation and provides accurate negative coupling with an inductive postwall iris. Figure 2 shows the geometric configura-tion, structural topology, and frequency response of a fourth-order, cross-coupled SIW filter with the negative coupling structure; coupling between the input and the output may be noted.

In [5] and [6], a balanced microstrip line with a pair of metalized via-holes was replaced by a section of grounded coplanar waveguide (CPW) for the realiza-tion of electrical coupling. To avoid a spurious reso-nance induced by the grounded CPW line that was too close to the passband, the coupling coefficient was controlled by the linewidth rather than the line length.

Another method to realize negative coupling on the single-layer SIW platform with only a magnetic iris is to use a higher-order resonance mode of the SIW cavity resonator [7][10], [13]. If the magnetic iris is properly located, the TE102 resonant mode can be used to obtain the 180 phase change, which is equivalent to the gen-eration of negative coupling in other signal paths with a TE101 resonant mode.

In [13], a Ka-band fourth-order boxlike filter with one transmission zero on the left or right side of the passband was presented. The TE102 mode resonance excited in one SIW cavity is used for the transforma-tion between 0 and 180 phases of coupling on a sin-gle-layer SIW, where only a magnetic postwall iris can be implemented for positive coupling. Figure 3 shows the geometric configuration, structural topology, and frequency responses of the proposed SIW filter. Note that the FTZ can move from one side to the other of the passband only when the self-coupling is changed without the change of intercoupling. The measured in-band return loss of the two filter prototypes with a center frequency of 35 GHz and absolute bandwidth of 1.3 GHz is below -14 dB, while the measured mini-mum in-band insertion loss is about 1.2 dB.

Cross coupling can also be realized by a nonphysical method, in which the signal path is generated by a spu-rious resonant mode. If the main resonance and spuri-ous resonance are properly chosen, all of the coupling paths can be realized by a magnetic iris, enabling the generation of FTZs [11]. More importantly, nonphysical cross coupling can be tuned not only by the size of the coupling structure but also by the resonant frequencies of spurious resonance. Therefore, the FTZs can be made near the passband for generating a high selectivity or far away from the passband for achieving a wide stopband.

In [12], an oversized SIW TE101/TE301 cavity or TE101/TE201 cavity was used as a basic unit that may be called singlet to realize a direct-coupled SIW filter for

Ka-band satellite ground terminal receiver applica-tions. Every singlet produces a transmission pole in the desired passband of the filter and a transmission zero at the defined location for yielding sufficient stopband attenuation. Figure 4 shows photographs and responses of two different fourth-order SIW filters, one using three oversized TE101/TE301 SIW cavities and one oversized TE101/TE201 SIW cavity and the other using two over-sized TE101/TE301 SIW cavities and two oversized TE101/TE201 SIW cavities. The Ka-band of 29.530 GHz needs to be isolated from the K-band passband operation for satellite ground terminal applications. The second filter exhibited a measured in-band insertion loss of 0.8 dB, a stopband attenuation (rejection) better than 40 dB over a wide frequency range of 23.9431.48 GHz, while the attenuation (rejection) in the targeted transmit band of 29.530 GHz is better than 52 dB. Note that to effectively measure the stopband rejection, appropriately placed absorbers are used to reduce the spurious coupling between two launching pins when a universal test fix-ture (Wiltron 3680K) is used. In Figure 4(b), the differ-ence between the measured S11 and the simulated S11 at 27.530GHz may be caused by the absorbers.

Because of the inherent structural flexibility of SIW technology, some cross-coupled filter topologies that are difficult or even impossible to physically realize in metallic waveguide can be easily implemented on an SIW platform [13][16]. In [13], two Ka-band fourth-order SIW filters were proposed with two transmission

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Figure 2. The (a) geometric configuration, (b) structural topology, and (c) frequency responses of a fourth-order cross-coupled SIW filter with negative coupling structure and sourceload coupling [4].


zeros on the left or right side of the passband, respec-tively, due to the diagonal coupling. All the coupling between resonators is realized by magnetic postwall irises that provide positive coupling. For the filter of two transmission zeros on the left side of the passband, the TE101 mode is used for the first or third SIW cav-ity resonator, while the TE102 resonant mode is excited in the second or fourth SIW cavity for the equivalent realization of a negative coupling in signal path 13. All of the resonators in the filter with two transmis-

sion zeros on the right side of the passband operate with the TE101 mode. Both filters are directly excited by a 50-X conductor-backed CPW with coupling slots so that a thick substrate can be used for the loss reduc-tion related to the top and bottom metals. In [14], an SIW transversal bandpass filter using a modified dou-blet with high selectivity was presented. The modified doublet contains two resonators that are not coupled to each other. The source and load are directly coupled to generate two transmission zeros. By changing the sign

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Figure 3. The (a) geometric configuration, (b) structural topology, and (c) frequency responses of the first prototyped fourth-order cross-coupled SIW filter with negative coupling structure and sourceload coupling [13]. (d) The frequency responses of the second prototyped fourth-order cross-coupled SIW filter with negative coupling structure and sourceload coupling [13]. (continued)


of one coupling coefficient and detuning the resonators in the modified doublet, two transmission zeros can be simultaneously placed below or above the passband, or one can be placed below and the other above the passband. Figure 5 shows the geometric configuration and structural topology as well as the simulated and measured frequency responses. Since the first electri-cal resonance is realized by the TE201 mode, both the coupling between the source and the second resonator and the sourceload coupling are negative. One FTZ is placed below and the other above the passband. In [15], a sixth-order SIW filter composed of two cascaded extended doublets was proposed. Each extended dou-blet consists of a main doublet with an additional res-onator grown in one of the branches. The source and the load are coupled to both branches of the doublet to generate the required two transmission zeros. The two consecutive building blocks are coupled by an inverter between two nonresonating nodes (NRNs). In [16], four FTZs were generated in a third-order cross-coupled SIW filter using a frequency-variant mixed coupling between the source and the load.

Dual-Mode SIW FiltersDual-mode filters can be made in various technologies including planar microstrip line and metallic wave-guide schemes. The primary reason for developing such filters is to reduce the filter size by at least 50% compared with cascaded resonator filters while mak-ing transmission loss as low as possible. Based on the all-inductive dual-mode filters presented in [17] and [18], a dual-mode SIW filter directly excited by a 50-X microstrip line was proposed for the first time in [19]. This work was then extended in [20][22] by the analy-sis and synthesis of a schematic topology.

An oversized SIW cavity with two degenerate modes, TE102 and TE201, is used as the basic building block with an inline input/output structure. Two consecutive basic blocks are coupled by a section of evanescent SIW. The source/load is simultaneously coupled with two elec-trical resonances within the first/last oversized SIW cavity. Each electrical resonance in one oversized SIW cavity is coupled with two electrical resonances in the other oversized SIW cavity without intercavity cou-pling. An inductive post placed where the maximum

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Figure 3. (continued) The (a) geometric configuration, (b) structural topology, and (c) frequency responses of the first prototyped fourth-order cross-coupled SIW filter with negative coupling structure and sourceload coupling [13]. (d) The frequency responses of the second prototyped fourth-order cross-coupled SIW filter with negative coupling structure and sourceload coupling [13].


electric field is located can be used to tune the resonant frequencies of degenerate modes. This can be used to compensate for fabrication error and tolerance. There-fore, an inline SIW filter with pseudoelliptical response can be obtained. Although the metallic post or slot line can be used to perturb the field distribution of the two degenerate modes for the generation of intracoupling

[23][25], no additional FTZs can be obtained due to the intracoupling. In addition, nonphysical cross coupling for additional FTZs can be generated when the input/output is perpendicular to each other [11]. Note that there is no coupling between the source and the load.

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Figure 4. Photographs and responses of two different fourth-order SIW filters with nonphysical cross coupling using (a) three oversized TE101/TE301 SIW cavities and one oversized TE101/TE201 SIW cavity and (b) two oversized TE101/TE301 SIW cavities and two oversized TE101/TE201 SIW cavities [12].

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Figure 6. The (a) geometric configuration, (b) structural topology, and (c) frequency responses of a basic building block for a dual-mode SIW circular cavity filter [26].


Another family of dual-mode SIW filters is based on the SIW circular cavity where there are two polarization-degenerated modes TM110 [26][28]. Similar to the case of an SIW rectangular cavity, an oversized SIW circular cavity with two polarization degenerate modes TM110 is used as the basic building block; the geometric configu-ration, structural topology, and frequency responses of which are shown in Figure 6. Note that the FTZs distant from the passband are generated due to the nonphysical cross coupling of spurious resonant modes [11]. The FTZ near the passband can move from one side to the other side of the passband when the resonant frequencies of degenerate modes are changed. This can be realized by changing the cavity from a circular shape to an ellipti-cal shape with more design freedom. However, the FTZ distant from the passband is always on the right side, and its position can be changed by changing the input/out-put coupling angle a [11]. The basic building block can be cascaded to design higher-order pseudoelliptical filters. In [29], a canonical folded topology was realized using an LTCC process based on two dual-mode SIW circular cavities that are excited by a 50-X microstrip through coupling slots. Intercoupling is implemented by a cross-shaped slot. Figure 7 shows the geometric configura-tion, structural topology, and frequency responses. Note that the source/load is coupled with only one electrical resonance and that each electrical resonance in one dual-mode SIW circular cavity is coupled with only one electri-cal resonance in the other dual-mode SIW cavity through the cross-shaped slot. By letting two pairs of coupling vias used for coupling inside the cavity perpendicular to each other, the field of the first and fourth resonators can have opposite direction. Therefore, negative nonadjacent coupling between the first and fourth resonators can be produced in the proposed filter structure.

Wideband SIW FiltersWideband filters have been designed and realized with TEM mode transmission lines due to their dispersionless, broadband, and single-mode performance. As mentioned earlier, SIW structures support only the transmission or propagation of TE modes that makes the design of wide-band SIW filters easier and more feasible without the design consideration of parasitic TM modes. To increase the coupling for wideband applications, a zigzag filter topology with 28% bandwidth for European ultrawide-band (UWB) applications was proposed in [30]. Additional controllable cross-coupling networks are realized by both physical and nonphysical methods to achieve sharper responses and more flexible tuning of the transmission zeros. A fast and accurate full-wave electromagnetic anal-ysis method based on the boundary integral-resonant mode expansion technique was proposed and developed to design the filter. Figure 8 shows the geometric configu-ration, structural topology, and frequency response of the filter. With the consideration that a realizable coupling is limited by the physical size of a coupling structure,

an alternative method to design a wideband SIW filter is to use the high-pass characteristic of an SIW and the bandstop behavior of planar periodic structures, such as uniplanar compact photonic bandgap, uniplanar compact defect ground structure, and CPW periodic structures [31]. Very compact wideband filters with a relative bandwidth of more than 50% can be obtained through the inherent integration of an SIW and a planar structure within the same processing technique and on the same substrate. Fig-ure 9 shows the geometric configuration and responses of the SIW-CPW filter. In addition, an SIW also can be cascaded with a planar low-pass filter for the generation of a UWB frequency response from 3.1 to 10.6 GHz [33].

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Figure 7. The (a) geometric configuration, (b) structural topology, and (c) frequency responses of a dual-mode SIW circular cavity filter based on an LTCC process [29].


Notch bands for the removal of interferences from the existing systems can be generated using bandstop reso-nators [32] or open stubs [33].

Multiband SIW FiltersTwo types of techniques are often used to generate dual-passband or multiband responses for multifunctional system applications. The first method is to use novel resonant structures with a great degree of design free-dom, such as stepped-impedance or dual-behavior reso-nators, because their two dominant resonances coincide with the two center frequencies of two designated pass-bands by adjusting their geometric parameters. The res-onators are placed so that appropriate coupling in the structure can be established. This method is very suit-able for the case where the passbands are distant from each other in frequency.

The other method is to use transmission zeros that are produced by cross-coupling or bandstop resonators to split a single passband into dual passbands or mul-tibands based on a single-filter circuit. This method is

very useful for the case where the passbands are close to each other.

Multiband filters with Cheby-shev or pseudoelliptical frequen-cy responses were proposed on an SIW platform in [34] for the first time. Bandstop SIW cav-ity resonators are used here to generate the transmission zeros to split the single passband into two or three subpassbands. The filters, consisting of the invert-er-coupled resonator sections with side-by-side horizontally oriented SIW cavities coupled by postwall irises, are analyti-cally synthesized from the gen-eralized low-pass prototypes having Chebyshev or quasi-elliptical responses. The filters are directly excited using 50-X microstrip lines. Figure10 shows the geometric configuration and frequency responses over the K-band of the dual-/triple-band SIW filters. In [35] and [36], the inverter-coupled resonator sec-tion was modified for a novel triple-passband filter. Multiple resonant modes in an SIW cav-ity produced by an LTCC process were utilized to generate multi-band responses that are distant from each other [37], [38]. The fre-quency bands are controlled by adequately choosing an appro-

priate geometric shape for the SIW cavity resonators. The desired coupling coefficients and external quality factors for both bands are realized by the positions of open slots and feeding probes.

Reconfigurable SIW FiltersReconfigurable filters are essential for future multifunc-tional radio and radar systems such as smart and cogni-tive radio and radar techniques across the commercial, defense, and civilian sectors to control and better use the RF spectrum. These techniques can eliminate interfer-ence while preserving good dynamic range under any signal receiving condition. Tunable resonators are a cru-cial building block in the design and realization of tun-able RF and microwave filters. In [39], an original tuning solution was proposed based on the insertion of vertical capacitive posts integrated within SIW cavities. One ex-tremity of each post is connected to a floating metallic ring, located on the substrate supporting the SIW cav-ity. Frequency agility is obtained once the metallic ring is connected to the ground plane by short-circuiting

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Figure 8. The (a) geometric configuration, (b) structural topology, and (c) frequency responses of a wideband SIW filter in zigzag topology [30].


the corresponding annular slot using surface actuators. Such a combination of cavity and planar activation leads to reconfigurable filters with high Q values and simpli-fied tuning control conditions. More capacitive posts can be used to increase frequency states.

A novel tunable second-order SIW filter was imple-mented on three-layer Rogers RT/duroid substrate using p-i-n diode switching elements [40]; other switching techniques such as RF microelectromechanical sys-tems (MEMS) could be used. The tuning mechanism is obtained by connecting or disconnecting perturbing via posts to/from the top metal layer of the cavity. The tuning location where subsequent perturbing vias are placed is empirically analyzed, and an optimum tuning location is obtained. Figure 11 shows the geometric con-figuration and measured responses of a reconfigurable SIW filter. The filter provides six states ranging from 1.55 to 2.0 GHz (25% tuning). The fractional bandwidth ranges from 2.3% to 3.0% with an insertion loss less than 5.4 dB and a return loss greater than 14 dB over the entire tuning range.

Based on the proposed combline resonator in SIW tech-nology [41], a low-loss tunable resonator based on a com-bline SIW cavity loaded with gallium arsenide varactor is presented [42]. The 2.63.1-GHz tunable center frequency was obtained with a Qu between 180 and 70, a capacitance variation between 0.25 and 1.25 pF. A two-pole filter has

been demonstrated on a low-cost substrate showing a tunable center frequency between 2.64 and 2.88 GHz with 1.273.63-dB insertion loss across the tuning range [43]. The IIP3 of +24dBm has been measured for a bias voltage of 5 V and a two-tone separation f3 of 50 kHz. The 1-dB compression point of the filter at the same bias voltage is only +10 dBm. In [44], switchable planar p-i-n diodes are used to realize the digitally tuned SIW filter. The center frequencies of three different states are 6.17, 5.83, and 5.58 GHz, with insertion loss varying between 1.6 and 2.4 dB. The equal ripple fractional bandwidth is reduced from 3.6% to 2.6% along the tuning range.

Another method to tune the SIW cavity is to embed a frequency agile material into an SIW cavity. A spe-cific switchable post constructed using plasma material (argon) was introduced in the SIW cavity [45]. The plasma conductivity can reach a high value when the medium is excited by strong dc voltage (on state). Whereas for the off state, the plasma behaves like a vacuum. Therefore,

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Figure 10. The geometric configuration of (a) dual-band and (b) triple-band SIW filters as well as (c) and (d) their frequency responses [34].


plasma acts as a switchable metallic post that can change the resonant frequency of an SIW cavity.

In [46], a dielectric post exhibiting dispersion provides electromagnetic agility and adaptability for the filter by introducing a vertical material perturbation. Low-loss dispersion of nanoparticles with variable concentrations is used as the enabling mechanism for electromagnetic agility and adaptation. The post diameter determines the amount of perturbation achieved. However, it is still a very challenging task to simultaneously tune the reso-nant frequency of an SIW cavity and coupling between two SIW cavities for constant absolute bandwidth or relative bandwidth during the tuning process.

Arts of Miniaturization for SIW FiltersAn SIW filter is still bulky for low-microwave-frequency applications because of their strong frequency depen-dence and non-TEM mode operation. Various techniques have been proposed to reduce the size of an SIW filter, such as SIW filters loaded with complementary split-ring resonators [47][51], folded SIW filters [52][55], combline

SIW filters [41], [56], [57], and so-called half-mode SIW filters [58][60]. However, these techniques may deterio-rate the Qu of an SIW resonator, which leads to high in-band insertion loss of the filter that, in turn, limits the applications of SIW filters in engineering. This is because the field singularity known for conventional planar lines may partially appear in the filter circuits.

An effective method to miniaturize an SIW filter is to utilize low-loss substrates that have high permittiv-ity, such as alumina ceramic substrates, by using the miniature hybrid microwave integrated circuits process or the photoimageable process [61][64] or on a silicon substrate by means of the MEMS process [65]. The low-loss high-permittivity substrates can be found in com-mon fabrication techniques such as alumina ceramic substrates using the miniature hybrid microwave inte-grated circuits process or the photoimageable process [61][64]. However, the best method for the miniaturiza-tion of SIW filters uses multilayer circuit processes with three-dimensional integration features. LTCC processes with the substrates of high permittivity and a nearly arbitrary number of layers are cost effective, and, most importantly, almost all of the filter topologies can be realized with great freedom through the LTCC process.

The first SIW filter based on a multilayer PCB sub-strate was proposed in [66]. A C-band fourth-order SIW filter with a canonical folded topology for an elliptical frequency response was designed and fabricated. The required negative coupling between resonators 1 and 4 is easily realized by adding the square slots at the cen-ter of the SIW cavities, while inductive postwall irises

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Layer 2

Layer 3(GND)

ThroughVias

P'

P'P

P

RF In

Figure 11. (a) and (b) The geometric configuration and (c) measured frequency response of a reconfigurable SIW filter [40].

50-45

-40

-35

-30

-25

S-P

aram

eter

s (d

B)

-20

-15

-10

-5

0

52 54 56 58 60

S11 (Simulated)

S

+ ++

1 4

2 3

L

S21 (Simulated)S11 (Measured)S21 (Measured)

62 64 66 68 70

Frequency (GHz)

(c)

(b)(a)

Figure 12. The (a) geometric configuration, (b) structural topology, and (c) response of an SIW filter based on LTCC [67].


for positive coupling are made between two SIW cavi-ties in the same layer. Rectangular slots at the edge of the cavity are made for positive coupling between two SIW cavities in the different layers. The filter area is reduced by one-half with a vertically stacked SIW cav-ity using a multilayer PCB process. The input and out-put microstrip lines are located in different layers, and via transitions are required for practical measurement and applications. To assign the input/output struc-tures to the top layer, a novel fourth-order SIW filter for 60-GHz applications was proposed for fabrication in an LTCC process [67], [68]. Resonators 1 and 4 are situated in the same layer, and the negative coupling between them is realized by a mixed coupling structure. The fil-ter, whose geometric configuration, structural topology, and frequency response are shown in Figure 12, has a very good performance at the 60-GHz band.

Another topology with a pseudoelliptical response was realized using an SIW triangular cavity in [69]. In this article, a negative coupling between resonators 1 and 2 on different layers is achieved using a circular slot at the center of the two cavities. All the other couplings are positive. A slot on the top metal layer is introduced to suppress the two higher spurious modes. The filter presents a 65% size reduction in comparison with its pla-nar counterpart. Figure 13 shows its geometric configu-

ration and structural topology and frequency response over the Ku-band.

Along with the realization of SIW filters based on different topologies using the LTCC process [70][76], techniques for further size reduction of SIW filters were studied. These techniques include dual-mode SIW fil-ters on LTCC [77], [78], folded SIW filters on LTCC [79][81], and evanescent-mode SIW filters on LTCC [82][86]. Although the evanescent-mode SIW filter has a size reduction of more than 50% and a higher spectral sepa-ration to the next spurious passband, the in-band inser-tion loss is increased because the Qu value is reduced. The filter is very sensitive to fabrication tolerances, especially to the compression or shrinking effects of the LTCC layers before and during the cofiring, because the electric field is concentrated above the capacitive stubs in only one thin LTCC layer [86].

ConclusionsAdvanced and innovative SIW filter structures are reviewed and discussed in the second of this three-part series with regard to various types of structure realiza-tions and design approaches. Physical or nonphysical coupling can be used for the realization of cross cou-pling in SIW filters with FTZs. Circular or square cavi-ties can be easily designed for dual-mode SIW filters. Techniques for wideband or multiband SIW filters were also discussed together with reconfigurable SIW filter platforms. Loaded SIW cavity and folded SIW cavity can be used to reduce the filter size for low-gigahertz applications. LTCC technology may present an effective method to miniaturize SIW filters with good Qu values.

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13.5-70

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S-P

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