Hindawi Publishing CorporationActive and Passive Electronic ComponentsVolume 2013 Article ID 953498 10 pageshttpdxdoiorg1011552013953498

Research ArticleA Low-Power Ultrawideband Low-Noise Amplifier in018 120583m CMOS Technology

Jun-Da Chen

National Quemoy University Department of Electronic Engineering University Road Jinning Township Kinmen 892 Taiwan

Correspondence should be addressed to Jun-Da Chen chenjdnquedutw

Received 11 July 2013 Revised 17 October 2013 Accepted 4 November 2013

Academic Editor Rezaul Hasan

Copyright copy 2013 Jun-Da ChenThis is an open access article distributed under the Creative Commons Attribution License whichpermits unrestricted use distribution and reproduction in any medium provided the original work is properly cited

This paper presents an ultrawideband low-noise amplifier chip using TSMC 018120583m CMOS technology We propose a UWB lownoise amplifier (LNA) for low-voltage and low-power application The present UWB LNA leads to a better performance in termsof isolation chip size and power consumption for low supply voltage This UWB LNA is designed based on a current-reusedtopology and a simplified RLC circuit is used to achieve the input broadband matching Output impedance introduces the LCmatching method to reduce power consumption The measured results of the proposed LNA show an average power gain (S

21) of

9 dB with the 3 dB band from 3 to 56GHz The input reflection coefficient (S11) less than minus9 dB is from 3 to 11 GHz The output

reflection coefficient (S22) less than minus8 dB is from 3 to 75 GHz The noise figure 46ndash53 dB is from 3 to 56GHz Input third-order-

intercept point (IIP3) of 2 dBm is at 53 GHz The dc power consumption of this LNA is 9mW under the supply of a 1 V supply

voltage The chip size of the CMOS UWB LNA is 103 times 078mm2 in total

1 Introduction

The ultrawideband (UWB) system has become one of themajor technologies for wireless communication systemsand local area networks The IEEE 802153a ultrawideband(UWB) system uses a specific frequency band (31 GHzsim106GHz) to access data and employs the system of orthogo-nal frequency-divisionmultiplexing (OFDM)modulation [1ndash3]The frequency band consists of four groups A B C andDwith thirteen channels Each channel bandwidth is 528MHzas shown in Figure 1The system operates across a wide rangeof frequency 31ndash5GHz or 31ndash106GHz The low frequencyband from 31 to 5GHz has been allocated for developingthe first generation of UWB systems [4] The system hasseveral advantages such as low complexity low cost and ahigh data rate for the wireless system In the front-end systemdesign a low-noise amplifier (LNA) is the first block in thereceiver path of a communication systemThe chief objectiveof the LNA is to reach the low-noise figure to improve theoverall system noise [5] The LNA must minimize the noisefigure over the entire bandwidth feature flat gain goodlinearity wideband input-output matching and low power

consumption In the radio frequency circuit designGaAs andbipolar transistors have performed fairly well Neverthelessthese processes lead to increased cost and greater complexityThe RF front-end circuits using CMOS technology canprovide a single-chip solution which greatly reduces the cost[6] With the rapid improvement of CMOS technology itcan implement RF ICs with CMOS Three major topologiesof the present wideband low-noise amplifiers are used Firstdistributed amplifiers [7ndash9] can provide good linearity andwidebandmatching but require high power consumption anda large chip area because of the multiple amplifying stagesSecond resistive shunt feedback amplifiers [4 10] providegoodwidebandmatching andflat gain but require high powerconsumption Last Chebyshev filter amplifiers [11 12] adopt awideband LC bandpass filter for input matching and a sourcefollower for outputmatchingThis type of wideband amplifierdissipates a small amount of dc power The Chebyshevfilter matching design requires three inductors Therefore avery large chip area is required This paper proposes RLCbroadband matching for LNA design The primary goal isto obtain wideband input-output matching and to reducethe chip area and power consumption The LNA topology is

2 Active and Passive Electronic Components

proposed in Section 2 The complete analysis of the designmethodology of the widebandmatching network is presentedin Section 3 In Section 4 implementation and measurementresults are presented The conclusion is presented in the lastsection

2 Low-Voltage WidebandCascode LNA Design

Figure 2 shows the basic cascode LNA The current-reusedconfiguration can be considered as two common-sourceamplifiers (119872

1and119872

2)The current shared cascode amplifier

provides a low-power characteristic under the low voltagesupply Because 119872

1and 119872

2share the same bias current

the total power consumption is minimized [6 12 16] For acascode amplifier the overall noise figure can be obtainedin terms of the noise figure (NF) and gain of each stage asfollows

119865total = 1198651+1198652minus 1

1198661198601

(1)

where 119865total is the total NF and 1198651 and 1198652 are the NF values ofthe first and second stage respectively 119866

1198601is the power gain

of this first stage therefore in the LNA circuits the noise ofthe first stage is more critical Consequently the gain of thetransistorrsquos 119872

1must be high enough to suppress the noise

The dominant noise sources for active MOSFET transistorsare flicker and thermal noisesThe flicker noise is modeled asthe voltage source in series with the gate of value

1198812119892(119891) =

119870

119882119871119862119900119909119891 (2)

where the constant 119870 is dependent on device characteristicsand can vary widely for different devices in the same processThe variables 119882 119871 and 119862

119900119909represent the width length

and gate capacitance per unit area respectively The flickernoise is inversely proportional to the transistor area119882119871 Inother words a larger device reduces this noise The noiseprocess of thermal noise is random The MOSFET thermalnoise includes threemain noise sources as shown in Figure 3distributed gate resistance noise (1198812rg) drain current noise(1198942nd) and gate current noise (1198942nd)

The thermal noise source mainly comes from the draincurrent noise and gate-induced noise of the transistorsMultifinger layout technology is applied to reduce the gate-induced noise According to the MOSFET noise analysis in[17 18] we must express the noise figure in a way that explic-itly considers power consumption In order to determine thewidth of119872

1and find out the best NF we use the dependency

among noise factor (119865) power dissipation (PD) and qualityfactor (119876

119871) to find out the optimal value [17] Based on

these parameters the related curves NF (dB) versus 119876119871 are

plotted in Figure 4 by MATLAB simulation software Thereis a compromise between noise performance and power gainin Figure 4 Therefore the 119876

119871can be determined to be 4 as

PD equals to 10mW The optimal width of MOSFET can beobtained by [17 18]

119882opt =3

2

1

120596119900119871119862119900119909119877119878119876119871

asymp1

3120596119900119871119862119900119909119877119878

(3)

where 120596119900is the operation frequency 119871 is the gate length and

119862ox is the gate capacitance per unit area Equation (3) showsthat the optimal width for 119872

1is 240120583m for 120596o = 34GHz

119871 = 018 120583m 119862119900119909= 862 Fm2 119877

119878= 50Ω and 119876

119871= 4

Figure 5 shows the proposed UWB LNA schematicThe current sharing cascode amplifier provides low-powercharacteristics with low voltage supply Note that 119872

2is

the common-gate stage of the cascode configuration whicheliminates theMiller effect and provides better isolation fromthe output return signal [18 19] The passive components 119862

1

1198622 1198623 1198771 and 119871

1are adopted for the matching network

at the input to resonate over the entire frequency band Theresistors 119877

2and 119877

119892are used to provide bias voltage for the

transistors 1198721and 119872

2 The capacitor 119862

4provides signal

coupling between the two stages The capacitor 1198625bypasses

the ac current to the ground and avoids coupling to the firststage This affects gain flatness therefore it is possible toprovide an ideal ac ground but gain flatness is not affectedin the design 119871

2is the inductor load of the first stage The

output matching network is composed of 1198713 1198714 1198626 and 119862

7

In the cascode stages the first stagersquos noise figure contributionis more than the second stagersquos The first stagersquos transistorsize and bias point should be optimized for the low NF [17]The second stagersquos transistor size and bias point should beoptimized for high linearity The cascode LNA design is fullof trade-offs between optimal gain low noise figure inputand output matching linearity and power consumptionMoreover to avoid oscillation the parasitic couplings haveto be minimized [16] The sizes of the devices of the LNA areshown in Table 1

3 Proposed Wideband MatchingNetwork Analysis

31 Input Impedance Matching As shown in Figure 2 in theconventional narrow band LNA design the input impedanceof the input stage (119872

1) can be written as

119885in = 119904 (119871119892+ 119871119878) +

1

119904119862gs1+ (

1198921198981

119862gs1)119871119904 (4)

where 119862gs1 is the gate-source capacitance and 1198921198981

is thetransconductance of the input transistor119872

1 We choose ap-

propriate values of inductance (119871119892+119871119904) and capacitance119862gs1

which resonate at a certain frequency The real term can bemade equal to 50Ω Equation (3) determines the width of119872

1

and finds the best NF For wideband design it is difficult tolet the imaginary part of (4) remain zero for a wide rangeTo discuss UWB input impedance matching we need toconsider the standard form of the second-order filter

119879 (119878) =11988621198782+ 1198861119878 + 119886119900

1198782 + 119878 (120596119900119876in) + 120596

2

119900

=11988621198782+ 1198861119878 + 119886119900

1198782 + 119878B + 1205962119900

(5)

Active and Passive Electronic Components 3

3432 3960 4488(MHz)

5016 5808 6336 6864 7392 7920 8448 8976 9504 10032

Group A Group B Group C Group D

Figure 1 The IEEE 802153a spectrum

Table 1 Device sizes of the proposed UWB LNA

Device 1198621

(pF)1198622

(pF)1198711

(nH)1198623

(pF)1198771

(kΩ)119877119892

(kΩ)1198624

(pF)1198712

(nH)1198625

(pF)1198772

(kΩ)1198713

(nH)1198714

(nH)1198626

(pF)1198627

(pF)1198721-1198722

(120583m)Size 951 011 091 76 014 384 57 553 57 384 119 059 016 123 240018

M2

M1

Ls

Lg

Vbias

Zin

Figure 2 Cascode low noise amplifier

V2rg

gsi2ng i2ndCgs rogmV

Rg

Figure 3 CMOS noise model

where 1198862 1198861 and 119886

119900are numerator coefficients determining

the type of second-order filter function 120596119900is called the pole

frequency119876in is termed the pole factor and 119861 is called band-width According to 119861 = 120596

119900119876in the bandwidth is inversely

proportional to the 119876in if the value of the pole frequency 120596119900is fixed

0 1 2 3 4 5 6 7 800

05

10

15

20

25

30

35

NF

(dB)

PD = 5 mWPD = 10 mW

PD = 15 mWPD = 20 mW

QL

Figure 4 Noise Figure versus 119876119871for several power dissipations at

34GHz

R2

L3 L4 C7

C6

C5

C4L2

M2

M1

C3

R1C2

C1 L1

Rg

Vbias

RFin

RFout

VDD

Figure 5 Schematic of the proposed UWB LNA

4 Active and Passive Electronic Components

Rs = 50Ω

Vs

Zin

C1

C2

C3

R1

L1Cgs1

Figure 6 The proposed wideband matchingapproximate first stageinput matching small-signal equivalent circuit

Figure 6 shows the proposed wideband matching and asmall-signal equivalent circuit in the first stage The inputimpedance of the RLC network can be written as

119885in =1

1198781198621

+1

1198781198622

[1198781198711+ (1198771+

1

1198781198623

) 1

119878119862gs1]

= 119877in + 119895119883in

(6)

Because 1198623capacitance value is much larger than 119862gs(1198623 ≫

119862gs) the (1198771 + (11198781198623))(1119878119862gs1) will approximate (119877

1+

(11198781198623))

119885in =1

1198781198621

+1

1198781198622

(1198781198711+ (1198771+

1

1198781198623

)) = 119877in + 119895119883in

119885in asymp(119878 + (119877

11198711)) ((119862

1+ 1198622) 11986211198622)

1198782 + 119878 (11987711198711) + ((119862

2+ 1198623) 1198711(11986221198623))

(7)

where 119862gs1 is the gate-source capacitance of1198721 This equiv-alent circuit can roughly be an RLC second-order filterstructure In (7) the quality factor of the filter circuit can begiven by

119891119900asymp

1

2120587radic

1198622+ 1198623

1198711(11986221198623) (8)

119876 asymp1

1198771

radic1198711(1198622+ 1198623)

11986221198623

(9)

where 119891119900is the resonant frequency In (8) and (9) to obtain

broader bandwidth a low-quality factor needs to be added tothe passive devices The narrowband LNA can be convertedinto a wideband amplifier by properly selecting passivedevices According to (8) the capacitors 119862

2 1198623and inductor

1198711will be optimized at the resonant frequency We use the

CAD of Agilent ADS to analyze the circuit performanceof input impedance matching According to (8) and (9) 119891

119900

is inversely proportional to 1198711while 119876 is proportional to

1198711 Therefore bandwidth is inversely proportional to 119871

1

Figure 7 is fixed as 1198621(951 pF) 119862

2(011 pF) 119862

3(76 pF) and

1198771(140Ω) showing that the resonant frequency is inversely

proportional to the 1198711 Bandwidth is inversely proportional

3 4 5 6 7 8 9 102 11Frequency (GHz)

minus20

minus25

minus15

minus10

minus5

0

L1 = 13nHL1 = 115nHL1 = 091 nH

S 11

(dB)

Figure 7 Fixed 1198621 1198622 1198623and 119877

1and simulated 119871

1variation input

reflection coefficient

to 1198711 According to (9) the input matching circuit uses 119877

1

and 1198623to reduce the number of inductors and make the 119876

factor smaller 119876 is inversely proportional to 1198771 Therefore

bandwidth is proportional to 1198771 Figure 8 is fixed as 119862

1

(951 pF) 1198622(011 pF) 119862

3(76 pF) and 119871

1(091 nH) showing

that the bandwidth is proportional to the 1198771 Figure 9 is fixed

as 1198621(951 pF) 119862

2(011 pF) 119862

3(76 pF) and 119871

1(091 nH)

showing that the noise figure is inversely proportional to the1198771 119891119900is inversely proportional to 119862

3while 119876 is inversely

proportional to 1198623 Therefore the bandwidth is proportional

to the 1198623 Figures 10 and 11 are fixed as 119862

1(951 pF) 119862

2

(011 pF) 1198771(140Ω) and 119871

1(091 nH) showing that the

bandwidth and noise figure are proportional to the 1198623 The

size of the transistors 1198721 1198771 and 119862

3must be carefully

selected There is a trade-off in the design between widebandmatching and the noise figure On the other hand thedevice size must yield a sufficient noise performance andpower gain As can be seen the second-order filter of1198711(091 nH) 119877

1(140Ω) and 119862

3(76 pF) is employed as the

inputmatching networkThe inputmatching network has lesscomplexity and better reflection coefficient from 3GHz to 10GHz

32 Output Impedance Matching Low-noise amplifiers relyon output impedance matching to achieve maximum powergain A source follower technique has been widely used toprovide wideband output matching Output impedance issimilar to 1119892

119898119904 in which 119892

119898119904is a gate-source transconduc-

tance of the source follower [4 10] However it consumesmore power For these reasons we introduce an outputimpedance matching method suitable for wideband LNAdesign Figure 12 shows the approximate output impedancesmall-signal equivalent circuit The output impedance of theRLC network can be written as

Active and Passive Electronic Components 5

1 2 3 4 5 6 7 8 9 100 11Frequency (GHz)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

minus20

minus15

minus10

minus5

0

S 11

(dB)

Figure 8 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation input

reflection coefficient

1 2 3 4 5 6 7 8 9 100 11

3456789

2

10

Frequency (GHz)

NF

(dB)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

Figure 9 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation noise

figure

119885out = ((1199031198892

1

1198781198621198892

1198781198713) + 119878119871

4)

1

1198781198626

+1

1198781198627

= 119877out + 119895119883out

(10)

119885out

asymp(119878+(1119903

11988921198621198892)) ((119862

7+1198626) 11986261198627)

1198782+119878 (111990311988921198621198892)+((119862

61198714+11986261198713+11986211988921198713) 1198621198892119862611987131198714)

(11)

where 1198621198892

is the parasitic capacitance of 1198722at the drain

node and 1199031198892

is the resistor of 1198722at the drain node This

equivalent circuit can approximately be an RLC second-order

1 2 3 4 5 6 7 8 90 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

C3 = 76 pFC3 = 095 pFC3 = 015 pF

S 11

(dB)

Figure 10 Fixed119862111986221198771 and 119871

1and simulated119862

3variation input

reflection coefficient

1 2 3 4 5 6 7 8 90

3456789

2

10

10 11

NF

(dB)

Frequency (GHz)C3 = 76 pFC3 = 095 pFC3 = 015 pF

Figure 11 Fixed119862111986221198771 and 119871

1and simulated119862

3variation noise

figure

filter structure In (11) the quality factor of the filter circuitcan be given by

119891119900asymp

1

2120587radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(12)

119876 asymp 11990311988921198621198892radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(13)

where 119891119900is the resonant frequency In (12) and (13) in order

to obtain broader bandwidth a low-quality factor needs tobe added to passive devices 119891

119900is inversely proportional to

1198621198892

while bandwidth is inversely proportional to 11990311988921198621198892 The

1198621198892capacitance is proportional to the width of the transistor

The 1199031198892

resistance is inversely proportional to the width ofthe transistor UWB system uses a specific frequency band

6 Active and Passive Electronic Components

Rs = 50 Ω

outZ

C7L4

C6L3

rd2 Cd2

Figure 12 The wideband output matching small-signal equivalentcircuit

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus30

minus20

minus25

minus15

minus10

minus5

0

S 22

(dB)

M2 = 160018 120583mM2 = 240018 120583mM2 = 320018 120583m

Figure 13 Fixed 1198626 1198627 1198713and 119871

4 simulated119872

2variation output

reflection coefficient

(31ndash5GHz or 31ndash106GHz) to access data Generally theresults of UWB return loss are less than minus10 dB The powerconsumption is proportional to the width of the transistorAccording to (13) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting 119872

2 This

provides good wideband matching and flat gain Figure 13is fixed as 119862

6(016 pF) 119862

7(123 pF) 119871

3(119 nH) and 119871

4

(059 nH) showing that the simulated 1198722variation output

reflection coefficient As can be seen the second-order filterof 1198722(240018 120583m) is employed as the output matching

network According to (12) the capacitor 1198626and inductor

1198713-1198714will be optimized at the resonant frequency Figure 14

is fixed as 1198626(016 pF) 119862

7(123 pF) 119872

2(240018120583m)

and 1198714(059 nH) showing that the resonant frequency is

inversely proportional to the 1198713 The output network has less

complexity and a better reflected coefficient from 31 GHz to106GHz

33 Gain At a high frequency the current gain of commonsource configured transistor 119872

1is 1198921198981119904119862gs1 [11] Figure 15

shows the approximate first stage 1198721load of the simplified

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

S 22

(dB)

L3 = 119nHL3 = 129nH

L3 = 109nH

Figure 14 Fixed 1198626 1198627 1198722 and 119871

4and simulated 119871

3variation

output reflection coefficient

Id1

Vout1

C5

L2

Cd11gm2

Figure 15 The first stage simplified small-signal equivalent circuit

small-signal equivalent circuit [12] The first stage load canbe approximated as

119885load1 asymp1

119878119871211986251198621198891

sdot119878211987121198625+ 11987811989211989821198712+ 1

1198782 + 119878 (11989211989821198625) + ((119862

5+ 1198621198891) (1198712(11986251198621198891)))

(14)

The transfer function of the input matching network is119885in(119904) The first stage voltage gain can be approximated as

1198601198811

asymp minus1198921198981119885in

119878119862gs1119877119904sdot 119885Load1 (15)

where 119877119904is the source resistance in which voltage gain is

determined by the load inductor 1198712 the total capacitance at

the drain of 1198721 and bypass capacitor 119862

5 The second stage

voltage gain can be estimated as

1198601198812

asymp minus1198921198982

119878119862gs2119877in2sdot 119885out (16)

Active and Passive Electronic Components 7

1 2 3 4 5 6 7 8 9 10

10

12

110

2

4

6

8

0

Frequency (GHz)

Δ

K

Δ K

Figure 16 119870 (stability factor) and Δ simulation result

where 119877in2 is the input resistance at the gate of1198722 and 119885outis the output load impedance The LNA voltage gain can beapproximated as

119860119881asymp11989211989811198921198982119885in119885Load1119885out

1198782119862gs1119862gs2119877119904119877in2 (17)

According to (17) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting the devicesize This provides good wideband matching noise figureoptimization design and flat gain

34 Stability Amplifier stability is important to preventnatural oscillation The stability can be determined by the 119878-parameters input and output matching network and circuitterminations The simpler tests show whether or not a devicecan be unconditionally stable [20] One of these is the 119870-Δtest which shows if a device can be unconditionally stable ifRolletrsquos condition is defined as

119870 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

minus100381610038161003816100381611987822

10038161003816100381610038162

+ |Δ|2

210038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1

|Δ| =10038161003816100381610038161198781111987822 minus 1198781211987821

1003816100381610038161003816 lt 1

(18)

The 119870-Δ test cannot be used to compare the relative sta-bility of twoormore devices because it involves constraints ontwo separate parameters A new criterion has been proposedwhich combines the 119878-parameters in a test involving only asingle parameter 120583 defined as

120583 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

100381610038161003816100381611987822 minus Δ119878lowast

11

1003816100381610038161003816 +10038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1 (19)

The performance of LNA is simulated using the advancedesign system (ADS) To consider the stability of the LNAthe119870 factor andΔ should be considered Figure 16 shows thatthe 119870 factor is always larger than 1 and the Δ is smaller than1 all the time Hence this circuit is stable for all frequency

1 2 3 4 5 6 7 8 9 10 11

11

0Frequency (GHz)

3

5

7

9

1

120583load120583source

120583lo

ad120583

sour

ce

Figure 17 120583 factor simulation result

RFin RFout

Vbias

DC power supply(HP 4142B)

Network analyzer(HP 8510C)

S-parameters test set(HP 8517B)

VDD

Figure 18 Test setup used for the LNA 119878-parameter measurements

bands There is another way to diagnose whether the LNAis unconditionally stable The 120583 factor at input and outputshould be larger than 1 in all frequency bands as shown inFigure 17

4 Measurement Results

On-wafer measurement is performed by an HP 8510C net-work analyzer and an HP 8517B is to test the 119878-parameter asshown in Figure 18 A network analyzer is used to measurethe frequency response and input matching of the LNAThe input and output impedance matchings are both 50ΩTo ensure that the LNA still provides a conversion gainwhen process deviation occurs the other process cornersnamely typical-NMOS typical-PMOS (TT) fast-NMOS fast-PMOS (FF) and slow-NMOS slow-PMOS (SS) are also usedto simulate this LNA The results are shown in Figure 19Figure 19 shows the measurement forward gain (119878

21) from

3 to 56GHz the measured gain is about 7ndash10 dB Themeasurement data is 6 dB lower than the pre-simulation data

8 Active and Passive Electronic Components

201816141210

86420

28 33 38 43 48 53 58Frequency (GHz)

S 21

(dB)

Pre-simulation TTPost-simulation FFPost-simulation TT

Post-simulation SSMeasurement

Figure 19 Simulation and measurement results of conversion gainversus frequency of the proposed LNA RF stands for simulationresults of modifiedMOSFET RFmodel FF TT and SS represent thesimulation results of fast-NMOS fast-PMOS typical-NMOS typical-PMOS and slow-NMOS slow-PMOS process corners

minus10

0

minus20

minus30

minus40

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 11

(dB)

Figure 20 Measured and simulated input reflection coefficient

because of the process drift and parasitic effect in the layoutThe result of conversion gain measurement is situated in thepost-simulation (SS) process corner

Figure 20 shows the input return loss (S11) from 3 to

11 GHz and the measured S11

is less than minus9 dB Figure 21shows output return loss (S

22) from 3 to 75GHz and the

measured S22is less thanminus8 dB Figure 22 shows reverse isola-

tion (S12) from 3 to 11 GHz and the measured S

12is less than

minus40 dB The simulation measured input of 1 dB compressionpoint at 53 GHz shown in Figure 23 is about minus8 dB Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz and the IIP

3is 2 dBm at 53 GHz The

IM3 is measured using two Agilnet E8247C continuous wave

minus10

minus15

minus20

minus25

minus30

minus5

0

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 22

(dB)

Figure 21 Measured and simulated output reflection coefficient

minus10

minus20

minus30

minus40

minus50

minus60

minus70

minus80

MeasurementSimulation

3 4 5 6 7 8 9 102 11

0

Frequency (GHz)

S 12

(dB)

Figure 22 Measured and simulated reverse isolation

minus40 minus35 minus30 minus25 minus20 minus15 minus10 minus5 0 5 100

1

2

3

4

5

6

7

8

9

Gai

n (d

B)

Input power (dBm)

P1dB

Figure 23 Measured input 1 db compression point at 53 GHz

Active and Passive Electronic Components 9

Table 2 Recently reported performance of UWB LNAs

Reference Process CMOS(120583m) 119878

11(dB) Avg gain

(db) Freq (GHz) NF (dB) IIP3(dBm) Die area

(mm2)Power(mW) Topology

[4] 018120583m CMOS ltminus78 119 2minus65 41minus46 4 (4GHz) 088 27 Feedback[7] 06120583m CMOS ltminus7 61 05minus55 54minus82 NA 112 834 Distributed[8] 018 120583m CMOS ltminus20 10 0minus11 31minus61 NA 144 196 Distributed[9] 018120583m CMOS ltminus12 14 3minus6 47minus67 minus5 (45GHz) 11 594 Distributed[10] 018120583m CMOS ltminus9 98 2ndash46 23minus5 minus7 (4GHz) 09 126lowast Feedback[11] 018120583m CMOS ltminus99 93 24minus95 4minus9 minus67 (6GHz) 11 9lowast Chebyshev filter[12] 018120583m CMOS ltminus10 86 24minus94 41minus10 minus35 (6GHz) 176 71lowast Chebyshev filter

[13] 018120583m CMOS ltminus5 191 28minus72 32minus38 minus1 (6GHz) 163 32 Feedback networksynthesis

[14] 018120583m CMOS ltminus10 17 2ndash11 38 NA 0635 1056lowastlowast Feedback[15] 015 120583mHEMT NA 18 085ndash1335 25 NA 1162 70lowastlowast FeedbackThis work 018120583m CMOS ltminus9 9 3ndash56 46ndash53 2 (53 GHz) 08 9 ProposedlowastOnly core LNA lowastlowastSimulation

minus25 minus20 minus15 minus10 minus5 0 5minus80

minus70

minus60

minus50

minus40

minus30

minus20

minus10

0

10

Input power (dBm)

Out

put p

ower

(dBm

)

FundamentalIMD3

IIP3

Figure 24 Measured IIP3at 53 GHz

(CW) generators and an Agilent E4407B spectrum analyzerA noise figure (NF) is measured using an Agilent N8975ANF meter with an Agilent 346C noise source The simulationnoise figure minimum is 41 dB as shown in Figure 25 Dueto the parasitic effect in the layout the minimum of themeasurement is 46 dB Figure 26 shows the micrograph ofthe LNA Table 2 summarizes the measurement results andcompares them with previous literature In the proposedLNA topology the simplified RLC input matching networkand the current-reused configuration benefit from the low-power design The distributed amplifier [7ndash9] requires highpower consumption and a large chip area Compared withthe Chebyshev filter and feedback network synthesis [11ndash13] the proposed input matching network which has onlyone spiral inductor simplifies the circuit complexity and

30 35 40 45 50 55 60 65 70 75 800

1

2

3

4

5

6

7

8

9

10N

oise

figu

re (d

B)

Measured NFSimulated NF

Frequency (GHz)

Figure 25 Simulation and measurement NF

Figure 26Micrograph of the proposed LNA (size 103times 078mm2)

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

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DistributedSensor Networks

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2 Active and Passive Electronic Components

proposed in Section 2 The complete analysis of the designmethodology of the widebandmatching network is presentedin Section 3 In Section 4 implementation and measurementresults are presented The conclusion is presented in the lastsection

2 Low-Voltage WidebandCascode LNA Design

Figure 2 shows the basic cascode LNA The current-reusedconfiguration can be considered as two common-sourceamplifiers (119872

1and119872

2)The current shared cascode amplifier

provides a low-power characteristic under the low voltagesupply Because 119872

1and 119872

2share the same bias current

the total power consumption is minimized [6 12 16] For acascode amplifier the overall noise figure can be obtainedin terms of the noise figure (NF) and gain of each stage asfollows

119865total = 1198651+1198652minus 1

1198661198601

(1)

where 119865total is the total NF and 1198651 and 1198652 are the NF values ofthe first and second stage respectively 119866

1198601is the power gain

of this first stage therefore in the LNA circuits the noise ofthe first stage is more critical Consequently the gain of thetransistorrsquos 119872

1must be high enough to suppress the noise

The dominant noise sources for active MOSFET transistorsare flicker and thermal noisesThe flicker noise is modeled asthe voltage source in series with the gate of value

1198812119892(119891) =

119870

119882119871119862119900119909119891 (2)

where the constant 119870 is dependent on device characteristicsand can vary widely for different devices in the same processThe variables 119882 119871 and 119862

119900119909represent the width length

and gate capacitance per unit area respectively The flickernoise is inversely proportional to the transistor area119882119871 Inother words a larger device reduces this noise The noiseprocess of thermal noise is random The MOSFET thermalnoise includes threemain noise sources as shown in Figure 3distributed gate resistance noise (1198812rg) drain current noise(1198942nd) and gate current noise (1198942nd)

The thermal noise source mainly comes from the draincurrent noise and gate-induced noise of the transistorsMultifinger layout technology is applied to reduce the gate-induced noise According to the MOSFET noise analysis in[17 18] we must express the noise figure in a way that explic-itly considers power consumption In order to determine thewidth of119872

1and find out the best NF we use the dependency

among noise factor (119865) power dissipation (PD) and qualityfactor (119876

119871) to find out the optimal value [17] Based on

these parameters the related curves NF (dB) versus 119876119871 are

plotted in Figure 4 by MATLAB simulation software Thereis a compromise between noise performance and power gainin Figure 4 Therefore the 119876

119871can be determined to be 4 as

PD equals to 10mW The optimal width of MOSFET can beobtained by [17 18]

119882opt =3

2

1

120596119900119871119862119900119909119877119878119876119871

asymp1

3120596119900119871119862119900119909119877119878

(3)

where 120596119900is the operation frequency 119871 is the gate length and

119862ox is the gate capacitance per unit area Equation (3) showsthat the optimal width for 119872

1is 240120583m for 120596o = 34GHz

119871 = 018 120583m 119862119900119909= 862 Fm2 119877

119878= 50Ω and 119876

119871= 4

Figure 5 shows the proposed UWB LNA schematicThe current sharing cascode amplifier provides low-powercharacteristics with low voltage supply Note that 119872

2is

the common-gate stage of the cascode configuration whicheliminates theMiller effect and provides better isolation fromthe output return signal [18 19] The passive components 119862

1

1198622 1198623 1198771 and 119871

1are adopted for the matching network

at the input to resonate over the entire frequency band Theresistors 119877

2and 119877

119892are used to provide bias voltage for the

transistors 1198721and 119872

2 The capacitor 119862

4provides signal

coupling between the two stages The capacitor 1198625bypasses

the ac current to the ground and avoids coupling to the firststage This affects gain flatness therefore it is possible toprovide an ideal ac ground but gain flatness is not affectedin the design 119871

2is the inductor load of the first stage The

output matching network is composed of 1198713 1198714 1198626 and 119862

7

In the cascode stages the first stagersquos noise figure contributionis more than the second stagersquos The first stagersquos transistorsize and bias point should be optimized for the low NF [17]The second stagersquos transistor size and bias point should beoptimized for high linearity The cascode LNA design is fullof trade-offs between optimal gain low noise figure inputand output matching linearity and power consumptionMoreover to avoid oscillation the parasitic couplings haveto be minimized [16] The sizes of the devices of the LNA areshown in Table 1

3 Proposed Wideband MatchingNetwork Analysis

31 Input Impedance Matching As shown in Figure 2 in theconventional narrow band LNA design the input impedanceof the input stage (119872

1) can be written as

119885in = 119904 (119871119892+ 119871119878) +

1

119904119862gs1+ (

1198921198981

119862gs1)119871119904 (4)

where 119862gs1 is the gate-source capacitance and 1198921198981

is thetransconductance of the input transistor119872

1 We choose ap-

propriate values of inductance (119871119892+119871119904) and capacitance119862gs1

which resonate at a certain frequency The real term can bemade equal to 50Ω Equation (3) determines the width of119872

1

and finds the best NF For wideband design it is difficult tolet the imaginary part of (4) remain zero for a wide rangeTo discuss UWB input impedance matching we need toconsider the standard form of the second-order filter

119879 (119878) =11988621198782+ 1198861119878 + 119886119900

1198782 + 119878 (120596119900119876in) + 120596

2

119900

=11988621198782+ 1198861119878 + 119886119900

1198782 + 119878B + 1205962119900

(5)

Active and Passive Electronic Components 3

3432 3960 4488(MHz)

5016 5808 6336 6864 7392 7920 8448 8976 9504 10032

Group A Group B Group C Group D

Figure 1 The IEEE 802153a spectrum

Table 1 Device sizes of the proposed UWB LNA

Device 1198621

(pF)1198622

(pF)1198711

(nH)1198623

(pF)1198771

(kΩ)119877119892

(kΩ)1198624

(pF)1198712

(nH)1198625

(pF)1198772

(kΩ)1198713

(nH)1198714

(nH)1198626

(pF)1198627

(pF)1198721-1198722

(120583m)Size 951 011 091 76 014 384 57 553 57 384 119 059 016 123 240018

M2

M1

Ls

Lg

Vbias

Zin

Figure 2 Cascode low noise amplifier

V2rg

gsi2ng i2ndCgs rogmV

Rg

Figure 3 CMOS noise model

where 1198862 1198861 and 119886

119900are numerator coefficients determining

the type of second-order filter function 120596119900is called the pole

frequency119876in is termed the pole factor and 119861 is called band-width According to 119861 = 120596

119900119876in the bandwidth is inversely

proportional to the 119876in if the value of the pole frequency 120596119900is fixed

0 1 2 3 4 5 6 7 800

05

10

15

20

25

30

35

NF

(dB)

PD = 5 mWPD = 10 mW

PD = 15 mWPD = 20 mW

QL

Figure 4 Noise Figure versus 119876119871for several power dissipations at

34GHz

R2

L3 L4 C7

C6

C5

C4L2

M2

M1

C3

R1C2

C1 L1

Rg

Vbias

RFin

RFout

VDD

Figure 5 Schematic of the proposed UWB LNA

4 Active and Passive Electronic Components

Rs = 50Ω

Vs

Zin

C1

C2

C3

R1

L1Cgs1

Figure 6 The proposed wideband matchingapproximate first stageinput matching small-signal equivalent circuit

Figure 6 shows the proposed wideband matching and asmall-signal equivalent circuit in the first stage The inputimpedance of the RLC network can be written as

119885in =1

1198781198621

+1

1198781198622

[1198781198711+ (1198771+

1

1198781198623

) 1

119878119862gs1]

= 119877in + 119895119883in

(6)

Because 1198623capacitance value is much larger than 119862gs(1198623 ≫

119862gs) the (1198771 + (11198781198623))(1119878119862gs1) will approximate (119877

1+

(11198781198623))

119885in =1

1198781198621

+1

1198781198622

(1198781198711+ (1198771+

1

1198781198623

)) = 119877in + 119895119883in

119885in asymp(119878 + (119877

11198711)) ((119862

1+ 1198622) 11986211198622)

1198782 + 119878 (11987711198711) + ((119862

2+ 1198623) 1198711(11986221198623))

(7)

where 119862gs1 is the gate-source capacitance of1198721 This equiv-alent circuit can roughly be an RLC second-order filterstructure In (7) the quality factor of the filter circuit can begiven by

119891119900asymp

1

2120587radic

1198622+ 1198623

1198711(11986221198623) (8)

119876 asymp1

1198771

radic1198711(1198622+ 1198623)

11986221198623

(9)

where 119891119900is the resonant frequency In (8) and (9) to obtain

broader bandwidth a low-quality factor needs to be added tothe passive devices The narrowband LNA can be convertedinto a wideband amplifier by properly selecting passivedevices According to (8) the capacitors 119862

2 1198623and inductor

1198711will be optimized at the resonant frequency We use the

CAD of Agilent ADS to analyze the circuit performanceof input impedance matching According to (8) and (9) 119891

119900

is inversely proportional to 1198711while 119876 is proportional to

1198711 Therefore bandwidth is inversely proportional to 119871

1

Figure 7 is fixed as 1198621(951 pF) 119862

2(011 pF) 119862

3(76 pF) and

1198771(140Ω) showing that the resonant frequency is inversely

proportional to the 1198711 Bandwidth is inversely proportional

3 4 5 6 7 8 9 102 11Frequency (GHz)

minus20

minus25

minus15

minus10

minus5

0

L1 = 13nHL1 = 115nHL1 = 091 nH

S 11

(dB)

Figure 7 Fixed 1198621 1198622 1198623and 119877

1and simulated 119871

1variation input

reflection coefficient

to 1198711 According to (9) the input matching circuit uses 119877

1

and 1198623to reduce the number of inductors and make the 119876

factor smaller 119876 is inversely proportional to 1198771 Therefore

bandwidth is proportional to 1198771 Figure 8 is fixed as 119862

1

(951 pF) 1198622(011 pF) 119862

3(76 pF) and 119871

1(091 nH) showing

that the bandwidth is proportional to the 1198771 Figure 9 is fixed

as 1198621(951 pF) 119862

2(011 pF) 119862

3(76 pF) and 119871

1(091 nH)

showing that the noise figure is inversely proportional to the1198771 119891119900is inversely proportional to 119862

3while 119876 is inversely

proportional to 1198623 Therefore the bandwidth is proportional

to the 1198623 Figures 10 and 11 are fixed as 119862

1(951 pF) 119862

2

(011 pF) 1198771(140Ω) and 119871

1(091 nH) showing that the

bandwidth and noise figure are proportional to the 1198623 The

size of the transistors 1198721 1198771 and 119862

3must be carefully

selected There is a trade-off in the design between widebandmatching and the noise figure On the other hand thedevice size must yield a sufficient noise performance andpower gain As can be seen the second-order filter of1198711(091 nH) 119877

1(140Ω) and 119862

3(76 pF) is employed as the

inputmatching networkThe inputmatching network has lesscomplexity and better reflection coefficient from 3GHz to 10GHz

32 Output Impedance Matching Low-noise amplifiers relyon output impedance matching to achieve maximum powergain A source follower technique has been widely used toprovide wideband output matching Output impedance issimilar to 1119892

119898119904 in which 119892

119898119904is a gate-source transconduc-

tance of the source follower [4 10] However it consumesmore power For these reasons we introduce an outputimpedance matching method suitable for wideband LNAdesign Figure 12 shows the approximate output impedancesmall-signal equivalent circuit The output impedance of theRLC network can be written as

Active and Passive Electronic Components 5

1 2 3 4 5 6 7 8 9 100 11Frequency (GHz)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

minus20

minus15

minus10

minus5

0

S 11

(dB)

Figure 8 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation input

reflection coefficient

1 2 3 4 5 6 7 8 9 100 11

3456789

2

10

Frequency (GHz)

NF

(dB)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

Figure 9 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation noise

figure

119885out = ((1199031198892

1

1198781198621198892

1198781198713) + 119878119871

4)

1

1198781198626

+1

1198781198627

= 119877out + 119895119883out

(10)

119885out

asymp(119878+(1119903

11988921198621198892)) ((119862

7+1198626) 11986261198627)

1198782+119878 (111990311988921198621198892)+((119862

61198714+11986261198713+11986211988921198713) 1198621198892119862611987131198714)

(11)

where 1198621198892

is the parasitic capacitance of 1198722at the drain

node and 1199031198892

is the resistor of 1198722at the drain node This

equivalent circuit can approximately be an RLC second-order

1 2 3 4 5 6 7 8 90 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

C3 = 76 pFC3 = 095 pFC3 = 015 pF

S 11

(dB)

Figure 10 Fixed119862111986221198771 and 119871

1and simulated119862

3variation input

reflection coefficient

1 2 3 4 5 6 7 8 90

3456789

2

10

10 11

NF

(dB)

Frequency (GHz)C3 = 76 pFC3 = 095 pFC3 = 015 pF

Figure 11 Fixed119862111986221198771 and 119871

1and simulated119862

3variation noise

figure

filter structure In (11) the quality factor of the filter circuitcan be given by

119891119900asymp

1

2120587radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(12)

119876 asymp 11990311988921198621198892radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(13)

where 119891119900is the resonant frequency In (12) and (13) in order

to obtain broader bandwidth a low-quality factor needs tobe added to passive devices 119891

119900is inversely proportional to

1198621198892

while bandwidth is inversely proportional to 11990311988921198621198892 The

1198621198892capacitance is proportional to the width of the transistor

The 1199031198892

resistance is inversely proportional to the width ofthe transistor UWB system uses a specific frequency band

6 Active and Passive Electronic Components

Rs = 50 Ω

outZ

C7L4

C6L3

rd2 Cd2

Figure 12 The wideband output matching small-signal equivalentcircuit

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus30

minus20

minus25

minus15

minus10

minus5

0

S 22

(dB)

M2 = 160018 120583mM2 = 240018 120583mM2 = 320018 120583m

Figure 13 Fixed 1198626 1198627 1198713and 119871

4 simulated119872

2variation output

reflection coefficient

(31ndash5GHz or 31ndash106GHz) to access data Generally theresults of UWB return loss are less than minus10 dB The powerconsumption is proportional to the width of the transistorAccording to (13) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting 119872

2 This

provides good wideband matching and flat gain Figure 13is fixed as 119862

6(016 pF) 119862

7(123 pF) 119871

3(119 nH) and 119871

4

(059 nH) showing that the simulated 1198722variation output

reflection coefficient As can be seen the second-order filterof 1198722(240018 120583m) is employed as the output matching

network According to (12) the capacitor 1198626and inductor

1198713-1198714will be optimized at the resonant frequency Figure 14

is fixed as 1198626(016 pF) 119862

7(123 pF) 119872

2(240018120583m)

and 1198714(059 nH) showing that the resonant frequency is

inversely proportional to the 1198713 The output network has less

complexity and a better reflected coefficient from 31 GHz to106GHz

33 Gain At a high frequency the current gain of commonsource configured transistor 119872

1is 1198921198981119904119862gs1 [11] Figure 15

shows the approximate first stage 1198721load of the simplified

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

S 22

(dB)

L3 = 119nHL3 = 129nH

L3 = 109nH

Figure 14 Fixed 1198626 1198627 1198722 and 119871

4and simulated 119871

3variation

output reflection coefficient

Id1

Vout1

C5

L2

Cd11gm2

Figure 15 The first stage simplified small-signal equivalent circuit

small-signal equivalent circuit [12] The first stage load canbe approximated as

119885load1 asymp1

119878119871211986251198621198891

sdot119878211987121198625+ 11987811989211989821198712+ 1

1198782 + 119878 (11989211989821198625) + ((119862

5+ 1198621198891) (1198712(11986251198621198891)))

(14)

The transfer function of the input matching network is119885in(119904) The first stage voltage gain can be approximated as

1198601198811

asymp minus1198921198981119885in

119878119862gs1119877119904sdot 119885Load1 (15)

where 119877119904is the source resistance in which voltage gain is

determined by the load inductor 1198712 the total capacitance at

the drain of 1198721 and bypass capacitor 119862

5 The second stage

voltage gain can be estimated as

1198601198812

asymp minus1198921198982

119878119862gs2119877in2sdot 119885out (16)

Active and Passive Electronic Components 7

1 2 3 4 5 6 7 8 9 10

10

12

110

2

4

6

8

0

Frequency (GHz)

Δ

K

Δ K

Figure 16 119870 (stability factor) and Δ simulation result

where 119877in2 is the input resistance at the gate of1198722 and 119885outis the output load impedance The LNA voltage gain can beapproximated as

119860119881asymp11989211989811198921198982119885in119885Load1119885out

1198782119862gs1119862gs2119877119904119877in2 (17)

According to (17) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting the devicesize This provides good wideband matching noise figureoptimization design and flat gain

34 Stability Amplifier stability is important to preventnatural oscillation The stability can be determined by the 119878-parameters input and output matching network and circuitterminations The simpler tests show whether or not a devicecan be unconditionally stable [20] One of these is the 119870-Δtest which shows if a device can be unconditionally stable ifRolletrsquos condition is defined as

119870 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

minus100381610038161003816100381611987822

10038161003816100381610038162

+ |Δ|2

210038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1

|Δ| =10038161003816100381610038161198781111987822 minus 1198781211987821

1003816100381610038161003816 lt 1

(18)

The 119870-Δ test cannot be used to compare the relative sta-bility of twoormore devices because it involves constraints ontwo separate parameters A new criterion has been proposedwhich combines the 119878-parameters in a test involving only asingle parameter 120583 defined as

120583 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

100381610038161003816100381611987822 minus Δ119878lowast

11

1003816100381610038161003816 +10038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1 (19)

The performance of LNA is simulated using the advancedesign system (ADS) To consider the stability of the LNAthe119870 factor andΔ should be considered Figure 16 shows thatthe 119870 factor is always larger than 1 and the Δ is smaller than1 all the time Hence this circuit is stable for all frequency

1 2 3 4 5 6 7 8 9 10 11

11

0Frequency (GHz)

3

5

7

9

1

120583load120583source

120583lo

ad120583

sour

ce

Figure 17 120583 factor simulation result

RFin RFout

Vbias

DC power supply(HP 4142B)

Network analyzer(HP 8510C)

S-parameters test set(HP 8517B)

VDD

Figure 18 Test setup used for the LNA 119878-parameter measurements

bands There is another way to diagnose whether the LNAis unconditionally stable The 120583 factor at input and outputshould be larger than 1 in all frequency bands as shown inFigure 17

4 Measurement Results

On-wafer measurement is performed by an HP 8510C net-work analyzer and an HP 8517B is to test the 119878-parameter asshown in Figure 18 A network analyzer is used to measurethe frequency response and input matching of the LNAThe input and output impedance matchings are both 50ΩTo ensure that the LNA still provides a conversion gainwhen process deviation occurs the other process cornersnamely typical-NMOS typical-PMOS (TT) fast-NMOS fast-PMOS (FF) and slow-NMOS slow-PMOS (SS) are also usedto simulate this LNA The results are shown in Figure 19Figure 19 shows the measurement forward gain (119878

21) from

3 to 56GHz the measured gain is about 7ndash10 dB Themeasurement data is 6 dB lower than the pre-simulation data

8 Active and Passive Electronic Components

201816141210

86420

28 33 38 43 48 53 58Frequency (GHz)

S 21

(dB)

Pre-simulation TTPost-simulation FFPost-simulation TT

Post-simulation SSMeasurement

Figure 19 Simulation and measurement results of conversion gainversus frequency of the proposed LNA RF stands for simulationresults of modifiedMOSFET RFmodel FF TT and SS represent thesimulation results of fast-NMOS fast-PMOS typical-NMOS typical-PMOS and slow-NMOS slow-PMOS process corners

minus10

0

minus20

minus30

minus40

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 11

(dB)

Figure 20 Measured and simulated input reflection coefficient

because of the process drift and parasitic effect in the layoutThe result of conversion gain measurement is situated in thepost-simulation (SS) process corner

Figure 20 shows the input return loss (S11) from 3 to

11 GHz and the measured S11

is less than minus9 dB Figure 21shows output return loss (S

22) from 3 to 75GHz and the

measured S22is less thanminus8 dB Figure 22 shows reverse isola-

tion (S12) from 3 to 11 GHz and the measured S

12is less than

minus40 dB The simulation measured input of 1 dB compressionpoint at 53 GHz shown in Figure 23 is about minus8 dB Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz and the IIP

3is 2 dBm at 53 GHz The

IM3 is measured using two Agilnet E8247C continuous wave

minus10

minus15

minus20

minus25

minus30

minus5

0

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 22

(dB)

Figure 21 Measured and simulated output reflection coefficient

minus10

minus20

minus30

minus40

minus50

minus60

minus70

minus80

MeasurementSimulation

3 4 5 6 7 8 9 102 11

0

Frequency (GHz)

S 12

(dB)

Figure 22 Measured and simulated reverse isolation

minus40 minus35 minus30 minus25 minus20 minus15 minus10 minus5 0 5 100

1

2

3

4

5

6

7

8

9

Gai

n (d

B)

Input power (dBm)

P1dB

Figure 23 Measured input 1 db compression point at 53 GHz

Active and Passive Electronic Components 9

Table 2 Recently reported performance of UWB LNAs

Reference Process CMOS(120583m) 119878

11(dB) Avg gain

(db) Freq (GHz) NF (dB) IIP3(dBm) Die area

(mm2)Power(mW) Topology

[4] 018120583m CMOS ltminus78 119 2minus65 41minus46 4 (4GHz) 088 27 Feedback[7] 06120583m CMOS ltminus7 61 05minus55 54minus82 NA 112 834 Distributed[8] 018 120583m CMOS ltminus20 10 0minus11 31minus61 NA 144 196 Distributed[9] 018120583m CMOS ltminus12 14 3minus6 47minus67 minus5 (45GHz) 11 594 Distributed[10] 018120583m CMOS ltminus9 98 2ndash46 23minus5 minus7 (4GHz) 09 126lowast Feedback[11] 018120583m CMOS ltminus99 93 24minus95 4minus9 minus67 (6GHz) 11 9lowast Chebyshev filter[12] 018120583m CMOS ltminus10 86 24minus94 41minus10 minus35 (6GHz) 176 71lowast Chebyshev filter

[13] 018120583m CMOS ltminus5 191 28minus72 32minus38 minus1 (6GHz) 163 32 Feedback networksynthesis

[14] 018120583m CMOS ltminus10 17 2ndash11 38 NA 0635 1056lowastlowast Feedback[15] 015 120583mHEMT NA 18 085ndash1335 25 NA 1162 70lowastlowast FeedbackThis work 018120583m CMOS ltminus9 9 3ndash56 46ndash53 2 (53 GHz) 08 9 ProposedlowastOnly core LNA lowastlowastSimulation

minus25 minus20 minus15 minus10 minus5 0 5minus80

minus70

minus60

minus50

minus40

minus30

minus20

minus10

0

10

Input power (dBm)

Out

put p

ower

(dBm

)

FundamentalIMD3

IIP3

Figure 24 Measured IIP3at 53 GHz

(CW) generators and an Agilent E4407B spectrum analyzerA noise figure (NF) is measured using an Agilent N8975ANF meter with an Agilent 346C noise source The simulationnoise figure minimum is 41 dB as shown in Figure 25 Dueto the parasitic effect in the layout the minimum of themeasurement is 46 dB Figure 26 shows the micrograph ofthe LNA Table 2 summarizes the measurement results andcompares them with previous literature In the proposedLNA topology the simplified RLC input matching networkand the current-reused configuration benefit from the low-power design The distributed amplifier [7ndash9] requires highpower consumption and a large chip area Compared withthe Chebyshev filter and feedback network synthesis [11ndash13] the proposed input matching network which has onlyone spiral inductor simplifies the circuit complexity and

30 35 40 45 50 55 60 65 70 75 800

1

2

3

4

5

6

7

8

9

10N

oise

figu

re (d

B)

Measured NFSimulated NF

Frequency (GHz)

Figure 25 Simulation and measurement NF

Figure 26Micrograph of the proposed LNA (size 103times 078mm2)

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

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DistributedSensor Networks

International Journal of

Active and Passive Electronic Components 3

3432 3960 4488(MHz)

5016 5808 6336 6864 7392 7920 8448 8976 9504 10032

Group A Group B Group C Group D

Figure 1 The IEEE 802153a spectrum

Table 1 Device sizes of the proposed UWB LNA

Device 1198621

(pF)1198622

(pF)1198711

(nH)1198623

(pF)1198771

(kΩ)119877119892

(kΩ)1198624

(pF)1198712

(nH)1198625

(pF)1198772

(kΩ)1198713

(nH)1198714

(nH)1198626

(pF)1198627

(pF)1198721-1198722

(120583m)Size 951 011 091 76 014 384 57 553 57 384 119 059 016 123 240018

M2

M1

Ls

Lg

Vbias

Zin

Figure 2 Cascode low noise amplifier

V2rg

gsi2ng i2ndCgs rogmV

Rg

Figure 3 CMOS noise model

where 1198862 1198861 and 119886

119900are numerator coefficients determining

the type of second-order filter function 120596119900is called the pole

frequency119876in is termed the pole factor and 119861 is called band-width According to 119861 = 120596

119900119876in the bandwidth is inversely

proportional to the 119876in if the value of the pole frequency 120596119900is fixed

0 1 2 3 4 5 6 7 800

05

10

15

20

25

30

35

NF

(dB)

PD = 5 mWPD = 10 mW

PD = 15 mWPD = 20 mW

QL

Figure 4 Noise Figure versus 119876119871for several power dissipations at

34GHz

R2

L3 L4 C7

C6

C5

C4L2

M2

M1

C3

R1C2

C1 L1

Rg

Vbias

RFin

RFout

VDD

Figure 5 Schematic of the proposed UWB LNA

4 Active and Passive Electronic Components

Rs = 50Ω

Vs

Zin

C1

C2

C3

R1

L1Cgs1

Figure 6 The proposed wideband matchingapproximate first stageinput matching small-signal equivalent circuit

Figure 6 shows the proposed wideband matching and asmall-signal equivalent circuit in the first stage The inputimpedance of the RLC network can be written as

119885in =1

1198781198621

+1

1198781198622

[1198781198711+ (1198771+

1

1198781198623

) 1

119878119862gs1]

= 119877in + 119895119883in

(6)

Because 1198623capacitance value is much larger than 119862gs(1198623 ≫

119862gs) the (1198771 + (11198781198623))(1119878119862gs1) will approximate (119877

1+

(11198781198623))

119885in =1

1198781198621

+1

1198781198622

(1198781198711+ (1198771+

1

1198781198623

)) = 119877in + 119895119883in

119885in asymp(119878 + (119877

11198711)) ((119862

1+ 1198622) 11986211198622)

1198782 + 119878 (11987711198711) + ((119862

2+ 1198623) 1198711(11986221198623))

(7)

where 119862gs1 is the gate-source capacitance of1198721 This equiv-alent circuit can roughly be an RLC second-order filterstructure In (7) the quality factor of the filter circuit can begiven by

119891119900asymp

1

2120587radic

1198622+ 1198623

1198711(11986221198623) (8)

119876 asymp1

1198771

radic1198711(1198622+ 1198623)

11986221198623

(9)

where 119891119900is the resonant frequency In (8) and (9) to obtain

broader bandwidth a low-quality factor needs to be added tothe passive devices The narrowband LNA can be convertedinto a wideband amplifier by properly selecting passivedevices According to (8) the capacitors 119862

2 1198623and inductor

1198711will be optimized at the resonant frequency We use the

CAD of Agilent ADS to analyze the circuit performanceof input impedance matching According to (8) and (9) 119891

119900

is inversely proportional to 1198711while 119876 is proportional to

1198711 Therefore bandwidth is inversely proportional to 119871

1

Figure 7 is fixed as 1198621(951 pF) 119862

2(011 pF) 119862

3(76 pF) and

1198771(140Ω) showing that the resonant frequency is inversely

proportional to the 1198711 Bandwidth is inversely proportional

3 4 5 6 7 8 9 102 11Frequency (GHz)

minus20

minus25

minus15

minus10

minus5

0

L1 = 13nHL1 = 115nHL1 = 091 nH

S 11

(dB)

Figure 7 Fixed 1198621 1198622 1198623and 119877

1and simulated 119871

1variation input

reflection coefficient

to 1198711 According to (9) the input matching circuit uses 119877

1

and 1198623to reduce the number of inductors and make the 119876

factor smaller 119876 is inversely proportional to 1198771 Therefore

bandwidth is proportional to 1198771 Figure 8 is fixed as 119862

1

(951 pF) 1198622(011 pF) 119862

3(76 pF) and 119871

1(091 nH) showing

that the bandwidth is proportional to the 1198771 Figure 9 is fixed

as 1198621(951 pF) 119862

2(011 pF) 119862

3(76 pF) and 119871

1(091 nH)

showing that the noise figure is inversely proportional to the1198771 119891119900is inversely proportional to 119862

3while 119876 is inversely

proportional to 1198623 Therefore the bandwidth is proportional

to the 1198623 Figures 10 and 11 are fixed as 119862

1(951 pF) 119862

2

(011 pF) 1198771(140Ω) and 119871

1(091 nH) showing that the

bandwidth and noise figure are proportional to the 1198623 The

size of the transistors 1198721 1198771 and 119862

3must be carefully

selected There is a trade-off in the design between widebandmatching and the noise figure On the other hand thedevice size must yield a sufficient noise performance andpower gain As can be seen the second-order filter of1198711(091 nH) 119877

1(140Ω) and 119862

3(76 pF) is employed as the

inputmatching networkThe inputmatching network has lesscomplexity and better reflection coefficient from 3GHz to 10GHz

32 Output Impedance Matching Low-noise amplifiers relyon output impedance matching to achieve maximum powergain A source follower technique has been widely used toprovide wideband output matching Output impedance issimilar to 1119892

119898119904 in which 119892

119898119904is a gate-source transconduc-

tance of the source follower [4 10] However it consumesmore power For these reasons we introduce an outputimpedance matching method suitable for wideband LNAdesign Figure 12 shows the approximate output impedancesmall-signal equivalent circuit The output impedance of theRLC network can be written as

Active and Passive Electronic Components 5

1 2 3 4 5 6 7 8 9 100 11Frequency (GHz)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

minus20

minus15

minus10

minus5

0

S 11

(dB)

Figure 8 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation input

reflection coefficient

1 2 3 4 5 6 7 8 9 100 11

3456789

2

10

Frequency (GHz)

NF

(dB)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

Figure 9 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation noise

figure

119885out = ((1199031198892

1

1198781198621198892

1198781198713) + 119878119871

4)

1

1198781198626

+1

1198781198627

= 119877out + 119895119883out

(10)

119885out

asymp(119878+(1119903

11988921198621198892)) ((119862

7+1198626) 11986261198627)

1198782+119878 (111990311988921198621198892)+((119862

61198714+11986261198713+11986211988921198713) 1198621198892119862611987131198714)

(11)

where 1198621198892

is the parasitic capacitance of 1198722at the drain

node and 1199031198892

is the resistor of 1198722at the drain node This

equivalent circuit can approximately be an RLC second-order

1 2 3 4 5 6 7 8 90 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

C3 = 76 pFC3 = 095 pFC3 = 015 pF

S 11

(dB)

Figure 10 Fixed119862111986221198771 and 119871

1and simulated119862

3variation input

reflection coefficient

1 2 3 4 5 6 7 8 90

3456789

2

10

10 11

NF

(dB)

Frequency (GHz)C3 = 76 pFC3 = 095 pFC3 = 015 pF

Figure 11 Fixed119862111986221198771 and 119871

1and simulated119862

3variation noise

figure

filter structure In (11) the quality factor of the filter circuitcan be given by

119891119900asymp

1

2120587radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(12)

119876 asymp 11990311988921198621198892radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(13)

where 119891119900is the resonant frequency In (12) and (13) in order

to obtain broader bandwidth a low-quality factor needs tobe added to passive devices 119891

119900is inversely proportional to

1198621198892

while bandwidth is inversely proportional to 11990311988921198621198892 The

1198621198892capacitance is proportional to the width of the transistor

The 1199031198892

resistance is inversely proportional to the width ofthe transistor UWB system uses a specific frequency band

6 Active and Passive Electronic Components

Rs = 50 Ω

outZ

C7L4

C6L3

rd2 Cd2

Figure 12 The wideband output matching small-signal equivalentcircuit

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus30

minus20

minus25

minus15

minus10

minus5

0

S 22

(dB)

M2 = 160018 120583mM2 = 240018 120583mM2 = 320018 120583m

Figure 13 Fixed 1198626 1198627 1198713and 119871

4 simulated119872

2variation output

reflection coefficient

(31ndash5GHz or 31ndash106GHz) to access data Generally theresults of UWB return loss are less than minus10 dB The powerconsumption is proportional to the width of the transistorAccording to (13) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting 119872

2 This

provides good wideband matching and flat gain Figure 13is fixed as 119862

6(016 pF) 119862

7(123 pF) 119871

3(119 nH) and 119871

4

(059 nH) showing that the simulated 1198722variation output

reflection coefficient As can be seen the second-order filterof 1198722(240018 120583m) is employed as the output matching

network According to (12) the capacitor 1198626and inductor

1198713-1198714will be optimized at the resonant frequency Figure 14

is fixed as 1198626(016 pF) 119862

7(123 pF) 119872

2(240018120583m)

and 1198714(059 nH) showing that the resonant frequency is

inversely proportional to the 1198713 The output network has less

complexity and a better reflected coefficient from 31 GHz to106GHz

33 Gain At a high frequency the current gain of commonsource configured transistor 119872

1is 1198921198981119904119862gs1 [11] Figure 15

shows the approximate first stage 1198721load of the simplified

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

S 22

(dB)

L3 = 119nHL3 = 129nH

L3 = 109nH

Figure 14 Fixed 1198626 1198627 1198722 and 119871

4and simulated 119871

3variation

output reflection coefficient

Id1

Vout1

C5

L2

Cd11gm2

Figure 15 The first stage simplified small-signal equivalent circuit

small-signal equivalent circuit [12] The first stage load canbe approximated as

119885load1 asymp1

119878119871211986251198621198891

sdot119878211987121198625+ 11987811989211989821198712+ 1

1198782 + 119878 (11989211989821198625) + ((119862

5+ 1198621198891) (1198712(11986251198621198891)))

(14)

The transfer function of the input matching network is119885in(119904) The first stage voltage gain can be approximated as

1198601198811

asymp minus1198921198981119885in

119878119862gs1119877119904sdot 119885Load1 (15)

where 119877119904is the source resistance in which voltage gain is

determined by the load inductor 1198712 the total capacitance at

the drain of 1198721 and bypass capacitor 119862

5 The second stage

voltage gain can be estimated as

1198601198812

asymp minus1198921198982

119878119862gs2119877in2sdot 119885out (16)

Active and Passive Electronic Components 7

1 2 3 4 5 6 7 8 9 10

10

12

110

2

4

6

8

0

Frequency (GHz)

Δ

K

Δ K

Figure 16 119870 (stability factor) and Δ simulation result

where 119877in2 is the input resistance at the gate of1198722 and 119885outis the output load impedance The LNA voltage gain can beapproximated as

119860119881asymp11989211989811198921198982119885in119885Load1119885out

1198782119862gs1119862gs2119877119904119877in2 (17)

According to (17) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting the devicesize This provides good wideband matching noise figureoptimization design and flat gain

34 Stability Amplifier stability is important to preventnatural oscillation The stability can be determined by the 119878-parameters input and output matching network and circuitterminations The simpler tests show whether or not a devicecan be unconditionally stable [20] One of these is the 119870-Δtest which shows if a device can be unconditionally stable ifRolletrsquos condition is defined as

119870 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

minus100381610038161003816100381611987822

10038161003816100381610038162

+ |Δ|2

210038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1

|Δ| =10038161003816100381610038161198781111987822 minus 1198781211987821

1003816100381610038161003816 lt 1

(18)

The 119870-Δ test cannot be used to compare the relative sta-bility of twoormore devices because it involves constraints ontwo separate parameters A new criterion has been proposedwhich combines the 119878-parameters in a test involving only asingle parameter 120583 defined as

120583 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

100381610038161003816100381611987822 minus Δ119878lowast

11

1003816100381610038161003816 +10038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1 (19)

The performance of LNA is simulated using the advancedesign system (ADS) To consider the stability of the LNAthe119870 factor andΔ should be considered Figure 16 shows thatthe 119870 factor is always larger than 1 and the Δ is smaller than1 all the time Hence this circuit is stable for all frequency

1 2 3 4 5 6 7 8 9 10 11

11

0Frequency (GHz)

3

5

7

9

1

120583load120583source

120583lo

ad120583

sour

ce

Figure 17 120583 factor simulation result

RFin RFout

Vbias

DC power supply(HP 4142B)

Network analyzer(HP 8510C)

S-parameters test set(HP 8517B)

VDD

Figure 18 Test setup used for the LNA 119878-parameter measurements

bands There is another way to diagnose whether the LNAis unconditionally stable The 120583 factor at input and outputshould be larger than 1 in all frequency bands as shown inFigure 17

4 Measurement Results

On-wafer measurement is performed by an HP 8510C net-work analyzer and an HP 8517B is to test the 119878-parameter asshown in Figure 18 A network analyzer is used to measurethe frequency response and input matching of the LNAThe input and output impedance matchings are both 50ΩTo ensure that the LNA still provides a conversion gainwhen process deviation occurs the other process cornersnamely typical-NMOS typical-PMOS (TT) fast-NMOS fast-PMOS (FF) and slow-NMOS slow-PMOS (SS) are also usedto simulate this LNA The results are shown in Figure 19Figure 19 shows the measurement forward gain (119878

21) from

3 to 56GHz the measured gain is about 7ndash10 dB Themeasurement data is 6 dB lower than the pre-simulation data

8 Active and Passive Electronic Components

201816141210

86420

28 33 38 43 48 53 58Frequency (GHz)

S 21

(dB)

Pre-simulation TTPost-simulation FFPost-simulation TT

Post-simulation SSMeasurement

Figure 19 Simulation and measurement results of conversion gainversus frequency of the proposed LNA RF stands for simulationresults of modifiedMOSFET RFmodel FF TT and SS represent thesimulation results of fast-NMOS fast-PMOS typical-NMOS typical-PMOS and slow-NMOS slow-PMOS process corners

minus10

0

minus20

minus30

minus40

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 11

(dB)

Figure 20 Measured and simulated input reflection coefficient

because of the process drift and parasitic effect in the layoutThe result of conversion gain measurement is situated in thepost-simulation (SS) process corner

Figure 20 shows the input return loss (S11) from 3 to

11 GHz and the measured S11

is less than minus9 dB Figure 21shows output return loss (S

22) from 3 to 75GHz and the

measured S22is less thanminus8 dB Figure 22 shows reverse isola-

tion (S12) from 3 to 11 GHz and the measured S

12is less than

minus40 dB The simulation measured input of 1 dB compressionpoint at 53 GHz shown in Figure 23 is about minus8 dB Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz and the IIP

3is 2 dBm at 53 GHz The

IM3 is measured using two Agilnet E8247C continuous wave

minus10

minus15

minus20

minus25

minus30

minus5

0

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 22

(dB)

Figure 21 Measured and simulated output reflection coefficient

minus10

minus20

minus30

minus40

minus50

minus60

minus70

minus80

MeasurementSimulation

3 4 5 6 7 8 9 102 11

0

Frequency (GHz)

S 12

(dB)

Figure 22 Measured and simulated reverse isolation

minus40 minus35 minus30 minus25 minus20 minus15 minus10 minus5 0 5 100

1

2

3

4

5

6

7

8

9

Gai

n (d

B)

Input power (dBm)

P1dB

Figure 23 Measured input 1 db compression point at 53 GHz

Active and Passive Electronic Components 9

Table 2 Recently reported performance of UWB LNAs

Reference Process CMOS(120583m) 119878

11(dB) Avg gain

(db) Freq (GHz) NF (dB) IIP3(dBm) Die area

(mm2)Power(mW) Topology

[4] 018120583m CMOS ltminus78 119 2minus65 41minus46 4 (4GHz) 088 27 Feedback[7] 06120583m CMOS ltminus7 61 05minus55 54minus82 NA 112 834 Distributed[8] 018 120583m CMOS ltminus20 10 0minus11 31minus61 NA 144 196 Distributed[9] 018120583m CMOS ltminus12 14 3minus6 47minus67 minus5 (45GHz) 11 594 Distributed[10] 018120583m CMOS ltminus9 98 2ndash46 23minus5 minus7 (4GHz) 09 126lowast Feedback[11] 018120583m CMOS ltminus99 93 24minus95 4minus9 minus67 (6GHz) 11 9lowast Chebyshev filter[12] 018120583m CMOS ltminus10 86 24minus94 41minus10 minus35 (6GHz) 176 71lowast Chebyshev filter

[13] 018120583m CMOS ltminus5 191 28minus72 32minus38 minus1 (6GHz) 163 32 Feedback networksynthesis

[14] 018120583m CMOS ltminus10 17 2ndash11 38 NA 0635 1056lowastlowast Feedback[15] 015 120583mHEMT NA 18 085ndash1335 25 NA 1162 70lowastlowast FeedbackThis work 018120583m CMOS ltminus9 9 3ndash56 46ndash53 2 (53 GHz) 08 9 ProposedlowastOnly core LNA lowastlowastSimulation

minus25 minus20 minus15 minus10 minus5 0 5minus80

minus70

minus60

minus50

minus40

minus30

minus20

minus10

0

10

Input power (dBm)

Out

put p

ower

(dBm

)

FundamentalIMD3

IIP3

Figure 24 Measured IIP3at 53 GHz

(CW) generators and an Agilent E4407B spectrum analyzerA noise figure (NF) is measured using an Agilent N8975ANF meter with an Agilent 346C noise source The simulationnoise figure minimum is 41 dB as shown in Figure 25 Dueto the parasitic effect in the layout the minimum of themeasurement is 46 dB Figure 26 shows the micrograph ofthe LNA Table 2 summarizes the measurement results andcompares them with previous literature In the proposedLNA topology the simplified RLC input matching networkand the current-reused configuration benefit from the low-power design The distributed amplifier [7ndash9] requires highpower consumption and a large chip area Compared withthe Chebyshev filter and feedback network synthesis [11ndash13] the proposed input matching network which has onlyone spiral inductor simplifies the circuit complexity and

30 35 40 45 50 55 60 65 70 75 800

1

2

3

4

5

6

7

8

9

10N

oise

figu

re (d

B)

Measured NFSimulated NF

Frequency (GHz)

Figure 25 Simulation and measurement NF

Figure 26Micrograph of the proposed LNA (size 103times 078mm2)

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

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DistributedSensor Networks

International Journal of

4 Active and Passive Electronic Components

Rs = 50Ω

Vs

Zin

C1

C2

C3

R1

L1Cgs1

Figure 6 The proposed wideband matchingapproximate first stageinput matching small-signal equivalent circuit

Figure 6 shows the proposed wideband matching and asmall-signal equivalent circuit in the first stage The inputimpedance of the RLC network can be written as

119885in =1

1198781198621

+1

1198781198622

[1198781198711+ (1198771+

1

1198781198623

) 1

119878119862gs1]

= 119877in + 119895119883in

(6)

Because 1198623capacitance value is much larger than 119862gs(1198623 ≫

119862gs) the (1198771 + (11198781198623))(1119878119862gs1) will approximate (119877

1+

(11198781198623))

119885in =1

1198781198621

+1

1198781198622

(1198781198711+ (1198771+

1

1198781198623

)) = 119877in + 119895119883in

119885in asymp(119878 + (119877

11198711)) ((119862

1+ 1198622) 11986211198622)

1198782 + 119878 (11987711198711) + ((119862

2+ 1198623) 1198711(11986221198623))

(7)

where 119862gs1 is the gate-source capacitance of1198721 This equiv-alent circuit can roughly be an RLC second-order filterstructure In (7) the quality factor of the filter circuit can begiven by

119891119900asymp

1

2120587radic

1198622+ 1198623

1198711(11986221198623) (8)

119876 asymp1

1198771

radic1198711(1198622+ 1198623)

11986221198623

(9)

where 119891119900is the resonant frequency In (8) and (9) to obtain

broader bandwidth a low-quality factor needs to be added tothe passive devices The narrowband LNA can be convertedinto a wideband amplifier by properly selecting passivedevices According to (8) the capacitors 119862

2 1198623and inductor

1198711will be optimized at the resonant frequency We use the

CAD of Agilent ADS to analyze the circuit performanceof input impedance matching According to (8) and (9) 119891

119900

is inversely proportional to 1198711while 119876 is proportional to

1198711 Therefore bandwidth is inversely proportional to 119871

1

Figure 7 is fixed as 1198621(951 pF) 119862

2(011 pF) 119862

3(76 pF) and

1198771(140Ω) showing that the resonant frequency is inversely

proportional to the 1198711 Bandwidth is inversely proportional

3 4 5 6 7 8 9 102 11Frequency (GHz)

minus20

minus25

minus15

minus10

minus5

0

L1 = 13nHL1 = 115nHL1 = 091 nH

S 11

(dB)

Figure 7 Fixed 1198621 1198622 1198623and 119877

1and simulated 119871

1variation input

reflection coefficient

to 1198711 According to (9) the input matching circuit uses 119877

1

and 1198623to reduce the number of inductors and make the 119876

factor smaller 119876 is inversely proportional to 1198771 Therefore

bandwidth is proportional to 1198771 Figure 8 is fixed as 119862

1

(951 pF) 1198622(011 pF) 119862

3(76 pF) and 119871

1(091 nH) showing

that the bandwidth is proportional to the 1198771 Figure 9 is fixed

as 1198621(951 pF) 119862

2(011 pF) 119862

3(76 pF) and 119871

1(091 nH)

showing that the noise figure is inversely proportional to the1198771 119891119900is inversely proportional to 119862

3while 119876 is inversely

proportional to 1198623 Therefore the bandwidth is proportional

to the 1198623 Figures 10 and 11 are fixed as 119862

1(951 pF) 119862

2

(011 pF) 1198771(140Ω) and 119871

1(091 nH) showing that the

bandwidth and noise figure are proportional to the 1198623 The

size of the transistors 1198721 1198771 and 119862

3must be carefully

selected There is a trade-off in the design between widebandmatching and the noise figure On the other hand thedevice size must yield a sufficient noise performance andpower gain As can be seen the second-order filter of1198711(091 nH) 119877

1(140Ω) and 119862

3(76 pF) is employed as the

inputmatching networkThe inputmatching network has lesscomplexity and better reflection coefficient from 3GHz to 10GHz

32 Output Impedance Matching Low-noise amplifiers relyon output impedance matching to achieve maximum powergain A source follower technique has been widely used toprovide wideband output matching Output impedance issimilar to 1119892

119898119904 in which 119892

119898119904is a gate-source transconduc-

tance of the source follower [4 10] However it consumesmore power For these reasons we introduce an outputimpedance matching method suitable for wideband LNAdesign Figure 12 shows the approximate output impedancesmall-signal equivalent circuit The output impedance of theRLC network can be written as

Active and Passive Electronic Components 5

1 2 3 4 5 6 7 8 9 100 11Frequency (GHz)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

minus20

minus15

minus10

minus5

0

S 11

(dB)

Figure 8 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation input

reflection coefficient

1 2 3 4 5 6 7 8 9 100 11

3456789

2

10

Frequency (GHz)

NF

(dB)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

Figure 9 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation noise

figure

119885out = ((1199031198892

1

1198781198621198892

1198781198713) + 119878119871

4)

1

1198781198626

+1

1198781198627

= 119877out + 119895119883out

(10)

119885out

asymp(119878+(1119903

11988921198621198892)) ((119862

7+1198626) 11986261198627)

1198782+119878 (111990311988921198621198892)+((119862

61198714+11986261198713+11986211988921198713) 1198621198892119862611987131198714)

(11)

where 1198621198892

is the parasitic capacitance of 1198722at the drain

node and 1199031198892

is the resistor of 1198722at the drain node This

equivalent circuit can approximately be an RLC second-order

1 2 3 4 5 6 7 8 90 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

C3 = 76 pFC3 = 095 pFC3 = 015 pF

S 11

(dB)

Figure 10 Fixed119862111986221198771 and 119871

1and simulated119862

3variation input

reflection coefficient

1 2 3 4 5 6 7 8 90

3456789

2

10

10 11

NF

(dB)

Frequency (GHz)C3 = 76 pFC3 = 095 pFC3 = 015 pF

Figure 11 Fixed119862111986221198771 and 119871

1and simulated119862

3variation noise

figure

filter structure In (11) the quality factor of the filter circuitcan be given by

119891119900asymp

1

2120587radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(12)

119876 asymp 11990311988921198621198892radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(13)

where 119891119900is the resonant frequency In (12) and (13) in order

to obtain broader bandwidth a low-quality factor needs tobe added to passive devices 119891

119900is inversely proportional to

1198621198892

while bandwidth is inversely proportional to 11990311988921198621198892 The

1198621198892capacitance is proportional to the width of the transistor

The 1199031198892

resistance is inversely proportional to the width ofthe transistor UWB system uses a specific frequency band

6 Active and Passive Electronic Components

Rs = 50 Ω

outZ

C7L4

C6L3

rd2 Cd2

Figure 12 The wideband output matching small-signal equivalentcircuit

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus30

minus20

minus25

minus15

minus10

minus5

0

S 22

(dB)

M2 = 160018 120583mM2 = 240018 120583mM2 = 320018 120583m

Figure 13 Fixed 1198626 1198627 1198713and 119871

4 simulated119872

2variation output

reflection coefficient

(31ndash5GHz or 31ndash106GHz) to access data Generally theresults of UWB return loss are less than minus10 dB The powerconsumption is proportional to the width of the transistorAccording to (13) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting 119872

2 This

provides good wideband matching and flat gain Figure 13is fixed as 119862

6(016 pF) 119862

7(123 pF) 119871

3(119 nH) and 119871

4

(059 nH) showing that the simulated 1198722variation output

reflection coefficient As can be seen the second-order filterof 1198722(240018 120583m) is employed as the output matching

network According to (12) the capacitor 1198626and inductor

1198713-1198714will be optimized at the resonant frequency Figure 14

is fixed as 1198626(016 pF) 119862

7(123 pF) 119872

2(240018120583m)

and 1198714(059 nH) showing that the resonant frequency is

inversely proportional to the 1198713 The output network has less

complexity and a better reflected coefficient from 31 GHz to106GHz

33 Gain At a high frequency the current gain of commonsource configured transistor 119872

1is 1198921198981119904119862gs1 [11] Figure 15

shows the approximate first stage 1198721load of the simplified

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

S 22

(dB)

L3 = 119nHL3 = 129nH

L3 = 109nH

Figure 14 Fixed 1198626 1198627 1198722 and 119871

4and simulated 119871

3variation

output reflection coefficient

Id1

Vout1

C5

L2

Cd11gm2

Figure 15 The first stage simplified small-signal equivalent circuit

small-signal equivalent circuit [12] The first stage load canbe approximated as

119885load1 asymp1

119878119871211986251198621198891

sdot119878211987121198625+ 11987811989211989821198712+ 1

1198782 + 119878 (11989211989821198625) + ((119862

5+ 1198621198891) (1198712(11986251198621198891)))

(14)

The transfer function of the input matching network is119885in(119904) The first stage voltage gain can be approximated as

1198601198811

asymp minus1198921198981119885in

119878119862gs1119877119904sdot 119885Load1 (15)

where 119877119904is the source resistance in which voltage gain is

determined by the load inductor 1198712 the total capacitance at

the drain of 1198721 and bypass capacitor 119862

5 The second stage

voltage gain can be estimated as

1198601198812

asymp minus1198921198982

119878119862gs2119877in2sdot 119885out (16)

Active and Passive Electronic Components 7

1 2 3 4 5 6 7 8 9 10

10

12

110

2

4

6

8

0

Frequency (GHz)

Δ

K

Δ K

Figure 16 119870 (stability factor) and Δ simulation result

where 119877in2 is the input resistance at the gate of1198722 and 119885outis the output load impedance The LNA voltage gain can beapproximated as

119860119881asymp11989211989811198921198982119885in119885Load1119885out

1198782119862gs1119862gs2119877119904119877in2 (17)

According to (17) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting the devicesize This provides good wideband matching noise figureoptimization design and flat gain

34 Stability Amplifier stability is important to preventnatural oscillation The stability can be determined by the 119878-parameters input and output matching network and circuitterminations The simpler tests show whether or not a devicecan be unconditionally stable [20] One of these is the 119870-Δtest which shows if a device can be unconditionally stable ifRolletrsquos condition is defined as

119870 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

minus100381610038161003816100381611987822

10038161003816100381610038162

+ |Δ|2

210038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1

|Δ| =10038161003816100381610038161198781111987822 minus 1198781211987821

1003816100381610038161003816 lt 1

(18)

The 119870-Δ test cannot be used to compare the relative sta-bility of twoormore devices because it involves constraints ontwo separate parameters A new criterion has been proposedwhich combines the 119878-parameters in a test involving only asingle parameter 120583 defined as

120583 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

100381610038161003816100381611987822 minus Δ119878lowast

11

1003816100381610038161003816 +10038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1 (19)

The performance of LNA is simulated using the advancedesign system (ADS) To consider the stability of the LNAthe119870 factor andΔ should be considered Figure 16 shows thatthe 119870 factor is always larger than 1 and the Δ is smaller than1 all the time Hence this circuit is stable for all frequency

1 2 3 4 5 6 7 8 9 10 11

11

0Frequency (GHz)

3

5

7

9

1

120583load120583source

120583lo

ad120583

sour

ce

Figure 17 120583 factor simulation result

RFin RFout

Vbias

DC power supply(HP 4142B)

Network analyzer(HP 8510C)

S-parameters test set(HP 8517B)

VDD

Figure 18 Test setup used for the LNA 119878-parameter measurements

bands There is another way to diagnose whether the LNAis unconditionally stable The 120583 factor at input and outputshould be larger than 1 in all frequency bands as shown inFigure 17

4 Measurement Results

On-wafer measurement is performed by an HP 8510C net-work analyzer and an HP 8517B is to test the 119878-parameter asshown in Figure 18 A network analyzer is used to measurethe frequency response and input matching of the LNAThe input and output impedance matchings are both 50ΩTo ensure that the LNA still provides a conversion gainwhen process deviation occurs the other process cornersnamely typical-NMOS typical-PMOS (TT) fast-NMOS fast-PMOS (FF) and slow-NMOS slow-PMOS (SS) are also usedto simulate this LNA The results are shown in Figure 19Figure 19 shows the measurement forward gain (119878

21) from

3 to 56GHz the measured gain is about 7ndash10 dB Themeasurement data is 6 dB lower than the pre-simulation data

8 Active and Passive Electronic Components

201816141210

86420

28 33 38 43 48 53 58Frequency (GHz)

S 21

(dB)

Pre-simulation TTPost-simulation FFPost-simulation TT

Post-simulation SSMeasurement

Figure 19 Simulation and measurement results of conversion gainversus frequency of the proposed LNA RF stands for simulationresults of modifiedMOSFET RFmodel FF TT and SS represent thesimulation results of fast-NMOS fast-PMOS typical-NMOS typical-PMOS and slow-NMOS slow-PMOS process corners

minus10

0

minus20

minus30

minus40

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 11

(dB)

Figure 20 Measured and simulated input reflection coefficient

because of the process drift and parasitic effect in the layoutThe result of conversion gain measurement is situated in thepost-simulation (SS) process corner

Figure 20 shows the input return loss (S11) from 3 to

11 GHz and the measured S11

is less than minus9 dB Figure 21shows output return loss (S

22) from 3 to 75GHz and the

measured S22is less thanminus8 dB Figure 22 shows reverse isola-

tion (S12) from 3 to 11 GHz and the measured S

12is less than

minus40 dB The simulation measured input of 1 dB compressionpoint at 53 GHz shown in Figure 23 is about minus8 dB Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz and the IIP

3is 2 dBm at 53 GHz The

IM3 is measured using two Agilnet E8247C continuous wave

minus10

minus15

minus20

minus25

minus30

minus5

0

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 22

(dB)

Figure 21 Measured and simulated output reflection coefficient

minus10

minus20

minus30

minus40

minus50

minus60

minus70

minus80

MeasurementSimulation

3 4 5 6 7 8 9 102 11

0

Frequency (GHz)

S 12

(dB)

Figure 22 Measured and simulated reverse isolation

minus40 minus35 minus30 minus25 minus20 minus15 minus10 minus5 0 5 100

1

2

3

4

5

6

7

8

9

Gai

n (d

B)

Input power (dBm)

P1dB

Figure 23 Measured input 1 db compression point at 53 GHz

Active and Passive Electronic Components 9

Table 2 Recently reported performance of UWB LNAs

Reference Process CMOS(120583m) 119878

11(dB) Avg gain

(db) Freq (GHz) NF (dB) IIP3(dBm) Die area

(mm2)Power(mW) Topology

[4] 018120583m CMOS ltminus78 119 2minus65 41minus46 4 (4GHz) 088 27 Feedback[7] 06120583m CMOS ltminus7 61 05minus55 54minus82 NA 112 834 Distributed[8] 018 120583m CMOS ltminus20 10 0minus11 31minus61 NA 144 196 Distributed[9] 018120583m CMOS ltminus12 14 3minus6 47minus67 minus5 (45GHz) 11 594 Distributed[10] 018120583m CMOS ltminus9 98 2ndash46 23minus5 minus7 (4GHz) 09 126lowast Feedback[11] 018120583m CMOS ltminus99 93 24minus95 4minus9 minus67 (6GHz) 11 9lowast Chebyshev filter[12] 018120583m CMOS ltminus10 86 24minus94 41minus10 minus35 (6GHz) 176 71lowast Chebyshev filter

[13] 018120583m CMOS ltminus5 191 28minus72 32minus38 minus1 (6GHz) 163 32 Feedback networksynthesis

[14] 018120583m CMOS ltminus10 17 2ndash11 38 NA 0635 1056lowastlowast Feedback[15] 015 120583mHEMT NA 18 085ndash1335 25 NA 1162 70lowastlowast FeedbackThis work 018120583m CMOS ltminus9 9 3ndash56 46ndash53 2 (53 GHz) 08 9 ProposedlowastOnly core LNA lowastlowastSimulation

minus25 minus20 minus15 minus10 minus5 0 5minus80

minus70

minus60

minus50

minus40

minus30

minus20

minus10

0

10

Input power (dBm)

Out

put p

ower

(dBm

)

FundamentalIMD3

IIP3

Figure 24 Measured IIP3at 53 GHz

(CW) generators and an Agilent E4407B spectrum analyzerA noise figure (NF) is measured using an Agilent N8975ANF meter with an Agilent 346C noise source The simulationnoise figure minimum is 41 dB as shown in Figure 25 Dueto the parasitic effect in the layout the minimum of themeasurement is 46 dB Figure 26 shows the micrograph ofthe LNA Table 2 summarizes the measurement results andcompares them with previous literature In the proposedLNA topology the simplified RLC input matching networkand the current-reused configuration benefit from the low-power design The distributed amplifier [7ndash9] requires highpower consumption and a large chip area Compared withthe Chebyshev filter and feedback network synthesis [11ndash13] the proposed input matching network which has onlyone spiral inductor simplifies the circuit complexity and

30 35 40 45 50 55 60 65 70 75 800

1

2

3

4

5

6

7

8

9

10N

oise

figu

re (d

B)

Measured NFSimulated NF

Frequency (GHz)

Figure 25 Simulation and measurement NF

Figure 26Micrograph of the proposed LNA (size 103times 078mm2)

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

International Journal of

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RoboticsJournal of

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Active and Passive Electronic Components

Control Scienceand Engineering

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Submit your manuscripts athttpwwwhindawicom

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Acoustics and VibrationAdvances in

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Electrical and Computer Engineering

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Advances inOptoElectronics

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Modelling amp Simulation in EngineeringHindawi Publishing Corporation httpwwwhindawicom Volume 2014

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Chemical EngineeringInternational Journal of Antennas and

Propagation

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DistributedSensor Networks

International Journal of

Active and Passive Electronic Components 5

1 2 3 4 5 6 7 8 9 100 11Frequency (GHz)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

minus20

minus15

minus10

minus5

0

S 11

(dB)

Figure 8 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation input

reflection coefficient

1 2 3 4 5 6 7 8 9 100 11

3456789

2

10

Frequency (GHz)

NF

(dB)

R1 = 500 ΩR1 = 140 ΩR1 = 50 Ω

Figure 9 Fixed 1198621 1198622 1198623 and 119871

1and simulated 119877

1variation noise

figure

119885out = ((1199031198892

1

1198781198621198892

1198781198713) + 119878119871

4)

1

1198781198626

+1

1198781198627

= 119877out + 119895119883out

(10)

119885out

asymp(119878+(1119903

11988921198621198892)) ((119862

7+1198626) 11986261198627)

1198782+119878 (111990311988921198621198892)+((119862

61198714+11986261198713+11986211988921198713) 1198621198892119862611987131198714)

(11)

where 1198621198892

is the parasitic capacitance of 1198722at the drain

node and 1199031198892

is the resistor of 1198722at the drain node This

equivalent circuit can approximately be an RLC second-order

1 2 3 4 5 6 7 8 90 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

C3 = 76 pFC3 = 095 pFC3 = 015 pF

S 11

(dB)

Figure 10 Fixed119862111986221198771 and 119871

1and simulated119862

3variation input

reflection coefficient

1 2 3 4 5 6 7 8 90

3456789

2

10

10 11

NF

(dB)

Frequency (GHz)C3 = 76 pFC3 = 095 pFC3 = 015 pF

Figure 11 Fixed119862111986221198771 and 119871

1and simulated119862

3variation noise

figure

filter structure In (11) the quality factor of the filter circuitcan be given by

119891119900asymp

1

2120587radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(12)

119876 asymp 11990311988921198621198892radic11986261198714+ 11986261198713+ 11986211988921198713

1198621198892119862611987131198714

(13)

where 119891119900is the resonant frequency In (12) and (13) in order

to obtain broader bandwidth a low-quality factor needs tobe added to passive devices 119891

119900is inversely proportional to

1198621198892

while bandwidth is inversely proportional to 11990311988921198621198892 The

1198621198892capacitance is proportional to the width of the transistor

The 1199031198892

resistance is inversely proportional to the width ofthe transistor UWB system uses a specific frequency band

6 Active and Passive Electronic Components

Rs = 50 Ω

outZ

C7L4

C6L3

rd2 Cd2

Figure 12 The wideband output matching small-signal equivalentcircuit

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus30

minus20

minus25

minus15

minus10

minus5

0

S 22

(dB)

M2 = 160018 120583mM2 = 240018 120583mM2 = 320018 120583m

Figure 13 Fixed 1198626 1198627 1198713and 119871

4 simulated119872

2variation output

reflection coefficient

(31ndash5GHz or 31ndash106GHz) to access data Generally theresults of UWB return loss are less than minus10 dB The powerconsumption is proportional to the width of the transistorAccording to (13) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting 119872

2 This

provides good wideband matching and flat gain Figure 13is fixed as 119862

6(016 pF) 119862

7(123 pF) 119871

3(119 nH) and 119871

4

(059 nH) showing that the simulated 1198722variation output

reflection coefficient As can be seen the second-order filterof 1198722(240018 120583m) is employed as the output matching

network According to (12) the capacitor 1198626and inductor

1198713-1198714will be optimized at the resonant frequency Figure 14

is fixed as 1198626(016 pF) 119862

7(123 pF) 119872

2(240018120583m)

and 1198714(059 nH) showing that the resonant frequency is

inversely proportional to the 1198713 The output network has less

complexity and a better reflected coefficient from 31 GHz to106GHz

33 Gain At a high frequency the current gain of commonsource configured transistor 119872

1is 1198921198981119904119862gs1 [11] Figure 15

shows the approximate first stage 1198721load of the simplified

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

S 22

(dB)

L3 = 119nHL3 = 129nH

L3 = 109nH

Figure 14 Fixed 1198626 1198627 1198722 and 119871

4and simulated 119871

3variation

output reflection coefficient

Id1

Vout1

C5

L2

Cd11gm2

Figure 15 The first stage simplified small-signal equivalent circuit

small-signal equivalent circuit [12] The first stage load canbe approximated as

119885load1 asymp1

119878119871211986251198621198891

sdot119878211987121198625+ 11987811989211989821198712+ 1

1198782 + 119878 (11989211989821198625) + ((119862

5+ 1198621198891) (1198712(11986251198621198891)))

(14)

The transfer function of the input matching network is119885in(119904) The first stage voltage gain can be approximated as

1198601198811

asymp minus1198921198981119885in

119878119862gs1119877119904sdot 119885Load1 (15)

where 119877119904is the source resistance in which voltage gain is

determined by the load inductor 1198712 the total capacitance at

the drain of 1198721 and bypass capacitor 119862

5 The second stage

voltage gain can be estimated as

1198601198812

asymp minus1198921198982

119878119862gs2119877in2sdot 119885out (16)

Active and Passive Electronic Components 7

1 2 3 4 5 6 7 8 9 10

10

12

110

2

4

6

8

0

Frequency (GHz)

Δ

K

Δ K

Figure 16 119870 (stability factor) and Δ simulation result

where 119877in2 is the input resistance at the gate of1198722 and 119885outis the output load impedance The LNA voltage gain can beapproximated as

119860119881asymp11989211989811198921198982119885in119885Load1119885out

1198782119862gs1119862gs2119877119904119877in2 (17)

According to (17) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting the devicesize This provides good wideband matching noise figureoptimization design and flat gain

34 Stability Amplifier stability is important to preventnatural oscillation The stability can be determined by the 119878-parameters input and output matching network and circuitterminations The simpler tests show whether or not a devicecan be unconditionally stable [20] One of these is the 119870-Δtest which shows if a device can be unconditionally stable ifRolletrsquos condition is defined as

119870 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

minus100381610038161003816100381611987822

10038161003816100381610038162

+ |Δ|2

210038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1

|Δ| =10038161003816100381610038161198781111987822 minus 1198781211987821

1003816100381610038161003816 lt 1

(18)

The 119870-Δ test cannot be used to compare the relative sta-bility of twoormore devices because it involves constraints ontwo separate parameters A new criterion has been proposedwhich combines the 119878-parameters in a test involving only asingle parameter 120583 defined as

120583 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

100381610038161003816100381611987822 minus Δ119878lowast

11

1003816100381610038161003816 +10038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1 (19)

The performance of LNA is simulated using the advancedesign system (ADS) To consider the stability of the LNAthe119870 factor andΔ should be considered Figure 16 shows thatthe 119870 factor is always larger than 1 and the Δ is smaller than1 all the time Hence this circuit is stable for all frequency

1 2 3 4 5 6 7 8 9 10 11

11

0Frequency (GHz)

3

5

7

9

1

120583load120583source

120583lo

ad120583

sour

ce

Figure 17 120583 factor simulation result

RFin RFout

Vbias

DC power supply(HP 4142B)

Network analyzer(HP 8510C)

S-parameters test set(HP 8517B)

VDD

Figure 18 Test setup used for the LNA 119878-parameter measurements

bands There is another way to diagnose whether the LNAis unconditionally stable The 120583 factor at input and outputshould be larger than 1 in all frequency bands as shown inFigure 17

4 Measurement Results

On-wafer measurement is performed by an HP 8510C net-work analyzer and an HP 8517B is to test the 119878-parameter asshown in Figure 18 A network analyzer is used to measurethe frequency response and input matching of the LNAThe input and output impedance matchings are both 50ΩTo ensure that the LNA still provides a conversion gainwhen process deviation occurs the other process cornersnamely typical-NMOS typical-PMOS (TT) fast-NMOS fast-PMOS (FF) and slow-NMOS slow-PMOS (SS) are also usedto simulate this LNA The results are shown in Figure 19Figure 19 shows the measurement forward gain (119878

21) from

3 to 56GHz the measured gain is about 7ndash10 dB Themeasurement data is 6 dB lower than the pre-simulation data

8 Active and Passive Electronic Components

201816141210

86420

28 33 38 43 48 53 58Frequency (GHz)

S 21

(dB)

Pre-simulation TTPost-simulation FFPost-simulation TT

Post-simulation SSMeasurement

Figure 19 Simulation and measurement results of conversion gainversus frequency of the proposed LNA RF stands for simulationresults of modifiedMOSFET RFmodel FF TT and SS represent thesimulation results of fast-NMOS fast-PMOS typical-NMOS typical-PMOS and slow-NMOS slow-PMOS process corners

minus10

0

minus20

minus30

minus40

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 11

(dB)

Figure 20 Measured and simulated input reflection coefficient

because of the process drift and parasitic effect in the layoutThe result of conversion gain measurement is situated in thepost-simulation (SS) process corner

Figure 20 shows the input return loss (S11) from 3 to

11 GHz and the measured S11

is less than minus9 dB Figure 21shows output return loss (S

22) from 3 to 75GHz and the

measured S22is less thanminus8 dB Figure 22 shows reverse isola-

tion (S12) from 3 to 11 GHz and the measured S

12is less than

minus40 dB The simulation measured input of 1 dB compressionpoint at 53 GHz shown in Figure 23 is about minus8 dB Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz and the IIP

3is 2 dBm at 53 GHz The

IM3 is measured using two Agilnet E8247C continuous wave

minus10

minus15

minus20

minus25

minus30

minus5

0

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 22

(dB)

Figure 21 Measured and simulated output reflection coefficient

minus10

minus20

minus30

minus40

minus50

minus60

minus70

minus80

MeasurementSimulation

3 4 5 6 7 8 9 102 11

0

Frequency (GHz)

S 12

(dB)

Figure 22 Measured and simulated reverse isolation

minus40 minus35 minus30 minus25 minus20 minus15 minus10 minus5 0 5 100

1

2

3

4

5

6

7

8

9

Gai

n (d

B)

Input power (dBm)

P1dB

Figure 23 Measured input 1 db compression point at 53 GHz

Active and Passive Electronic Components 9

Table 2 Recently reported performance of UWB LNAs

Reference Process CMOS(120583m) 119878

11(dB) Avg gain

(db) Freq (GHz) NF (dB) IIP3(dBm) Die area

(mm2)Power(mW) Topology

[4] 018120583m CMOS ltminus78 119 2minus65 41minus46 4 (4GHz) 088 27 Feedback[7] 06120583m CMOS ltminus7 61 05minus55 54minus82 NA 112 834 Distributed[8] 018 120583m CMOS ltminus20 10 0minus11 31minus61 NA 144 196 Distributed[9] 018120583m CMOS ltminus12 14 3minus6 47minus67 minus5 (45GHz) 11 594 Distributed[10] 018120583m CMOS ltminus9 98 2ndash46 23minus5 minus7 (4GHz) 09 126lowast Feedback[11] 018120583m CMOS ltminus99 93 24minus95 4minus9 minus67 (6GHz) 11 9lowast Chebyshev filter[12] 018120583m CMOS ltminus10 86 24minus94 41minus10 minus35 (6GHz) 176 71lowast Chebyshev filter

[13] 018120583m CMOS ltminus5 191 28minus72 32minus38 minus1 (6GHz) 163 32 Feedback networksynthesis

[14] 018120583m CMOS ltminus10 17 2ndash11 38 NA 0635 1056lowastlowast Feedback[15] 015 120583mHEMT NA 18 085ndash1335 25 NA 1162 70lowastlowast FeedbackThis work 018120583m CMOS ltminus9 9 3ndash56 46ndash53 2 (53 GHz) 08 9 ProposedlowastOnly core LNA lowastlowastSimulation

minus25 minus20 minus15 minus10 minus5 0 5minus80

minus70

minus60

minus50

minus40

minus30

minus20

minus10

0

10

Input power (dBm)

Out

put p

ower

(dBm

)

FundamentalIMD3

IIP3

Figure 24 Measured IIP3at 53 GHz

(CW) generators and an Agilent E4407B spectrum analyzerA noise figure (NF) is measured using an Agilent N8975ANF meter with an Agilent 346C noise source The simulationnoise figure minimum is 41 dB as shown in Figure 25 Dueto the parasitic effect in the layout the minimum of themeasurement is 46 dB Figure 26 shows the micrograph ofthe LNA Table 2 summarizes the measurement results andcompares them with previous literature In the proposedLNA topology the simplified RLC input matching networkand the current-reused configuration benefit from the low-power design The distributed amplifier [7ndash9] requires highpower consumption and a large chip area Compared withthe Chebyshev filter and feedback network synthesis [11ndash13] the proposed input matching network which has onlyone spiral inductor simplifies the circuit complexity and

30 35 40 45 50 55 60 65 70 75 800

1

2

3

4

5

6

7

8

9

10N

oise

figu

re (d

B)

Measured NFSimulated NF

Frequency (GHz)

Figure 25 Simulation and measurement NF

Figure 26Micrograph of the proposed LNA (size 103times 078mm2)

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

International Journal of

AerospaceEngineeringHindawi Publishing Corporationhttpwwwhindawicom Volume 2014

RoboticsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Active and Passive Electronic Components

Control Scienceand Engineering

Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

International Journal of

RotatingMachinery

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporation httpwwwhindawicom

Journal ofEngineeringVolume 2014

Submit your manuscripts athttpwwwhindawicom

VLSI Design

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Shock and Vibration

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Civil EngineeringAdvances in

Acoustics and VibrationAdvances in

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Electrical and Computer Engineering

Journal of

Advances inOptoElectronics

Hindawi Publishing Corporation httpwwwhindawicom

Volume 2014

The Scientific World JournalHindawi Publishing Corporation httpwwwhindawicom Volume 2014

SensorsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Modelling amp Simulation in EngineeringHindawi Publishing Corporation httpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Chemical EngineeringInternational Journal of Antennas and

Propagation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Navigation and Observation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

DistributedSensor Networks

International Journal of

6 Active and Passive Electronic Components

Rs = 50 Ω

outZ

C7L4

C6L3

rd2 Cd2

Figure 12 The wideband output matching small-signal equivalentcircuit

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus30

minus20

minus25

minus15

minus10

minus5

0

S 22

(dB)

M2 = 160018 120583mM2 = 240018 120583mM2 = 320018 120583m

Figure 13 Fixed 1198626 1198627 1198713and 119871

4 simulated119872

2variation output

reflection coefficient

(31ndash5GHz or 31ndash106GHz) to access data Generally theresults of UWB return loss are less than minus10 dB The powerconsumption is proportional to the width of the transistorAccording to (13) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting 119872

2 This

provides good wideband matching and flat gain Figure 13is fixed as 119862

6(016 pF) 119862

7(123 pF) 119871

3(119 nH) and 119871

4

(059 nH) showing that the simulated 1198722variation output

reflection coefficient As can be seen the second-order filterof 1198722(240018 120583m) is employed as the output matching

network According to (12) the capacitor 1198626and inductor

1198713-1198714will be optimized at the resonant frequency Figure 14

is fixed as 1198626(016 pF) 119862

7(123 pF) 119872

2(240018120583m)

and 1198714(059 nH) showing that the resonant frequency is

inversely proportional to the 1198713 The output network has less

complexity and a better reflected coefficient from 31 GHz to106GHz

33 Gain At a high frequency the current gain of commonsource configured transistor 119872

1is 1198921198981119904119862gs1 [11] Figure 15

shows the approximate first stage 1198721load of the simplified

2 3 4 5 6 7 8 9 10 11Frequency (GHz)

minus20

minus15

minus10

minus5

0

S 22

(dB)

L3 = 119nHL3 = 129nH

L3 = 109nH

Figure 14 Fixed 1198626 1198627 1198722 and 119871

4and simulated 119871

3variation

output reflection coefficient

Id1

Vout1

C5

L2

Cd11gm2

Figure 15 The first stage simplified small-signal equivalent circuit

small-signal equivalent circuit [12] The first stage load canbe approximated as

119885load1 asymp1

119878119871211986251198621198891

sdot119878211987121198625+ 11987811989211989821198712+ 1

1198782 + 119878 (11989211989821198625) + ((119862

5+ 1198621198891) (1198712(11986251198621198891)))

(14)

The transfer function of the input matching network is119885in(119904) The first stage voltage gain can be approximated as

1198601198811

asymp minus1198921198981119885in

119878119862gs1119877119904sdot 119885Load1 (15)

where 119877119904is the source resistance in which voltage gain is

determined by the load inductor 1198712 the total capacitance at

the drain of 1198721 and bypass capacitor 119862

5 The second stage

voltage gain can be estimated as

1198601198812

asymp minus1198921198982

119878119862gs2119877in2sdot 119885out (16)

Active and Passive Electronic Components 7

1 2 3 4 5 6 7 8 9 10

10

12

110

2

4

6

8

0

Frequency (GHz)

Δ

K

Δ K

Figure 16 119870 (stability factor) and Δ simulation result

where 119877in2 is the input resistance at the gate of1198722 and 119885outis the output load impedance The LNA voltage gain can beapproximated as

119860119881asymp11989211989811198921198982119885in119885Load1119885out

1198782119862gs1119862gs2119877119904119877in2 (17)

According to (17) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting the devicesize This provides good wideband matching noise figureoptimization design and flat gain

34 Stability Amplifier stability is important to preventnatural oscillation The stability can be determined by the 119878-parameters input and output matching network and circuitterminations The simpler tests show whether or not a devicecan be unconditionally stable [20] One of these is the 119870-Δtest which shows if a device can be unconditionally stable ifRolletrsquos condition is defined as

119870 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

minus100381610038161003816100381611987822

10038161003816100381610038162

+ |Δ|2

210038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1

|Δ| =10038161003816100381610038161198781111987822 minus 1198781211987821

1003816100381610038161003816 lt 1

(18)

The 119870-Δ test cannot be used to compare the relative sta-bility of twoormore devices because it involves constraints ontwo separate parameters A new criterion has been proposedwhich combines the 119878-parameters in a test involving only asingle parameter 120583 defined as

120583 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

100381610038161003816100381611987822 minus Δ119878lowast

11

1003816100381610038161003816 +10038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1 (19)

The performance of LNA is simulated using the advancedesign system (ADS) To consider the stability of the LNAthe119870 factor andΔ should be considered Figure 16 shows thatthe 119870 factor is always larger than 1 and the Δ is smaller than1 all the time Hence this circuit is stable for all frequency

1 2 3 4 5 6 7 8 9 10 11

11

0Frequency (GHz)

3

5

7

9

1

120583load120583source

120583lo

ad120583

sour

ce

Figure 17 120583 factor simulation result

RFin RFout

Vbias

DC power supply(HP 4142B)

Network analyzer(HP 8510C)

S-parameters test set(HP 8517B)

VDD

Figure 18 Test setup used for the LNA 119878-parameter measurements

bands There is another way to diagnose whether the LNAis unconditionally stable The 120583 factor at input and outputshould be larger than 1 in all frequency bands as shown inFigure 17

4 Measurement Results

On-wafer measurement is performed by an HP 8510C net-work analyzer and an HP 8517B is to test the 119878-parameter asshown in Figure 18 A network analyzer is used to measurethe frequency response and input matching of the LNAThe input and output impedance matchings are both 50ΩTo ensure that the LNA still provides a conversion gainwhen process deviation occurs the other process cornersnamely typical-NMOS typical-PMOS (TT) fast-NMOS fast-PMOS (FF) and slow-NMOS slow-PMOS (SS) are also usedto simulate this LNA The results are shown in Figure 19Figure 19 shows the measurement forward gain (119878

21) from

3 to 56GHz the measured gain is about 7ndash10 dB Themeasurement data is 6 dB lower than the pre-simulation data

8 Active and Passive Electronic Components

201816141210

86420

28 33 38 43 48 53 58Frequency (GHz)

S 21

(dB)

Pre-simulation TTPost-simulation FFPost-simulation TT

Post-simulation SSMeasurement

Figure 19 Simulation and measurement results of conversion gainversus frequency of the proposed LNA RF stands for simulationresults of modifiedMOSFET RFmodel FF TT and SS represent thesimulation results of fast-NMOS fast-PMOS typical-NMOS typical-PMOS and slow-NMOS slow-PMOS process corners

minus10

0

minus20

minus30

minus40

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 11

(dB)

Figure 20 Measured and simulated input reflection coefficient

because of the process drift and parasitic effect in the layoutThe result of conversion gain measurement is situated in thepost-simulation (SS) process corner

Figure 20 shows the input return loss (S11) from 3 to

11 GHz and the measured S11

is less than minus9 dB Figure 21shows output return loss (S

22) from 3 to 75GHz and the

measured S22is less thanminus8 dB Figure 22 shows reverse isola-

tion (S12) from 3 to 11 GHz and the measured S

12is less than

minus40 dB The simulation measured input of 1 dB compressionpoint at 53 GHz shown in Figure 23 is about minus8 dB Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz and the IIP

3is 2 dBm at 53 GHz The

IM3 is measured using two Agilnet E8247C continuous wave

minus10

minus15

minus20

minus25

minus30

minus5

0

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 22

(dB)

Figure 21 Measured and simulated output reflection coefficient

minus10

minus20

minus30

minus40

minus50

minus60

minus70

minus80

MeasurementSimulation

3 4 5 6 7 8 9 102 11

0

Frequency (GHz)

S 12

(dB)

Figure 22 Measured and simulated reverse isolation

minus40 minus35 minus30 minus25 minus20 minus15 minus10 minus5 0 5 100

1

2

3

4

5

6

7

8

9

Gai

n (d

B)

Input power (dBm)

P1dB

Figure 23 Measured input 1 db compression point at 53 GHz

Active and Passive Electronic Components 9

Table 2 Recently reported performance of UWB LNAs

Reference Process CMOS(120583m) 119878

11(dB) Avg gain

(db) Freq (GHz) NF (dB) IIP3(dBm) Die area

(mm2)Power(mW) Topology

[4] 018120583m CMOS ltminus78 119 2minus65 41minus46 4 (4GHz) 088 27 Feedback[7] 06120583m CMOS ltminus7 61 05minus55 54minus82 NA 112 834 Distributed[8] 018 120583m CMOS ltminus20 10 0minus11 31minus61 NA 144 196 Distributed[9] 018120583m CMOS ltminus12 14 3minus6 47minus67 minus5 (45GHz) 11 594 Distributed[10] 018120583m CMOS ltminus9 98 2ndash46 23minus5 minus7 (4GHz) 09 126lowast Feedback[11] 018120583m CMOS ltminus99 93 24minus95 4minus9 minus67 (6GHz) 11 9lowast Chebyshev filter[12] 018120583m CMOS ltminus10 86 24minus94 41minus10 minus35 (6GHz) 176 71lowast Chebyshev filter

[13] 018120583m CMOS ltminus5 191 28minus72 32minus38 minus1 (6GHz) 163 32 Feedback networksynthesis

[14] 018120583m CMOS ltminus10 17 2ndash11 38 NA 0635 1056lowastlowast Feedback[15] 015 120583mHEMT NA 18 085ndash1335 25 NA 1162 70lowastlowast FeedbackThis work 018120583m CMOS ltminus9 9 3ndash56 46ndash53 2 (53 GHz) 08 9 ProposedlowastOnly core LNA lowastlowastSimulation

minus25 minus20 minus15 minus10 minus5 0 5minus80

minus70

minus60

minus50

minus40

minus30

minus20

minus10

0

10

Input power (dBm)

Out

put p

ower

(dBm

)

FundamentalIMD3

IIP3

Figure 24 Measured IIP3at 53 GHz

(CW) generators and an Agilent E4407B spectrum analyzerA noise figure (NF) is measured using an Agilent N8975ANF meter with an Agilent 346C noise source The simulationnoise figure minimum is 41 dB as shown in Figure 25 Dueto the parasitic effect in the layout the minimum of themeasurement is 46 dB Figure 26 shows the micrograph ofthe LNA Table 2 summarizes the measurement results andcompares them with previous literature In the proposedLNA topology the simplified RLC input matching networkand the current-reused configuration benefit from the low-power design The distributed amplifier [7ndash9] requires highpower consumption and a large chip area Compared withthe Chebyshev filter and feedback network synthesis [11ndash13] the proposed input matching network which has onlyone spiral inductor simplifies the circuit complexity and

30 35 40 45 50 55 60 65 70 75 800

1

2

3

4

5

6

7

8

9

10N

oise

figu

re (d

B)

Measured NFSimulated NF

Frequency (GHz)

Figure 25 Simulation and measurement NF

Figure 26Micrograph of the proposed LNA (size 103times 078mm2)

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

International Journal of

AerospaceEngineeringHindawi Publishing Corporationhttpwwwhindawicom Volume 2014

RoboticsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Active and Passive Electronic Components

Control Scienceand Engineering

Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

International Journal of

RotatingMachinery

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporation httpwwwhindawicom

Journal ofEngineeringVolume 2014

Submit your manuscripts athttpwwwhindawicom

VLSI Design

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Shock and Vibration

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Civil EngineeringAdvances in

Acoustics and VibrationAdvances in

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Electrical and Computer Engineering

Journal of

Advances inOptoElectronics

Hindawi Publishing Corporation httpwwwhindawicom

Volume 2014

The Scientific World JournalHindawi Publishing Corporation httpwwwhindawicom Volume 2014

SensorsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Modelling amp Simulation in EngineeringHindawi Publishing Corporation httpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Chemical EngineeringInternational Journal of Antennas and

Propagation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Navigation and Observation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

DistributedSensor Networks

International Journal of

Active and Passive Electronic Components 7

1 2 3 4 5 6 7 8 9 10

10

12

110

2

4

6

8

0

Frequency (GHz)

Δ

K

Δ K

Figure 16 119870 (stability factor) and Δ simulation result

where 119877in2 is the input resistance at the gate of1198722 and 119885outis the output load impedance The LNA voltage gain can beapproximated as

119860119881asymp11989211989811198921198982119885in119885Load1119885out

1198782119862gs1119862gs2119877119904119877in2 (17)

According to (17) the narrowband LNA can be convertedinto a wideband amplifier by properly selecting the devicesize This provides good wideband matching noise figureoptimization design and flat gain

34 Stability Amplifier stability is important to preventnatural oscillation The stability can be determined by the 119878-parameters input and output matching network and circuitterminations The simpler tests show whether or not a devicecan be unconditionally stable [20] One of these is the 119870-Δtest which shows if a device can be unconditionally stable ifRolletrsquos condition is defined as

119870 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

minus100381610038161003816100381611987822

10038161003816100381610038162

+ |Δ|2

210038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1

|Δ| =10038161003816100381610038161198781111987822 minus 1198781211987821

1003816100381610038161003816 lt 1

(18)

The 119870-Δ test cannot be used to compare the relative sta-bility of twoormore devices because it involves constraints ontwo separate parameters A new criterion has been proposedwhich combines the 119878-parameters in a test involving only asingle parameter 120583 defined as

120583 =1 minus

10038161003816100381610038161198781110038161003816100381610038162

100381610038161003816100381611987822 minus Δ119878lowast

11

1003816100381610038161003816 +10038161003816100381610038161198781211987821

1003816100381610038161003816

gt 1 (19)

The performance of LNA is simulated using the advancedesign system (ADS) To consider the stability of the LNAthe119870 factor andΔ should be considered Figure 16 shows thatthe 119870 factor is always larger than 1 and the Δ is smaller than1 all the time Hence this circuit is stable for all frequency

1 2 3 4 5 6 7 8 9 10 11

11

0Frequency (GHz)

3

5

7

9

1

120583load120583source

120583lo

ad120583

sour

ce

Figure 17 120583 factor simulation result

RFin RFout

Vbias

DC power supply(HP 4142B)

Network analyzer(HP 8510C)

S-parameters test set(HP 8517B)

VDD

Figure 18 Test setup used for the LNA 119878-parameter measurements

bands There is another way to diagnose whether the LNAis unconditionally stable The 120583 factor at input and outputshould be larger than 1 in all frequency bands as shown inFigure 17

4 Measurement Results

On-wafer measurement is performed by an HP 8510C net-work analyzer and an HP 8517B is to test the 119878-parameter asshown in Figure 18 A network analyzer is used to measurethe frequency response and input matching of the LNAThe input and output impedance matchings are both 50ΩTo ensure that the LNA still provides a conversion gainwhen process deviation occurs the other process cornersnamely typical-NMOS typical-PMOS (TT) fast-NMOS fast-PMOS (FF) and slow-NMOS slow-PMOS (SS) are also usedto simulate this LNA The results are shown in Figure 19Figure 19 shows the measurement forward gain (119878

21) from

3 to 56GHz the measured gain is about 7ndash10 dB Themeasurement data is 6 dB lower than the pre-simulation data

8 Active and Passive Electronic Components

201816141210

86420

28 33 38 43 48 53 58Frequency (GHz)

S 21

(dB)

Pre-simulation TTPost-simulation FFPost-simulation TT

Post-simulation SSMeasurement

Figure 19 Simulation and measurement results of conversion gainversus frequency of the proposed LNA RF stands for simulationresults of modifiedMOSFET RFmodel FF TT and SS represent thesimulation results of fast-NMOS fast-PMOS typical-NMOS typical-PMOS and slow-NMOS slow-PMOS process corners

minus10

0

minus20

minus30

minus40

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 11

(dB)

Figure 20 Measured and simulated input reflection coefficient

because of the process drift and parasitic effect in the layoutThe result of conversion gain measurement is situated in thepost-simulation (SS) process corner

Figure 20 shows the input return loss (S11) from 3 to

11 GHz and the measured S11

is less than minus9 dB Figure 21shows output return loss (S

22) from 3 to 75GHz and the

measured S22is less thanminus8 dB Figure 22 shows reverse isola-

tion (S12) from 3 to 11 GHz and the measured S

12is less than

minus40 dB The simulation measured input of 1 dB compressionpoint at 53 GHz shown in Figure 23 is about minus8 dB Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz and the IIP

3is 2 dBm at 53 GHz The

IM3 is measured using two Agilnet E8247C continuous wave

minus10

minus15

minus20

minus25

minus30

minus5

0

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 22

(dB)

Figure 21 Measured and simulated output reflection coefficient

minus10

minus20

minus30

minus40

minus50

minus60

minus70

minus80

MeasurementSimulation

3 4 5 6 7 8 9 102 11

0

Frequency (GHz)

S 12

(dB)

Figure 22 Measured and simulated reverse isolation

minus40 minus35 minus30 minus25 minus20 minus15 minus10 minus5 0 5 100

1

2

3

4

5

6

7

8

9

Gai

n (d

B)

Input power (dBm)

P1dB

Figure 23 Measured input 1 db compression point at 53 GHz

Active and Passive Electronic Components 9

Table 2 Recently reported performance of UWB LNAs

Reference Process CMOS(120583m) 119878

11(dB) Avg gain

(db) Freq (GHz) NF (dB) IIP3(dBm) Die area

(mm2)Power(mW) Topology

[4] 018120583m CMOS ltminus78 119 2minus65 41minus46 4 (4GHz) 088 27 Feedback[7] 06120583m CMOS ltminus7 61 05minus55 54minus82 NA 112 834 Distributed[8] 018 120583m CMOS ltminus20 10 0minus11 31minus61 NA 144 196 Distributed[9] 018120583m CMOS ltminus12 14 3minus6 47minus67 minus5 (45GHz) 11 594 Distributed[10] 018120583m CMOS ltminus9 98 2ndash46 23minus5 minus7 (4GHz) 09 126lowast Feedback[11] 018120583m CMOS ltminus99 93 24minus95 4minus9 minus67 (6GHz) 11 9lowast Chebyshev filter[12] 018120583m CMOS ltminus10 86 24minus94 41minus10 minus35 (6GHz) 176 71lowast Chebyshev filter

[13] 018120583m CMOS ltminus5 191 28minus72 32minus38 minus1 (6GHz) 163 32 Feedback networksynthesis

[14] 018120583m CMOS ltminus10 17 2ndash11 38 NA 0635 1056lowastlowast Feedback[15] 015 120583mHEMT NA 18 085ndash1335 25 NA 1162 70lowastlowast FeedbackThis work 018120583m CMOS ltminus9 9 3ndash56 46ndash53 2 (53 GHz) 08 9 ProposedlowastOnly core LNA lowastlowastSimulation

minus25 minus20 minus15 minus10 minus5 0 5minus80

minus70

minus60

minus50

minus40

minus30

minus20

minus10

0

10

Input power (dBm)

Out

put p

ower

(dBm

)

FundamentalIMD3

IIP3

Figure 24 Measured IIP3at 53 GHz

(CW) generators and an Agilent E4407B spectrum analyzerA noise figure (NF) is measured using an Agilent N8975ANF meter with an Agilent 346C noise source The simulationnoise figure minimum is 41 dB as shown in Figure 25 Dueto the parasitic effect in the layout the minimum of themeasurement is 46 dB Figure 26 shows the micrograph ofthe LNA Table 2 summarizes the measurement results andcompares them with previous literature In the proposedLNA topology the simplified RLC input matching networkand the current-reused configuration benefit from the low-power design The distributed amplifier [7ndash9] requires highpower consumption and a large chip area Compared withthe Chebyshev filter and feedback network synthesis [11ndash13] the proposed input matching network which has onlyone spiral inductor simplifies the circuit complexity and

30 35 40 45 50 55 60 65 70 75 800

1

2

3

4

5

6

7

8

9

10N

oise

figu

re (d

B)

Measured NFSimulated NF

Frequency (GHz)

Figure 25 Simulation and measurement NF

Figure 26Micrograph of the proposed LNA (size 103times 078mm2)

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

International Journal of

AerospaceEngineeringHindawi Publishing Corporationhttpwwwhindawicom Volume 2014

RoboticsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Active and Passive Electronic Components

Control Scienceand Engineering

Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

International Journal of

RotatingMachinery

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporation httpwwwhindawicom

Journal ofEngineeringVolume 2014

Submit your manuscripts athttpwwwhindawicom

VLSI Design

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Shock and Vibration

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Civil EngineeringAdvances in

Acoustics and VibrationAdvances in

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Electrical and Computer Engineering

Journal of

Advances inOptoElectronics

Hindawi Publishing Corporation httpwwwhindawicom

Volume 2014

The Scientific World JournalHindawi Publishing Corporation httpwwwhindawicom Volume 2014

SensorsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Modelling amp Simulation in EngineeringHindawi Publishing Corporation httpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Chemical EngineeringInternational Journal of Antennas and

Propagation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Navigation and Observation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

DistributedSensor Networks

International Journal of

8 Active and Passive Electronic Components

201816141210

86420

28 33 38 43 48 53 58Frequency (GHz)

S 21

(dB)

Pre-simulation TTPost-simulation FFPost-simulation TT

Post-simulation SSMeasurement

Figure 19 Simulation and measurement results of conversion gainversus frequency of the proposed LNA RF stands for simulationresults of modifiedMOSFET RFmodel FF TT and SS represent thesimulation results of fast-NMOS fast-PMOS typical-NMOS typical-PMOS and slow-NMOS slow-PMOS process corners

minus10

0

minus20

minus30

minus40

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 11

(dB)

Figure 20 Measured and simulated input reflection coefficient

because of the process drift and parasitic effect in the layoutThe result of conversion gain measurement is situated in thepost-simulation (SS) process corner

Figure 20 shows the input return loss (S11) from 3 to

11 GHz and the measured S11

is less than minus9 dB Figure 21shows output return loss (S

22) from 3 to 75GHz and the

measured S22is less thanminus8 dB Figure 22 shows reverse isola-

tion (S12) from 3 to 11 GHz and the measured S

12is less than

minus40 dB The simulation measured input of 1 dB compressionpoint at 53 GHz shown in Figure 23 is about minus8 dB Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz and the IIP

3is 2 dBm at 53 GHz The

IM3 is measured using two Agilnet E8247C continuous wave

minus10

minus15

minus20

minus25

minus30

minus5

0

MeasurementSimulation

3 4 5 6 7 8 9 102 11Frequency (GHz)

S 22

(dB)

Figure 21 Measured and simulated output reflection coefficient

minus10

minus20

minus30

minus40

minus50

minus60

minus70

minus80

MeasurementSimulation

3 4 5 6 7 8 9 102 11

0

Frequency (GHz)

S 12

(dB)

Figure 22 Measured and simulated reverse isolation

minus40 minus35 minus30 minus25 minus20 minus15 minus10 minus5 0 5 100

1

2

3

4

5

6

7

8

9

Gai

n (d

B)

Input power (dBm)

P1dB

Figure 23 Measured input 1 db compression point at 53 GHz

Active and Passive Electronic Components 9

Table 2 Recently reported performance of UWB LNAs

Reference Process CMOS(120583m) 119878

11(dB) Avg gain

(db) Freq (GHz) NF (dB) IIP3(dBm) Die area

(mm2)Power(mW) Topology

[4] 018120583m CMOS ltminus78 119 2minus65 41minus46 4 (4GHz) 088 27 Feedback[7] 06120583m CMOS ltminus7 61 05minus55 54minus82 NA 112 834 Distributed[8] 018 120583m CMOS ltminus20 10 0minus11 31minus61 NA 144 196 Distributed[9] 018120583m CMOS ltminus12 14 3minus6 47minus67 minus5 (45GHz) 11 594 Distributed[10] 018120583m CMOS ltminus9 98 2ndash46 23minus5 minus7 (4GHz) 09 126lowast Feedback[11] 018120583m CMOS ltminus99 93 24minus95 4minus9 minus67 (6GHz) 11 9lowast Chebyshev filter[12] 018120583m CMOS ltminus10 86 24minus94 41minus10 minus35 (6GHz) 176 71lowast Chebyshev filter

[13] 018120583m CMOS ltminus5 191 28minus72 32minus38 minus1 (6GHz) 163 32 Feedback networksynthesis

[14] 018120583m CMOS ltminus10 17 2ndash11 38 NA 0635 1056lowastlowast Feedback[15] 015 120583mHEMT NA 18 085ndash1335 25 NA 1162 70lowastlowast FeedbackThis work 018120583m CMOS ltminus9 9 3ndash56 46ndash53 2 (53 GHz) 08 9 ProposedlowastOnly core LNA lowastlowastSimulation

minus25 minus20 minus15 minus10 minus5 0 5minus80

minus70

minus60

minus50

minus40

minus30

minus20

minus10

0

10

Input power (dBm)

Out

put p

ower

(dBm

)

FundamentalIMD3

IIP3

Figure 24 Measured IIP3at 53 GHz

(CW) generators and an Agilent E4407B spectrum analyzerA noise figure (NF) is measured using an Agilent N8975ANF meter with an Agilent 346C noise source The simulationnoise figure minimum is 41 dB as shown in Figure 25 Dueto the parasitic effect in the layout the minimum of themeasurement is 46 dB Figure 26 shows the micrograph ofthe LNA Table 2 summarizes the measurement results andcompares them with previous literature In the proposedLNA topology the simplified RLC input matching networkand the current-reused configuration benefit from the low-power design The distributed amplifier [7ndash9] requires highpower consumption and a large chip area Compared withthe Chebyshev filter and feedback network synthesis [11ndash13] the proposed input matching network which has onlyone spiral inductor simplifies the circuit complexity and

30 35 40 45 50 55 60 65 70 75 800

1

2

3

4

5

6

7

8

9

10N

oise

figu

re (d

B)

Measured NFSimulated NF

Frequency (GHz)

Figure 25 Simulation and measurement NF

Figure 26Micrograph of the proposed LNA (size 103times 078mm2)

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

International Journal of

AerospaceEngineeringHindawi Publishing Corporationhttpwwwhindawicom Volume 2014

RoboticsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Active and Passive Electronic Components

Control Scienceand Engineering

Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

International Journal of

RotatingMachinery

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporation httpwwwhindawicom

Journal ofEngineeringVolume 2014

Submit your manuscripts athttpwwwhindawicom

VLSI Design

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Shock and Vibration

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Civil EngineeringAdvances in

Acoustics and VibrationAdvances in

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Electrical and Computer Engineering

Journal of

Advances inOptoElectronics

Hindawi Publishing Corporation httpwwwhindawicom

Volume 2014

The Scientific World JournalHindawi Publishing Corporation httpwwwhindawicom Volume 2014

SensorsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Modelling amp Simulation in EngineeringHindawi Publishing Corporation httpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Chemical EngineeringInternational Journal of Antennas and

Propagation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Navigation and Observation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

DistributedSensor Networks

International Journal of

Active and Passive Electronic Components 9

Table 2 Recently reported performance of UWB LNAs

Reference Process CMOS(120583m) 119878

11(dB) Avg gain

(db) Freq (GHz) NF (dB) IIP3(dBm) Die area

(mm2)Power(mW) Topology

[4] 018120583m CMOS ltminus78 119 2minus65 41minus46 4 (4GHz) 088 27 Feedback[7] 06120583m CMOS ltminus7 61 05minus55 54minus82 NA 112 834 Distributed[8] 018 120583m CMOS ltminus20 10 0minus11 31minus61 NA 144 196 Distributed[9] 018120583m CMOS ltminus12 14 3minus6 47minus67 minus5 (45GHz) 11 594 Distributed[10] 018120583m CMOS ltminus9 98 2ndash46 23minus5 minus7 (4GHz) 09 126lowast Feedback[11] 018120583m CMOS ltminus99 93 24minus95 4minus9 minus67 (6GHz) 11 9lowast Chebyshev filter[12] 018120583m CMOS ltminus10 86 24minus94 41minus10 minus35 (6GHz) 176 71lowast Chebyshev filter

[13] 018120583m CMOS ltminus5 191 28minus72 32minus38 minus1 (6GHz) 163 32 Feedback networksynthesis

[14] 018120583m CMOS ltminus10 17 2ndash11 38 NA 0635 1056lowastlowast Feedback[15] 015 120583mHEMT NA 18 085ndash1335 25 NA 1162 70lowastlowast FeedbackThis work 018120583m CMOS ltminus9 9 3ndash56 46ndash53 2 (53 GHz) 08 9 ProposedlowastOnly core LNA lowastlowastSimulation

minus25 minus20 minus15 minus10 minus5 0 5minus80

minus70

minus60

minus50

minus40

minus30

minus20

minus10

0

10

Input power (dBm)

Out

put p

ower

(dBm

)

FundamentalIMD3

IIP3

Figure 24 Measured IIP3at 53 GHz

(CW) generators and an Agilent E4407B spectrum analyzerA noise figure (NF) is measured using an Agilent N8975ANF meter with an Agilent 346C noise source The simulationnoise figure minimum is 41 dB as shown in Figure 25 Dueto the parasitic effect in the layout the minimum of themeasurement is 46 dB Figure 26 shows the micrograph ofthe LNA Table 2 summarizes the measurement results andcompares them with previous literature In the proposedLNA topology the simplified RLC input matching networkand the current-reused configuration benefit from the low-power design The distributed amplifier [7ndash9] requires highpower consumption and a large chip area Compared withthe Chebyshev filter and feedback network synthesis [11ndash13] the proposed input matching network which has onlyone spiral inductor simplifies the circuit complexity and

30 35 40 45 50 55 60 65 70 75 800

1

2

3

4

5

6

7

8

9

10N

oise

figu

re (d

B)

Measured NFSimulated NF

Frequency (GHz)

Figure 25 Simulation and measurement NF

Figure 26Micrograph of the proposed LNA (size 103times 078mm2)

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

International Journal of

AerospaceEngineeringHindawi Publishing Corporationhttpwwwhindawicom Volume 2014

RoboticsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Active and Passive Electronic Components

Control Scienceand Engineering

Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

International Journal of

RotatingMachinery

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporation httpwwwhindawicom

Journal ofEngineeringVolume 2014

Submit your manuscripts athttpwwwhindawicom

VLSI Design

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Shock and Vibration

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Civil EngineeringAdvances in

Acoustics and VibrationAdvances in

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Electrical and Computer Engineering

Journal of

Advances inOptoElectronics

Hindawi Publishing Corporation httpwwwhindawicom

Volume 2014

The Scientific World JournalHindawi Publishing Corporation httpwwwhindawicom Volume 2014

SensorsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Modelling amp Simulation in EngineeringHindawi Publishing Corporation httpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Chemical EngineeringInternational Journal of Antennas and

Propagation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Navigation and Observation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

DistributedSensor Networks

International Journal of

10 Active and Passive Electronic Components

reduces the chip area References [10ndash12] only show core LNApower consumption excluding the output source followerReferences [14 15] only show simulation It should show thelinearity of the experiment The LNA plays an important rolein improving overall system linearity Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption

5 Conclusion

An RLC wideband matching UWB LNA has been presentedin the above results which can operate at a supply of 1 Vvoltage in a 018 120583m CMOS technology In the proposedtopology the narrowband LNA can be converted into awideband amplifier by the RLC matching method The RLCinput matching circuit reduces the chip area The outputmatching method reduces power consumption The mainadvantages of the LNA topology are low power use moderatenoise linearity power gain and a small chip area

Conflict of Interests

The author indicated no potential conflict of interests

Acknowledgments

The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport

References

[1] R Harjani J Harvey and R Sainati ldquoAnalogRF physical layerissues for UWB systemsrdquo in Proceedings of the 17th InternationalConference on VLSI Design pp 941ndash948 January 2004

[2] Z Bai S Xu W Zhang and K Kwak ldquoPerformance analysis ofa novel m-ary code selected DS-BPAM UWB communicationsystemrdquo Journal of Circuits Systems and Computers vol 15 no3 pp 455ndash466 2006

[3] Q YangH Zhu andK S Kwak ldquoSuperimposed training-basedchannel estimation for MIMO-MB-OFDM UWB systemsrdquoJournal of Circuits Systems and Computers vol 17 no 5 pp865ndash870 2008

[4] J Jung T Yun and J Choi ldquoUltra-wideband low noise amplifierusing a cascode feedback topologyrdquo Microwave and OpticalTechnology Letters vol 48 no 6 pp 1102ndash1104 2006

[5] S M R Hasan and J Li ldquoA 12dB 07V 850W CMOS LNAfor 866MHz UHF RFID readerrdquo Active and Passive ElectronicComponents vol 2010 Article ID 702759 5 pages 2010

[6] S Toofan A R Rahmati A Abrishamifar and G RoientanLahiji ldquoA low-power and high-gain fully integrated CMOSLNArdquo Microelectronics Journal vol 38 no 12 pp 1150ndash11552007

[7] B M Ballweber R Gupta and D J Allstot ldquoA fully integrated05ndash55-GHz CMOS distributed amplifierrdquo IEEE Journal ofSolid-State Circuits vol 35 no 2 pp 231ndash239 2000

[8] X Guan and C Nguyen ldquoLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively

coupled common-source gain cells for UWB systemsrdquo IEEETransactions on Microwave Theory and Techniques vol 54 no8 pp 3278ndash3283 2006

[9] C-P Chang and H-R Chuang ldquo018 120583m 3-6 GHz CMOSbroadband LNA for UWB radiordquo Electronics Letters vol 41 no12 pp 696ndash698 2005

[10] C-W KimM-S Kang P T Anh H-T Kim and S-G Lee ldquoAnultra-wideband CMOS low noise amplifier for 3-5-GHz UWBsystemrdquo IEEE Journal of Solid-State Circuits vol 40 no 2 pp544ndash547 2005

[11] A Bevilacqua and A M Niknejad ldquoAn ultrawideband CMOSlow-noise amplifier for 31-106-GHz wireless receiversrdquo IEEEJournal of Solid-State Circuits vol 39 no 12 pp 2259ndash22682004

[12] M-F Chou W-S Wuen C-C Wu K-A Wen and C-Y Chang ldquoA CMOS low-noise amplifier for ultra widebandwireless applicationsrdquo IEICE Transactions on Fundamentals ofElectronics Communications and Computer Sciences vol e88-A no 11 pp 3110ndash3117 2005

[13] Y-J E Chen and Y-I Huang ldquoDevelopment of integratedbroad-band CMOS low-noise amplifiersrdquo IEEE Transactions onCircuits and Systems I vol 54 no 10 pp 2120ndash2126 2007

[14] C Wang J Jin and F Yu ldquoA CMOS 2-11 GHz continuousvariable gain UWB LNArdquo IETE Journal of Research vol 56 no6 pp 367ndash372 2010

[15] A Marzuki A Md Shakaff and Z Sauli ldquoA 15 V 085-1335GHZ MMIC low noise amplifier design using optimizationtechniquerdquo IETE Journal of Research vol 55 no 6 pp 309ndash3142009

[16] A Telli S Demir and M Askar ldquoCMOS planar spiral inductormodeling and low noise amplifier designrdquo MicroelectronicsJournal vol 37 no 1 pp 71ndash78 2006

[17] D K Shaeffer and T H Lee ldquoA 15-V 15-GHz CMOS low noiseamplifierrdquo IEEE Journal of Solid-State Circuits vol 32 no 5 pp745ndash759 1997

[18] T H Lee ldquoThe design of CMOS radio-frequency integratedcircuitsrdquo Cambridge 2004

[19] B Razavi Design of Analog CMOS Integrated Circuit McGraw-Hill 2001

[20] D M PozarMicrowave and RF Design of Wireless System JohnWiley amp Sons 2001

International Journal of

AerospaceEngineeringHindawi Publishing Corporationhttpwwwhindawicom Volume 2014

RoboticsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Active and Passive Electronic Components

Control Scienceand Engineering

Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

International Journal of

RotatingMachinery

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporation httpwwwhindawicom

Journal ofEngineeringVolume 2014

Submit your manuscripts athttpwwwhindawicom

VLSI Design

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Shock and Vibration

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Civil EngineeringAdvances in

Acoustics and VibrationAdvances in

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Electrical and Computer Engineering

Journal of

Advances inOptoElectronics

Hindawi Publishing Corporation httpwwwhindawicom

Volume 2014

The Scientific World JournalHindawi Publishing Corporation httpwwwhindawicom Volume 2014

SensorsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Modelling amp Simulation in EngineeringHindawi Publishing Corporation httpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Chemical EngineeringInternational Journal of Antennas and

Propagation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Navigation and Observation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

DistributedSensor Networks

International Journal of

International Journal of

AerospaceEngineeringHindawi Publishing Corporationhttpwwwhindawicom Volume 2014

RoboticsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Active and Passive Electronic Components

Control Scienceand Engineering

Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

International Journal of

RotatingMachinery

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporation httpwwwhindawicom

Journal ofEngineeringVolume 2014

Submit your manuscripts athttpwwwhindawicom

VLSI Design

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Shock and Vibration

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Civil EngineeringAdvances in

Acoustics and VibrationAdvances in

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Electrical and Computer Engineering

Journal of

Advances inOptoElectronics

Hindawi Publishing Corporation httpwwwhindawicom

Volume 2014

The Scientific World JournalHindawi Publishing Corporation httpwwwhindawicom Volume 2014

SensorsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Modelling amp Simulation in EngineeringHindawi Publishing Corporation httpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Chemical EngineeringInternational Journal of Antennas and

Propagation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

Navigation and Observation

International Journal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014

DistributedSensor Networks

International Journal of