reflexive field containment in dynamic inductive power transfer systems

4592 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 9, SEPTEMBER 2014

Reflexive Field Containment in Dynamic InductivePower Transfer Systems

Kibok Lee, Student Member, IEEE, Zeljko Pantic, Student Member, IEEE, and Srdjan M. Lukic, Member, IEEE

Abstract—We present a new topology appropriate for “dy-namic” wireless charging. Possible applications include chargingof electric vehicles or robots moving in a large, predesignated area.We propose a system with a transmitter made from multiple coilscommensurable with the moving receiver(s), and powered by a sin-gle inverter. The proposed system uses the reactance reflected bythe receiver to automatically increase the field strength in coupledportions of the transmitter-receiver system, thus allowing efficientpower transfer and adherence to electromagnetic field emissionstandards without complex shielding circuits, switches, electron-ics, and communication. The power transfer is at its peak when thetransmitting and receiving coils approach their maximum coupling(as defined by the geometrical constraints of the system), resultingin improved system-level efficiency. The presented analysis is sup-ported with simulations and experiments.

Index Terms—Inductive power transfer (IPT), resonant cou-pling, roadway-powered electric vehicles.

I. INTRODUCTION

INDUCTIVE power transfer (IPT) via magnetic couplingis an effective way to transfer power over relatively large

air gaps. Numerous applications of this technology have beenconsidered, from powering portable electronics, to charging bat-teries of electric vehicles (EVs). If the transmitter and receivercoils are commeasurable in size and perfectly aligned, resultingin a relatively large coupling coefficient (k > 0.2), the systemefficiency reaches 90%, with the coil-to-coil efficiency as highas 95% [1]–[3]. In many emerging applications of IPT systems,it is necessary to provide the receiver with lateral and longi-tudinal freedom of movement while being charged. Possibleapplications include charging the battery of an EV while it ismoving (the concept of “dynamic EV charging”) [4]–[7], pow-ering a swarm of robots moving on a flat surface [8]–[10], andpowering automated-guided vehicles or overhead conveyers. Inthese applications, a receiver moves laterally and longitudinallyin a plane parallel to a surface on which the transmitter coil ismounted. The state of the art for powering such systems can be

Manuscript received April 29, 2013; revised July 31, 2013; accepted Octo-ber 7, 2013. Date of current version April 30, 2014. This work was partiallysupported by the National Science Foundation under Award EEC-0812121.Recommended for publication by Associate Editor U. K. Madawala.

K. Lee and S. M. Lukic are with the Department of Electrical and ComputerEngineering, North Carolina State University, Raleigh, NC 27695 USA (e-mail:[email protected]; [email protected]).

Z. Pantic is with the Department of Electrical and Computer Engineering,Utah State University, Logan, UT 84322 USA (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2013.2287262

Fig. 1. Conceptual visualization of the proposed field-focusing approach.

categorized by the transmitter design as single-coil designs [4],[7], [11], where the transmitter coil is substantially larger thanthe receiver, or segmented coil designs [9], [10], [12] where thetransmitter is made of multiple lumped coils that are commea-surable in size with the receiving coil (see Fig. 1).

The single-coil designs simplify system control, and providea relatively constant coupling coefficient as the vehicle movesin the design space, but they suffer from three key drawbacks.First, for high-power applications, the elongated coil requirescompensation capacitors to be distributed along the coil to main-tain the voltage below prescribed standards [4]. Second, the fieldemitted in uncoupled sections of the coil needs to be containedto ensure that emission standards are not violated. Third, theresulting coupling coefficient is fairly low, due to the large un-coupled flux of the transmitter coil, which results in low totalefficiency. These issues can be partially addressed by appro-priate magnetic designs that contain the leakage field, and bydistributing the compensation capacitance. For example, an in-novative solution [13] to the leakage flux problem is to usea nonlinear flux guide by covering the transmitter coil with aferromagnetic material, and using a permanent magnet on thereceiver to locally saturate the ferromagnetic material and forma flux link with the receiving coil. Another approach [5] is touse ferromagnetic materials and coils of alternating polarity tominimize the field exposure in the uncoupled sections by chan-neling the leakage flux. However, both approaches require theuse of ferromagnetic material and/or permanent magnets, anddo not reduce the produced field, but simply channel it.

With a segmented transmitter coil design, the issues of fieldcontainment, large transmitter coil self-inductance, and diffi-culties with coil impedance compensation can be addressed.Reduction in coil segment size exacerbates the issues and ad-vantages associated with coil segmentation: small coils can fur-ther contain the leakage flux, thus improving the coupling andefficiency, but results in a complicated design with many bypassswitches and sensors. Still, developing a segmented transmit-ter coil design that contains the field by only powering thecoils that are coupled with a receiver is quite challenging. First,

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LEE et al.: REFLEXIVE FIELD CONTAINMENT IN DYNAMIC INDUCTIVE POWER TRANSFER SYSTEMS 4593

there is a need for precise receiver position feedback to identifywhich coils should be powered. Second, a method to energizeor bypass coils is needed. One option is to power each coilwith a dedicated inverter, which would allow complete controlover the power transfer; however, this approach is typically costprohibitive especially in applications where a large area needsto be powered, as for example, in the EV dynamic chargingapplication.

The segmented transmitter coil design with a comeasurabletransmitter and receiver coils has found application in low-power designs, where the bypass switches are not as complexor as expensive. For example, in [14] and [15], a mesh of coilshas been proposed for wireless charging pads of portable elec-tronics. The transmitter is composed of multiple unloaded coilsthat cover the entirety of the area where the receiver is expectedto be placed, with each unloaded coil commeasurable with thereceiving coil. This system requires a position identification al-gorithm and uses a set of relays to energize only the coils thatare best coupled with the receiver, allowing free-positioning ofthe receiving coil.

In [5], the authors present an effective method to selectivelycancel the field in the sections of the transmitter coil wherethere is no receiver. However, this approach requires: switchingboxes between the transmitter coil sections; vehicle detectionsensors; and cables to control the action of the switching boxes.Furthermore, the system needs to be de-energized for the switchboxes to turn ON and OFF. In addition, each switching boxconsists of eight bidirectional thyristor-based switches and twocoupling transformers with eight capacitors; these componentssignificantly increase the cost and reduce the reliability of thesystem. In [9], multiple transmitter pads are used to transferpower to moving robots. Here, the authors power the entireprimary coil (i.e., all pads), and utilize alternating magneticpoles to contain the leakage field within a couple centimetersabove the pad surface.

In this paper, we propose a novel method for segmentationand compensation of transmitter coils for dynamic IPT applica-tions that results in high-power transfer efficiency to the receiver,and low leakage flux in noncoupled sections, while employinga single power converter and preserving simplicity and reliabil-ity of single-coil designs. The system uses a single inverter topower multiple coil segments, by connecting coil segments inparallel. Power is limited by compensating the coil segmentsso that the coil resonance occurs at a frequency offset from thesystem operating frequency. Due to the large uncompensatedreactance, the current in a given coil segment is limited whenthe coil is uncoupled. The result is a relatively weak field inthe uncoupled segments of the sectionalized transmitter coil.By designing the receiver to reflect a reactance that brings thetransmitter coil into resonance at the operating frequency, thefield generated by the transmitter coil is automatically increasedwhen the receiving coil becomes aligned with it. Therefore, asthe title of this paper suggests, the field produced by the seg-mented transmitter coil can be contained by the position of thereceiver in a reflexive manner (i.e., without any control action).This approach eliminates the use of switching boxes, proposedin [5], does not require any receiver position tracking, presented

in [14] and [15], and eliminates the need for the use of mag-netic materials for field containment, suggested in [4], [5], [9],and [10]. The proposed method is especially effective in appli-cations where 1) one inverter is used to power a transmitter witha segmented coil design, 2) multiple receivers may be coupledwith a single transmitter, and 3) the movement of the receivers ishighly dynamic, making position identification algorithms morechallenging to implement. The remainder of this paper describesthe concept in more detail.

II. RECEIVER DESIGN

An IPT transmitter typically consists of a power electronicsconverter that drives an externally compensated coil, tuned tobe resonant at the switching frequency of the converter. The re-ceiver contains a resonant coil, a conditioning circuit in the formof a rectifier and filter, and a controllable load. The compensationcircuits used at the transmitter and receiver are typically simpleseries or parallel resonant topologies, with more complex com-pensations used when additional operating criteria need to besatisfied. The receiver compensation is tuned at the frequency ofthe field emitted by the transmitter coil, and reflects resistive andreactive impedances back to the transmitter coil. The resistivecomponent is a function of the receiver loading, and correspondsto the power delivered to the load. The reactive component istypically unwanted, since it effectively detunes the resonant coilon the transmitter. In the case of a series-compensated receiver,the reactive component reflected on the transmitter is zero whenthe receiver is in perfect resonance at the operating frequency,and the effect of coupling to stray objects is ignored. In the caseof a parallel-compensated receiver, the reflected impedance Zr

at resonance can be described as [16]

Zr =Vr

Is=

ω0M2

L2(Q − j) (1)

where Vr is the voltage induced in the transmitter coil, Is isthe transmitter coil current, ω0 is the resonant frequency, Mis the mutual inductance between the transmitter coil and thereceiving coil, L2 is the self-inductance of the receiving coil,and Q is the quality factor of a parallel-compensated receivercircuit defined as

Q =Req

ω0L2(2)

where Req is the effective load of receiver. In the case of a dcload RLoad connected to the resonant tank through a rectifier, therelation between the two resistances for parallel-compensatedreceiver is found to be

Req =π2

8RLoad . (3)

Based on (1), the parallel-compensated receiver reflects areal component proportional to the coupling coefficient and thequality factor. The reactive component is proportional to themutual coupling. As the quality factor increases, the real com-ponent reflected back onto the transmitter segment is increasedwhile the reflected reactive component is unchanged. The series-compensated receiver, on the other hand, reflects a purely real


Fig. 2. Transmitter coil segment coupled to the series–parallel-compensatedLCC receiver.

load. In this study, we seek to design a compensation topol-ogy that intentionally increases the reactance reflected onto thetransmitter and uses it to control the current flow in segmentedtransmitter coils. We strive to limit the current (and, therefore,the emitted field intensity) in the uncoupled segments, whileincreasing the current as the receiver aligns with the transmittercoil segment.

A. Proposed Receiver Design

Fig. 2 shows a single transmitter coil segment coupled to areceiver that is series–parallel-compensated using two capac-itors and will be named the LCC-compensated receiver. Thistopology has been proposed in [17] and [18], where C1 partiallycompensates the inductance of the receiving coil (thus increas-ing the short-circuit current of the receiver), while C2 boosts thevoltage at the receiver. In fact, authors in [17] define the currentand voltage boost factor (QI and QV ) using this analogy, andwe adopt the same terminology. The compensation capacitorsC1 and C2 are chosen to form a resonant tank at the operatingfrequency

C1 =n

n − 1· C; C2 = n · C; C =

C1C2

C1 + C2(4)

with the resonant frequency of the receiver defined as

ω0 =1√L2C

. (5)

Here, the value of an auxiliary variable C is determined from(5), and ω0 is the operating frequency of the circuit. The ratioof C1 to C2 is n − 1. We name the parameter n the tappingcoefficient, realizing that this coefficient represents the fictitioustapping of the parallel compensation capacitor. The reactancevalue of each branch of the resonant circuit in Fig. 2 is given by

ω0L2 −1

ω0C1=

1ω0C2

. (6)

The quality factor of the system at the resonant frequencycan be calculated by finding the current and voltage boost factor(QI and QV .). The current boost factor (QI ) is found from (4)and (6) as described in [17]

QI =IL

Isc=

ω0L2

ω0L2 − 1ω0 C1

=C2

C1+ 1 = n (7)

where Isc=Voc /jω0L2 is the short-circuit current of the receivingcoil and IL is the input current of the rectifier. The voltage boost

Fig. 3. Transmitter design.

factor of receiver (QV ) is defined as

QV =Vac

Voc= ω0ReqC2 =

Req

ω0L2· n = Q · n. (8)

Here, Vac is the input voltage of the rectifier, and Voc is theopen-circuit voltage induced by the transmitter coil current. Q isthe quality factor of the parallel-compensated receiver definedin (2), with Req defined in (3). The quality factor (Qtotal) of anLCC receiver is the product of QI and QV

Qtotal = QI · QV =Req

ω0L2· n2 = Q · n2 . (9)

From (9), we conclude that the quality factor of the LCCreceiver depends on the tapping coefficient n, and it is differentfrom that of a traditional parallel-compensated receiver. Thecurrents of the LCC receiver also depend on n

Iin = IL + j · IC 2 = Isc · n + j · Isc · Qtotal (10)

where Iin is the coil current, IC 2 is the current of parallel capac-itor. If n is chosen to be one, the currents of the LCC receiverare consistent with that of a parallel-compensated receiver.

Finally, we derive the impedance reflected onto the transmitterfrom the LCC receiver. The impedance (Zreceiver) seen by theopen-circuit voltage Voc in Fig. 2 is calculated as

Zreceiver = jω0L2 +1

jω0C1+

(Req

∥∥∥∥ 1jω0C2

)

=ω0L2

n

QV + j

1 + Q2V

. (11)

The load impedance reflected back onto the transmitter coilsegment is

Zr =(ω0M)2

Zreceiver=

ω0M2

L2(Qtotal − n · j). (12)

From (12), we conclude that the reactive component reflectedonto the transmitter is now controllable by choosing the appro-priate value of n.

III. TRANSMITTER DESIGN

The proposed transmitter coil design consists of a bandpassfilter formed by LF and CF , a parallel compensation capacitorCcomp , and multiple series-compensated transmitter coils con-nected in parallel (see Fig. 3). The branch impedance of a singletransmitter coil is formed by the coil inductance Ls1 and the


compensation capacitor Cs1

Zs1 = jω0Ls1 +1

jω0Cs1= jΔX (13)

where ΔX is set to equal the reactance which will be reflectedfrom the LCC receiver in the predetermined coupled condition,as defined in (12)

ΔX = nω0M

20

L2. (14)

Here, M0 is a constant mutual inductance and is determinedbefore designing the transmitter coil. Since the reflected reac-tance is the function of the mutual coupling M , the target valueof reflected impedance ΔX is only achieved for a predeterminedcoupling, which we define as “perfectly aligned” condition thatcorresponds to M0 . Based on (12), (13), and (14), when thetransmitter coil is perfectly aligned with the LCC receiver, theseries-compensated transmitter coil becomes purely resistive

Zs1 = jω0Ls1 +1

jω0Cs1+ Zr =

ω0M20

L2Qtotal = Rr . (15)

To reduce inverter switching loss generated by the reactivecurrent flowing through the uncoupled coils when multiple coilsare connected to the inverter in parallel, we introduce a com-pensation circuit composed of a parallel capacitor Ccomp and abandpass filter LF CF . The circuit should be designed consider-ing the case when none of the segmented coils is coupled withthe receiver. In this mode, the impedance seen by the invertershould tend toward infinity. Looking at the impedance Zc asdefined in Fig. 3, while ignoring the parasitic resistances of theinductors and capacitors, we arrive at

Zc,no−load =1

N t o t a ljΔX + jω0Ccomp

. (16)

Here, Ntotal is the number of transmitter coils connectedto the inverter in parallel, and ΔX is the impedance of eachtransmitter branch as defined in (14). Since the denominator of(16) should be set to zero to obtain infinite impedance, Ccompequates to

Ccomp =Ntotal

ω0ΔX. (17)

The bandpass filter LF CF simply serves to eliminate the highorder current harmonics.

Considering that a number of transmitter coil segments maycouple with multiple identical receivers, the impedance of thesecoupled segments becomes purely resistive for perfectly alignedconditions, and the expression for the impedance Zc in Fig. 3becomes

Zc,loaded =1

N c o u p le dRr

+ N t o t a l−N c o u p le djΔX + jω0Ccomp

=Rr

Ncoupled− jω0Ccomp

R2r

Ntotal · Ncoupled. (18)

Here, we assume that R2r � ΔX2 , where Rr is the real re-

flected load defined in (15), and Ncoupled is the number of cou-pled transmitter coil segments. In (18), the impedance Zc,loaded

Fig. 4. Current flow in the transmitter coils.

has a small capacitive component, which is a function of thenumber of uncoupled coil segments. The LC filter should, there-fore, be designed to maintain a lagging inverter current withrespect to the voltage, which results in more efficient operationof the inverter switches [19].

Fig. 4 shows the general case when some of the transmittercoil segments are coupled with the receiver while the others areuncoupled. The current in each segment of the transmitter coilis found as

Is = Vs ·(

jω0CcompNcoupled

Ntotal+

Ncoupled

Rr

)(19)

IC comp = Vs · jω0Ccomp (20)

It,coupled =Vs

Rr; It,uncoupled =

Vs

jΔX. (21)

In (19)–(21), the voltage drop across the LF CF filterimpedance is neglected to simplify the analysis. From (19),the inverter output current is proportional to the number of cou-pled coils, and almost purely resistive. The inverter current willbe close to zero when all coils are uncoupled with the receiver.The circulating current flowing in the compensation capacitorCcomp is determined by the total number of the transmitter coilsconnected to the inverter. When the transmitter coil is perfectlyaligned with the receiver, the current is determined by reflectedreal load, while the current of the uncoupled coils is determinedby the reactive component ΔX.

IV. SYSTEM DESIGN CONSIDERATIONS

A. Field Focusing Considerations

In this section, we investigate the field focusing capabilities ofthe proposed system, and compare the proposed system opera-tion with the state-of-the-art compensation approaches: LCL-compensated-transmitter with parallel-compensated-receiver(LCL/LCp) and series-compensated-transmitter with parallel-compensated-receiver (LCs/LCp). We present the field focusingcapability of the three systems by considering three equivalentdesigns that are all nominally designed to transfer 300 W to theload. The compensation circuit parameters for the three systemsare given in Table I.

As described earlier, our goal is to minimize the magneticfield emissions in the uncoupled transmitter coils, while max-imizing the field in the section of the transmitter where thereceiver is perfectly aligned with a transmitter coil. Since thefield strength emitted by the transmitter coil is proportional toits current, an appropriate metric of “field focusing” is the ratio


TABLE IDESIGN PARAMETERS ∗(TRANSMITTER TOPOLOGY/RECEIVER TOPOLOGY)

of the transmitter coil currents in the perfectly aligned and inuncoupled condition, or the “current gain” between the alignedand uncoupled conditions. Using parameters in Table I, the cur-rent gain for the three systems is shown in Fig. 5 by plotting thecurrent magnitude as a function of frequency for three positionsof the receiver (uncoupled −M = 0, midpoint M = M0 /2, andperfectly aligned M = M0).

First, the LCL topology current is independent of the couplingwith the receiver, as shown in Fig. 5(a). This is a well-knownproperty of the LCL compensation [20], [21]. Next, we comparethe LCs/LCC and LCs/LCp topologies. In the aligned position(M = M0), the transmitter currents are set to be equal in bothsystems since the quality factors and power transfer is set tobe the same. When the receiver is removed (M = 0), in thecase of the parallel-compensated receiver, the transmitter currentincreases substantially, due to the proximity of the operatingfrequency and the unloaded resonance of the transmitter coil.On the other hand, in the case of the LCC-compensated receiver,the resonance in the uncoupled case is shifted further from theoperating point of 100 kHz, resulting in a lower current for theuncoupled condition.

It is interesting to note system behavior when the receiver islocated in between the perfectly aligned and uncoupled positions(M = M0 /2): the transmitter current in the parallel-compensatedreceiver is still much larger than for the LCC-compensation.A large field current in the relatively poor coupling conditionis detrimental to system efficiency, as well as safety, since theleakage field could be substantial, leading to safety concerns,or substantial losses in the shielding material. Therefore, weconclude that the goal of “field focusing” is achieved with theproposed method of LCC compensation.

In calculating the ratio of current magnitudes between cou-pled and uncoupled coil segments, we note that the current in theuncoupled segment will be the ratio of the input voltage and thereactance of the coil segment. Similarly, the current in coupledsegment will be the ratio of the input voltage and the reflectedresistance of the coil, since the reactive component is canceledout by the reflected load. Therefore, the “current gain” of thetransmitter coil, when using the parallel-compensated receivercan be found from (1) as

Igain,parallel =It,coupled

It,uncoupled=

|Vin/Re(Zr )||Vin/Im(Zr )|

=1Q

(22)

Fig. 5. Transmitter coil current as a function of the receiver position andoperating frequency. Top: LCL-compensated transmitter with parallel LC re-ceiver (LCL/LCp). Middle: Series-compensated LC transmitter with parallel LCreceiver (LCs/LCp). Bottom: Series-compensated LC transmitter with series–parallel LCC-compensated receiver (LCs/LCC).

where Vin is the rms value of the voltage applied to the coil,which is a known constant regardless of the coupling condi-tion. Similarly, the current gain in case of an LCC receiver iscalculated from (12) as

Igain,LCC =n

Qtotal=

QI

Qtotal=

1QV

. (23)

Parasitic resistances of the transmitter coil and compensationcapacitor are neglected to simplify the analysis. Comparing (22)and (23), the LCC compensation provides an additional degreeof freedom over the LC compensation: by choosing a sufficientlylarge value of current boost factor QI (n), the overall quality


Fig. 6. Effect of Qtota l and n on the current gain.

factor can be set to the desired value while providing the desiredcurrent gain. Fig. 6 shows that the current gain increases as nincreases and Qtotal decreases. It should be noted from (9) thatQtotal is a function of n2 .

B. Power Transfer Considerations

Next, we look at the power transferred to the receiver forthe three topologies listed in Table I. Looking at the proposedcompensation system first, the transferred power is a functionof both the transmitter coil current and real load reflected fromthe receiver, which varies with the coupling condition

P =V 2

s

n2 (k20 − k2)2 + (k2Qtotal)

2k2Qtotal

ω0Ls. (24)

Here, k0 is the coupling coefficient between the transmitterand receiver for perfectly coupled condition, while k is thecoupling coefficient for arbitrary coupling condition, and Ls isthe transmitter coil inductance. Similarly, the power deliveredto the load for the LCs/LCp topology is found to be

P =V 2

s

(k20 − k2)2 + (k2Q)2

k2Q

ω0Ls. (25)

The LCL-compensated transmitter coil has constant currentregardless of the position of the receiver [20], [21]. There-fore, the transferred power to the parallel-compensated receiverbecomes

P = k2 (It)2 ω0LsQ (26)

where Q is the quality factor of the parallel-compensated re-ceiver, and Ls is the transmitter coil inductance. The powertransfer for the LCL-compensated transmitter is, therefore, pro-portional to k2 .

Fig. 7 shows the transmitter coil current and power transferredto the load as a function of the receiver position for the threetopologies. The relation between the receiver position and the

Fig. 7. Top: Effect of misalignment on transmitter coil current and Bottom:power transfer. Pad width is 350 mm.

coupling coefficient was obtained experimentally. Transmitterand receiver windings are identical and the size of designed coilpads are 350 mm × 350 mm. Air gap between the transmitterand receiver pads is set at 170 mm. Coil inductance and qualityfactor and power transfer for all three systems are the same atthe aligned position, with parameters provided in Table I.

Analyzing results in Fig. 7(a), it is apparent that the proposeddesign limits the field emitted by the transmitter as the receiveris misaligned. In contrast, the current (and therefore the emittedfield) is relatively constant for the LCL design regardless ofthe transmitter–receiver mutual coupling, while the current forthe LC/LC system reaches its minimum when the receiver isaligned. Therefore, flux focusing is achieved using the proposedcompensation. Looking at the power transfer to the receiver,shown in Fig. 7(b), the power delivered to the receiver is almostconstant for a misalignment of about 30 mm, after which thepower transfer quickly drops.

Effectively, the proposed design transfers power when thetransmitter–receiver coupling is good, while limiting it whenthe receiver is misaligned. The advantage of the proposed ap-proach is that the power transfer is limited in the poor couplingcondition, when the power transfer is inefficient, and when thereis a higher probability that the receiver will couple with strayobjects (for example, the metallic vehicle chassis in vehicle dy-namic charging application). Note that the power profile as afunction of misalignment for the proposed system, shown inFig. 7(b), is not unique: the high power transfer area can bewidened or narrowed by the appropriate coil design and selec-tion of the operating quality factor. The optimal power profileis application-specific, and is beyond the scope of this paper.


Fig. 8. Transmitter and receiver equivalent circuit with compensation circuitlosses included.

C. Efficiency Considerations

An important aspect of the system design is to consider overallsystem efficiency, keeping in mind that a subset of coil segmentswill couple the receiver, while other coils will remain uncoupled.For comparison and to determine system design parameters, weanalyze a single coil segment in the coupled and uncoupledcondition. The overall system efficiency is determined by thenumber of coupled and uncoupled segments at any given time.At this point, we only consider the losses in the compensationcircuit, realizing that the losses in the power electronics can beminimized with the appropriate choice of switching schemes.

Looking at the uncoupled condition first, the “idle” loss willbe the result of the circulating currents in the transmitter com-pensation circuit, and it can be evaluated from (19)–(21)

Ploss = I2s (RLF + RC F ) + I2

C compRC comp

+ I2t,uncoupled (RC s1 + RLs1) (27)

with the parasitic resistances presented in Fig. 8. The current Is

will be relatively small for the uncoupled condition, allowingthe losses of LC filter to be neglected. We note that the losses areminimized by increasing the reactive component ΔX , which isproportional to the parameter n, as shown in (14).

Similarly, for the perfectly coupled condition, efficiency canbe calculated by evaluating the losses in the parasitic resistancesof the transmitter and the receiving coils. Calculating the trans-mitter efficiency first

ηtransmitter

=ω0 M 2

0L2

QtotalI2t,coupled(

ω0 M 20

L2QtotalI

2t,coupled + (RLs1 + RC s1)I2

t,coupled

+RC compI2C comp + (RLF + RC F )I2

s

) . (28)

The receiver efficiency is

ηreceiver =n2Req

n2Req + Q2totalRC 2 + (n2 + Q2

total)(RL2 + RC 1).

(29)The overall system efficiency is obtained by multiplying (28)

and (29)

ηtotal = ηtransmitter · ηreceiver . (30)

Based on the discussion above, we realize that increasingn directly reduces the loss in the uncoupled sections, whilemaximizing (30) reduces the losses in coupled sections. Evalu-ating (30) numerically for the parameters in Table I, we arrive at

Fig. 9. Efficiency, current gain and transfer power in relation to Qtota l . Top:efficiency and current gain and Bottom: transfer power.

Fig. 9(a), which shows that for higher values of Qtotal efficiencyincreases, while the current gain reduces. System efficiency re-duces with increasing tapping coefficient n, while the currentgain increases. The peak power occurs at relatively low values ofQtotal , as shown in Fig. 9(b). The operation at a higher Qtotalresults in higher system efficiency while reducing the powertransfer and the current gain. The system design, therefore, re-quires a tradeoff between efficiency, current gain, and powertransfer. Consequently, using the parameters given in Table I,results in the calculated efficiency of 93%, a current gain of 3.7,at the operating Qtotal of 1.5. This analysis is experimentallyverified in Section V.

D. Sensitivity to Capacitance Variation

Since the proposed method relies on the reactance reflectedfrom the receiver back onto the transmitter coil to achieve thedesired field control, precise tuning of the resonant tank isparamount to realizing the expected system performance. Here,we investigate the sensitivity of the proposed compensation sys-tem to variations in resonant tank parameters. Inductance vari-ations in the coils could occur due to magnetic coupling withstray objects, while the variations in the capacitances could re-sult from manufacturing or temperature tolerances [22]. Thetransmitter coil design is a simple series LC circuit, and anydetuning in the parameters would result in a shift in the resonantfrequency ω = 1/

√LC. Since the receiver coil is designed to

inject reactance back into the transmitter coil, any variation in


Fig. 10. Tapping coefficient n as a function of series branch impedancevariation.

the resonant frequency will result in transmitter coil resonanceoccurring at a coupling that differs from the perfectly alignedcondition. Generally, any shift in the transmitter resonance ismore easily tolerated for systems that rely on a large reactanceinjection from the receiver, thus minimizing the relative influ-ence of transmitter detuning.

On the receiver, as described in Section II, the series capaci-tance C1 and inductance L2 determines the tapping coefficient,or current gain n = QI , while the parallel capacitance C2 de-termines the voltage gain QV . Since the voltage gain must besmaller than one to achieve good field focusing (i.e., large in-crease in the transmitter current when the receiver is alignedwith it), the system is quite insensitive to the variations in C2 .Focusing on the variation of C1 and L2 , and referring to (7), wenote that the ratio of the impedances of these two elements deter-mines the tapping coefficient n. System sensitivity to impedancedetuning rises with n as shown in Fig. 10. As an example, forthe system described in Table I with n = 5.9, a 5% variationin C1 results in 4.78 < n< 7.93, while 10% variation in C1results in 4.07 < n< 12.9. In turn, a change in n affects thecoupling condition at which the transmitter current reaches itspeak, since the reflected impedance on the transmitter was foundto be nω0M

2 /L2 . In other words, deviation of n from the de-signed value could result in the peak transmitter current whenthe receiver is not perfectly aligned, leading to lower systemefficiency and power transfer. Note that the effect of the cou-pling M is stronger than that of tapping coefficient n, reducingsystem sensitivity.

Still, if the goal is to transfer a fixed amount of power to thereceiver, even as n deviates from the designed value, the loadresistance can be controlled via a boost converter to deliver thenominal power. Fig. 11 shows that by changing Req , the powerdelivered to the load at perfect alignment can be maintained atthe nominal value in face of a 5% variation in C1 . Note thatthe variation in L2 is equivalent to the variation in C1 , in termsof the effect on the tapping coefficient n. The plot also provesthat the delivered power is relatively insensitive to variations inC2 . Changing Req controls the voltage gain QV , and thereforeQtotal , allowing the delivered power to stay at the nominal valuein face of variations in n or, equivalently, C1 . After a certaindegree of detuning, increasing the quality factor further can nolonger maintain the delivered power at the rated value, due tothe higher selectivity of the circuit, as described in [22] for theparallel-compensated receiver.

Fig. 11. Power transfer to the load as C1 and C2 are varied; Req is chosen tomaintain the power transfer at 300 W. Top: power delivered to the load. Bottom:choice of Req that result in 300 W power transfer.

Fig. 12. Experimental setup with three transmitter coil windings made of Litzwire, a receiver with ferromagnetic material embedded in a plastic holder andcovered with an aluminum shield; a small receiver powering an LED is placedon each coil to visually demonstrate the field containment as the receiver moves.

V. EXPERIMENTAL VALIDATION

A set of experiments was conducted to validate the perfor-mance of the proposed system. As shown in Fig. 12, the practicalprototype consists of a transmitter power supply, three identicaltransmitter coils, an LCC receiver, rectifier, filter, and a resis-tive load. The transmitter coils are designed as simple inductiveloops with no ferromagnetic material used to channel the flux(see Fig. 12). The receiver on the other hand contains both ferro-magnetic material to channel the flux, and an aluminum plate tominimize the interaction with the vehicle chassis or other objectslocated above the receiver. The parameters of the experimental


TABLE IISYSTEM PARAMETERS

TABLE IIIEXPERIMENT PARAMETERS AND RESULTS

setup are listed in Tables II, while III shows the power transfermeasurements when the receiver is aligned with a transmittercoil and the receiver quality factor (i.e., loading) is varied to threedifferent values. The power transfer is maintained at 300 W forall three cases by varying the inverter input voltage. The tableshows that the efficiency of the power transfer improves as thecurrent gain reduces, in line with the analysis in Section IV-C.Figs. 13 and 14 show the current and voltage of the inverterand the transmitter coil when n = 5.9. The plots show that theinverter current is close to zero in the uncoupled condition; thecurrent can be further reduced by increasing the inductance ofthe bandpass filter to further attenuate the high-frequency har-monics. The current in the transmitter coil is higher than throughthe inverter, and almost entirely inductive, in line with the factthat little real power is consumed by the coil.

To verify the field focusing capability of the system, we mea-sured the power transfer to the receiver as it moves with respectto the transmitter coils, as shown in Fig. 15. The experimentalmeasurements, along with the expected power transfer, calcu-lated as described in Section IV-B, are shown in Fig. 15. Thesystem is designed to transfer 300 W when the receiver coil isperfectly aligned with any of the three transmitters (i.e., whenΔx = 0, 350, and 700 mm). In the intermediate positions, thepower to the receivers drops very quickly to near zero, in linewith the calculations in Section IV-B. Comparing the predictedand measured values, there is a slight mismatch when the re-ceiver is in between the two transmitter coils, and the transferredpower is larger than the calculations. This mismatch is attributedto neglecting the mutual coupling terms between the variouscoils that participate in the power transfer in the analysis [23],

Fig. 13. Top: Inverter waveforms inverter current and voltage for perfectlyaligned conditions CH1: inverter voltage (50 V/div) CH2: inverter current(5 A/div). Bottom: Inverter current and voltage for uncoupled conditions CH1:inverter voltage (50 V/div) CH2: inverter current (5 A/div).

Fig. 14. Transmitter coil waveforms. Top: Transmitter coil current and voltagefor perfectly aligned conditions CH1: inverter voltage (50 V/div) CH2: invertercurrent (5 A/div). Bottom: Transmitter coil current and voltage for uncoupledconditions CH1: inverter voltage (50 V/div) CH2: inverter current (5 A/div).

[24]. An important note here is that the experimental setup isnot magnetically optimized for a given application, and mag-netic design optimization is considered to be beyond the scope


Fig. 15. Power transferred to the receiver as a function of the receiver positionwith respect to the transmitter coil (top), system layout (bottom) simulationsand experimental measurements.

of this paper. The magnetic design of the transmitter coil witha segmented coil design is looked at in detail in, for example,[8].

VI. CONCLUSION

A new IPT system was designed by using a series-compensated transmitter coil and a series–parallel-compensatedLCC receiver. The advantage of using the LCC receiver hasbeen described and compared to parallel-compensated receiverin terms of reflected load onto the transmitter coil. A compen-sation circuit is developed to reduce the inverter losses result-ing from the reactive current of the uncoupled transmitter coilswhen multiple transmitter coils are connected to the inverter.The experimental results verify that this proposed system canautomatically control the current in the transmitter coils as afunction of the coupling with the receiver.

The main benefit of the proposed system over other methodsof powering a sectionalized transmitter coil is that it obviates theneed for receiver position sensors and for switches required toenergize selected portions of the sectionalized coil. As a result,the field strength in the uncoupled potions is reduced, and thereis no need for complex methods of field containment proposed inother work. It should be noted that this paper simply introduces anew compensation circuit, and that therefore this approach canbe used in conjunction with other segmentation methods thathave been proposed, as well as methods to contain the leakageflux in dynamic IPT systems. We envision this dynamic IPTsystem to operate by delivering power in pulses to a constant dcload as the receiver moves along the sectionalized transmittercoil. Power flow regulation can then be controlled by modulatingthe number of pulses rather than by changing the power andenergy transferred in each pulse.

REFERENCES

[1] H. H. Wu, A. Gilchrist, K. D. Sealy, and D. Bronson, “A high efficiency5 kW inductive charger for EVs using dual side control,” IEEE Trans. Ind.Inform., vol. 8, no. 3, pp. 585–595, Aug. 2012.

[2] M. Budhia, G. A. Covic, and J. T. Boys, “Design and optimization ofcircular magnetic structures for lumped inductive power transfer sys-tems,” IEEE Trans. Power Electron., vol. 26, no. 11, pp. 3096–3108, Nov.2011.

[3] M. Pinuela, D. C. Yates, S. Lucyszyn, and P. D. Mitcheson, “MaximizingDC-to-Load efficiency for inductive power transfer,” IEEE Trans. PowerElectron., vol. 28, no. 5, pp. 2437–2447, May 2013.

[4] J. Huh, S. W. Lee, W. Y. Lee, G. H. Cho, and C. T. Rim, “Narrow-width in-ductive power transfer system for online electrical vehicles,” IEEE Trans.Power Electron., vol. 26, no. 12, pp. 3666–3679, Dec. 2011.

[5] J. Huh, W. Lee, S. Choi, and C. Rim, “A new cross-segmented powersupply rail for roadway powered electric vehicles,” in Proc. IEEE 3rdInt. Symp. Power Electron. Distrib. Generation Syst., 2012, pp. 291–296.

[6] M. L. G. Kissin, H. Hao, and G. A. Covic, “A practical multiphase IPTsystem for AGV and roadway applications,” in Proc. IEEE Energy Con-vers. Congr. Expo., 2010, pp. 1844–1850.

[7] J. G. Bolger, F. A. Kirsten, and L. S. Ng, “Inductive power coupling for anelectric highway system,” in Proc. 28th IEEE Veh. Technol. Conf., 1978,pp. 137–144.

[8] J. P. C. Smeets, T. T. Overboom, J. W. Jansen, and E. A. Lomonova,“Comparison of position-independent contactless energy transfer sys-tems,” IEEE Trans. Power Electron., vol. 28, no. 4, pp. 2059–2067, Apr.2013.

[9] L. Byunghun, K. Hyunjae, R. Chun-Taek, L. Sungwoo, andP. Changbyung, “Resonant power shoes for humanoid robots,” in Proc.IEEE Energy Convers. Congr. Expo., 2011, pp. 1791–1794.

[10] P. Changbyung, L. Sungwoo, C. Gyu-Hyeong, C. Su-Yong, and R. Chun-Taek, “Omni-directional inductive power transfer system for mobile robotsusing evenly displaced multiple pick-ups,” in Proc. IEEE Energy Convers.Congr. Expo., 2012, pp. 2492–2497.

[11] G. A. Covic, J. T. Boys, M. L. G. Kissin, and H. G. Lu, “A three-phaseinductive power transfer system for roadway-powered vehicles,” IEEETrans. Ind. Electron., vol. 54, no. 6, pp. 3370–3378, Dec. 2007.

[12] M. Yilmaz, V. T. Buyukdegirmenci, and P. T. Krein, “General design re-quirements and analysis of roadbed inductive power transfer system fordynamic electric vehicle charging,” in Proc. IEEE Transp. ElectrificationConf. Expo., 2012, pp. 1–6.

[13] D. Baarman, J. R. Stoner, T. William, J. Schwanneck, K. Turner, andB. Mose, “Selectively controllable electromagnetic shielding,” U.S. Patent0 112 552, Mar. 29, 2012.

[14] S. Y. R. Hui and W. W. C. Ho, “A new generation of universal contact-less battery charging platform for portable consumer electronic equip-ment,” IEEE Trans. Power Electron., vol. 20, no. 3, pp. 620–627, May2005.

[15] W. X. Zhong, X. Liu, and S. Y. R. Hui, “A novel single-layer windingarray and receiver coil structure for contactless battery charging systemswith free-positioning and localized charging features,” IEEE Trans. Ind.Electron., vol. 58, no. 9, pp. 4136–4144, Sep. 2011.

[16] W. Chwei-Sen, O. H. Stielau, and G. A. Covic, “Design considerations fora contactless electric vehicle battery charger,” IEEE Trans. Ind. Electron.,vol. 52, no. 5, pp. 1308–1314, Oct. 2005.

[17] S. Raabe and G. A. Covic, “Practical design considerations for contactlesspower transfer quadrature pick-ups,” IEEE Trans. Ind. Electron., vol. 60,no. 1, pp. 400–409, Jan. 2013.

[18] N. A. Keeling, G. A. Covic, and J. T. Boys, “A unity-power-factor IPTpickup for high-power applications,” IEEE Trans. Ind. Electron., vol. 57,no. 2, pp. 744–751, Feb. 2010.

[19] R. L. Steigerwald, “A comparison of half-bridge resonant convertertopologies,” IEEE Trans. Power Electron., vol. 3, no. 2, pp. 174–182,Apr. 1988.

[20] Z. Pantic, S. Bai, and S. M. Lukic, “ZCS LCC-compensated resonantinverter for inductive-power-transfer application,” IEEE Trans. Ind. Elec-tron., vol. 58, no. 8, pp. 3500–3510, Aug. 2011.

[21] W. Chwei-Sen, G. A. Covic, and O. H. Stielau, “Investigating an LCL loadresonant inverter for inductive power transfer applications,” IEEE Trans.Power Electron., vol. 19, no. 4, pp. 995–1002, Jul. 2004.

[22] Z. Pantic and S. M. Lukic, “Framework and topology for active tuning ofparallel compensated receivers in power transfer systems,” IEEE Trans.Power Electron., vol. 27, no. 11, pp. 4503–4513, Nov. 2012.


[23] L. Chi Kwan, W. X. Zhong, and S. Y. R. Hui, “Effects of magnetic cou-pling of nonadjacent resonators on wireless power domino-resonator sys-tems,” IEEE Trans. Power Electron., vol. 27, no. 4, pp. 1905–1916, Apr.2012.

[24] H. Matsumoto, Y. Neba, K. Ishizaka, and R. Itoh, “Comparison of charac-teristics on planar contactless power transfer systems,” IEEE Trans. PowerElectron., vol. 27, no. 6, pp. 2980–2993, Jun. 2012.

Kibok Lee (S’13) received the B.S. and M.S. de-grees in electrical engineering from Korea University,Seoul, Korea, in 2005 and 2007, respectively. He iscurrently working toward the Ph.D. degree at NorthCarolina State University, Raleigh, NC, USA.

From 2007 to 2011, he was with LG electronicsR&D Center, Seoul, Korea. Since 2011, he has beenwith the Future Renewable Electric Energy Deliv-ery and Management Systems Center, North CarolinaState University, where he is currently a ResearchAssistant. His primary areas of interest are inductive

power transfer and motor drives.

Zeljko Pantic (S’09) received his B.Sc. and M.Sc.degrees from the School of Electrical Engineering,Belgrade University, Belgrade, Serbia, in 1998 andin 2007, respectively.

In 2001, he joined the group for Power Convertersand Motor Drives as a teaching and research assistantfor motor drive and electric vehicle related coursesand served as a co-instructor. In 2009, he becamea Research Assistant at the NSF-funded Future Re-newable Electric Energy Delivery and Management(FREEDM) Systems Center at North Carolina State

University in Raleigh, North Carolina. As a Ph.D. student at NC State, he ex-plored the emerging topic of wireless inductive power transfer (WIPT). Aftergraduation in 2013, he joined the Department of Electrical and Computer En-gineering, Utah State University, Logan, UT, USA, as an Assistant Professor.His primary areas of interest are systems for wireless inductive power transfer,power electronics, electric and hybrid electric vehicles, and motor drives.

Srdjan M. Lukic (S’02–M’07) received the Ph.D.degree in electrical engineering from the Illinois In-stitute of Technology, Chicago, IL in 2008.

He is currently an Assistant Professor in the De-partment of Electrical and Computer Engineering,North Carolina State University, Raleigh, NC, USA.He serves as the Distributed Energy Storage DevicesSubthrust Leader in the National Science Founda-tion Future Renewable Electric Energy Delivery andManagement Systems Engineering Research Center.His research interests include design, and control of

power electronic converters and electromagnetic energy conversion with appli-cation to wireless power transfer, energy storage systems, and electric automo-tive systems.

Dr. Lukic serves as an Associate Editor of the IEEE TRANSACTIONS ON IN-DUSTRY APPLICATIONS; he has served as a Guest Editor for the Special Sectionof the IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS ON ENERGY STOR-AGE SYSTEMS—Interface, Power Electronics and Control, and has been selectedas the Distinguished Lecturer of the IEEE Vehicular Technology Society for the2011–2015 term.

reflexive field containment in dynamic inductive power transfer systems

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