power electronics

Efficiency Comparison of Power Electronic Convertersused in Grid-Connected Permanent-Magnet Wind

Energy Conversion System Based on SemiconductorsPower Losses

by

Md. Arifujjaman and L. Chang

REPRINTED FROM

WIND ENGINEERINGVOLUME 35, NO. 4, 2011

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Efficiency Comparison of Power ElectronicConverters used in Grid-Connected Permanent-Magnet Wind Energy Conversion System Based onSemiconductors Power Losses

Md. Arifujjaman, Post Doctoral Fellow, and L. Chang, ProfessorDepartment of Electrical and Computer Engineering, University of New BrunswickFredericton, NB, Canada, E3B 5A3E-mail: [email protected]

WIND ENGINEERING VOLUME 35, NO. 4, 2011 PP 445-464 445

ABSTRACTAn efficiency comparison of power converters is presented for a permanent magnet

generator based grid-connected wind energy conversion system. The power converters

examined are: the intermediate boost converter (IBC), the intermediate buck-boost

converter (IBBC), the back-to-back converter (BBC) and the matrix converter (MC). The

aim is to determine which power electronic converter yields the highest efficiency in terms

of power losses of the semiconductor devices with the varying wind speeds. In view of this, a

furled wind turbine model is developed that generates power for different wind speeds.

Afterwards, a relation between the wind speed and power loss is established to evaluate the

efficiency of the power electronic converters for discrete wind speeds. The power loss model

presented in this paper has taken into account the conduction and switching losses of the

semiconductor devices within each converter. Simulation results are presented showing the

power loss characteristics with the variation in wind speeds. Finally, with regard to the

global efficiency for the power electronic converters of the considered wind speed regime,

the IBC is found to be the most favourable choice considering the typical wind conditions

encountered by a wind energy conversion system operating at the kWatt level.

1. NOMENCLATURESθ : Furling angle

d : Duty cycle of the boost and buck-boost converter

ESR : Rated off-state switching loss energy of the diode

EON, EOFF : Rated on and off-state switching loss energy of the IGBT respectively

fWT , fSW : Frequency of the wind turbine rotor and switching frequency of the

semiconductors respectively

Ir,d, Ir,i : Reference commutation current od diode and IGBT respectively

rd, rce : On-state resistance of the diode and IGBT respectively

Vf0, Vce0 : On-state voltage of the Diode and IGBT respectively

Vr,d, Vr,i : Reference commutation voltage of the diode and IGBT respectively

2. INTRODUCTIONGrid-connected wind energy conversion system (WECS) is a cost effective renewable

technology and provides a continuously growing contribution to climate change goals, energy

diversity and security. However, the most important aspect for connecting a WECS with the

electric grid requires a power electronic converter (PEC) that allows variable-speed operation,

reduces mechanical stress, and increases efficiency. Among various performances and design

criteria for the converter, the overall efficiency is one of the most significant factors because of

a growing concern regarding the energy savings and cost. However, a considerable lack in

comparison of a series of converters is observed in the previous literatures that practically

discussed with the precise model of semiconductor devices power loss that varies with the

wind speed and finally, the global efficiency of the converters in the wind energy domain.

The PECs are composed of semiconductor devices and indeed, power losses occur during

operation. Moreover, as the PECs are connected to a wind turbine; the power losses will vary with

the wind speeds. As a result, while calculating the efficiency of a converter, one should also

consider the variation of the losses for each wind speed to clarify the effectiveness of a specific

converter for the entire wind speed regime. A calculation of converter efficiency for a given

operating condition is performed in [1–19] in terms of the total semiconductor power losses.

However, either calculating converter efficiency or individual semiconductor power loss model

lacks a considerable valid justification. This is because, firstly, a non-linear loss model approach is

unable to reflect the switching losses of the semiconductor devices, which could be a dominant

factor during the high switching state [1–8]. Secondly, power loss model based on the data provided

by the manufacturers is ambiguous and pessimistic [9–13]. Thirdly, physics-based simulation

models of semiconductor devices power losses requires implicit integration methods, leading to an

increased simulation time. Furthermore, it requires detail knowledge of the dimensions of the

devices [14, 15]. There have been numerous other efforts for modeling the power losses of

converters used in wind turbine systems. In [16–19] presented the concept of maximum device

rating power loss model of the converter, however, switching losses is often ignored.

Based on the above discussions, it can be asserted that most of the attempts for the

converter efficiency calculation in terms of the power loss model have been developed based

on several assumptions and often neglected a fraction of the entire converter power losses.

This discrepancy could affect the preference of an efficient grid-connected WECS that is in a

great need for high penetration of the wind power. As a consequence, this research aims at

advancing the use of grid-connected WECS by modeling the wind turbine (WT) power

generation and power losses of the semiconductor devices for the most useable converters

with the variation in wind speeds. Based on the power generation and loss for individual wind

speed, a global efficiency is calculated for the considered wind speed regime of each

converter and an efficient converter is determined.

This paper is organized as follows: Followed by a detail literature review in the second

section, the selected converter systems are presented in the third section. The fourth section

describes the global efficiency calculation of the converters. The modeling approach to obtain

the power generation of the furled WT along with the operating conditions is described in the

fifth section, while the sixth section contains the mathematical model for power loss

calculations in the semiconductor devices. Simulation results and discussions are presented

and discussed in the seventh section and finally, the findings of the investigations are

highlighted in the conclusions.

3. POWER ELECTRONIC CONVERTERS FOR GRID CONNECTED WIND ENERGYCONVERSION SYSTEMThe design concept of wind turbine has progressed from induction generator based fixed

speed, flapping/passive pitching-controlled drive train with gearbox to permanent magnet

generator (PMG)-based variable-speed, furling/soft stall-controlled systems with or without

446 EFFICIENCY COMPARISON OF POWER ELECTRONIC CONVERTERS USED IN GRID-CONNECTED

PERMANENT-MAGNET WIND ENERGY CONVERSION SYSTEM BASED ON SEMICONDUCTORS POWER LOSSES

gearbox. The PMG-based grid connected WECS is favorable due to the flexibility in design

which allows smaller and lighter diameter, higher output level may be achieved without the

need to increase generator size, lower maintenance and operating costs and most

importantly, eliminates the need for separate excitation or cooling systems. Fig. 1 has shown

the four potential converters that have been selected for study and which are very common

with the PMG-based WECS. Firstly, an intermediate boost converter (IBC), employs a 3-phase

WIND ENGINEERING VOLUME 35, NO. 4, 2011 447

Figure 1: Typical power electronic converters a) Intermediate boost converter (IBC), b) Intermediate

buck-boost converter (IBBC), c) Back-to-back converter (BBC), d) Matrix converter (MC)

bridge rectifier, a boost converter and a grid-connected pulse width modulated (PWM)

inverter (Fig. 1a) [10, 20]. Secondly, intermediate buck-boost converter (IBBC), employs a 3-

phase bridge rectifier, a buck-boost converter and a grid-connected PWM inverter (Fig. 1b) [21,

22]. Thirdly, back-to-back converter (BBC), composed of a rectifier, DC link and a PWM

inverter (Fig. 1c) [23, 24]. And lastly, matrix converter (MC), employs 9 bidirectional switches

for three phase input and output connection between the generator and grid (Fig. 1d) [25, 26].

Each of the converters has their own advantages and disadvantages and a brief literature

review can be found in [27].

4. EFFICIENCY CALCULATIONIn order to calculate the efficiency of the converters, the relation between the wind speed and

power generation/loss for each wind speed is needed. Once the power generation is known

for discrete wind speed using Section V, the total power generation, Pg is expressed as

(1)

where wi represents a particular wind speed within the wind speed regime and Pg,i (wi) is the

power generation for wi wind speed. . The power loss for the wind speed of each converter can

be found as described in Section VI and the total power loss, Pl is mathematically expressed as

(2)

The global efficiency, η of the converters is then calculated as,

(3)

5. WIND TURBINE MODELA wind turbine can be characterized by the non-dimensional curve of power coefficient, Cp as

a function of tip speed ratio, λ, where, λ is given in terms of rotor speed, ωs (rad/s), wind speed,

w (m/s), and rotor radius, Rw (m) as

(4)

The relationship between Cp and λ can be approximated by a quartic equation. In this

research, the curve is obtained from the literature [28]. A model for Cp as a function of λ is

calculated and the curve generated by the approximate model and the actual data are

presented in Fig. 2a. Statistical analysis shows that the R2 value of the model is 99.8% and the

p-value from the chi-square goodness-of-fit test is less than 0.0001, which shows that the

predicted model for Cp with the fitted coefficients is acceptable. The resulting equation is

found to be

(5)

The curve between wind speed and furling angle is derived from published data [29]. An

approximate model is used to determine the relation between wind speeds and furling angle.

It is found that a fifth order model is sufficient to represent the relationship. The R2 value and

the p-value from the chi-square goodness-of-fit test of the expected model are found to be

C p λ λ λ λ λ( ) = − + − +0 00044 0 012 0 097 0 2 0 114 3 2. . . . .

λ ω= R

ww s

η =−

×P P

P

g l

g

100%

P P wl l i i

i w

wn

= ( )=∑ ,

1

P P wg g i i

i w

wn

= ( )=∑ ,

1



98.19% and less than 0.0001 respectively, thus validating the modeling approach. The modeling

equation for the wind turbine is determined as

(6)

where θ is the furling angle in degree. The actual data and approximated model curve are

represented in Fig. 2b.

The output power of the wind turbine can be expressed as

(7)

where ρ is the air density (kg.m�3) and A is the rotor rotational area, i.e., πRw2.

When the wind speed increases, wind turbines yaw to an angle θ along its horizontal axis

because of the furling action. The effective wind velocity at the rotor plane in that case will be

P AC waero p= ( )0 5 3. ρ λ

θ = − + − + +0 00019282 011317 0 21115 1 179 1 4072 3 16285 4 3 2. . . . . .w w w w w


Figure 2: a) Power coefficient as a function of tip speed ratio, b) Furling angle versus wind speed

wcosθ [30]. Incorporating the furling action, the theoretical power of the wind turbine can be

written from (7) as

(8)

Equation (8) represents the power for varying wind speed of the wind turbine rotor without

accounting for maximum power production. Maintaining a constant optimum tip speed ratio

can ensure the maximum power production. It should be noted that the value of the optimum

tip speed ratio varies from one turbine to another. The relation between power coefficient and

tip speed ratio used in this study requires to assume an optimum tip speed ratio of 7 for the

wind turbine and thus (8) is expressed in terms of the optimum power coefficient as

(9)

where Cp,OPT(λ) is the optimum value of power co-efficient at the optimum TSR.

The theoretical maximum power of the wind turbine described in (9) serves as the dynamic

power reference for all four converters. In the variable-speed systems the speed increases

linearly with the wind speed up to the rated speed, which in this case is reached at 13 m/s. And

from rated to cut-out wind speed, i.e., 17 m/s the turbine speed is lowered using furling action.

The current at the output of the generator is determined by the power generation by the WT

and the generator voltage.

6. POWER LOSS MODELA mathematical model of the power losses in the semiconductor devices (diodes/IGBTs) is

required in order to compare the efficiency of the converters. The losses for the

semiconductor devices are strongly dependent on the voltage and current waveforms.

Simplified analytical derivation of voltage and current equations associated with the

individual semiconductor devices are derived to determine the power losses. The power loss

model presented in this investigation focus on the losses generated during the conduction and

switching states of the semiconductor devices.

6.1. Intermediate Boost Converter (IBC)For the 3-phase diode bridge rectifier, the losses are calculated for a single diode from the known

voltage and current equations. It is assumed that the current and voltage in the 3-phase diode

bridge rectifier are equally distributed in the diodes. As a consequence, considering each diode

current, Id1IBC

and forward voltage, Vfthe conduction losses, Pc1,d-RIBC

for the diode is expressed as [31]

(10)

Under the assumption of a linear loss model for the diodes, the switching loss in each diode,

Ps1,d-RIBC

is given by [32]

(11)

where VdcIBC

and IdcIBC

are the output current and voltage of the bridge rectifier.

The total losses of the 3-phase diode bridge rectifier, Pt,d-RIBC

is expressed as

(12)P P Pt d RIBC

c d RIBC

s d RIBC

, , ,− − −= +( )6 1 1

P f EV

V

I

Is d R

IBCWT SR

dcIBC

r d

dcIBC

r d1,

, ,

. .− =

P V Ic d RIBC

f dIBC

1 1, − =

P = 0.5R AC wmax w p,OPT

3λ θ( )( ) cos

P AC waero p= ( )( )0 53

. cosρ λ θ



The conduction and switching loss of the boost converter is calculated by assuming an ideal

inductor (LDIBC

) at the input. For a boost converter, the IGBT is turned on for a duration d,

while the diode conducts for the duration (1–d). The on-state current of the IGBT is the input

current, IdcIBC

, while the inverter input current, Idc2IBC

is given by [33]

(13)

The conduction loss for the diode, Pc,d-B

IBC and IGBT, Pc,i-B

IBC can be obtained by multiplying

their on-state voltage and current with the respective duty cycle and is given by [33]

(14)

(15)

The actual commutation voltage and current for the boost converter are the DC link voltage,

Vdc2IBC

and input current to the converter, IdcIBC

. The switching loss of the diode, Ps,d-BIBC

and

IGBT, Ps,i-BIBC

in the boost converter are given by [33]

(16)

(17)

The sum of (14) to (17) gives the total losses, Pt,(d+i)-BIBC

(18)

With the exclusion of snubber circuit, the PWM inverter consists of 6 IGBTs and 6 anti parallel

diodes. The conduction losses of a diode, Pc1,d-IIBC

and IGBT, Pc1,i-IIBC

for the inverter can be

expressed as [34],

(19)

(20)

where IomIBC

is the maximum value of the sinusoidal output current, iomIBC

.

For the inverter, the commutation voltage and current are the DC link voltage and output

current. An approximated solution for the switching loss of a diode, Ps1,d-IIBC

and IGBT, Ps1,i-IIBC

is

given by [35]

(21)

(22)

The loss, Pt,(d+i)-IIBC

of the PWM inverter is obtained as the sum of (19) to (22) and expressed by

(23), while the total loss, PtIBC

for the IBC is expressed by (24).

P f E EV

V

I

Is i I

IBCsw ON OFF

dcIBC

r i

omIBC

r i1

21,

, ,− = +[ ]π

P f EV

V

I

Is d I

IBCsw SR

dcIBC

r d

omIBC

r d1

21,

, ,− =

π

P r I V Ic i IIBC

ce omIBC

ce omIBC

1

2

0

1

8

1

3

1

2

1

8, − = +

( ) + +

π π

P r I V Ic d IIBC

d omIBC

f omIBC

1

2

0

1

8

1

3

1

2

1

8, − = −

( ) + −

π π

P P P P Pt d i BIBC

c d BIBC

c i BIBC

s d BIBC

s i BIBC

,( ) , , , ,+ − − − − −= + + +

P f E EV

V

I

Is i B

IBCsw ON OFF

dcIBC

r i

dcIBC

r i,

, ,

. .− = +( ) 2

P f EV

V

I

Is d B

IBCsw SR

dcIBC

r d

dcIBC

r d,

, ,

. .− = 2

P I V r I dc i BIBC

dcIBC

ce ce dcIBC

, .− = +( )0

P I V r I dc d BIBC

dcIBC

f d dcIBC

, .− = +( ) −( )0 1

I I ddcIBC

dcIBC

2 1= −( )


(23)

(24)

6.2. Intermediate Buck-Boost Converter (IBBC)Equation (10) and (11) are used to calculate the conduction and switching losses for the 3-

phase bridge rectifier and the total loss, Pt,d-RIBBC

is expressed by (25)

(25)

The relationship between the average inductor current, IL and the output current, Idc2IBBC

for

the continuous mode buck-boost converter is given by

(26)

Average source current, IdcIBBC

is related to the average inductor current, IL by

(27)

The conduction loss for the diode, Pc,d-BBIBBC

and IGBT, Pc,i-BBIBBC

can be expressed as

(28)

(29)

IdcIBBC

serves as the reference commutation current while, Vdc2IBBC

is the reference

commutation voltage of the buck-boost converter. As a result, the switching loss for the diode,

Ps,d-BBIBBC

and IGBT, Ps,i-BBIBBC

is expressed as

(30)

(31)

The sum of (28) to (31) gives the total losses, Pt,(d+i)-BBIBBC

as

(32)

The PWM inverter used for the IBBC maintains the same characteristic as used with IBC. This

conclusion leads that equation (19) to (22) can be used to calculate the conduction and

switching losses for the PWM inverter of the IBBC and is given by (33), and the total loss, PtIBBC

is expressed by (34)

(33)

(34)P P P PtIBBC

t d RIBBC

t d i BBIBBC

t d i IIBBC= + +− + − +( )−, ,( ) ,

P P P P Pt d i IIBBC

c d IIBBC

c i IIBBC

s d IIBBC

s i IIBBC

, , , , ,+( )− − − − −= + + +( )6 1 1 1 1

P P P P Pt d i BBIBBC

c d BBIBBC

c i BBIBBC

s d BBIBBC

s i BBIBBC

,( ) , , , ,+ − − − − −= + + +

P f E EV

V

I

Is i BB

IBBCsw ON OFF

dcIBBC

r i

dcIBBC

r i,

, ,

. .− = +( ) 2

P f EV

V

I

Is d BB

IBBCsw SR

dcIBBC

r d

dcIBBC

r d,

, ,

. .− = 2

P I V rI

dc i BB

IBBCdc ce ce

dcIBBC

, − = +

0

P I dV

dr

I

dc d BB

IBBCdc

IBBC fd

dcIBBC

, − = −( ) +

1 0

2

I I ddcIBBC

L=

− = −( )I I ddcIBBC

L2 1

P P Pt d RIBBC

c d RIBBC

s d RIBBC

, , ,− − −= +( )6 1 1

P P P PtIBC

t d RIBC

t d i BIBC

t d i IIBC= + +− + − +( )−, ,( ) ,

P P P P Pt d i IIBC

c d IIBC

c i IIBC

s d IIBC

s i IIBC

, , , , ,+( )− − − − −= + + +( )6 1 1 1 1



6.3. Back-to-Back Converter (BBC)It is well understood that the modulation algorithm is always unique for a specific

conventional PWM converter regardless of DC-AC or AC-DC operation mode, the model of

conduction and switching losses is also unique to a specific conventional PWM converter as

far as the conditions of physical devices keep unchanged. In view of these the PWM rectifier

results (35) and (36) for the conduction, Pcc1,d-RBBC

and switching loss, Pcc1,i-RBBC

of the anti

parallel diode respectively. In addition, (37) and (38) presents the conduction, Pss1,d-RBBC

and

switching, Pss1,i-RBBC

loss of the IGBT respectively [36].

(35)

(36)

(37)

(38)

where IamBBC

is the maximum value of the PMG line current at the input of the PWM rectifier.

The total loss, Ptt(d+i),-RBBC

for the PWM rectifier is given by

(39)

Assuming the DC link capacitor average voltage is constant and remains in proper

coordination to work the inverter properly. This also implies that the mean capacitor current

is zero so as the power losses of the capacitor. The power loss of the DC link capacitor due to

leakage current is ignored for the time being.

In a similar manner as applied for the PWM inverter power loss calculation of the IBC and

IBBC, (19) to (22) are adapted to calculate the conduction and switching loss for the PWM

inverter of the BBC and the PWM inverter loss, Ptt(d+i),-IBBC

is expressed by (40), while the total

losses, PtBBC

of the BBC is expressed by (41).

(40)

(41)

6.4. Matrix Converter (MC)Since a suitable bidirectional semiconductor switch can be realized using several ways, this

research considered the back-to-back IGBT arrangement as it allows independent control of

the current in both directions within each switch. In this paper, the control method reported

[37] has been employed because of good input current control characteristics under balanced

input voltage conditions. In this analysis, a 3-phase MC is used and a sinusoidal current output

is assumed. The conduction losses of the diode and IGBT for a single phase are presented by

(42) and (43) respectively.

(42)P V I r Ic dMC

f omMC

d omMC

, = + ( )2 20

2

π

P P PtIBBC

tt d i RBBC

t d i IBBC= ++( )− +( )−, ,

P P P P Pt d i IBBC

c d IBBC

c i IBBC

s d IBBC

s i IBBC

, , , , ,+( )− − − − −= + + +( )6 1 1 1 1

P P P P Ptt d i RBBC

cc d RBBC

cc i RBBC

ss d RBBC

ss i RBBC

, , , , ,+( )− − − − −= + + +( )6 1 1 1 1

P f E EV

V

I

Iss i R

BBCsw ON OFF

dcBBC

r i

amBBC

r i1

1,

, ,− = +[ ]π

P f EV

V

I

Iss d R

BBCsw SR

dcBBC

r d

amBBC

r d1

1,

, ,− =

π

P r I V Icc i RBBC

ce amBBC

ce amBBC

1

2

0

1

8

1

3

1

2

1

8, − = +

( ) + +

π π

P r I V Icc d RBBC

d amBBC

f amBBC

1

2

0

1

8

1

3

1

2

1

8, − = −

( ) + −

π π


(43)

where IomBBC

is the maximum value of the line current at the output of the MC.

The switching losses when normalized with the reference voltage and current results (44)

and (45) for the diode and IGBT respectively.

(44)

(45)

where VaBBC

is the maximum value of the line current at the output of the MC.

The total losses of the MC is then found as

(46)

7. RESULTS AND DISCUSSIONSThe analytical calculations illustrated in the above sections were carried out to determine the

total power generation/losses, and afterwards efficiency in four power electronic converters

under varying wind speed conditions. The rated power for the wind turbine is assumed to be

1.5 kW. The inverter switching frequency is considered as 20 kHz and to investigate the worst-

case scenario of the power loss in this numerical simulation study, the modulation index is

assumed unity and load current is assumed in phase with the output voltage. In addition, the

variation of duty cycle of the IBC and IBBC is considered as the maximum value. This is due to

the fact that the semiconductor devices will transmit the maximum indirect power, i.e.,

maximum stress on the devices thus resembles the worst case scenario [38]. The thermal

model of the converters is neglected provided that the heat sink is adequate enough to

maintain the semiconductors proper working. Power wasted in the power supplies for the

control of the converters is also ignored (It may be between 10–20W). The analytical

calculation is based on the EUPEC IGBT module FP15R12W1T4_B3 [39].

Fig. 3 shows the power generation by the wind turbine based on the modeling equations

described in Section 5 and operating from a wind speed of 2 m/s (cut-in) to 17 m/s (cut-out),

while the rated wind speed is 13 m/s. The furling control has been established through the

furling model (6) and effectively reduces the power above rated wind speed. This power

serves as the dynamic power reference for all power electronic converters and summed up to

calculate the total power generation by the WT for the considered wind speed regime.

The conduction and switching losses for the diodes and IGBTs of the IBC, which composed of

a 3-phase diode bridge rectifier, a boost converter and a PWM inverter is presented in Fig. 4a–c

respectively for a similar wind variation. In an analogous fashion, power losses for the IBBC, BBC

and MC are presented through Fig. 5, Fig. 6 and Fig. 7 respectively. The total loss of each

converter is shown in Fig. 8 and can be found by summing up all the losses (conduction and

switching) for each converters. The results of the power losses for the converters show that the

power loss is higher for a low wind speed (12m/s) than for the rated wind speed (13 m/s) except

for the MC and are due to the furling action. The furling angle varies abruptly from 9 m/s to 13

m/s with negligible change close to the maximum speed. Meanwhile, the voltage remains linear

with the wind speed. As a result, the captured aerodynamic power as well as the current are

asymmetrical on either side of the rated wind speed and are reflected on the power loss curve.

P P P P PtIBBC

c dMC

c iMC

s dMC

s iMC= + + +( )3 , , , ,

P f E EV

V

I

Is i

MCsw ON OFF

aMC

r d

amIBBC

r d,

, ,

. .= +( ) 122π

P f EV

V

I

Is d

MCsw SR

aMC

r d

amIBBC

r d,

, ,

. .= 122π

P V I r Ic iMC

ce omMC

ce omMC

, = + ( )2 20

2

π



The power generation by the wind turbine and total power loss of the converters within

the considered wind speed regime is shown in Fig. 9. It is obvious that the IBC resulted lower

power losses compared to any other converters. The IBBC exhibits a small amount of higher

losses than the IBC. This is because to achieve the same voltage level, a boost converter

requires less duty cycle than a buck-boost converter and can be seen from Fig. 10. This

implies that due to a higher duty cycle, the conduction losses of the buck-boost converter

will eventually increase and reflect on the total loss calculation for the IBBC. However,

compared to the IBC and IBBC, the BBC and MC have higher power losses due to the

involvement of higher number of semiconductor devices. Furthermore, higher values of

internal resistance of the IGBTs increase the conduction losses and so as the total power

losses of the BBC and MC. In addition, on-state and off-state switching loss energies of the


Figure 3: Variation of power generation with wind speed by the wind turbine.

Figure 4: Variation of power losses with wind speed of the IBC.

IGBTs associated with the BBC and MC increase switching losses, while the off-state

switching loss energy of a diode is lower and the IBC or IBBC adapts a 3-phase bridge

rectifier at the input for the AC–DC conversion. The efficiency calculation of the converters

reveal that the IBC maintains an efficiency of 86%, while the MC has the lowest efficiency of

80%. The IBBC has an efficiency of 83% which is lower in value than the IBC and is due to the

higher conduction losses involve with the buck-boost converter. Compared to the boost

converter of the IBC, IBBC requires higher voltage and current rating components for the

buck-boost converter of the IBBC. The BBC maintains efficiency of 81%, and finally as a

whole, the IBC has received the highest efficiency than any other converters under

consideration. However, before selecting a particular converter, several other issues should

also needs to be taken into consideration. The total number of semiconductor devices



Figure 5: Variation of power losses with wind speed of the IBC.

Figure 6: Variation of power losses with wind speed of the BBC.

associated with the BBC and MC are more than the IBC and IBBC. This signifies that the cost

of the IBC and IBBC is lower than the BBC and MC as more diodes and IGBTs are involved

with these two converters. Furthermore, typically IGBTs are 6.6 times more expensive than

diodes and also supports the cost assumptions. The input voltage phase and amplitude of the

BBC is independent of the generator emf phase and amplitude. This facilitates the reverse

power flow characteristic of the BBC by altering the phase of the generated voltage. The

reverse power flow then can be used to start the wind turbine rotor thus could enhance the

energy generation by a WECS. This bidirectional power flow characteristic is absent in the

IBC and IBBC due to the involvement of rectifier at the input which composed of

unidirectional diodes. On the other hand, inclusion of a 3-phase bridge rectifier with the IBC

and IBBC provide unity power factor capability without using any extra control circuitry,


Figure 7: Variation of power losses with wind speed of the MC.

Figure 8: Variation of power losses with wind speed for the converters.

which is an essential part of the BBC and MC and thus greatly reduce the control complexity.

Moreover, In contrast to other power electronic converters, the MC might not be a suitable

choice. This is due to the fact that increase number of semiconductor devices will not only

decrease the efficiency but also increase the cost of the system. Nonetheless, using reverse

blocking IGBT instead of conventional back-to-back bidirectional IGBTs could improve the

efficiency of the MC. However, in either way, control complexity will greatly increase and

might make the system less preferable. A qualitative comparison is presented in Table 1 for

the converters by considering the efficiency, device count, cost, power flow direction, and

control complexity, while the IBC serves as a standard. It is notable that the IBC not only

maintains the highest efficiency, but also occupies a higher position in terms of the cost and



Figure 9: Variation of total power losses for the converters at 20 kHz.

Figure 10: Variation of power losses of the BBC.

control complexity, however, exhibits unidirectional power flow characteristic. In

conclusion, it is concluded that even though the BBC and MC provides the reverse power

flow characteristic, IBC still holds a better position and could be a suitable choice for high

penetration of wind power.

The emphasis is then given to clarify the converter performances under the influence of

switching frequency. This is of particular importance because the losses of power

electronic converters typically increase with an increase in switching frequency and

subsequently increase the power loss leading to a decrease in the efficiency and vice versa.

For a low switching frequency, however, the ripple in the current is substantial, and there

could be a considerable difference in turn-on and turn-off losses. A trade-off should be

made between the operating point of the switching frequency and the desired

performance of the system.

Fig. 11 and Fig. 12 present the variation of the total power losses with wind speeds of the

converters for a 10 KHz and 30 KHz switching frequency. It is clear that high switching

frequency produce more losses to the converters and vice versa. Corresponding to these

power losses, Fig. 13 and Fig. 14 presents the total power losses for the converters from cut-in to

cut-out wind speed. In either graphs, the IBC still maintains the lowest losses compared to any

other converters. Afterwards, efficiency is calculated and Table 2 presents the variation of the

efficiency with the variation in switching frequency with the IBC@20kHz as a standard. It is


Figure 11: Variation of power losses with wind speed for the converters at 10 kHz

Table 1 Comparison of power electronic converters

Power ElectronicConcerters IBC IBBC BBC MC

Efficiency 86% 83% 81% 80%Device count Diode-7 Diode-7 Diode-12 Diode-18

IGBT-7 IGBT-7 IGBT-12 IGBT-18Cost Low High Higher Highest

Power flow direction Unidirectional Unidirectional Bidirectional BidirectionalControl complexity Low Low High Highest

obvious that the IBC provides a higher efficiency for both 10 kHz and 30 kHz operation.

Furthermore, the efficiency (91%) at 10 kHz switching frequency is 5% more than the

efficiency (86%) that occurred at 20 kHz. On the other hand, the efficiency (80%) at 30 kHz

switching frequency is about 6% less than the efficiency that occurred at 20 kHz. The same




Figure 12: Variation of power losses with wind speed for the converters at 30 kHz.

Table 2 Efficiency variation of power electronic converters with switching frequency

Efficiency

Power Electronic Concerters IBC IBBC BBC MC

10 kHz 91% 88% 87% 84%20 kHz 86% 83% 81% 80%30 kHz 80% 78% 74% 73%

conclusion can be drawn for all the other converters in consideration. This leads to an

imperative observation that operating the system at a lower (higher) switching frequency

either than 20 kHz, will increase (decrease) the power loss without a significant change in the

efficiency. But decrease in switching frequency augments the cost due to the installation of a

filter circuit at the output of the PCS and commence lower order harmonics to the system. In

contrast to the lower switching frequency (10 kHz), a higher switching frequency (30 kHz)

will introduce higher order harmonics and can be easily filtered out using less costly filter

circuit with an expanse of low efficiency. In conclusion, lower switching frequency leads

higher efficiency of the converters with an introduction of lower order harmonics and high

cost, while the opposite picture can be portrayed for the higher switching frequency, however,

in either scenario, there is no significant change is observed in the system efficiency. As a

whole, a 20 kHz switching frequency is a reasonable one to consider and the IBC is an

optimum choice for further development of grid-connected wind energy conversion system.

In conclusion, the scope of research for a permanent magnet generator based grid-connected

wind energy conversion system could be aimed for an IBC as a power electronic converter.

This converter could be an optimum choice considering low power losses, high efficiency as

well as less complex architecture and consequently more reliable and less costly during the

operation.

8. CONCLUSIONSIn this paper an efficiency comparison of intermediate boost converter, intermediate buck-

boost converter, back-to-back converter and matrix converter converters has been carried

out taking into account the semiconductor devices conduction and switching losses. The same

wind turbine, generator, IGBTs and diodes have been used for the comparison. The power

generation of a wind turbine and power losses of the power electronic converters are

calculated with the variation in wind speed. It has been found that an intermediate boost

converter is more efficient than any other converters, however, lacks the characteristic of

bidirectional power flow. Nonetheless, considering the device count, cost and control

complexity, and intermediate boost converter could be an optimum choice for connecting a

wind energy conversion system with the grid.



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