Sensors and Actuators A 214 (2014) 25– 30

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High-Q and miniaturized complementary split ring resonator-loaded substrate integrated waveguide microwave sensor for crack

detectionin metallic materialsTaehwa Yun, Sungjoon Lim ∗School of Electrical and Electronics Engineering, Chung-Ang University, Seoul, Republic of Korea

a r t i c l e i n f o a b s t r a c tArticle history:Received 20 November 2013Received in revised form 1 April 2014Accepted 2 April 2014Available online 13 April 2014

Keywords:Nondestructive testing (NDT)Substrate integrated waveguide (SIW) Complementary split ring resonator (CSRR) Microwave sensorCrack detection

In this paper, a high-Q and miniaturized complementary split ring resonator (CSRR)-loaded substrateintegrated waveguide (SIW) microwave sensor for the detection of cracks in metallic materials is pre-sented. A SIW and CSRR with a high Q are used for achieving high sensitivity. In addition, by loading theCSRR on the SIW, the sensor is miniaturized and can be operated below the cut-off frequency of the SIW.The proposed sensor exhibits a frequency shift of 220 MHz for a 100- m crack and 600 MHz for a 1-mmcrack.

© 2014 Elsevier B.V. All rights reserved.

1. Introduction

Fatigue and corrosion of metal materials endanger mechani- cal systems such as aircraft fuselages, energy production plants, and steel bridges. Such fatigue and corrosion can cause sub- millimeter-size cracks or pits. These cracks are difficult to detect because of the size and coatings such as paint, composite lami- nates, and protective substances on the surfaces of the metallic materials. Several nondestructive testing (NDT) techniques have been developed to detect the cracks in metal materials. Most of the NDT solutions are based on eddy currents [1] and ultrasound[2]. Over the last few years, many studies on crack detection have been based on microwave and millimeter-wave technologies[3–7].

In Ref. [3], a near-field open-ended waveguide was developed

on basis of the perturbation in the surface current. When a crackexists in a metallic surface, the surface current flowing insidethe aperture of the waveguide is perturbed. The perturbed sur-face current leads to a variation in the reflection coefficient andrenders crack detection possible. The open-ended waveguide sen-

sor can detect cracks of less than 1mm. For crack detection, thelength of the crack should be greater than that of the long section

∗ Corresponding author. Tel.: +82 28205827.E-mail address: sungjoon@cau.ac.kr (S. Lim).

of the waveguide, and expensive measurement equipment isrequired because it is operated at a high frequency (24 GHz).Despite operating at a high frequency, the variations in the mag-nitude and phase of the reflection coefficient owing to the crackare very small. For enhancing the resolution and sensitivity, adielectric and dielectric slab-loaded waveguide are used as in Ref.[6], and the variations in the reflection coefficient are still verylow, while the operating frequencies are still high (greater than20 GHz).

In Ref. [7], a sensor for detecting microcracks in a metal surface

using a dual-behavior resonator filter was suggested. This sensordetects cracks in a metallic surface by the variation in the resonantfrequency and is operated at 10 GHz. Thus, expensivemeasurementequipment is required. The variations in the resonant

frequencyand quality (Q) factor from the existence of cracks are 50 MHz and45, respectively. These values enable the sensor to detect cracks inmetallic materials. However, the sensitivity and resolution of thissensor are low.

In this paper, we present a high-Q and miniaturized sensor for

crack detection in metal materials using a substrate integratedwaveguide (SIW) cavity and a complementary split ring resonator(CSRR). This sensor exhibits high sensitivity achieved by the high-Qcharacteristicof the CSRR and isminiaturizedusing theevanescent-wave amplification characteristics of the CSRR [8–10]. Further, thesensor operates a low frequency (∼5 GHz) while increasing the sen-sitivity and resolution.

http://dx.doi.org/10.1016/j.sna.2014.04.0060924-4247/© 2014 Elsevier B.V. All rights reserved.

26 T. Yun, S. Lim / Sensors and Actuators A 214 (2014) 25–30

V W

V L

Fig. 1. Planar view of the CSRR-loaded SIW: (a) top view, (b) bottom view, (c) CSRR, and (d) lossless equivalent circuit model of the CSRR. L = 11.2mm (0.191 o), W = 12mm(0.205 o), D = 1 mm, Vp = 1.8 mm, IW = 2.4 mm, IL = 1.8 mm, a1 = a2 = 3.3 mm, a3 = a5 = 0.2 mm, a4 = 0.5 mm, and g = 0.5 mm.

2. Sensor design

For replacing the conventional waveguide that is difficult to fab- ricate and has high cost, we use an SIW cavity resonator with a high Q factor. The SIW, initially reported in Ref. [11], has not only the advantage of low cost but is also convenient to combine with planar structures while maintaining the advantages of the conven- tional waveguide cavity (high Q factor, high power capacity). The SIW is composed of two parallel rows of metal vias embedded in the substrate, as shown Fig. 1(a). The SIW cavity has design param- eters of the center-to-center width and length (Wc and Lc) between the rows of vias, the diameter (D) of the vias, and the spacing (Vp) between the vias.

The resonant frequency of the SIW cavity is determined by the effective width and length [12]:

For high sensitivity and resolution, we combine a CSRR and the

SIW cavity. The CSRR reported in Ref. [14] consists of two rectangu-lar slots, a metal strip between the slots, and a conducting island,as shown in Fig. 1(c). When a voltage difference is excited in theCSRR, a capacitance (Cr) is induced in the rectangular slot by theE-field, and an inductance (Lr) is induced in the metal strip andconducting island by the H-field. Lr and Cr are modeled as a high-Q shunt-connected LC resonator tank [15]. The equivalent circuitmodel of the CSRR is illustrated in Fig. 1(d). According to the theoryof evanescent-wave amplification [8], this LC tank resonates at alower frequency than in the dominant mode of the SIW cavity. Thischaracteristic renders the sensor compact. In addition, when theSIW is combined with a CSRR, the CSRR mode and SIW mode aregenerated. At the CSRR mode, the Q value achieved is higher thanthat of the SIW without the CSRR. Using this feature of the CSRRmode, the resolution and sensitivity of the sensor are improved.

1fmn0 = 2√ ε

m

2Weff

n

2+

Leff(1) We have designed the CSRR to operate at approximately

5 GHz. This particular choice is to avoid the use of expensive measurement equipment. For the design of the CSRR, the LC resonator tank inwhere m, n, Weff, ε, and Leff are the mode indices,

permittivity,permeability and the effective width and length of the SIW cavity,respectively. The effective width and length of the SIW cavity canbe approximated as follows [12]:

Fig. 1(d) is used. When the loss is neglected, its resonant frequency is determined by

1

D2Weff = Wc − 1.08

p

D2+ 0.1

c(2)

fCSRR =2 Lr Cr

(7)

D2Leff = Lc − 1.08

p

D2+ 0.1

c(3)

Cr is calculated using the approximate formula corresponding

to a metallic disk surrounded by a ground plane, while Lr can becalculated using the duality of the SRR model corresponding

T. Yun, S. Lim / Sensors and Actuators A 214 (2014) 25–30 27

to the capacit an ce of the SRR model [15]. We determine the dimensions ofThe radiation losses due to the EM field leakage inside the SIW

occur when the spacing between the vias is too large because theSIW has rows of metal vias in the substrate. Additionally, the viadiameter D affects the return loss. For minimizing these losses,two design rules related to the diameter D and via spacing Vp areformulated as [13]

gD < (4)5Vp ≤ 2D (5)where g is the guided wavelength.

The SIW cavity is designed on the basis of a Rogers Duroid 6010

substrate with a thickness of 1.27 mm, a relative permittivity of10.2, and a loss tangent of 0.0023. The dimensions of the SIW cavityare determined to maintain the dominant resonant frequency of7.3 GHz from Eqs. (1)–(5) and optimized using ANSYS HFSS. Weuse gold vias, and the SIW cavity is excited by a 50- microstriptransmission line with a line width of 1.05 mm. The inset feedingmethod is used for impedance matching.

theouter slotof the CSRR as anelectric dipole [15] and otherdimen- sions of the CSRR are optimized using ANSYS HFSS to promptly obtain the resonant frequency of 5.12 GHz. The CSRR is embedded in the bottom of the SIW for avoiding a stripline-to-SIW transition.

When the CSRR is positioned close to the surface of the metallic material, the E-field and H-field are confined such that the res- onant frequency of the CSRR is changed by the variation in the effective capacitance. When the aluminum sample is loaded on the proposed sensor, the impedance and resonant frequencies are changed. In order to minimize the impedance mismatching, a top surface of the aluminum is coated with Teflon. A thicker Teflon coating can provide stable impedance matching while the sensitiv- ity is degraded. Teflon with 0.1mm thickness is chosen because it is the minimum realizable thickness while keeping stable impedance matching.

The simulated reflection coefficients of the SIW cavity with and without a CSRR are plotted in Fig. 2. The width and length of the SIW cavity is chosen to have the dominant resonant frequency of7.29 GHz. Fig. 3(a) and (c) shows the magnitude distributions of

Fig. 2. Simulated reflection coefficients of the SIW cavity (without a CSRR) andCSRR-loaded SIW cavity.

the electric field inside the SIW cavity and the CSRR-loaded SIW cavity at their dominant TE101 modes. For EM simulation, ANSYS high frequency structure simulator (HFSS) is used. When the CSRRis loaded on the SIW cavity, the impedance is changed. Therefore, the feeding lines are different for the impedance matching of both the SIW cavity without the CSRR (Fig. 3(a)) and the SIW cavity with the CSRR (Fig. 3(b) and (c)). However, the SIW cavity size is iden- tical for both SIW cavities with and without the CSRR. Although the size of two cavities is identical, their TE101 resonant frequen- cies are slightly different because of the CSRR perturbation on the CSRR-loaded SIW cavity. The electric coupling from the CSRR on the SIW surface decreases the TE101 resonant frequency of the SIW cav- ity. Therefore, Fig. 3(a) and (c) is plotted at 7.29 GHz and 6.83 GHz, respectively. On the other hand, when the CSRR is loaded on the SIW cavity, the additional resonance occurs at 5.12 GHz which is generated from the CSRR mode as shown in Fig. 3(b). When the width of the SIW is smaller, the resonant frequency at CSRR modeis affected because of the coupling between the SIW and CSRR. In addition, the radius of vias affects the impedance matching at the CSRR mode while via spacing shows less effects on the impedance of the CSRR mode.

3. Simulation and experimental results

Fig. 4 illustrates the measurement environment to sense the crack of the aluminum. The proposed microwave sensor is placed on the aluminum and the CSRR is patterned on the bottom of the

sensor. As mentioned earlier, the surfaces of the metallic struc-tures are covered with coatings. Therefore, a thin Teflon film with athickness of 0.1 mm is used to model this coating on an aluminumplate with a 5-mm thickness, as shown in Fig. 4. For crack model-ing, cracks only with 1-, 1.2-, and 2-mm widths and 1- and 2-mmdepths are fabricated in addition to a faultless plate. The frequencyresponses from the cracks with 0.1-mm and 0.2-mm widths areobserved only in the EM simulations because of the difficulty infabricating these cracks of 0.1-mm and 0.2-mm widths. The pictureof the fabricated sensor is shown in Fig. 5(a). For crack detection inmetallic materials, the sensor is scanned along the surfaces of themetallic materials. For determining the existence of cracks, a fault-less aluminum sample is set as a reference, as shown in Fig. 5(b),and the variation in the resonant frequency because of the crack isrecorded using an HP 8510C vector network analyzer.

As shown in Fig. 6, the variations in the reflection coefficient

are presented when the sensor is positioned on the cracks withvarious widths, and the depth of the crack is maintained at 1 mm.The shifts in the resonant frequency due to the crack widths of0.1mm and 2 mm are 220MHz and 980 MHz, respectively. As thecrack width increases, the resonant frequency tends to decrease.As shown in Fig. 7, the variations in the reflection coefficient arepresented when the sensor is positioned on the cracks with vari-ous widths, and the depth of the crack is maintained at 2 mm. Theshifts in the resonant frequency due to the crack widths of 0.1mmand 2 mm are 280MHz and 1080 MHz, respectively. It is observedthat the resonant frequency tends to decrease as the crack widthincreases. In addition, the resonant frequency tends to increase asthe crack depth increases. Therefore, the resonant frequency of theproposed sensor is dependent on both the crack width and depth.When the crack widths are equal to 0.1 mm, the difference of theresonant frequencies between the crack depths of 1mm and 2 mmis 60MHz. When the crack widths are equal to 2 mm, the differenceof the resonant frequencies between the crack depths of 1 mm and

2 mm is 170 MHz. The simulated and measured resonant frequen-cies for different widths and depths are summarized in Table 1. Asexpected, the resonant frequency tends to decrease when the crackdepth increases. The Q factor, which determines the resolution, isgreater than 148, as summarized in Table 1.

To confirm the effect of Teflon thickness, the simulated reflec-

tion coefficients are plotted at different thickness of Teflon(t= 0.1 mm, 0.3 mm, 0.5 mm, 0.7 mm, and 0.9 mm) in Fig. 8(a).

The resonant frequencies of the faultless aluminum are decreasedas the thickness of Teflon coating increases. For instance, theresonant frequencies are 5.12 GHz, 4.79 GHz, 4.67 GHz, 4.61 GHz,and 4.58 GHz at t = 0.1 mm, 0.3 mm, 0.5 mm, 0.7 mm, and 0.9 mm,respectively. In addition, these frequencies are compared withthe resonant frequencies of the fault aluminum (0.1 mm widthand 1 mm depth crack). Frequency difference between the fault-

Fig. 3. Magnitude of the E-field: (a) inside the SIW cavity at 7.29 GHz, (b) bottom of the CSRR-loaded SIW at 5.12 GHz, and inside the CSRR-loaded SIW at 6.83 GHz.

Fig. 4. Measurement environment: (a) side view and (b) cross-sectional view.

Fig. 5. Fabricated sensor and aluminum sample: (a) top and bottom views of the sensor and (b) aluminum sample.

Fig. 6. Simulated and measured results of the reflection coefficient for a crack depth of 1 mm.

Fig. 7. Simulated and measured results of the reflection coefficient for a crack depth of 2 mm.

Table 1Variations in the simulated and measured resonant frequencies and Q factors of the sensor for different widths and depths.

Faultless Depth: 1mm Depth: 2mmWidth [mm] 0 0.1 0.2 1 1.2 2 0.1 0.2 1 1.2 2Resonant frequency [GHz] (simulation) 5.12 4.90 4.72 4.52 4.38 4.22 4.84 4.61 4.43 4.24 4.05Resonant frequency [GHz] (measurement)

5.12 N/A N/A 4.49 4.39 4.21 N/A N/A 4.44 4.24 4.04Q-factor 256 245 157 224 219 210 242 153 148 212 202

Fig. 8. (a) Simulated reflection coefficients at different thickness of Teflon, (b) the simulated resonant frequencies at different roughness levels on the top of the Teflonsurface, and (c) the simulated reflection coefficients at 50- m, 100- m, 155- m, roughness levels on the top of the Teflon surface for the faultless and cracked (0.1-mm width and 1-mm depth) aluminum.

less and fault aluminum is 220 MHz, 50 MHz, 20MHz, 10 MHz,and 10MHz at t = 0.1 mm, 0.3 mm, 0.5 mm, 0.7 mm, and 0.9 mm,respectively. It cannot be observed the frequency shift after 0.9 mmthickness.

The frequency shifting is observed at different roughness of the

Teflon surface from full wave simulation. For the roughness setting,Hammerstand roughness model is used [16]. As shown in Fig. 8(b),the difference of the resonant frequencies with and without thecrack of 0.1-mm width and 1-mm depth is clear, although

theresonant frequencies are slightly increased with larger roughnesslevel. Fig. 8(c) shows the simulated reflection coefficientsat50- m,100- m, and 155- m roughness levels on the top of the Teflon

fw

reflectioncoefficientsat100- mroughness are less than−10 dBforboth faultless and cracked aluminum. However, when the rough-ness is 155 m, the reflection coefficients of the faultless andcracked aluminum are −6.3 dB at 5.47 GHz and −7 dB at 5.4 GHz,respectively. Although their resonant frequencies are different, the

sensitivity is poor at 155- m roughness level.Fig. 9 shows the relationship between the resonant frequency

and crack width at 1-mm and 2-mm depth of crack. The widthsensitivity (GHz/m) can be defined from Fig. 9 as the ratio of thefrequency shift to width difference at a fixed depth.

surface. As the roughness level is larger, the sensitivity becomes fWidth sensitivity (GHz/m) =

2 − fw1 (8)lower because of impedance mismatching and lower Q-factor. The

w

= w2 − w1

30 T. Yun, S. Lim / Sensors and Actuators A 214 (2014) 25–30

Fig. 9. Relationship between the resonant frequency and crack width at 1-mm and2-mm depth of crack.

When the depth of crack is 1 mm, the maximum and minimum width sensitivities are 220 GHz/m and 2.25 GHz/m, respectively. When the depth of crack is 2mm, the maximum and minimum width sensitivities are 280 GHz/m and 2.5 GHz/m. At both 1mm and 2mm depth, the maximum and minimum values are observed between faultless and 0.1 mm of crack width and between 1.2mm and 2 mm of crack width, respectively. Therefore, the width sensi- tivity decreases with wider crack because electromagnetic energy attenuates.

4. Conclusion

A high-Q and miniaturized CSRR-loaded SIW microwave sensor has beenpresented for crackdetection inmetallicmaterials.An SIW cavity and a CSRR have been used to detect the cracks in the metal- lic materials covered by dielectric layers. This sensor is operated at approximately 5 GHz, and the variation in the resonant frequency due to a crack of 1-mm width and 1-mm depth is 630 MHz. In addi- tion, the sensor exhibits a wide range of resonant frequency shifts and a high Q factor greater than 148. These properties increase the sensitivity and resolution of the sensor. When comparedwithother microwave sensors, this sensor operates at a lower frequency and exhibits higher sensitivity and resolution.

Acknowledgement

This research was supported by Basic Science Research Pro- gram through the National Research Foundation of Korea (NRF) funded by the Ministry of Education, Science and Technology (2012R1A1A2040160).

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Biographies

Taehwa Yun was born in Seoul, Korea, in 1984. He received the B.S. degree in electrical and electronics from Chung-Ang University, Seoul, Korea, in 2013. Currently, he is working toward the M.S. degree in electrical and elec- tronics engineering, Chung-Ang University. His research interests are focused on electromagnetic sensors, antenna arrays, and microwave circuit design.

Sungjoon Lim received the B.S. degree in electronic engi- neering from Yonsei University, Seoul, Korea, in 2002, and the M.S. and Ph.D. degrees in electrical engineering from the University of California at Los Angeles (UCLA), in 2004 and 2006, respectively. After

a postdoctoral position at the Integrated Nanosystem Research Facility (INRF), the University of California at Irvine, he joined the School of Electrical and Electronics Engineering, Chung-Ang Uni- versity, Seoul, Korea, in 2007, where he is currently an associate professor. He hasauthoredand coauthoredmore than 80 technical conference, letter and journal papers. His research interests include engineered electromag- netic structures (metamaterials, electromagneticbandgap

materials, and frequency selective surfaces), printed antennas, substrate integrated waveguide (SIW) components, inkjet-printedelectronicsand RF MEMS applications. He is also interested in the modeling and design of microwave circuits and systems. Dr.Lim received the InstitutionofEngineeringand Technology (IET)PremiumAward in 2009.