lumped components a thesis
TRANSCRIPT
FDTD MODELING OF RF AND MICROWAVE
CIRCUITS WITH ACTIVE AND
LUMPED COMPONENTS
by
BHARATHA YAJAMAN BE
A THESIS
IN
ELECTRICAL ENGINEERING
Submitted to the Graduate Faculty of Texas Tech University in
Partial Fulfillment of the Requirements for
the Degree of
MASTER OF SCIENCE
IN
ELECTRICAL ENGINEERING
Approved
Chairp^r^noftfe^ommittee
Accepted
Dean of the Graduate School
August 2004
ACKNOWLEDGEMENTS
I would like to thank everybody who has helped me during my graduate school
years and while I was working on my thesis
I would like to express my sincere gratitude to Dr Mohammad Saed my thesis
advisor for his support and guidance during my graduate studies research work and
thesis preparation This thesis would not have been possible without his continuous help
I would also like to express my sincere appreciation to Dr Jon G Bredeson
for serving on my thesis committee I would like to thank Sharath and Vijay for helping
me out in programming on quite a few occasions
Most of all I would like to thank my parents who have always stood by me and
made it possible for me to pmsue graduate studies My brother has also been a constant
source of motivation
I would like to thank my good friends Anu Dipen Nikhil Sharath Rohini Hari
Kanth Kiran and Swapnil who have constantly supported and encouraged me during my
Masters
To all of you I thank you
11
TABLE OF CONTENTS
ACKNOWLEDGEMENTS u
ABSTRACT vi
LIST OF FIGURES vii
CHAPTER
L INTRODUCTION 1
11 Background 1
12 Scope of the Thesis 3
13 Document Organization 4
IL FDTD ALGORITHM 5
21 Introduction 5
22 Maxwells Equation and hiitial Value Problem 5
23 FDTD Formulation 10
24 Numerical Dispersion 20
25 Numerical Stability 21
26 Frequency Domain Analysis 21
27 Source 22
28 Implementation of Basic FDTD Algorithm 23
m BOUNDARY CONDITIONS 24
31 Inttoduction 24
32 Perfect Electric Conductor 25
33 Perfectly Magnetic Conductor 26
34 Perfectly Matched Layer 28
35 Perfectly Matched Layer Formulation 28
111
IV MODELING LUMPED COMPONENTS 33
41 Inttoduction 33
42 Linear Lumped Components 33
43 Incorporating 2 Port Networks with S-Parameters 37
V FDTD SIMULATION PROGRAM 39
51 Introduction 39
52 Building a Three Dimensional (3D) model 39
53 FDTD Simulation Engine - An Overview 40
54 Software flow 42
55 Important Functions in Simulation 43
VL SIMULATION RESULTS 44
61 Inttoduction 44
62 Microstrip Line 44
621 Modeling of the materials 45
622 Source 45
623 Botmdary Condition 46
624 Results 46
63 Line Fed Rectangular Microstrip Anterma 49
631 Modeling of Materials 51
632 Somce 51
633 Botmdary Conditions 51
634 Results 52
64 Microstrip Low-pass Filter 55
641 Somce 56
642 Boimdary Conditions 56
643 Results 57
65 Branch Line Coupler 59
651 Somce 60
IV
652 Botmdary Conditions 60
653 Results 61
66 Series RLC Circuit 62
661 Somce 63
662 Boundary Conditions 63
663 Results 63
67 Parallel RL Circuit 65
671 Source 66
672 Boundary Conditions 66
673 Results 66
VIL CONCLUSION 68
71 Thesis Highlights 68
72 Futtire Work 69
REFERENCES 70
APPENDIX 72
FORMULATION OF PERFECTLY MATCHED LAYER 73
ABSTRACT
The objective of this thesis is to develop a systematic framework for modeling
electromagnetic wave propagation m RF and Microwave circuits with active and lumped
components using S-parameters using the Finite-Difference Time-Domain (FDTD)
approach which was originally proposed by K S Yee in 1966 The mediod
approximates the differentiation operators of the Maxwell equations with finite
difference operators m time and space This mediod is suitable for finding the
approximate electric and magnetic field in a complex three dimensional stmcture in the
time domain The computer program developed in this thesis can be used to simulate
various microwave circuits This thesis provides the theoretical basis for the method
the details of the implementation on a computer and fmally the software itself
VI
LIST OF FIGURES
21 Discretization of the model into cubes and the position of field components on the grid 11
22 Renaming the indexes of E and H field components
cortesponding to Cube(ijk) 16
23 Basic flow for implementation of Yee FDTD scheme 23
31 FDTD Lattice terminated by PML slabs 25
32 PEC on top surface of Cube (i j k) 26
33 PMC Boundary Conditions 27
51 Major software block of FDTD simulation program 41
52 Flow of FDTD Program 42
61 Microstrip Line 44
62 Distribution of Ez(xyz) at 200 time steps 47
63 Distribution of Ez(xyz) at 400 time steps 47
64 Distribution of Ez (xyz) at 1000 time steps 48
65 Amplitude of the input pulse 48
66 Amplitude of the Fourier ttansform of the input pulse 48
67 Line-fed rectangular microstrip antenna 50
68 Distiibution of E2(xyz) at 200 time steps 52
69 Distribution of Ez(xyz) at 400 time steps 53
610 DisttibutionofEz(xyz) at 600 time steps 53
611 Return loss ofthe rectangular antenna 54
VI1
612 Amplitudes of the Fourier ttansforms of the input
pulse and the reflected waveform 55
613 Low-pass Filter Detail 55
614 Retum loss of Low-pass Filter 57
615 Insertion loss of Low-pass Filter 58
616 Branch Line Coupler Detail 59
617 Scattering Parameters of Branch Lme Coupler 61
618 Series RLC Circuit Detail 62
619 Amplitude ofthe input and transmitted pulse 64
620 Amplitude of the Fourier transform of the input
reflected and ttansmitted pulse 64
621 Parallel LC Circuit Detail 65
622 Amplitude ofthe input and ttansmitted pulse 67
623 Amplitude of the Fourier ttansform of the input reflected and ttansmitted pulse 67
Vlll
CHAPTER I
INTRODUCTION
11 Background
In 1966 Kane Yee (1966) presented what we now refer to as the Finite-Difference
Time-Domain (FDTD) method for modelmg electtomagnetic phenomenon The Yees
algorithm as it is usually called in the literatme is well known for its robustness and
versatility The method approximates the differentiation operators of the Maxwell
equations with finite-difference operators in time and space This method is suitable for
finding the approximate electric and magnetic fields in a complex three-dimensional
stmcture in the time domain Many researchers have contributed immensely to extend the
method to many areas of science and engineering (Taflove 1995 1998) Initially the
FDTD method is used for simulating electtomagnetic waves scattering from object and
radar cross section measurement (Taflove 1975 Kunz and Lee 1978 Umashankar and
Taflove 1982) In recent years there is a proliferation of focus in using the method to
simulate microwave circuits and printed circuit board (Zhang and Mei 1988 Sheen et al
1990 Sui et al 1992 Toland and Houshmand 1993 Piket-May et al 1994 Ciampolini et
al 1996 Kuo et al 1997 Emili et al 2000) hiitially Zhang and Mei (1988) and Sheen et
al (1990) concenttated on modeling planar circuits using FDTD method Later Sui et al
(1992) and Toland and Houshmand (1993) inttoduced the lumped-element FDTD
approach This enabled the incorporation of ideal lumped elements such as resistors
capacitors inductors and PN junctions Piket-May et al (1994) refined the method
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
ACKNOWLEDGEMENTS
I would like to thank everybody who has helped me during my graduate school
years and while I was working on my thesis
I would like to express my sincere gratitude to Dr Mohammad Saed my thesis
advisor for his support and guidance during my graduate studies research work and
thesis preparation This thesis would not have been possible without his continuous help
I would also like to express my sincere appreciation to Dr Jon G Bredeson
for serving on my thesis committee I would like to thank Sharath and Vijay for helping
me out in programming on quite a few occasions
Most of all I would like to thank my parents who have always stood by me and
made it possible for me to pmsue graduate studies My brother has also been a constant
source of motivation
I would like to thank my good friends Anu Dipen Nikhil Sharath Rohini Hari
Kanth Kiran and Swapnil who have constantly supported and encouraged me during my
Masters
To all of you I thank you
11
TABLE OF CONTENTS
ACKNOWLEDGEMENTS u
ABSTRACT vi
LIST OF FIGURES vii
CHAPTER
L INTRODUCTION 1
11 Background 1
12 Scope of the Thesis 3
13 Document Organization 4
IL FDTD ALGORITHM 5
21 Introduction 5
22 Maxwells Equation and hiitial Value Problem 5
23 FDTD Formulation 10
24 Numerical Dispersion 20
25 Numerical Stability 21
26 Frequency Domain Analysis 21
27 Source 22
28 Implementation of Basic FDTD Algorithm 23
m BOUNDARY CONDITIONS 24
31 Inttoduction 24
32 Perfect Electric Conductor 25
33 Perfectly Magnetic Conductor 26
34 Perfectly Matched Layer 28
35 Perfectly Matched Layer Formulation 28
111
IV MODELING LUMPED COMPONENTS 33
41 Inttoduction 33
42 Linear Lumped Components 33
43 Incorporating 2 Port Networks with S-Parameters 37
V FDTD SIMULATION PROGRAM 39
51 Introduction 39
52 Building a Three Dimensional (3D) model 39
53 FDTD Simulation Engine - An Overview 40
54 Software flow 42
55 Important Functions in Simulation 43
VL SIMULATION RESULTS 44
61 Inttoduction 44
62 Microstrip Line 44
621 Modeling of the materials 45
622 Source 45
623 Botmdary Condition 46
624 Results 46
63 Line Fed Rectangular Microstrip Anterma 49
631 Modeling of Materials 51
632 Somce 51
633 Botmdary Conditions 51
634 Results 52
64 Microstrip Low-pass Filter 55
641 Somce 56
642 Boimdary Conditions 56
643 Results 57
65 Branch Line Coupler 59
651 Somce 60
IV
652 Botmdary Conditions 60
653 Results 61
66 Series RLC Circuit 62
661 Somce 63
662 Boundary Conditions 63
663 Results 63
67 Parallel RL Circuit 65
671 Source 66
672 Boundary Conditions 66
673 Results 66
VIL CONCLUSION 68
71 Thesis Highlights 68
72 Futtire Work 69
REFERENCES 70
APPENDIX 72
FORMULATION OF PERFECTLY MATCHED LAYER 73
ABSTRACT
The objective of this thesis is to develop a systematic framework for modeling
electromagnetic wave propagation m RF and Microwave circuits with active and lumped
components using S-parameters using the Finite-Difference Time-Domain (FDTD)
approach which was originally proposed by K S Yee in 1966 The mediod
approximates the differentiation operators of the Maxwell equations with finite
difference operators m time and space This mediod is suitable for finding the
approximate electric and magnetic field in a complex three dimensional stmcture in the
time domain The computer program developed in this thesis can be used to simulate
various microwave circuits This thesis provides the theoretical basis for the method
the details of the implementation on a computer and fmally the software itself
VI
LIST OF FIGURES
21 Discretization of the model into cubes and the position of field components on the grid 11
22 Renaming the indexes of E and H field components
cortesponding to Cube(ijk) 16
23 Basic flow for implementation of Yee FDTD scheme 23
31 FDTD Lattice terminated by PML slabs 25
32 PEC on top surface of Cube (i j k) 26
33 PMC Boundary Conditions 27
51 Major software block of FDTD simulation program 41
52 Flow of FDTD Program 42
61 Microstrip Line 44
62 Distribution of Ez(xyz) at 200 time steps 47
63 Distribution of Ez(xyz) at 400 time steps 47
64 Distribution of Ez (xyz) at 1000 time steps 48
65 Amplitude of the input pulse 48
66 Amplitude of the Fourier ttansform of the input pulse 48
67 Line-fed rectangular microstrip antenna 50
68 Distiibution of E2(xyz) at 200 time steps 52
69 Distribution of Ez(xyz) at 400 time steps 53
610 DisttibutionofEz(xyz) at 600 time steps 53
611 Return loss ofthe rectangular antenna 54
VI1
612 Amplitudes of the Fourier ttansforms of the input
pulse and the reflected waveform 55
613 Low-pass Filter Detail 55
614 Retum loss of Low-pass Filter 57
615 Insertion loss of Low-pass Filter 58
616 Branch Line Coupler Detail 59
617 Scattering Parameters of Branch Lme Coupler 61
618 Series RLC Circuit Detail 62
619 Amplitude ofthe input and transmitted pulse 64
620 Amplitude of the Fourier transform of the input
reflected and ttansmitted pulse 64
621 Parallel LC Circuit Detail 65
622 Amplitude ofthe input and ttansmitted pulse 67
623 Amplitude of the Fourier ttansform of the input reflected and ttansmitted pulse 67
Vlll
CHAPTER I
INTRODUCTION
11 Background
In 1966 Kane Yee (1966) presented what we now refer to as the Finite-Difference
Time-Domain (FDTD) method for modelmg electtomagnetic phenomenon The Yees
algorithm as it is usually called in the literatme is well known for its robustness and
versatility The method approximates the differentiation operators of the Maxwell
equations with finite-difference operators in time and space This method is suitable for
finding the approximate electric and magnetic fields in a complex three-dimensional
stmcture in the time domain Many researchers have contributed immensely to extend the
method to many areas of science and engineering (Taflove 1995 1998) Initially the
FDTD method is used for simulating electtomagnetic waves scattering from object and
radar cross section measurement (Taflove 1975 Kunz and Lee 1978 Umashankar and
Taflove 1982) In recent years there is a proliferation of focus in using the method to
simulate microwave circuits and printed circuit board (Zhang and Mei 1988 Sheen et al
1990 Sui et al 1992 Toland and Houshmand 1993 Piket-May et al 1994 Ciampolini et
al 1996 Kuo et al 1997 Emili et al 2000) hiitially Zhang and Mei (1988) and Sheen et
al (1990) concenttated on modeling planar circuits using FDTD method Later Sui et al
(1992) and Toland and Houshmand (1993) inttoduced the lumped-element FDTD
approach This enabled the incorporation of ideal lumped elements such as resistors
capacitors inductors and PN junctions Piket-May et al (1994) refined the method
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
TABLE OF CONTENTS
ACKNOWLEDGEMENTS u
ABSTRACT vi
LIST OF FIGURES vii
CHAPTER
L INTRODUCTION 1
11 Background 1
12 Scope of the Thesis 3
13 Document Organization 4
IL FDTD ALGORITHM 5
21 Introduction 5
22 Maxwells Equation and hiitial Value Problem 5
23 FDTD Formulation 10
24 Numerical Dispersion 20
25 Numerical Stability 21
26 Frequency Domain Analysis 21
27 Source 22
28 Implementation of Basic FDTD Algorithm 23
m BOUNDARY CONDITIONS 24
31 Inttoduction 24
32 Perfect Electric Conductor 25
33 Perfectly Magnetic Conductor 26
34 Perfectly Matched Layer 28
35 Perfectly Matched Layer Formulation 28
111
IV MODELING LUMPED COMPONENTS 33
41 Inttoduction 33
42 Linear Lumped Components 33
43 Incorporating 2 Port Networks with S-Parameters 37
V FDTD SIMULATION PROGRAM 39
51 Introduction 39
52 Building a Three Dimensional (3D) model 39
53 FDTD Simulation Engine - An Overview 40
54 Software flow 42
55 Important Functions in Simulation 43
VL SIMULATION RESULTS 44
61 Inttoduction 44
62 Microstrip Line 44
621 Modeling of the materials 45
622 Source 45
623 Botmdary Condition 46
624 Results 46
63 Line Fed Rectangular Microstrip Anterma 49
631 Modeling of Materials 51
632 Somce 51
633 Botmdary Conditions 51
634 Results 52
64 Microstrip Low-pass Filter 55
641 Somce 56
642 Boimdary Conditions 56
643 Results 57
65 Branch Line Coupler 59
651 Somce 60
IV
652 Botmdary Conditions 60
653 Results 61
66 Series RLC Circuit 62
661 Somce 63
662 Boundary Conditions 63
663 Results 63
67 Parallel RL Circuit 65
671 Source 66
672 Boundary Conditions 66
673 Results 66
VIL CONCLUSION 68
71 Thesis Highlights 68
72 Futtire Work 69
REFERENCES 70
APPENDIX 72
FORMULATION OF PERFECTLY MATCHED LAYER 73
ABSTRACT
The objective of this thesis is to develop a systematic framework for modeling
electromagnetic wave propagation m RF and Microwave circuits with active and lumped
components using S-parameters using the Finite-Difference Time-Domain (FDTD)
approach which was originally proposed by K S Yee in 1966 The mediod
approximates the differentiation operators of the Maxwell equations with finite
difference operators m time and space This mediod is suitable for finding the
approximate electric and magnetic field in a complex three dimensional stmcture in the
time domain The computer program developed in this thesis can be used to simulate
various microwave circuits This thesis provides the theoretical basis for the method
the details of the implementation on a computer and fmally the software itself
VI
LIST OF FIGURES
21 Discretization of the model into cubes and the position of field components on the grid 11
22 Renaming the indexes of E and H field components
cortesponding to Cube(ijk) 16
23 Basic flow for implementation of Yee FDTD scheme 23
31 FDTD Lattice terminated by PML slabs 25
32 PEC on top surface of Cube (i j k) 26
33 PMC Boundary Conditions 27
51 Major software block of FDTD simulation program 41
52 Flow of FDTD Program 42
61 Microstrip Line 44
62 Distribution of Ez(xyz) at 200 time steps 47
63 Distribution of Ez(xyz) at 400 time steps 47
64 Distribution of Ez (xyz) at 1000 time steps 48
65 Amplitude of the input pulse 48
66 Amplitude of the Fourier ttansform of the input pulse 48
67 Line-fed rectangular microstrip antenna 50
68 Distiibution of E2(xyz) at 200 time steps 52
69 Distribution of Ez(xyz) at 400 time steps 53
610 DisttibutionofEz(xyz) at 600 time steps 53
611 Return loss ofthe rectangular antenna 54
VI1
612 Amplitudes of the Fourier ttansforms of the input
pulse and the reflected waveform 55
613 Low-pass Filter Detail 55
614 Retum loss of Low-pass Filter 57
615 Insertion loss of Low-pass Filter 58
616 Branch Line Coupler Detail 59
617 Scattering Parameters of Branch Lme Coupler 61
618 Series RLC Circuit Detail 62
619 Amplitude ofthe input and transmitted pulse 64
620 Amplitude of the Fourier transform of the input
reflected and ttansmitted pulse 64
621 Parallel LC Circuit Detail 65
622 Amplitude ofthe input and ttansmitted pulse 67
623 Amplitude of the Fourier ttansform of the input reflected and ttansmitted pulse 67
Vlll
CHAPTER I
INTRODUCTION
11 Background
In 1966 Kane Yee (1966) presented what we now refer to as the Finite-Difference
Time-Domain (FDTD) method for modelmg electtomagnetic phenomenon The Yees
algorithm as it is usually called in the literatme is well known for its robustness and
versatility The method approximates the differentiation operators of the Maxwell
equations with finite-difference operators in time and space This method is suitable for
finding the approximate electric and magnetic fields in a complex three-dimensional
stmcture in the time domain Many researchers have contributed immensely to extend the
method to many areas of science and engineering (Taflove 1995 1998) Initially the
FDTD method is used for simulating electtomagnetic waves scattering from object and
radar cross section measurement (Taflove 1975 Kunz and Lee 1978 Umashankar and
Taflove 1982) In recent years there is a proliferation of focus in using the method to
simulate microwave circuits and printed circuit board (Zhang and Mei 1988 Sheen et al
1990 Sui et al 1992 Toland and Houshmand 1993 Piket-May et al 1994 Ciampolini et
al 1996 Kuo et al 1997 Emili et al 2000) hiitially Zhang and Mei (1988) and Sheen et
al (1990) concenttated on modeling planar circuits using FDTD method Later Sui et al
(1992) and Toland and Houshmand (1993) inttoduced the lumped-element FDTD
approach This enabled the incorporation of ideal lumped elements such as resistors
capacitors inductors and PN junctions Piket-May et al (1994) refined the method
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
IV MODELING LUMPED COMPONENTS 33
41 Inttoduction 33
42 Linear Lumped Components 33
43 Incorporating 2 Port Networks with S-Parameters 37
V FDTD SIMULATION PROGRAM 39
51 Introduction 39
52 Building a Three Dimensional (3D) model 39
53 FDTD Simulation Engine - An Overview 40
54 Software flow 42
55 Important Functions in Simulation 43
VL SIMULATION RESULTS 44
61 Inttoduction 44
62 Microstrip Line 44
621 Modeling of the materials 45
622 Source 45
623 Botmdary Condition 46
624 Results 46
63 Line Fed Rectangular Microstrip Anterma 49
631 Modeling of Materials 51
632 Somce 51
633 Botmdary Conditions 51
634 Results 52
64 Microstrip Low-pass Filter 55
641 Somce 56
642 Boimdary Conditions 56
643 Results 57
65 Branch Line Coupler 59
651 Somce 60
IV
652 Botmdary Conditions 60
653 Results 61
66 Series RLC Circuit 62
661 Somce 63
662 Boundary Conditions 63
663 Results 63
67 Parallel RL Circuit 65
671 Source 66
672 Boundary Conditions 66
673 Results 66
VIL CONCLUSION 68
71 Thesis Highlights 68
72 Futtire Work 69
REFERENCES 70
APPENDIX 72
FORMULATION OF PERFECTLY MATCHED LAYER 73
ABSTRACT
The objective of this thesis is to develop a systematic framework for modeling
electromagnetic wave propagation m RF and Microwave circuits with active and lumped
components using S-parameters using the Finite-Difference Time-Domain (FDTD)
approach which was originally proposed by K S Yee in 1966 The mediod
approximates the differentiation operators of the Maxwell equations with finite
difference operators m time and space This mediod is suitable for finding the
approximate electric and magnetic field in a complex three dimensional stmcture in the
time domain The computer program developed in this thesis can be used to simulate
various microwave circuits This thesis provides the theoretical basis for the method
the details of the implementation on a computer and fmally the software itself
VI
LIST OF FIGURES
21 Discretization of the model into cubes and the position of field components on the grid 11
22 Renaming the indexes of E and H field components
cortesponding to Cube(ijk) 16
23 Basic flow for implementation of Yee FDTD scheme 23
31 FDTD Lattice terminated by PML slabs 25
32 PEC on top surface of Cube (i j k) 26
33 PMC Boundary Conditions 27
51 Major software block of FDTD simulation program 41
52 Flow of FDTD Program 42
61 Microstrip Line 44
62 Distribution of Ez(xyz) at 200 time steps 47
63 Distribution of Ez(xyz) at 400 time steps 47
64 Distribution of Ez (xyz) at 1000 time steps 48
65 Amplitude of the input pulse 48
66 Amplitude of the Fourier ttansform of the input pulse 48
67 Line-fed rectangular microstrip antenna 50
68 Distiibution of E2(xyz) at 200 time steps 52
69 Distribution of Ez(xyz) at 400 time steps 53
610 DisttibutionofEz(xyz) at 600 time steps 53
611 Return loss ofthe rectangular antenna 54
VI1
612 Amplitudes of the Fourier ttansforms of the input
pulse and the reflected waveform 55
613 Low-pass Filter Detail 55
614 Retum loss of Low-pass Filter 57
615 Insertion loss of Low-pass Filter 58
616 Branch Line Coupler Detail 59
617 Scattering Parameters of Branch Lme Coupler 61
618 Series RLC Circuit Detail 62
619 Amplitude ofthe input and transmitted pulse 64
620 Amplitude of the Fourier transform of the input
reflected and ttansmitted pulse 64
621 Parallel LC Circuit Detail 65
622 Amplitude ofthe input and ttansmitted pulse 67
623 Amplitude of the Fourier ttansform of the input reflected and ttansmitted pulse 67
Vlll
CHAPTER I
INTRODUCTION
11 Background
In 1966 Kane Yee (1966) presented what we now refer to as the Finite-Difference
Time-Domain (FDTD) method for modelmg electtomagnetic phenomenon The Yees
algorithm as it is usually called in the literatme is well known for its robustness and
versatility The method approximates the differentiation operators of the Maxwell
equations with finite-difference operators in time and space This method is suitable for
finding the approximate electric and magnetic fields in a complex three-dimensional
stmcture in the time domain Many researchers have contributed immensely to extend the
method to many areas of science and engineering (Taflove 1995 1998) Initially the
FDTD method is used for simulating electtomagnetic waves scattering from object and
radar cross section measurement (Taflove 1975 Kunz and Lee 1978 Umashankar and
Taflove 1982) In recent years there is a proliferation of focus in using the method to
simulate microwave circuits and printed circuit board (Zhang and Mei 1988 Sheen et al
1990 Sui et al 1992 Toland and Houshmand 1993 Piket-May et al 1994 Ciampolini et
al 1996 Kuo et al 1997 Emili et al 2000) hiitially Zhang and Mei (1988) and Sheen et
al (1990) concenttated on modeling planar circuits using FDTD method Later Sui et al
(1992) and Toland and Houshmand (1993) inttoduced the lumped-element FDTD
approach This enabled the incorporation of ideal lumped elements such as resistors
capacitors inductors and PN junctions Piket-May et al (1994) refined the method
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
652 Botmdary Conditions 60
653 Results 61
66 Series RLC Circuit 62
661 Somce 63
662 Boundary Conditions 63
663 Results 63
67 Parallel RL Circuit 65
671 Source 66
672 Boundary Conditions 66
673 Results 66
VIL CONCLUSION 68
71 Thesis Highlights 68
72 Futtire Work 69
REFERENCES 70
APPENDIX 72
FORMULATION OF PERFECTLY MATCHED LAYER 73
ABSTRACT
The objective of this thesis is to develop a systematic framework for modeling
electromagnetic wave propagation m RF and Microwave circuits with active and lumped
components using S-parameters using the Finite-Difference Time-Domain (FDTD)
approach which was originally proposed by K S Yee in 1966 The mediod
approximates the differentiation operators of the Maxwell equations with finite
difference operators m time and space This mediod is suitable for finding the
approximate electric and magnetic field in a complex three dimensional stmcture in the
time domain The computer program developed in this thesis can be used to simulate
various microwave circuits This thesis provides the theoretical basis for the method
the details of the implementation on a computer and fmally the software itself
VI
LIST OF FIGURES
21 Discretization of the model into cubes and the position of field components on the grid 11
22 Renaming the indexes of E and H field components
cortesponding to Cube(ijk) 16
23 Basic flow for implementation of Yee FDTD scheme 23
31 FDTD Lattice terminated by PML slabs 25
32 PEC on top surface of Cube (i j k) 26
33 PMC Boundary Conditions 27
51 Major software block of FDTD simulation program 41
52 Flow of FDTD Program 42
61 Microstrip Line 44
62 Distribution of Ez(xyz) at 200 time steps 47
63 Distribution of Ez(xyz) at 400 time steps 47
64 Distribution of Ez (xyz) at 1000 time steps 48
65 Amplitude of the input pulse 48
66 Amplitude of the Fourier ttansform of the input pulse 48
67 Line-fed rectangular microstrip antenna 50
68 Distiibution of E2(xyz) at 200 time steps 52
69 Distribution of Ez(xyz) at 400 time steps 53
610 DisttibutionofEz(xyz) at 600 time steps 53
611 Return loss ofthe rectangular antenna 54
VI1
612 Amplitudes of the Fourier ttansforms of the input
pulse and the reflected waveform 55
613 Low-pass Filter Detail 55
614 Retum loss of Low-pass Filter 57
615 Insertion loss of Low-pass Filter 58
616 Branch Line Coupler Detail 59
617 Scattering Parameters of Branch Lme Coupler 61
618 Series RLC Circuit Detail 62
619 Amplitude ofthe input and transmitted pulse 64
620 Amplitude of the Fourier transform of the input
reflected and ttansmitted pulse 64
621 Parallel LC Circuit Detail 65
622 Amplitude ofthe input and ttansmitted pulse 67
623 Amplitude of the Fourier ttansform of the input reflected and ttansmitted pulse 67
Vlll
CHAPTER I
INTRODUCTION
11 Background
In 1966 Kane Yee (1966) presented what we now refer to as the Finite-Difference
Time-Domain (FDTD) method for modelmg electtomagnetic phenomenon The Yees
algorithm as it is usually called in the literatme is well known for its robustness and
versatility The method approximates the differentiation operators of the Maxwell
equations with finite-difference operators in time and space This method is suitable for
finding the approximate electric and magnetic fields in a complex three-dimensional
stmcture in the time domain Many researchers have contributed immensely to extend the
method to many areas of science and engineering (Taflove 1995 1998) Initially the
FDTD method is used for simulating electtomagnetic waves scattering from object and
radar cross section measurement (Taflove 1975 Kunz and Lee 1978 Umashankar and
Taflove 1982) In recent years there is a proliferation of focus in using the method to
simulate microwave circuits and printed circuit board (Zhang and Mei 1988 Sheen et al
1990 Sui et al 1992 Toland and Houshmand 1993 Piket-May et al 1994 Ciampolini et
al 1996 Kuo et al 1997 Emili et al 2000) hiitially Zhang and Mei (1988) and Sheen et
al (1990) concenttated on modeling planar circuits using FDTD method Later Sui et al
(1992) and Toland and Houshmand (1993) inttoduced the lumped-element FDTD
approach This enabled the incorporation of ideal lumped elements such as resistors
capacitors inductors and PN junctions Piket-May et al (1994) refined the method
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
ABSTRACT
The objective of this thesis is to develop a systematic framework for modeling
electromagnetic wave propagation m RF and Microwave circuits with active and lumped
components using S-parameters using the Finite-Difference Time-Domain (FDTD)
approach which was originally proposed by K S Yee in 1966 The mediod
approximates the differentiation operators of the Maxwell equations with finite
difference operators m time and space This mediod is suitable for finding the
approximate electric and magnetic field in a complex three dimensional stmcture in the
time domain The computer program developed in this thesis can be used to simulate
various microwave circuits This thesis provides the theoretical basis for the method
the details of the implementation on a computer and fmally the software itself
VI
LIST OF FIGURES
21 Discretization of the model into cubes and the position of field components on the grid 11
22 Renaming the indexes of E and H field components
cortesponding to Cube(ijk) 16
23 Basic flow for implementation of Yee FDTD scheme 23
31 FDTD Lattice terminated by PML slabs 25
32 PEC on top surface of Cube (i j k) 26
33 PMC Boundary Conditions 27
51 Major software block of FDTD simulation program 41
52 Flow of FDTD Program 42
61 Microstrip Line 44
62 Distribution of Ez(xyz) at 200 time steps 47
63 Distribution of Ez(xyz) at 400 time steps 47
64 Distribution of Ez (xyz) at 1000 time steps 48
65 Amplitude of the input pulse 48
66 Amplitude of the Fourier ttansform of the input pulse 48
67 Line-fed rectangular microstrip antenna 50
68 Distiibution of E2(xyz) at 200 time steps 52
69 Distribution of Ez(xyz) at 400 time steps 53
610 DisttibutionofEz(xyz) at 600 time steps 53
611 Return loss ofthe rectangular antenna 54
VI1
612 Amplitudes of the Fourier ttansforms of the input
pulse and the reflected waveform 55
613 Low-pass Filter Detail 55
614 Retum loss of Low-pass Filter 57
615 Insertion loss of Low-pass Filter 58
616 Branch Line Coupler Detail 59
617 Scattering Parameters of Branch Lme Coupler 61
618 Series RLC Circuit Detail 62
619 Amplitude ofthe input and transmitted pulse 64
620 Amplitude of the Fourier transform of the input
reflected and ttansmitted pulse 64
621 Parallel LC Circuit Detail 65
622 Amplitude ofthe input and ttansmitted pulse 67
623 Amplitude of the Fourier ttansform of the input reflected and ttansmitted pulse 67
Vlll
CHAPTER I
INTRODUCTION
11 Background
In 1966 Kane Yee (1966) presented what we now refer to as the Finite-Difference
Time-Domain (FDTD) method for modelmg electtomagnetic phenomenon The Yees
algorithm as it is usually called in the literatme is well known for its robustness and
versatility The method approximates the differentiation operators of the Maxwell
equations with finite-difference operators in time and space This method is suitable for
finding the approximate electric and magnetic fields in a complex three-dimensional
stmcture in the time domain Many researchers have contributed immensely to extend the
method to many areas of science and engineering (Taflove 1995 1998) Initially the
FDTD method is used for simulating electtomagnetic waves scattering from object and
radar cross section measurement (Taflove 1975 Kunz and Lee 1978 Umashankar and
Taflove 1982) In recent years there is a proliferation of focus in using the method to
simulate microwave circuits and printed circuit board (Zhang and Mei 1988 Sheen et al
1990 Sui et al 1992 Toland and Houshmand 1993 Piket-May et al 1994 Ciampolini et
al 1996 Kuo et al 1997 Emili et al 2000) hiitially Zhang and Mei (1988) and Sheen et
al (1990) concenttated on modeling planar circuits using FDTD method Later Sui et al
(1992) and Toland and Houshmand (1993) inttoduced the lumped-element FDTD
approach This enabled the incorporation of ideal lumped elements such as resistors
capacitors inductors and PN junctions Piket-May et al (1994) refined the method
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
LIST OF FIGURES
21 Discretization of the model into cubes and the position of field components on the grid 11
22 Renaming the indexes of E and H field components
cortesponding to Cube(ijk) 16
23 Basic flow for implementation of Yee FDTD scheme 23
31 FDTD Lattice terminated by PML slabs 25
32 PEC on top surface of Cube (i j k) 26
33 PMC Boundary Conditions 27
51 Major software block of FDTD simulation program 41
52 Flow of FDTD Program 42
61 Microstrip Line 44
62 Distribution of Ez(xyz) at 200 time steps 47
63 Distribution of Ez(xyz) at 400 time steps 47
64 Distribution of Ez (xyz) at 1000 time steps 48
65 Amplitude of the input pulse 48
66 Amplitude of the Fourier ttansform of the input pulse 48
67 Line-fed rectangular microstrip antenna 50
68 Distiibution of E2(xyz) at 200 time steps 52
69 Distribution of Ez(xyz) at 400 time steps 53
610 DisttibutionofEz(xyz) at 600 time steps 53
611 Return loss ofthe rectangular antenna 54
VI1
612 Amplitudes of the Fourier ttansforms of the input
pulse and the reflected waveform 55
613 Low-pass Filter Detail 55
614 Retum loss of Low-pass Filter 57
615 Insertion loss of Low-pass Filter 58
616 Branch Line Coupler Detail 59
617 Scattering Parameters of Branch Lme Coupler 61
618 Series RLC Circuit Detail 62
619 Amplitude ofthe input and transmitted pulse 64
620 Amplitude of the Fourier transform of the input
reflected and ttansmitted pulse 64
621 Parallel LC Circuit Detail 65
622 Amplitude ofthe input and ttansmitted pulse 67
623 Amplitude of the Fourier ttansform of the input reflected and ttansmitted pulse 67
Vlll
CHAPTER I
INTRODUCTION
11 Background
In 1966 Kane Yee (1966) presented what we now refer to as the Finite-Difference
Time-Domain (FDTD) method for modelmg electtomagnetic phenomenon The Yees
algorithm as it is usually called in the literatme is well known for its robustness and
versatility The method approximates the differentiation operators of the Maxwell
equations with finite-difference operators in time and space This method is suitable for
finding the approximate electric and magnetic fields in a complex three-dimensional
stmcture in the time domain Many researchers have contributed immensely to extend the
method to many areas of science and engineering (Taflove 1995 1998) Initially the
FDTD method is used for simulating electtomagnetic waves scattering from object and
radar cross section measurement (Taflove 1975 Kunz and Lee 1978 Umashankar and
Taflove 1982) In recent years there is a proliferation of focus in using the method to
simulate microwave circuits and printed circuit board (Zhang and Mei 1988 Sheen et al
1990 Sui et al 1992 Toland and Houshmand 1993 Piket-May et al 1994 Ciampolini et
al 1996 Kuo et al 1997 Emili et al 2000) hiitially Zhang and Mei (1988) and Sheen et
al (1990) concenttated on modeling planar circuits using FDTD method Later Sui et al
(1992) and Toland and Houshmand (1993) inttoduced the lumped-element FDTD
approach This enabled the incorporation of ideal lumped elements such as resistors
capacitors inductors and PN junctions Piket-May et al (1994) refined the method
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
612 Amplitudes of the Fourier ttansforms of the input
pulse and the reflected waveform 55
613 Low-pass Filter Detail 55
614 Retum loss of Low-pass Filter 57
615 Insertion loss of Low-pass Filter 58
616 Branch Line Coupler Detail 59
617 Scattering Parameters of Branch Lme Coupler 61
618 Series RLC Circuit Detail 62
619 Amplitude ofthe input and transmitted pulse 64
620 Amplitude of the Fourier transform of the input
reflected and ttansmitted pulse 64
621 Parallel LC Circuit Detail 65
622 Amplitude ofthe input and ttansmitted pulse 67
623 Amplitude of the Fourier ttansform of the input reflected and ttansmitted pulse 67
Vlll
CHAPTER I
INTRODUCTION
11 Background
In 1966 Kane Yee (1966) presented what we now refer to as the Finite-Difference
Time-Domain (FDTD) method for modelmg electtomagnetic phenomenon The Yees
algorithm as it is usually called in the literatme is well known for its robustness and
versatility The method approximates the differentiation operators of the Maxwell
equations with finite-difference operators in time and space This method is suitable for
finding the approximate electric and magnetic fields in a complex three-dimensional
stmcture in the time domain Many researchers have contributed immensely to extend the
method to many areas of science and engineering (Taflove 1995 1998) Initially the
FDTD method is used for simulating electtomagnetic waves scattering from object and
radar cross section measurement (Taflove 1975 Kunz and Lee 1978 Umashankar and
Taflove 1982) In recent years there is a proliferation of focus in using the method to
simulate microwave circuits and printed circuit board (Zhang and Mei 1988 Sheen et al
1990 Sui et al 1992 Toland and Houshmand 1993 Piket-May et al 1994 Ciampolini et
al 1996 Kuo et al 1997 Emili et al 2000) hiitially Zhang and Mei (1988) and Sheen et
al (1990) concenttated on modeling planar circuits using FDTD method Later Sui et al
(1992) and Toland and Houshmand (1993) inttoduced the lumped-element FDTD
approach This enabled the incorporation of ideal lumped elements such as resistors
capacitors inductors and PN junctions Piket-May et al (1994) refined the method
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
CHAPTER I
INTRODUCTION
11 Background
In 1966 Kane Yee (1966) presented what we now refer to as the Finite-Difference
Time-Domain (FDTD) method for modelmg electtomagnetic phenomenon The Yees
algorithm as it is usually called in the literatme is well known for its robustness and
versatility The method approximates the differentiation operators of the Maxwell
equations with finite-difference operators in time and space This method is suitable for
finding the approximate electric and magnetic fields in a complex three-dimensional
stmcture in the time domain Many researchers have contributed immensely to extend the
method to many areas of science and engineering (Taflove 1995 1998) Initially the
FDTD method is used for simulating electtomagnetic waves scattering from object and
radar cross section measurement (Taflove 1975 Kunz and Lee 1978 Umashankar and
Taflove 1982) In recent years there is a proliferation of focus in using the method to
simulate microwave circuits and printed circuit board (Zhang and Mei 1988 Sheen et al
1990 Sui et al 1992 Toland and Houshmand 1993 Piket-May et al 1994 Ciampolini et
al 1996 Kuo et al 1997 Emili et al 2000) hiitially Zhang and Mei (1988) and Sheen et
al (1990) concenttated on modeling planar circuits using FDTD method Later Sui et al
(1992) and Toland and Houshmand (1993) inttoduced the lumped-element FDTD
approach This enabled the incorporation of ideal lumped elements such as resistors
capacitors inductors and PN junctions Piket-May et al (1994) refined the method
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
fiirther witii the inttoduction of Eber-Molls ttansistor model and die resistive voltage
source All these components comcide with an electiic field component in the model
Ciampolini et al (1996) introduced a more realistic PN junction and ttansistor model
taking into account junction capacitance Furthermore adaptive time-stepping is used to
prevent non-convergence of the nonlinear solution particularly when there is rapid
change in the voltage across the PN junction In 1997 Kuo et al presented a paper
detailing a FDTD model with metal-semiconductor field effect ttansistor (MESFET)
Recentiy Emili et al (2000) inttoduced an extension ofthe lumped-element FDTD which
enabled a PN junction resistor inductor and capacitor to be combined into a single
component Parallel to the development in including lumped circuit elements into FDTD
formulation another significant advance occmed with the development of frequency
dependent FDTD method (Luebbers and Hunsberger 1992) This development allowed
modeling of system with linearly dispersive dielectric media Further developments by
Gandhi et al (1993) and Sullivan (1996) enabled the inclusion of general dispersive
media where in addition to being dispersive the dielectric can also behave nonlinearly
under intense field excitation A detailed survey in this area can also be found in Chapter
9 ofthe book by Taflove (1995)
FDTD method is performed by updating the Electric field (E) and Magnetic field
(H) equations throughout the computational domain in terms of the past fields The
update equations are used in a leap-frog scheme to incrementally march Electric field (E)
and Magnetic field (H) forward in time This thesis uses the formulation initially
proposed by Yee (1966) to find the approximate electric and magnetic fields in a complex
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
three-dimensional stiiicture in the time domain to model planar circuits and active and
lumped components
12 Scope ofthe Thesis
The object ofthe thesis is to come up with a systematic framework for modeling
electtomagnetic propagation in RF and Microwave circuits with active and lumped
components using S-parameters The FDTD simulation engine is written in C and a
graphing utility (gnuplot) is used to view the electric and magnetic fields The software
mns on Windowsreg platform The simulation engine is developed in a systematic manner
so that it can be extended in fiiture to incorporate more features A convenient and
flexible notation for describing complex three-dimensional models is also inttoduced
This notation describes the model cube-by-cube using lines of compact syntax With this
notation many different types of problems can be rapidly constmcted without having to
modify the declaration in the FDTD simulation engine The description of the model is
stored in a file the FDTD simulation engine simply reads the model file sets up
necessary memory requirements in the computer and the simulation is ready to proceed
Since it is not possible to embody all the progresses and advancements of FDTD
into the thesis the model considered is limited to the foUowings
bull Active and lumped components include resistors capacitors inductors diodes
and ttansistors
bull All conductors are assumed to be perfect electric conductor
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
bull Dielectric material is isottopic non-dispersive and non-magnetic (|i =io)
However the permittivity 8 of the dielectric is allowed to vary as a function of
location time
13 Document Organization
This chapter presented a background for the Finite Difference Time Domain
(FDTD) method The following is a brief account ofthe rest ofthe contents ofthe thesis
Chapter 2 provides some fimdamental concepts of FDTD and inttoduces the Yees FDTD
formulation and its implementation Chapter 3 inttoduces the boimdary conditions and
formulation of a Perfectly Matched Layer Chapter 4 discusses the method to incorporate
lumped elements and devices into FDTD framework Chapter 5 discusses the simulation
and the results to illustrate the effectiveness of the program Finally Chapter 6
summarizes the work and provides a brief conclusion ofthe thesis
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
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degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
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Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
CHAPTER II
FDTD ALGORITHM
21 Inttoduction
This chapter inttoduces the procedure for applying finite-difference time domain
(FDTD) method to time-domain Maxwell equations Specifically the Yees FDTD
formulation will be illusttated hi this chapter the Yee FDTD scheme will be restiicted to
a model with linear non-dispersive and non-magnetic dielectric Various considerations
such as numerical dispersion stability ofthe model will be discussed
22 Maxwells Equation and Initial Value Problem
Consider the general Maxwells equations in time domain including magnetic
curtent density and magnetic charge density
VxE = -M-mdashB (221a) dt
VxH = J + mdashD (221b) dt
V3 = p^ (221c)
V5 = pbdquo (22 Id)
M and pbdquo are equivalent somces since no magnetic monopole has been discovered to
the best of die autiiors knowledge The other parameters for (221a)-(221d) are listed
as follows
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
E - Electiic field intensity
H - Magnetic field intensity
D - Electric flux density
B - Magnetic flux density
p^ - Electric charge density
J - Electric curtent density
Usually we will just call the 1 electiic field and die ~H magnetic field In
(221a)-(221d) if the media is linear we could inttoduce additional constitutive
relations
D = Efl (222a)
^ = ^fH (222b)
where indicates convolution operation in time [3] We would usually call poundf the
permittivity and ii the permeability of the media both parameters are ftinction of
frequency co For a dispersionless medium gf (co) = e and f (co) = i (222a)-(222b)
reduce to the form
3 = eE (223a)
B = HH (223b)
From the Maxwells equations continuity relations for electric and magnetic current
density can be derived For instance from (221a)
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
V(VxE) = -VM-V[mdashB] = 0 dt )
=gtmdash(v5)=-vi7 =mdashp dt^ dt
^ bull ^ = - ^ P trade (224a)
Note that the vector identity VV x = 0 is used Similarly from (22 lb)
V(V X i ) = VJ + v | - ^ 5 1 = 0 dt )
dt^ dt^
V J = - - P (224b)
Notice that in (221a)-(221d) only the cml equations (221a) for pound and (221b) for H
determine how the fields will change over time This provides a clue that perhaps only
these two equations are all that is needed to formulate a FDTD scheme To obtain a
unique solution for the E and H fields of a system described by (221a)-(221d)
additional information in the form of initial conditions and boundary conditions (if the
domain is bounded) is required First assuming an unbounded domain then (221a)-
(22Id) together with the initial conditions for all the field components and sources
iJMp^p^) constitute the initial value problem (IVP) for Maxwells equations
For simplicity suppose the model is a three-dimensional (3D) model with linear
isottopic non-dispersive dielectiics and there is no magnetic charge and magnetic
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
curtent (M = 0p^ =0) Also assume that initially all field components and sources are
0 Thus the following equations describe the Initial Value Problem for the model
VxE = B dt
WxH=J+mdashD dt
VD = p^
VB = 0
E(xyzt)
(225a)
(225b)
(225c)
(225d)
=0 = H(xyzt)
1=0 = J(xyzt)
1=0 = 0 and p^(xyzt)[^^ =0
for all (jcyzt)eR^ and tgt0 (225e)
From (224b)
0 _^
p^(xyzt) = -jVJ(xyzT)dT + C^ where Ci is a constant Using pXxyzt)^^^ =0
and J(xyzt) = 0 we see that Ci =0 Thus =0
p(xyzt) = -^VJ(xyzT)dT
From (225a)
V(Vxpound) = -V dt J dt^ V
Using the initial condition (225e) V5 r=0
= fNH = 0 this implies that (=0
VB = VB = 0 oO (=0
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
Therefore the divergence equation for ~B is implicit in (225a) hi other words (225d)
can be obtained from (225a) provided initial conditions (225e) apply Now consider
(225b)
(vx )= VJ + V (d-^ mdashD
dt J
^ = 0 =gt -VJ = V
( d-mdashD
dt
VD = -jVJdT + C2
Using the initial conditions (225e) VD
Hence from (226)
= euroVE 1=0 t=0
= 0 this implies that C2 =0
VD = -jVJdT = p^
Again this shows that divergence equation for D is implicit in (225b) provided initial
conditions (225e) applies From this argument we see that the divergence equations
(225c) and (225d) are redundant thus die PP of (225a)-(225e) can be reduced to
mdashbull a
VxE = B dt
VxH = J + mdashD dt
(227a)
(227b)
E(xyzt) 1=0
= H(xyzt) 1=0
= J(xyzt) = 0 1=0
Pe(xyZt) = 0 1=0
In general this is also tme when the dielecttic is dispersive and nonlinear as long
as die initial conditions for magnetic and electiic flux densities fulfill
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
VZ) = V5 =0 =0
= 0 This can also be extended to the case when magnetic cmrent
density M and magnetic charge density p^ are present
23 FDTD Formulation
Equations (227a)-(227c) form tfie basis of Yees FDTD scheme [10] There are
a number of finite-difference schemes for Maxwells equations but die Yee scheme
persists as it is very robust and versatile Allowing for conduction electiic curtent
density J = crE (227a)-(227b) can be written as
dE 1 mdash CT-mdash = _ V x E dt s e
= V xE dt jLi
Under Cartesian coordinate system these can be further expanded as
dH^
dt
dH^
dt
dH^
dt
dt
(SE^_^^ dy dz
ifdE^ dE
u dz dx
ifdE^ dE^]
p dy dx
I fdH^ dH^
pound dy dz crE^
(231a)
(231b)
(231c)
(23Id)
10
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
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agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
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Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
dt
I fdH^ dH ^ ^ -oE
dz dx (23 le)
dE^
dt
1 (dH dH^ ^ aE dy dx
(23If) y
Let us introduce the notation
^(ijk) = E^iAxjAykAznAt)) and so on for Ey Ez and Hx Hy H components In
Yees scheme the model is first divided into many small cubes The edges of each cube
will form the three-dimensional space grid The Yees scheme can be generalized to
variable cube size and non-orthogonal grid The position ofthe E and field components
in the space grid is shown in Figure 21
(i4-ijk+l)
^ ^
(ijk-t-l)
(i7Jl-Tgt t (iikiL
X
Cube(ijk)
i
^ H
x^iAxA)
t ^Ul -^jt l
ne^yi-i bullU-)-^X)
(i+ljk)
(ij+lk+li
M
i^
i l - l k)
Av (i+1 j + l k i
Figure 21 Discretization of die model into cubes and the position of field components on the grid
11
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
The cube in Figme 21 is called the Yee Cell From Figure 21 it is observed that
each E field component is smrounded by four H field components similarly each H field
component is surtounded by fom E field components (If die field components on
adjacent cubes are taken into account) For example the component H 2 2
IS
surtounded hy E ^ E E and pound z(iJM-) z(ij-HM-) y(ij+-k)
The inspiration for
choosing this artangement stems from the cml equations (227a)-(227b) As an example
converting (227a) into integral form after using Stokes Theorem
T T ^ d rr-JEdl = -mdashj^Bds dt
This equation states that a changing magnetic flux will generate a circular electric
field surtotmding the flux tube Similarly the integral form of (227b) also states that a
changing electric flux and an electric cmrent will generate a circular magnetic field
surtounding the flux tube Using the center difference operator to replace the time and
( 1 1 ^ space derivatives at time-step n and space lattice point zAx(y + mdash)Agt(A + mdash)Az on
(231a)
_1_ At
f n+mdash
H 2 - H 2 1 0 rbullbull 1 0 bull 1 2 ) 2 2)
1_ Ay E E
V f^^-^^iJ V-2jj i] __
Az
(233a)
12
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
H 2
x lJ+
n
= H ^ A^
poundbull - pound
Ay Az
Repeating this procedure for (33Id) at time step laquo + - and space lattice point
^ 1 (i + -)AxjAykAz
At
(
xi^^ik]^ ii+iytj eAy H
V I 2
1
- 2
2 j 1 2 )
sAz H 2
1 n+-
2 +-_+- V i+-jkmdash
I 2 2j 1 2 2J
- - o pound ^ f x(i+-^Jk)
H-1
Substituting E^^^j^-^ with average time step laquo and n+I
Af
^ pn+l _ pn
x i+mdashJk x i+-Jk I 2- J I 2^ j
sAy H
z + -V I 2
-
2 J L 2 ^ 2 j
sAz
1
2
V
- H 2
1 1 mdashamdash s 2
-n+l r 1 p
xii+^jkj x^i+-jk]
13
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
After some algebraic manipulation we obtained
^n+l
bullbull]
( o-At^
2s
1 + cjAt
V 2s
At_
2s
pn+
J ( 1
H 2
1 + oAt
2s
-H 2 H 2 (bullv4) (H -] ^(4-^-4] +4^--i)
Ay Az
V
(233b)
By the same token the update field equations for the other field components can be
derived
1 1 nmdash
= H 2 H 2 =H 2 - mdash
E -E E -Er jMn 4+^7tj zf+iy+ j h-j-k+^
Az Ax
(233c)
1 n+-
H 2
^E At
2 4] = (44)
y iJ+^M -E E -E
i i ) [^y] x[jk]
Ax Av
(233d)
14
poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
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poundbull 1-
cjAt
~2i 1 + crAt
2s
pn+
y[u-k]
A^
2s
1 + aAt
n+- n+- 2 _ W 2
Az 2s
1 1 n + - n+mdash
Ax
(233e)
pn+ 1 -
aAt ~2s
aAt 1-1-
V 2s At_
Ys
71+1
z(4)
1 + aAt
J
lt J n^mdash n^mdash
A^-i^2] lt V-^ Ax
2s
1 1
H ^ - H 2
Agt
(233f)
Examination of (233a)-(233f) shows that all the field components fall on the
locations accounted for by the space grid of Figure 21 Equations (233a)-(233f) are
explicit in natme thus computer implementation does not require solving for determinant
or inverse of a large matrix To facilitate the implementation in digital computer the
indexes of the field components are renamed as shown in Figure 22 so that all the
indexes become integers This allows the value of each field component to be stored in a
three-dimensional artay in the software with the artay indexes cortespond to the spatial
indexes of Figme 22 In the figme additional field components are drawn to improve the
clarity ofthe convention
15
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
Cubed jk)
t i ( i H i ki
-yiH-lpc)
Figure 22 Renaming the indexes of E and H field components cortesponding to Cube(ijk)
Using the new spatial indexes for field components as in Figure 22 (233a)-
(233f) become
1 n + -
1 nmdash
HxLjk) - ^ x ( A ) -At
Ay Az (234a)
+ - 1 mdash At fj 2 mdash TT 2 _ _
( f _ P P mdash P ^x(iJk-H) ^x(ijk) ^bull( + jk) ^zUjk)
Az Ax
(234b)
ffzijk) - ^z(ijk)
n-- l^t P P P _ P ^y(jk + l) ^y(iJk) ^x(jk ^xUjk)
Ax Ay (234c)
pn+ ^x(lJk)
_oA^^ 2s
1 + crAr 2s J
E+
At_
2s
(
1 + 2s
n + mdash i + I - - n + I N
^jk) ^ - ( V l ) ^ i ( A ) ^ v ( A
Av Ar
(234d)
16
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
rn+l ^y(uk)
1-oAt ~2s
1 + aAt 71+1
^(bullM)+
A^
2s
1 + aAt 2s
1
n + -
^x^jk)
+ -
- 2 x(-7-l)
Az
1 1 A IHmdash nHmdash
Hz(i^jk)-H(Xjk)
Ax
(234e)
71+1
2 ( i J )
1 -aAt
~2s aAt 1 +
V 2s
71+1
z(JA) + bull
A^
2pound
1 + aAt 2s
1 IHmdash
^yljk)
1 i + -
mdash W 2
1 1 n + - +mdash
t^x^jk)-Hx(Lj-k)
Ax Ay
(234f)
Equations (234a)-(234f) form the basis of computer implementation of Yees
FDTD scheme for FVP of Maxwells equations Since (234a)-(234f) compute the new
field components from the field components at previous time-steps these equations are
frequently called update equations Notice that in the equations the temporal location of
the E and H field components differs by half time-step (mdashAr) In a typical simulation
flow one would determine the new H field components at n + - from the previous field
components using (234a)-(234c) Then the new E field components at laquo + l will be
calculated using (234d)-(234f) The process is tiien repeated as many times as required
until the last time-step is reached Because of diis die scheme is sometimes called
leapfrog scheme To show tiiat die divergence equations (225c) and (225d) are implicit
in (234a)-(234f) considermdash(^X)-^(M)) Assuming ^ E and a are
homogeneous and using (234d) diis can be written as
17
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
-^Kljk)-Kt^jk))=
_a^^ 2pound_
aAt 1 +
V 2s At f
K(iik)-K(ij-k) Ax
2s
1 + aAt
2s
A
2s
n+-mdash M 2 _ pf 2
zjk) ^^zij-k) y(ijk) ^yijk-)
Ay Az
( 1
1 + aAt
2s
1 n+-
1 A
_1_
Ax
J
1 A
zi-jk) ^zi-j-k) yi-jk] ^y(-ljk-
Ay Az Ax
Similar terms can be vmtten for Ex and Ey components
(235a)
Ay (pn+l _ pn+i ^ _ ^yiJk) ^y(i]-k))-
_aAty 2s
1 + aAt
2s )
WyiJk)-K[iJ-k)) Ay
2E
1 + crA^
2s
V
2E
I 1 n^mdash fHmdash
^x(ijk) ~^x(jk-) ^lk) ~ ^-U-jk)
Az Ax
__
Ay
A f bdquol
1 + crA
2e
1 n+mdash
I A
Hj mdash H_i I H mdash H (ij-Xk) ~-x(-IA-l) _ --|-IO -(-l-)
AT AX Av
(235b)
18
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
^^zijk)-^z(ijk-)r 2s_ aAt
+ -
1 + V 2s
A^
2s
^zijk)-Eiij^k-i))mdash
( 1
1 + aAt
2s V
2s
1 Pf T- mdashPI 2 PI 2 PI 1 bullyijk) ^y(i-ljk) ^x(ijk)-^x(ij-lk)
Ax Ay V
_1_ Az
1 + aAt 2s
)
1 At ( 1 1 laquo+mdash n^mdash n^mdash nmdash
^yuik) - -^ j ( - i j t - l ) ^xiJk)-^x(ij-lk-)
Ax Ay Az
(235c)
Svimming up (235a)-(235c) the following equation is obtained
pn+l P^ F+l C^+l 17+ 17+ ^xijk)-^x(i-ijk) ^ ^y(iJk)-^y(iHk) ^ ^z(Uk)-^zijk-)
Ax ^y Az
1 -crA
2s ( TT
aAt 1 +
V 2s )
p _ p P _ p P p ^x(ijk) ^x(i-jk) ^y(jk) ^y(ij-lk) _^^zjk) ^z(ijk-)
V Ax Ay Az
(236)
From (236) it is observed that if initiallypounddeg( ) = pounddeg(t) = - (y) = 0 for all j k
then
zpn+l i rlaquo+l i r + 77 P _ P^^ ^x(iJk)-^xi-Jk) ^ ^yijk)~^y(iJ-k) ^ ^zijk) ^ziJk-) _
Ax Ay Az = 0 (237)
for all time-steps n It is immediately recognized that (237) is actually the finite-
difference equivalent of the divergence equation (225c) in free space A similar
procedme can also be applied to the H field components obtaining
19
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
PERMISSION TO COPY
In presenting this thesis m partial fulfilhnent ofthe requirements for a masters
degree at Texas Tech University or Texas Tech University Healdi Sciences Center I
agree tiiat the Library and my major department shall make it freely available for
research purposes Permission to copy this thesis for scholarly purposes may be
granted by die Dhector of the Library or my major professor It is understood that any
copying or publication of this thesis for fmancial gam shall not be allowed widiout my
further written permission and that any user may be liable for copyright infrmgement
Agree (Pemussion is granted)
Student Sigdature Date
Disagree (Permission is not granted)
Stiident Signature Date
1+i bdquo+i _ i _l 1 PI 2 PI 2 PI2 --^^ ^ xiMik)-t^xijk) ^^(iy-
Ax Ay Az
(+iylt) ^(-yt) ^ ^^(i+it)--^^(-yA) ^ ^zijM)-Ezujk) _ Q
Finally it should be pointed out that die Yees FDTD scheme is second order
accurate due to the center-difference operator employed for approximatmg the
differential operators This means the tiiincation ertor for die center difference scheme is
proportional to (AX^ Ay^ A Z ^ A^ )
24 Numerical Dispersion
The numerical algorithms for Maxwells cml equations as defined by (234a)-
(234f) causes dispersion ofthe simulated wave modes in the computational domain For
instance in vacuum the phase velocity of the numerical wave modes in the FDTD grid
can differ from vacuum speed of light In fact the phase velocity of the numerical wave
modes is a fimction of wavelength the direction of propagation and the size ofthe cubes
This numerical dispersion can lead to nonphysical results such as broadening and ringing
of single-pulse waveforms imprecise canceling of multiple scattered wave and pseudo-
refraction A detailed analysis of this numerical dispersion is presented in Chapter 5 of
Taflove (1995) [1] and dius will not be discussed here It is shown in [1 2] that to limit
the amount of numerical dispersion the edges of each cube must be at least ten times
smaller than the shortest wavelengtii expected to propagate in the computational domain
Of course the numerical dispersion will be suppressed even further if smaller cube size is
maintained However using a cube size that is too small will increase the number of cubes
20
needed to fill die computational domain and hence increase computational demand of die
model The mle-of-thumb of Ax Ay Az lt -^ where A is die wavelengtii cortesponding
to the expected highest significant harmonics in the model is adequate for most purposes
25 Numerical Stability
The numerical algorithm for Maxwells cml equations as defined by (234a)-
(234f) requires that the time increment At have a specific bound relative to die spatial
discretization Ax Ay and Az For an isotropic FDTD analysis based on centtal
differencing to remain stable the following mathematical statement must be tme
At lt C((AJC) +(Ay)^ + (AZ) ) 2 C = - ^ (241)
Equation (241) is known as Comant-Freidrichs-Lewy (CFL) Stability Criterion
In an unstable model the computed result for E and H field components will increase
without limit as the simulation progresses
26 Frequency Domain Analysis
The FDTD method is conducted in the time domain In order to obtain frequency
domain generalities such as scattering parameters reflection coefficients and input
impedance Fourier Transform is used to post process the results Oppenheim and Schafer
[12] Zhang and Mei [15] as well as Sheen Ali Abouzahra and Kong [4] present die
details on how such an analysis is performed using Fast Fourier Transform The Fourier
ttansform of an electric field E(t) at frequency fi is calculated using
21
deg (251)
27 Somce
To simulate a voltage source excitation it is necessary to impose an electric field
in a rectangular region Somce can either be sinusoidal or a Gaussian pulse A Gaussian
pulse is desirable as the excitation because its frequency spectrum is also Gaussian and
will provide frequency-domain information from dc to the desired cut off frequency by
adjusting the width of the pulse There are two types of somces the hard somce and the
soft source If a source is assigned a value to E it is referred to as hard source If a value
is added to E at a certain point it is referted to as soft source With a hard somce a
propagating pulse will see that value and be reflected because a hard value of E looks
like a metal wall to the FDTD but with a soft source a propagating pulse will just pass
through
i-gt
28 hnplementation of Basic FDTD Algorithm
Figme 23 illusttates the basic flow of implementing Yees FDTD scheme on a
computer
8 H
E = H = 0
t - n + 05
t = n + l
Tni ti al 1 zati on
1 r
Set excitation
1
^ iL
bull Compute E fields components at interior
bull Compute E fields components at boundary
r
Compute new H field component values
C 1
Max time-steps (n) reached
^ Yes
n = n+l
i k
^ No
y
Post process Results
Figure 23 Basic flow for implementation of Yee FDTD scheme
23
CHAPTER III
BOUNDARY CONDITIONS
31 Introduction
A basic consideration with implementation of FDTD approach on computer is the
amount of data generated Clearly no computer can store an unlimited amount of data
therefore the domain must be limited in size The computational domain must be large
enough to enclose the stmcture of interest and a suitable absorbing boundary condition
on the outer perimeter ofthe domain must be used to simulate its extension to infinity If
the computational domain is extended sufficiently far beyond all somces and scatterers
all waves will be outgoing at the boimdary A suitably formulated boundary condition
will absorb majority ofthe incident wave energy allowing only a small amount to reflect
back into the computational domain This type of boundary condition is known as the
Absorbing Boundary Condition (ABC) Realization of an ABC to terminate the outer
boundary of the space lattice in an absorbing material medium is analogous to the
physical tteatment ofthe walls of an anechoic chamber
Ideally the absorbing medium is a only a few cells thick does not reflect all
impinging waves over their fiill frequency specttami highly absorbing and effective in
near field of a source or a scatterer
24
PML Comer Region
Figure 31 FDTD Lattice terminated by PML slabs
The theories and formulations of some highly effective ABC can be found in
Taflove (1995) and Kunz and Luebbers (1993) Here only the formulation concerning the
ABC used in this thesis is discussed The FDTD algorithm in this thesis uses the Uniaxial
Perfectly Matched Layer ABC terminated by a perfect electric conductor
32 Perfect Electric Conductor
The term PEC is an acronym for perfect electiic conductor and is used to model a
perfectly conductive metal surface The boundary conditions at a perfect electric
conductor require the tangential electric field components to be zero at the boundary
A perfect electiic conductor is modeled by simply setting the tangential electric
field components equal to zero at every time step where die perfect electric conductor is
25
located For example if there is a PEC on one ofthe surface of Cube (ijk) in Figure 32
the following E field components will be zero at all time-steps
^xdjMx) = iy+u+i) = KiijMi) = K(MjMi) =0 (321)
In addition to being used to tertninate the PML layers PEC type conditions can be
assigned to the surface of Yee Cells that are inside the mesh The use of PEC surfaces
inside the mesh allows us to model perfectiy conductive metal surface The use of such
conditions is used to model conductive surfaces and ground planes
PEC surface
Cube(ijk)
Figure 32 PEC on top surface of Cube (i j k)
33 Perfectly Magnetic Conductor
The term PMC is an acronym for perfect magnetic conductor The boundary
conditions at a perfect magnetic conductor require the tangential magnetic field
components to be zero at the boimdary stated mathematically
nxH = 0 where n is a surface normal unit vector
26
It should be clear by examining die figure 22 that there are no tangential
magnetic fields specified on die surface of a Yee Cell To handle this sittiation Li
Tassoudji Shin and Kong recommend use ofthe image method As we shall see using
the image method implies a certain restiiction Shen and Kong [11] provide a clear
inttoduction to the image method Figure 34 part (a) illustrates the interface between
media and a perfect magnetic conductor Part (b) is an illusttation of tiie image problem
copy - bull x
Boundary
Hbdquo
copy Ex
^
Hbdquo copy
Ex
V
Hbdquo
copy Ex
-Hbdquo copy
Hbdquo copy
Ex
-H
copy
(a) PMC Material Shaded
(b) Image Problem
Figure 33 PMC Boundary Conditions
Assigning image field values that are opposite the values of the cortesponding
fields in the media ensure zero tangential magnetic fields on the cell boundary
PMC and PEC surfaces are often used in the modeling certain problems that
posses symmetry to reduce computational time
27
34 Perfectly Matched Layer
A new fervor has been created in this area by JP Berengers inttoduction of a
highly effective absorbing-material ABC designated as the Perfectiy Matched Layer or
PML [8] The itmovation of Berengers PML is that plane waves of arbitrary incidence
polarization and frequency are matched at the boundary and subsequently absorbed in
the PML layer
The PML formulation introduced by Berenger (split-field PML) is a hypothetical
medium based on a mathematical model In the split-field PML if such a medium were to
exist it must be an anisotropic medium An anisotropic medium was first discussed by
Sacks et al [14] For a single interface the anisottopic medium is uniaxial and is
composed of both electric and magnetic permittivity tensors
35 Perfectly Matched Layer Formulation
The idea of Berengers PML [8] is if a wave is propagating in medium A and it
impinges medium B the amount of reflection is dictated by die inttinsic impedance of die
two media
r = mdashmdashmdash for normal incidence (3-51)
The inttinsic impedance of a medium is determined by tiie dielectric constant e and
permeability i
28
If the value of p changes with e so that ri remains a constant F would be equal to
zero and there would be no reflection but the pulse will continue to propagate in the new
medium So the new medium must be lossy so that the pulse will die out before it hits the
boundary This can be accomplished by making e and i complex Sacks et al [14] shows
that there are two conditions to form a PML
1 The impedance going from the background medium to the PML must be a
constant
^ = 1 (353) = 1 ^Fx
2 In the direction perpendicular to the boundary (the x direction for instance) die
relative dielectiic constant and relative permeability must be inverse of those in
the other directions
(354) poundFX
MFX
Spy
1
^^Fy
And each of die complex quantities will be of die form
^Fm = ^ l F m + 7 ^Hm
(3 i gt) form=xoryorz o--i
s and u are fictitious values to implement the PML and have nothing to do with the
real values of s and n which specify die medium
Following selection of parameters satisfies Equation (354) [7]
29
^Fm =MFn=^ (356a)
Dm Hm
o
CTr (356b)
Substittiting Equations (356a)-(356b) in (355) the value in Equation (353) becomes
fin = M Fx
bullFx
1 + -7raquo^0
| l + - ^ I laquo^0
= 1
This fulfills the above mentioned requirements
The development of the PML is done for two dimensional version and then it is
developed for three dimensions as it closely follows the two dimensional version The
arrival for three dimensional PML equations from two dimensional PML equations is
derived in the appendix A This section only goes as far as presenting the general form of
three dimensional PML equations
curl h =
H(i+^fk+^)-H(i-^jk+^)
-H(ij+^k+^)+H(ij-^k+^)
I rn-Xi- - bull 1 Io(iJk + -) = ir(ifk + -) + curl_h
laquo+-- 1 laquo mdash 1 D HiJk + -) = gi3(i)gj3(i)D^ ilJk + -)
gi2(i)gj2(if (^^^^ _ ^^j^^)^^ ( ^ 1)^
30
The one dimensional gparameters are defined as
gm2(m) = -mdash where m = iik 1 -I- fml(m)
1-fml(m) gm3(m)= ) where m = ijk
1 + fml(m)
where
fil(i)=^
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scaling
a(x) = ^ xY
7 -
And optimal value of cr is calculated using the relation
m + 1 ^opt =
1507ryjs^Ax
For polynomial scaling the optimal value ofm is typically between 3 and 4
Similarly
DrHi + fk) = g3(j)gk3(k)DrHi + jk)
+ gk2(k)gj2(j) ^curl_h + gil(k)r^(i + ^jk)
31
1 1 1
laquo + - 1 laquo--- 1 Dy HhJ + -k) = gi3(i)gk3(k)D^ HiJ + -^k)
gi2(i)gk2(k) curl_h + gjl(j)Il(ij + k)
2 J
HT ii + j k) = fj3(j)fk3(k)H (f + 1 j k)
+ fj2(J)fk2(k)
curl _e + filii)Jbdquo (i + - j k)
1 1 H(ij+-k)^fi3(i)fk3(k)H(ij+-k)
fi2(i)fk2(k) 1 n+-
curl _e + fjl(j)Jffy Hij + -k)
Hf (U j k + ^) = fi3(i)fj3(j)H (i j A + i )
+ fi2(i)fj2(j)
curl _e + fkl(k)Jbdquo^ ^ (i Jk + -)
32
CHAPTER IV
MODELING LUMPED COMPONENTS
41 Introduction
Chapter 2 and Chapter 3 provided the algorithms of the FDTD method for
modeling a three dimensional distributed passive problems hi this chapter die inclusion
of active and passive component models such as resistor capacitor inductor and bipolar
junction transistor (BJT) will be presented The inclusion of resistor capacitor and
inductor was first proposed by [13] The formulation for resistor capacitor and inductor
follows an altemative approach using S-parameters proposed by [16]
42 Linear Lumped Components
In FDTD method a lumped component is assumed to coincide with an E field
component Linear lumped components encompass elements such as resistor capacitor
and inductor Associated with each lumped element is the I-V relation A certain potential
difference imposed across the lumped element will result in a certain amount of curtent
flowing through the lumped element with negligible time delay Since the lumped
element coincide with an E field component the curtent along the element and the rate of
change of the E field across the element determine die values of the H field components
surtounding the E field Thus to include the lumped element into die FDTD model only
the cortesponding E field component needs to be modified to reflect the presence of an
electiic current density due to die lumped element The basis of formulating the update
33
equation for the E field component with the presence of lumped element is Maxwells
curl equation (227b) The following procedures are essentially based on Piket-May et al
(1994) [13] Here it is assumed the lumped element to coincide with E field component
Ezijk) generalization to other E field components can be easily carried out In all cases
the voltage V across the lumped element at time-step n is defined by
F =-pound Az (421)
Resistor
Assuming a z-directed resistor in a dielectric with permittivity e Let the
resistance be R From V = IR the V-I relationship for the resistor at time-step n + mdash can
be written as
i+- _ Az bdquo+] p (A^f) bullz(yt) - 2jj- = (M) (bully)
The cortesponding cmrent density is given by
1 laquo+-
T^2 _hAlplusmn (423) ^ bull ^ bull bullgt A x A y
From Maxwells equation we have
V x = J + mdashpound) dt
Reartanging the equation (424) will lead to an update equation for E^^jj^^ which
coincides with a lumped resistor R
34
^z(ijk) -
AtAz
2RsAxAy
AtAz 1- 2RsAxAy
F +
At^
s
1 + AtAz V X ^ Z ( 0 (425)
2ipoundAxAy
Similar update equations can be derived for E for a lumped resistor along x and y
direction
Capacitor
1-V relationship for an ideal capacitor is given by
dV I = C-
dt
Let the capacitor with capacitance C be z oriented the I-V relationship at time step n + -
can be written as
+- CAz bdquobdquo+ 2 - z(iyi) A^
Pz(ijk) EzUJk)) (426)
The cortesponding cmrent density is given by
n+- 2 =
2
AxAy (427)
Using the similar procedure we used above die update equation for die associated E field
is given by
^ A ^
Z7+ _ p A ^zUJk) - ^zUjk) ^
1 + CAz
poundAxAy ^
1 n + -
Vxz(^y (428)
35
Inductor
I-V relationship for an ideal inductor is given by
I = jv(T)dT
Let the inductor with inductance L be z oriented the I-V relationship at time step laquo + -
can be written as
n+- AzAt ^ ^zojk)=mdashr-ZEujk) (429)
The cortesponding curtent density is given by
1 laquo + -
1 2 -2 _ ^z(ijk)
AxAy
Using the similar procedme we used above the update equation for the associated E field
is given by
EZ-Kraquo -^vx^i -^t^ (laquo-gto)
36
43 hicorporating 2 Port Networks with S-Parameters
Simple networks may have an equivalent lumped circuit element model and they
can be included in the FDTD modeling using the algorithms described in the previous
section In this section a general algorithm proposed by [16] is presented
The electric field time-stepping equation is modified to allow the addition of the
two-port network From Maxwell equation we have
V x = J bdquo + | ^ Z ) (431)
dt
where Jnet is an impressed curtent density through which the network will be
incorporated Let the network be oriented in z-direction so the curtent density is given by
J bdquo = - ^ (432)
AxAy
At port 1 ofthe network
Iea VbdquoeaYuit)^VbdquoenY2(t) bull^ net ~ AxAy AxAy
^ j _^EbdquobdquoY(t)^Ebdquo^2ynit) (4 3 3) ^bull^^ AxAy
where indicates convolution
To incorporate the network characteristics in die time-domain the S-parameters is
converted to Y-parameters and then an IFFT or hiverse Discrete Fourier Transfonn is
performed to ttanslate die Y-parameters in the frequency domain into time domain
The time-marching equation for the z-component ofthe electtic field at Port i (i =
1 2) is given by
37
At 1
EL =E+mdashVXH 2 ^zPorti -zPorti
AtAz
1+-2
z(y)
^ oAxAy Zi^i(opoundbdquo(-+Ei^2(opound =0 1=0
zPort2(n-l)
(434)
Similar update equations can be derived for electric field E along x and y
direction
38
CHAPTER V
FDTD SIMULATION PROGRAM
51 Inttoduction
In this chapter the FDTD simulation program will be described in detail The
flow the dynamics and important algorithms of die FDTD simulation engine will be
shown Simulation examples to verify the accuracy of the program are presented in
Chapter 6
52 Building a Three Dimensional (3D) model
A three-dimensional model consists of many characteristics and elements To
describe to the FDTD simulation program the natme of the model obviously a computer
file with certain syntax is required Syntax is developed in this thesis and the file
describing the model This file defining the model is a text file consisting of lines of text
which describe the properties of a single cell or a collection of cells or for all cells in the
3D model
Syntax
Identifier XI Yl Zl X2 yen2 Z2 Material Property
XI = Start coordinate along x-axis
Yl = Start coordinate along y-axis
Zl = Start coordinate along z-axis
X2 - End coordinate along x-axis
39
Y2 - End coordinate along y-axis
Z2 = End coordinate along z-axis
Material Property = Characteristic value of die identifier for die specified range of cells
Identifier = Characteristic of the cell or collection of cells 1 represents Dielectiic
material 2 represents MetalPEC 3 represents PMC 4 represents a Lumped element
Example
2 0 0 0 60 120 3 22
100060120000
4303953041500
The first line declares a range delimited by (0 0 0) and (60 120 3) with
dielectric constant Zr - 22 The second line specifies a range delimited by (0 0 0) and
(60120 0) is a perfect conducting plane (specifying a ground plane) The third line
specifies it is a lumped element port 1 specified by co-ordinates (30 39 5) and port 2
specified by co-ordinates (30 39 5)
53 FDTD Simulation Engine - An Overview
The basic simulation engine is shown in Figure 51 In this chapter only an
overview of the program will be given The actual somce code for the complete software
contains more dian 3000 lines of C code FDTD is coded in C language widi ftittire
upgrade and expansion in mind
The output of the software contains the following files the electiic and magnetic
components and voltages at specified points The output files are dat files (data files)
40
Thus visualizing software is required to view the output file Gnuplot [17] is employed in
visualizing the output file This software can be integrated with the curtent FDTD
software but the C language integration works only on LINUX platform This integration
is not provided in my thesis as it will predominantiy be used on Windows platform
Input File(s)
FDTD Simulation Engine jo bull Read and parse input file(s) bull^^ bull Setup all data-types and
variables bull Preprocessing and
determining computational coefficients
bull Enforce Boundary conditions bull Perform E and H updates
Output Files
Figure 51 Major software block of FDTD simulation program
41
54 Software flow
High level flow of FDTD module is shown in figure 52
Standard FDTD
Routine
Start
hiitialize FDTD cells
Parse the input file(s)
Initialize the Field obiects
Update Boundary Conditions
Update E field Components
Update H field Components
End of Iteration
Yes
Stop
Figure 52 Flow of FDTD Program
No
42
There are more steps in the initialization portion as compared to the basic flow of
Figure 52 as the FDTD simulation engine needs to accommodate different types of
elements with various update functions
55 Important Functions in Simulation
A number of functions are declared for the FDTD Simulation Engine Among the
more important fiinctions are the electric field update function (fncUpdateE) magnetic
field update function (fncUpdateH) and Absorbmg Boundary Condition (fncPML) The
electric field update function contains the variables to store all the E field components in
the model [Equations (233a)-(233f)] Similarly the magnetic field function contains the
variables to store all H field components in the model [Equations (233a)-(233f)] The
Absorbing Boundary Condition function contains some variables and artays which store
the values calculated to implement a PML [Equations (3524a)-(2524c) (3525a)-
(2525c)] The other functions are the Discrete Fourier Transform function (fncDFT)
which is used to obtain the frequency domain quantities of the electric field components
S parameter function (fncsparam) which is used to find Y parameters from an S-
parameter file to model lumped elements Complex number arithmetic operations
functions (f nccomplex) is used to perform all die complex number arithmetic operations
as S-parameter values are complex numbers
43
CHAPTER VI
SIMULATION RESULTS
61 Inttoduction
In this chapter several simulation examples using the FDTD software developed
will be shown to verify the accuracy and effectiveness ofthe program Simulation results
of a microstrip line line fed rectangular patch antenna low-pass filter a branch line
coupler series RLC circuit and parallel LC modeled using the FDTD software will be
presented
62 Microstrip Line
The model of the Microstrip line used for this numerical simulation is shown in
Figure 61
L2= 128
246 mm
Figure 61 Microstiip Line
In order to cortectly model the diickness of the substtate Az is chosen so that
tiiree nodes exactly match die diickness hi order to cortectly model the dimensions ofthe
44
ttansmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the ship
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
dimensions ofthe metallic strip are thus 6 Ax x 3 00 Ay The reference plane for the port 1
is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 60 x 320 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero
621 Modeling ofthe materials
In this case we are dealing with only three materials free space the dielectric
material ofthe substtate and the metal The circuit shown in Figme 61 is constmcted on
Dmoid substtate with Sr = 22 and no appreciable loss term The metal is specified by
ensuring the E fields within diose points cortesponding to die metal remain as zero
622 Somce
The source is initialized by a uniform E field between die smface and the ground
plane at a point Vi shown in Figure 61
45
A hard source cannot be used as the main information is the reflection coming
back the opposite way A hard source would look like a metal barrier which will make
the source to decay almost to zero so a soft source is used since we want the propagating
pulse to pass through
pulse = exp(-l X (pow((T -tjl spread20)))
dz[i][J][k] = dz[i][j][k] +pulse
623 Boundary Condition
We are using PML for our absorbmg boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
624 Results
The simulation is performed for 6000 time steps The spatial distiibution of E (x
y t) just beneath the microstiip at 200 400 and 1000 time steps is shown in Figure 62-
64 The amplitude of the input pulse and the amplitude of the Fourier ttansform of the
input pulse are shown in the Figme 65 Figme 66
46
Ez
Ez 200 Stepsdat
- 200 -- 150 100
250 300
350
60
Ez
Figure 62 Distribution of Ez(xyz) at 200 time steps
Ez 400 Stepsdat
300 350
Figure 63 Distiibution of Ez(xyz) at 400 time steps
47
Ez
Ez 1000 Stepsdat
250 300
350
Figure 64 Distiibution of Ez (xyz) at 1000 time steps
00005
00004
1000 2000 3000 4000
Time Steps
5000 6000
Figure 65 Amplittide of die input pulse
48
u u o
0028
0026 bull D
1 0024 pound
t 0-022 3
pound 002
0018
0016
n m I
rmdash--___ _ -
-
-
-
Outputdat -
~
-
bull
-
bull
8 10 12
Frequency (GHz)
14 16 18 20
Figure 66 Amplitude ofthe Fourier ttansform ofthe input pulse
63 Line Fed Rectangular Microstrip Antenna
The model of the line fed rectangular patch antenna used for this numerical
simulation is showm in Figme 67 [4] The operating resonance approximately
cortesponds to the frequency where Li = 1245nim = XI2
49
Li = 1245 mm ylt gt
0794 mm
246 mm
Figme 67 Line-fed rectangular microstrip antenna
In order to cortectly model the thickness of the substrate Az is chosen so tiiat
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
antenna Ax and Ay are chosen such that an integral number of nodes will exactly fit the
rectangular patch
The space steps used are Ax = 0389 mm Ay = 04mm and Az = 0265mm The
rectangular patch is thus 32Ax ^ 40Ay the length ofthe microstrip from the source plane
is 50Ay and the reference plane for port 1 is chosen 1 OAy from the edge of the FDTD
wall The line width of microstrip line is modeled as 6Ax A 10 cell PML is used and the
total mesh dimensions are 60 x 120 x 25 in x y and z directions respectively
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so diat the Gaussian will start at approximately
zero
50
631 Modeling of Materials
In this case we are dealing with only three materials free space the dielectiic
material ofthe substtate and the metal The circuit shown in Figure 67 is consttoicted on
Duroid substrate with s = 22 and no appreciable loss term The metal is specified by
ensuring the tangential E fields within those points cortesponding to the metal remam as
zero
632 Source
The source is defined by a uniform Ez field between the anterma surface and the
ground plane at a point Vi shown in Figure 67
A hard source carmot be used as the as the main information is the reflection
coming back the opposite way A hard source would look like a metal barrier so soft
source is used since we want the propagating pulse to pass through
pulse = exp(-l X (pow((T -1^) I spread20)))
dz[i][j][k] = dz[i][j][k] +pulse
633 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of die PML are 20x20x 10 in x y and z directions
respectively
51
634 Resuhs
The simulation is performed for 4000 time steps The spatial distribution of Ez (x
y t) just beneath the microstrip at 200 400 and 600 time steps is shown in Figure 69 -
611 which show good agreement with [4]
Ez 200 Stepsdat
Ez
120 100
Figure 68 Distribution of Ez(xyz) at 200 time steps
52
Ez
Ez 400 Stepsdat
120 100
Figure 69 Distribution of Ez (xyz) at 400 time steps
Ez 600 Stepsdat
Ez
120 100
50 - g5 0
Figure 610 Distiibution of Ez (xyz) at 600 time steps
53
The scattering coefficient |Sii| and the amplitude ofthe Fourier ttansforms ofthe
input pulse and the reflected wave form shown in Figme 611 612 show good
agreement with [4]
8 10 12 14 Frequency (GHz)
16 18
Figure 611 Retum loss ofthe rectangular antenna
54
003
o I 0015
pound 001
0005
Inputdat Outputdat
0 2 4 6 8 10 12 14 16 18 20
Frequency (GHz)
Figure 612 Amplitudes ofthe Fourier transforms ofthe input pulse and the reflected
waveform
64 Microstrip Low-pass Filter
The acttial dimensions ofthe low pass filter to be simulated are shown by Figure-
613 [4]
2032 mm
254 mm
2413 mm
Figme 613 Low-pass filter detail
55
The space steps used are Ax = 04064 mm Ay = 04233nmi and Az = 0265mm
The dimensions of the long rectangular patch are thus 50Ax x 6Ay The reference plane
for port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 80 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
641 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point Vi shown in Figure 613 Soft source is used since we want the
propagating pulse to pass through
642 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation tiius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
56
643 Resuhs
The simulation is performed for 6000 time steps The scattering coefficient
results |Sii| (Retum Loss) and IS12I (Insertion Loss) shown in Figure 614-615 show
good agreement with [4]
8 10 12
Frequency (GHz)
14 16
Figure 614 Retum loss of Low-pass filter
18
57
T3
fM
-30
-40
8 10 12 14
Frequency (GHz)
16 18 20
Figure 615 Insertion loss of Low-pass filter
58
65 Branch Line Coupler
The acttial dimensions of the branch line coupler to be simulated are shown by
Figure 616 [4]
975 mm
mm
396 mm 2413 mm
Figure 616 Branch Line Coupler Detail
To model this circuit Ax Ay and Az are chosen to match the dimensions of the
circuit as effectively as possible In order to correctly model the thickness of the
substtate Az is chosen so that three nodes exactly match the thickness The space steps
Ax and Ay are chosen to match the center-to-center distance of 975mm exactly Again
small ertors in the other x and y dimensions occm
The space steps used are Ax = 0406 mm Ay = 0406 mm and Az = 0265nim and
the total mesh dimensions are 60 x 140 x 25 in x y and z directions respectively and a
10 cell PML is used dius the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively The center-to-center distances are 24Ax x 24Ay The distance
59
from the somce to the edge of the coupler is 50Ay and the reference planes are chosen
lOAy from the edge ofthe coupler The strip width is modeled as 6 Ay and die wide ship
ofthe coupler is modeled as lOAy
The time step used is At = 0441 picoseconds The Gaussian half-width is T = 15
picoseconds and time delay to is set to 3T so that the Gaussian will start at approximately
zero
651 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane at a point VI shown in Figme 616 Soft source is used since we want the
propagating pulse to pass through
652 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus die dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
60
653 Results
The simulation is performed for 4000 time steps The scattering coefficient
results shown in Figure 617 show good agreement with [4]
4 5 6 Frequency (GHz)
Figure 617 Scattering Parameters of Branch Line Coupler
61
66 Series RLC Circuit
The layout ofthe series RLC circuit to be simulated are shown by Figure 618
Lumped Element
Ground plane
Figure 618 Series RLC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exactly match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04mm and Az = 0265mm The
dimensions of the metallic ship are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for die port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions of die PML are 20 x 20 x 10 in x y and
z directions respectively
The time step used is At = 0441 picoseconds A Gaussian pulse is used for the
somce Its half-width is T = 15 picoseconds and time delay to is set to 3T so diat die pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
62
661 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft somce is used since we want the propagating pulse to pass through
662 Boundary Conditions
We are using PML for our absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
663 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 619 Figure 620
The results do not satisfy the theory and large variations occur between the
simulations and the theory The simulation results show that the series RLC circuit is
behaving like an open circuit
63
00005
00004
00003
00002
00001
-00001
-00002
Vtotdat Vtransdat
0 200 400 600 800 1000 1200 1400 1600
Time Steps
Figure 619 Amplitude ofthe input pulse and ttansmitted pulse
0035
003
0025 01 TJ 3
t 002 E lt
^ 0015 3
001
0005 h
bull
DFT_Vincdat DFT_Vrefdat
DFT Vtransdat
0 10 20 30 40 50 60 70 80 90 100
Figure 620 Fourier amplittide of die input pulse reflected and ttansmitted pulse
64
67 Parallel LC Circuit
The layout ofthe parallel LC circuit to be simulated are shown by Figme 621
Lumped Element
Ground plane
Figure 621 Parallel LC Circuit Detail
In order to cortectly model the thickness of the substtate Az is chosen so that
three nodes exacdy match the thickness In order to cortectly model the dimensions ofthe
transmission line Ax and Ay are chosen such that an integral number of nodes will
exactly fit the strip
The space steps used are Ax = 0386 mm Ay = 04nim and Az = 0265mm The
dimensions of the metallic strip are thus 6Ax x 50Ay and the lumped element is 1 cell
thick The reference plane for the port 1 is lOAy from the edge ofthe FDTD wall
The total mesh dimensions are 30 x 130 x 25 in x y and z directions respectively
and a 10 cell PML is used thus the dimensions ofthe PML are 20 x 20 x 10 in x y and z
directions respectively
65
The time step used is At = 0441 picoseconds A Gaussian pulse is used for die
source Its half-width is T = 15 picoseconds and time delay to is set to 3T so that the pulse
will start at approximately zero The cutoff frequency is chosen as 7GHz
671 Source
The source is initialized by a uniform Ez field between the surface and the ground
plane Soft source is used since we want the propagating pulse to pass through
672 Boundary Conditions
We are using PML for om absorbing boundary condition A 10 cell PML is used
in the simulation thus the dimensions of the PML are 20 x 20 x 10 in x y and z
directions respectively
673 Results
The simulation is performed for 1500 time steps The amplitude of the input
pulse ttansmitted pulse and the amplitude ofthe Fourier ttansform ofthe input pulse and
transmitted pulse are shown in the Figure 622 Figme 623
The results do not satisfy the theory and large variations occur between the
simulations and the dieory The simulation results show that the parallel LC circuit is
behaving like a short circuit
66
00006
00005
00004
00003
00002
00001
0
-00001
-00002
-00003
-00004
-00005
V
Vtotdat Vtransdat
200 400 600 800 1000 1200 1400 1600
Time Steps
Figure622 Amplitude ofthe input pulse and transmitted pulse
004
0035
003
0025
002
0015
001
0005
DFr_Vincdat DFr_Vrefdat
DFT Vtransdat
- L 1mdash
0 10 20 30 40 50 60 70 80 90 100
Figure623 Fourier amplittide ofthe input pulse reflected pulse and ttansmitted pulse
67
CHAPTER VII
CONCLUSION
This chapter brings this thesis to a close by drawmg from all odier chapters m
this thesis to summarize the nuggets of knowledge that we revealed A few of die
highlights from this diesis and suggestions for fiiture research are discussed
71 Thesis Highlights
A systematic framework for modeling electtomagnetic propagation in RF and
Microwave circuits with active and lumped components using Finite-difference Time-
domain (FDTD) approach has been developed in this thesis Chapter 2 provided the
algorithms to approximate a three-dimensional model and all the update equations
needed to find the E and H fields in the model as a function of time-step Chapter 3
provided the algorithms to implement a three-dimensional PML absorbing boundary
condition Chapter 4 provided algorithms to include the presence of lumped
components using S-parameters in the diree-dimensional FDTD model Chapter 5
introduced a convenient notation to describe a complex three-dimensional model cube-
by-cube This notation is very flexible and compact with ample capacity to include
new feattires in the fiiture In Chapter 5 die algoridims presented in Chapter 2
Chapter 3 and Chapter 4 are systematically converted mto a computer program The
architecttire of die software is also illustrated clearly in Chapter 5 Aldiough the
algorithm can be implemented in any programming language it was coded using C
68
programming in this thesis Verification ofthe effectiveness ofthe FDTD framework is
performed in Chapter 6 where many simulations were carried out
72 Futtire Work
The following are some remarks for futme improvement
bull The present work uses Cartesian co-ordinate system so solving problems in other
co-ordinates system can be implemented
bull Include radiation characteristics for antermas
bull Include formulation of dispersive and magnetic materials
bull More verifications for active circuits
bull Improve the speed ofthe simulation and optimizing the code further
bull Provide a GUI
Finally it should be cautioned that as with all computer simulations the results are
only as good as the models This means diat if the models are not accurate or do not
reflect the actual physics of the components then the result will not match well with
measurements
69
REFERENCES
[I] Allen Taflove Advances in computational electrodynamics The Finite-Difference Time-Domain method 1995
[2] Allen Taflove Computational electtodynamics The Finite-Difference Time-Domain method 1998
[3] Constantine A Balanis Advanced Engineering Electtomagnetics Wiley Text Books New York 1989
[4] David M Sheen and Sami L Ali Application ofthe Three-Dimensional Finite-Difference Time-Domain Method to the Analysis of Planar Microstiip Circuits IEEE Trans Microwave Theory Tech vol38 No7 July 1900
[5] Dermis M Sullivan A simplified PML for use with the FDTD mediod IEEE Microwave and Guided Wave Letters Vol 6 Feb 1996 pp 97-98
[6] Dennis M Sullivan An unsplit step 3-D PML for use widi the FDTD method IEEE Microwave and Guided Wave Letters Vol 7 July 1997 pp 184-186
[7] Dermis M Sullivan Electromagnetic Simulation using the FDTD Method IEEE Press
[8] J P Berenger A perfectly matched layer for the absorption of electtomagnetic bullwaes^ Journal of Computational Physics vol 114 no 2 pp 185-200 1994
[9] KS Kunz and RJ Luebbers The Finite Difference Time Domain Method for Electromagnetics Boca Raton FL CRC Press 1993
[10] Kane S Yee Numerical Solution of hiitial Boundary Value Problems Involving Maxwells Equations in Isottopic Media IEEE Transactions on Antennas and Propagation vol 14 no 3 May 1966
[II] L Shen and J Kong Applied Electtomagnetism PWS Publishers Boston Massachusetts 1983
[12] Oppenheim A V Schafer R W Discrete-time signal processing Prentice Had 1989
[13] Piket-May M J Taflove A Baron J (1994) FD-TD modeling of digital signal propagation in 3-D circuits witii passive and active loads IEEE Trans Microwave Theory and Techniques Vol 42(No 8) 1514-1523
70
[14] Sacks Z S D M Kingsland R Lee and J-F Lee A perfectiy matched anisottopic absorber for use as an absorbing boundary condition IEEE Transactions on Antennas and Propagation Vol 43 No 12 1460-1463 December 1995
[15] X Zhang and K K Mei Time-Domain Finite Difference Approach to the Calculation of the Frequency-Dependent Characteristics of Microstrip Discontinuities IEEE Transactions on Microwave Theory and Technique vol36 no 12 December 1988 pp 1775-1787
[16] Xiaoning Ye James L Drewnaik Incorporating Two-Port Networks with S-Parameters into FDTD IEEE Microwave and Wireless Components Vol 11 No2 Febmary2001
[17] httpwvv^wgnuplotinfo Gnuplot
71
APPENDIX
72
This appendix contains die derivation of three dimensional PML equations we
used The development ofthe PML is done for two dimensional version and dien it is
developed for three dimensions as it closely follows the two dimensional version
We begin by using Maxwells equation
^ = V x i (Ala) dt
D(CD) = Sosl(co)E(o)) (A lb)
dH 1 ^ ^ = Vxpound (Alc)
dt ju
where D is the electric flux density We will now normalize the above equations using
E^ ^E (A2a)
D= -^D (A2b)
Using (A2a)-(A2b) in (Ala-Alc) will lead to
^ = mdash L - V x (A3a) dt is^pi
D(CO) = SI(CO)F(CO) (A-3b)
dH I -VxE (A-3c) dt V ^oMo
In three-dimensional case we will be dealing with six different fields E^ Ey pound- H Hy
H hi two-dimensional simulation diere are two groups of vectors (1) the ttansverse
73
magnetic (TM) mode which is composed ofE^ Hx Hy (2) ttansverse electiic (TE) mode
which is composed of Ex Ey_ Hx
We will work here with TM mode and equation (A3) is reduced to
dD^
dt = -Cr
dH dH^
dy dx
D((o) = s^(o))E(co)
dt
dH^
dt
Co =
dt
dt
1
(A4a)
(A4b)
(A4c)
(A4d)
We now have eliminated s and n from the spatial derivatives for the normalized
units Instead of putting them back to implement the PML we can add fictitious dielectric
constants and permeabilitiesf^^ JUFX and i^^[1315]
(dH dH joD^Sp^(x)sl^(y) = Co
D(CO) = SI((O)I(O))
H(co)MFxix)-MFx(y) = -^c
Hy(co)Mry(x)MFyiy) = ^o
dy dx
dE^
dt
dE^
dt
(A5a)
(A5b)
(A5c)
(A5d)
74
We will begin by implementing PML only in x direction Therefore we will retain
only the x dependent values of sp and p^ in equations (A5a) - (A5d)
joD^sp^(x) = c^ dH dH^
V dy dx
Hx((o)pp(x) = -c^ dE^
dt
dE ^yi(0)MFy(x) = Cbdquo^
Using the values of equation (355)
Jed 1 +
joy
J(o
1 +
o-o(x)
J03poundo )
o-p(^)
D=c dH^JJ[A dy dx
H^(co) = -c^ dE
dy
HAco) = cbdquo dE^
dx
(A6a)
(A6b)
(A6c)
The penneability of Hx in equation (A6b) is the inverse of H in equation (A6c)
which satisfies the second die requirement of PML mentioned in equation (354)
Equations (A6a) - (A6c) have to be put in FDTD fortnulation Moving equation
(A6a) to the time domain along with the spatial derivatives we get
75
1 nmdash
D (ij) = gi3(i)D^ Hij)
f 1
-f gi2(i)AtCQ
H(i + -J)-H(i--j)
Ax
1 ^
2
1 1 H(ij+-)-H(ij--)
Ax
(A7)
where the parameters gi2 and gi3 are given by
1 gi2(i) =
1 + o-rf(0-A
2poundn
(A8a)
1 -oAi)At
gm) = ^- 1 +
2S
(A8b)
Almost identical treatinent of equation (A6c) gives
Hy=fi3(i + ^)H(i + j)
+ fi2(i + ^Atcbdquo E ~Hi + lj)-E^ Hij)
Ax
(A9)
where
fi2(i + ) = 1
a(i + -)At 1 +
2S 0 y
(A 10a)
76
a(i + -)At 1 2
fi3(i +-) = ^fo 2 1
a(i + -)At l + -2Sn
(A 10b)
Equation (A6b) needs a different tteattnent so it is rewritten as
jaH^ = -cbdquo dE^^a^(x) 1 dE^
_ dy jo)s^ jo) dy
The spatial derivative will be written as
1 1 n + - n+-
dE^ ^E^ ^ij + l)-E^ HiJ)^ curl_e dy Ay Ay
Implementing this into FDTD formulation gives
Hf(ij+^)-H(ij+^)
A = -c curl e a^ (x) J^ curl e
Ay n=o Ay
1 1 HT (i J + -) = H (i j + -) + c^At mdash - mdash +
curl_e c^At a[j(x) n^ 1
^y ^y poundo IHXHIJ + T)
Equation (A6b) is implemented as the following series of equations
curl_e = E^ ~Hhj)-Ez HUj + l) (A 11a)
1 1 -- 1
Jfix (i^J + ^) = hx iiJ + 2^ + ^^-^ (A lib)
Hf a j^b=H a y -F i )+ - ^+ fHiDO (u j+) (A lie)
77
with filii) = oii)At
2s
In calculatingand g parameters the conductivity is calculated using polynomial
or geometric scaling relation Using polynomial scalmg
aix) = 11 ^-
And optimal value of cris calculated using the relation
w-i-1 ^opt =
150nJsAx
For polynomial scaling the optimal value ofm is typically between 3 and 4
So far we have shown the implementation in the x direction Obviously it must
also be done in the y direction Therefore we must also go back and add y dependent
terms from equations (A5) we set aside So instead of equations (A6) we have
JO) 1-F-o-z)(^)Yi ^oiy)
JO)
JO)
1 +
JO)S )
q- f l (^ )
JO)So
1+-J0)pound
D =Cr d_H^_dH^
dy dx ^
1 + jo)poundo
dE HAo)) = -cbdquo-^
dy
Y
Jo)poundo A J^^o
(A 12a)
(A 12b)
(A12c)
Using the same procedme as before equation (A7) will be replaced by
78
1 nmdash
A (^7) = gi^(i)gj3(i)DrHiJ)
+ gi2(i)gj2(i)Atc
1 H(i + -j)-HUi-^j)
Ax
H(ij+^)-H(ij-^)^
Ax
In y direction Hy will need an implementation which is similar to the one used for
Hv in the x direction
1 n+-curl_e = E^ Ai + lj)-E^ Hij) (A 13a)
1 1 _i 1 Cx ( + -^) = ^ 1 (i + -J) + curl_e (A 13b)
Hri + ^j) = fi3(i + ^)H(i + ^j)
1 curl pound bullmdash 1
+ y 2 ( - F i ) ^ ^ + i i ( 0 ^ ( - F - y )
(A 13c)
Finally the Hx in the x direction becomes
curl e = E^ ~HiJ)-E Hij + )
n mdash 1
2
79
HT(iJ^)=mj^)H(ij^)
^^ l^curl e ^^r-si^^- bull U + i2(7 + - ) ^ = - + MO (27 + - )
We will now implement the PML for three directions the only difference being
we will be dealing with three directions instead of two [14]
So equation (A 12a) will now become
r JO)
a^(x) r 1 +
V J^^o J
O-D(gt) (
(
JO)
(
JO) V JO)So
1 + V Jo)poundo J
Y
A
1 + O-D(^) D=Cr
1 + go(j)
Jo)e
JOgtpoundo J
(
dH dH
D =Cr 1-F
V
1-F JO)Sbdquo
D = Cr curl h + Cr
^ dy dx
a(z)YdH dH^^
jo)poundo A ^ dx ^
o-(z)
(A14)
curl h (A 15)
= Jmdashcurl_h is an integration when it goes to the time domain and equation (A 15) JO)
now becomes
JO) Y
V Jogtpoundo ) Jo)So J D=Cr bdquo h+^^I
Dz ^o J
curl h mdash
Hy(i^fk^]^-H](i-fk^^-^
-H(ij^k^^^H(ij-k^--^^
(A 16)
Il(ifk^) = r ih j ^ + ) + curl _ h (A 17)
80
1 n+mdash
A HiJk + ^) = gi3(i)gj3(i)D Hijk + ^)
gi2(i)gj2(i) curl_h + gkl(k)Jl(ijk + -)
(A 18a)
The one dimensional gparameters are defined as
gm2(m) = 1
1 + fml(m) where m = i j k
- 1 - finl(m) gm3(m)^-mdashmdashmdashmdash where m = ijk
1 + fml(m)
Similarly
DT (f -F i j k) = gj3(j)gk3(k)Dr ( + i j k)
-F gk2(k)gj2(j) curl_h + gil(k)Jl(i + -Jk)
(A 18b)
D~HiJ + k) = gi3(i)gk3(k)DrHij + k)
+ ga(i)gk2(k) 1 ^
curl__h + gjliJ)JaiiJ + -k)
(A 18c)
Hi + jk) = fj3(j)fk3(k)H(i + jk)
+ fj2(j)fk2(k) curl_e + fil(i)jljii + -jk)
(A 19a)
81
KhJ + k) = fi3(i)fk3(k)H(ij + -k) bull 2
-F fi2(i)fk2(k) 1
curl _e + fjl(j)Jny i j + -k)
(A 19b)
Hf (i j c +1) = fi3(i)fj3(j)H a j k + i )
fi2(i)fj2(j) ( 1
cur _ e + fkl(k)Ibdquo^ ^ (i j k + -) (A 19c)
82
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