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LINEAR TECHNOLOGYLINEAR TECHNOLOGYLINEAR TECHNOLLINEAR TECHNOLOGOGYSEPTEMBER 1999 VOLUME IX NUMBER 3
Four New Amplifiers ServeMany Applications
, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,FilterCAD, Hot Swap, LinearView, Micropower SwitcherCAD, No Latency ∆Σ, No RSENSE, Operational Filter, OPTI-LOOP,PolyPhase, PowerSOT, SwitcherCAD and UltraFast are trademarks of Linear Technology Corporation. Other productnames may be trademarks of the companies that manufacture the products.
IN THIS ISSUE…Four New AmplifiersServe Many Applications ................. 1Issue Highlights .............................. 2LTC® in the News….......................... 2DESIGN FEATURESJFET Op Amps Equal Low NoiseBipolars and Have PicoampCurrent Noise .................................. 3Alexander Strong
LT®1813: 100MHz, 750V/µs AmplifierDraws Only 3mA ............................. 5George Feliz
Micropower, Precision Current SenseAmplifier Operates from 2.5V to 60V....................................................... 7
Richard Markell, Glen Brisebois andJim Mahoney
Triple 300MHz Current FeedbackAmplifiers Drive RGB/ComponentVideo and LCD Displays ................ 12Brian Hamilton
LTC1702/LTC1703 SwitchingRegulator Controllers Set a NewStandard for Transient Response..................................................... 16
Dave Dwelley
An 8-Channel, High-Accuracy,No Latency ∆Σ™ 24-Bit ADC ........... 21Michael K. Mayes
Micropower 5-Lead SOT-23 SwitchingRegulators Extend Battery Life inSpace-Sensitive Applications ........ 24Bryan Legates
Versatile Ring Tone Generator FindsUses in Motor Drivers and Amplifiers..................................................... 28
Dale Eagar
DESIGN IDEAS35 Watt Isolated DC/DC ConverterReplaces Modules at Half the Cost..................................................... 30
Robert Sheehan
Comparator Circuit ProvidesAutomatic Shutdown of the LT1795High Speed ADSL Power Amplifier..................................................... 33
Tim Regan
(More Design Ideas on pages 35–37;complete list on page 30)
New Device Cameos ....................... 38Design Tools ................................. 39Sales Offices ................................. 40
This issue of Linear Technologyfeatures a quartet of exciting newamplifiers from LTC: The LT1792/LT1793 low noise JFET op amps, theLT1813 high slew rate op amp, theLT1787 high-side current senseamplifier and the LT1399/LT1399HVtriple current feedback amplifiers.
The LT1792/LT1793 single JFETop amps offer both very low voltagenoise and low current noise, provid-ing the lowest total noise over a widerange of transducer impedances.These op amps are unconditionallystable for gains of one or more, evenwith capacitive loads of 1000pF. Theirlow offset voltage and high DC gainallow the LT1792/LT1793 to fit intoprecision applications, especiallythose involving high impedance,capacitive transducers.
The LT1813 is a 100MHz, 750V/µ sdual operational amplifier. Requiringonly 3mA of supply current, it usesLTC’s advanced, low voltage comple-mentary bipolar process and a fewdesign tricks to exceed the perfor-mance of its older siblings. A keyfigure of merit for amplifiers is theratio of gain bandwidth to supplycurrent (expressed as MHz/mA). Thenew process employed by the LT1813forsakes high supply voltage operationfor a 3×–4× increase in MHz/mA.Blazing speed from such a modestamount of supply current (3mA) isextremely attractive for low powerapplications. The LT1813 extends thefrequency response of applicationssuch as filters, instrumentationamplifiers and buffers.
The LT1787 employs precisiontechnology to build a superior high-side current sense amplifier. TheLT1787 will find uses in cellularphones, portable instruments andwireless telecom devices for preciselymonitoring the current into or out ofa battery. The LT1787 monitors bidi-rectional currents via the voltageacross an external sense resistor. Itfeatures a minute input offset voltageof 40µ V with a full-scale input of up to500mV. This translates to a dynamicrange of over 12 bits. The LT1787HVfeatures a 60V maximum input, whichallows it to be used in telecom andindustrial applications that requirethe sensing of higher voltages. Thedevice is self-powered from the sup-ply that it monitors and requires only60µA supply current.
For video and computer displayapplications, LTC introduces theLT1399/LT1399HV triple currentfeedback amplifiers. These devicescontain three independent 300MHzCFAs, each with a shutdown pin.Each CFA has 0.1dB gain flatness of150MHz and a slew rate of 800V/µs.The LT1399 operates on supplies from4V to ±6V. The LT1399HV operateson supplies ranging from 4V to ±7.5V.Each amplifier can be enabled in 30nsand disabled in 40ns, making themideal in spread-spectrum and por-table equipment applications. Withthe addition of a small series resistor,the parts can drive large capacitiveloads. This feature, combined withthe LT1399HV’s high voltage opera-tion, makes it ideal for driving LCDdisplays.
Linear Technology Magazine • September 19992
EDITOR’S PAGE
Issue HighlightsIn addition to the four new ampli-
fiers described on page one, this issueof Linear Technology features a vari-ety of power and data converterproducts.
In the power area, we introduce theLTC1702/LTC1703. The LTC1702 isthe first in a new family of low voltage,high speed switching regulator con-trollers. It is designed to operate fromthe standard 5V logic rail and gener-ate two lower voltage, high currentregulated outputs. Running at a fixed550kHz switching frequency, eachside of the LTC1702 features a volt-age feedback architecture using a25MHz gain-bandwidth op amp asthe feedback amplifier, allowing loop-crossover frequencies in excess of50kHz. Large onboard MOSFET driv-ers allow the LTC1702 to drive highcurrent external MOSFETs efficientlyat 550kHz and beyond, allowing theuse of small external inductors andcapacitors while maintaining excel-lent output ripple and transientresponse. The LTC1703 is a modifiedLTC1702 with a 5-bit DAC control-ling the output voltage at side 1. TheDAC conforms to the Intel Mobile VIDspecification.
Two other new switchers debutedin this issue are the micropowerLT1615 and LT1617. These devicescan be used in a number of topolo-gies, including boost, SEPIC andpositive-to-negative. With an inputvoltage range of 1.2V to 15V, they areideal for a wide range of applicationsand work with a variety of inputsources. An internal 36V switch al-lows them to easily generate outputvoltages of up to ±34V without theuse of costly transformers. TheLT1615 is designed to regulate posi-tive output voltages, whereas theLT1617 is designed to directly regu-late negative output voltages withoutthe need for level-shifting circuitry.Both parts use a current-limited, fixed
off-time control scheme, which helpsachieve high efficiency operation overa wide range of load currents. Bothdevices use tiny, low profile inductorsand capacitors to minimize the over-all system footprint and cost.
A new ∆Σ ADC is also featured inthis issue: the LTC2408 is the off-spring of the LTC2400 24-bit ∆Σconverter, introduced in our Novem-ber 1998 issue. The LTC2408combines the high performanceLTC2400 with an 8-channel analoginput multiplexer. The resulting de-vice offers many unique features. Thesingle-cycle settling characteristicslead to simplified multiplexer hook-up and channel selection, without theadded overhead seen with otherconverters. The exceptional noise per-formance of the device eliminates theneed for a programmable gain amp-lifier (PGA). The unprecedented 10ppmabsolute accuracy of the device allowsmeasurement of microvolt signalssuperimposed upon large DC volt-ages. A wide range of sensor inputsand voltage levels can be applied si-multaneously to the LTC2408.
Our final Design Feature spotlightssome unusual applications for theLT1684 ring tone generator, intro-duced in the June issue. The LT1684was specifically designed for OEMtelephone equipment. Because of theversatility of the device, the LT1684ring tone chip finds itself at home inmotor drives, digital input amplifiedspeakers, alarm systems and sinewave UPS systems.
Our Design Ideas section featuresan isolated 35 watt DC/DC converterdesigned to replace “half-brick” powermodules at half the cost and acomparator circuit that providesautomatic shutdown for the LT1795high speed ADSL power amplifier.
We conclude with a trio of NewDevice Cameos.
LTC in the News…On July 20, Linear Technology Cor-poration announced its financialresults for fiscal year 1999. RobertH. Swanson, Chairman & CEO,stated, “We had record levels ofsales and profits as we finished theyear strongly. As we commence ourfirst fiscal year in the new millen-nium, we are encouraged by theincreasing opportunity for highperformance analog circuits.” TheCompany reported net sales for thefourth quarter of $140,524,000 (a6% increase over net sales for thefourth quarter of the previous year).Net income for the fourth quarterwas $54,179,000 compared with$49,503,000 a year ago. For theyear, the Company’s net sales were$506,669,000 (up 5% over the pre-vious year), with net income of$194,293,000 (up 7% versus theprior year).
The Company was also featuredin “America’s Favorite Stocks,” anarticle in the July issue of Moneymagazine that detailed which stocksare most popular among invest-ment clubs in the United States.Linear Technology ranked 94th interms of number of clubs holdingthe stock, and 68th in terms of thevalue of Linear Technology stockheld by clubs.
Linear Technology was includedin Business Week’s annual Global1,000 rankings. The Companyranked 289th among U.S. compa-nies and 570th among companiesinternationally.
For more information on parts featured in this issue, seehttp://www.linear-tech.com/go/ltmag
For more information on parts featured in this issue, seehttp://www.linear-tech.com/go/ltmag
http://www.linear-tech.com/ezone/zone.htmlArticles, Design Ideas, Tips from the Lab…
Linear Technology Magazine • September 1999 3
DESIGN FEATURES
JFET Op Amps Equal Low NoiseBipolars and Have PicoampCurrent Noise by Alexander Strong
The LT1792 and LT1793 are singleJFET op amps that offer both very lowvoltage noise (4nV/√Hz for the LT1792and 6nV/√Hz for the LT1793) and lowcurrent noise (10fA/√Hz for theLT1792 and 0.8fA/√Hz for theLT1793), providing the lowest totalnoise over a wide range of transducerimpedance. Traditionally, op ampusers have been faced with a choice:which op amp will have the lowestnoise for the transducer at hand. Forhigh transducer impedance, theLT1792/LT1793 JFET op amps willwin over the lowest voltage noisebipolar op amps due to lower currentnoise. The current noise (2qIB) of anamplifier is a function of the inputbias current (IB). For lower trans-ducer impedance, bipolar op ampsusually win over typical JFET op ampsdue to lower voltage noise for thesame tail current of the differentialinput pair. The LT1792/LT1793 opamps are designed to have voltagenoise that approaches that of bipolarop amps. All of these op amps areunconditionally stable for gains ofone or more, even with capacitive
loads of 1000pF. The low offset volt-age of 250µ V and high DC gain of fourmillion allow the LT1792/LT1793 tofit into precision applications. Voltagenoise, slew rate and gain-bandwidthproduct are 100% tested. All of thespecifications are maintained in theSO-8 package.
The combination of low voltage andcurrent noise offered by the LT1792/LT1793 makes them useful in a widerange of applications, especially withhigh impedance, capacitive transduc-ers such as hydrophones, precisionaccelerometers and photo diodes. Thetotal noise in such systems is the gaintimes the square root of the sum ofthe op amp input–referred voltagenoise squared, the thermal noise ofthe transducer (4kTR) and the opamp’s bias current noise times thetransducer resistance squared (2qIB× R2). Figure 1 shows total inputvoltage noise versus source resistance.In a low source resistance application(<5k), the op amp’s voltage noise willdominate the total noise. In this regionof low source resistance, the LT1792/LT1793 JFET op amps are way aheadof other JFET op amps; only very lownoise bipolar op amps such as theLT1007 and LT1028 have the edge.As source resistance increases from5k to 50k, the LT1792/LT1793 willmatch the best bipolar or JFET opamp for noise performance, since thethermal noise of the transducer (4kTR)will dominate the total noise. A fur-ther increase in source resistance, toabove 50k, brings us to the regionwhere the op amp’s current noise(2qIB × RSOURCE) will dominate thetotal noise. At these high sourceresistances, the LT1792/LT1793 willoutperform the lowest noise bipolarop amp due to the inherently lowcurrent noise of FET input op amps.In some conditions it may be neces-
sary to add a capacitor in parallelwith a source resistor to cancel thepole that is caused by the sourceimpedance and the input capacitance(14pF for the LT1792 and 1.5pF forthe LT1793). Observe what happensto noise with source resistances over100k; the overall noise for the LT1792and LT1793 actually decreases.
The high input impedance JFETfront end makes the LT1792 andLT1793 suitable for applicationswhere very high charge sensitivity isrequired. Figure 2 illustrates theLT1792 and LT1793 in inverting andnoninverting modes of operation. Acharge amplifier is shown in theinverting mode example; here the gaindepends on the principle of chargeconservation at the input of theamplifier. The charge across the trans-ducer capacitance, CS, is transferredto the feedback capacitor, CF, result-ing in a change in voltage, dV, equal todQ/CF, resulting in a gain of CF/CS.For unity gain, the CF should equalthe transducer capacitance plus theinput capacitance of the amplifier andRF should equal RS. In the non-invert ing mode example, thetransducer current is converted to achange in voltage by the transducercapacitance; this voltage is then buff-
TOTAL SOURCE RESISTANCE (Ω)1001
10
1k
10k
1k 100M 1G
LT1169 • F02
100k
100
10M10k 1M
INPU
T NO
ISE
VOLT
AGE
(nV/
√Hz)
en = AV en2(OP AMP) + 4kTR + 2qIBR2
SOURCE RESISTANCE = 2RS = R* PLUS RESISTOR† PLUS RESISTOR 1000pF CAPACITOR
RESISTOR NOISE ONLY
LT1793
LT1007*
LT1007†
LT1792/†
LT1793
–
+
CS
CS
RS
RS
VO
LT1792
LT1007
LT1792*
(AND RESISTORNOISE)
LT1793*
retemaraP 2971TL 3971TL stinU
V SO )xaM( 65.0 37.0 Vm
IB )xaM( 054 01 Ap
eN )zHk1( 2.4 6 zH√/Vn
iN )zHk1( 01 8.0 zH√/Af
PWBGf( O )zHk001= 6 5 zHM
IS 2.4 2.4 Am
Table 1. LT1792/LT1793 specifications
Figure 1. Comparison of LT1792/LT1793 andLT1007 input voltage noise vs sourceresistance
Linear Technology Magazine • September 19994
DESIGN FEATURES
ered by the amplifier, with a gain of1 + R2/R1. A DC path is provided byRS, which is either the transducerimpedance or an external resistor.Since RS is usually several orders ofmagnitude greater than the parallelcombination of R1 and R2, RB is addedto balance the DC offset caused bythe noninverting input bias currentand RS. The input bias currents,although small at room temperature,can create significant errors overincreasing temperature, especiallywith transducer resistances of up to1000M or more. The optimum valuefor RB is determined by equating thethermal noise (4kTRS) to the currentnoise times RS, (2qIB) RS, resulting inRB = 2VT/IB (VT = kT/q = 26mV at25°C). A parallel capacitor, CB, is usedto cancel the phase shift caused bythe op amp input capacitance and RB.
The LT1792 has the lowest voltagenoise (4nV/√Hz) of the two, whichmakes it the best choice for trans-ducer impedances of 5k or less. Fortransducer impedances over 100M,the LT1793, with a typical input biascurrent of only 3pA, will have loweroutput noise than the LT1792. TheLT1793 has the additional advantageof very high input resistance (1013
ohms). Unlike most JFET op amps,the LT1792 and LT1793 have inputbias currents that remains almostconstant over the entire commonmode range. The specifications forthe LT1792 and LT1793 are summa-rized in Table 1.
The low noise of the LT1792 andLT1793 is achieved by maximizingthe gm of the input pair. The polygateJFETs have a higher gm-to-area ratiothan standard, single-gate JFETs.This is done by maximizing the tailcurrent and the size of the input JFETgeometeries. Forty percent of the totalsupply current is used as the tailcurrent for the LT1792 and LT1793.These op amps are best used withvery high impedance transducers. Thelow noise hydrophone amplifier inFigure 3 is an application where theLT1792 excels. The AC current out-put of the hydrophone is converted toa voltage output by the 100M inputresistor (R8). This signal is amplified
R2
OUTPUT
RB
CB
R1
CS RS CB ≅ CSRB = RSRS > R1 OR R2
TRANSDUCER
OUTPUT
CF
CBRB
CB = CF⎪⎪CSRB = RF⎪⎪RS
RF
CS RS
TRANSDUCER
–
+
–
+LT1792 LT1792
Q = CV; = I = CdQdt
dVdt
–
+
–
+
8
4
5V TO 15V
1
2
3
–5V TO –15V
C20.47µF
6
5
7
R8100M R6
100k
R41M
R51M
LT1792
1/2LT1464
R33.9k
C1*
R1*100M
R71M
R2200Ω
CTHYDRO-PHONE
OUTPUT
DC OUTPUT ≤ 2.5mV FOR TA < 70°COUTPUT VOLTAGE NOISE = 128nV/√Hz AT 1kHz (GAIN = 20)C1 ≈ CT ≈ 100pF TO 5000pF; R4C2 > R8CT; *OPTIONAL
by the R3/R2 ratio. DC leakage cur-rents at the output of the hydrophoneare subtracted by the servo action ofthe feedback amplifier. This amplifierneed not have the low voltage noise ofthe LT1792; therefore, it can be cho-sen to minimize the overall systemsupply current. The LT1464 has lessthan an order of magnitude of supplycurrent of the LT1792 and LT1793and picoampere input bias current.This allows the time constant of thisloop to be set using high value resis-tors and less expensive low valuecapacitors.
The LT1792 and LT1793 op ampsare in a class by themselves whenamplifying low level signals from highimpedance sources. The design andprocess have been optimized to pro-duce both low power consumptionand low current and voltage noise.Most competing JFET op amps willhave higher voltage noise or muchhigher supply current. Practically allbipolar op amps will have higher cur-rent noise. No other op amp will deliverthe noise performance for a givensupply current. For applicationswhere low noise and power are issues,the LT1792 and LT1793 are the bestchoices.
Figure 2. LT1792/LT1793 inverting and noninverting gain configurations
Figure 3. Low noise hydrophone amplifier with DC servo
Authors can be contactedat (408) 432-1900
Linear Technology Magazine • September 1999 5
DESIGN FEATURES
LT1813: 100MHz, 750V/µs AmplifierDraws Only 3mA by George Feliz
IntroductionThe LT1813 is a 100MHz dual opera-tional amplifier that has beenoptimized for supply voltages under12V. It features an easy-to-use voltagefeedback topology with high imped-ance inputs, yet it slews 750V/µ swith only 3mA supply current. DCperformance has not been neglected—the device has a 1.5mV maximumVOS and a 400nA maximum IOS.
PerformanceA summary of important specifica-tions of the LT1813, compared to itshigher voltage brethren, is shown inTable 1. A key figure of merit is theratio of gain bandwidth to supplycurrent (GBW/ISUPPLY, expressed inunits of MHz/mA). The new processemployed by the LT1813 forsakes highsupply voltage operation for a 3×–4×increase in MHz/mA compared to theLT1361 and LT1364. Blazing speedfrom such a modest amount of supplycurrent is extremely attractive for lowpower applications. The LT1813 alsopropagates the family traits ofmatched, high input impedance in-puts and low VOS, IB, IOS and inputnoise. The improved common modeinput range of the LT1813 adds to itsutility in low supply voltage applica-
tions. Stability with capacitive load-ing is another distinctive and desirablefeature. Although the LT1813 is notstable with unlimited capacitive loads,it is stable with nearly two orders ofmagnitude more capacitance thancompetitors’ high speed amplifiers.The small-signal transient response
in unity gain with CLOAD =100pF,500pF and 1000pF is shown inFigure 1.
The LT1813 extends the frequencyresponse of applications such as activefilters, instrumentation amplifiers andbuffers. Figure 2 shows the LT1813converting a single-ended signal to adifferential drive for the LTC141714-bit analog-to-digital converter(ADC). Note that the top amplifierprovides unity voltage gain, but theamplifier is configured in a noise-gainof 2 to match the phase response ofthe bottom amplifier, which has again of –1. The filter in front of theADC reduces broadband noise. Thespurious free dynamic range (SFDR)of this circuit is –79dB for a 425kHz,2VP-P input.
Circuit DesignA simplified schematic of the circuitis shown in Figure 3. The circuit lookssimilar to a current feedback ampli-fier, but both inputs are high
100pF
500pF
1000pF
100m
V/DI
V
200ns/DIV
3181TL 4631TL 1631TL
htdiwdnaBniaG zHM001 zHM05 zHM73
reifilpmAreptnerruCylppuS Am0.3 Am0.6 Am8.3
/WBG I YLPPUS Am/zHM3.33 Am/zHM3.8 Am/zHM7.9
etaRwelS sµ/V057 sµ/V054 sµ/V053
egnaRedoMnommoCtupnI V0.4± V2.3–,V4.3+ V2.3–,V4.3+
gniwStuptuO V0.4± V1.4± V0.4±tnerruCtuptuO Am06 Am54 Am83
V SO )xaM( Vm5.1 Vm5.1 Vm0.1
IB )xaM( Aµ0.4 Aµ0.2 Aµ0.1
I SO )xaM( An004 An053 An052
A LOV )niM( Vm/V5.1 Vm/V5.3 Vm/V3
egatloVesioNtupnI zH/Vn8 zH/Vn9 zH/Vn9
tneruCesioNtupnI zH/Ap1 zH/Ap1 zH/Ap9.0
C DAOL Fp0001 ∞ ∞(egatloVylppuSxaM V+ ot V–) V6.21 V63 V63
Table 1. Comparison of dual, high speed op amps (VS = ±5V, 25°C)
(VOUT = ±3V)
Figure 1. LT1813 in a gain-of-one configuration, no RL; CL = 100pF, 500pF or 1000pF
Linear Technology Magazine • September 19996
DESIGN FEATURES
impedance as in a traditional voltagefeedback amplifier. A complementarycascade of emitter followers, Q1–Q4,buffers the noninverting input anddrives one side of resistor R1. Theother side of the resistor is driven byQ5–Q8, which form a buffer for theinverting input. The input voltageappears across the resistor, generat-ing currents in Q3 and Q4 that aremirrored by Q9–Q11 and Q13–Q15into the high impedance node. Tran-sistors Q17–Q24 form the outputstage. Bandwidth is set by R1, thegm’s of Q3, Q4, Q7 and Q8 and thecompensation capacitor, CT.
The voltage drops of Q1–Q4 andthe diodes Q10 and Q14 set the inputcommon mode range of the amplifier.The emitters of Q3 and Q4 follow thenoninverting input. As the inputapproaches either supply rail, thelimiting voltage is determined by thesaturation of Q3 or Q4, which occursat approximately a VBE plus a VSATfrom the supply rail. Typically, theinput common mode range is 1V fromeither supply rail, and is guaranteedby the CMRR specification to be 1.5Vfrom either rail. This excellent inputrange is achieved without compro-mising the output impedance of themirrors Q9–Q11 and Q13–Q15,because Q25 and Q26 provide float-ing bias points for cascode devices Q9and Q13. Lower bandwidth processescannot successfully use this tech-
nique and maintain high bandwidth,due to phase shift in the mirror.
The current available to slew com-pensation capacitor CT is proportionalto the voltage that appears across R1.This method of “slew boost” achieveslow distortion due to its inherent lin-earity with input step size. Large slewcurrents can be generated withoutincreasing quiescent current. A lowvalue for R1 reduces the input noisevoltage to 8nV/√Hz and helps reduceinput offset voltage and drift. TheLT1813 is built with small-geometry,multi-GHz transistors that produceabundant bandwidth with meageroperating currents and allow for fur-ther reduction of idling supply current.
The output stage buffers the highimpedance node from the load by
–
+
–
+
1k
1k
1k
1k100Ω
100Ω
500pF
IN
LT1813
LT1813
LTC1417SERIALOUT14 BITS
V+
–IN
V–
Q5 Q6
Q7
Q8
R1
Q4
Q3
Q2Q1 +IN
Q10Q25
Q14Q26
Q13
Q15
Q11
Q9
Q12
Q17
CT
Q18
Q16
Q22
Q24
Q19
Q21Q20
Q23
RCCC
C1
C2
OUT
providing current gain. The simplestoutput stage would be two pairs ofcomplementary emitter followers,which would provide a current gain ofBetaNPN × BetaPNP. Unfortunately, thisgain is insufficient for driving evenmodest loads. Adding another emitter-follower or a Darlington configurationreduces output swing and createsinstability with large capacitive loads.
The solution used on the LT1813was to create a pair of compositetransistors formed by transistorsQ19–Q21 and Q22–Q24. The currentmirrors attached to the collectors ofemitter followers Q19 and Q22 pro-vide additional current gain. The ratioof transistor geometries Q20 to Q21and Q23 to Q24 increase the currentgain by approximately fifteen. There
Figure 2. Single-ended to differential ADC buffer: 2VP-P input at 425kHz yields –79dB SFDR
Figure 3. LT1813 simplified schematic
continued on page 15
Linear Technology Magazine • September 1999 7
DESIGN FEATURES
Micropower, Precision Current SenseAmplifier Operates from 2.5V to 60V
IntroductionThe LT1787 is a precision, high-sidecurrent sense amplifier designed formonitoring of the current either intoor out of a battery or other elementcapable of sourcing or sinking cur-rent. The LT1787 features a miniscule40µV (typical) input offset voltage witha 128mV full-scale input voltage. (Thepart is generally used at ±128mV full-scale, although it is specified for500mV minimum full-scale.) Thistranslates to a 12-bit dynamic rangein resolving currents. A hefty 60Vmaximum input voltage specificationallows the part to be used not only inlow voltage battery applications butalso in telecom and industrial appli-cations where higher voltages may bepresent.
The device is self-powered from thesupply that it is monitoring andrequires only 60µA of supply current.The power supply rejection ratio ofthe LT1787 is in excess of 130dB.
The LT1787 allows the use of auser-selectable sense resistor, thevalue of which depends on the cur-rent to be monitored. This allows theinput voltage to be optimized to the128mV value, maximizing dynamic
range. The part has a fixed voltagegain of eight from input to output.
Additional LT1787 features includeprovisions for input noise filtering(both differential and common mode)and the ability to operate over a verywide supply range of 2.5V to 60V. Thepart is available in both 8-lead SOand MSOP packages.
Operation of the LT1787Figure 1 shows a function diagram ofthe LT1787 integrated circuit. Whencurrent is flowing from VS
+ to VS–, a
sense voltage of VSENSE = ISENSE •RSENSE is generated. Because ampli-fier A1’s positive and negative inputsare forced equal by feedback, VSENSEalso appears across the RG side withthe higher VS potential. Hence, for thesituation where VS
+ is greater thanVS
–, VSENSE appears across RG2A andRG2B. This current flows through Q2and becomes the output current, IOUT.Q1 is kept off by the amplifier anddoes not contribute to IOUT.
For split-supply operation, whereVS
+ and VEE range from ±2.5V to±30V; with VBIAS at ground, VOUTbecomes (IOUT • ROUT). In this case,and for the above described directionof sense current, VOUT is positive or“above” VBIAS; thus Q2 is sourcingcurrent.
When current flows from VS– to
VS+, input VS
– is at a higher potentialthan VS
+, so VSENSE will appear acrossRG1A and RG1B. The current, IOUT =VSENSE/(RG1A + RG1B), will be conductedthrough Q1, while Q2 remains off.IOUT then duplicates itself through aone-to-one current mirror at VOUT.VOUT is negative or “below” VBIAS; Q1sources current and the mirror sinksthe current at the VOUT node.
The output voltage across ROUT isrelated to the input sense voltage bythe following relationships:
VOUT – VBIAS = IOUT • ROUT
VSENSE = ISENSE • RSENSE
IOUT = VSENSE/RG,RG = RG1A + RG1B = RG2A + RG2B
RG(TYP) = 2.5k
VOUT – VBIAS = (ROUT) (VSENSE)/RG, ROUT/RG = 8ROUT(TYP) = 20k
VOUT – VBIAS = 8 (VSENSE),VSENSE = VS
+ – VS–
VOUT = 8 (VSENSE) + VBIAS
Selection of RSENSEMaximum sense current can vary foreach application. To sense the widestdynamic range, it is necessary to selectan external sense resistor that fitseach application. 12-bit dynamicrange performance can be achievedregardless of the maximum current tobe sensed, whether it is 10mA or 10A.The correct RSENSE value is derived sothat the product of the maximumsense current and the sense resistorvalue is equal to the desired maxi-mum sense voltage (usually 128mV).For instance, the value of the senseresistor to sense a maximum currentof 10mA is 128mV/10mA = 12.8Ω.Since the LT1787 is capable of 12-bitresolution, the smallest measurablecurrent is 10mA/4096 counts =2.44µA/count. In terms of sense volt-age, this translates to 128mV/4096= 31.25µ V/count. Other currentranges can be accommodated by asimple change in value in the senseresistor. Care should also be taken toensure that the power dissipated inthe sense resistor, IMAX.
2 • RSENSE,does not exceed the maximum powerrating of the resistor.
by Richard Markell, Glen Brisebois and Jim Mahoney
RSENSE
1787 F 01
RG2
RG2A
RG1
RG1A
VOUT
VBIAS
ROUT
VS–
– +A1
Q1 Q2
1:1 MIRRORVEE
FIL–
VS+
FIL+
Figure 1. LT1787 function diagram
Linear Technology Magazine • September 19998
DESIGN FEATURES
Application Circuits
Dual-Supply,Bidirectional Current SenseFigure 2 shows the schematic dia-gram of the LT1787 operated withdual supplies. This circuit can sensecurrent in either direction, positivecurrent flow being from VS
+ to VS–,
where the output ranges from 0V to1.024V for VSENSE = 0mV to 128mV,
and negative current flow being fromVS
– to VS+, where the output varies
from 0V to –1.024V for VSENSE = 0mVto –128mV. Figure 3 shows the out-put voltage versus VSENSE for thisconfiguration.
Operation with BiasIf a negative supply is not available, avoltage reference may be connectedto the VBIAS pin of the LT1787. Thevalue of the reference is not critical, it
simply biases the output of the part toa new “zero” point (for VSENSE = 0V);zero is now at VBIAS = VREF, which, forthe case of Figure 4, is equal to 1.25V.This configuration can be used forboth unipolar and bipolar currentsensing, with VOUT ranging either“above” or “below” VBIAS, dependingon the direction of the current flow.This can be seen from the graph shownin Figure 5. (Note that the referencemust be able to both source and sink
1
2
3
4
8
7
6
5
C11µF
C21µF–5V
RSENSE
15V
OUTPUT
LT1787
FIL+
VS+
VBIAS
VOUT
FIL–
VS–
DNC
VEE
1000pF
TOCHARGER/
LOAD
Figure 2. Split-supply, bidirectional operation
1
2
3
4
8
7
6
5
C11µF
C21µF
RBIAS20k5%
RSENSE
3.3VTO60V 3.3V
LT1787HV
FIL+
VS+
VBIAS
VOUT
FIL–
VS–
DNC
VEE LT1634-1.251000pF
TOCHARGER/
LOAD
OUTPUT
Figure 4. Bidirectional operation with a reference
–
+
FIL–
VS+
DNC
VEE
8
2
6
5
FIL+
VS–
VBIAS
VOUT
1
7
3
4
LTC1404LT1495
LT1634-1.25
36k
100pF
1ΩIS
5V
VCC
VCC
LT1787
Figure 6. Operation with a buffer
SENSE VOLTAGE (VS+ – VS
–) (mV)
OUTP
UT V
OLTA
GE (V
)
1.5
1.0
0.5
0
–0.5
–1.0
–1.5–128 –64 0 32–96 –32 64 96 128
VS = ±1.65V TO ±30V TA = –40°C TO 85°C
Figure 3. VOUT vs VSENSE in bipolar mode
SENSE VOLTAGE (VS+ – VS
–) (mV)
OUTP
UT V
OLTA
GE (V
)
2.75
2.25
1.75
1.25
0.75
0.25
–0.25–128 –64 0 32–96 –32 64 96 128
VS = 3.3V TO 60V TA = –40°C TO 85°C
Figure 5. VOUT vs VSENSE in bipolar mode
4500
4000
3500
3000
2500
2000
1500
1000
500
0–200 –100 0 100
COUN
TS
INPUT CURRENT (mA)
SINGLE 5V SUPPLYVREF = 1.25VI–V CONVERTERUSING INTERNAL ROUT
–250 –150 –50 50
Figure 7. Output counts vs input currentfor Figure 6’s circuit
Linear Technology Magazine • September 1999 9
DESIGN FEATURES
current from the VBIAS pin—refer tothe block diagram in Figure 1.)
Operation with a BufferFigure 6 uses a rail-to-rail op amp,the LT1495, as an I/V converter tobuffer the LT1787’s output. TheLT1634-1.25 reference is used to biasthe LT1787’s output so that zero cur-rent is now represented by a 1.25Voutput. This allows the device to moni-tor current in either direction whilethe circuit operates on a single sup-ply. This also allows lower voltageoperation, since VOUT of the LT1787 isheld constant by the op amp. Figure7 shows input current versus outputcounts (from the LTC1404 A/D con-verter), showing excellent linearity.
Single-Supply Current SenseThe circuit in Figure 8 provides goodaccuracy near full-scale, but has alimited dynamic range. In this circuit,the LT1787 is operated from a singlesupply of 2.5V minimum to 60V maxi-mum. Current is allowed to flowthrough RSENSE in both directions but
is measured in a single direction only,with current flow from VS
+ to VS–. In
this connection, VBIAS and VEE aregrounded. The output voltage (VOUT )of the LT1787 for this circuit is equalto 8 • VSENSE, for VSENSE = VS
+ – VS– =
0mV to 128mV. The dynamic rangelimits of this circuit can be seen in thegraph shown in Figure 9.
Operation withan A/D ConverterFigure 10 shows the details of anLT1787 connected to a LTC1404 12-bit serial A/D converter. Details of thecircuit are similar to those shownpreviously in Figures 2 and 8 and inthe text detailing these circuits. Themain difference in the applications isthat the circuits in Figures 2 and 8provide an analog output voltage pro-portional to the current, whereas thecircuit shown in Figure 10 digitizesthat analog voltage to provide a digi-tal output. Figures 11 and 12 showthe output of the LTC1404 A/D con-verter. The data in Figure 11 wascollected with VEE operated from –5V;in other words, both the LT1787 andthe LTC1404 used a negative supplyof –5V. Similarly, the data in Figure12 was taken with VEE connected toground.
Connecting the optional section ofthe schematic (still operating the cir-cuit from a single supply) allows theA/D’s reference to “bias up” theLT1787 exactly as shown in Figure 5.Of course, the graph of the outputwould then be recast as similar toFigure 11 (counts versus VSENSE).
Auto ShutdownLinear RegulatorFigure 13 shows the details of a linearregulator with high-side currentsensing and latched shutdown capa-bility. The circuit shuts down powerto the load when the current reachesits overcurrent trip point. Power canthen be restored only by cycling themain power off and on again. Thiscircuit features the LT1787 and theLTC1440 precision comparator withon-chip reference. The LT1528 is a 3Alow dropout linear regulator withshutdown.
The circuit uses the LT1787, U1,as a precision current sensor; thegain of the LT1787 allows the use of a0.05Ω sense resistor, which dissi-pates a mere 0.312W of power. TheLTC1440 ultralow power compara-tor, U2, with its internal reference, isused as a precision trigger, followedby a 74HC00 connected as an RS flip-flop (U3C and U3D) to latch the errorcondition until power is removed andreapplied. The other two NAND gates
1
2
3
4
8
7
6
5
C0.1µF
RSENSE
VOUT
2.5V TO60V
TOLOAD
LT1787HV
FIL+
VS+
VBIAS
VOUT
FIL–
VS–
DNC
VEE
OU
TPU
T VO
LTAG
E (V
)
0.30
0.25
0.20
0.15
0.10
0.05
0IDEAL
VSENSE (mV)
0 5 10 15 20 25 30
FIL–
VS+
DNC
VEE
8
2
5
6
FIL+
VS–
VOUT
VBIAS
1
7
3
4LTC1404
1Ω 1%IS
VCC = 5V
VEE
VEE
LT1787
10k
10k
10µF16V
10µF16V
10µF16V
CONV
CLK
DOUT
CLOCKINGCIRCUITRY
OPTIONAL:CONNECT TO VBIAS;DISCONNECT VBIAS
FROM GROUND
7
6
5
1
8
2
3
4
AINVREF
GND
DOUT
VOUT (±1V)
~20k
1000
800
600
400
200
0
–200
–400
–600
–800
–1000–128 –96 –64 –32 0 32 64 96 128
VSENSE (mV)
COUN
TS
VCC = 5VVEE = –5V
Figure 8. Output voltage referred to ground—unidirectional sensing mode
Figure 9. Expanded scale of VOUT vs VSENSE,unidirectional current sensing mode
Figure 10. Connection to an A/D converter (current-to-counts converter) Figure 11. LT1787 input to LTC1404 ADC
Linear Technology Magazine • September 199910
DESIGN FEATURES
are used to provide a power-on resetof the RS flip-flop. The 1M resistor,R7, and the 0.33µ F capacitor, C3, atpin 2 of U3A provide a long enoughtime constant to properly initializethe flip-flop.
As shown, the circuit’s trip pointfor shutdown is just under 2.5A. Thismay be changed by altering the valueof current sense resistor R1. Consultthe LT1787 data sheet for details onhow to alter this resistor to sensedifferent current ranges.
Battery Fuel GaugeThe industry standard method forgauging battery charge is to keeptrack of the endpoints, the full chargeand discharged states, and, inbetween, to measure how much dis-charge has occurred since the lastfull charge (and vice versa). In anautomobile, this would be analogousto having only a “full” reading and an“empty” reading on the gas tank, and,in between, to keep track of mileage.Applying this strategy to batteries iscalled “Coulomb-counting”; it isachieved by measur ing (andnumerically accumulating in amicrocontroller) the current flow fromthe battery over time. Keeping trackof the history of battery currents andvoltages allows the present state ofthe battery charge to be determined.Hence the need for an accurate cur-rent sense amplifier like the LT1787.
Figure 14 shows a schematic formeasuring battery current using theLT1787 and measuring battery volt-age using the micropower LT1635with 200mV internal reference. TheLT1787 is configured for single-sup-ply, bidirectional operation, with the
2.0V reference coming from U2B, theLT1635 (created by amplifying its in-ternal 200mV reference by a gain often.) Note that the reference voltage isfed to the ADC, so its absolute valueis not critical, except in that it willform the center point for the batteryvoltage measurement and will thusdetermine the valid battery input volt-age range. Resistors R1 and R2 forma divide-by-five, bringing the batteryvoltage down from ~10.8V to ~2.1V toput it within the input range of adownstream ADC. Op amp U2A withresistors R3 and R4 level shift this bythe reference voltage and apply a gainof five. If a 12-bit ADC with a 5Vreference is used, the following equa-tions apply:
VBATT = 4 (VREF ) + VB= (5/4095) (4(Ch1) + Ch2)
IBATT = VS / RS= (1/8) (VI – VREF) / 0.05= 2.5 (VI – VREF)= 2.5 (5/4095) (Ch0 – Ch1)
where Ch0, Ch1 and Ch2 are in countsfrom 0 to 4095.
1000
800
600
400
200
00 20 40 60 80 100 120
VSENSE (mV)
COUN
TS
VCC = 5VVEE = –5V
–
+
VIN
SHDN
OUTPUT
SENSE
5
4
1
2
GND3
U4 LT1528
+
FIL–
VS+
DNC
VEE
8
2
5
6
FIL+
VS–
VOUT
VBIAS
1
7
3
4
R1 0.05Ω
U1 LT1787
VIN 5V C1 0.1µF R2
20k
R4 3.24k
R5 20kR32.4M
R6430Ω
C21µF
IN+
IN–
HYST
REF
V–
V+
OUT
GND
U2 LTC1440
C70.1µF
3
4
5
6
2
7
8
2
C40.01µF
D11N5712
5V
5V
5V
5V
R10180k
U3C
U3B
U3A
U3D
R810k
R71M
R910k
12
3
654
109
8
13
1211
VOUT3.3V/2.5A
C6100µF
R11330Ω
C51µF
LATCHED SHUTDOWN
POWER-ONRESET
U3 = 74HC00
C30.33µF
Figure 12. LT1787 input to LTC1404 ADC—single supply
Figure 13. LT1787 auto shutdown with latch
Linear Technology Magazine • September 1999 11
DESIGN FEATURES
Figure 15 shows a typical dischargeand charge cycle for a 10.8V, 4A-hourLi-Ion battery.
ConclusionThe LT1787 high-side current senseamplifier provides an easy-to-usemethod of sensing current with 12-bit resolution for a multiplicity ofapplication areas. The part can oper-
FIL–
VS–
DNC
VEE
8
7
5
6
FIL+
VS+
VOUT
VBIAS
1
2
3
4
RS 0.05Ω
U1 LT1787
C1 0.22µF
–
+
–
+
VBATT
SONY LIP902010.8V, 4AH
Li-Ion BATTERYSTAR GROUND
AT NEGATIVEBATTERY
TERMINAL
IBATT
LOAD CHARGE(12.5V MAX)
R9 1k ADC CH0VI
C60.22µF
C40.22µF
C50.22µF
R8 1kADC CH1VREF
5V
C2 100pF
R5 9.09k
U2B1/2 LT1635
1
7
8
4
R61k1%
U2A1/2 LT1635
6
3
2R31M1%
C3100pF
R4249k 1%
ADC CH2
VB
R11M1%
R2249k
1%
VBATT
200mV
R7 1k
SENSE VBATT AT THE POSITIVE BATTERYTERMINAL
– VS +
14
12
10
8
6
4
2
0
–2
–40 30 60 90 120 150 180
TIME (MINUTES)
VOLT
AGE
(V)
CURR
ENT
(A)
VBATT
IBATT
ate to 60V, making it ideal for highervoltage topologies such as might beused in telecom or industrial applica-tions. Additionally, the part can findhomes in battery-powered, handheldequipment and computers, where theneed for gauging the amount of cur-rent consumed and/or the amount ofcharge remaining in the battery iscritical.
Figure 14. Battery “fuel gauging” system
Figure 15. Discharge and charge cycleof a 10.8V Li-Ion battery
http://www.linear-tech.com/ezone/zone.htmlArticles, Design Ideas, Tips from the Lab…
For more information on parts featured in this issue, seehttp://www.linear-tech.com/go/ltmag
For more information on parts featured in this issue, seehttp://www.linear-tech.com/go/ltmag
Linear Technology Magazine • September 199912
DESIGN FEATURES
Triple 300MHz Current FeedbackAmplifiers Drive RGB/ComponentVideo and LCD Displays by Brian Hamilton
IntroductionWith the advent of HDTV and DVDvideo, there is a renewed focus onRGB/component video to maximizepicture quality. LCD displays havealso entered the mainstream for highend and portable computer displays.Both of these applications requirehigh speed triple amplifiers for rout-ing and conditioning video signals.LCD displays also require voltageswings of over 10V, with fast settling,into large capacitive loads. With theseapplications in mind, Linear Tech-nology has introduced the LT1399and LT1399HV triple current feed-back amplifiers. The LT1399 andLT1399HV contain three independent300MHz current feedback amplifiers,each with a shutdown pin. Eachamplifier has exceptional 0.1dB gainflatness of 150MHz and a slew rate of800V/µs. Output drive current is aminimum of 80mA over temperature.
The LT1399 operates on all sup-plies from a single 4V to ±6V. TheLT1399HV supports higher supplyvoltages and will operate on suppliesranging from a single 4V to ±7.5V.Each of the three amplifiers draws4.6mA when active. When disabled,
each amplifier draws zero supply cur-rent and its output becomes highimpedance. Each amplifier can beenabled in 30ns and disabled in 40ns,making the LT1399 and LT1399HVideal in spread-spectrum and por-table equipment applications.
With the addition of a small seriesresistor at the output, the LT1399and LT1399HV are capable of driv-ing large capacitive loads. TheLT1399HV’s high voltage operation,when combined with its ability todrive capacitive loads, makes it idealfor driving LCD displays.
What’s InsideFigure 1 is a simplified schematic forone of the three amplifiers found inthe LT1399/LT1399HV. Tying EN lowallows current to flow through J1,Q1–Q4 and R1. Q3 and Q6 mirror thiscurrent on top, while Q5 and Q7mirror the current on the bottom. Q6and Q7 thus act as current sourcesfor input-stage transistors Q8–Q11.+IN is a high-impedance input, driv-ing the bases of Q8 and Q9. Theemitters of these transistors then drivethe bases of Q10 and Q11, which
have their emitters tied together andform a buffered representation of +IN.This node is the inverting input –IN.Any current flowing into or out of –INmodulates the collector currents ofQ12 and Q14. This, in turn, modu-lates the collector currents of Q13and Q15, which drive the high imped-ance node. Transistors Q16–Q21 andresistors R2 and R3 form the outputstage that buffers the signal at thehigh impedance node from the output.
Using the LT1399HVto Drive LCD DisplaysDriving present-day XGA and UXGALCD displays can be a difficult prob-lem because they require drivevoltages of up to 12V, usually presenta capacitive load of over 300pF andrequire fast settling. The LT1399HVis particularly well suited for drivingthese LCD displays because it iscapable of swinging more than ±6Von ±7.5V supplies and, with a smallseries resistor RS at the output, candrive large capacitive loads with goodsettling time. This circuit topologycan be seen in Figure 2. As seen inFigure 3, at a gain of three, with a16.9Ω output series resistor and a330pF load, the LT1399HV is capableof settling to 0.1% in 30ns for a 6Vstep. Similarly, as seen in Figure 4, a12V output step will settle in 70ns.
EN
+IN –IN OUT
V+
V –
Q1
Q2
J1
Q3 Q12
Q14Q5
R1
Q4
Q6
Q8
Q9
Q7
Q10
Q11
Q13Q16
Q18
HI-Z
Q15
Q19
Q17
Q20
R2
R3
Q21
–
+VIN
RG
RF
RS
CLOAD
VOUT1/3
LT1399HV
Figure 1. LT1399 simplified schematic (one amplifier)Figure 2. Adding an output series resistor fordriving capacitive loads
Linear Technology Magazine • September 1999 13
DESIGN FEATURES
A 3-Input Video Multiplexerand Cable DriverFigure 5 shows a low cost, 3-inputvideo MUX cable driver. The scopephoto in Figure 6 displays the cableoutput of a 30MHz square wave driv-ing 150Ω. In this circuit, the activeamplifier is loaded by the sum of RFand RG of each disabled amplifier.Resistor values have been chosen tokeep the total back termination at75Ω while maintaining a gain of oneat the 75Ω load. Figure 7 shows theenvelope of the output signal as themultiplexer is switched from channelA to channel B. Channel A is beingdriven by a 2VP-P, 3.58MHz sine wave.The switching envelope of the outputis well behaved and the switchingtime is approximately 32ns.
Buffered RGB toColor-Difference MatrixTwo LT1399s can be used to createbuffered color-difference signals fromRGB inputs. In the application shownin Figure 8, a total of four amplifiersis used to create color-difference sig-nals. The luminance signal Y is createdusing amplifiers A2 and A3. Theremaining color-difference signalseach use a single amplifier and thenewly created Y output to perform theappropriate difference function.
The R input arrives via 75Ω coaxand is routed to 1082Ω resistor R8and the noninverting input of LT1399amplifier A1. There is also an 80.6Ωtermination resistor, R11, whichyields a 75Ω input impedance at theR input when considered in parallelwith R8. R8 connects to the inverting
input of a second amplifier, A2, whichalso sums the weighted G and B in-puts to create a –0.5 • Y output.Amplifier A3 then takes the –0.5 • Youtput and amplifies it by a gain ofminus two, resulting in the Y output.Amplifier A1 is configured in a nonin-verting gain-of-two configuration, withthe bottom of the gain resistor R2 tiedto the Y output. The output of amplifierA1 thus results in the color-differ-ence output R – Y.
The B input is similar to the Rinput. It arrives via 75Ω coax and isrouted to 2940Ω resistor R10 and thenoninverting input of amplifier A4.There is also a 76.8Ω terminationresistor R13, which yields a 75Ω input
impedance when considered in paral-lel with R10. R10 also connects to theinverting input of amplifier A2, add-ing the B contribution to the Y signalas discussed above. Amplifier A4 isconfigured in a noninverting gain-of-two configuration with the bottom ofthe gain resistor R4 tied to the Youtput. The output of amplifier A4thus results in the color-differenceoutput B – Y.
The G input also arrives via 75Ωcoax and adds its contribution to theY signal via a 549Ω resistor R9 that istied to the inverting input of amplifierA2. There is also an 86.6Ω termina-tion resistor, R12, which yields a 75Ωtermination when considered in par-
–
+1/3 LT1399RG
200Ω
RF324Ω
A
EN AVIN A
–
+1/3 LT1399RG
200Ω
RF324Ω
EN BVIN B
B CCHANNELSELECT
97.6Ω
97.6Ω
–
+1/3 LT1399RG
200Ω
RF324Ω
EN CVIN C
97.6Ω
75Ω
VOUT
75Ω CABLE
1399 TA01
VS = ±5VRF = 324ΩRG = 162ΩRS = 16.9ΩCL = 330pF
VIN1V/DIV
VOUT2V/DIV
20ns/DIV
Figure 3. LT1399/LT1399HV large-signal pulse responsedriving 330pF (typical LCD loading)
VS = ±7VRF = 324ΩRG = 162ΩRS = 16.9ΩCL = 330pF
VIN2V/DIV
VOUT5V/DIV
50ns/DIV
Figure 4. LT1399HV output-voltage swing is>12V with ±7V supplies
Figure 5. 3-input video mux/cable driver
Linear Technology Magazine • September 199914
DESIGN FEATURES
allel with R9. Using superposition, itis straightforward to determine theoutput of amplifier A2. Althoughinverted, it sums the R, G and Bsignals in the standard proportions of0.3R, 0.59G and 0.11B, which areused to create the Y signal. AmplifierA3 then inverts and amplifies thesignal by two, resulting in the Y out-put. Two additional LT1399 amplifiersremain unused, available for addi-tional signal conditioning as needed.
Buffered Color-Differenceto RGB MatrixThe LT1399 can also be used to cre-ate buffered RGB outputs fromcolor-difference signals. As seen inFigure 9, the R output is a back-terminated 75Ω signal created usingresistor R5 and LT1399 amplifier A1configured for a gain of two via 324Ωresistors R3 and R4. The noninvert-ing input of amplifier A1 is connectedvia 1k resistors R1 and R2 to the Yand R – Y inputs, respectively, result-ing in cancellation of the Y signal atthe amplifier input. The remaining Rsignal is then amplified by A1.
The B output is also a back-termi-nated 75Ω signal created usingresistor R16 and amplifier A3configured for a gain of two via 324Ωresistors R14 and R15. The non-inverting input of amplifier A3 isconnected via 1kΩ resistors R12 andR13 to the Y and B – Y inputs respec-tively, resulting in cancellation of theY signal at the amplifier input. Theremaining B signal is then amplifiedby A3.
The G output is the most compli-cated of the three. It is a weightedsum of the Y, R – Y and B – Y inputs.The Y input is attenuated via resis-tors R6 and R7 such that amplifierA2’s noninverting input sees 0.83Y.Using superposition, we can calcu-late the positive gain of A2 by assumingthat R8 and R9 are grounded. Thisresults in a gain of 2.41 and a contri-bution at the output of A2 of 2Y. TheR – Y input is amplified by A2 andresistors R8 and R10, giving a gain of–1.02. This results in a contributionat the output of A2 of 1.02Y – 1.02R.The B – Y input is amplified by A2 andresistors R9 and R10, giving a gain of–0.37. This results in a contributionat the output of A2 of 0.37Y – 0.37B.
If we now sum the three contribu-tions at the output of A2, we get:
A2OUT = 3.40Y – 1.02R – 0.37BIt is important to remember though,
that Y is a weighted sum of R, G, andB such that:
Y = 0.3R + 0.59G + 0.11B.If we substitute for Y at the output
of A2, we then get:
A2OUT = (1.02R – 1.02R) + 2G +(0.37B – 0.37B) = 2GThe back-termination resistor R16
then halves the output of A2, result-ing in the G output.
OUTPUT200mV/DIV
5nS/DIV
RL = 150ΩRF = RG = 340Ωf = 30MHz
EN A
EN B VS = ±5VVINA = VINB =2VP-P AT 3.58MHz
OUTPUT5V/DIV
5V/DIV
10ns/DIV
–
+A2
1/3 LT1399
–
+A3
1/3 LT1399
–
+A1
1/3 LT1399
R7324Ω
R6162Ω
R5324Ω
R102940Ω
R9549Ω
R1180.6Ω
R
G
B
R1286.6Ω
R1376.8Ω
ALL RESISTORS 1%VS = ±5V
R81082Ω
75ΩSOURCES
R1324Ω
R2324Ω
R4324Ω
R3324Ω
B – Y
Y
R – Y
–
+A4
1/3 LT1399
Figure 6. 30MHz square wave response Figure 7. 3-input video mux switching response
Figure 8. Buffered RGB to color-difference matrix
Linear Technology Magazine • September 1999 15
DESIGN FEATURES
–
+A2
1/3 LT1399R21k
B-Y
R-Y
Y
R10324Ω
R1175Ω
R6205Ω
R21k
R11k
R8316Ω
R9845Ω
–
+A3
1/3 LT1399R14324Ω
B
G
R1675Ω
R121k
R131k
R15324Ω
ALL RESISTORS 1%VS = ± 5V
–
+A1
1/3 LT1399R3324Ω
R
R575Ω
R4324Ω
–
+A1
1/3 LT1399
D11N4148
C14.7µF
D21N4148
R975Ω
R8324Ω
VS6V TO 12V
R7324ΩR5
2.32Ω
R475Ω
VIN
R21300Ω
R11000Ω
5V
R6107Ω
R3160Ω
VOUT
+
Single-Supply RGBVideo AmplifierThe LT1399 can be used with a singlesupply voltage of 6V or more to driveground-referenced RGB video. As seenin Figure 10, two 1N4148 diodes, D1and D2, have been placed in serieswith the output of the amplifier A1,but within the feedback loop formedby resistor R8. These diodes effec-tively level-shift A1’s output downwardby 2 diodes, allowing the circuit out-put to swing to ground.
Amplifier A1 is used in a positivegain configuration. The feedbackresistor R8 is 324Ω. The gain resistoris created from the parallel com-bination of R6 and R7, giving a
Thevenin-equivalent 80.4Ω connectedto 3.75V. This gives an AC gain of fivefrom the noninverting input of ampli-fier A1 to the cathode of D2. However,the video input is also attenuatedbefore arriving at A1’s positive input.Assuming a 75Ω source impedancefor the signal driving VIN, the Thev-enin-equivalent signal arriving at A1’spositive input is 3V + (0.4 • VIN), witha source impedance of 714Ω. Thecombination of these two inputs givesan output at the cathode of D2 of 2 •VIN with no additional DC offset. The75Ω back termination resistor R9halves the signal again such thatVOUT equals a buffered version of VIN.
It is important to note that the4.7µ F capacitor C1 is required tomaintain the voltage drop acrossdiodes D1 and D2 when the circuitoutput drops low enough that thediodes might otherwise be reversebiased. This means that this circuitworks fine for continuous video input,but will require that C1 be chargedafter a period of inactivity at the input.
ConclusionLinear Technology has introduced theLT1399 and LT1399HV triple 300MHzcurrent feedback amplifiers. Both ofthese products are well suited for usein component video applications. Thehigher supply voltage rating of theLT1399HV makes it an excellentchoice for LCD driver applications.Both products feature 4.6mA of sup-ply current per amplifier, 300MHz–3dB bandwidth, an exceptional0.1dB gain flatness of 150MHz, 800V/µ s slew rate and a shutdown pin foreach channel.
Figure 9. Buffered color-difference to RGB matrix Figure 10. Single-supply RGB video amplifier (one of three channels)
is no output swing penalty as theswing is limited at the collectors of Q9and Q13. The dynamics of the com-posites are not as benign as those ofemitter followers, so compensation isrequired and is provided by C1 andC2.
The stability with capacitive loadsis provided by the RC, CC networkbetween the output stage and the
LT1813, continued from page 6 gain node. When the amplifier is driv-ing a light or moderate load, the outputcan follow the high impedance nodeand the network is bootstrapped andhas no effect. When driving a heavyload such as a capacitor or small-value resistor, the network isincompletely bootstrapped and addsto the compensation provided by CT.The added capacitance provided byCC slows down the amplifier and the
zero created by RC adds phase marginto increase stability.
ConclusionThe combination of a high slew rate,DC accuracy and a frugal 3mA-per-amplifier supply current make theLT1813 a compelling choice for lowvoltage and low power, high speedapplications.
Linear Technology Magazine • September 199916
DESIGN FEATURES
LTC1702/LTC1703 Switching RegulatorControllers Set a New Standard forTransient ResponseIntroductionThe LTC1702 dual switching regula-tor controller uses a high switchingfrequency and precision feedback cir-cuitry to provide exceptional outputregulation and transient responseperformance. Running at a fixed550kHz switching frequency, eachside of the LTC1702 features a volt-age feedback architecture using a25MHz gain-bandwidth op amp asthe feedback amplifier, allowing loop-crossover frequencies in excess of50kHz. Large onboard MOSFET driv-ers allow the LTC1702 to drive highcurrent external MOSFETs efficientlyat 550kHz and beyond. The highswitching frequency allows the use ofsmall external inductors and capaci-tors while maintaining excellentoutput ripple and transient response,even as load currents exceed the 10Alevel. The dual-output LTC1702 ispackaged in a space-saving 24-pinnarrow SSOP, minimizing board spaceconsumed.
Mobile PCs using the most recentIntel Pentium® III processors requireLTC1702-level performance coupledwith a DAC-controlled voltage at thecore supply output. The LTC1703 isdesigned specifically for this applica-tion and consists of a modifiedLTC1702 with a 5-bit DAC control-ling the output voltage at side 1. TheDAC conforms to the Intel Mobile VID
specification. Figure 6 shows anexample of a complete mobile Pen-tium III power supply solution usingthe LTC1703. The LTC1703 is pack-aged in the 28-pin SSOP package,conserving valuable PC board realestate in cramped mobile PC designs.
LTC1702/LTC1703ArchitectureThe LTC1702/LTC1703 each consistof two independent switching regula-tor controllers in one package. Eachcontroller is designed to be wired as avoltage feedback, synchronous step-down regulator, using two externalN-channel MOSFETs per side aspower switches (Figure 1). A smallexternal charge pump (DCP and CCP inFigure 1) provides a boosted supplyvoltage to keep M1 turned fully on.The switching frequency is setinternally at 550kHz. A user-program-mable current limit circuit uses thesynchronous MOSFET switch, M2,as a current sensing element, elimi-nating the need for an external lowvalue current sensing resistor. TheLTC1702/LTC1703 are designed tooperate from a 5V or 3.3V input sup-ply, provided either by the mainoff-line supply in an AC powered sys-tem or a primary switching regulatorin battery powered systems. Maxi-mum input voltage is 7V.
Synchronous operation maximizesefficiency at full load, where resistivedrops in the switching MOSFET andthe synchronous rectifier dominatethe power losses. As the load dropsand switching losses become a largerfactor, the LTC1702/LTC1703 auto-matically shifts into discontinuousmode, where the synchronous recti-fier MOSFET turns off before the endof a switching cycle to prevent reversecurrent flow in the inductor. As theload current continues to decrease,the LTC1702/LTC1703 switchesmodes again and enters Burst Mode™,where it will only switch as requiredto keep the output in regulation, skip-ping cycles whenever possible toreduce switching losses to a bareminimum. With no output load inBurst Mode, the supply current forthe entire system drops to the 3mAquiescent current drawn by each sideof the LTC1702/LTC1703. Each sidecan be shutdown independently; withboth sides shut down, the LTC1702/LTC1703 enters a sleep mode whereit draws less than 50µA.
Inside the LTC1702/LTC1703The LTC1702/LTC1703 featurespeerless regulation and transientresponse, due to both to its highswitching frequency and a carefullydesigned internal architecture (Figure2). Much of the transient responseimprovement comes from a new feed-back amplifier design. Unlikeconventional switching regulatordesigns, the LTC1702/LTC1703 usea true 25MHz gain-bandwidth op ampas the feedback amplifier (FB in Fig-ure 2). This allows the use of anoptimized compensation scheme thatcan tailor the loop response moreprecisely that the traditional RC fromCOMP to ground. A “type 3” feedbackcircuit (Figure 3) typically allows the
by Dave Dwelley
+
TG
BOOST
SW
BG
PGND
PVCCDCP CIN
+COUT
VOUT
LEXT
VIN
QT
QB
CCP1µF
LTC1702/LTC1703
Figure 1. LTC1702/LTC1703 switching architecture
Pentium is a registered trademark of Intel Corp.
Linear Technology Magazine • September 1999 17
DESIGN FEATURES
loop to be crossed over beyond 50kHzwhile maintaining good stability, sig-nificantly enhancing load transientresponse. Two additional high speedcomparators (MIN and MAX in Figure2) run in parallel with the main feed-back amplifier, providing virtuallyinstantaneous correction to suddenchanges in output voltage. In a typi-cal application, the LTC1702/LTC1703 will correct the duty cycleand have the output voltage headedback in the right direction the verynext switching cycle after a transientload is applied.
The positive input of the feedbackop amp is connected to an onboardreference trimmed to 800mV ±3mV.DC output error due to the referenceand the feedback amplifier are inside0.5% and DC load and line regulationare typically better than 0.1%, pro-viding excellent DC accuracy. The800mV reference level allows the
LTC1702/LTC1703 to provide regu-lated output voltages as low as 0.8Vwithout additional external compo-nents. This reference performance,combined with the high speed inter-nal feedback amplifier and properlychosen external components, allowsthe LTC1702 to provide output regu-lation tight enough for virtually anymicroprocessor, today or in the future.For those Intel processors that don’tknow what voltage they want untilthey actually get powered up, theLTC1703 with its onboard 5-bit VIDoutput voltage control is the bestsolution.
Another architecture trick insidethe LTC1702/LTC1703 reduces therequired input capacitance with vir-tually no performance penalty. TheLTC1702/LTC1703 includes a singlemaster clock, which drives the twosides such that side 1 is 180° out ofphase from side 2. This technique,
known as 2-phase switching, has theeffect of doubling the frequency of theswitching pulses seen by the inputcapacitor and significantly reducingtheir RMS value. With 2-phase switch-ing, the input capacitor is sized asrequired to support a single side atmaximum load. As the load increasesat the other side, it tends to cancel,rather than add to, the RMS currentseen by the input capacitor; hence,no additional capacitance needs to beadded.
External ComponentsThe other half of the performanceequation is made up by the externalcomponents used with the LTC1702/LTC1703. The 550kHz clock frequencyand the low 5V input voltage allow theuse of external inductors in the 1µHrange or lower (LEXT in Figure 1) whilestill keeping inductor ripple currentunder control. This low inductance
BURSTLOGIC
SOFTSTART
90% DC
GAIN
RUN/SS
COMP
500mV
800mV 760mV 840mV
IMAX
DRIVELOGIC
100µsDELAY
OSC550kHz
+ –FB MIN MAX
920mV
FLT
DISFCB
FB
1702 BD
BOOST
TG
FROMOTHERHALF
SD TO THISHALF
SD TO ENTIRE CHIP
FAULT
PVCC
10µsDELAY
FROMOTHERHALF
SW
BG
PGND
PGOODD
40k
VID0
VCC
40k
VID1
VCC
40k
VID2
VCC
40k
VID3
VCC
40k
VID4
VCC
R11
RB1
SENSE
FB1SWITCH
CONTROLLOGIC
LTC1703 ONLY
ILIM
Figure 2. LTC1702/LTC1703 block diagram
Linear Technology Magazine • September 199918
DESIGN FEATURES
value helps in two ways: it reducesthe energy stored in the inductor dur-ing each switching cycle, reducingthe physical core size required; and itraises the attainable di/dt at the out-put of the circuit, decreasing the timethat it takes for the circuit to correctfor sudden changes in load current.This, in turn, reduces the amount ofoutput capacitance (COUT in Figure 1)required to support the output volt-age during a load transient. Togetherwith the reduced capacitance at theinput due to the LTC1702/LTC1703’s2-phase internal switching, thissignificantly reduces the amount oftotal capacitance needed, comparedto a conventional design running at300kHz or less.
Each side of an LTC1702/LTC1703circuit requires a pair of N-channelpower MOSFETs to complete thepower switching path. These are cho-sen for low RDS(ON) and minimum gatecharge, to minimize conductive losseswith heavy loads and switching lossesat lighter loads. MOSFET types thatwork well with the LTC1702/LTC1703include the IRF7805 from Interna-tional Rectifier, the Si9802 and Si9804from Siliconix and the FDS6670A fromFairchild.
The compensation componentsround out the list of external partsrequired to complete an LTC1702/LTC1703 circuit. Because theLTC1702/LTC1703 uses an op ampas the feedback amplifier, the com-pensation network is connectedbetween the COMP pin (at the output
of the op amp) and the FB pin (theinverting input) as a traditional opamp integrator (Figure 3). A biasresistor is added to set the DC outputvoltage and two pole/zero pairs areadded to the circuit to compensate forphase shift caused by the inductor/output capacitor combination. Cur-rent limit and soft start time for eachside are programmed with a singleresistor (RIMAX) at each IMAX pin and asingle capacitor (CSS) at each RUN/SS pin. Optional FAULT (LTC1702/LTC1703) and PWRGD (LTC1702only) flags are available to providestatus information to the host system.
Applications
Dual Outputs from a 5V SupplyA typical LTC1702 application isshown in Figure 4. The input is takenfrom the 5V logic supply. Side 1 is setup to provide 1.8V at 10A and side 2is set to supply 3.3V at a lower 3A loadlevel. System efficiency peaks atgreater than 90% at each side. Thiscircuit shows examples of both highpower and lower power output designspossible with the LTC1702 controller.Side 1 uses a pair of ultralow RDS(ON)Fairchild FDS6670A SO-8 MOSFETs
0.8V
VOUT
RB
COMP
–
+FB
FB
C2
C3
C1R2
R1
R3
BOOST1
TG1
SW1
BG1
IMAX1
COMP1
FB1
RUN/SS1
BOOST2
TG2
SW2
BG2
IMAX2
COMP2
FB2
RUN/SS2
LTC1702
VCC PVCC
GND PGND
+ +
+
VOUT11.8V/10A
COUT1470µF
×2
R1110k
0.1%
C11330pF
RB17.96k0.1%
C21680pF
C31560pF
LEXT11µH 12A
LEXT22.2µH 6A
R31 4.7k
R21 13k
QT1
QB1
DCP2MBR0530T
DCP1MBR0530T
CCP11µF
CCP21µF
1µF
D2 MBR330T
RIMAX1 22k
CSS10.1µF
CSS20.1µF
10Ω
CIN470µF×2
QT2
QB2
VOUT23.3V/3A
COUT2470µF
C12120pF
C22270pF
R22 20k
RB21.62k0.1%
R12 4.99k
0.1%C32820pF
R32 2.2k
QT1, QB1: FAIRCHILD FDS6670A (207) 775-4502QT2, QB2: 1/2 SILICONIX Si9802 (800) 544-5565
RIMAX2 47k
VIN 5V
LEXT1:LEXT2:
MURATA LQT12535C1ROM12 (814) 237-1431COILTRONICS UP2B-2R2 (561) 241-7876
Figure 3. Type-3 feedback loop
Figure 4. Dual outputs from a 5V supply
Linear Technology Magazine • September 1999 19
DESIGN FEATURES
and a large 1µ H/12A Murata surfacemount inductor. CIN consists of two470µF, low ESR tantalum capacitorsto support side 1 at full load, andCOUT1 uses two more of the same toprovide better than 5% regulationwith 0A–10A transients.
Side 2 uses a single SO-8 packagewith two smaller MOSFETs inside(the Siliconix Si9402) and a smaller2.2µ H/6A inductor. COUT2 is a single
470µF tantalum to support 0A–3Atransients while maintaining betterthan 5% regulation. As the load cur-rent at side 2 increases, the LTC17022-phase switching actually reducesthe RMS current in CIN, removing theneed for additional capacitance at theinput beyond what side 1 requires.Both sides exhibit exceptional tran-sient response (Figure 5). The entirecircuit can be laid out in less than 2square inches when a double-sidedPC board is used.
2-Step Converter forNotebook ComputersFigure 6 is a complete power supplyfor a typical notebook computer usingthe next generation of Intel mobilePentium III processor. The circuit usesthe LTC1628 to generate 5V and 3.3Vfrom the input battery and uses theLTC1703 to generate the processorcore voltage (with 5-bit VID control)
and the CPU I/O ring supply voltage.Both the LTC1628 and the LTC1703use 2-phase switching to minimizecapacitance required by the circuit;the entire 4-output circuit requiresbarely 2000µ F while generating 60Wof output power.
The 2-step conversion used in thiscircuit provides improved transientresponse compared to the traditionalsingle-step approach where each
SIDE 1VOUT = 1.8VVIN = 5VILOAD = 0A–10A–0A2% MAX DEVIATION
SIDE 2VOUT = 3.3VVIN = 5VILOAD = 0A–3A–0A2.2% MAX DEVIATION
2-Step ConversionAs microprocessor operating volt-ages continue to decrease, powerconversion for CPU core power isbecoming a daunting challenge. Acore power supply must have fasttransient response, good efficiencyand low heat generation in the vicin-ity of the processor. These factorswill soon force a move away from1-step power conversion directlyfrom battery or wall adapter to pro-cessor, to 2-step conversion, wherethe CPU core power is obtained fromthe 5V or 3.3V supply.
Several benefits result from 2-stepconversion: more symmetrical tran-sient response, lower heat generationin the vicinity of the processor andeasy modification for lower proces-sor voltages in the future. Peakcurrents taken from the battery arealso reduced, which leads toimproved battery chemical efficiencythat can often compensate for theslight difference in electrical effi-ciency measured using laboratorypower supplies. Battery life in a realnotebook computer is virtuallyidentical for 1-step and 2-steparchitectures.
The duty cycle for a step-downswitching regulator is given by theratio of VOUT to VIN. In 1-step powerconversion, the duty cycle must bevery low because the step-down ratiois large. This gives a very fast inductorcurrent rise time and a much slowercurrent decay time. The inductor sizemust be large enough to keep thecurrent under control during theramp-up. Fast current rise and slowcurrent decay mean that the tran-sient response of the regulator is goodfor load increases but poor for loaddecreases. The lower, constant inputvoltage for a 2-step conversion pro-cess yields a more symmetricaltransient response and allows smaller,lower cost external components to beused. Because there is less switchingloss due to the lower voltage swings,the switching frequency may also beincreased.
Thermal concerns are also easedwith the 2-step approach. To mini-mize high current PCB trace lengths,the core supply must be located nearthe processor. Core-voltage-level1-step converters usually run at mid-
80% efficiencies, while the secondstep of a 2-step solution (like theLTC1703) runs near 90% efficiency,minimizing heat generation near theprocessor.
The biggest argument against2-step conversion is the perceiveddrop in efficiency. “Off the cuff” cal-culations give a false impressionthat efficiency decreases. In fact,accurate calculations of efficiencyfor 2-step power conversion basedon actual circuit measurementsshow efficiency numbers within 1%of 1-step, high efficiency converters.As time goes forward, microproces-sor fabrication lithography willcontinue to shrink and force stilllower CPU core operating voltagesand higher operating currents; 1.1Vsupplies and 15A operating cur-rents are already on the horizon forportable systems. These demandswill render the traditional 1-stepconversion approaches unworkableas a result of infinitesimal duty cyclesand severely skewed transientbehavior.For more information on 2-step conversion,see www.linear-tech.com/ezone/2-step.html
Figure 5a. Transient response, side 1 Figure 5b. Transient response, side 2
Linear Technology Magazine • September 199920
DESIGN FEATURES
1
2
3
4
5
6
7
8
9
10
11
12
13
14
28
27
26
25
24
23
22
21
20
19
18
17
16
15
PVCC
BOOST1
BG1
TG1
SW1
IMAX1
FCB
RUN/SS1
COMP1
SGND
FB1
SENS
VID0
VID1
IMAX2
BOOST2
BG2
TG2
SW2
PGND
FAULT
RUN/SS2
COMP2
FB2
VCC
VID4
VID3
VID2
LTC1703
++
+
VOUT41.5V12A
1µF 180µF, 4V× 6
L3, 0.8µHETQP6F0R8L
D5MBRD-835L
QB1AIRF7811
QB1BIRF7811
QT1AIRF7811
QT1BIRF7811 D6
MBR0520LT1
1µF
18.7k, 1%
R18, 1M
0.22µF
200pF 15pF100k
220pF10k
1µF 24.9k, 1% D7MBR0520LT1
QT2NDS8926
QB2NDS8926
L4 2.2µFDO3316P-222
1µF150µF6V× 2
POINT A
L5, 0.33µHDO3316P-331HC
VOUT31.8V3A180µF
4V
8.06k1%
10.2k1%
2200pF
10Ω
1k
1µF
100k
15pF 220pF
0.22µF
1µF
VID0VID1VID2VID3VID4COREVENABLE
1.8VENABLEFAULT
0.1µF
NOTE: PLACE LTC1703 CLOSE TO PROCESSOR
1
2
3
4
5
6
7
8
9
10
11
12
13
14
28
27
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RUN/SS1
SENSE1+
SENSE1–
VOSENSE1
FREQSET
STBYMD
FCB
ITH1
SGND
3.3VOUT
ITH2
VOSENSE2
SENSE2–
SENSE2+
FLTCPL
TG1
SW1
BOOST1
VIN
BG1
EXTVCC
INTVCC
PGND
BG2
BOOST2
SW2
TG2
RUNSS2
LTC1628
+
+
+
5VENABLE
STBYMD
STDBY3.3V
STDBY5V
3.3VENABLE
0.1µF
0.1µF
0.1µF
330pF 330pF
33k
56pF
56pF
33k
1000pF
0.1µF
0.1µF
D2 CMDSH-3
1µF
0.01µF
4.7µF
D1 CMDSH-3
0.22µF
Q5IRF7807
Q4IRF7807
D4MBRS130T3
L24.6µH
ETQP6F4R6H
D3MBRD835L
Q3IRF7805
Q2IRF7805
Q1IRF7805
L12.9µH
ETQP6F2R9L
0.1µF50V
0.004Ω
150µF6V
× 2
180µF4V
0.1µF50V
22µF50V
VIN7V TO20V
VOUT15V4A
105k1%
20.0k1%
20.0k1%
63.4k1%
100pF
100pF
47pF
47pF
10µF6.3V
10µF6.3V
VOUT23.3V5A
0.1µF50V
1000pF
TO POINTA
0.01Ω
voltage is derived directly from thebattery voltage. 2-step also allows theuse of smaller external componentswithout paying an efficiency or per-formance penalty and it eases layoutand thermal management concerns.See the “2-Step Conversion” sidebarfor more information.
ConclusionThe LTC1702 and LTC1703 achieveDC and AC regulation performancethat tops the best switching regulatorcontrollers available today. As logicdensities continue to climb, moreapplications are appearing where theinput voltage is limited to below 7Vand the output voltage is low, theoutput current is high and multiple
outputs are required. The LTC1702and LTC1703 provide the best com-bination of regulation performance,high efficiency, small size and lowsystem cost for such applications,whether they appear in advancednotebook computers or complex logicsystems.
Figure 6. 4-output notebook computer power supply
D1–D7: MOTOROLA(800) 441-2447Q1–Q5, QT1A/1B, QB1A/1B:INTERNATIONAL RECTIFIER(310) 322-3331QT2, QB2: FAIRCHILD(207) 775-4503L1–L3: PANASONIC(201) 348-7522L4, L5: COILTRONICS(561) 241-7876
Linear Technology Magazine • September 1999 21
DESIGN FEATURES
An 8-Channel, High-Accuracy,No Latency ∆∆∆∆∆ΣΣΣΣΣ 24-Bit ADC by Michael K. Mayes
IntroductionRecently, Linear Technology intro-duced the world’s most accurate,simplest to use, 24-bit analog-to-digi-tal converter, the LTC2400. With itson-chip oscillator, 120dB line-fre-quency rejection, user-transparentoffset/full-scale calibration, 10 parts-per-million (ppm) total unadjustederror and 1.5µ VRMS noise, theLTC2400 has become a key buildingblock in many system designs. TheLTC2400’s ease of use and high per-formance enable faster design cyclesand better performance than other∆Σ converters.
This article introduces theLTC2408, a device combining the highperformance LTC2400 ADC core withan 8-channel analog input multiplexer(see Figure 1). This device offers manyunique features. The single-cycle set-tling characteristics lead to simplifiedmultiplexer hook up and channelselection, without the added overheadrequired other converters. The excep-tional noise performance of the deviceeliminates the need for a program-mable gain amplifier (PGA). This allowsdirect digitization of a variety of volt-age levels. Its 10ppm absoluteaccuracy ensures a minimum perfor-
mance in excess of 16 bits. A uniqueanalog modulator implementation al-lows measurement of microvoltsignals superimposed upon large DCvoltages. A wide range of sensor inputsand voltage levels can be appliedsimultaneously to the LTC2408. Thesesignals can extend below ground,above VCC or anywhere in between,with the same 10ppm absolute accu-racy.
Single-Cycle SettlingEnsures No LatencyMany applications requiring 16-bit to24-bit resolution use delta-sigma (∆Σ)ADCs. These applications typicallymeasure slow-moving signals, suchas those found in temperature mea-surement, weight scales, strain-gagetransducers, gas analyzers, batterymonitoring circuits and DVMs. Oneadvantage delta-sigma converterarchitectures offer over conventionalADCs is on-chip digital filtering. Forthe low frequency applicationsdescribed above, this filter is designedto provide rejection of line frequenciesat 50Hz or 60Hz and their harmonics.
A disadvantage of conventionaldigital filters, prior to the release of
the LTC2400, was digital filter set-tling time. If the input signal changesabruptly, the conversion result isinvalid for the following 3–4 conver-sion cycles (see Figure 2a). This makesmultiplexing the input difficult. TheLTC2400 does not exhibit a filter set-tling time; hence, it is easy to multiplex(see Figure 2b); There is a one-to-onecorrespondence between the conver-sion result and the applied inputsignal. Each conversion result isindependent from the previous con-version result. The 10ppm total erroris maintained for each conversioncycle, even in the extreme case ofsequentially measuring 0V and 5V onadjacent channels.
The Advantages ofNot Using a ProgrammableGain Amplifier (PGA)The exceptional noise performance ofthe LTC2408 (1.5µVRMS) correspondsto an effective resolution of 21.6 bitsfor a 5V input range. Low level inputsignals within a 100mV range achievebetter than 16-bit effective resolutionwithout the use of a PGA. On theother hand, conventional ∆Σ ADCsare significantly noisier than theLTC2408 for a 5V input range. Theseconverters require internal PGAs inorder to improve the noise perfor-mance for low level input voltages.
The LTC2408 offers several sig-nificant advantages over thoseconverters requiring a PGA. Oneadvantage the LTC2408 offers is theability to measure small signals(microvolts) superimposed upon largeDC voltages (volts). For example (seeFigure 3), a 100mV signal sitting on2V (2V to 2.1V) can be measured withthe same accuracy and noise perfor-mance as a 100mV signal sitting onground (0V to 0.1V). Conversely, anADC operating with a programmablegain of 50 is limited to an input rangeof 0V to 0.1V with a 5V reference (see
CH0
CH1
CH2
CH3
CH4
CH5
CH6
CH7
8-CHANNELMUX
24-BIT ∆-Σ ADC
CSADC
CSMUX
CLK/SCK
DIN
SDO
FO
ADCINMUXOUT VREF+
VREF–
LTC2408
9
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12
13
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15
17
23
20
19, 25
21
24
26
7 4 3
2.7V–5.5V
2, 8
1µF
6GND
1, 5, 16, 18 22, 27, 28
VCC
Figure 1. LTC2408 block diagram
Linear Technology Magazine • September 199922
DESIGN FEATURES
Figure 3). It cannot digitize any signallarger than 100mV full-scale.
A second advantage the LTC2408offers is full-scale accuracy. Since thetotal unadjusted error is less than10ppm, the absolute accuracy of anyinput voltage within the 0V to 5Vrange is within 10ppm or 16 bits.Alternatively, devices using PGAsexhibit full-scale errors limited by thematching of internal components. Theuser is burdened with removal of theseerrors. The user must first apply thesystem’s full-scale voltage to the deviceand then perform a system calibra-tion.
The use of a PGA in conventional∆Σ adds complexity. Each channelrequires a system full-scale and off-set calibration. Each channel mayhave a different PGA gain and input-signal range settings, correspondingto different offset and full-scale cali-bration coefficients. This requires
programming and maintaining con-figuration/status registers, gain/offset registers and channel/PGA-gainregisters. The LTC2408 does not re-quire any registers. The offset andfull-scale error corrections are per-formed during each conversion cycleand are transparent to the user.
“Microvolts on Volts”In order to measure a small levelsignal (microvolts) superimposedupon a large signal (volts), the con-verter must exhibit extremely goodDC performance. The device musthave very low offset and full-scaleerrors and excellent linearity perfor-mance in order to accurately digitizesmall signals with large fixed DC lev-els. Additionally, the temperaturecoefficients of offset, full-scale andlinearity errors must be low. TheLTC2408’s offset error is less than1ppm and its offset drift is less than
N N+1
CONVERT CONVERT CONVERT CONVERT CONVERT
N XXXX XXXX XXXX N+1
VIN
DATA OUT
INVALID DATAOR INDETERMINATE RESULTS
N N+1 N+2
CONVERT N+1 CONVERT N+2
N N+1 N+2
VIN
CS
SDO(VALID
DATA OUT)
ADCOPERATION
0.01ppm/°C (see Figure 4a). The full-scale error is less than 4ppm while itsdrift is less than 0.02ppm/°C (seeFigure 4b). Combined with an inte-gral nonlinearity error of 4ppm, theLTC2408 can consistently resolve lowlevel signals in the microvolt range,regardless of the fixed DC level (withinthe 0V to VREF range).
The accuracy, noise performance,and temperature stability of theLTC2408 enable the converter tomeasure input signals from a multi-tude of sensors (see Figure 5). Inaddition to the LTC2408’s ability tomeasure signals from 0 to VREF, thedevice also has overrange/underrangecapabilities. The device can measurean input signal 100mV below groundand 100mV above VREF, even if VREF isequal to VCC.
A Simple 4-WireSPI InterfaceInterfacing to the LTC2408 is simple.The individual CS and CLK signals(see Figure 6) can be common to boththe ADC and multiplexer or drivenindependently to allow separate con-
0V
2V
VREF (5V)
0V
100mV
0V
100mV
0V
100mV
USE UPPER 8 BITSTO DETERMINE RANGEV I
N
0V
VREF (5V)
0V
100mV
INFORMATIONNOT AVAILABLEV I
N
5.0
2.5
0
–2.5
–5.0–55 –30 –5 20 45 75 95 120
TEMPERATURE (°C)
FULL
-SCA
LE E
RROR
(PPM
)
VCC = 5VVREF = 5VVIN = 0V
5.0
2.5
0
–2.5
–5.0–55 –30 –5 20 45 75 95 120
TEMPERATURE (°C)
FULL
-SCA
LE E
RROR
(PPM
)
VCC = 5VVREF = 5VVIN = 5V
Figure 2a. Effect of conventional digital filter settling time
Figure 2b. The LTC2408 has no digital filter settling time.
Figure 3. Full range without PGA (left); limited range with PGA (right)
Figure 4a. Offset error drift
Figure 4b. Full-scale error drift
Linear Technology Magazine • September 1999 23
DESIGN FEATURES
–
+5.9µV/°C
40.6µV/°C
5V ±µV
VREF
VREF
12k
100Ω
350Ω
350Ω
VIN10k
BIPOLAR SENSOR±100mV
VCC TO VCC + 100mV
CH0
CH1
VREF–
CH2
CH3
CH4
CH5
CH6
CH7
MICROVOLTS AROUND 0V(WITH FIXED GAIN TO REDUCE NOISE)
MICROVOLTS AROUND VCC
COLD-JUNCTION TEMPERATUREMEASUREMENTS (MICROVOLTS)
MICROVOLTS AT MIDSCALE
INPUT VOLTAGE—10k SERIES RESISTOR FOR ISOLATION/FAULT PROTECTION
"LIVE-AT-ZERO" INPUT
OVERRANGE
DIRECT TEMPERATURE MEASUREMENTS
VCC
VREF+
2.7V–5.5V
LTC2408
CHOPPER-STABILIZED
OP AMP
AMPLIFIEDLOW LEVEL
THERMOCOUPLES(TYPES R, S)
Pt RTD
HALF-BRIDGE
DIRECT THERMOCOUPLEVOLTAGE CONVERSION
(TYPES J, K)
CS
SDO
CLK
DIN
VIN
EN D2 D1 D0 DON'T CARE
EOC "0" MSB MSB
66ms BUILT-IN SETTLING TIME FOR THE MUX
HI-ZHI-Z
INTERNALOFFSET CAL
(66ms)
CONVERT
(66ms)
trol of the ADC and the mux. DIN isserially programmed to select thedesired input channel; SDO is theserial output data of the converter.DIN and SDO may be shared by usingan external driver with a high imped-ance output state. Since the LTC2408exhibits single-cycle settling, there isno overhead associated with digitalfilter settling time. At the conclusionof each conversion, a new channelmay be selected by a 4-bit serial inputword, or the same channel can beretained by not shifting in a newword. A new input channel may beselected up to 66ms after the data-output read has been completed. This66ms period may be used to allow theinput signal to settle or offer the userflexibility in the timing of the muxchannel selection.
ConclusionThe LTC2408 is a highly accurate NoLatency ∆Σ converter capable of digi-tizing a variety of input signals. Itsexceptional noise performance allowsdirect digitization of sensors. Thedevice can measure microvolts onone channel and volts on another, allwith 10ppm accuracy. The LTC2408requires no user calibration or PGA,and there is no overhead associatedwith the input multiplexer. TheLTC2408’s exceptional accuracy,ease-of-use and eight input channelsmake it an ideal multichannel ADCfor complete system monitoring.
Figure 5. The LTC2408 simultaneously measures many input devices.
Figure 6. Mux/ADC timing and look ahead
Linear Technology Magazine • September 199924
DESIGN FEATURES
Micropower 5-Lead SOT-23Switching Regulators Extend BatteryLife in Space-Sensitive Applications
by Bryan LegatesIntroductionThe LT1615 and LT1617 are designedfor portable electronics that need apower solution with a minimum foot-print and long battery life. Thesedevices can be used as step-up orboost converters, single-ended pri-mary inductance converters (SEPIC)or positive-to-negative converters.With an input voltage range of 1.2V to15V, these devices are ideal for a widerange of applications and work with avariety of input sources. An internal36V switch allows the two devices toeasily generate output voltages of upto ±34V without the use of costlytransformers. The LT1615 is designedto regulate positive output voltages,whereas the LT1617 is designed todirectly regulate negative output volt-ages without the need for level-shiftingcircuitry. Both parts use a current-limited, fixed off-time control scheme,which helps achieve high efficiencyoperation over a wide range of loadcurrents. With a no-load quiescentcurrent of only 20µ A (with the outputin regulation) and a shutdown quies-cent current of 0.5µA, these devicessqueeze the most life out of any bat-tery application. Both devices usetiny, low profile inductors andcapacitors to minimize the overallsystem footprint and cost.
The LT1615 and LT1617 are pincompatible with two other membersof the PowerSOT™ family, the LT1613and LT1611, respectively. This allowsthe same board layout to be used toevaluate the performance of multipledevices. The LT1613 and LT1611 areboth current mode, constant fre-quency devices, capable of producinglarger output currents than theLT1615 and LT1617.1
LT1615 2-Cell to 3.3VBoost ConverterA popular supply for many portableelectronic devices, a 2-cell alkaline to3.3V converter with the LT1615 candeliver 60mA of load current. Thecircuit is shown in Figure 1 and thesystem efficiency appears in Figure 2.The efficiency peaks at 84% with afresh 2-cell battery and averages 78%over the entire 1.5V to 3V input volt-age range. Switching waveforms withan input voltage of 2.8V and a 60mAload appear in Figure 3. This photoillustrates the Burst Mode operationof the LT1615, as the system deliversenergy to the output in short burstsabout every 15µs. The device is instandby (drawing only 20µ A of quies-
cent current) for about 12µ s of each15µ s burst cycle, which greatlyincreases the overall converterefficiency.
LT1615 1-Cell Li-Ionto 15V Boost ConverterThe internal 36V switch of the LT1615makes it an attractive choice forapplications that need a high outputvoltage at a relatively low current.Figure 4’s circuit is a typical systemthat provides 15V at 15mA from asingle-cell Li-Ion battery. Efficiency,shown in Figure 5, reaches 82% froma fully charged battery.
VIN SW
FB
LT1615
VIN1.5V TO 3V
L14.7µH D1
SHDN
604k
1MC222µF
VOUT3.3V60mA
1615 TA03
GNDC14.7µF
C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150C2: TAIYO YUDEN JMK325BJ226 (408) 573-4150L1: MURATA LQH3C4R7M24 (814) 237-1431D1: MOTOROLA MBR0520 (800) 441-2447
4 3
2
154.7pF
90
85
80
75
70
65
60
55
500.1 1 10 100
LOAD CURRENT (mA)
EFFI
CIEN
CY (%
)
VIN = 1.5V
VIN = 3.0V
VOUT20mV/DIV
AC COUPLED
IL1500mA/DIV
VSW2V/DIV
5µs/DIV
Figure 1. 2-cell to 3.3V boost converter
Figure 2. 3.3V boost converter efficiencyreaches 84%
Figure 3. Switching waveforms of 3.3V boost converter with 60mA load
Linear Technology Magazine • September 1999 25
DESIGN FEATURES
LT1615 33V Boost ConverterThe converter in Figure 6 displays theexceptional input and output voltagerange of the LT1615. A 33V output iseasily generated from a wide ranginginput voltage using a simple boosttopology. A small, 1µ F ceramiccapacitor is all that is needed at theoutput, making the total footprintmuch smaller than other systems thatneed much larger tantalum capaci-tors. The efficiencies for inputs of3.3V, 5V and 12V are shown in Figure
7. With a 12V input, this convertercan deliver up to 1.32W (40mA at33V) of power at an efficiency of 85%,all from a tiny SOT-23 package.
LT1615 1-Cell Li-Ionto 3.3V SEPICLithium-ion (Li-Ion) is the battery ofchoice for systems needing the mostenergy with the lightest weight, butwith a cell voltage ranging from 4.2Vdown to 2.5V, a simple boost topologycannot be used to provide a 3.3V
VOUT20mV/DIV
AC COUPLED
IL1200mA/DIV
VSW5V/DIV
2µs/DIV
output. Figure 8’s circuit is a SEPICconverter that can easily do the job,providing 100mA of load current. Thecircuit shown uses two separateinductors, but a single, dual-windinginductor (a 1:1 transformer) can besubstituted. Figure 9 shows theswitching waveforms for this SEPICconverter with an input voltage of2.7V and a 50mA load. Notice thatthis circuit uses the same basic BurstMode operation as the boost con-verter, but the inductor current is
VIN SW
FB
LT1615
VIN2.5V TO 4.2V
L110µH D1
SHDN
88.7k
1MC24.7µF
15V15mA
1615 TA03
GNDC14.7µF
C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150C2: TAIYO YUDEN EMK316BJ475 (408) 573-4150L1: MURATA LQH3C100KZ4 (814) 237-1431D1: MOTOROLA MBR0520 (800) 441-2447
4 3
2
15
90
85
80
75
70
65
60
55
500.1 1 10 100
LOAD CURRENT (mA)
EFFI
CIEN
CY (%
)
VIN = 2.5V
VIN = 4.2V
VIN SW
FB
LT1615
VIN3.3V TO 12V
L110µH D1
SHDN
76.8k
2MC21µF
33V8mA
1615 TA03
GNDC14.7µF
C1: TAIYO YUDEN EMK316BJ475 (408) 573-4150C2: TAIYO YUDEN GMK316BJ105 (408) 573-4150L1: MURATA LQH3C100KZ4 (814) 237-1431D1: MOTOROLA MBR0540 (800) 441-2447
4 3
2
15
90
85
80
75
70
65
60
55
500.1 1 10 100
LOAD CURRENT (mA)
EFFI
CIEN
CY (%
) VIN = 5V
VIN = 3.3V
VIN = 12V
VIN SW
FB
LT1615
VIN2.5V TO
4.2V
L110µH D1
C31µF
SHDN
604k
1MC210µF
VOUT3.3V100mA
1615 TA07
GNDC14.7µF
C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150C2: TAIYO YUDEN JMK316BJ106 (408) 573-4150C3: TAIYO YUDEN JMK107BJ105 (408) 573-4150L1, L2: MURATA LQH3C100K24 (814) 237-1431D1: MOTOROLA MBR0520 (800) 441-2447
L210µH
4 3
2
154.7pF
Figure 4. Li-Ion to 15V boost converter
Figure 6. 33V boost converter
Figure 5. Efficiency of Figure 4’sboost converter
Figure 7. 33V boost converter efficiency
Figure 8. Li-Ion to 3.3V SEPIC converter Figure 9. Switching waveforms of the 3.3V SEPIC converter with 50mA load
Linear Technology Magazine • September 199926
DESIGN FEATURES
about one half that of the 3.3V boostcircuit whose waveforms are shownin Figure 3. For the SEPIC, the switchcurrent is split equally between thetwo inductors, with both inductorsproviding current to the load whenthe switch is turned off. Typical effi-ciency for this converter is 70%.
LT1615 ±20VDual-Output ConverterFigure 10 shows a single-inductor,dual-output converter ideal for use inapplications needing both a positiveand a negative voltage. The positiveoutput is generated using a tradi-tional boost converter, whereas thenegative output is generated using aninverting charge pump. Regulation isachieved by sensing the positive out-put, but by using identical outputcapacitors and rectifying diodes, thenegative output is also very well regu-lated. For a 2× difference in output
currents, the positive and negativeoutput voltages differ less than 3%;for a 10× difference, they differ lessthan 5%. This converter provides 8mAtotal output current from a 1.5V input(two fully discharged alkaline batter-ies) and 12mA total output currentfrom a 2.5V input (a fully dischargedsingle-cell Li-Ion battery). Increasingthe value of L1 to 22µ H increases theavailable output current by about15%. If even larger load currents areneeded, the same converter can beimplemented using the LT1613 inplace of the LT1615. If the accuracy ofthe negative output is more criticalthan the accuracy of the positive out-put, try the same topology using theLT1617 to regulate the negative out-put. Efficiency for this dual-outputconverter is quite good, reaching 77%with a single Li-Ion battery. See Fig-ure 11 for efficiency curves at severaldifferent input voltages.
LT1615 ±20VDual-Output Converterwith Load DisconnectOne drawback to the circuit shown inFigure 10 is that during shutdown,the positive output is one diode dropbelow the input voltage. This is anundesirable condition for many sys-tems, and can be easily corrected withthe circuit in Figure 12. This is a dual-output converter where both outputsare developed using charge pumps,so that both are disconnected fromthe input when the LT1615 is turnedoff. An additional benefit is that crossregulation is improved because bothoutputs are generated in the samemanner. For a 5× difference in outputcurrents, the positive and negativeoutput voltages differ less than 1%;for a 10× difference, they differ lessthan 2%. The improvements this cir-cuit provides do come at a cost: slightlylower efficiency. Figure 13 shows thatthe efficiency curves are about 3%lower for load currents greater than1mA, but the efficiency still reaches arespectable 74%.
VIN SW
FB
LT1615
VIN1.5V TO 5V
L110µH D1
SHDN
130k
2MC21µF
20V4mA
–20V4mA
1615 TA04
GND
C31µF
C14.7µF
C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150C2, C3, C4: TAIYO YUDEN TMK316BJ105 (408) 573-4150L1: MURATA LQH3C100K24 (814) 237-1431D1, D2, D3: MOTOROLA MBR0530 (800) 441-2447
4 3
2
154.7pF
C41µF
D3
D2
80
75
70
65
60
55
500.1 1 10 100
LOAD CURRENT (mA)
EFFI
CIEN
CY (%
)
VIN = 3V
VIN = 1.8V
VIN = 4.2V
VIN SW
FB
LT1615
VIN1.5V TO 5V
L110µH D1
D4
SHDN
130k
2MC21µF
20V4mA
–20V4mA
1615 TA05
GND
C31µF
C14.7µF
C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150C2, C3, C4: TAIYO YUDEN TMK316BJ105 (408) 573-4150C5: TAIYO YUDEN LMK212BJ105 (408) 573-4150L1: MURATA LQH3C100K24 (814) 237-1431D1, D2, D3, D4: MOTOROLA MBR0530 (800) 441-2447
4 3
2
154.7pF
C41µF C5
1µF
D3
D2
80
75
70
65
60
55
500.1 1 10 100
LOAD CURRENT (mA)
EFFI
CIEN
CY (%
)
VIN = 3V
VIN = 1.8V
VIN = 4.2VFigure 10. ±20V dual-output converter
Figure 12. ±20V dual-output converter with load disconnect
Figure 11. Efficiency of Figure 10’s circuit
Figure 13. Efficiency of Figure 12’s circuit
Linear Technology Magazine • September 1999 27
DESIGN FEATURES
VIN SW
NFB
LT1617
VIN2.5V TO 4.2V
L122µH
D1
C30.22µF
SHDN
24.9k
267kC24.7µF
–15V15mA
1615 TA07
GNDC14.7µF
C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150C2: TAIYO YUDEN EMK316BJ475 (408) 573-4150C3: TAIYO YUDEN TMK1316BJ224 (408) 573-4150L1, L2: MURATA LQH3C220K24 (814) 237-1431D1: MOTOROLA MBR0530 (800) 441-2447
L222µH
4 3
2
15
80
75
70
65
60
55
500.1 1 10 100
LOAD CURRENT (mA)
EFFI
CIEN
CY (%
)
VIN = 2.5V
VIN = 4.2V
VIN SW
NFB
LT1617
VIN5V
L110µH
D1
C30.22µF
SHDN
24.9k
619kC21µF
–33V20mA
1615 TA07
GNDC14.7µF
C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150C2: TAIYO YUDEN GMK316BJ105 (408) 573-4150C3: TAIYO YUDEN GMK1212BJ224 (408) 573-4150L1: MURATA LQH3C100K24 (814) 237-1431D1: MOTOROLA MBR0540 (800) 441-2447
4 3
2
15
D280
75
70
65
60
55
500.1 1 10 100
LOAD CURRENT (mA)
EFFI
CIEN
CY (%
)
VIN = 5V
LT1617 1-Cell Li-Ionto –15V Inverting ConverterMany electronic systems need a nega-tive supply but have only a positiveinput voltage to work with. A wellregulated, positive-to-negative con-verter can be easily designed usingthe LT1617. A Li-Ion to –15V invert-ing converter capable of providing15mA of load current is detailed inFigure 14. Efficiency for this inverter,shown in Figure 15, peaks at 76%.
LT1617 5V to –33VInverting Charge PumpFor the previous inverting converter,the maximum voltage seen by the
power switch is equal to the sum ofthe input and output voltages; this,along with the 36V switch rating,limits the output voltage that can beprovided using the inverting topol-ogy. If higher negative voltages areneeded, use an inverting charge pump,in which the maximum voltage seenby the switch is equal to the outputvoltage. Figure 16 shows a –33V, 20mAinverting charge pump that can pro-vide 20mA of load current. Efficiencyreaches 74%, as seen in Figure 17.
ConclusionThe applications presented show theversatility of the LT1615 and LT1617.These devices are capable of produc-ing a wide range of positive andnegative outputs from a variety ofinput sources. Their tiny SOT-23packages, along with small externalcomponents, combine to minimizefootprint and cost in space-consciousapplications.
References1. Pietkiewicz, Steve. “SOT-23 Switch-ing Regulators Deliver Low NoiseOutputs in a Small Footprint.” LinearTechnology IX:1 (February 1999),pp.11–13, 23.
Figure 14. Li-Ion to –15V inverting converter
Figure 16. –33V inverting charge pump
Figure 15. Li-Ion to –15V inverting converterefficiency
Figure 17. 5V to –33V inverting charge pumpefficiency
forthe latest information
on LTC products, visit
www.linear-tech.com
Linear Technology Magazine • September 199928
DESIGN FEATURES
Versatile Ring Tone Generator FindsUses in Motor Drivers and Amplifiers
OverviewThe LT1684 was specifically designedfor OEM telephone equipment. Itsfunction is to interface between thedigital control logic and the high volt-age analog phone line. When used inthe design of a phone system, theLT1684 allows software to control thefrequency, voltage and cadence of thering signal. Because of similarities ofapplication to telephone systems, theLT1684 ring tone chip finds itself athome in motor drives, digital inputamplified speakers, alarm systemsand sine wave UPS systems.
Introducing the LT1684The LT1684 provides the tool set toeasily implement a digital, pulse widthmodulated (PWM) signal to DC-coupled voltage converter (at highcurrents), while providing isolation,switching frequency filtering andoutput protection. Figure 1 is a sys-tem-level diagram of the LT1684 inaction. By controlling a pair of exter-nal MOSFETs, the LT1684 utilizestheir inherent robustness while pro-
viding control of the output voltageand current. In its telecom applica-tion circuit, the LT1684 provides upto ±240V of smooth, clean output atup to 200mA of output current. Higheroutput voltages are obtained by cas-coding MOSFETs, while highercurrents are readily achieved by usingthe LT1166 MOSFET automatic biasgenerator chip as a companion.
The CircuitFigure 2 is the schematic of theLT1684 implementing a digital-PWM-to-ringing-telephone converter. Thisis something like a high power, highvoltage, isolated, output filtered D/Aconverter. Like its DAC counterpart,the LT1684 has a precision reference,switches and an output amplifier.Unlike its DAC counterpart, it includespost-conversion ripple filtering, iso-lation and a robust high voltageoutput.
In addition to the isolation, filter-ing and amplification, the LT1684provides the gate-bias control andgate voltage protection for the twoexternal MOSFETs. Providing such aplethora of functions from a single,monolithic IC requires the use of asomewhat tricky circuit. This circuit
by Dale Eagar
LT1684STUFF
DIGITALSTUFF
VDIGITAL
DIGITAL GROUND
PWM
5V+ HIGH VOLTAGE
–HIGH VOLTAGE HIGHVOLTAGEGROUND
ISOLATIONBARRIER
HIGHVOLTAGELOAD
IN A GATE+
IN B
C86.8nF
Q2IRF9610
Q1IRF610
R10100k
V+
P1
µC
P2
LIM+
OUTBGOUT ATREF
COMP1
COMP2AMPIN LIM–
V–
GATE–
C6100pF
D11N4001
C720pF
C2100pF
C1100pF
DCISOLATION
R9100Ω
R110k
R210k
D21N5817
C90.1µF
C31µF
R33k
R6 5.1kR4 2k
R5300k
C56.8nF
R7100k
R8100Ω
RING-TONEOUTPUT
+100V
+
–
–100V
LT1684
C44700pF
1684 TA01
FB1
FB1: FERRONICS FMB1601 (716) 388-1020
–
+
C4
R5
R4R3
C3
Figure 1. The LT1684 uses differential pulse width modulation to provide isolation for digitallycontrolled analog power solutions.
Figure 2. Typical LT1684 digital-PWM-to-ringing-telephone converter applicationFigure 3. The basic 2nd order lowpass MFBfilter, as implemented by the LT1684
Linear Technology Magazine • September 1999 29
DESIGN FEATURES
is arrived at by applying a circuittransformation to a simple filter cir-cuit. This transformation is performedon the basic 2nd order lowpass mul-tiple feedback filter (MFB) circuit andends up looking somewhat like thefilter/amplifier shown in Figure 3.The filter/amplifier components inFigure 2 are R3–R5 and C3 and C4.Looking backwards through thecircuit transformation, these compo-
nents, in fact, form the 2nd orderMFB filter shown in Figure 3. Thevalues chosen for these componentsin Figure 2 implement a 2nd orderButterworth MFB lowpass filter witha cutoff frequency of 100Hz and a DCgain of 100. These were chosen toprovide ±80V of output swing withPWM duty factors of 10% to 90%,while filtering the 10kHz PWM rippleto meet telephone specifications.
PWMIN
1000pF
1000pF
10k
10k
3k
300k
0.1µF
470pF
5.1k2k
100k
100k
6800pF
6800pF
MTP-2N50E
MTP-2N50E
120V
–120V
100Ω
100Ω
IN A
IN B
BGOUT
AMPIN
V+
COMP1
LIM+
OUT
ATREF
COMP2
LIM–
V–
GATE+
GATE–
14
1
13
12
10
2
9
8
7
3
4
5
11
6
LT1684
FB1
100pF
20pF
180µH
180µH
3.9k
1nF
1nF
2N3904
2N3906
2N3906
2N3904
100Ω
47Ω
47Ω
100Ω
100Ω
IRF9240
–100V
1k
1k
1µF
1µF
0.22Ω
0.22Ω
0.22Ω
IRF230
100V
2k 1µHVIN
SENSE+
ILIM+
VOUT
ILIM–
SENSE–
VBOTTOM
VTOP
LT11662
4
1
8
7
3
6
5
TYPICAL POWER SLICE(1 OF 13 IN PARALLEL)
100Ω
FB1: FERRONICS FMB1601 (716) 388-1020
5kWLOAD
Stealing the LT1684for Use In Other ApplicationsThe LT1684, used as shown in Figure1, outputs a ring signal that meetsBelcore specification. This means wecan ring a phone 22,000 feet away.The LT1684 fits another role, where22,000 feet of separation would be anice minimum. This is the applica-tion of the LT1684 in the scaleablepower amplifier, as detailed in Figure4. This amplifier can be used to drivemotors, simulate the power companyin sine wave UPS systems and oper-ate large audio drivers. Because of itsscaleable nature, this design can beused at any power level. The circuit inFigure 4 is shown implementing a5000W bits-to-decibels converter.When this converter is implementedwith the appropriate audio driversand enclosures, the output soundpressure level can be significant—sosignificant, in fact, that the authorsuggests giving it a wide berth of atleast 22,000 feet.
–
+
3kHz LPF
TRIANGLE IN10kHz
5V 5V
–5V
LT1671 LT1784
C1100pF R1
10k
C2100pF
R210k
RTERM120Ω
IN A
IN B
LT1684
14
1
–1V
1V0
ANALOGIN
0.8V
0
–0.8V
RS485
Figure 4. 5kW PWM-to-analog converter
Figure 5. A remote, isolated, analog input amplifier using a robust RS485 driver and aterminated, twisted-pair line continued on page 35
Linear Technology Magazine • September 199930
35 Watt Isolated DC/DC ConverterReplaces Modules at Half the Cost
by Robert Sheehan
IntroductionThe choice between building or buy-ing an isolated DC/DC converter canbe a complex decision. If you use anoff-the-shelf, module you are con-strained by what the module makersoffer in their catalogs. In many cases,this may not precisely meet the re-quirements for a particular project.Also, while simple to use, the cost ofthese modules can be significantlyhigher than the cost of “rolling yourown.” The complexity of the DC/DC
design can be daunting and leadsmany to the decision to buy. Demon-stration circuit DC227 provides aDC/DC solution that can serve theneeds of many “standard” moduleapplications and offers the designerthe option of customizing the designto suit any slightly unusual systemrequirements. The power supply nowbecomes merely another collection ofparts in the system.
FeaturesDemonstration circuit DC227 is aboard level replacement for “half-brick” DC/DC converters. It canprovide 5V or 3.3V at up to 7A from anisolated 48V (36V to 72V) input. Theisolation voltage is 500VDC with anoption for 1500VDC. The circuit haslow input capacitance, fast turn-ontime, low shutdown power consump-tion and overtemperature protection.Continuous short-circuit protectioneliminates any restriction on maxi-mum capacitive load. The outputovervoltage circuit provides protec-tion for open or short circuits on theoutput power or sense lines. The stan-dard footprint allows the circuit to fit
directly into the module’s socket. Fig-ure 1 shows a typical layout for a2.28" by 2.40" circuit board.
DC227A-A is designed for 500VDCisolation and lowest cost; it uses astandard Coiltronics VERSA-PAC™transformer and a Pulse Engineeringinductor for the output filter. DC227A-B has 1500V isolation and uses asemicustom transformer, also fromCoiltronics. DC227A-C has 500VDCisolation and achieves the highestefficiency using a Panasonic type PCC-S1 inductor for the output filter. Theefficiency curves in Figures 2–5 arequite competitive, reaching 85% forthe DC227A-C with a 5V output. Theefficiency at 3.3V out is somewhatlower, due to the fixed losses of theoutput rectifier.
Circuit DescriptionThis single-ended forward converteroperates at a nominal switching fre-quency of 200kHz. Referring to theschematic in Figure 6, pulse widthmodulation is controlled by U1, anLT1247 current mode PWM control-ler. Transformer T2 and optocouplerQ7 provide galvanic isolation. C2 is a
Figure 1. Control (left) and power component (right) views of demonstration circuit DC227, a complete 35W DC/DC converterin a 2.28" by 2.40" footprint
VERSA-PAC is a trademark of Coiltronics, Inc.
DESIGN IDEAS35 Watt Isolated DC/DC ConverterReplaces Modules at Half the Cost................................................... 30
Robert Sheehan
Comparator Circuit ProvidesAutomatic Shutdown of the LT1795High Speed ADSL Power Amplifier................................................... 33
Tim Regan
SMBus ControlledCCFL Power Supply ..................... 35Jim Williams
Triple Output TFT-LCD Bias SupplyUses All Ceramic Capacitors ....... 36Gary Shockey
Low Noise Boost VCO Power Supply forPortable Applications .................. 37 Ted Henderson
Linear Technology Magazine • September 1999 31
local bypass cap to reduce commonmode–induced current.
To achieve fast start-up time, ahysteretic buck regulator is used forthe bias supply power. U2, an LT1431shunt voltage regulator, provides con-trol for this function, with Q1 actingas the switch element; L2 and C21provide output filtering. Q2 and Q4protect the circuit during a hot plug,making this a very robust design; it isalso impervious to output short cir-cuits. The input surge voltage islimited to 80V by the rating of Q1–Q4.
The main switching power paththrough T2 comprises L1 and C18 asthe input filter, Q6 as the primaryswitch, D7 as the secondary rectifierand L3 and C14, C16, C17 and C20as the secondary filter. Transient volt-
90
85
80
75
70
65
60
55
500 1 2 3 4 5 6 7
EFFI
CIEN
CY (%
)
IOUT (A)
VIN = 36V
VIN = 48V
VIN = 72V
90
85
80
75
70
65
60
55
500 1 2 3 4 5 6 7
EFFI
CIEN
CY (%
)
IOUT (A)
VIN = 36V
VIN = 48V
VIN = 72V
90
85
80
75
70
65
60
55
500 1 2 3 4 5 6 7
EFFI
CIEN
CY (%
)
IOUT (A)
VIN = 36V
VIN = 48V
VIN = 72V
90
85
80
75
70
65
60
55
500 1 2 3 4 5 6 7
EFFI
CIEN
CY (%
)
IOUT (A)
VIN = 36V
VIN = 48V
VIN = 72V
age suppressor D8 is used to protectSchottky diode D7 during large-sig-nal transient conditions. Power istransferred during the on cycle of Q6and integrated by the output filter,just as in a buck regulator. The inputfilter component values for L1 andC18 are optimal and should not bechanged without careful evaluation.C19 damps the input filter and willprovide adequate stability for largevalues of input inductance. See LTCApplication Note 19 for a discussionof input filter stability analysis.
Output voltage feedback is con-trolled using U3, another LT1431shunt voltage regulator, as an erroramplifier. In the event of a fault on theoutput power or sense lines, Z1/Q5will override U3 and provide overvolt-age protection. R10 and R21 are sizedto handle any overvoltage condition.
During an output short-circuit con-dition, the LT1247 is able to decreasethe on time of Q6 to less than 200ns.This results in good control of theoutput short-circuit current, keepingpower dissipation to a manageablelevel.
The demonstration circuit usessurface mount devices for Q6 and D7.For elevated temperature operationat the full rated load, TO-220 devicescan by mounted on a standard half-brick heat sink.
For –48V inputs that require hotswap capability, the LT1640H nega-tive voltage HotSwap™ controllerprovides a seamless interface.Demonstration circuit DC223A-Busing the LT1640HCS8 is the recom-mended solution for use with theDC227A.
ConclusionAt 35 watts, the topology presentedhere is one of the most common usedby the module manufacturers. This isonly one solution for isolated power,and opens up many possibilities forother input and output voltagecombinations. For lower power, dem-onstration circuit DC211 using theLT1425 isolated flyback switchingregulator is designed for 10 watts.Demonstration circuit DC259 usingthe LT1339 adds synchronous recti-fication, providing a high efficiencysolution for 50 watts. See the DC/DCConverter Module section of LTC’sVolume 1 1999 New Products Catalogfor additional information.
Figure 3. DC227A-A/B 5V output efficiency(typical)
Figure 4. DC227A-C 3.3V output efficiency(typical)
Figure 5. DC227A-A/B 3.3V output efficiency(typical)
Figure 2. DC227A-C 5V output efficiency(typical)
http://www.linear-tech.com/ezone/zone.htmlArticles, Design Ideas, Tips from the Lab…
Linear Technology Magazine • September 199932
DESIGN IDEAS
COLL
ECTO
R
COM
P
R MID
GND-
F
V+
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S
1 2 7 6
3 4 8 5
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R MID
GND-
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R TOP
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1 2 7 6
3 8 4 5
U3 L
T143
1CS8
+VIN
36V–
72V C1
912
µF10
0VR1
810
0k
R16
10k
Q2 FMM
TA56
R11
1k
Q1 FMM
TA56
R12
20k
Q3FM
MTA
06
D3 BAT5
4
D4FM
MD9
14 C12
1000
pF10
00V R7 10
0k
D2 B
AT54
R19
10Ω
D5 BAS2
1
R13
47k
R21.
30k
R3 1k
RT1
NTC
10k
AT 2
5°C
C6 3300
pFC1 47
00pF
C30.
01µF
R6 845Ω
R4 8.2k
C5 0.1µ
F
C4 1µF
–VIN
ON/O
FF
L14.
7µH
Q4FM
MTA
06R25
4.7
Ω
R23
16.2
kC1
00.
01µF
C18
2.2µ
F10
0V
L2 1mH
C15
470p
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C7 1500
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2545
CT(D
UAL)
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3µH
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C16
, C1
7, C
2022
0µF,
10V
×4
R14
220Ω
1/4W
+VOU
T
–VOU
T
Z1 5.6V
R31
100Ω
R30
1k
Q5FM
MT3
904
+SEN
SE
TRIM –S
ENSE
R21
100Ω
1/2W
R27
15.4
k0.
1%
R28
7.41
k0.
1%JP3
R20
10k
R22
4.99
k0.
1%R10
100Ω
1/2W
R26
100Ω
C9 0.04
7µF
C11
0.02
2µF
C13
0.02
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MD9
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Q7M
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7
R29
330Ω
C210
00pF
, 100
0VOR
2200
pF S
AFET
YRE
COGN
IZED
JP1JP2
R33
33Ω
1/4W
1 2 1112 10
4 9
5 8
6 73 8 7
46
REPMUJ
DNA2T
3L2C
A-A722CDNOITALOSI
CDV0051PJ;0021-5PV
SCINORTLIOC36635-EP
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TAGXK201Y8081JVNO
MIRTI V
B-A722CDNOITALOSI
CDV00512PJ;2X-18241-20XTC
SCINORTLIOC36635-EP
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O A
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ER F
OR 3
.3V
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P3: O
PEN
= 3.
3V; S
HORT
ED =
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CHAS
SIS
R15
1k
C21
10µF
25V
R17
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, 1/4
W
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941
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SAN
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X(6
19) 6
61-6
835
Figure 6. 35W isolated DC/DC converter schematic diagram
Linear Technology Magazine • September 1999 33
DESIGN IDEAS
Comparator Circuit Provides AutomaticShutdown of the LT1795 High SpeedADSL Power Amplifier by Tim Regan
IntroductionData transmission standards suchas HDSL2, ADSL (both Full Rate andG.Lite) and VDSL (collectively knownas xDSL) require the combined speed,output power and dynamic rangecapabilities of the LT1795 to drive thetelephone line. In a typical data com-munications installation, in both thetelephone central office and officebuilding sites, hundreds of telephone-wire pairs are brought together into aline multiplexer. These multiplexerscompact the individual line driver andreceiver circuits to save space. Eightlines per PC card are often imple-mented. This tight partitioning raisesthe challenges of both power and heatmanagement in each installation.
A simple comparator circuit can beused to monitor the activity of anindividual phone line and completelyshut down the line driver when not inuse. This provides a means of imp-lementing activity-based powerconsumption with reduced overallheat generation.
Controlling Power DissipationThe LT1795 is a dual, high speed,current feedback amplifier with highoutput current capability. With a
50MHz Gain Bandwidth product,900V/ms slew rate and an outputstage that can source and sink 500mA,the LT 1795 is ideal for use as a lowdistortion differential line driver invery high data rate modem applica-tions.For the LT1795 to obtain full perfor-mance, a fair amount of quiescentoperating current is required—typi-cally 60mA. When supplied by ±15Vpower supplies to obtain maximumdynamic output range, the quiescentpower dissipation with no load is typi-cally 1.8 watts (30V × 60mA). In theseapplications, however, full perfor-mance is not required at all times orfor all applications. To address thisissue, the LT1795 provides the abilityto completely shut down the driver orto tailor the quiescent operating cur-rent to match the actual requirementsof a particular application. Figure 1illustrates how this control isimplemented.
Pins 10 and 11 of the LT1795 com-bine to control the operating currentof the amplifiers. If the ShutdownReference (pin 11) is grounded andthe Shutdown input (pin 10) is drivento a voltage greater than two diodedrops above ground, both amplifiersare biased to “full speed ahead” withmaximum AC performance and alsomaximum quiescent power dissipa-tion. The current through the twodiodes shown in Figure 1 is internallylimited to 300µ A and results in 60mAof quiescent current for the amplifi-ers. Resistance can be added betweenthe Shutdown Reference pin andground to limit the maximum operat-ing current of the amplifiers. Thisprogrammability can optimize thetrade-off between AC performance(primarily slew rate and bandwidth)and quiescent package power dissi-pation. Many applications require the
peak current drive capability of theLT1795 but do not need the full band-width and slew performance. Detailsof this control can be found on theLT1795 data sheet.
If the Shutdown input is grounded,both amplifiers are disabled and thetotal quiescent current drain for thepackage is reduced to only 200µA.This feature can be used to save asubstantial amount of power whenthe line drivers are not in use at alltimes. A simple comparator circuit,as shown in Figure 2, provides atimed, automatic shutdown when noinput signals are applied to theamplifiers.
Timed Automatic ShutdownIn this circuit, an LT1795 is config-ured as a unity gain differential driverfor a 100Ω transformer-coupled wirepair. The two comparators in anLT1720 package monitor the signalson each of the input lines to the driveramplifier. If the signal on either inputexceeds the threshold set by the sen-sitivity adjustment (shown to providea range from 65mV to 500mV peak),the output of one or the other of thecomparators goes high immediatelyand puts the LT1795 into action. TheLT1720 comparators feature a propa-gation delay of only 4.5ns, allowingthem to respond to input signals wellbeyond 10MHz in frequency.
The LT1720 outputs are wire-ORedthrough a simple timing network.When either output is high, timingcapacitor CT charges quickly to 5Vthrough the diodes shown. While asignal is present, one or the othercomparator output keeps CT charged.The voltage on the timing capacitor isbuffered by a third comparator, anLTC1440, to provide a sharp 0V to 5Vshutdown/enable signal to theLT1795.
10
11
SHUTDOWN
SHUTDOWNREFERENCE
IQ ADJUST
0µA–300µA
TO INTERNALAMPLIFIERCIRCUITRY
0V
5V ON
OFF
Figure 1. Shutdown/IQ adjust for the LT1795
Linear Technology Magazine • September 199934
DESIGN IDEAS
–
+
–
+
–
+
–
+
–
+
+eIN
–eIN
49.9Ω
49.9Ω
49.9Ω
49.9Ω
1k 1k
100Ω
A11/2 LT1795
A21/2 LT1795
1:1
±eOUT
15V
9
8
2, 19
3
12
13
–15
11S/D REF
18
10 S/D
1k 1k
5V
1k
1k
1k
0.1µ
0.1µ
6.2k
RT2100k
IN4148
RT1100k
IN4148
50k*
CT1µF
C11/2 LT1720
C21/2 LT1720
1
2
3
45
8
510Ω
1µF
2.61k649k
3
4
5HYST
C3LTC1440
5V
1
6
2
ON
SHUTDOWN
1.2V
INPUT SENSITIVITY ADJUSTMENT65mV TO 500mV PEAK
*
1.3V
7
67
of the LTC1440—that is, when nofurther signals are applied to the inputof the circuit—the output of theLTC1440 snaps low and immediatelyshuts down the LT1795 power ampli-fier. The discharge voltage of CT isfairly slow moving but the input hys-teresis set between pins 4 and 5 of theLTC1440 allows a clean ON-to-OFFtransition.
With the component values pro-vided in Figure 2, the circuit “wakesup” and properly amplifies the inputsignal in approximately 50µs. Most ofthis delay is in the charging of CT andthe propagation delay of 8µ s throughthe LTC1440. With the Shutdown pindriven high, the LT1795 is fully up tospeed in only 1µ s. For a DSL applica-tion, this wake-up time occurs duringthe initial line “training-up” intervalof the data transmission sequence.
When the input signal is removed,CT discharges in approximately 65msto put the driver into a low powerdissipation state until the next datatransmission. This time-out intervaland wake-up time can be easily tai-lored through the selection of CT andRT1 and RT2. If the two timing resis-tors are of equal value, the OFF timeinterval is set by the relationship: t =0.713 • RTX • CT.
ConclusionThis simple comparator sensing andtiming circuit provides automatic con-trol over the power dissipation of ahigh speed power amplifier line driver.With no signal present there is nowasted quiescent power. This samesystem enhancement can also beachieved through direct logic controlof the LT1795 shutdown feature. Thiswould require a 0V to 5V controlsignal for each line, which is set orcleared in synchronization with eachdata transmission interval.
The LTC1440 is a CMOS compara-tor with a built-in voltage referenceand programmable hysteresis. The1.18V reference is used for DC bias-ing to set a stable input threshold forthe circuit. The input current of thiscomparator, 10pA, does not present asignificant load on the timingcapacitor.
When the signals of both inputs tothe driver amplifier drop below thethreshold, both outputs of the LT1720go low. While low, the timing capaci-tor discharges exponentially throughthe two 100k resistors, RT1 and RT2. Ifthe voltage on CT is given time todischarge below the 1.18V threshold
Figure 2. Automatic shutdown of the LT1795 power amplifier
For more information on parts featured in this issue, seehttp://www.linear-tech.com/go/ltmag
For more information on parts featured in this issue, seehttp://www.linear-tech.com/go/ltmag
Linear Technology Magazine • September 1999 35
DESIGN IDEAS
SMBus Controlled CCFL Power Supplyby Jim Williams
Figure 1 shows a cold cathode fluo-rescent lamp (CCFL) power supplythat is controlled via the popularSMBus interface. The LT1786 CCFLswitching regulator receives theSMBus instruction. The IC convertsthis instruction to a current, whichappears at the IOUT pin. This current,routed to the ICCFL pin, provides a setpoint for switching regulator opera-tion. The resultant duty cycle at theVSW pin pulls current through L2. L2,acting as a switched current sink,drives a resonant Royer convertercomposed of Q1–Q2, C1 and L1. Thehigh voltage sine wave produced at
L2’s secondary drives the floatinglamp.
Current flow into the Royer con-verter is monitored by the IC at pin 13(“Royer” in Figure 1).1 Royer currentcorrelates tightly with lamp current,which, in turn, is proportional tointensity. The IC compares the Royercurrent to the SMBus-derived cur-rent, closing a lamp-intensity controlloop. The SMBus permits wide-rangeregulated lamp-intensity control andallows complete IC shutdown. Opti-mal display and lamp characteristicspermit 90% efficiency. The circuit iscalibrated by correlating SMBus
instruction codes with attendant RMSlamp current. Detailed informationon circuit operation and measure-ment techniques appears in thereferences below.
BAT8V TO 28V
16
15
14
13
12
11
10
9
ICCFL
DIO
CCFL VC
AGND
SHDN
SMBSUS
ADR
CCFL VSW
BULB
BAT
ROYER
VCC
IOUT
SCL
SDA
CCFLPGND
LT1786F
LAMP
10 6
L1
R1750Ω
L2100µH
3 2 1 54
+
+
C7, 1µF
C1*0.068µF
C51000pF
R2220k
R3100k
C3A2.2µF35V
C42.2µF
3V ≤ VCC≤ 6.5V
TOSMBusHOST
C3B2.2µF35V
C227pF3kV
+
Q2* Q1*SHUTDOWN
C1 MUST BE A LOW LOSS CAPACITOR, C1 = WIMA MKI OR MKP-20(914) 347-2474
= PANASONIC ECH-U(201) 348-7522
Q1, Q2 = ZETEX ZTX849
OR ROHM 2SC5001
0µA TO 50µA ICCFL CURRENT GIVES 0mA TO 6mA LAMP CURRENT FOR A TYPICAL DISPLAY.
FOR ADDITIONAL CCFL/LCD CONTRAST APPLICATION CIRCUITS, REFER TO THE LT1182/83/84/84F DATA SHEET
D1BAT85
1
2
3
4
5
6
7
8D1
1N5818
L1 = COILTRONICS CTX210605
L2 = COILTRONICS CTX100-4
*DO NOT SUBSTITUTE COMPONENTS
COILTRONICS (561) 241-7876
(516) 543-7100
(800) 955-7646
Figure 1. 90% efficient floating CCFL with 2-wire SMBus lamp-current control
References:1. Williams, Jim. Linear Technology
Application Note 65: A FourthGeneration of LCD BacklightTechnology. November 1995.
2. LT1786F Data Sheet. LinearTechnology Corp. 1998.
1 Local historians can’t be certain, but this may bethe only IC pin ever named after a person.
Analog Inputs WelcomeThe scaleable amplification systemdetailed in Figure 4 can be drivenwith analog inputs while still main-taining full isolation. Such a systemis detailed in Figure 5, where theanalog input is filtered (to preventailiasing) and converted to PWM. Fig-ure 5 goes on to show the use of an
RS485 differential driver to drive atwisted pair line. The receiver end ofthe twisted pair line is terminatedwith a resistor and put across theisolation barrier. This provides verygood ESD protection on both ends ofthe line.
ConclusionThe LT1684 is useful in a wide varietyof applications. The LT1684 is a highlyintegrated solution for use in anysystem that requires digital control ofhigh output voltage or high outputpower.
Ring Tone, continued from page 29
Linear Technology Magazine • September 199936
DESIGN IDEAS
Triple Output TFT-LCD Bias SupplyUses All Ceramic Capacitors by Gary Shockey
Current power supply require-ments for TFT-LCD panels call for an8V or 10V main supply plus two ormore auxiliary outputs. The overalllayout must be small and meet tightheight requirements of under 2mm.Bulky inductors and capacitors mustbe eliminated if the design is to meetspace requirements. The circuit de-scribed in this design idea featuresthe new LT1949 and delivers 8V at200mA from 3.3V while generatingauxiliary 24V and –8V outputs capableof 10mA of output current.
The LT1949 is a boost switchingregulator that comes in the MSOP-8package and has an integrated 1.1ANPN power transistor. For this appli-
cation to be small and have a lowprofile, the boost converter mustswitch at a high frequency, whichallows compact inductors to be used,and must be able to work with ceramic
output capacitors. The LT1949 doesboth of these things. The switchingfrequency is fixed at 600kHz and theexternal compensation pin allows forloop characteristics to be tuned sothat tiny ceramic output capacitorscan be used.
To understand circuit operation,refer to Figure 1. The LT1949 gener-ates the 8V output in the normalboost mode configuration, while usingcharge pumps for the 24V and –8Voutputs. During boost operation, theSW pin is switching between VOUTand ground. When at VOUT, capacitorC6 is charged to VOUT through D5.When the SW pin flies to ground, C6holds its charge, causing D6 to beforward biased, charging C5 to –8V.The positive 24V output is developedin a similar fashion except that VOUTis tripled. Figure 2 details the tran-sient response of VOUT to an 80mA to200mA load step.
VOUT100mV/DIV
INDUCTORCURRENT
500mA/DIV
LOADSTEP
200mA80mA
50µs/DIV
Figure 2. Transient response for an 80mA to 200mA load step
Figure 1. 3.3V to 8V/200mA DC/DC converter with auxiliary 24V and –8V outputs
SHDN
LBI
FB
LBO
VIN SW
VC GND
U1LT1949
24V/10mA
VOUT8V/200mA
–8V/10mA
C44.7µF
C54.7µF
C210µF
R240.2k1%
R37.5k1%
R147k
C3680pF
C110µF
L110µH
D1 D2 D3 D4
C70.1µF
C80.1µF
C90.1µF
VIN3.3V
C64.7µF
D5
D6
SHUTDOWN
L1: SUMIDA CDRH628B-100(847) 956-0666
C1, C2: TAIYO YUDEN TMK432BJ106MN X7R 1210(408) 573-4150
C4: TAIYO YUDEN TMK325BJ475MN X5R 1210C5, C6: TAIYO YUDEN LMK316BJ475ML X5R 1206C7–C9: 0.1µF, 50V CERAMICD1–D6: ZETEX FMMD7000 DUAL DIODE
(516) 543-7100D7: MBRM120LT3R3: 5.76k FOR 10V/–10V/30V OUTPUTS
D7
http://www.linear-tech.com/ezone/zone.htmlArticles, Design Ideas, Tips from the Lab…
Linear Technology Magazine • September 1999 37
DESIGN IDEAS
Low Noise Boost VCO Power Supplyfor Portable Applications
IntroductionMany portable RF products use volt-age controlled oscillators (VCOs) togenerate the RF carrier frequency.These applications often require lownoise VCO power supply voltages thatare greater than the primary batterysupply. A DC/DC converter poweringa low noise linear regulator is oftenused. Unfortunately, there are sev-eral disadvantages to this solution.The DC/DC converter tends to pro-duce noise that may not be rejectedby the regulator, resulting in regula-tor output noise far greater than thethermal noise levels. The linear regu-lator may require a large output
compensation capacitor with specificESR requirements. The board areafor both devices and support compo-nents can be large. The LTC1682charge pump DC/DC voltage con-verter has been designed to minimizethese issues. The charge pump andlinear regulator have been mutuallyoptimized for minimum regulator out-put noise. The linear regulator wasdesigned to operate with several dif-ferent types of output capacitorsincluding small, low value, low ESRceramic capacitors. The charge pumpvoltage converter and low dropoutlinear regulator are combined on one
die and assembled in an extremelysmall MS8 package. Both fixed andadjustable output voltage versionsare available to cover the widest pos-sible output voltage range.
VCO Power SupplyFigure 1 shows the LTC1682 generat-ing a 4.2V low noise power supply fora 900MHz VCO with an input voltagerange of 2.5V to 4.4V. Figure 2 showsthe close-in phase noise of the VCOoperating open loop and Figure 3shows the typical peak-to-peak noisevoltage at VOUT.
ConclusionThe new LTC1682 family of chargepump DC/DC voltage converters rep-resents a complete low noise solutionfor VCO power supplies. A wide inputvoltage range of 1.8V to 4.4V, lowdropout voltage, low quiescent cur-rent, low external parts count andsmall board area make these devicesideal for portable applications.
36k
SHUTDOWN
15k
100k
4.7µF
0.22µF
VIN2.5V TO
4.4V
4.7µF
4.2V1
2
3
4
8
7
6
5
VOUT
SHDN
FB
GND
CPO
C+
VIN
C–
LTC1682
VCOMURATA
MQE001-902
1µF
4.7µF 1000pF
1k
B
M
P
C
1000pF1000pF 4.7µF
VC
VCOOUTPUT902MHz
AMPLITUDE10dB/DIV
CENTER = 902MHzSPAN = 100kHzSWP = 10sRESBW = 1kHzVBW = 30HzREF = 0dBm
VOUT200µV/DIV
100µs/DIV
COUT = 4.7µFIOUT = 10mAVOUT = 4.0VBW = 10Hz–2MHz
Figure 1. 4.2V VCO power supply
Figure 2. Close-in phase noise Figure 3. Output noise voltage
by Ted Henderson
For more information on parts featured in this issue, seehttp://www.linear-tech.com/go/ltmag
For more information on parts featured in this issue, seehttp://www.linear-tech.com/go/ltmag
Linear Technology Magazine • September 199938
NEW DEVICE CAMEOS
LT1762/LT1763 Low Noise,Micropower, Low DropoutRegulators Save Current inBattery-Powered ApplicationsThe LT1762 and LT1763 are low noise,low dropout linear regulators. TheLT1762 is rated for 150mA of outputcurrent, whereas the LT1763 is ratedfor 500mA. The typical dropout volt-age for either regulator at the ratedoutput current is 300mV. The reg-ulators are designed for use inbattery-powered systems, with 25µAoperating current for the LT1762 and30µA for the LT1763; both regulatorsfeature a 0.1µ A shutdown state. Qui-escent current is well controlled forthese devices; it does not rise in drop-out as is the case with many otherregulators.
The LT1762 and LT1763 regula-tors feature low noise operation. Withthe addition of an external 0.01µFbypass capacitor, output voltage noiseover the 10Hz to 100kHz bandwidthis reduced to 30µ VRMS for bothregulators. Both regulators can oper-ate with small capacitors, as low as2.2µF for the LT1762 and 4.7µ F forthe LT1763. Small ceramic capaci-tors can be used with either devicewithout the need for additional seriesresistance as is common with otherregulators. Internal protection cir-cuitry on both regulators includesreverse-battery protection, current
limiting, thermal limiting and reverse-current protection.
Both regulators are available infixed output voltages of 2.5V, 3V,3.3V and 5V, or as an adjustabledevice with an output voltage range of1.22V to 20V. The LT1762 regulatorsare packaged in the 8-lead MSOPpackage and the LT1763 regulatorsare available in the 8-lead SO package.
LTC2050:Zero-Drift OperationalAmplifier in SOT-23The LTC2050 is the latest zero-driftoperational amplifier in the LTC fam-ily. Available in the SOT-23 and SO-8,the LTC2050 permits single-supplyoperation down to 2.7V. The op ampconsumes 800µA of current and the6-lead SOT-23 and SO-8 packagesinclude a shutdown pin that dropssupply current below 10µ A. Inputcommon mode range has beenextended from the negative rail towithin 1V (typical) of the positive rail.
The LTC2050’s specifications rivalthose of the other members of thezero-drift op amp family, with typicaloffset voltages of 1µV, offset drifts of10nV/°C, DC to 10Hz noise of1.5µ VP-P, and a gain-bandwidth of3MHz. The LTC2050 uses the indus-try-standard op amp pinout andrequires no external components.
LTC1563-2/LTC1563-3:Easy-to-Use 3V, Rail-to-Rail,DC Accurate, Active RCLowpass Filter FamilyThe LTC1563-2 and LTC1563-3 areactive RC 4th order lowpass filterssuitable for systems with resolutionof 16 bits or more. They operate on asupply voltage as ranging from 2.7Vto ±5V. They support cutoff frequen-cies from 5kHz to 256kHz withrail-to-rail input and output. Eachpart comes in the narrow SSOP-16package (SO-8 footprint).
The LTC1563-2 and LTC1563-3are extremely easy to use: unlike con-ventional 4th order, discrete RC activelowpass filters, which require the cal-culation of a minimum of six differentexternal resistors and four differentexternal capacitors, they require onlysix equal-valued resistors to producea unity gain, 4th order Butterworth(LTC1663-2) or Bessel (LTC1563-3)filter. The calculation of the resistorvalues is trivial—complex algorithmsor filter design software is not needed.By simply allowing the six externalresistors to be of different values,gain and other transfer functions (forexample, Chebyshev, Gaussian tran-sitional and linear-phase equiripple)are achieved. Cascading two devicesforms an 8th order lowpass filter.Simple resistor-value tables make thedesign of any all-pole lowpass filterelementary. Complicated design al-gorithms are a thing of the past.
The LTC1563-2 and LTC1563-3also feature excellent DC offset (typi-cally less than 1mV); they are DCaccurate and their broadband noiseranges from 30µ VRMS to 60µVRMS de-pending on the cutoff frequency. Forcutoff frequencies below 25.6kHz, theparts have a low power mode wherethe supply current is typically 1mA.For higher frequencies, the supplycurrent is 10mA typically. A shut-down mode is also provided to limitthe supply current to less than10µ A.
New Device Cameos
For further information on anyof the devices mentioned in thisissue of Linear Technology, usethe reader service card or callthe LTC literature servicenumber:
1-800-4-LINEAR
Ask for the pertinent data sheetsand Application Notes.
Linear Technology Magazine • September1999 39
DESIGN TOOLS
Applications on DiskFilterCAD™ 2.0 CD-ROM — This CD is a powerful filterdesign tool that supports all of Linear Technology’s highperformance switched capacitor filters. Included is Fil-terView™, a document navigator that allows you toquickly find Linear Technology monolithic filter datasheets, the FilterCAD manual, application notes, designnotes and Linear Technology magazine articles. It doesnot have to be installed to run FilterCAD. It is notnecessary to use FilterView to view the documents, asthey are standard .PDF files, readable with any versionof Adobe Acrobat™. FilterCAD runs on Windows® 3.1 orWindows 95. FilterView requires Windows 95. TheFilterCAD program itself is also available on the web andwill be included on the new LinearView™ CD.
Available at no charge.
Noise Disk — This IBM-PC (or compatible) programallows the user to calculate circuit noise using LTC opamps, determine the best LTC op amp for a low noiseapplication, display the noise data for LTC op amps,calculate resistor noise and calculate noise using specsfor any op amp. Available at no charge
SPICE Macromodel Disk — This IBM-PC (or compat-ible) high density diskette contains the library of LTC opamp SPICE macromodels. The models can be used withany version of SPICE for general analog circuit simula-tions. The diskette also contains working circuit examplesusing the models and a demonstration copy of PSPICE™by MicroSim. Available at no charge
SwitcherCAD™ — The SwitcherCAD program is a pow-erful PC software tool that aids in the design andoptimization of switching regulators. The program cancut days off the design cycle by selecting topologies,calculating operating points and specifying componentvalues and manufacturer’s part numbers. 144 pagemanual included. $20.00
SwitcherCAD supports the following parts: LT1070 se-ries: LT1070, LT1071, LT1072, LT1074 and LT1076.LT1082. LT1170 series: LT1170, LT1171, LT1172 andLT1176. It also supports: LT1268, LT1269 and LT1507.LT1270 series: LT1270 and LT1271. LT1371 series:LT1371, LT1372, LT1373, LT1375, LT1376 and LT1377.
Micropower SwitcherCAD™ — The MicropowerSCADprogram is a powerful tool for designing DC/DC convert-ers based on Linear Technology’s micropower switchingregulator ICs. Given basic design parameters,MicropowerSCAD selects a circuit topology and offersyou a selection of appropriate Linear Technology switch-ing regulator ICs. MicropowerSCAD also performs circuitsimulations to select the other components which sur-round the DC/DC converter. In the case of a batterysupply, MicropowerSCAD can perform a battery lifesimulation. 44 page manual included. $20.00
MicropowerSCAD supports the following LTC micro-power DC/DC converters: LT1073, LT1107, LT1108,LT1109, LT1109A, LT1110, LT1111, LT1173, LTC1174,LT1300, LT1301 and LT1303.
Technical Books1990 Linear Databook, Vol I —This 1440 page collec-tion of data sheets covers op amps, voltage regulators,references, comparators, filters, PWMs, data conver-sion and interface products (bipolar and CMOS), in bothcommercial and military grades. The catalog featureswell over 300 devices. $10.00
1992 Linear Databook, Vol II — This 1248 page supple-ment to the 1990 Linear Databook is a collection of allproducts introduced in 1991 and 1992. The catalogcontains full data sheets for over 140 devices. The 1992Linear Databook, Vol II is a companion to the 1990Linear Databook, which should not be discarded.
$10.00
1994 Linear Databook, Vol III —This 1826 page supple-ment to the 1990 and 1992 Linear Databooks is acollection of all products introduced since 1992. A totalof 152 product data sheets are included with updatedselection guides. The 1994 Linear Databook Vol III is acompanion to the 1990 and 1992 Linear Databooks,which should not be discarded. $10.00
1995 Linear Databook, Vol IV —This 1152 page supple-ment to the 1990, 1992 and 1994 Linear Databooks is acollection of all products introduced since 1994. A totalof 80 product data sheets are included with updatedselection guides. The 1995 Linear Databook Vol IV is acompanion to the 1990, 1992 and 1994 Linear Databooks,which should not be discarded. $10.00
1996 Linear Databook, Vol V —This 1152 page supple-ment to the 1990, 1992, 1994 and 1995 Linear Databooksis a collection of all products introduced since 1995. Atotal of 65 product data sheets are included with updatedselection guides. The 1996 Linear Databook Vol V is acompanion to the 1990, 1992, 1994 and 1995 LinearDatabooks, which should not be discarded. $10.00
1997 Linear Databook, Vol VI —This 1360 page supple-ment to the 1990, 1992, 1994, 1995 and 1996 LinearDatabooks is a collection of all products introducedsince 1996. A total of 79 product data sheets are in-cluded with updated selection guides. The 1997 LinearDatabook Vol VI is a companion to the 1990, 1992, 1994,1995 and 1996 Linear Databooks, which should not bediscarded. $10.00
1990 Linear Applications Handbook, Volume I —928 pages full of application ideas covered in depth by40 Application Notes and 33 Design Notes. This catalogcovers a broad range of “real world” linear circuitry. Inaddition to detailed, systems-oriented circuits, this hand-book contains broad tutorial content together with liberaluse of schematics and scope photography. A specialfeature in this edition includes a 22-page section onSPICE macromodels. $20.00
1993 Linear Applications Handbook, Volume II —Continues the stream of “real world” linear circuitryinitiated by the 1990 Handbook. Similar in scope to the1990 edition, the new book covers Application Notes 40through 54 and Design Notes 33 through 69. Referencesand articles from non-LTC publications that we havefound useful are also included. $20.00
1997 Linear Applications Handbook, Volume III —This 976 page handbook maintains the practical outlookand tutorial nature of previous efforts, while broadeningtopic selection. This new book includes ApplicationNotes 55 through 69 and Design Notes 70 through 144.
Subjects include switching regulators, measurementand control circuits, filters, video designs, interface,data converters, power products, battery chargers andCCFL inverters. An extensive subject index referencescircuits in LTC data sheets, design notes, applicationnotes and Linear Technology magazines. $20.00
1998 Data Converter Handbook — This impressive1360 page handbook includes all of the data sheets,application notes and design notes for LinearTechnology’s family of high performance data converterproducts. Products include A/D converters (ADCs), D/Aconverters (DACs) and multiplexers—including the fast-est monolithic 16-bit ADC, the 3Msps, 12-bit ADC withthe best dynamic performance and the first dual 12-bitDAC in an SO-8 package. Also included are selectionguides for references, op amps and filters and a glossaryof data converter terms. $10.00
Interface Product Handbook — This 424 page hand-book features LTC’s complete line of line driver andreceiver products for RS232, RS485, RS423, RS422,V.35 and AppleTalk® applications. Linear’s particularexpertise in this area involves low power consumption,high numbers of drivers and receivers in one package,mixed RS232 and RS485 devices, 10kV ESD protectionof RS232 devices and surface mount packages.
Available at no charge
Power Management Solutions Brochure — This 96page collection of circuits contains real-life solutions forcommon power supply design problems. There are over70 circuits, including descriptions, graphs and perfor-mance specifications. Topics covered include batterychargers, desktop PC power supplies, notebook PCpower supplies, portable electronics power supplies,distributed power supplies, telecommunications andisolated power supplies, off-line power supplies andpower management circuits. Selection guides are pro-vided for each section and a variety of helpful designtools are also listed for quick reference.
Available at no charge.
Data Conversion Solutions Brochure — This 64 pagecollection of data conversion circuits, products andselection guides serves as excellent reference for thedata acquisition system designer. Over 60 products areshowcased, solving problems in low power, small sizeand high performance data conversion applications—with performance graphs and specifications. Topicscovered include ADCs, DACs, voltage references andanalog multiplexers. A complete glossary defines dataconversion specifications; a list of selected applicationand design notes is also included.
Available at no charge
Telecommunications Solutions Brochure —This 76page collection of application circuits and selectionguides covers a wide variety of products targeted fortelecommunications. Circuits solve real life problemsfor central office switching, cellular phones, high speedmodems, base station, plus special sections covering–48V and Hot SwapTM applications. Many applicationshighlight new products such as Hot Swap controllers,power products, high speed amplifiers, A/D converters,interface transceivers and filters. Includes a telecom-munications glossary, serial interface standards, protocolinformation and a complete list of key application notesand design notes. Available at no charge.
continued on page 40
DESIGN TOOLS
Information furnished by Linear Technology Corporationis believed to be accurate and reliable. However, LinearTechnology makes no representation that the circuitsdescribed herein will not infringe on existing patent rights.
Linear Technology Magazine • September 1999© 1999 Linear Technology Corporation/Printed in U.S.A./
LINEAR TECHNOLOGY CORPORATION1630 McCarthy BoulevardMilpitas, CA 95035-7417(408) 432-1900 FAX (408) 434-0507www.linear-tech.comFor Literature Only: 1-800-4-LINEAR
Acrobat is a trademark of Adobe Systems, Inc.; Windowsis a registered trademark of Microsoft Corp.; Macintoshand AppleTalk are registered trademarks of Apple Com-puter, Inc. PSPICE is a trademark of MicroSim Corp.
CD-ROM CatalogLinearView — LinearView™ CD-ROM version 3.01 isLinear Technology’s latest interactive CD-ROM. It al-lows you to instantly access thousands of pages ofproduct and applications information, covering LinearTechnology’s complete line of high performance analogproducts, with easy-to-use search tools.
The LinearView CD-ROM includes the complete productspecifications from Linear Technology’s Databook li-brary (Volumes I–VI) and the complete ApplicationsHandbook collection (Volumes I–III). Our extensivecollection of Design Notes and the complete collectionof Linear Technology magazine are also included.
A powerful search engine built into the LinearView CD-ROM enables you to select parts by various criteria,such as device parameters, keywords or part numbers.All product categories are represented: data conversion,references, amplifiers, power products, filters and inter-face circuits. Up-to-date versions of Linear Technology’ssoftware design tools, SwitcherCAD, Micropower
DESIGN TOOLS, continued from page 39 SwitcherCAD, FilterCAD, Noise Disk and Spice Macro-model library, are also included. Everything you need toknow about Linear Technology’s products and applica-tions is readily accessible via LinearView. LinearViewruns under Windows 95 and Macintosh® System 8.0 orlater.
Available at no charge.
World Wide Web SiteLinear Technology Corporation’s customers can nowquickly and conveniently find and retrieve the latesttechnical information covering the Company’s productson LTC’s internet web site. Located at www.linear-tech.com, this site allows anyone with internet accessand a web browser to search through all of LTC’stechnical publications, including data sheets, applica-tion notes, design notes, Linear Technology magazineissues and other LTC publications, to find informationon LTC parts and applications circuits. Other areaswithin the site include help, news and information aboutLinear Technology and its sales offices.
Other web sites usually require the visitor to downloadlarge document files to see if they contain the desiredinformation. This is cumbersome and inconvenient. Tosave you time and ensure that you receive the correctinformation the first time, the first page of each datasheet, application note and Linear Technology maga-zine is recreated in a fast, download-friendly format.This allows you to determine whether the document iswhat you need, before downloading the entire file.
The site is searchable by criteria such as part numbers,functions, topics and applications. The search is per-formed on a user-defined combination of data sheets,application notes, design notes and Linear Technologymagazine articles. Any data sheet, application note,design note or magazine article can be downloaded orfaxed back. (files are downloaded in Adobe Acrobat PDFformat; you will need a copy of Acrobat Reader to viewor print them. The site includes a link from which youcan download this program.)
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