linear technology magazine circuit collection, volume...
TRANSCRIPT
AN84-1
INTRODUCTION
Application Note 84 is the fourth in a series that excerptsuseful circuits from Linear Technology magazine to pre-serve them for posterity. This application note highlights“power” circuits from issue VI:1 (February 1996) throughissue VIII:4 (November 1998). Another application notewill feature data conversion, interface and signal process-ing circuits from the same era. Like its predecessor, AN 66,this Application Note includes circuits that can powermost any system you can imagine, from “server” powersupplies that generate in excess of 50 amps to micro-power systems for portable and handheld equipment.Also included are power converters that can be voltage
programmed using Intel’s VID code. Charge pump con-verters, linear regulators and battery charger circuits areincluded here, with Li-Ion batteries receiving extra atten-tion. There are, of course, circuits that cannot be so simplycategorized. Come browse. I’ll get out of the way and letthe authors describe their creations.
Note: Article Titles appear in this application note exactlyas they originally appeared in Linear Technology maga-zine. This may result in some inconsistency in the usageof terminology.
Linear Technology Magazine Circuit Collection, Volume IVPower Products
Richard Markell, Editor
Application Note 84
April 2000
TABLE OF CONTENTSIntroduction ......................................................................................................................................................... 1
REGULATORS—SWITCHING (BUCK)New LTC®1435–LTC1439 DC/DC Controllers Feature Value and Performance ..................................................... 4The LTC1266 Operates From ≥12V and Provides 3.3V Out at 12A ....................................................................... 7The New LTC1435 Makes a Great Microprocessor Core Voltage Regulator ......................................................... 8LTC1433/LTC1434: High Efficiency, Constant-Frequency Monolithic Buck Converter ........................................ 1024 Volt to 14 Volt Converter Provides 15 Amps ................................................................................................. 12LTC1553 Synchronous Regulator Controller Powers Pentium® Pro and Other Big Processors ......................... 13Synchronizing LTC1430s for Reduced Ripple .................................................................................................... 16Combine a Switching Regulator and an UltraFast™ Linear Regulator for a High Performance 3.3V Supply ....... 18The LTC1624: a Versatile, High Efficiency, SO-8 N-Channel Switching Regulator Controller ............................. 19Low Cost 3.3V to 1.xV 6 Amp Power Supply ..................................................................................................... 21The LT®1374: New 500kHz, 4.5A Monolithic Buck Converter ............................................................................. 23LTC1504: Flexible, Efficient Synchronous Switching Regulator Can Source or Sink 500mA .............................. 24High Efficiency Distributed Power Converter Features Synchronous Rectification ............................................. 26Fixed Frequency, 500kHz, 4.5A Step-Down Converter in an SO-8 Operates from a 5V Input ............................. 29VID Voltage Programmer for Intel Mobile Processors........................................................................................ 32New DC/DC Controller Enables High Step-Down Ratios ..................................................................................... 34LTC1627 Monolithic Synchronous Step-Down Regulator Maximizes Single or Dual Li-Ion Battery Life ............ 36The LTC1625 Current Mode DC/DC Controller Eliminates the Sense Resistor ................................................... 38PolyPhase™ Switching Regulators Offer High Efficiency in Low Voltage, High Current Applications ................. 39LTC1622: Low Input Voltage, Current Mode PWM Buck Converter .................................................................... 43
Application Note 84
AN84-2
Wide Input Range, High Efficiency Step-Down Switching Regulators ................................................................ 46REGULATORS—SWITCHING (BOOST)
±12 Volt Output from the LT1377 ...................................................................................................................... 51The LT1370: New 500kHz, 6A Monolithic Boost Converter ................................................................................ 53Bootstrapped Synchronous Boost Converter Operates at 1.8V Input ................................................................. 55
REGULATORS (SWITCHING)—BUCK-BOOST500kHz Buck-Boost Converter Needs No Heat Sink ........................................................................................... 56Battery-Powered Buck-Boost Converter Requires No Magnetics ....................................................................... 57
REGULATORS—SWITCHING (INVERTING)Making –5V 14-Bit Quiet .................................................................................................................................... 57Negative-to-Positive Telecommunication Supply ............................................................................................... 60Positive-to-Negative Converter Powers –48V Telecom Circuits ......................................................................... 61Low Noise LT1614 DC/DC Converter Delivers –5V at 200mA from 5V Input ..................................................... 62–48V to 5V DC/DC Converter Operates from the Telephone Line ....................................................................... 63
REGULATORS—SWITCHING (FLYBACK)The LT1425 Isolated Flyback Controller ............................................................................................................. 65High Isolation Converter Uses Off-the-Shelf Magnetics ..................................................................................... 68Wide-Input-Range, Low Voltage Flyback Regulator ........................................................................................... 69
REGULATORS—SWITCHING (LOW NOISE)The LT1533 Heralds a New Class of Low Noise Switching Regulators ............................................................... 70LT1533 Ultralow Noise Switching Regulator for High Voltage or High Current Applications .............................. 74
REGULATORS—SWITCHING (MULTIOUTPUT)LTC1538-AUX: a New Addition to LTC’s Adaptive Power™ Controller Family .................................................... 77High Efficiency, Low Power, 3-Output DC/DC Converter .................................................................................... 77Dual-Output Voltage Regulator ........................................................................................................................... 78Switcher Generates Two Bias Voltages without Transformer ............................................................................. 80New IC Features Reduce EMI from Switching Regulator Circuits ....................................................................... 81
REGULATORS—SWITCHING (MICROPOWER)Power Management and High Efficiency Switcher Maximize Nine-Volt Battery Life ........................................... 85LT1307 Micropower DC/DC Converter Eliminates Electrolytic Capacitors .......................................................... 86An Ultralow Quiescent Current, 5V Boost Regulator........................................................................................... 89Capacitive Charge Pump Powers 12V VPP from 5V Source ............................................................................... 90LTC1474 and LTC1475 High Efficiency Switching Regulators Draw Only 10µA Supply Current ........................ 91Free Digital Panel Meters from the Oppressive Yoke of Batteries ....................................................................... 94The LTC1514/LTC1515 Provide Low Power Step-Up/Step-Down DC/DC Conversion without Inductors ........... 95LTC1626 Low Voltage Monolithic Step-Down Converter Operates from a Single Li-Ion Cell ............................. 9612V Wall Cube to 5V/400mA DC/DC Converter is 85% Efficient......................................................................... 99Micropower 600kHz Fixed-Frequency DC/DC Converters Step Up from a 1-Cell or 2-Cell Battery ................... 100LT1610 Micropower Step-Up DC/DC Converter Runs at 1.7MHz ..................................................................... 103Low Noise 33V Varactor Bias Supply ............................................................................................................... 105The LTC1516 Converts Two Cells to 5V with High Efficiency at Extremely Light Loads ................................... 106
REGULATORS—LINEARLow Dropout Regulator Driver Handles Fast Load Transients and Operates on A Single 3V–10V Input ........... 107
Application Note 84
AN84-3
The LT1575/LT1577 UltraFast Linear Regulator Controllers Eliminate Bulk Tantalum/Electrolytic OutputCapacitors ........................................................................................................................................................ 108LT1579 Battery-Backup Regulator Provides Uninterruptible Power ................................................................. 111
BATTERY CHARGERSThe LT1511 3A Battery Charger Charges All Battery Types, Including Lithium-Ion .......................................... 114LT1512/LT1513 Battery Chargers Operate with Input Voltages Above or Below the Battery Voltage ............... 116Li-Ion Battery Charger Does Not Require Precision Resistors .......................................................................... 118LT1510 Charger with –∆V Termination ............................................................................................................ 119Constant-Voltage Load Box for Battery Simulation........................................................................................... 121High Efficiency, Low Dropout Lithium-Ion Battery Charger Charges Up to Five Cells at 4 Amps or More ........ 122Battery Charger IC Can Also Serve as Main Step-Down Converter ................................................................... 127LT1635 1A Shunt Charger ................................................................................................................................ 129800mA Li-Ion Battery Charger Occupies Less Volume than Two Stacked Quarters ......................................... 130Single-Cell Li-Ion Battery Supervisor ............................................................................................................... 132
POWER MANAGEMENTLTC1479 PowerPath™ Controller Simplifies Portable Power Management Design .......................................... 134The LTC1473 Dual PowerPath Switch Driver Simplifies Portable Power Management Design ........................ 137Short-Circuit-Proof Isolated High-Side Switch ................................................................................................. 139Tiny MSOP Dual Switch Driver is SMBus Controlled ........................................................................................ 140LTC1710: Two 0.4Ω Switches with SMBus Control Fit into Tiny MSOP-8 Package ......................................... 141
MISCELLANEOUSVID Voltage Programmer for Intel Mobile Processors...................................................................................... 141Battery Charger IC Doubles as Current Sensor ................................................................................................. 145100V, 2A, Constant-Voltage/ Constant-Current Bench Supply ......................................................................... 146A Complete Battery Backup Solution Using a Rechargeable NiCd Cell ............................................................. 147What Efficiency Curves Don’t Tell ..................................................................................................................... 149
APPENDIX A: COMPONENT VENDOR CONTACTS ......................................................................... 153INDEX............................................................................................................................ 157
, LTC, and LT are registered trademarks of Linear Technology Corporation; Adaptive Power, Burst Mode, No RSENSE, PolyPhase, PowerPath and UltraFast are trademarks of Linear TechnologyCorporation. Gelcell is a trademark of Johnson Controls, Inc.; Kool Mµ is a registered trademark of Magnetics, Inc.; Pentium is a registered trademark of Intel Corp.; VERSA-PAC is a trademark ofCoiltronics, Inc.
Application Note 84
AN84-4
Regulators—Switching (Buck)
NEW LTC1435–LTC1439 DC/DC CONTROLLERSFEATURE VALUE AND PERFORMANCEby Randy Flatness, Steve Hobrecht and Milton Wilcox
Introduction
The new LTC1435–LTC1439 multiple-output DC/DC con-trollers bring unprecedented levels of value to supplies fornotebook computers and other battery-powered equip-ment, while eliminating previous performance barriers.
10k0.01µF
10k
51pF
VIN 28V (MAX)
COSC
PLL IN
VPROG
PLL LPF
RUN/SS
LBO
POR
SFB
ITH
LBI
BG
SENSE+
SW
TGS
TGL
SENSE–
VOSENSE
EXT VCC
AUX FB
AUX DR
BOOSTINT VCC VIN DR VCC
SGND PGND AUX ON
T1 = DALE LPE-8562-A092(650) 665-9301
*CENTRAL SEMICONDUCTOR(516) 435-1110
LTC1437
0.1µF
EXT.CLOCK
56pF
510pF
IRF7403
IRLML2803
IRF7403
MBRS140
*CMDSH-3
1000pF
2.2µF
0.033Ω
VOUT15V/3A
VOUT212V/200mA
100µF10V×2
4.7µF25V
0.1µF
3.3µF35V
22µF35V×20.1µF
100
100
T1MBRS1100
100k
47k
1MEG
ZETEXFZT749
26V
+
+
+
+
+
For example, a new Adaptive Power™ output stage allowstwo previously incompatible parameters, constant fre-quency operation and good low current efficiency, tocoexist in the same power supply. A second breakthroughallows N-channel power MOSFETs to be used exclusively,while maintaining low dropout operation previously avail-able only with P-channel MOSFETs. Other innovationsinclude an auxiliary linear regulator loop, a phase-lockedloop (PLL) to synchronize the oscillator to an externalsource, a self-contained power-on-reset (POR) timer andprogrammable run delays useful for staging outputvoltages.
Figure 1. High Efficiency, Constant Frequency, Dual-Output Supply Delivers 3A at 5V and 250mA at 12V
Application Note 84
AN84-5
Cost Effective LTC1437 Switcher/Linear Combinationwith 5V/3A and 12V/200mA Outputs
The main switcher loop, shown in the schematic in Fig-ure 1, is set to 5V by strapping the VPROG pin high. Otheroutput options include 3.3V (VPROG low) and adjustable(VPROG open).
The 12V output in Figure 1’s circuit is provided by theauxiliary linear regulator operating in conjunction with asecondary winding feedback loop using the SFB pin. Theturns ratio for the transformer is 1:2.2, resulting in a
secondary output voltage of approximately 15V. The sec-ondary resistive divider causes the SFB pin voltage to dropbelow the internal 1.19V reference if the secondary outputis loaded and the 5V output has little or no load. This forcescontinuous operation as necessary to guarantee sufficientheadroom for the linear regulator to maintain 12V regula-tion independent of the 5V load. The auxiliary output isturned on and off with the AUX ON pin.
The auxiliary regulator can also be used in an adjustablemode, determined by the voltage on the AUX DR pin.When the AUX DR voltage is higher than 9.5V, as is the
10
VIN 5.2V-25V
BG1
SENSE1+
SW1
BG1
TGS1
TGL1
SENSE1–
EXT VCC
PLL IN
ITH1
PLL LPF
RUN/SS1
COSC
BOOST1
BG2
SENSE2+
SW2
TGS2
TGL2
SENSE2–
VOSENSE2
AUX ON
ITH2
AUX FB
RUN/SS2
AUX DR
BOOST2SFB1 INT VCC VIN VPROG1 VPROG2
LB1 LB0 SGND PGND POR2
Si4412DY
IRLML2803
Si4412DY
MBRS140
*CMDSH-3
1000pF
0.1µF
1000pF
0.033Ω
100µF10V×2
22µF35V×2
0.1µF
100
100
10k1000pF
LTC1439
0.1µF
EXT.CLOCK
10µH
220pF
10k0.01µF
56pF
Si4410DY
IRLML2803
Si4410
MBRS140
*CMDSH-3
1000pF
4.7nF
2.2µF
0.02Ω
VOUT23.3V/3A
VOUT32.9V/2.5A
VOUT15V/3A
221k
47k
100
100µF10V×2
330µF6.3V
0.1µF
22µF35V×2
0.1µF
100
100
10k1000pF
0.05µF
10µH
220pF
51pF
316k
20
ZETEXZTX849
MMBT2907ALT1
*CENTRAL SEMICONDUCTOR (516) 435-1110
+
+
+
+
+
+
Figure 2. High Efficiency, Constant-Frequency, Triple-Output Supply Features 200mV Dropout
Application Note 84
AN84-6
case in Figure 1, the regulator automatically configuresitself for fixed 12V operation using an internal AUX FBresistive divider. When AUX DR is less than 8.5V, theinternal divider is removed and the user can adjust theoutput voltage via an external divider referenced to 1.19V.The external auxiliary regulator PNP pass transistor issized for the desired output current; in this case a SOT-223device is used to deliver up to 200mA.
Synchronizable, Triple-Output, Low Dropout Supply
The LTC1439-based supply shown in Figure 2 is anexample of how three logic supply voltages, 5V, 3.3V and2.9V, can be easily derived using only two simple induc-tors. The two main DC/DC controller loops are used tosupply 5V/3A and 3.3V/5.5A. Up to 2.5A of the 3.3V outputcurrent is then used to supply a 2.9V output using theadjustable capability of the auxiliary linear regulator.
The 2.9V output also illustrates the use of an external NPNpass transistor with the auxiliary regulator. Because only0.4V is dropped across the NPN transistor, 2.9V efficiencyremains in the 85% range. And thanks to the 99% dutycycle capability of the switcher loops, Figure 2’s supplycan maintain all three output voltages in regulation downto VIN = 5.2V with a 2A load on the 5V output.
The phase-locked loops built into the LTC1437/LTC1436-PLL and LTC1439 offer a convenient means of synchroni-zation for the applications in Figures 1 and 2. The internaloscillator is actually a voltage-controlled oscillator (VCO)controlled by the voltage on the PLL LPF pin. When no
PLL IN signal is present, the PLL LPF pin goes low,causing the oscillator to run at its minimum frequency(fMIN = 180kHz with COSC = 56pF). Applying a 3.3V or 5Vlogic signal of any duty cycle to the PLL IN pin will causethe oscillator frequency to lock to the external frequencyand to track it up to a maximum of fMAX = 2 • fMIN. A logicsignal may also be coupled to PLL LPF to effect a 2:1frequency shift, provided that the initial frequency hasbeen set to less than 200kHz.
Figure 3 is a photograph showing the 3.3V output stagedto start 10ms before the 5V output when power is firstapplied to Figure 2’s circuit.
An internal regulation monitor is continually monitoringthe main controller output in the LTC1436/LTC1437, andthe controller 2 output (3.3V in Figure 2) in the LTC1438/LTC1439. When out of regulation or in shutdown mode,the POR open drain output pulls low. At start-up, once theoutput voltage has reached 5% of its final value, an internaltimer is started, after which the POR pin is released. Thetimer is accomplished by counting 216 oscillator cycles,yielding a delay-to-release reset of approximately 300msin a typical application.
The EXT VCC pin is normally connected to the 5V output toallow INT VCC power to be derived from the regulator itself.Quiescent current is then reduced because driver andcontrol currents are scaled by a factor approximately equalto the 5V controller duty cycle. EXT VCC can also beconnected to other external high efficiency sources, up toa maximum of 10V.
Figure 3. Start-Up of 3.3V and 5V Supplies is Easily StagedUpon Initial Application of Input Power
Application Note 84
AN84-7
THE LTC1266 OPERATES FROM ≥12V AND PROVIDES3.3V OUT AT 12Aby Craig Varga
Circuit Description and Operation
The design in Figure 4 relies on a floating high-side driverthat provides enough gate-drive capability to easily switcha large power MOSFET. The LTC1266 is configured todrive a P-channel MOSFET by tying pin 3 (PINV) to ground.This is required because there will be a net inversion by thefloating driver. Q4 controls the driver stage and providesgate-discharge capability through D3. When the low-sideswitches are on, C16 charges to 12V through D1. Whenthe LTC1266 signals Q1 to turn on, Q4 is turned off. R11provides base current for Q6, which, in conjunction withQ5, acts like an SCR. Once fired, the regenerative behaviorof Q5 and Q6 rapidly charges the gate of Q1. Since C16 isreferenced to the source of Q1, the top of C16 rises abovethe 12V supply rail as Q1 turns on, forcing the gate of Q1
TDRV
Q4VN2222LL
Q6MPS2222
PWRVIN
PINV
BINH
VIN
CT
ITH
SENSE–
ALL POLARIZED CAPACITORS ARE AVX TYPE TPS (207) 282-5111 OR EQUIVALENT
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
BDRV
PGND
LBO
U1LTC1266
LBIN
SGND
E1S/D
S/D
VFB
SENSE+
C11000pF
C131µF
C14300pF
C31000pF
C160.1µF
R61k
12V
R114.3k
R10220Ω
R1010k1%
R9220Ω
C23300pF
R851Ω
R125.1Ω
R131.0Ω
R40.015Ω
R50.015Ω
R1100Ω
R2100Ω
R36.04k, 1%
C170.001µF
C4 TO C6, C15330µF, 6.3V×4
VOUT3.3V12A
+
+
5Q1
Si4410
Q52N3906
6 7 8
1
4
2 3
D1MBR120T3
D3MBR0520LT3
5Q3
Si4410 6 7 8
1
4
2 3
5Q2
Si4410 6 7 8
1
4
2 3
D2MBRS320T3
L14µH
C7 TO C12100µF, 16V×6
to nearly 24V above ground. When the LTC1266 takespin 1 high, Q4 turns on, pulling charge from the gatecapacitance of Q1 through D3. This back biases the base-emitter junction of Q6, forcing the pull-up circuit, andtherefore Q1, off.
Since the input voltage is high relative to the output, thenominal duty factor of the high-side switch is small (in thiscase approximately 31%). As a result, the RMS currentthrough Q1 is relatively low. By contrast, the low-sideswitches are on nearly 70% of the time, and therefore seea much higher RMS current. This explains why the low-side switch employs two MOSFETs, whereas the high-sideswitch uses only one. Schottky diode D2 is used to helpkeep the body diodes of Q2 and Q3 from turning on duringthe short dead time before switching transitions. Thesebody diodes exhibit relatively long reverse recovery times,contributing to commutation losses. The Schottky diodeimproves overall efficiency several percent, but the circuitwill function correctly without it. Switching losses in the
Figure 4. 12V In, 3.3V/12A Out Supply
Application Note 84
AN84-8
TDRV
Q4VN2222LL
Q6MPS2222
Q7MPS2222A
PWRVIN
PINV
BINH
VIN
CT
ITH
SENSE–
1. ALL POLARIZED CAPACITORS ARE AVX TYPE TPS OR EQUIVALENT UNLESS NOTED OTHERWISE.(207) 282-5111
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
BDRV
PGND
LBO
U1LTC1266
LBIN
SGND
E1S/D
S/D
VFB
SENSE+
C11000pF
C111µF
C7470pF
C31000pF
C90.1µF
R61k
R114.3k
R1420k
R10220Ω
R710k1%
R9220Ω
C23300pF
R851Ω
R125.1Ω
R131.0Ω, 1/4W
R40.015Ω
R50.015Ω
R1100Ω
R2100Ω
R36.04k, 1%
+C12330µF35VSEE NOTE 3
C100.001µF
+C13330µF35VSEE NOTE 3
+
C4 TO C6, C15330µF 6.3V×4
VOUT3.3V12A
5Q1
Si4410
Q52N3906
6 7 8
1
4
2 3
D2MBR0520LT3
D4MBR0540LT3
5Q3
Si4410 6 7 8
1
4
2 3
5Q2
Si4410 6 7 8
1
4
2 3
D1MBRS340T3
L17µH
24V IN
D31N75912V
3. C12 AND C13 ARE PANASONIC TYPE HF OR EQUIVALENT(201) 348-7522
2. L1 CONSISTS OF 15 TURNS OF #16 AWG ON MAGNETICS, INC. 77848-A7 Kool Mµ CORE (800) 245-3984
two low-side switches are nearly zero, since these devicesare turned on and off into nearly zero volts (the forwarddrop of the Schottky).
There is no fundamental limitation on how high the maxi-mum input voltage can be with this approach. The drivelevel shift is limited by the breakdown rating of Q4.Obviously, the power transistors and input capacitorsmust be rated for the intended input voltage. A low power12V supply is needed to provide power for the LTC1266and voltage for the bootstrap supply.
Figure 5 shows a 24V input design. As the input supplyvoltage is increased, one thing to watch for is the potentialfor overlap in the high- and low-side turn-on/turn-offtransitions. The LTC1266 is designed to prevent shoot-through by actually waiting until the gate voltage of oneswitch is low before allowing the other switch to be turnedon. Using the floating driver defeats this capability, so thiscondition must be checked for. The high-side drive turn-on time may be reduced by lowering the value of R11.Using a larger device for Q4 will speed up the turn-offtransition. The value of C16 may also need to be a bit largerif R11 is reduced to limit drooping of the bootstrap supplyvoltage.
Figure 5. 24V In, 3.3V/12A Out Supply
THE NEW LTC1435 MAKES A GREATMICROPROCESSOR CORE VOLTAGE REGULATORby John Seago
Current microprocessor architectures require differentvoltages for the core and the I/O ring. For portable com-puter applications, the microprocessor core voltage isreduced for lower power consumption. Three high current
regulated voltages, 5V, 3.3V and 2.9V, are commonlyrequired. Several IC manufacturers offer two-output con-trollers, like the LTC1438, which are normally used for 5Vand 3.3V. Another controller is required to generate the2.9V. Figure 6 shows a simple circuit using the LTC1435to provide 2.9V at 2.65 amps for the Intel portable Pentium®
processor.
Application Note 84
AN84-9
The circuit’s 165kHz switching frequency was selected asa compromise between transient response and circuitefficiency. This frequency is determined by the value of C1.Output voltage transient response is shown in Figure 7.The transient response can be adjusted for other applica-tions by changing the values of compensation compo-nents R1, C3 and C14. Efficiency curves for different inputvoltages and load currents up to 3.2 amps are shown inFigure 8.
1
2
3
LTC1435
C168pF
C3330pF
L110µH
C4100pF
C1447pF
R110k
R20.033Ω
C20.1µF
C50.001µF
4
5
6
7
8
COSC
RUN/SS
C922µF35V
C1022µF35V
ITH
SFB
SGND
VOSENS
SENSE–
C9, C10 =C12, C13 =
D1 =D2 =
AVX, TPSE226M035AVX, TPSD107M010MOTOROLA, MBRS0530MOTOROLA, MBRS140T3
SENSE+
TG
5.5V-28V
BOOST
SW
VIN
INT VCC
BG
PGND
EXT VCC
16
15
14
Q1SI4412
13
12
11
10
9
C60.1µF
C84.7µF
C12100µF10V
C13100µF10VC7
0.1µFC11470pF
D1MBRS0530
D2MBRS140T3
2.9V/2.65A
R335.7k
R424.9k
Q2SI4412
+ +
+
+
+
L1 =Q1 = Q2 =
R2 =
SUMIDA, CDRH125-10SILICONIX, SI4412DYIRC, LR2010-01-R033-F
Another feature of the LTC1435 is the option to maintainconstant switching frequency under all load conditions orto select Burst Mode™ operation for the highest efficiencyat light loads. Pulling the SFB pin high enables Burst Modewhen load current drops to a low value. However, BurstMode can degrade transient response at low input voltagesand should not be used for pulsed load applications wheregood transient response at low input voltage is required.
Figure 6. 2.9V Regulator for Portable Pentium Processor
50
100
90
80
70
60
EFFI
CIEN
CY (%
)10A0.01A 0.1A 1.0A
5.5V INPUT
10V INPUT
15V INPUT
20V INPUT
28V INPUT
Figure 8. LTC1435 Efficiency Curves for Different Input Voltages
0.0A
4A
100mVP-P
500µs/DIV
50mV/DIV
2A/DIV
Figure 7. Output Voltage vs Transient Response
Application Note 84
AN84-10
The SFB pin in the circuit of Figure 6 is grounded, whichwill defeat the Burst Mode and ensure constant frequencyoperation.
It is sometimes necessary to shut down power to the load.RUN/SS is a dual-function pin on the LTC1435 that pro-vides both output voltage on/off control and output cur-rent soft-start capability. When RUN/SS (pin 2) is pulledlow by an open collector or open drain device, the outputvoltage is turned off and the controller shuts down. Thesoft-start feature takes over when the low is removed frompin 2. Figure 9 shows the output voltage under no-loadconditions at turn-on, with the soft-start capacitor C2equal to 0.1µF. This simulates the start up conditions of amicroprocessor held in standby until after the input volt-age has stabilized. If the regulator is started under full-loadconditions, the output current ramp time will be approxi-mately 0.5s/µF of soft-start capacitance. The output volt-
age during this soft-start period depends on the loadimpedance. If soft-start is not required, capacitor C2 is notused and the current limit setting of the regulator deter-mines the maximum load current during start-up.
In order to properly enhance the top MOSFET (Q1), INTVCC is level shifted by charge pumping capacitor C6 to INTVCC minus one diode drop. C6 provides the power to turnQ1 on and off. The INT VCC of the LTC1435 is regulated to5V, but will increase with higher voltage applied to EXTVCC, up to a maximum of 10V. For outputs between 5V and10V, the output should be connected to EXT VCC. Thepower loss of the INT VCC linear regulator will be replacedby the more efficient switcher output and the gate-drivevoltage of both MOSFETs will be increased for lower “ON”resistance. Figure 10 shows L1 input voltage and currentwith a 10 volt input, 2.9 volt output, and 2.65 amp loadcurrent.
LTC1433/LTC1434: HIGH EFFICIENCY, CONSTANT-FREQUENCY MONOLITHIC BUCK CONVERTERby San-Hwa Chee
Typical Application: Buck ConverterSupplies 3.3V at 600mA
Figure 11 shows a practical LTC1433 circuit that can beused for cellular telephone applications. Efficiency curvesfor this circuit at various input voltages are shown inFigure 12. Note that the efficiency reaches 93% at a supplyvoltage of 5V and a load current of about 150mA. This highefficiency makes the LTC1433 and LTC1434 attractive forpower-sensitive applications. The circuit works all the waydown to 3.6V at a load current of 250mA before droppingout and the oscillator frequency is a constant 210kHzdown to 20mA load current.
Typical Application: Positive-to-Negative Converter
Both the LTC1433 and LTC1434 can easily be set up for anegative output voltage. Figure 13 shows the schematicusing the LTC1433. The efficiency curve is shown in Figure14. This circuit is set up so that the output is taken from thedevice ground. Components that are normally referencedback to the device ground, such as the Run/SS capacitor,oscillator frequency capacitor and the ITH compensationnetwork, are connected to the output instead of to thecircuit ground.
2µs/DIV
10V
0.0V
4A
0.0A 2A/DIV
5V/DIV0.0V
0.0A
200µs/DIV
2.9V
1.25A
1V/DIV
1A/DIV
Figure 10. Soft-Start Output Voltage and Inductor CurrentFigure 9. Inductor Input Voltage and Current Waveforms
Application Note 84
AN84-11
0.001 0.01 0.10 1.0040
50
60
70
80
90
100
EFFI
CIEN
CY (%
)
LOAD CURRENT (A)
VIN = 5V
VIN = 9V
VIN = 12V
0.1µF100µF††
16V100µF†
10V
0.01µF
L1**68µH
D1*
VOUT–5.0V
INPUT VOLTAGE3V TO 7.5V
6800pF
5.1k680pF
VIN (V) IOUT MAX (mA)
3.0
4.0
5.0
6.0
7.0
7.5
180
240
290
340
410
420
* MOTOROLA MBRS130LT3** COILCRAFT DO3316 SERIES
† AVX TPSD107M010R0100†† AVX TPSE107M016R0100
1
2
3
4
5
6
7
8 9
10
16
15
14
13
12
11
BSW
NC
SSW
LBI
SGND
LBO
RUN/SS
NC
SVIN
POR
PWRVIN
PGND
ITH
COSC
VOSENSE
VPROG
LTC1433
100pF+
+
0.001 0.01 0.10 1.0040
50
60
70
80
90
100
EFFI
CIEN
CY (%
)
LOAD CURRENT (A)
VIN = 3.5V
VIN = 7V
VOUT = –5.0VCOSC = 100pF
1
23
4
56
78 9
10
16
1514
13
1211
BSW
NCSSW
LBI
SGND
LBO
RUN/SS
NC
PGND
SVIN
POR
PWRVIN
ITH
COSC
VOSENSEVPROG
100µHL1**
10k
5.1k680pF
6800pF
0.1µF
D1*
100µF†
10V
0.1µF
INPUT VOLTAGE3.6V TO 12V
68µF††
20V
VOUT3.3V
POWER ON RESET
LTC1433
47pF
* MBRS130LT3** COILCRAFT DO3316-104
† AVX TPSD107M010R0100†† AVX TPSE686M020R0150
+
+
Figure 11. LTC1433 Typical Application: 3.3V Output at 600mA Figure 12. Efficiency vs Load Currentfor Figure 11’s Circuit
Figure 13. Positive-to-Negative (–5.0V) Converter
Figure 14. Efficiency Curves for Figure 13’s Positive-to-Negative Converter
Application Note 84
AN84-12
24 VOLT TO 14 VOLT CONVERTERPROVIDES 15 AMPSby John Seago
Combining the LTC1435 with a large geometry powerMOSFET and good PCB layout allows large currents to beprocessed easily and efficiently. With the use of a currentsense transformer, output voltages greater than 10V canbe implemented. The circuit in Figure 15 shows an LTC1435configured as a conventional buck regulator using a singleN-channel MOSFET to control an output voltage greaterthan 10V with load current exceeding 15 amps. Theefficiency of the breadboard measured 94% with a 24Vinput, 14V output and 15A of load current. If maximum
+ +
+
+
+ +
TG
BOOST
SW
VIN
INT VCC
BG
PGND
EXT VCC
1
2
3
4
5
6
7
8
COSC
RUN/SS
ITH
SFB
SGND
VOSENS
SENSE–
SENSE+
16
15
14
13
12
11
10
9
U1LTC1435
C5, 100pF
C4, 47pF
C1120pF
C3, 330pF
C94.7µF
C111000µF35V
C8, 0.1µF
C200.001µF
C181µF
R15470Ω
R132.2k
C190.01µF
Q62N3904
R1616k
R6127k
R14430Ω
C7, 0.1µF
D1 D2
Q12N3904
100T1T
T1
C15100pF
R71K
R310Ω
R81.2Ω
C160.001µF
C170.001µF
C13470µF
25V
C14470µF25V
14VAT 15A
GND
Q42N3906
R90.62Ω
R10100Ω
R11100k
Q5VN2222LL
Q3
D3
D4
D81N4148
Q22N3906
1N758
D6
D71N751
C12470pF
D51N4148
C2, 0.1µF
R211.8k
R12, 100Ω
R4, 100Ω
R5, 100Ω
R1, 10k
C6, 0.001µF
L110µH
C101000µF35V
INPUT18V TO 28V
C10, C11 = NICHICON, UPL1V102MHH6(847) 843-7500
C13, C14 = NICHICON, UPL1E471MHH6D1, D3, D6 = MOTOROLA, MBRS0540
(800) 441-2447D4 = MOTOROLA MBR2045 WITH
THERMALLOY #7020 HEAT SINKQ3 = INTERNATIONAL RECTIFIER, IRL3803
(310) 322-3331WITH THERMALLOY #6299 HEAT SINK(972) 243-4321
L1: CORE = MAGNETICS, 55930-AZ(800) 245-3984WINDING = 8T #14 BIFILAR
T1: CORE = MAGNETICS W-41406-TCWINDING = PRI = 1T #18 SEC = 100T #32
efficiency is required, adding a second power MOSFET forsynchronous switching will improve efficiency byabout 1%.
This circuit’s 100kHz switching frequency was selected toreduce switching losses so that PCB mounted heat sinkscould be used without requiring additional air flow. Theswitching frequency can be set from 50kHz to 400kHz byselecting an appropriate value for C1. The current sensetransformer T1 uses a 1:100 turns ratio to scale down thebuck inductor input current and develop the voltage acrossR9, used by the ±SENSE inputs for regulation. Short-circuit protection is provided by Q4 and Q5. When thecurrent transformer secondary voltage developed across
Figure 15. 14V, 15A Buck Regulator
Application Note 84
AN84-13
A = Q3 SWITCH VOLTAGE20V/DIV
B = L1 CURRENT10A/DIV
C = T1 PRIMARY CURRENT10A/DIV
D = OUTPUT VOLTAGE RIPPLE0.2V/DIV
2µS/DIV
0.0V
0.0A
0.0A
14VDC
R8 and R9 is enough to turn on Q4, Q5 temporarily pullsthe RUN/SS pin low, turning off the regulator. Outputcurrent soft-starts when Q5 releases the RUN/SS pin. Thisresults in frequent attempts to establish output voltage ifa short exists, without high current continuously flowingthrough the power elements. The power elements consistof input capacitors C10 and C11, Current sense trans-former T1, buck inductor L1, power MOSFET Q3, commu-tating diode D4 and output capacitors C13 and C14.
Although the wide 3.6V–36V input voltage range and 99%duty cycle operation of the LTC1435 are ideal for battery/wall adapter input applications, operating above 95% dutycycle causes problems for the current sense transformer.To avoid transformer saturation, the Q6 stage limits dutycycle to approximately 90%. Current through R16 tries to
charge C20 to the 3V base voltage of Q6. If the switch cycleterminates at less than a 90% duty cycle, C20 is reset byD8. If the duty cycle exceeds 90%, C20 charges until Q6turns on, ending the switch cycle.
Switch voltage, inductor current, T1 primary current, andoutput voltage ripple waveforms are shown in Figure 16.These waveforms were measured with a 24V input, 14Voutput, and 15A load current. When MOSFET Q3 turns on,the switch voltage (Trace A) goes high, the inductorcurrent (Trace B) increases, as does the T1 primarycurrent (Trace C) and the output ripple voltage (Trace D).When Q3 turns off, the switch voltage goes low, inductorcurrent decreases as its stored energy supplies loadcurrent through D4, T1 primary current goes to zero andthe output voltage decreases slightly.
Figure 16. Buck Regulator Circuit Waveforms
LTC1553 SYNCHRONOUS REGULATOR CONTROLLERPOWERS PENTIUM
® PRO AND
OTHER BIG PROCESSORSby Y.L. Teo, S.H. Lim and Craig Varga
The LTC1553 provides current-limit and short-circuit pro-tection without the use of an external sense resistor. It hasexcellent (±1%) output regulation over temperature, linevoltage and load current variations. To compliment themain voltage-feedback loop, the LTC1553 includes twoadditional feedback loops to provide good large-signaltransient response. The LTC1553 adds additional internalcircuits to conform to the Intel Pentium Pro processor
power converter requirements while minimizing the num-ber of external components. An on-chip 5-bit digital-to-analog converter (DAC) provides output voltages con-forming to Intel’s specifications. This allows the LTC1553to read the code sent by the processor and provide it withthe requested voltage. The LTC1553 also provides apower-good indication (PWRGD) to the system. There isalso an on-chip overvoltage protection circuit that latchesthe regulator in an off state if the output voltage ever rises15% or more above the DAC-requested voltage.
In applications with other processors, the four DAC inputscan be routed to a jumper block, zero ohm resistors or a
Application Note 84
AN84-14
DIP switch, or hard wired, to set the desired outputvoltage. This allows the output voltage to be programmedeasily in steps while eliminating the need to stock anassortment of precision resistors. This flexibility in outputvoltage setting is cheap insurance against last-minutepower supply voltage changes by microprocessor manu-facturers.
LTC1553 Overview
The on-chip, 5-bit digital-to-analog converter (DAC) allowsthe output voltage to be adjusted from 1.80V to 3.5V, asshown in Table 1. Current limiting is maintained by sens-ing the voltage drop across the RDS(ON) of the high-sideMOSFET. The DAC accuracy, initial reference voltagetolerance and internal feedback resistor tolerances resultin a maximum initial output voltage error of ±1% of theselected output voltage. The line and load regulation plustemperature drift over the 0°C to 70°C temperature rangewill contribute another ±1% to the output error budget.This gives a total static operating error of less than ±2%,providing sufficient headroom (3%) for the dynamicresponse to remain within a ±5% output voltage tolerance,while still requiring a reasonable amount of outputcapacitance.
G1
IFB
G2
1552_06.eps
Pentium ProProcessorSYSTEM LTC1553
COMP
Q1A, Q1B, Q2: MOTOROLA MTD20N03HDL(800) 441-2447
LO2.0µH/18A
VCC
VIN = 5V
VOUT
PWRGD
Q1A, Q1B(2 IN PARALLEL)
Q2
DALE NTHS-1206N02(605) 665-9301
CONNECTINGVID0–VID4
TO DIP SWITCHTO SET VOUT
SS GND PGND SENSE
IMAX PVCC
5.6k5.6k RIMAX5.6k
1.8k
5V C02310µF
7 × 330µF
0.1µF
0.1µF
1N581710µF CIN990µF3 × 330µF
VID0–VID4
OUTEN
C1100pF
CSS0.01µF
0.1µFCC0.01µF
RC20k
FAULTOT
+
+
+
Typical Application
A typical application for LTC1553 is converting 5V to1.8V–3.5V in a Pentium Pro processor based personalcomputer. The supply may be in the form of a voltageregulator module (VRM) or may be implemented directlyon the motherboard. The output is used to power thePentium Pro processor and the input is taken from thesystem’s 5V supply. The circuit shown in Figure 17 pro-vides 1.80V–3.5V at 14A while maintaining output regula-tion within ±1%. The output voltage is determined byconnecting the five DAC inputs to the VID pins of theprocessor. The power MOSFETs are sized to minimizeboard space and allow operation without the need of a heatsink. With proper airflow, ambient temperature conditionsof up to 50° Celsius are acceptable. Typical efficiency isabove 90% from 1A to 10A at 3.3V out. (see Figure 18).Achieving higher output currents from LTC1553 baseddesigns is simply a matter of selecting appropriate MOS-FETs and passive components.
It pays to look at the regulator design from two perspec-tives: electrical and thermal. Most processor applicationsoperate at average currents that are approximately 80% orless of the specified peak current. As such, the thermal
Figure 17. Typical 5V to 1.8V–3.5V/14A LTC1553 Application
Application Note 84
AN84-15
LOAD CURRENT (A)
0
30
40
50
60
10
20
70
80
90
100
EFFI
CIEN
CY (%
)
100
1552_07.eps
0.1 1 10
design can be based on the lower current level. Highercurrents, while present, are typically not of sufficientduration to significantly heat the power devices. Thedesign does, however, need to be capable of delivering thepeak current without entering current limit or resulting indevice failures. Keep in mind that the power dissipation ina resistive element, such as a MOSFET, varies as thesquare of load current. As such, raising the load currentfrom 80% to 100% translates to approximately 56% morepower dissipation (1/0.82). Designing for this higher ther-
mal load results in a huge, and most likely unnecessary,design margin. A good understanding of your systemrequirements can result in substantial savings in the sizeand cost for the power supply.
RIMAX sets current limit to the desired level. Add one-halfof the inductor ripple current to the maximum load currentto determine the peak switch current. Multiply this currentby the maximum on-resistance of the selected MOSFETswitch to determine the minimum current limit thresholdvoltage. It’s a good idea to add at least a 10% margin to thislimit. Also, be sure to use the hot on-resistance of theMOSFET. A multiplier of about 1.4 times the room tem-perature RDS(ON) should be used to determine the hotresistance. In the case of two parallel MTD20N03HDLs(Q1A and Q1B), the cold resistance is approximately0.035Ω each; therefore, assume the hot resistance to beapproximately 0.050Ω. Divide this by two because theFETs are in parallel. The threshold voltage is programmedby multiplying the IMAX pin’s sink current by the value ofRIMAX. Since we now can determine the required thresh-old, we need to calculate the value of RIMAX. Use thespecified minimum sink current, 150µA, to calculate theresistor value.
The soft-start time is programmed by the 0.01µF capconnected to the SS pin. The larger the value of thiscapacitor, the slower the turn-on ramp.
Inductor LO is sized to handle the full load current, up to theonset of current limit, without saturating. A value ofbetween 2µH and 3µH is adequate for most processorsupply designs. Be careful not to overspecify the inductor.
Figure 18. Efficiency Plot for Figure 17’s Circuit
4DIV 3DIV 2DIV 1DIV 0DIV )CDV(0 1 1 1 1 *0 1 1 1 0 *0 1 1 0 1 *0 1 1 0 0 *0 1 0 1 1 *0 1 0 1 0 *0 1 0 0 1 *0 1 0 0 0 *0 0 1 1 1 *0 0 1 1 0 *0 0 1 0 1 08.10 0 1 0 0 58.10 0 0 1 1 09.10 0 0 1 0 59.10 0 0 0 1 00.20 0 0 0 0 50.21 1 1 1 1 UPCoN1 1 1 1 0 1.21 1 1 0 1 2.21 1 1 0 0 3.21 1 0 1 1 4.21 1 0 1 0 5.21 1 0 0 1 6.21 1 0 0 0 7.21 0 1 1 1 8.21 0 1 1 0 9.21 0 1 0 1 0.31 0 1 0 0 1.31 0 0 1 1 2.31 0 0 1 0 3.31 0 0 0 1 4.31 0 0 0 0 5.3
noisnapxeerutufrofdevreseR*
Table 1. Output Voltage vs VIDx Code
Application Note 84
AN84-16
The inductor need not retain its no-load inductance up tothe current-limit threshold. If the inductor still retains onthe order of 25% to 30% of its initial inductance underworst-case short-circuit current conditions, the circuitshould prove reliable. However, you do want to ensurethat approximately 60% to 75% of the initial inductance isretained at nominal full load. Excessive inductance roll-offwill result in higher than expected output ripple voltage athigh loads, along with increased dissipation in the powerFETs and the inductor itself.
Proper loop compensation is critical for obtaining opti-mum transient response while ensuring good stabilitymargins. The compensation network shown here givesgood response when used with the inductor and theoutput capacitors values shown in Figure 17. Several lowESR capacitors are placed in parallel to reduce the totaloutput ESR, resulting in lower output ripple and improvedtransient performance. Generally speaking, low ESR, highvalue output capacitors should be chosen to optimize theuse of board space. However, if the ESR value is too lowfor a given capacitor value, loop stability problems canoccur. The feedback loop depends on the frequency of theESR “zero” being well below the loop crossover fre-quency. There is 45° of positive phase shift at the fre-quency where the capacitive reactance equals the ESR ofthe capacitor. Without this phase shift, the loop would beimpossible to stabilize. Low ESR, AVX TPS-series tanta-lum capacitors are a very good compromise betweenESR, capacitance value and physical size.
Input capacitors are included to suppress the input switch-ing noise and to keep the input 5V supply variation to aminimum during the Q1 ON/OFF cycle. Excessive con-
ducted emissions are usually traced back to inadequateinput capacitance or poor layout of the power-path traces.The crucial parameter for the input capacitors is ripplecurrent rating. A reasonable rule of thumb says that theinput capacitor ripple current is going to be approximately50% of the load current. Therefore, in a typical PentiumPro processor application, the input capacitors should berated for close to 7ARMS. An excellent choice for the inputcapacitors are Sanyo OS-CONs or the equivalent. Theyhave extremely high ripple current ratings for their sizeand have demonstrated excellent reliability in this type ofapplication. Low ESR aluminium electrolytic capacitorsare a viable option from both input and output. Althoughlower in cost than OS-CONs or tantalum capacitors, theirlong-term reliability is not as good. Using 105°C capaci-tors and keeping operating temperatures low will help toobtain reasonable capacitor life.
The combination of the Dale NTHS-1206N02 thermistorand the 1.8k resistor are for overtemperature monitoring.The OT flag trips if the ambient temperature at Q1 reachesabout 90°C; at 100°C the G1 and G2 drivers stop operat-ing. If the system monitors the OT flag, there should beample time to take precautions, saving data and systemconfiguration information prior to an overtemperatureshutdown. Alternatively, CPU activity could be reduced,lowering power supply current and allowing the supply tocool down.
The PWRGD pin gives the CPU rail-voltage OK indication.If, for any reason, the output regulation falls out of the ±5%limit (including an overtemperature shutdown), PWRGDwill provide a logic low signal to the system monitor.
SYNCHRONIZING LTC1430s FOR REDUCED RIPPLEby Craig Varga
The recent move to split-plane microprocessors by severalCPU makers has led to the inclusion of multiple switchingregulators on many motherboard designs. These regula-tors typically provide 3.3V for system logic and a separatesupply for the processor core. Current requirements of5A–10A or more per supply are not unusual. The LTC1430synchronous buck regulator is commonly used to providethese tightly regulated supplies. By nature, the inputcurrent waveform in the buck topology is discontinuous,
resulting in large input ripple current. By synchronizing apair of supplies out of phase, it is possible to achieve adegree of ripple current cancellation. This results in lessstress on the input capacitors (the number of input capaci-tors could be reduced) and lower EMI. The ripple is easierto filter since the frequency is effectively doubled and thepeak-to-peak current is reduced.
It is extremely simple to synchronize a pair of LTC1430sin an appropriate phase relationship. Simply connect aresistor divider from the low gate drive of a “master”
Application Note 84
AN84-17
regulator to the sync pin of a “slave” regulator. Theresistors should divide the gate-drive voltage down tosomething slightly less than the VCC supply of the slaveregulator, typically from 12V down to approximately 4.5V.Total divider resistance of 20k to 30k is adequate. Also, theslave regulator must be set up to free run slower than themaster regulator. If, for example, the master is configured
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8
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16
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5
7
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G1
IFB
G2
PGND
–SENS
+SENS
FB
U1LTC1430
12V 5V
C3
C2
Q1
Q2
L1
R1130k
PVCC2
Vcc
FSET
IMAX
SD
COMP
SS
SGND
R215k
R310k
15
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8
10
9
4
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1
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6
PVCC1
G1
IFB
G2
PGND
–SENS
+SENS
FB
U2LTC1430
12V 5V
OUT 2
OUT 1
C4
C1
Q3
Q4
L2
PVCC2
Vcc
FSET
IMAX
SD
COMP
SS
SGND
DI1430_01.eps
to run at approximately 300kHz (a 130k resistor fromFSET to ground) the slave can be left to run at its naturalfrequency of 200kHz. The slave frequency will be forcedup to that of the master.
The sync function on the LTC1430 works as follows: whenthe shutdown pin is pulled low, the high-side switch turnsoff; normal duty factor control determines when the high-side switch will turn back on. As long as the shutdown pinis held low for less than approximately 40µs, the chip willnot shut down.
The simplified schematic (Figure 19) shows the synchro-nization circuitry. For a detailed description of LTC1430-based regulator designs, see the LTC1430 data sheet. Thescope photo (Figure 20) shows the voltage at the commonconnection of the two FETs of each regulator.
Figure 19. Simplified Schematic Diagramof Synchronization Circuitry
MASTER
SLAVE
Figure 20. Phase Relations Between the Switching Nodesof the Two Regulators
Application Note 84
AN84-18
COMBINE A SWITCHING REGULATOR AND ANULTRAFAST LINEAR REGULATOR FOR A HIGHPERFORMANCE 3.3V SUPPLYby Craig Varga
Introduction
It is becoming increasingly necessary to provide lowvoltage power to microprocessor loads at very highcurrent levels. Many processors also exhibit high speedload transients. The Pentium® Pro processor from Intelexhibits both of these requirements. This processorrequires 3.3V ±5% at approximately 14A peak (9A aver-age) and is capable of making the transition from a low
+
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+
DI1575_01.eps
TG
SW
BOOST
INTVCC
BG
S+
S–
EXTVCC
COSC
RUN/SS
ITH
SFB
SGND
VOS
9
1
2
3
4
5
6
16
14
15
12
11
8
7
13
10 C4, 4.7µFC50.1µF
D2MBRS330T3
R815K
R3100
R4100
C181000µF10V
C201000µF
10V
C191000µF10V
R67.5mΩ
L14µH
C2, 1000pF
VIN
U2LTC1435
IPOS
INEG
GATE
COMP
S/D
VIN
GND
FB
C21, 10pF
C221000pF
R21.21k
1%
C1, 470pF
R92k
Q1IRLZ44
R12.1k, 1% 3.3V
VCORE
1
2
3
4
8
7
6
5
U1LT1575
PGND
D1, CMDSH-3 Q3
Q2
C161µF
C14, 150µF, 16V
C151µF
C171µF
12V
C11150µF16V
C12150µF16V
C13150µF16V
C3, 0.1µF
C91500pF
R516.5k
C10, 1000pF
C8, 68pF
C7, 0.1µF
R735.7k
+
C231µF
C60.1µF
12V
40 × 1µFX7RCERAMIC0805 CASE
+
+
L1 = COILTRONICS CTX01-13199-X2(561) 241-7876
Q2, Q3 = SILICONIX SUD50N03-10(800) 544-5565
power state to full load in several clock cycles. Generally,switching regulators are used to supply such high powerdevices, because of the unacceptable power losses asso-ciated with linear regulators. Unfortunately, switchingregulators exhibit much slower transient response thanlinear regulators. This greatly increases the output capaci-tor requirements for switchers.
Circuit Operation
The circuit shown in Figure 21 takes advantage of a new,ultrahigh speed linear regulator combined with a switch-ing regulator to get the best of both worlds. An LTC1435synchronous buck regulator is combined with an LT1575
Figure 21. 12V to 3.3V/9A (14A Peak) Hybrid Regulator
Application Note 84
AN84-19
LOAD CURRENT (A)
50
90
80
70
60
100EF
FICI
ENCY
(%)
14
DI1575_02.eps
0 2 4 6 8 10 12
SWITCHER EFFICIENCY
TOTAL EFFICIENCY
linear regulator to generate a 3.3V output from a 12V inputwith an overall conversion efficiency of approximately72%. The output is capable of current slew rates ofapproximately 20A per microsecond.
The LT1575 uses an IRLZ44 MOSFET as the pass transis-tor, allowing the dropout voltage to be less than 550mV.Setting the switching supply’s output to only 700mVabove the output of the linear regulator ensures outputregulation. The switcher is therefore set up to deliver 4.0Vat 14A from the 12V supply. Conversion efficiency of theswitcher is around 90% (depending on load), whereas theLT1575’s efficiency is 82.5% (see Figure 22). The 12Vinput current is only about 5.5A. At an average current of9A, the power dissipation in the linear pass transistor isonly 6.3W. A small stamped aluminum heat sink is adequate.
Figure 23 shows the transient response to a 10A load stepwith a rise time of approximately 50ns. The only outputcapacitance is 40, 1µF ceramic capacitors. No additionalbulk capacitance is required at the processor. The circuiteliminates approximately a dozen low ESR tantalumcapacitors at the load, which would be required withoutthe linear postregulator. The switching supply’s output isdecoupled with three aluminum electrolytic capacitors.Because the transient response at this point is much lesscritical than at the load, the long-term degradation of thealuminum capacitors will not be as detrimental to thecircuit’s performance as it would be if they were used forload decoupling.
Figure 22. Efficiency of Figure 21’s CircuitFigure 23. Transient Response of Figure 21’s Circuitto a 10A Load Step
200µs/DIV
50mV/DIV
THE LTC1624: A VERSATILE, HIGH EFFICIENCY, SO-8N-CHANNEL SWITCHING REGULATOR CONTROLLERby Randy G. Flatness
Introduction
The LTC1624 is a current mode switching regulator con-troller operating at an internally set frequency of 200kHz.This versatile 8-pin controller uses the same constantfrequency current mode architecture and Burst Modeoperation as the LTC1435–LTC1439 controllers, but with-out the synchronous switch. The LTC1624, like the othermembers of the family, drives a cost-effective, external N-channel MOSFET for the topside switch and maintains lowdropout operation previously available only with P-chan-nel MOSFETs.
The LTC1624 can be configured to operate in all standardswitching configurations, including boost, step-down,inverting, SEPIC and flyback, without a limitation on theoutput voltage. A wide input voltage range of 3.5V to 36Vallows operation from a variety of power sources, from asfew as four NiCd cells up though high voltage wall adapt-ers. Tight load regulation, coupled with a reference voltagetrimmed to 1%, provides very accurate output voltagecontrol.
Application Circuits
The LTC1624 can be used in a wide variety of switchingregulator applications, the most common being the step-down converter. Other switching regulator architecturesdiscussed here include step-up and SEPIC converters.
Application Note 84
AN84-20
+
+
1624_06.eps
LTC1624
SENSE–
ITH/RUN
VFB
CC, 570pFRC5.1k
100pF
GND
VIN
BOOST
TGCB0.1µF
1000pF
M1Si4412DY
L110µH
RSENSE0.05Ω
R235.7k
VOUT3.3V/2A
R120k
CIN22µF35Vx2
COUT100µF10Vx2
VIN4.5V TO 25V
D1MBRS340T3
SW4 5
6
7
8
3
2
1
+
+
1624_08.eps
LTC1624
SENSE–
ITH/RUN
VFB
CC, 330pFRC5k
100pF
GND
1
2
3
4
8
7
6
5
VIN
BOOST
TGCB0.1µF
1000pF
M1Si4412DY
L120µH
D1MBRS130LT3
RSENSE0.04Ω
R235.7k1%
VOUT12V/1A
R13.92k1%
CIN22µF
35V x2
COUT100µF16Vx2
VIN5V
SW
The basic step-down converter is shown in Figure 24. Thisapplication shows a 3.3V/2A converter operating from aninput voltage range of 4.5V to 25V. The efficiency for thiscircuit is shown in Figure 25.
Step-up and SEPIC applications require a low-side switchpulling the inductor to ground (see Figures 26 and 28).Since the source of the MOSFET must be grounded, theswitch pin (SW) on the LTC1624 is also grounded in orderfor the driver to supply a gate-to-source signal to controlthe MOSFET. In these applications, the voltage on theboost pin is a constant 5V, resulting in a 0V–5V gate-drivelevel. A capacitor from boost to switch is still required,since this capacitor supplies the gate-charge currents.
The basic step-up converter is shown in Figure 26. TheLTC1624 is used to create 12V/1A from a 5V source withthe efficiency shown in Figure 27. Efficiency is above 90%from 20mA up to close to full load, dropping only to 89%at 1A.
In order to allow input voltages both above and below theoutput voltage, a SEPIC converter can be used. An exampleof the LTC1624 used as a 12V/0.5A SEPIC converteroperating from an input range of 5V to 20V is shown inFigure 28.
Figure 24. High Performance 3.3V/2A Step-Down DC/DC Converter
Figure 26. 12V/1A Step-Up Converter
LOAD CURRENT
50
60
70
80
90
100
EFFI
CIEN
CY (%
)
10A
1624_07.eps
1mA 1A100mA10mA
VIN = 5V
VIN = 10V
VIN = 20V
Figure 25. Efficiency Plotof Figure 24’s Circuit
Application Note 84
AN84-21
LOAD CURRENT
50
60
70
80
90
100
EFFI
CIEN
CY (%
)1A
1624_09.eps
1mA 100mA10mA
++
+
1624_10.eps
LTC1624
SENSE–
ITH/RUN
VFB
CC, 330pFRC10k
100pF
GND
1
2
3
4
8
7
6
5
VIN
BOOST
TGCB0.1µF
L1a, L1b:CTX50-4
1000pF
M1Si4412DY
L1a
L1b
D1MBRS130LT3RSENSE
0.082Ω
R235.7k1%
VOUT12V/0.5A
R13.92k1%
CIN22µF
35V x2
22µF35V
COUT100µF16Vx2
VIN5V TO 15V
SW
Figure 27. Efficiency Plot for Figure26’s Circuit
Figure 28. 12V/0.5A DC/DC Converter Operates from 5V–15V Inputs
LOW COST 3.3V TO 1.XV 6 AMP POWER SUPPLYby Sam Nork
As voltage requirements for microprocessors drop, theneed for high power DC/DC conversion from a 3.xV supplyto a lower voltage keeps growing. The LTC1430 is a veryattractive choice for such DC/DC applications, due to itslow cost, high efficiency and high output power capability.However, there are two problems: first, 3.xV does notprovide enough gate drive to ensure low RDS(ON) usingexternal logic-level FETs; and second, the LTC1430 has a4V minimum input requirement. These obstacles are bothovercome by using an LTC1517-5 regulated charge pumpto generate the input voltage for the LTC1430.
The circuit shown in Figure 29 uses the LTC1430 toproduce a synchronous 3.3V to 1.9V step-down DC/DCconverter. The circuit achieves 90.5% efficiency at 3 ampsof output current and has a 6 amp maximum outputcapability. (Refer to the LTC1430 data sheet for detaileddescription of LTC1430-based designs). Power for theLTC1430 is derived from the output of the LTC1517-5.
The LTC1517-5 is a switched capacitor charge pumpavailable in a tiny, 5-pin SOT-23 package. The part usesBurst Mode operation to generate a 5V output from a 2.7Vto 5V input.The regulated 5V supply powers the internalcircuitry of the LTC1430 and ensures that the LTC1430 can
Application Note 84
AN84-22
LOAD CURRENT (A)
40
70
100
90
80
50
60EFFI
CIEN
CY (%
)
10
1517_02.EPS
0.1 1
VIN = 3.3VVOUT = 1.9V
provide adequate gate drive to the external N-channelFETs. With insufficient gate drive, output power and effi-ciency will be significantly reduced due to high RDS(ON) ofthe FETs. In this circuit, typical supply current drawn by theLTC1430 is between 25mA and 30mA, the vast majority ofwhich is needed to charge and discharge the external FETs.Because the LTC1517-5 has a maximum effective outputimpedance of 50Ω, this current can be comfortably sup-plied from a 3.3V input. If the input voltage drops to 3V orlower, the LTC1517-5 output may also drop. However,with the FETs shown in Figure 29, the LTC1517-5 willprovide a 4.5V minimum supply to the LTC1430 at inputvoltages down to 3V. The circuit’s efficiency is shown inFigure 30.
Pulling the SHDN pin on the LTC1430 low will shut downthe power supply. Q1 and Q2 will be forced off and theLTC1430 quiescent current will drop to 1µA. Although theLTC1517-5 does not have a shutdown feature, the no-loadoperating current is an extremely low 6µA. This keeps the
LTC1517-5
LTC1430CS
G1PVCC1PGNDGNDSENSE–
FBSENSE+
SHDN
G2PVCC2
VCCIFB
IMAXFREQSET
COMPSS
12345678
161514131211109
C30.22µF
51
2
3 4
C6 TO C9*330µF6.3V× 4
C1510µF10VC13
390pF
C140.012µFC12
0.1µF
C160.018µF
VOUT1.9V6A
C40.1µF
5V
C51µF
D1BAT54
D2MBRS120
R54.99K1%
R610K1%
L12.4µH, 8ASUMIDA
CDRH127-2R4
C111µF
Q1Si4410
Q2Si4410
R224k
R45.2k
C100.1µF
C1Y5V CERAMIC
3.3µF
VIN3.3V
C2Y5V CERAMIC
10µF
+
C17 TO C21 *330µF6.3V× 5
*AVX TPS TANTALUM(207) 282-5111
1517 TA03
+
R1100Ω
R31k
ON OFF +
overall shutdown current below 10µA plus external FETleakage. (For further reductions in shutdown current, an 8-pin LTC1522 may be used in place of the LTC1517-5; theLTC1522 is the same as an LTC1517-5 with shutdown.)The additional LTC1517-5 circuitry will not take up muchboard space. The entire circuit consumes only 0.045 in2.
Figure 29. 3.3V to 1.9V/6A Power Supply
Figure 30. Efficiency of Figure 29’s Circuit
Application Note 84
AN84-23
THE LT1374: NEW 500kHz, 4.5A MONOLITHICBUCK CONVERTERby Karl Edwards
Introduction
The LT1374 is a 4.5A buck converter using an on-chip80mΩ switch. With its 500kHz operating frequency andintegral switch, only a few external, surface mount compo-nents are required to produce a complete switching regu-lator. The LT1374’s features include current mode control,external synchronization and a low current (typically 20µA)shutdown mode. Improvements have been made to reducestart-up headroom and switching noise. A novel powerdevice layout makes it possible to fit a high speed, bipolar,80mΩ switch into a surface mount SO-8 package. TheLT1374 is also available in DD and TO-220 packages forhigher power applications.
Application: 5V/4.25A Buck Converter
With its 25V input and 4.5A minimum switch current, theLT1374 will fit into a wide range of applications. Figure 31shows a typical buck converter with a 6V to 25V inputrange, a 5V output and 4.25A of output current capability.Due to the low on-resistance of the switch, efficiencyremains high over a wide range of currents, as shown inFigure 32. To reduce power dissipation, both the BIAS pinand boost circuit are supplied from the 5V output.
BOOST
LT1374-5
VINOUTPUT**5V/4.25A
* RIPPLE CURRENT RATING > IOUT/2** L1 = COILTRONICS UP2-4R7; (561) 241-7876
INCREASE L1 TO 10µH FOR LOAD CURRENTS ABOVE 3.5A AND TO 20µH ABOVE 4A
INPUT6V TO 25V
1374_02.EPS
C20.27µF
CC3.3nF
D1MBRS330T3
C1100µF, 10VSOLIDTANTALUM
C3*10µF TO
50µF
D21N914
L1**5µH
VSW
FB
BIAS
GND VCSHDN
R156k
R233k
+
+
LOAD CURRENT (A)0
EFFI
CIEN
CY (%
) 90
95
100
1.5 2.5
1374_03
85
80
0.5 1.0 2.0 3.0 3.5
75
70
Several factors, including maximum current, core andcopper losses, size and cost, affect the choice of inductor,L1. A high value, high current inductor gives the highestoutput current with the lowest ripple, at the expense of alarge physical size and cost. Lower inductance values tendto be physically smaller, have higher current ratings andare cheaper, but output ripple current, and hence ripplevoltage, increases.
The input capacitor, C3, experiences very high ripplecurrents, up to IOUT/2, so low ESR tantalum capacitors areneeded. At 4.25A output current, two capacitors in parallelare required to meet the ripple current requirement. Theripple current in the output capacitor, C1, is lower, but itsESR still needs to be low to limit output voltage ripple. Thevoltage drop across the catch diode, D1, has a significanteffect on overall converter efficiency, especially at higherinput voltages when the switch duty cycle is low. Its abilityto survive short-circuit conditions may increase its powerrating. For good electrical performance, D1 must be placedclose to the LT1374. The power dissipated in D1 will raisethe PC board’s temperature around the LT1374. This mustbe taken into account when modeling or taking benchmeasurements of die temperature.
Figure 31. 5V Buck ConverterFigure 32. Efficiency of Figure 31’s Circuit:10V In, 5V Out
Application Note 84
AN84-24
The loop compensation capacitor, CC, produces a pole inthe frequency response at 240Hz. Unity-gain phase mar-gin can be further improved with the addition of a resistor,typically 2k, in series with CC, adding a zero to thefrequency response. This, however, can cause a large-signal subharmonic problem in the loop. The output ripplevoltage feeds back through the error amplifier to the VCpin, changing the current trip point of the next cycle. Thischanges the voltage ripple at the output, and the loop isclosed. Adding a second capacitor directly from the VC pinto ground to form a pole at one-fifth the switching fre-quency solves the problem.
PCB Layout
All high current, high speed circuits require careful layoutto obtain optimum performance. When laying out thePCB, keep the trace length around the high frequencyswitching components as short as possible. This mini-mizes the EMI and RFI radiation from the loop created bythis path. These traces have a parasitic inductance ofapproximately 20nH/inch, which can cause an additionalproblem at higher operating voltages. At switch-off, thecurrent flowing in the trace inductance causes a voltagespike. This is in addition to the input voltage across theswitch transistor. At higher currents, the additional volt-age can potentially cause the output switching transistorto exceed its absolute maximum voltage rating.
LTC1504: FLEXIBLE, EFFICIENTSYNCHRONOUS SWITCHING REGULATORCAN SOURCE OR SINK 500mAby Dave Dwelley
Introduction
The LTC1504 is an 8-pin step-down switching regulator.It consists of a 200kHz fixed frequency, voltage-feedback,buck-mode switching regulator controller and a pair of1.5Ω power switches in an 8-pin SO package. The LTC1504also includes a synchronous rectifier on-chip, maximizingefficiency and minimizing external parts count while al-lowing the output to both sink and source current: it cansource or sink up to 500mA with input voltages from 3.3Vto 10V and output voltages as low as 1.26V. The LT1504can achieve 100% duty cycle at the output switch, maxi-mizing dropout performance with low input-to-outputvoltage differentials. The LTC1504 includes an onboardprecision reference and user-programmable current-limit and soft-start circuits, allowing implementation offull-featured power conversion circuits with a minimum ofexternal components.
The LTC1504 architecture is optimized for maximumefficiency at loads above 50mA and does not include alight-load Burst Mode™ circuit. This penalizes efficiencyat very light loads but allows the device to seamlessly shiftbetween sourcing and sinking current, opening up a wholenew class of applications. A micropower shutdown mode
is included. The diminutive SO-8 package minimizes theamount of space the LTC1504 fills while allowing adequatethermal dissipation for 500mA load current levels. TheLTC1504 allows previously impossible (or at least awk-ward) tasks to be completed with ease.
Minimum Component-Count Circuits
Figure 33 shows a fully functional LTC1504 5V to 3.3Vregulator, including current limit and soft-start, using thefixed-output LTC1504-3.3 and only six external compo-nents. Efficiency is above 90% with load currents between
+
+
IMAX SHDN
VCC
GND
SW
SENSE
SS COMP
LTC1504-3.3
SHUTDOWN
VIN5V
CIN22µF
RIMAX**68k
CC1000pF
CSS*O.1µF
COUT100µF
LEXT 47µHVOUT3.3V/500mA
CIN = AVX TPSC226M016R0375COUT = SANYO 16CV100GXLEXT = SUMIDA CD54-470
* OPTIONAL: DELETE TO DISABLE SOFT START** OPTIONAL: DELETE TO DISABLE CURRENT LIMIT
1504_01.EPS
Figure 33. Minimum Parts-Count 5V–3.3V Converter
Application Note 84
AN84-25
50mA and 200mA, peaking at 92% at 100mA and remain-ing above 82% all the way to the maximum 500mA load.Current limit is set at 500mA in this example; it can bereduced by lowering the value of RIMAX. CSS sets the start-up time at approximately 25ms.
The circuit in Figure 33 relies on the ESR of the outputcapacitor to maintain loop stability with just a singlecapacitor at the COMP pin. Figure 33 uses a surface mountelectrolytic capacitor with about 400mΩ ESR. A low ESRtantalum output capacitor can improve the transientresponse at the output but requires a more complexcompensation network at the COMP pin (Figure 34). Thereis a tradeoff to be made here: the minimum componentcount solution is the simplest and uses the least expensivecomponents but pays a penalty in transient response. Thelow ESR circuit in Figure 34 has improved transientresponse and actually uses less board space: the tantalumoutput capacitor is smaller than the electrolytic deviceused in Figure 33 and the additional compensation compo-nents are tiny 0603 surface mount devices.
Note that the input bypass capacitor in both Figures 33 and34 is an AVX TPS type, a relatively costly surge-testedtantalum capacitor. This is a small, surface mount devicethat has a surge current rating adequate to support the500mA maximum load current of the LTC1504. Buckregulators (like the LTC1504) inherently draw large RMScurrents from the input bypass capacitor, and the capacitortype chosen must be capable of withstanding this current
without overheating. As with all switching regulator cir-cuits, layout is critical to obtaining maximum perfor-mance; if in doubt, contact the LTC Applications Depart-ment for component selection and layout advice.
Sink/Source Capability Improves SCSI Terminatorsand Supply Splitters
Figure 35 shows an adjustable-output LTC1504 con-nected as a 2.85V regulator for use as a SCSI terminator.The ability of the LTC1504 circuit to sink current makes itideal for use in terminator applications, where the load isjust as likely to be putting current into the regulator astaking it out. The synchronous-buck architecture of theLTC1504 allows it to shift cleanly between sourcing andsinking current, making it ideal for such applications. Thesmall number of tiny external components requiredminimizes the space used by the terminator circuit. A lowESR output capacitor is used along with an optimizedcompensation network to improve output transientresponse and maintain maximum data fidelity.
+
+
IMAX SHDN
VCC
GND
SW
SENSE
SS COMP
LTC1504-3.3
SHUTDOWN
VIN5V
RIMAX68k
CC0.01µF
CF220pF
CSSO.1µF
VOUT3.3V/500mA
CIN = AVX TPSC226M016R0375COUT = AVX TPSE107M016R0125LEXT = SUMIDA CD54-470
1504_02EPS
RC7.5k
CIN22µF
COUT100µF
LEXT 47µHIMAX SHDN
SS COMP
VCC
GND
SW
FB
10µFCERAMIC
TERMPWR
7.5k
0.01µF
220pF
15k
12k
+
COUT = AVX TPSC107M006R0150LEXT = SUMIDA CD54-470 1540_03.EPS
LTC1504
110Ω
110Ω
110Ω
110Ω 18TO27LINES
IMAX SHDN
VCC
GND
SW
SENSE
SS COMP
LTC1504
COUT100µF
LEXT 47µH
IMAX SHDN
SS COMP
VCC
GND
SW
FB
10µFCERAMIC
5V
7.5k
0.01µF
220pF
11.8k
12.1k
+
COUT = TAJC476M016RLEXT = SUMIDA CDRH73-470 (LOWER RIPPLE/HIGHER EFFICIENCY)
CDRH73-220 (FASTER TRANSIENT RESPONSE)
1540_04.EPS
LTC1504
IMAX SHDN
VCC
GND
SW
SENSE
SS COMP
LTC1504
SHUTDOWN
SPLIT SUPPLY2.5V ±500mA
COUT47µF
LEXT 47µH (22µH)*
*
Figure 34. Improved Transient Response
Figure 35. SCSI-2 Active Terminator
Figure 36. 5V Supply Splitter
Application Note 84
AN84-26
HIGH EFFICIENCY DISTRIBUTEDPOWER CONVERTER FEATURES SYNCHRONOUSRECTIFICATIONby Dale Eagar
Introducing the LT1339
The LT1339 is the buck/boost converter that needs nosteroids. As a full-featured switching controller, the LT1339incorporates the features needed for system-level solu-tions. The LT1339 has an innovative slope-compensationfunction that allows the circuit designer freedom in con-trolling both the slope and offset of the slope-compensa-tion ramp. Additionally, the LT1339 has an average cur-rent limit loop that yields a constant output current limit,regardless of input and/or output voltage. The LT1339’sRUN pin is actually the input to a precision comparator,giving the designer freedom to select an undervoltagelockout point and hysteresis appropriate for the design.The SYNC and SS (soft-start) pins allow simple solutionsto system-level design considerations. Like all LinearTechnology controllers, the LT1339 has anti-shoot-through circuitry that ensures the robustness that isdemanded in real-world applications for medium and highpower conversion.
For input voltages ranging from 12V to 48V and outputvoltages ranging from 1.3V to 36V, the LT1339 is a simple,robust solution to your power-conversion problems. TheLT1339 is ideal for power levels ranging from tens of wattsto tens of kilowatts. The LT1339 is straightforward andremarkably easy to use. This is one power converter that’snot afraid of 20A, 50A or even 150A of load current.
Distributed Power
Figure 36 details a typical low voltage buck converter. Thiscircuit has a VIN range of 10V to 18V with configurableoutput current and voltage. This simple circuit delivers250W of load power into a 5V load while maintainingefficiencies in the mid-nineties.
Higher Input Voltages
The circuit shown in Figure 37 is limited to 20V because ofthe maximum rating (Abs Max) of the LT1339 VIN pin. Theinput voltage can be extended above 20V by inserting a10V Zener diode where the asterisk (*) is shown in Figure37. This will extend the input voltage of Figure 37’s circuitup to 30V (the Abs Max rating of the MOSFETs).
Substituting a different set of feedback resistors (Figure35) creates a 5V supply splitter, which creates a 2.5V“ground” to allow analog circuitry to operate from splitsupplies. Op amp circuits and data converters like to
operate from dual supplies, and the sink/source capabilityof the LTC1504 allows load currents to be returned directlyto the 2.5V “ground” supply.
+
1 8
C20.1µF
C11µF
15
RREF1k
*SEE TEXT1339_01.EPS
SYNC
IAVG
VC
VREF
SENSE+
SENSE–
FB
TG
TS
BG
13
2
4
3
5
7
10
19
18
16
17 20
11
12
9
RUN
5VREF
SLOPE
CT
SGND
12VIN BOOST
D1 1N914
LT1339
PGNDCCOMP2200pF
RCOMP4.7k
R1100k
*
10V TO18V
RT15k
CAVG 2200pF
CT 1500pF
C31µF
COUTPUT2200µF6.3VOS-CON
5V50A
Q1IRL3803
Q2–Q5IRL3103D2×4
L110µH50A
RFB3k
RSENSE0.002Ω
D31N5817
D21N5817
+ CINPUT1000µF16V ×2OS-CON
RFB3K
1.66K1.25K450Ω40Ω
VOUT5V
3.3V2.8V1.8V1.3V
RSENSE
0.01Ω0.005Ω0.002Ω
ILIMIT
10A20A50A
Figure 37. 10V–18V In, 5V/50A Out Buck Converter
Application Note 84
AN84-27
+
1 8
C30.1µF
C11µF
15
R110k
D3 1N914
1339_02.EPS
SYNC
IAVG
VC
VREF
SENSE+
SENSE–
FB
TG
TS
BG
13
2
4
3
5
7
10
19
18
16
11
12
9
RUN
5VREF
SLOPE
CT
SGND
12VIN BOOST
D11N914 48V
LT1339
PGNDC22200pF
RCOMP4.7k
R1100k
12V(SUPPLIED SEPARATELY)
RT15k
CAVG 2200pF
CT1500pF
C31µF
C41µF
COUT2200µF 6.3VOS-CON×4
5V50A
Q1–Q2IRFZ44×2
Q3–Q6IRFZ44×4
L110µH 50A
RFB
3k
RSENSE0.002Ω
Q7FMMT619
Q9FMMT619
Q10FMMT720
Q8FMMT720
RREF1k
+ C547µF
17 20
+ CINPUT1500µF63V×4
RFB18.2K8.66K
3K1.66k1.25k
VOUT24V12V5V
3.3V2.8V
RSENSE
0.01Ω0.005Ω0.002Ω
ILIMIT
10A20A50A
12V D2 3.3V
D4 TO D143A ×10
VIN15V–25V
CIN1000µF 35V
×2
R132k
D112V
Q7 FZT849
C100.1µF
C1122µF35V
C10.1µF
R1100k
RT47k
R21k
R3560Ω
CT230pF
CAVG2.2nF
R433k
C4220pF
U3CNY17-3
RSENSE0.02Ω1/2W
D2IN4148
C31µF
12V
12V
Q1
D4MURS120
D3 MURS120
Q2
Q3 Si4450
Q4 Si4450C3
3.3µFR1010Ω
R1110Ω
R910ΩR14 10Ω
L1 7µH
Q6
T2**
T1*
10
8
6
9
4
2
R81k
R12 1k
D6 1N914D51N914
C7 1µF
C51µF
C9 0.1µF
C61µF
R5
3.3Ω
U2LT1431
COLL REF
GND-S GND-F
V+
1
5 6
3
8
RFB2.49k
RREF2.49k
R7 1k
R6 560Ω
COUT220µF 10VOS-CON
VOUT5V/6A11
12
20
19
18
SENSE+
SENSE–
BOOST
TG
TS
16
7
BG
VC
2
4
3
6
5
9
14
10
5VREF
SLOPE
CT
SS
IAVG
FB
PHASE
VREF
SYNC SGND PGND
U1LT1339
1 8 15
RUN 12VIN
13 17
ISOLATIONBARRIER
PRIMARYGROUND
SECONDARYGROUND
Q1, Q2 = SUD50N03L1 = 15 TURNS AWG20 77130-A7
T1 = POWER TRANSFORMERT2 = GATE-DRIVE TRANSFORMER(SEE FIGURE 4 FOR DETAILS)
Q5
***
Si4539DY
+
+
+
Figure 38. 48V In, 5V/50A Out, High Power Buck Converter
Figure 39. Galvanically Isolated Synchronous Forward Converter (see Figure 40 for Details of T1 and T2)
Application Note 84
AN84-28
Blame it on the Physicists
As the input voltage approaches 30V, the bottom MOS-FETs will begin to exhibit “phantom turn-on.” Thisphenomenon is driven by the instantaneous voltage stepon the drain, the ratio of CMILLER to CINPUT, and yields
localized gate voltages above VT, the threshold voltage ofthe bottom MOSFET. To defeat the physicists, we add 3.3Vof negative offset to the bottom gate drive, effectivelymaking the threshold of the bottom MOSFETs 3.3V harderto reach (see Figure 38). This offset is provided by the 3.3VZener, 1µF capacitor, 10k resistor and the 1N914 diodepreceding the gate of the bottom MOSFETs.
The Synchronous Forward Converter
Figure 39 details a Galvanically isolated LT1339 synchro-nous forward converter. Operating at its rated load of 6Vat 5A, this circuit achieves 87% efficiency with a 15V inputand 85% efficiency with a 24V input. Figure 40 showsdetails of the transformers used in Figure 39’s circuit.
The Synchronous Boost Converter
The LT1339 becomes a synchronous boost controllerwhen the PHASE pin is grounded. Figure 41 details a 250Wboost converter that outputs 28V at 9A from a 5V supply.
SECONDARY, 9 TURNS TRIFILAR 26AWG
PRIMARY, 9 TURNS TRIFILAR 26AWG
2MILPOLYESTERFILM
1500VDC ISOLATIONTUCK TAPE ENDS
T1: PHILIPS EFD20-3F3 CORELp = 93µH, Al = 1150nH/T2 (NO GAP)
T2: COILTRONICS VP1-1400 (500V ISOLATION)
1339 04 .eps
4
1
5
2
10
7
11
8
6
3
12
9
BOOST 12VIN
Q1 TO Q2IRF3205 ×2
Q3 TO Q6IRF3205 ×4
Q8 FMMT619Q7 FMMT720
C31µF
C41µF
Q10FMMT720
Q9FMMT619
L140µH
RSENSE0.002Ω
CINPUT220µF 6.3V
×4
VIN5V/60A
R2 100Ω
R3 100Ω
12V
12V (SUPPLIED SEPARATELY)D11N914
R1100k
C747µF16V
COUT2200µF35V ×6
RFB27k
RREF1.2k
RT10k
C11µF
CT2200pF
RCOMP
7.5k
CAVG2.2nF
CCOMP1.5nFCSS
10µFC60.1µF
TG
TS
BG
SENSE+
SENSE–
VREF
PHASE
CT
VC
IAVG
SS
5VREF
SLOPE
FB
RUN
U1LT1339
SYNC PGND SGND
19
18
16
12
11
10
14
3
7
5
6
9
2
4
13
20 17
1 15 8
VOUT28V/8.5A
L1 = 12T 4× AWG12 ON 77437-A7
C510pF
+
+ ++
+
Figure 40. Transformer Details of Figure 39’s Circuit
Figure 41. This 5V to 28V Synchronous Boost Converter Limits Input Current at 60A (DC)
Application Note 84
AN84-29
FIXED FREQUENCY, 500kHz, 4.5A STEP-DOWNCONVERTER IN AN SO-8 OPERATES FROM A 5V INPUTby Karl Edwards
Introduction
The LT1506 is a 500kHz monolithic buck mode switchingregulator, functionally identical to the LT1374 but opti-mized for lower input voltage applications. Its high 4.5Aswitch rating makes this device suitable for use as theprimary regulator in small to medium power systems. Thesmall SO-8 footprint and input operating range of 4V to15V is ideal for local onboard regulators operating from 5Vor 12V system supplies. The 4.5A switch is included on thedie, along with the necessary oscillator, control and logiccircuitry to simplify design. The part’s high switchingfrequency allows a considerable reduction in the size ofexternal components, providing a compact overall solution.
The LT1506 is available in standard 7-pin DD and fused-lead SO-8 packages. It maintains high efficiency over awide output current range by keeping quiescent supplycurrent to 4mA and by using a supply-boost capacitor tosaturate the power switch. The topology is current modefor fast transient response and good loop stability. Fullcycle-by-cycle short-circuit protection and thermal shut-down are provided. Both fixed 3.3V and adjustable outputvoltage parts are available.
5V to 3.3V Buck Converter
The circuit in Figure 42 is a step-down converter suitablefor use as a local regulator to supply 3.3V logic from a 5Vpower bus. The high efficiency, shown in Figure 43, re-moves the need for bulky heat sinks or separate powerdevices, allowing the circuit to be placed in confinedlocations. Since the boost circuit only needs 3V to operate,the boost diode can still be connected to the output,improving efficiency. Figure 42’s circuit shows the shut-down pin option. If this pin is pulled to a logic low, theoutput is disabled and the part goes into shutdown mode,reducing supply current to 20µA. An internal pull-up ensurescorrect operation when the pin is left open. The SYNC pin,an option for the DD package, can be used to synchronizethe internal oscillator to a system clock. A logic-level clocksignal applied to the SYNC pin can synchronize the switch-ing frequency in the range of 580kHz to 1MHz.
Current Sharing Multiphase Supply
The circuit in Figure 44 uses multiple LT1506s to producea 5V, 12A power supply. There are several advantages tousing a multiple switcher approach compared to a singlelarger switcher. The inductor size is considerably reduced.Inductor size is proportional to the energy that needs to bestored in the core. Three 4A inductors store less energy(1/2Li2) than a single 12A coil, so they are much smaller.In addition, synchronizing three converters 120° out ofphase with each other reduces input and output ripplecurrents. This reduces the ripple rating, size and cost ofthe filter capacitors.
BOOST
LT1506-3.3
VIN
OUTPUT3.3V4A
INPUT5V
1506 TA01
C20.68µF
CC1.5nF D1
MBRS330T3
C1100µF, 10VSOLIDTANTALUM
C310µF TO
50µFCERAMIC
D21N914
L15µH
VSW
SENSESHDN
OPENOR
HIGH= ON GND VC
+
+ EFFI
CIEN
CY (%
)
0 4LOAD CURRENT (A)
21 3
90
85
80
75
70
Figure 42. 5V to 3.3V Step-Down Converter Figure 43. Efficiency vs Load Current for Figure 42’s Circuit
Application Note 84
AN84-30
Current Sharing/Split Input Supplies
Current sharing is accomplished by connecting the VCpins to a common compensation capacitor. The output ofthe error amplifier is a gm stage, so any number of devicescan be connected together. The effective gm of the com-posite error amplifier is the product of the individualdevices. In Figure 44, the compensation capacitor, C4, hasbeen increased by 3×. Tolerances in the reference voltagescause small offset currents to flow between the VC pins.The overall effect is that the loop regulates the output at avoltage somewhere between the minimum and maximumreferences of the devices used. Switch-current matchingbetween devices will be typically better than 300mA overthe full current range. The negative temperature coefficientof the VC-to-switch-current transconductance preventscurrent hogging.
A common VC voltage forces each LT1506 to operate at thesame switch current, not at the same duty cycle. Eachdevice operates at the duty cycle defined by its inputvoltage. This is a useful feature in a distributed powersystem. The input voltage to each device could vary due todrops across the backplane, copper losses, connectorsand so on. The common VC signal ensures that loading isstill shared between the devices.
Synchronized Ripple Currents
A ring counter generates three synchronization signals at600kHz, 33% duty cycle, phased 120° apart. The syncinput will operate over a wide range of duty cycles, so nofurther pulse conditioning is needed. At full load, eachdevice’s input ripple current is a 4A trapezoidal wave at600kHz, as shown in Figure 45. Summing these wave-forms gives the effective input ripple for the completesystem. The resultant waveform, shown at the bottom ofFigure 45, remains at 4A but its frequency has increasedto 1.8MHz. The higher frequency eases the requirementson the value of input filter without the 3× increase in ripplecurrent rating that would normally occur. Although only asingle input capacitor is required, practical layout restric-tions usually dictate an individual capacitor at each device.Figure 46 shows the output ripple current waveforms. Theresultant 1.8MHz triangular waveform has a maximumamplitude of 350mA at an input voltage of 10V. This issignificantly lower than would be expected for a 12Aoutput. Interestingly, at inputs of 7.6V and 15V, thetheoretical summed output ripple current cancels com-pletely. To reduce board space and ripple voltage, C1 andC3 are ceramic capacitors. Loop compensation capacitorC4 must be adjusted when using ceramic output capaci-tors, due to the lack of effective series resistance (ESR).
+
+
C468nF25V
C1, C3: MARCON THCS50E1E106ZD1: ROHM RB051L-40D2: 1N914L1: DO3316P-682
+ C3C10µF25V
C2C330nF10V
D1C
D2C
1506 F15
L1C6.8µH
+ C110µF25V
5V12AR1
5.36k1%
R24.99k1%
1.8MHz
3-BIT RINGCOUNTER
+
+ C3B10µF25V
C2B330nF10V
D1B
D2BL1B6.8µH
+
+ C3A10µF25V
INPUT6V TO 15V
C2A330nF10V
D1A
D2AL1A6.8µH
VC SYNC SW GND
LT1506-SYNC
VIN BOOST FBVC SYNC SW GND
LT1506-SYNC
VIN BOOST FBVC SYNC SW GND
LT1506-SYNC
VIN BOOST FB
Figure 44. Current-Sharing 5V/12A Supply
Application Note 84
AN84-31
The typical tantalum compensation value of 1.5nF isincreased to 22nF (×3) for the ceramic output capacitor. Ifsynchronization is not used and the internal oscillatorsfree run, the circuit will operate correctly, but ripplecancellation will not occur. Input and output capacitorsmust be ripple rated for the individual output currents.
TIME
CURR
ENT
TIME
CURR
ENT
TIME
CURR
ENT
TIME
CURR
ENT
PHASE 1
PHASE 2
PHASE 3
TOTAL
TIMECU
RREN
T
TIME
CURR
ENT
TIME
CURR
ENT
TIME
CURR
ENT
PHASE 1
PHASE 2
PHASE 3
TOTAL
Redundant Operation
The circuit shown in Figure 44 is fault tolerant whenoperating at less than 8A of output current. If one powerstage fails open circuit, the output will remain in regula-tion. The feedback loop will compensate by raising thevoltage on the VC pin, increasing the switch current of thetwo remaining devices.
Figure 45. Input Current Figure 46. Output Current
Application Note 84
AN84-32
VID VOLTAGE PROGRAMMERFOR INTEL MOBILE PROCESSORSby Peter Guan
Figure 47 shows a VID-programmed DC/DC converter foran Intel mobile processor that uses the LTC1435A andLTC1706-19 to deliver 7A of output current with a pro-grammable VOUT of 1.3V to 2.0V from a VIN of 4.5V to 22V.Simply connecting the LTC1706-19’s FB and SENSE pinsto the LTC1435A’s VOSENSE and SENSE– pins, respec-tively, closes the loop between the output voltage senseand the feedback inputs of the LTC1435A regulator withthe appropriate resistive divider network, which is con-trolled by the LTC1706-19’s four VID input pins.
Each VID pin must be grounded or driven low to producea digital low input, whereas a digital high input can begenerated by either floating the VID pin or connecting it toVCC. The LTC1706-19 is fully TTL compatible and opera-tional over a VID input voltage range that is much higherthan VCC.
Table 2 shows the VID inputs and their correspondingoutput voltages. VID3 is the most significant bit (MSB) andVID0 is the least significant bit (LSB). When all four inputsare low, the LTC1706-19 sets the regulator output voltageto 2.00V. Each increasing binary count is equivalent todecreasing the output voltage by 50mV. Therefore, to
+
+ +
M1Si4410DY
M2 Si4410DY
CIN10µF, 30V×2
L1 3.3µHDB*
RSENSE0.015Ω
COUT820µF4V×2
D1MBRS
-140T3
4.7µF
1000pF
0.22µF
COSC 43pF
CSS0.1µF
CC1000pF
CC2220pF
51pF
VIN
TG
SW
INTVCC
BOOST
BG
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
13
16
14
12
15
11
10
1
2
3
5
6
VOUT1.30V TO2.00V/7A
VIN4.5V TO 22V
SENSE– SENSE+
7 8
LTC1435A
*DB = CMDSH-3
RC10k
RF4.7Ω
CF0.1µF
VCC
FB
VID0
VID1
VID2
VID3 GND
SENSE
LTC1706-19
FROM µP
L1: COILCRAFT D05022P-332HC
7
8
1
2
VID0
VID1
VID2
VID3
6
5
SENSE
FBGND
VCC
4
3LTC1706-19
0.1µF
VCC 2.7V TO 5.5V
+
+
1
2
3
4
8
7
6
5
SENSE–
ITH/RUN
VFB
GND
VIN
BOOST
TG
SW
LTC1624
470pF
6.8k
100pF
1000pF
0.1µF
RSENSE0.033Ω
Si4412DY
L1 10µH
MBRS340T3
CIN22µF35V×2
VOUT1.3V–3.0V
COUT100µF10V×2
VIN4.8V TO 20V
L1: SUMIDA CDRH125-10
Figure 47. Intel Mobil Pentium II VID Power Converter
Figure 48. High Efficiency SO-8, N-Channel 3A Switching Regulator with Programmable Output
Application Note 84
AN84-33
+
+ +
M1Si4412DY
M2 Si4412DY
CIN22µF, 35V×2
L13.3µH
DB*
RSENSE0.02Ω
COUT100µF10V×2
D1MBRS
-140T3
4.7µF
1000pF
0.22µF
COSC 39pF
CSS0.1µF
CC510pF
100pF
VIN
TGL
TGS
SW
INTVCC
BOOST
BGL
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
18
21
19
20
17
22
16
15
2
3
4
6
8
VOUT1.30V–2.00V/5A
VIN4.5V–22V
SENSE– SENSE+
9 10
LTC1436A-PLL
*DB = CMDSH-3
RC10k
VCC
FB
VID0
VID1
VID2
VID3 GND
SENSE
LTC1706-19
FROM µP
M3IRLML2803
PLL LPF PLLIN
10k 0.1µF
EXTERNALFREQUENCY
SYNCHRONIZATION
1 24
edoC 3DIV 2DIV 1DIV 0DIV tuptuO
0000 DNG DNG DNG DNG V00.2
1000 DNG DNG DNG taolF V59.1
0100 DNG DNG taolF DNG V09.1
1100 DNG DNG taolF taolF V58.1
0010 DNG taolF DNG DNG V08.1
1010 DNG taolF DNG taolF V57.1
0110 DNG taolF taolF DNG V07.1
1110 DNG taolF taolF taolF V56.1
0001 taolF DNG DNG DNG V06.1
1001 taolF DNG DNG taolF V55.1
0101 taolF DNG taolF DNG V05.1
1101 taolF DNG taolF taolF V54.1
0011 taolF taolF DNG DNG V04.1
1011 taolF taolF DNG taolF V53.1
0111 taolF taolF taolF DNG V03.1
Table 2. VID Inputs and Coresponding Output Voltages Figure 48 shows a combination of the LTC1624 and theLTC1706-19 configured as a high efficiency step-downswitching regulator with a programmable output of 1.3Vto 2.0V from an input of 4.8V to 20V. Using only oneN-channel power MOSFET, the two SO-8 packaged LTCparts offer an extremely versatile, efficient, compact regu-lated power supply.
Figure 49 shows the LTC1436A-PLL and the LTC1706-19,a combination that yields a high efficiency low noisesynchronous step-down switching regulator with pro-grammable 1.3V to 2V outputs and external frequencysynchronization capability.
Besides the LTC family of 1.19V-referenced DC/DC con-verters, the LTC1706-19 can also be used to program theoutput voltages of regulators with different onboard refer-ences. Figure 50 shows the LTC1706-19 programmingthe output of the LT1575, an UltraFast™ transient response,low dropout regulator that is ideal for today’s power-hungry desktop microprocessors. However, since theLT1575 has a 1.21V reference instead of a 1.19V refer-ence, the output will range from 1.27V to 2.03V in steps of50.8mV.
obtain a 1.30V output, the three MSBs are left floatingwhile only VID0 is grounded. In cases where all four VIDinputs are tied high or left floating, such as when noprocessor is present in the system, a regulated 1.25Voutput is generated at VSENSE.
Figure 49. High Efficiency, Low Noise, Synchronous Step-Down Switching Regulator with Adjustable Output Voltage
Application Note 84
AN84-34
+
7
8
1
2
VID0
VID1
VID2
VID3
1
2
3
4
8
7
6
5
SHDN
VIN
GND
FB
IPOS
INEG
GATE
COMP
6
5
SENSE
FBKGND
VCC
4
3LTC1706-19
1µF
VCC 3.3V
VIN12V
10pF7.5k
1000pF
24µF
5.1Ω
220µFIRFZ24
3.3V
VOUT1.27V TO 2.03IN 50.8mV STEPS
LT1575
Figure 50. UltraFast Transient Response, Low Dropout Regulator with Adjustable Output Voltage
NEW DC/DC CONTROLLER ENABLESHIGH STEP-DOWN RATIOSby Greg Dittmer
Capabilities of the LTC1435
The LTC1435 high efficiency synchronous DC/DC control-ler has been extremely popular for notebook computersand other battery-powered equipment due to its low noise,constant-frequency operation and its dual N-channel drivefor outstanding high current efficiency without sacrificinglow dropout operation. However, its 400ns to 500nsminimum on-time requires lower operating frequencies(<150kHz) to regulate output voltages below 2.0V if VIN is
high. This occurs because tON = VOUT/(VIN • f); thus, at lowduty ratios, frequency must be decreased to keep tON >tON(MIN). Lowering the operating frequency is usually notdesirable because it increases noise and componnentsize.
What happens if minimum on-time is violated in theLTC1435? If VIN is increased so that the on-time fallsbelow tON(MIN), the LTC1435 will begin to skip cycles toremain in regulation. During this “cycle-skipping” mode,the output remains in regulation but the operating fre-quency decreases, causing the inductor ripple current andoutput ripple voltage to increase.
OUTPUT VOLTAGE (V)1.25 1.5
MAX
IMUM
VIN
(V)
2.0 2.5
35
30
25
20
15
10
5
0
AN70 F52
1.75 2.25
f = 250kHzILOAD = 0AL = 4.7µHT = 25°C
LTC1435
LTC1435A
MOSFET VDS LIMIT
INDUCTOR RIPPLE CURRENT (% OF IMAX)0
MIN
IMUM
ON-
TIM
E (n
s)
40 70
400
350
300
250
200
AN70 F52
20 6010 30 50
RECOMMENDED REGION FOR MIN ON-TIME AND MAX EFFICIENCY
IMAX = 0.1RSENSE
Figure 51. LTC1435/LTC1435A Maximum VIN ComparisonFigure 52. LTC1435A Minimum On-Timevs Inductor Ripple Current
Application Note 84
AN84-35
Enter the LTC1435A
The operating envelope has been substantially expandedwith the introduction of the new LTC1435A DC/DC controller,which has all the outstanding features of the LTC1435 witha reduced minimum on-time of 300ns or less and improvednoise immunity at low output voltages. With these im-provements, high performance at output voltages down to1.3V can be achieved with operating frequencies in excessof 250kHz from input supply voltages above 22V. Figure 51shows the resulting improvement of maximum VIN vsoutput voltage as a result of the reduced minimum on-time.
The LTC1435A’s minimum on-time is dependent on thespeed of the internal current comparator, which in turn isdependent on the amplitude of the signal the comparatoris monitoring: inductor ripple current. Thus, the higher theripple current, the lower the minimum on-time. Figure 52shows how minimum on-time varies as a function of theinductor ripple amplitude. At higher amplitudes, tON(MIN)is less than 250ns; at low amplitudes it can be 350ns ormore. This means that for low duty cycle applicationswhere the on-time is approaching tON(MIN), there may bea minimum ripple current amplitude, and hence, a maxi-mum inductance necessary to prevent cycle skipping. Or,expressed differently, the lower the inductance, the higherthe maximum VIN that can be achieved before the mini-mum on-time is violated and cycle skipping occurs. Formost applications, 40% ripple not only reduces the mini-mum on-time but also optimizes efficiency.
22V to 1.6V Converter at 250kHz
Figure 53 shows the LTC1435A configured in an allN-channel synchronous buck topology as a 22V to 1.6V/3A converter running at 250kHz. The 43pF COSC capacitorsets the internal oscillator frequency at 250kHz and the33mΩ sense resistor sets the maximum load current at3A. For a 22V to 1.6V converter, the on-time required is:
tON = 1.6/(22 × 250kHz) = 291ns
Can the LTC1435A do this? At maximum VIN the inductorripple is
∆IL =VOUT • (1 – VOUT/VIN)
F • L
1.6 • (1 – 1.6/22)250kHz • 4.7µH
= = 1.3A
which is 43% of the 3A maximum load. From Figure 52,43% ripple gives a minimum on-time of 235ns, which iswell below the 291ns required by this application, so nocycle skipping will occur. If a 10µH inductor is used, theripple amplitude drops to 0.6A or 20% and the minimumon-time increases to 280ns. This does not provide muchmargin below the 291ns on-time required, and thus the4.7µH inductor is a better choice.
+
+
+
M1Si44412DY
M2 Si44412DY
CIN10µF, 30V×2
L1 4.7µHDB*
RSENSE0.033Ω 35.7k
1%
102k1%
COUT100µF, 6.3V×2
D1MBRS
-140T3
4.7µF
1000pF
0.1µF
COSC 43pF
CSS0.1µF
CC330pF
CC251pF
100pF
VIN
TG
SW
INTVCC
BOOST
BG
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
13
16
14
12
15
11
10
1
2
3
5
6
VOUT1.60V/3A
VIN4.5V TO 22V
SENSE– SENSE+
7 8
LTC1435A
*DB = CMDSH-3CENTRAL(516) 435-1110
RC10k
Figure 53. LTC1435A 22V to 1.6V/3A Converter (f = 250kHz)
Application Note 84
AN84-36
Intel Mobile Processor VID Power Converter
Figure 54 shows the LTC1435A used with an LTC1706-19to implement an Intel Mobile Pentium® II Processor VIDpower converter. This DC/DC converter provides digitallyselectable output voltages over the range of 1.3V to 2.0V
+
+ +
M1Si4410DY
M2 Si4410DY
CIN10µF, 30V×2
L1 3.3µHDB*
RSENSE0.015Ω
COUT820µF4V×2
D1MBRS
-140T3
4.7µF
1000pF
0.22µF
COSC 43pF
CSS0.1µF
CC1000pF
CC2220pF
51pF
VIN
TG
SW
INTVCC
BOOST
BG
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
13
16
14
12
15
11
10
1
2
3
5
6
VOUT1.30V TO2.00V/7A
VIN4.5V TO 22V
SENSE– SENSE+
7 8
LTC1435A
*DB = CMDSH-3CENTRAL(516) 435-1110
RC10k
RF4.7Ω
CF0.1µF
VCC
FB
VID0
VID1
VID2
VID3 GND
SENSE
LTC1706-19
FROM µP
3
5
7 8 1 2 4
6
in 50mV increments at 250kHz and a 7A load current. Theselectable output voltage is implemented by replacing theconventional feedback resistor network with theLTC1706-19, which provides the appropriate feedbackresistor ratios internally. The proper ratio is selected withthe 4-bit digital input pins.
FIgure 54. Intel Mobil Pentium II VID Power Converter
LTC1627 MONOLITHIC SYNCHRONOUSSTEP-DOWN REGULATOR MAXIMIZES SINGLEOR DUAL LI-ION BATTERY LIFEby Jaime Tseng
Introduction
The LTC1627 is a new addition to a growing family ofpower management products optimized for Li-Ion batter-ies. Li-Ion batteries, with their high energy density, arebecoming the chemistry of choice for many handheldproducts. As the demand for longer battery operating timecontinues to increase and the operating voltages of submi-cron DSPs and microcontrollers decreases, more demandsare placed on DC/DC conversion. The LTC1627 mono-lithic, current mode synchronous buck regulator wasspecifically designed to meet these demands.
The LTC1627, with its operating supply range of 2.65V to8.5V, can operate from one or two Li-Ion batteries as wellas 3- to 6-cell NiCd and NiMH battery packs.
The LTC1627 incorporates power saving Burst Modeoperation and 100% duty cycle for low dropout to maxi-mize the battery operating time. In Burst Mode operation,both power MOSFETs are turned off for increasing inter-vals as the load current drops. Along with the gate-chargesavings, unused circuitry is shut down between burstintervals, reducing the quiescent current to 200µA. Thisextends operating efficiencies exceeding 90% to over twodecades of output load range.
Typical Applications
The LTC1627, with its synchronous switching and atten-dant circuitry, provides the means of easily constructing asecondary flyback regulator, as shown in Figure 55. Thisflyback regulator is regulated by the secondary feedbackresistive divider tied to the SYNC/FCB pin. This pin forcescontinuous operation whenever it drops below its ground-referenced threshold of 0.8V. Power can then be drawnfrom the secondary flyback regulator whether the mainoutput is loaded or not.
Application Note 84
AN84-37
1 or 2 Li-Ion Step-Down Converter
Figure 56 is a schematic diagram showing the LTC1627being powered by one or two Li-Ion batteries. All thecomponents shown in this schematic are surface mountand have been selected to minimize the board space andheight. The output voltage is set at 3.3V, but is easilyprogrammed to other voltages.
Single Li-Ion Step-Down Converter
The circuit in Figure 57 is intended for input voltages below4.5V, making it ideal for single Li-Ion battery applications.Diodes D1 and D2 and capacitors C1 and C2 comprise thebootstrapped charge pump to realize a negative supply atthe VDR pin, the return pin for the top P-channel MOSFETdriver. This allows Figure 57’s circuit to maintain lowswitch RDS(ON) all the way down to the UVLO trip voltage.
22µF***6.3V
25µH†
1:1
D1MBR0520LT1
VSEC†††
3.3V/100mA
ITH
RUN/SS
VFB
GND
SYNC/FCB
VDR
VIN
SW
LTC1627
CITH47pF
CSS0.1µF
COUT**100µF6.3V
R1100k1%
R280.6k
1%
1
2
3
4 5
6
7
8
VIN ≤ 8.5V
CIN*22µF16V
VOUT1.8V/0.3A
R3249k1%R480.6k1%
++
+
AVX TPSC226M016R0375AVX TPSC107M006R0150AVX TAJA226M006R(207) 282-5111
***
***
D1††
1.8V
COILTRONICS CTX25-1(561) 241-7876MMSZ4678T110mA MIN LOAD CURRENTRECOMMENDED
†
†††††
ITH
RUN/SS
VFB
GND
SYNC/FCB
VDR
VIN
SW
LTC1627
CITH47pF
CSS0.1µF
COUT†100µF6.3V
25µH*
R1249k
1%
R280.6k
1%
1
2
3
4 5
6
7
8
VIN ≤ 8.4V
CIN††22µF16V
VOUT3.3V/0.5A+
+SUMIDA CD54-250(847) 956-0666AVX TPSC107M006R0150AVX TPSC226M016R0375(207) 282-5111
*
†††
ITH
RUN/SS
VFB
GND
SYNC/FCB
VDR
VIN
SW
LTC1627
CITH47pF
CSS0.1µF
COUT100µF6.3V
D1
C10.1µF
15µH*
R1169k
1%
R280.6k
1%
1
2
3
4 5
6
7
8
VIN2.8V–4.5V
CIN††22µF16V
C20.1µF
VOUT2.5V/0.5A
D2
BAT54S**
++
SUMIDA CD54-150(847) 956-0666ZETEX BAT54S(516) 543-7100AVX TPSC107M006R0150AVX TPSC226M016R0375(207) 282-5111
*
**
†††
Figure 55. Dual-Output 1.8V/0.3A and 3.3V/100mA Application
Figure 56. Lithium-Ion to 3.3V/0.5A regulator
Figure 57. Single Lithium-Ion to 2.5V/0.5A Regulator
Application Note 84
AN84-38
THE LTC1625 CURRENT MODEDC/DC CONTROLLER ELIMINATESTHE SENSE RESISTOR
by Christopher B. Umminger
Introduction
Power supply designers have a new tool in their quest forever higher efficiencies. In the past, when designing astep-down DC/DC converter, one had to choose betweenthe high efficiency of voltage mode control and the manybenefits of current mode control. Although voltage modecontrol offers high efficiency and a simple topology, it isdifficult to compensate, has poor rejection of input-voltage transients and does not inherently limit outputcurrent under fault conditions, such as an output shortcircuit. Current mode control overcomes these problemsby adding a control loop to regulate the inductor currentin addition to the output voltage. Unfortunately, a senseresistor is required to measure this current, which addscost and complexity while reducing converter efficiency.However, with the new LTC1625 No RSENSE™ controller,one can enjoy all of the benefits of current mode controlwithout the penalties of using a sense resistor.
The LTC1625 is a step-down DC/DC switching regulatorcontroller that is capable of a wide range of operation withinputs from 3.7V to 36V. Fixed output voltages of 5V and3.3V can be selected or an external resistive divider can beused to obtain output voltages from 1.19V up to nearly the
full input voltage. The controller provides synchronousdrive for N-channel power MOSFETs and retains theadvantage of low dropout operation typically associatedwith P-channel MOSFETs. Burst Mode™ operation main-tains efficiency at low load currents, but can be overriddento assist secondary-winding regulation by forcing con-tinuous operation. In addition to eliminating the senseresistor, the LTC1625 further reduces the external partscount by incorporating the oscillator timing capacitor. Theoscillator frequency can be set to 150kHz, 225kHz, or canbe injection locked to any frequency between these points.
Design Examples
Figure 58 shows the LTC1625 in an application supplyinga 2.5V output using an external feedback divider. Si4410DYMOSFETs from Siliconix allow this converter to deliver up
+
+
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
EXTVCC
SYNC
RUN/SS
FCB
ITH
SGND
VOSENSE
VPROG
VIN
TK
SW
TG
BOOST
INTVCC
BG
PGND
LTC1625+
5V
CSS 0.1µF
RC1
10k
CC1 820pF
CC2 220pF
RF 1Ω
CF0.1µF
CIN10µF30V× 2
M1Si4410DY
M2Si4410DY
L1 7µH
R211k1%
R110k1%
COUT100µF10V0.065Ω ESR× 3
CBO.22µF
CVCC4.7µF
VOUT2.5V/5A
VIN5V TO 28V
D1MBRS140T3
*DB
DB = CMDSH-3*
1
EFFI
CIEN
CY (%
)
100
95
90
85
80
DI_1068_02a. EPS
3LOAD CURRENT (A)
2 4 50
VIN = 20VVOUT = 2.5V
LTC1625
LTC1435
Figure 58. 2.5V/5A Adjustable-Output Supply
Figure 59. Efficiency vs Load Current
Application Note 84
AN84-39
to 5A of load current. Ripple current is 1.8A (36% of fullload) and current limit occurs around 6A. Note also thatthe EXTVCC pin is connected to an external 5V supply. Thisincreases efficiency by drawing the roughly 7mA gatecharge current from a supply lower than VIN.
An efficiency plot of this circuit is shown in Figure 59. AnLTC1435 with identical components in the power path isalso plotted for comparison. At lower output voltages suchas this, the sense resistor is responsible for an increasingshare of the total power loss. By eliminating this source ofloss, the LTC1625 is easily able to deliver an efficiency
+
+
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
EXTVCC
SYNC
RUN/SS
FCB
ITH
SGND
VOSENSE
VPROG
VIN
TK
SW
TG
BOOST
INTVCC
BG
PGND
LTC1625+
CSS 0.1µF
RC1
10k
CC11nF
CC2 220pF
RF 4.7Ω
CF0.1µF
CIN22µF35V× 2
M1Si4412DY
M2Si4412DY
L1* 39µH
R235.7k1%
R13.92k1%
COUT100µF16V0.030Ω ESR
CBO.1µF
CVCC4.7µF
VOUT12V/2.2A
VIN12V TO 28V
L1 = SUMIDA CDRH127-390MCDB = CMDSH-3
INTVCC
D1MBRS140T3
**DB
***
greater than 90% at high load current. The benefit ofreduced I2R loss is readily apparent at the highest loads.The controller makes a transition to Burst Mode operationbelow around 1.1 A which keeps the efficiency high atmoderate loads.
A circuit demonstrating the wide output range of theLTC1625 is shown in Figure 60. This application usesSi4412DY MOSFETs to deliver a 12V output at up to 2.2A.Note that the SYNC pin is tied high for 225kHz operationin order to reduce the inductor size and ripple current.
PolyPhase SWITCHING REGULATORS OFFERHIGH EFFICIENCY IN LOW VOLTAGE,HIGH CURRENT APPLICATIONSby Craig Varga
Introduction
In recent years, there has been a tendency in the digitalworld toward smaller device geometries and higher gatecounts. This has led to requirements for lower voltagesand higher currents for logic supplies. As this trendcontinues, to levels under 2V and over 30A, the conven-tional buck regulator approach ceases to be viable. Switchcurrents are too high for a single device to handle, inductorenergy storage exceeds what is available in surface mounttechnology and ripple current requirements on input ca-pacitors dictate the use of many capacitors in parallel.Although all this may seem like enough of a challenge, the
transient response requirements also become much moresevere. The question that arises is: “is there a topology thatcan solve all of these problems simultaneously? ” Theanswer is “PolyPhase™.”
What is PolyPhase, Anyway?
Since it is apparent that multiple FETs need to be paralleledto handle the current requirements, the question is whetherthere is a way to drive them intelligently, rather than bybrute force. The solution is to stagger the turn-on times sothat the dead bands in the input current waveform are“filled up,” so to speak. In the simplest implementation,there are essentially two independent synchronous buckregulators operating 180° out of phase. The net effect ofthis is that the input and output ripple currents of the twochannels tend to cancel during steady-state operation.This results in significant reductions in both input and
Figure 60. 12V/2.2A Adjustable-Output Supply
Application Note 84
AN84-40
5 4 6 8 12
AST
AST
–T +T RET
RCC
3 1
Q
SYNC
1
SYNC
2
I SEN
SE2
I SEN
SE1
I SEN
SE1
I SEN
SE2
SYNC
1
SYNC
2
Q
OSC
10 11 13
5V
5V
12V
5V
C23
0.47
µF
CHAR
GE P
UMP
OPTI
ONAL
U4
CD40
47
(POW
ER F
ROM
5V)
2RX RS
T
CX
9
C6, 1
00pF
,NP
O, 5
%
R6 3.09
k1%
R3 1k
C22
1µF
10V
C322
µF 25V
C13
180p
F
C21
47µF
10V
C1 1µF
16VR1 51
Ω
R4 1k
C28
0.1µ
F
+C7 47
0µF
6.3V
+
R10
10Ω
R24
39k
+
C35
1µF
10V
C34
1µF
10V
C36
1µF
10V
+VIN
R13
0.00
2ΩTR
ACE
Q1M
MBT
3906
LT1
D1BA
W56
LT1
R12
10k
D4BA
T54
D2 BAT5
4
D3 BAT5
4
C18
1000
pF+
R59.
76k
1%+V
OUT
2.5V
/30A
OUTP
UTRT
N
+VIN
INPU
TRT
N
7 8 5 6
2 1 4 3
U2 LTC1
430A
CS8
C14
1500
pF
C11
470µ
F6.
3V
+
R14
10k
R28
1Ω
C25
1µF
16V
C19
6800
pF
C5 1µF
10V
PVCC
2G2 SH
DNCO
MP
PVCC
1G1 FB
GND
R22
1Ω R23
1Ω
5V12
V
CHAR
GE P
UMP
OPTI
ONAL
R20
1Ω
Q2Si
4410
DYQ3
Si44
10DY
L20.
8µH
ETQP
1F0R
8LB
Q5Si
4410
DY
Q4Si
4410
DY
R21
1Ω
C10
470µ
F6.
3V
C32
470µ
F6.
3V+
7 8 5 6
2 1 4 3
U3 LTC1
430A
CS8
C15
1500
pF
C8 470µ
F6.
3V
C17
1000
pF
+
R15
10k
C2 1µF
16V
R29
1Ω
R2 9.76
k1%
C16
180p
F
C24
1µF
16V
C20
6800
pF
C4 1µF
10V
PVCC
2G2 SH
DNCO
MP
PVCC
1G1 FB
GND
C12
470µ
F6.
3V
R26
1Ω R27
1ΩR19
1Ω
C27,
0.4
7µF
Q6Si
4410
DYQ7
Si44
10DY
L10.
8µH
ETQP
1F0R
8LB
Q9Si
4410
DY
Q8Si
4410
DY
R25
1Ω
R16
10Ω
C947
0µF
6.3V
C33
470µ
F6.
3V+
+
R17
10k,
1%
R11
0.00
2ΩTR
ACE
R18
10k
1%
R7 51k
(4)
C30
0.02
2µF
12V
R32
10Ω
R8 4.3k
C26
22µF 25V
+C2
91µ
F16
V C31
0.02
2µF
C38
3300
pF
C37
3300
pF
27
6
××
×
48
13
+–
U1 LT10
06SH
ARE
AMPL
IFIE
R
NOTE
S:1.
ALL
RES
ISTO
RS =
±5%
UNL
ESS
NOTE
D OT
HERW
ISE.
2. IN
PUT/
OUTP
UT C
APAC
ITOR
S =
KEM
ET T
510
SERI
ES(4
08) 9
86-0
424
3. T
RACE
RES
ISTO
RS R
11, R
13 =
0.1
" WID
E x
0.67
5" L
ONG
R9 4.3k
R31
1kR30
1k
+
CLO
CK
Figure 61. 2-Phase Synchronous Buck Regulator
Application Note 84
AN84-41
output capacitor requirements. There is also a fourfoldreduction in the total inductor energy storage require-ment, which means much smaller inductors and vastlyimproved transient dynamics. During a large load step, thetwo channels operate at maximum duty factor in anattempt to maintain the desired output voltage. Bothinductor currents slew rapidly and are now additive, sincethey are going in the same direction. Hence, the slew rateis double what a single channel could do for equal inductorvalues. However, due to the ripple current cancellationduring steady-state conditions, the two inductors can bereduced to approximately one-half the value that a singlechannel design would require for equal ripple currents.Since during slew they appear to be operating in parallel,the actual slew rate is four times that of a single channeldesign with equal steady-state output ripple current. Bothinput and output ripple frequencies are double those of asingle-channel design, further simplifying filteringrequirements.
CURRENT (A)0
EFFI
CIEN
CY (%
)100
95
90
85
80
75
705 10 15 20
DC201 F01b
25 30
VIN = 5V
VOUT = 2V
VOUT = 3.3V
VOUT = 2.5V
Figure 62. Efficiency of Figure 61’s Circuit, VIN = 5V
Why Stop at Two?
If two channels are good, aren’t more channels better? Ina word, yes. In principle, there is no limit to the number ofparallel channels that can be added. As the number ofchannels, n, increases, the ripple frequency increases to ntimes the single-channel frequency. Input and outputRMS ripple currents continue to decrease. Diminishingreturns are reached as n rises above three. At three stages,the ripple reductions are very substantial and dynamicperformance is excellent. Adding more channels producesslight improvements but the dramatic gains will have beenrealized by n = 3. The only real penalty is added complexity.
The bottom line is that PolyPhase designs offer a consid-erable reduction in the cost and volume of the powerdevices at the expense of a little added complexity in thecontrol circuitry.
Figure 64. Transient Response with10A Load Step (100ns Rise Time)
10µs/DIV
100mV/DIV ∆V =160mV
Figure 65. Ripple Cancellation—Input
2µs/DIV
CHANNEL A5A/DIV
CHANNEL A + B =TOTAL INPUT
RIPPLE CURRENT,UNFILTERED
CHANNEL B5A/DIV
IO = 15Af = 306kHz
2µs/DIV
20mV/DIV
VO = 2.5VVIN = 5V
Figure 63. Output Ripple with 30A Load
Application Note 84
AN84-42
2-Phase Design Example
The circuit shown in Figure 61 is a 2-phase, voltage mode–control, synchronous buck regulator designed for a 5Vinput and output voltages below 3.3V. It is intended topower large memory arrays, ASICs, FPGAs and the like inserver and workstation applications. The output is capableof more than 30 amps continuous at outputs of 2.5V andbelow, with peak current capability of greater than 40amps. The design is entirely surface mount and themaximum height above the board is 5.5mm. Overall boardarea is only 4.24 in2. Efficiency is excellent, as can be seenin the curve in Figure 62. Output ripple voltage is shown inFigure 63. The circuit’s dynamic response to a 10 amp loadstep is shown in Figure 64. The response is dominated bythe output capacitor’s ESR and shows the output voltagerecovered to the original level in under 10µs. Figures 65and 66 show how the input and output ripple currentscancel.
Circuit Operation
The basic design consists of two LTC1430CS8-basedsynchronous buck regulators connected in parallel andoperated 180° out of phase. U4, the CD4047 oscillator, isused to generate the required clock signals and synchro-
Figure 66. Ripple Cancellation—Output
nize the two LTC1430s. Unfortunately, simply connectingtwo regulators in parallel is a recipe for instant disaster.The output voltages of the two regulators will be slightlydifferent due to normal component tolerances. Therefore,the higher output voltage channel will attempt to supplythe full load current, while the lower voltage output willsink current from the output in a desperate attempt toreduce the output voltage to where it thinks it should be.The result is like a dog chasing its tail, with large currentsrunning around in a circle and going nowhere.
Op amp U1 solves this problem. Because the two channelsare identical, if the output currents are the same, the inputcurrents will be also. Low value sense resistors are in-cluded in the input power path to allow the circuit tomeasure input current. U1 then forces the input current ofchannel two to match the input current of channel one bymaking small adjustments in channel two’s output volt-age. It does this by adding or subtracting a small amountof current from channel two’s feedback divider. The twosense resistors are short lengths of PCB trace and onlyneed to be ratiometrically accurate. The absolute value ofthese resistors is not important (see Linear TechnologyApplication Note 69, Appendix A, for a discussion on howto design trace resistors).
The only remaining trick in the circuit is the role of Q1 andits associated circuitry. At start-up, the LTC1430’s clockfrequency is slowed down to approximately 10kHz untilthe output voltage rises to approximately 50% of thedesired level. If, during this start-up phase, an attempt ismade to synchronize the controller to a very high fre-quency, the oscillator ramp amplitude never rises to a levelsufficiently high to trip the PWM comparator and enablethe FET drivers. Therefore, the output gets stuck onground. Q1 fixes this by forcing the sync signals highduring the turn-on transient. Once the output voltagenears its final level, the clock signals are allowed tosynchronize the two PWM controllers.
2µs/DIV
A + B
CHANNEL ACHANNEL B
5A/DIV
IO = 25A
Application Note 84
AN84-43
LTC1622: LOW INPUT VOLTAGE,CURRENT MODE PWM BUCK CONVERTERby San-Hwa Chee
Introduction
The 8-pin LTC1622 step-down DC/DC controller is de-signed to help system designers harness all of the avail-able energy from lithium-ion batteries in several ways. Itswide operating input-voltage range (2.0V to an absolutemaximum of 10V) and 100% duty cycle allows lowdropout for maximum energy extraction from the battery.The part’s low quiescent current, 400µA, with a shutdowncurrent of 15µA, extends battery life. Its user-selectableBurst Mode operation enhances efficiency at low loadcurrent.
For portable applications where board space is a pre-mium, the LTC1622 operates at a constant frequency of550kHz and can be synchronized to frequencies of up to750kHz. High frequency operation allows the use of smallinductors, making this part ideal for communicationsproducts. The LTC1622 comes in a tiny 8-lead MSOPpackage, providing a complete power solution whileoccupying only a small area.
2.5V/1.5A Step-Down Regulator
A typical application circuit using the LTC1622 is shownin Figure 67. This circuit supplies a 1.5A load at 2.5V withan input supply between 2.7V and 8.5V. The 0.03Ω senseresistor is selected to ensure that the circuit is capable of
supplying 1.5A at a low input voltage. In addition, asublogic threshold MOSFET is used, since the circuitoperates at input voltages as low as 2.7V. The circuitoperates at the internally set frequency of 550kHz. A 4.7µHinductor is chosen so that the inductor’s current remainscontinuous during burst periods at low load current. Forlow output voltage ripple, a low ESR capacitor (100mΩ) isused.
Efficiency Considerations
The efficiency curves for Figure 67’s circuit are shown inFigures 68 and 69. Figure 68 shows the efficiency withBurst Mode enabled, whereas Figure 69 has Burst Modedefeated. (Burst Mode is defeated by connecting theSYNC/Mode pin to ground.) Note that, at low load cur-rents, the efficiency is higher with Burst Mode operation.However, constant frequency operation is still achievable
SENSE–
PDRV
GND
VFB
VIN
ITH
SYNC/MODE
RUN/SS
1
7
6
3
8
2
5
4
C110µF16V R1
10kC3
220pF
470pF
R2 0.03Ω
Si3443DV
D1
L1 4.7µH
R3159k
R475k
C247µF6V
VOUT2.5V/1.5A
LTC1622VIN2.5V–8.5V
C1: MURATA CERAMIC GRM235Y5V106Z(814) 236-1431
C2: SANYO POSCAP 6TPA47M(619) 661-6835
L1: MURATA LQN6C-4R7M04(814) 237-1431
D1: IR10BQ015(310) 322-3331
R2: DALE, 0.25W(605) 665-9301
+
+
Figure 67. LTC1622 Typical Application: 2.5V/1.5A Converter
LOAD CURRENT (A)
60EFFI
CIEN
CY (%
)
70
80
90
100
0.001 1.000
50
0.01040
0.100
VIN = 3.3V
VIN = 6V
VIN = 4.2V
VIN = 8.4V
VOUT = 2.5VRSENSE = 0.03Ω
Figure 68. Efficiency vs Load Current for Figure 67’sCircuit (Burst Mode Operation Enabled)
Application Note 84
AN84-44
SENSE–
PDRV
GND
VFB
VIN
ITH
SYNC/MODE
RUN/SS
1
7
6
3
8
2
5
4
C147µF16V R1
22kC3
100pF
470pF
R2 0.03Ω
Si3443DV
D1
L1 1.3µH
R3159k
R475k
C2100µF6V
VOUT2.5V/1.5A
LTC1622VIN2.5V TO
8.5V
C1: AVX TPSD476M016R0150(803) 946-0362
C2: AVX TPSD476M016R0065L1: MURATA LQN6C-1R5M04
(814) 237-1431
D1: IR10BQ015(310) 322-3331
R2: DALE, 0.25W(605) 665-9301
+
+
Figure 72. 2.5V/1.5A Converter with Improved Transient Response
at a lower load currents with Burst Mode operation de-feated. The kinks in the efficiency curves indicate thetransition out of Burst Mode operation.
The components of Figure 67 have been carefully chosento provide the amount of output power using a minimumof board space. Efficiency is also a prime consideration inselecting the components, as illustrated in Figures 68 and69. Figures 70 and 71 show the transient response of VOUTwith a load step from 50mA to 1.2A. Figure 70 has BurstMode enabled, while Figure 71 has it defeated. Note thatthe output voltage ripple (in the middle portion of thephotographs) is higher for Burst Mode operation than withBurst Mode disabled at 50mA load current.
Applications that require better transient response can usethe circuit in Figure 72, whose components are selectedspecifically for this requirement. Figures 73 and 74 showthe response with and without Burst Mode operation,respectively. Note that the transient response has been
Figure 70. Transient Response with Burst ModeOperation Enabled; Load Step = 50mA to 1.2A
Figure 71. Transient Response with Burst ModeOperation Inhibited; Load Step = 50mA to 1.2A
OUTPUT VOLTAGE(AC COUPLED)
0.1V/DIV
0.1ms/DIV
OUTPUTVOLTAGE
(AC COUPLED)0.1V/DIV
0.1ms/DIV
LOAD CURRENT (A)
EFFI
CIEN
CY (%
)
0.001 1.0000.010 0.100
VIN = 3.3V
VIN = 6V
VIN = 4.2V
VIN = 8.4V
VOUT = 2.5VRSENSE = 0.03Ω
100
90
80
70
60
50
40
Figure 69. Efficiency vs Load Current for Figure 66’s Circuit(Burst Mode Operation Disabled)
enhanced significantly. However, this comes at the expenseof slightly reduced efficiency at low load currents, asindicated by the efficiency curves of Figures 75 and 76.
Application Note 84
AN84-45
Figure 73. Transient Response with Burst ModeOperation Enabled; Load Step = 50mA to 1.2A
Figure 74. Transient Response with Burst ModeOperation Inhibited; Load Step = 50mA to 1.2A
0.1ms/DIV
OUTPUTVOLTAGE
(AC COUPLED)0.1V/DIV
OUTPUTVOLTAGE
(AC COUPLED)0.1V/DIV
0.1ms/DIV
LOAD CURRENT (A)
EFFI
CIEN
CY (%
)
100
90
80
70
60
50
400.001 1.0000.010 0.100
VIN = 3.3V
VIN = 6V
VIN = 8.4V
VOUT = 2.5VRSENSE = 0.03Ω
VIN = 4.2V
LOAD CURRENT (A)
EFFI
CIEN
CY (%
)
100
90
80
70
60
50
400.001 1.0000.010 0.100
VIN = 3.3V
VIN = 6V
VIN = 8.4V
VOUT = 2.5VRSENSE = 0.03Ω
VIN = 4.2V
Figure 75. Efficiency vs Load Current for Figure 72’sCircuit (Burst Mode Operation Enabled)
Figure 76. Efficiency vs Load Current for Figure 72’sCircuit (Burst Mode Operation Disabled)
Application Note 84
AN84-46
WIDE INPUT RANGE, HIGH EFFICIENCYSTEP-DOWN SWITCHING REGULATORSby Jeff Schenkel
Introduction
The LT1676/LT1776 are current mode switching regulatorICs optimized for high efficiency operation in high inputvoltage, low output voltage buck topologies. These twoparts are pin-for-pin compatible and virtually identical inoperation, the only difference being their internal oscilla-tor frequencies—100kHz for the LT1676 vs 200kHz forthe LT1776. They operate in a fixed frequency mode (asopposed to constant off-time or on-time, for instance) andcan be externally synchronized to a higher switchingfrequency.
The internal output switch is rated at a nominal peakcurrent of 700mA, which typically accommodates DCoutput currents of up to 500mA. The input voltage rangeis 7.4V to 60V. Maintaining acceptable efficiency in theupper half of this input voltage range requires very fastoutput-switch edge rates. The LT1676/LT1776 containspecialized output circuitry to deliver this performance.Additionally, they contain circuitry to monitor output loadlevel and reduce leading-edge switch rate (turn-on) whenthe output load is light. This arrangement helps avoidpulse skipping at light load, with its consequent subhar-monic behavior.
True current mode operation is supported, with all its wellknown advantages for switching regulator operation. Theshutdown pin implements a pair of functions. Pulling itdown to near ground turns off the part almost completelyand reduces the quiescent current to a few tens of micro-amperes. The second shutdown pin function acts at athreshold of roughly 1.25V. Below this level, the partoperates normally, except that output switching action isinhibited. This allows the implementation of an undervolt-age lockout function set by, for instance, an externalresistor divider. The LT1676/LT1776 are available in both8-pin SO and PDIP packages.
Applications
Minimum Component-Count Application
Figure 77 shows a basic “minimum component count”application using the LT1676. The circuit produces 5.0V atup to 500mA IOUT with input voltages in the range of 12Vto 48V. The typical POUT/PIN efficiency is shown in Figure78. No pulse skipping is observed down to zero externalload. (The several milliamperes drawn by the VCC pin actsas a sufficient preload.) As shown, the SHDN and SYNCpins are unused, however either (or both) can be option-ally driven by external signals as desired.
VINVCCVSW
LT1676FBVC
SHDN
SYNC
C1: PANASONIC HFQ(201) 348-7522
C2: AVX D CASE TPSD107M010R0080(803) 946-0362
C4, C5: X7R OR COG/NPOD1: MOTOROLA 100V, 1A, SMD SCHOTTKY
(800) 441-2447L1: COILCRAFT DO3316P-224
(847) 936-6400
2
C139µF63V 5
4
1676 F04a
3
7
8
1
VIN12V TO
48V
6
GND
+
C2100µF10V
+D1MBRS1100 R1
36.5k1%
VOUT5V0mA to 500mA
R212.1k1%
R322k5%
L1220µH
C32200pFX7R C4
100pF
C5100pF
FOR 3.3V VOUT VERSION: R1: 24.3k, R2: 14.7k L1: 150µH, DO3316P-154 IOUT: 0mA TO 500mA
LOAD CURRENT (mA)1
60
EFFI
CIEN
CY (%
)
70
80
90
10 100 1000
1676 F04b
50
40
30
20
VIN = 12V
VIN = 24V
VIN = 36VVIN = 48V
Figure 77. Minimum Component-Count Application Figure 78. Efficiency of Figure 77’s Circuit
Application Note 84
AN84-47
Minimum PC Board Area Application
The previous application example used the LT1676 todemonstrate simultaneously the maximum input voltageand output current capability. As such, the input bypasscapacitor choice was a high frequency aluminum electro-lytic type, rated to 63V. Also, the 100kHz switching rate ofthe LT1676 requires an inductor of about 220µH. TheDO3316 device size was chosen to support the outputcurrent requirements. However, both of these compo-nents are physically large.
The application example in Figure 79 shows a circuit thatis much smaller physically than the previous minimumcomponent count application. The nominal 200kHz switch-ing frequency of the LT1776 allows the use of a physicallysmaller 68µH inductor—a Coilcraft DO1608C-683. Thisinductor will support output current to 400mA at 5V.However, the part is incapable of withstanding an indefiniteshort circuit to ground. (Momentary shorts of a fewseconds or less can still be tolerated.) Additionally, thebulky aluminum electrolytic capacitor previously on VINhas been replaced by a compact 35V-rated tantalum type.The result is a postage-stamp-sized circuit with efficiencyas shown in Figure 80.
Burst Mode Application
The minimum component count application demonstratesthat power supply efficiency degrades with lower output
load current. This is not surprising, as the LT1676 itselfrepresents a fixed power overhead. A possible way toimprove light load efficiency is to use Burst Mode™operation.
Figure 81 shows the LT1676 configured for Burst Modeoperation. Output voltage regulation is now provided in a“bang-bang” digital manner, via comparator U2, anLTC1440. Resistor divider R4/R5 provides a scaled ver-sion of the output voltage, which is compared against U2’sinternal reference. Intentional hysteresis is set by the R6/R7 divider. As the output voltage falls below the regulationrange, the LT1676 is turned on. The output voltage risesand, as it climbs above the regulation range, the LT1676is turned off. Efficiency is maximized as the LT1676 is onlypowered up while it is providing heavy output current.Figure 82 shows that efficiency is typically maintained at75% or better down to a load current of 10mA. Even at aload current of 2mA, efficiency is still a respectable 65%to 75% (depending on VIN).
Resistor divider R1/R2 is still present, but does notdirectly influence output voltage. It is chosen to ensurethat the LT1676 delivers high output current throughoutthe voltage regulation range. Its presence is also requiredto maintain proper short-circuit protection. TransistorsQ1 and Q2 and resistor R7 form a high VIN, low quiescentcurrent voltage regulator to power U2.
VINVCCVSW
LT1776FBVC
SHDN
SYNC
C1: AVX D CASE 15µF 35VTPSD156M035R0300(803) 946-0362
C2: AVX D CASE 100µF 10V TPSD107M010R0080C3: 2200pF, X7RC4, C5: 100pF, X7R OR COG/NPO
2
C115µF35V
5
4
1776 F07a
3
7
8
1
VIN10V TO 30V
6
GND
+C2100µF10V
+D1MBRS-1100
R136.5k1%
VOUT5V0mA to400mA
R212.1k1%R3
22k5%
L168µH
C32200pF
C4100pF
C5100pF
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY MBRS1100
(800) 441-2447L1: COILCRAFT DO1608C-683
(847) 936-6400
FOR 3.3V VOUT VERSION: IOUT: 0mA TO 500mA L1: 47µH, DO1608C-473 R1: 24.3k, R2: 14.7k
LOAD CURRENT (mA)1
60
EFFI
CIEN
CY (%
)
70
80
90
10 100 1000
1776 F07b
50
40
30
20
VIN = 10V
VIN = 20V
VIN = 30V
Figure 79. Minimum PC Board ApplicationFigure 80. Efficiency of Figure 79’s Circuit
Application Note 84
AN84-48
Battery Charger Application
Figure 83 shows the LT1776 configured as a constant-current/constant-voltage battery charger. An LT1620 rail-to-rail current sense amplifier (U2) monitors the differen-tial voltage across current sense resistor R4. As thisequals and exceeds the voltage across resistor R5 in theR5/R6 divider, the LT1620 responds by sinking current atits IOUT pin. This is connected to the VC control node of theLT1776 and therefore acts to reduce the amount of powerdelivered to the load. The overall constant-current/con-stant-voltage behavior can be seen in Figure 84.
Target voltage and current limits are independently pro-grammable. The output voltage of 7.2V, which corre-sponds to the charging voltage of a 3-cell lead-acid bat-tery, is set by the R1/R2 divider and the internal referenceof the LT1776. Output current, presently 200mA, is set bycurrent sense resistor R4 and the R5/R6 divider. (A 16-pinversion of the LT1620 that implements end-of-cycle de-tection is also available. This is useful for implementinglead-acid battery “top-off” charger behavior or the like.See the LT1620 data sheet for further information.)
The circuit as shown accommodates an input voltagerange of 11V to 30V. The upper input voltage limit of 30Vis determined not by the LT1776, but by the LT1121-5regulator (U3). (A regulated 5V is required by the LT1620.)This regulator was chosen for its micropower behavior,which helps maintain good overall efficiency. However,the basic catalog part is only rated to 30V. Substitution ofthe industry standard LM317, for example, extends theallowable input voltage to 40V (or more with the HVversion), but its greater quiescent current drain degradesefficiency from that shown.
VIN
VIN12V TO 48V
VCCVSW
U1LT1676
FBVC
SYNC2
C139µF63V
NC
Q1PN2484
Q22N2369
5
4
1676 F06
3
7
8
6
1
GNDSHDN
V+
V–
IN–
IN+U2
LTC1440REF
HYST
OUT4
7
12
3
6
5
8
GND
+
C2100µF10V
+D1MBRS1100
R139k5%
VOUT5V
R210k5%
R622k
R72.4M
R3323k1%
R4100k1%
L1220µH
C3100pF
R710M
C1: PANASONIC HFQ(201) 348-2552
C2: AVX D CASE TPSD107M010R0080(803) 946-0362)
C4, C5: X7R OR COG/NPOD1: MOTOROLA 100V, 1A,
SMD SCHOTTKY(800) 441-2447
L1: COILCRAFT DO3316-224(847) 639-6400
LOAD CURRENT (mA)1
60
EFFI
CIEN
CY (%
)
70
80
90
10 100 1000
1676 F07b
50
40
30
20
VIN = 12V
VIN = 36VVIN = 48V
VIN = 24V
Figure 81. Burst Mode Operation Configuration
Figure 82. Efficiency of Figure 81’s Circuit
Application Note 84
AN84-49
Dual Output SEPIC Converter
All of the previous applications provide a single positiveoutput voltage. Real world situations often require dualsupply voltages. The SEPIC topology (single-ended pri-mary inductance converter) offers a cost-effective way tosimultaneously generate a negative voltage with a singlepiece of magnetics. The circuit in Figure 85 uses anLT1776 to generate both positive and negative 5V. The twoinductors shown are actually just two windings on astandard Coiltronics inductor. Capacitor C3 creates theSEPIC topology, which improves regulation and reducesripple current in L1.
For the best negative supply voltage regulation, this outputshould have a preload of at least 1% of the maximumpositive load. Total available current from both outputs islimited to 500mA. Maximum negative supply current islimited by the positive 5V load. A typical limit is one-half ofthe positive current, but a more exact calculation includesthe input voltage. For this and further details of thistopology, see Linear Technology Design Note 100.
Positive-to-Negative Converter
The previous example used a dual inductor to create a pairof output voltages, one positive and the other negative.The positive-to-negative converter topology illustrated inFigure 86 generates a single negative output voltage froma positive input voltage, using just an ordinary inductor.The topology is somewhat similar to the original step-down arrangement, but the inductor is grounded and theLT1776 ground is now referred to the negative outputvoltage. Note that the integrated circuit must now be rated
VIN
VCC
VSW
U1LT1776
U3LT1121-5
FB
VC
SHDN
SYNC
C1: PANASONIC HFQ(201) 348-7522
C2: AVX TPSD107M010R0080(803) 946-0362
L1: COILCRAFT DO3316P-104(847) 639-6400
7
C139µF63V
5
4
1776 TA02
2
3
8
1
VIN11V TO 30V(SEE TEXT)
6
GND
VCC
IN+PROG
U2LT1620
IOUT
IN–NC
AVG
SENSE
2
5
4
6
3
8
1
7
GND
+
D1MBRS1100
R157.6k1%
3-CELLLEAD-ACIDBATTERY
R212.1k1%
R322k
R53k
R612k
L1100µH
R40.5Ω
C42200pF
C3100pF
C70.1µF
C5100pF
C60.33µF
C2100µF10V
C81µF
+
+
7.2V
OUTPUT CURRENT (mA)0
OUTP
UT V
OLTA
GE (V
)
8
7
6
5
4
3
2
1
050 100 150 250200
1776 TA05
Figure 83. Wide VIN Range, High Efficiency Battery Charger
Figure 84. Battery Charger Output Voltage vsOutput Current for Figure 83’s Circuit
Application Note 84
AN84-50
+
+
+
+ C115µF35V
C7100pF
VCC
VSW
FB
VC
2
3
7
8
1
6
SHDN
SYNC
5
4
VIN
GND
C52200pFX7R
R322k5%
C6100pF
C3100µF10V
C4100µF10V
C2100µF10V
L1* 100µH
R136.5k1%
R212.1k1%
L1*100µH
D2MBR1100
D1MBR-1100
VOUT –5V†
VOUT 5V†
VIN10V TO 28V
LT1776
†TOTAL AVAILABLE CURRENT IS LIMITED TO 500mA (SEE TEXT)
AVX D CASE TPSD156M035R0300(803) 946-0362AVX D CASE TPSD107M010R0080X7R OR COG/NPOMOTOROLA MBRS1100 100V, 1A, SMD SCHOTTKY(800) 441-2447COILTRONICS CTX100-3 SINGLE CORE WITH 2 WINDINGS(561) 241-7876
C1:
C2, C3, C4:C6, C7:D1, D2:
*L1:
+
+ C115µF35V
C5100pF
VCC
VSW
FB
VC
2
3
7
8
1
6
SHDN
SYNC
5
4
VIN
GNDC32200pFX7RR3
22k5%
C4100pF
C2100µF10V
L1 100µHR136.5k1%
R212.1k1%D1
MBRS1100
VOUT–5V0mA TO 300mA
VIN10V TO 28V
AVX D CASE TPSD156M035R0300(803) 946-0362AVX D CASE TPSD107M010R0080X7R OR COG/NPOMOTOROLA MBRS1100 100V, 1A, SMD SCHOTTKY(800) 441-2447COILCRAFT D03316-104(847) 639-6400
C1:
C2:C4, C5:
D1:
L1:
LT1776
for the worst case sum of the input voltage plus theabsolute value of the output voltage. The relatively highinput voltage rating of the LT1676/LT1776 parts alongwith their good efficiency under such conditions makethem an excellent choice for implementing this topology.The circuit as shown converts an input voltage in the rangeof 10V to 28V to a –5V output. Available output current is300mA at the worst case VIN of 10V.
The user should exercise caution in modifying this circuitfor other applications. The positive-to-negative topologyis not as straightforward as the step-down topology. It is
actually more like a flyback topology, in that current isdelivered to the output in discrete pulses. The outputcapacitor must supply the entire load current for at leasta portion of the switching cycle, so output capacitor ripplecurrent rating and ESR may be an issue. Maximumavailable output current will usually be a strong functionof input voltage. Supporting low VIN-to-VOUT ratios mayrequire additional components for maintaining control-loop stability. A detailed theoretical analysis of this topol-ogy and its behavior can be found in Linear TechnologyApplication Note 44.
Figure 85. Dual-Output SEPIC Converter
Figure 86. Positive-to-Negative Converter
Application Note 84
AN84-51
Regulators—Switching (Boost)
±12 VOLT OUTPUT FROM THE LT1377by John Seago
Many applications use positive and negative voltages,with only one voltage requiring tight regulation. Often,cost and board space are more important than regulationof the second output. An equal output of opposite polaritycan be added to a boost configuration by means of anegative charge pump. This two-output configuration isshown in Figure 87. The 1MHz switching frequency of theLT1377 decreases required board space, and the availabil-ity of both positive and negative feedback amplifiersallows regulation of either positive or negative output.
In the circuit of Figure 87, the LT1377 with L1, D1, D2 andC6 make up a positive boost circuit. As the internal powerswitch in the IC turns on, the voltage at pin 8 goes low andenergy is stored in inductor L1. When the power switchturns off, L1 transfers energy through diodes D1 and D2to capacitor C6 and the positive output load. C6 suppliesload current when the power switch is on. Resistors R2and R3 provide feedback from the positive output. R1, C3and C4 provide loop compensation. C1 is the input capaci-tor and C2 provides local decoupling for the IC.
The charge pump consists of two capacitors, two diodesand a small inductor. When the power switch turns off, L1also replenishes the charge on C5, forward biasing D3.When the power switch turns on, the charge on C5 reversebiases D3, forward biases D4 and supplies energy to C7and the negative output load. L2 attenuates capacitivecurrent spikes. D2 was added so that the voltage dropacross both D1 and D2 would be approximately equal tothe sum of the voltage drops of D3, D4 and the saturationvoltage of the power switch in the LT1377. This makesboth output voltages approximately equal but opposite inpolarity. D1 and D2 can be replaced with a single Schottkydiode if equal outputs are not required.
Voltage and current waveforms of the internal powerswitch are shown in Figure 88. These measurements weretaken at pin 8 of the LT1377 with the circuit powered froma 5V supply. Figure 89 shows the ripple voltage from eachoutput. The high frequency spikes can be attenuated witha small LC filter if necessary.
The circuit of Figure 87 was intended to operate from a 5Vsupply and provide ±12V outputs at 100mA each. Itoperates over an input range of 4V to 10V and load currentvariations from 15mA to 100mA. The regulated positiveoutput voltage remains constant for changes in the input
VIN
L110µH
L20.047µH
D1MURS110
R286.6k
R310k
R12k
D2MURS110
SGND PGND
U1LT1377
S/S VSW8
5
4
1
2
3
6 7
FB
4V TO 10VINPUT
–12VOUTPUT
12VOUTPUT
C110µF25VY5U
C510µF25VY5U
C62.2µF25VY5U
C72.2µF25VY5U
D3MBRS130L
D4MBRS130L
C20.47µF
C30.0047µF
C1 = C5 = 1E106ZY5U-C205-M, TOKIN (408) 432-8020C6 = C7 = 1E225ZY5U-C203. TOKIN (408) 432-8020L1 = CTX10-1P, COILTRONICS (407) 241-7876L2 = PM20-R047M, GARRETT (805) 922-0594
C40.047µF
NFBVC
+
+
+
+
Figure 87. Positive Output Regulated Supply
Application Note 84
AN84-52
INPUT VOLTAGE
–10.75
–11.25
–11.00
–11.50
–11.75
–12.25
–12.00
–12.50
–12.75
–13.00
OUTP
UT V
OLTA
GE
100 2 31 4 5 8 96 7
±15mA LOAD
±50mA LOAD
±100mA LOAD
VIN
L110µH
L20.047µH
D1MURS110
R32.21k
R28.25k
R12k
D2MURS110
SGND PGND
U1LT1377
S/S VSW8
5
4
1
2
3
6 7
FB
4V TO 10VINPUT
–12VOUTPUT
12VOUTPUT
C110µF25VY5U
C510µF25VY5U
C62.2µF25VY5U
C72.2µF25VY5U
D3MBRS130L
D4MBRS130L
C20.47µF
C30.0047µF
C40.047µF
NFBVC
C1 = C5 = 1E106ZY5U-C205-M, TOKIN (408) 432-8020C6 = C7 = 1E225ZY5U-C203, TOKIN (408) 432-8020L1 = CTX10-1P, COILTRONICS (407) 241-7876L2 = PM20-R047M, GARRETT (805) 922-0594
+
+
+
+
INPUT VOLTAGE
10.75
11.25
11.00
11.50
11.75
12.25
12.00
12.50
12.75
13.00
OUTP
UT V
OLTA
GE
100 2 31 4 5 8 96 7
±100mA LOAD
±50mA LOAD±15mA LOAD
0.5µs/DIV
12V OUTPUTRIPPLE
0.1V/DIVAC COUPLED
–12V OUTPUTRIPPLE
0.1V/DIVAC COUPLED
0.5µs/DIV
0
0
SWITCH VOLTAGE5V/DIV
PIN 8
SWITCH CURRENT0.5A/DIV
PIN 8
Figure 88. Switch Voltage and Current Waveforms
Figure 89. Output Ripple Voltage
Figure 90. Unregulated Negative OutputVoltage with Positive Output Voltage Regulated
Figure 91. Unregulated Positive Output Voltagewith Negative Output Voltage Regulated
Figure 92. Negative Output Regulated Dual Supply
Application Note 84
AN84-53
THE LT1370: NEW 500kHz, 6AMONOLITHIC BOOST CONVERTERby Karl Edwards
Introduction
The LT1370 is a 500kHz, 6A boost converter. At 65mΩon-resistance, 42V maximum switch voltage and 500kHzswitching frequency, the LT1370 can be used in a widerange of output voltage and current applications.
The high efficiency switch is included on the die, alongwith the oscillator, control and protection circuitry neces-sary for a complete switching regulator. This part com-bines the convenience and low parts count of a monolithicsolution with the switching capabilities of a discretepower device and controller. The LT1370, features cur-
rent mode operation, external synchronization and lowcurrent shutdown mode (12µA typical). Only a few surfacemount components are needed to complete a small, highefficiency DC/DC converter. The LT1370 will operate in allthe standard switching configurations, including boost,buck, flyback, forward, inverting and SEPIC.
5V to 12V Boost Converter
Figure 93 shows a typical 5V to 12V boost application. Thefeedback divider network has been selected to give thedesired output voltage. As long as R2 is less than 7k, FBinput bias current can be ignored. The inductor needs to bechosen carefully to meet both peak and average currentvalues. The output capacitor can see high ripple cur-rents—often, as in this application, higher than the ripplerating of a single capacitor. This requires the use of twosurface mount tantalums in parallel; both capacitors shouldbe of the same value and manufacturer. The input capaci-tor does not have to endure such high ripple currents and
voltage and load current, while the voltage of the unregu-lated negative output changes as shown in Figure 90. Lineand load regulation of the unregulated output will improvewith smaller changes of input voltage or load current.
A common requirement is for the positive output toregulate the majority of power while the negative outputsupplies a much smaller, unregulated bias current. Mea-surements taken on the test circuit of Figure 87 showedthe unregulated −12V output had less than ±1% variationfor a fixed 15mA load while the input voltage changed from
4V to 10V with a load current change of 15mA to 200mAon the regulated positive output.
Occasionally, it is more important to regulate the negativeoutput than the positive output. The circuit in Figure 92 isthe same as that shown in Figure 87, except feedbackresistors R2 and R3 have different values and providefeedback from the negative output to the negative feed-back amplifier of the LT1377. Figure 91 shows the varia-tion in unregulated positive output for input voltage andload current variations.
LT1370
VIN
VC
5V
GND
FB
1370_02.EPS
VSWS/S
L1*
C1**22µF25V
C4**22µF25V× 2
C20.047µF
C30.0047µF
R32k
R26.19k1%
R153.6k1%
VOUT†
12V
D1MBRD835L
ONOFF
*
**
COILTRONICS(561) 241-7876UP2-4R7 (4.7µH)UP4-100 (10µH)AVX TPSD226M025R0200
+ +
L14.7µH10µH
IOUT1.8A2.0A
†MAX IOUT
OUTPUT CURRENT (A)0
70
EFFI
CIEN
CY (%
)
75
80
85
90
95
100
0.5 1.0 1.5 2.0
1370_03
Figure 93. 5V to 12V Boost Converter Figure 94. 12V Output Efficiency
Application Note 84
AN84-54
a single capacitor will normally suffice. The catch diode,D1, must be rated for the output voltage and averageoutput current. The compensation capacitor, C2, normallyforms a pole with the internal gm of the part in the 2Hz to20Hz range. It also creates a zero in conjunction withseries resistor R3, at 1kHz to 5kHz.
A second capacitor, C3, is sometimes required to preventerratic switching. Ripple current in the output capacitor’sESR causes voltage ripple. This feeds back through theerror amp to the VC pin, changing the current-trip thresh-old cycle-to-cycle. The problem appears as subharmonicoscillation. Adding C3, typically one-tenth the value ofthe main compensation capacitor, reduces the loop gainat the switching frequency, preventing the oscillation.
The ground return from the compensation network mustbe separate from the high current switch ground. If dropsin the ground trace due to switch current cause the VC pinto dip, premature switch-off will occur. This effect appearsas poor load regulation. A solution to this is to return thecompensation network to the FB pin. The S/S pin in thisexample is driven by a logical on/off signal, a low inputforcing the LT1370 into its 12µA shutdown mode. Figure94 shows the overall converter efficiency. Note that peakefficiency is over 90%; efficiency stays above 86% at thedevice’s maximum operating current.
Positive-to-Negative Converter
The NFB (negative feedback) pin allows negative outputregulators to be designed with direct feedback. In thecircuit shown in Figure 95, a 2.7V to 13V input, –5V outputconverter, the output is monitored by the NFB pin and asimple divider network. No complex level shifting orunusual grounding techniques are required. The regularFB pin is left open circuit and the divider network, R2, R3,is calculated based on the –2.49V NFB reference voltageand 30µA of input current. The switch-clamp diodes, D2and D3, prevent the leakage spike from the transformer,T1, from exceeding the switch’s absolute maximum volt-age rating. The Zener voltage of D2 must be higher than theoutput voltage, but low enough that the sum of inputvoltage and clamp voltage does not exceed the switchvoltage rating.
5V SEPIC Converter
Figure 96 shows a SEPIC converter. One of the advantagesof the SEPIC topology is that the input voltage can rangefrom below to above the output voltage. In Figure 96, theinput voltage range is from 4V to 9V, with a 5V output. Themagnetic coupling of inductors L1A and L1B is not criticalfor operation, but generally they are wound on the samecore. C2 couples the inductors together and eliminates the
LT1370
VIN
VC
VIN2.7V TO 13V
*PULSE PE-53719 (619) 674-8100
GND
NFB
1370_04.EPS
VSWS/S
D2P6KE-15AD31N4148
D1MBRD835L
C1100µF
C20.047µF
C30.0047µF
R12k
R32.49k1%
R22.49k1%
–VOUT†
–5V
C4100µF× 2
ONOFF
VIN3V5V9V
IOUT1.75A2.25A3.0A
2
1
4T1*
3•
•
†MAX IOUT
++
FB
LT1370
VIN
GND
VIN4V TO 9V
VC
FB
1370_05.EPS
VSWS/S
C133µF20V
C40.047µF
C50.0047µF
R12k
R36.19k1%
R218.7k1%
VOUT†
5V
C3100µF10V× 2
ONOFF
L1A*6.8µH
•
•L1B*6.8µH
C24.7µF
C1 = AVX TPSD 336M020R0200C2 = TOKIN 1E475ZY5U-C304C3 = AVX TPSD107M010R0100BH ELECTRONICS 501-0726(612) 894-9590INPUT VOLTAGE MAY BE GREATER ORLESS THAN OUTPUT VOLTAGE
D1MBRD835L
VIN4V5V7V9V
IOUT1.8A2A
2.6A2.9A
†MAX IOUT
*
**
++
Figure 95. Positive-to-Negative Converter with Direct Feedback
Figure 96. Two Li-Ion Cells to SEPIC Converter
Application Note 84
AN84-55
BOOTSTRAPPED SYNCHRONOUS BOOST CONVERTEROPERATES AT 1.8V INPUTby Tom Gross
Some applications, such as those powered by batteries orsolar cells, see their input voltage decrease as they oper-ate. Many regulators that could operate with high inputvoltages cease to function as the input voltage decreases.The circuit in Figure 97 maintains the maximum loadcurrent as the input voltage drops. The regulator boosts a2.5V–4.2V input to 5V at a maximum load current of 2A(10W of output power).
The circuit is a bootstrapped synchronous boost regulatorusing an LTC1266 synchronous regulator controller.Diodes D1 through D4 allow the circuit to start-up usingthe (low) input voltage and then to be powered duringnormal operation by the higher output voltage. The crucialelements in this circuit are the switches: two IRF7401 N-channel MOSFETs. These MOSFETs are fully enhanced atvery low gate-to-source voltages (at 2V of VGS, the peakdrain current is rated at 15A). The low enhancement
need for a switch snubber network. C2 must have a verylow ESR, because the ripple current is equal to ISW/2. Itscapacitance value is not critical and has no significanteffect on loop stability. The voltage across C2 is equal to
voltages allow the circuit to start up at low input voltages(crucial for low series-cell-count, battery-powered ap-plications). Diodes D3 and D4, along with capacitor C2,form a charge-pump circuit, which the controller uses forthe MOSFETs’ gate drive. The switches are driven by anLTC1266 synchronous regulator controller.
Because the circuit is powered from the 5V output, it willstill operate if the input supply voltage drops below theminimum input voltage of the IC. This bootstrapping
PWR VIN
PINV
BINH
VIN
CT
ITH
BDRIVE
LBOUT
LBIN
TDRIVE
VFB
SHDN
2
3
4
5
6
7
16
14
13
1
10
11
SGND PGND
SENSE+ SENSE–
++
+
VIN2.5V TO 4.2V
C1330µF
6.3VTANT
D1 D2 D3 D4
C510µF16V
C61µF
C7220pF
C80.012µF
R86.2kC90.033µF
R3 0.025Ω
R1100Ω R2
100Ω
C2 1µF
L122µH/7ACOILCRAFTDO-5022-223
R4 2Ω
R5 2Ω Q1IRF7401
D5 MBRS120T3
Q2IRF7401
R6100k1%
R733.2K1%
LTC1266
C41200pF
C33× 330µF6.3VTANT
VOUT5V/2A
12 15
89
D1 TO D4 = MBR0530T1
1000pF
VIN (V)
EFFI
CIEN
CY (%
)
1630_03.EPS
95
90
85
80
75
702.7 2.9 3.1 3.3 3.5 3.7 3.9 4.1 4.3
ILOAD = 0.25A
ILOAD = 1A
ILOAD = 2A
the input. A 4.7µF, 50V ceramic will work in most SEPICapplications. The S/S pin is used as a logical on/off signal.In the off state, there is no leakage to the output, and only12µA leakage from the input.
Figure 97. Bootstrapped Synchronous Boost Converter
Figue 98. Efficiency of Figure 97’s Circuit
Application Note 84
AN84-56
allows the circuit to start up even when the input voltageis below the minimum input voltage of the IC (3.5V). Witha 1A load, the regulator operates down to 1.8V.
Figure 98 shows the efficiency of the regulator versus theinput voltage at three different load currents. At 2A of loadcurrent, efficiency drops as the input voltage is decreased
due to the higher power losses in the inductor. A largerinductor will increase efficiency and/or allow for largerload currents. The efficiency with the indicated inductor isgood, averaging above 83% overall. Higher efficiency willhelp to increase the run time of battery-poweredapplications.
Regulators (Switching)—Buck-Boost
500kHz BUCK-BOOST CONVERTERNEEDS NO HEAT SINKby Mitchell Lee
Thanks to an efficient 0.25Ω switch, the LT1371 SEPICconverter shown in Figure 99 operates at full power withno heatsink. Up to 9W at 5V output is available, and thecircuit works over a wide range of input voltages extendingfrom the LT1371’s 2.7V minimum to 20V, limited by therating of the capacitors.
A 1:1 bifilar-wound toroid is used as the magnetic ele-ment. A careful analysis showed that, in spite of the500kHz operating frequency, a high permeability (mr =125) Magnetics Inc. Kool Mµ® core exhibited the bestefficiency when compared to powdered iron materials.Copper loss is minimized by the use of the high-perm KoolMµ material, with only a slight core-loss penalty.
FBNFB
S/S
NC
OFF ON
SWVIN
2.7V TO 20VINPUT
VCGND
LT1371
33µF20V
OS-CON
L1 HL-8798
100µF20V
OS-CON
150µF6.3VOS-CON5VOUTPUT
3.6k
1.2k
4.7nF
47nF
20k
MBRS340T3
L1 = HURRICANE ELECTRONICS LAB HL-8798 (801) 635-2003, FAX (801) 635-2495
COILTRONICS CTX10-4 (561) 241-7876, FAX (561) 241-9339
++
+
Maximum available output current varies with input volt-age, and is shown (for 3A peak switch current) in Figure100. Efficiencies for several input voltages are shown inFigure 101. At a 2.7V input, most of the loss is tied up inthe LT1371 switch, whereas the output diode is thedominant source of loss with high inputs. Because theselosses are small, surface mount construction providesadequate dissipation, eliminating the need for heat sinks.
In this application, the synchronization feature of theLT1371 is not used. When driven with an external clock atthe shutdown/sync pin (S/S), the chip can be synchro-nized to any frequency between 600kHz and 800kHz.
INPUT (V)
0
1000
500
1500
2000
OUTP
UT (m
A)
200 105 15
LOAD (mA)
50
70
60
80
90
EFFI
CIEN
CY (%
)
20000 1000500 1500
VIN = 12V
VIN = 5V
VIN = 2.7V
Figure 99. 5V, 9W Converter Operates Over a Wide Input Rangewith Good Efficiency
Figure 100. Maximum Avialable Output Current
Figure 101. Efficiency of Figure 99’s Circuit
Application Note 84
AN84-57
BATTERY-POWERED BUCK-BOOST CONVERTERREQUIRES NO MAGNETICSby John Seago
One of the problems that designers of portable equipmentface is generating a regulated voltage that is between thecharged and discharged voltage of a battery pack. As anexample, when generating a 3.3V output from a 3-cellbattery pack, the regulator input voltage changes fromabout 4.5V at full charge to about 2.7V when discharged.At full charge, the regulator must step down the inputvoltage, and when the battery voltage drops below 3.3V,the regulator must step up the voltage. The same problemoccurs when a 5V output is required from a 4-cell inputvoltage that varies from about 3.6V to 6V. Ordinarily, aflyback or SEPIC configuration is required to solve thisproblem.
The LTC1515 switched capacitor DC/DC converter, canprovide this buck-boost function for load currents up to50mA with only three external capacitors. The circuitshown in Figure 102 will provide a regulated 3.3V outputfrom a 3-cell input or a 5V output from a 4-cell input.Connecting the 5/3 pin to VIN will program the output to
ON OFF
RESET
5V 3.3V
SHDN
POR
5/3
VIN
VOUT
GND
CI+
CI–
1
2
3
7
8
4
6
5
3 OR 4CELLS
C210µF10V(1206)
C310µF10V(1206)C1
0.1µF(0603)
3.3V/50mAOR5V/50mA
LTC1515CS8-3.3/5
R1 100k
AVX 0603YC104MAT2ATAIYO YUDEN LMK316F106ZL
C1 =C2, C3 =
5V, whereas grounding the 5/3 pin programs the output to3.3V.
The absence of bulky magnetics provides another benefit:this circuit requires only 0.07 square inches of boardspace in those applications where components can bemounted on both sides of the board. The addition of R1provides a power-on-reset flag that goes high 200ms afterthe output reaches 93.5% of its programmed value. TheSHDN pin allows the output to be turned on or off with a3V logic signal.
Regulators—Switching (Inverting)
MAKING –5V 14-BIT QUIETby Kevin R. Hoskins
Many high performance data acquisition systems reapmultiple benefits when using ±5V supplies rather than asingle 5V supply. These benefits include the ability tohandle larger signal magnitudes than is possible with asingle 5V supply. This increases a system’s dynamic rangeand helps improve the signal-to-noise ratio. Operating on±5V also increases headroom, which is important forsignal conditioning. Compared to operating on 5V, condi-tioning circuitry operating on ±5V has twice the headroom,allowing it to easily handle ±2.5V signals without clipping.Additionally, the greater headroom avoids the limitationsof rail-to-rail operation and widens the selection of highperformance operational amplifiers and analog-to-digitalconverters, such as the LTC1419.
Although a switching or charge-pump power supply is anefficient way to create a –5V supply from a single 5V
RIPPLE FREQUENCY (Hz)
AMPL
ITUD
E OF
POW
ER S
UPPL
Y FE
EDTH
ROUG
H (d
B)
0
–20
–40
–60
–80
–100
–1201k 100k 1M 10M
1410 G08
10k
VRIPPLE = 0.1V
VSS
VDD
DGND
supply, they are not generally recommended for use withADCs. Typical ADCs have inadequate PSRR, which de-creases with increasing frequency. This poor PSRR per-formance cannot sufficiently attenuate the noise createdby switching or charge-pump supplies. However, LTC’s
Figure 102. Battery-Powered Buck-Boost Converter
Figure 103. The LTC1419’s Positive Supply PSRR of 90dB to200kHz is a Significant Contributor to this ADC’s WidebandConversion Performance and 80dB SINAD
Application Note 84
AN84-58
new family of ADCs, here represented by the LTC1419,has excellent PSRR. This family makes it easy to achievehigh performance data conversion, even at 14 bits, usinga switch-mode regulator for a –5V supply.
The LTC1419’s high PSRR is shown in Figure 103. Itshows that when operating on ±5V, the negative andpositive PSRR are typically 80dB and 90dB, respectively,up to 200kHz for a 100mV ripple voltage. Combined withproper layout, the LTC1419’s high PSRR allows it toconvert signals without signal degradation while usingswitching regulators and charge pumps to generate its–5V supply. Applications including high speed communi-cations, high resolution signal processing and widebandmultiplexing benefit from the LTC1419’s advantages—its20MHz S/H bandwidth, 800ksps conversion rate and 14-bit resolution. This article shows two supply designs thatare quiet enough to use with the LTC1419.
Low Noise Inverting Converter
The LT1373 switching regulator shown in Figure 104 isconfigured as an inverting converter, creating –5V from5V. This configuration has the advantage of a smalltriangular switching-current waveform through the sec-
ondary inductor. This current waveform is continuous,producing much less harmonic content than is created bya typical positive-to-negative voltage converter, with itsrectangular switching current waveform. With the compo-nents shown, the LT1373 operates continuously with loadcurrents above 10mA. Because the LTC1419 typicallydraws 18mA of negative supply current, the LT1373 willalways operate in the quiet continuous mode.
Regulated Charge Pump Converter
The LTC1419’s negative PSRR also allows the use ofcharge pumps to create –5V. The circuit shown in Figure105 uses the LT1054 regulated charge pump. This circuithas the advantage of reduced board space, since it lacks aninductor and requires fewer passive components. How-ever, the LT1373 circuit can supply more current (150mA)than the LT1054 circuit (100mA).
Performance Results
What is the effect of using either of these switch-basedsupplies on the LTC1419’s conversion performance? TheFFTs in Figures 106–108 show the excellent results. Figure
+
+
DI_1419_01.eps
AVDD28
+AIN1
DVDD27
–AIN2
VSS26
VREF3
BUSY25
COMP4
CS24
C71µF CER
C5
C822µF10V
TANT
C6
5V
ANALOG INPUT
U1LTC1419
AGND5
CONVST23
D13 (MSB)6
RD 22
MICROPROCESSOR/MICROCONTROLLERINTERFACED12
7
SHDN21
D118
D020
D109
D119
D910
D218
D811
D317
D712
D416
D613
D515
DGND14
VSW
1
2
4
3
L1
VIN5
NFB3
8
S/S4
GND7
VC
C90.01µF
D1
R34.99k
R54.99k1%
R6499Ω1%
R44.99k1%
C11100µF
10VTANT
C120.1µF
1GND S
6
U2LT1373
C1010µFCER
INVERTINGCONVERTER
C5, C6, C7 = 10µF CERAMICL1 = OCTAPAC CTX-100-1D1 = 1N5818
Figure 104. The LTC1419’s 80dB PSRR Allows the LTC1373 to Generatethe –5V and Power the ADC without Signal-Coversion Degradation
Application Note 84
AN84-59
DI_1419_02.eps
281
272
263
BUSY25
COMP4
CS24
C71µF CER
C5C1
10µFTANT
C6
5V
ANALOGINPUT
U2LTC1419
AGND5
CONVST23
D13 (MSB)6
RD 22
MICROPROCESSOR/MICROCONTROLLERINTERFACED12
7
SHDN21
D118
D020
D109
D119
D910
D218
D811
D317
D712
D416
D613
D515
DGND14
V+FB/SHDN1
VREF6
OSC7
8
CAP+2
GND3
VOUT5
CAP–4
U1LT1054
R2, 120k
R1, 30.1k C30.002µF
C4100µFTANT
C22µF
C5, C6, C7 = 10µF CERAMIC
AVDD+AIN
DVDD–AIN
VSSVREF
+
+
FREQUENCY (kHz)
–160
–80
–100
–120
–140
0
–20
–40
–60
AMPL
ITUD
E (d
B)
400
DI_1419_03.eps
0 50 100 150 200 250 300 350
LTC1419±5V LAB SUPPLIESfSAMPLE = 800kHz
fIN = 91kHzS/N = 80.5dB
FREQUENCY (kHz)
–160
–80
–100
–120
–140
0
–20
–40
–60
AMPL
ITUD
E (d
B)
400
DI_1419_05.eps
0 50 100 150 200 250 300 350
LTC14195V LAB SUPPLY –5V LT1054fSAMPLE = 800kHz
fIN = 91kHzS/N = 80.8dB
FREQUENCY (kHz)
–160
–80
–100
–120
–140
0
–20
–40
–60
AMPL
ITUD
E (d
B)
400
DI_1419_04.eps
0 50 100 150 200 250 300 350
LTC14195V LAB SUPPLY–5V LT1373fSAMPLE =800kHz
fIN = 91kHzS/N = 80.5dB
Figure 105. The LTC1419’s High Negative Supply PSRR alsoAllows the Use of the LT1054 Regulated Charge Pump toGenerate –5V without Loss of Performance
Figure 106. This FFT of an LTC1419 Powered by a ±5VLab Supply Shows a SINAD of 80.5dB for a 91kHz Inputsampled at 800ksps
Figure 107. When the –5V Supply is Generated by an LT1373Switching Regulator, the SINAD, the Noise Floor, and the 91kHzFundamental’s Harmonics Remain Essentially the Same asin Figure 106
Figure 108. When the –5V Supply is Generated by an LT11054Inverter, the SINAD, the Noise Floor, and the 91kHz Fundamental’sHarmonics Are Again Unchanged from Those in in Figure 106
106 is an FFT of a typical LTC1419 operating on ±5V froma lab supply and converting a full-scale 91kHz sine waveat 800ksps. The noise floor is approximately 114dB belowfull scale, the second harmonic’s amplitude is approxi-mately 90dB below full scale and the SINAD is 80.5dB.Figure 107 shows the FFT of the same LTC1419 operatingon a 5V lab supply and –5V from the LT1373 circuit. Thenoise floor and the second harmonic’s amplitude remainthe same relative to full scale and the SINAD remains thesame at 80.5dB. Figure 108 shows the LTC1419’s re-sponse when its –5V is generated by the LT1054 circuit.As with the LT1373 circuit, the noise floor and the ampli-tude of the harmonics remain the same and the SINAD is80.8dB.
Application Note 84
AN84-60
NEGATIVE-TO-POSITIVETELECOMMUNICATION SUPPLYby Kurk Mathews
Many telecommunication circuits require a positive sup-ply voltage derived from a –48V input. The traditionalapproach to negative-to-positive conversion has been touse a buck-boost converter (see Figure 109). Unfortu-nately, this topology suffers drawbacks as the power leveland input-to-output voltage difference increases.
A more appropriate solution for –48V to 5V conversion isshown in Figure 110. The LT1680 is used to implement aforward converter with its output referenced to the inputcommon. Compared to the buck-boost converter, switchcurrent is reduced by a factor of two and output capacitorripple is reduced by a factor of five.
The LT1680 is referenced to –48V and requires a 12V biassupply. The 12V is generated by using the RUN/SHDN and
+
+
–VIN
+VOUT
DI_1680_01.eps
9 10 13 15 1
7
8126543216
14
11
5VREF CT IAVG SS VC SGND PGND VREF
S+ S– GATE SYNC SL_ADJRUN/SHDN
12VINVFB
LT1680
+
++
+
+
+
T1
33Ω
33Ω
1.5nF
1.5nF
MBR2045CT
L1, OUTPUT20µH
50Ω1W
330µF6.3V
SANYO OSCON
1k
300pF
4.22k
Q32N5401
IRF64010Ω
MBR0520LT1
220µF*35V
220µF*35V
220µF*35V
220µF*35V
24k
24k
1µF63V
R124k
1.2k
0.1µF
0.1µF
0.22µF
1k
2.2nF
1nF
16k
20k1µF
R57.5k
R34.75k
Q22N3904
Q12N7000
20k
BAV21
R21M
C1220µF 35V
0.1µF
R478.7k
L1BIAS
–48V
INPUTCOM
5V/6A
T1 = COILTRONICS VP5-1200, 1:1:1:1:1:1(SIX WINDINGS, EACH 77µH)
L1 = PHILIPS EFD20-3F3-E63-S CORE SET (Al = 63nH/T2)OUTPUT 18T BIFILAR 22AWG, BIAS 54T BIFILAR 32 AWG
* = SANYO CV-GX
0.015Ω1W
a bootstrap winding on the output inductor, L1. When inputvoltage is first applied, R1 begins charging C1. As C1charges, Q1 is held on by R2, shorting R3. R4 and R5 forma voltage divider that holds the RUN/SHDN pin below its1.25V threshold until the 12VIN pin reaches approximately14V. Once out of standby, Q1 is turned off by Q2, reducingthe run threshold to approximately 9V and allowing C1 timeto discharge slightly before the overwinding on L1 takesover. The only remaining issue is feedback. Q3 translatesthe output voltage to a current, which flows to the VFB pin.
Figure 109. Buck-Boost Converter
Figure 110. –48V to 5V Telecommunications Supply
Application Note 84
AN84-61
POSITIVE-TO-NEGATIVE CONVERTER POWERS–48V TELECOM CIRCUITSby Mitchell Lee
If you’re designing a system that interfaces to telecomequipment, chances are you’ll need a –48V supply. Thecircuit in Figure 111 supplies up to 6W at –48V and scalesto more than 12W with higher power components. Basedon the inverting topology, the converter exhibits excellentefficiency over a wide range of loading conditions (seeFigure 112).
The LT1171’s error amplifier is designed for positive-boost applications, and hence its gain and reference are ofthe wrong phase and polarity for sensing an invertedoutput. In this application, the error amplifier is simplybypassed and feedback is applied at the compensation(VC) pin. Zener diode D2 senses the output, pulling downon Q1 and the VC pin, in response to small increases inoutput voltage. Pulling down on the VC pin reduces peakswitch current, and constitutes negative feedback. If theoutput is a little low, the Zener’s diminished feedbacksignal is overcome by an internal 200µA current source atthe VC pin, thereby increasing peak switch current andrestoring the output voltage.
The combination of the LT1171 and the VP-2 seriesVERSA-PAC™ coil (CTX02-13836) are suited for 120mAoutput current as shown. For lighter loads of up to 60mA,use the LT1172 and a VP-1-series equivalent to the coilshown. For up to 15W, use the LT1171 and a VP-5equivalent. High voltage versions of the LT1170 family(-HV) allow inputs of up to 20V without exceeding the peakswitch-voltage rating.
This converter starts working at 2.7V and will regulate–48V at reduced power. You can add undervoltage lockoutby inserting a Zener diode (VZ = VLOCKOUT – 2.7V) betweenthe input supply and the LT1172’s VIN pin.
The LT1680’s unique differential current sense amplifierhas an input common mode range of –0.3V to 60V. If VINis expected to exceed 60V, the sense resistor could berelocated in the main FET’s source and the input capaci-
tors’ voltage increased. Because the forward converter isfundamentally an isolated topology, an optocoupler andreference could be added to provide isolation between theinput and output of the supply.
VC GND FB
VIN VSW
+
CTX02-13836†VIN12V
220µF16V
220µF63V
150µF63V
D3**
100nF 1k
1µF
D1*
Q12N3904 1k
1k
D21N5261B47V
10nF
VOUT–48V/120mA
D1 = 1N4148D3 = MUR120COILTRONICS(561) 241-7876
***†
1
122
114
9
3
105
86
7
LT1172HV
GND
+
+
0
EFFI
CIEN
CY (%
)
100
90
80
70
60
50
DI_1171_02. EPS
12060LOAD CURRENT (mA)
30 90
Figure 111. 12V to –48V Converter Features GoodEfficiency Over a Wide Range of Loads
Figure 112. Converter EfficiencyRises to 80% at Only 20mA Load
Application Note 84
AN84-62
LOW NOISE LT1614 DC/DC CONVERTER DELIVERS–5V AT 200mA FROM 5V INPUTby Steve Pietkiewicz
The inverting DC/DC converter function is traditionallyrealized with a capacitor-based charge pump. Althoughsimple, the output impedances of the best charge pumpsolutions are in the 5Ω to 10Ω range, resulting in significantregulation issues when the load current increases be-yond a few tens of milliamperes. The LT1614 inductor-based inverting DC/DC converter uses closed-loop regu-lation to obtain an output impedance of 0.1Ω, eliminatingoutput voltage droop under load.
Figure 113 details the 5V to –5V converter circuit. TheLT1614 contains an internal 0.6Ω switch rated at 30V,allowing up to 28V differential between input and output.Quiescent current is 1mA and the device contains a low-battery detector with a 200mV reference voltage. Thedevice switches at 600kHz, allowing the use of small,inexpensive external inductors and capacitors. In fact, thetotal cost of the components specified in Figure 113(excluding the LT1614) is approximately $0.70 in 10,000-piece quantities.
The LT1614 operates by driving its NFB pin to a voltage of–1.24V, allowing direct regulation of the negative output.This converter topology, which consists of inductors in
series with both input and output, results in low outputnoise and also in low reflected noise on the 5V inputsupply. The output and switch nodes are shown in Figure114. Output ripple voltage of 40mV is due to the ESR of thetantalum output capacitor C2. Ripple voltage can be re-duced substantially by replacing output capacitor C2 witha 10µF ceramic unit, as pictured in Figure 115.
+
+ LT1614SHDN
VCGND
NFB
VIN SW
VIN5V
C133µF
L122µH
L222µH
C31µF
D1
VOUT–5V/200mA
C233µF
R169.8k
R224.9k
MBR0520 (800) 441-2447MURATA LQH3C220 (814) 237-1431AVX TAJB336M010 (803) 946-0362AVX1206YC105KAT (CERAMIC, X7R)
D1:L1, L2:C1, C2:
C3:
100k
1nF
Figure 113. 5V to –5V DC/DC Converter Uses an InvertingTopology and Delivers 200mA.
Figure 115. LT1614 Output and Switch Node with a 10µFCeramic Output Capacitor and 200mA Load Current
Figure 116. Improper Placement of D1’s CathodeResults in 60mV Switching Spikes at Output, Evenwith a 10µF Ceramic Output Capacitor
Figure 114. LT1614 Output and Switch Node with a 33µFTantalum Capacitor and 200mA Load Current
VOUT100mV/DIV
AC COUPLED
VSW5V/DIV
500ns/DIV
VOUT100mV/DIV
AC COUPLED
VSW5V/DIV
500ns/DIV
VOUT100mV/DIV
AC COUPLED
VSW5V/DIV
500ns/DIV
Application Note 84
AN84-63
LOAD CURRENT (mA)3
EFFI
CIEN
CY (%
)
90
80
70
60
50
4010 30 100 300
1610 TA02Figure 117. 5V to –5V Converter Efficiency Reaches 73%
In layout, be sure to tie D1’s cathode directly to theLT1614’s GND pin, as shown in Figure 113. This keeps theswitching current loops tight and prevents the introductionof high frequency spikes on the output. The low noise thatcan be achieved with a ceramic capacitor may be corruptedby noise spikes if proper layout practice is not followed. Toillustrate this point, output and switch waveforms fromFigure 113’s circuit, with a 10µF ceramic output capacitorand 200mA load, but with D1’s cathode arbitrarily con-nected to the ground plane, are shown in Figure 115. 60mVswitching spikes ruin an otherwise clean output.
Efficiency of the circuit is detailed in Figure 117. Efficiencyreaches 73% at a 50mA load, and is above 70% at a200mA load. Larger inductors with less copper resistance
–48V TO 5V DC/DC CONVERTER OPERATESFROM THE TELEPHONE LINEby Gary Shockey
DC/DC converters for use inside the telephone handsetrequire operation from the high source-impedance phoneline. Additionally, the CCITT specifications call for on-hook
power consumption of 25mW maximum. The DC/DCconverter circuit presented here is 70% efficient at aninput power of 25mW, providing 5V at 3.4mA. Controlled,low peak switch current ensures that the –48V input linedoes not experience excessive voltage drops duringswitching.
DI_48-5_01.eps
LT1316
LBI
LB0 FB
SHDN7
1
2
R3604k1%
Q32N3904
– 48V
R21.30M1%
R569.8k1%
R6121k1%
R7432k, 1%
Q2MPSA928
3 4
6R42M
R11.3M
C10.1µF
C20.022µF
C447µF
D21N4148
C347µF
D11N5817T1
10:1:1
L3L1
VOUT5V
5
Q1D3
1N4148
VIN SW
RSET GND
L2VA
R6, Q2 AND R7 MUST BE PLACED NEXT TO THE FB PIN
T1 =DALE LPE-4841-A313, LPRI = 2mH (605) 665-1627Q1 =ZETEX ZVN 4424A (516) 543-7100
+
+
can be used to increase efficiency, although such induc-tors are more expensive than the Murata units specified.
Figure 118. –48V to 5V Flyback Converter
Application Note 84
AN84-64
Figure 119. Switch Voltage and Current Waveforms Figure 120. Output Ripple Voltage and Current Waveforms
VOUTAC COUPLED
100mV/DIV
SECONDARYCURRENT
200mA/DIV
PRIMARYCURRENT50mA/DIV
SWITCH PINVOLTAGE10V/DIV
SECONDARYCURRENT
200mA/DIV
PRIMARYCURRENT50mA/DIV
1µS/DIV
The circuit shown in Figure 118 operates as a flybackregulator with an auxiliary winding to provide power forthe LT1316. To understand the operation of this circuit,examine Figure 118. When power is first applied, the LBIpin is low, causing the SHDN pin to be grounded throughLBO. This places the part in shutdown mode and only thelow-battery comparator remains active. During this state,VIN rises at a rate determined by R1 and C1. The LT1316draws only 6µA in shutdown mode; R1 needs to supplyonly this current, the current through R2 and R4, and C1’scharging current. When LBI reaches 1.17V (VIN ≈ 3.7V)the LBO pin lets go of SHDN and the part enters the activemode. Once this state is reached, switching action beginsand the output voltage begins to increase. As the deviceswitches, the LT1316 VIN pin draws current out of C1; VINthen decreases sufficiently to trip the low-battery detec-tor, stopping the switching. Start-up proceeds in thisirregular fashion until, eventually, the voltage at VA increasesto 5V. (VA is the same as VOUT, because L2 and L3 have thesame number of turns.) After start-up, current is suppliedto the LT1316 from VA rather than from the –48V rail,
increasing efficiency. VOUT must not be loaded until itreaches 5V or the circuit will not start.
During each switch cycle, current in the transformerprimary ramps up until current limit is reached (SeeFigures 119 and 120). This peak switch current can be setby adjusting R5. The circuit shown uses a 69.8kΩ resistorto give a peak switch current of 50mA. Increasing R5decreases the current limit. Secondary peak current will beapproximately equal to the primary peak current multi-plied by the transformer turns ratio. The FB pin has a sensevoltage of 1.23V and VOUT can be set by the followingformula:
VOUT = 1.23(R7/R6) + 0.6V.
Efficiency versus load current is detailed in Figure 121.Note that for the range of 4mA to 80mA, 70% efficiency orgreater is achieved. Figure 122 shows input current versusoutput power. Less than 80µA quiescent current flowswhen the converter supplies 0.5mW over the 36V–72Vrange.
LOAD CURRENT (mA)
40
70
60
50
80
90
EFFI
CIEN
CY (%
)
100
DI_48-5_04.eps
1 10
VIN = 48V VIN = 36V
VIN = 72V
POWER OUT (mW)
0
0.1
0.2
0.3
INPU
T CU
RREN
T (m
A)
5
DI_48-5_05.eps
0 1 2 3 4
VIN = 48V
VIN = 72V
VIN = 36V
Figure 121. Efficiency vs Load Current Figure 122. Input Current vs Power Out
50µS/DIV
Application Note 84
AN84-65
Regulators—Switching (Flyback)
THE LT1425 ISOLATED FLYBACK CONTROLLERby Kurk Mathews
Introduction
Low voltage circuitry, such as local area networks (LAN),isolation amplifiers and telephone interfaces, frequentlyrequires isolated power supplies. The flyback converter isoften the choice for these low power supplies because ofits simplicity, size and low parts count. Unfortunately,designers are forced to add optocouplers and refer-ences in order to achieve the desired output regulation andtransient response.
The LT1425 provides a one-chip solution for these andother applications. The LT1425 is a 275kHz current modecontroller with an integral 1.25A switch designed primarilyto provide well regulated, isolated voltages from 3V–20Vsources. The LT1425 is available in a 16-pin SO. Featuresinclude a new error amplifier and load compensationcircuitry that eliminate the need for optocouplers whilemaintaining output regulation typically within a few percent.
Figure 124. Transient Response of LT1425 5V to –9V Converter
200mV/DIV
100mA/DIV
5ms/DIV
1425_01.eps
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
GND
N/C
RFB
VC
RREF
SYNC
SGND
GND
3.01k1%
R3
R1R2
2
4
1
3
7
MBRS130LT3
T1
6
22.1k1%
0.1µF
LT1425
47pF
C5
C6
C310µF25V
C410µF25V
1.8k
OUTCOM
–9V
1000pF
C110µF25V
5V
INPUTCOM
C210µF25V
0.1µF
100k
GND
SD
ROCMP
RCMPC
VIN
VSW
PGND
GNDC1, C2, C3, C4 = MARCON THCS50E1E106Z CERAMIC
CAPACITOR, SIZE 1812. (847) 696-2000
D1
D2
1TREMROFSNART
L IRPSNRUTOITAR NOITALOSI
EZISL( × W × )H I TUO YCNEICIFFE 1D 2D 2R,1R 6C,5C 3R
ELAD703A-1484-EPL 63 µH 1:1:1 CAV005 7.01 × 5.11 × mm3.6 Am052 %67 DESUTON DESUTON 74 Ω Fp033 k3.31
SCINORTLIOC38431-20XTC 72 µH 1:1 CAV005 41 × 41 × mm2.2 Am002 %07 8425N1 1LT0450RBM 57 Ω Fp022 k9.5
Typical Applications
Figure 123 shows a typical flyback LAN supply using theLT1425. Figure 123 also includes details on an alternatetransformer for a complete PCMCIA type II height solu-tion. The output voltage is within 1% of –9V for loadcurrents of 0mA–250mA. Input current is limited to 0.35amps in the event the output is short circuited. The outputvoltage droops only 300mV during a 50mA to 250mA loadtransient (see Figure 124). The off-the-shelf transformersprovide 500VAC of isolation. The high switching frequencyallows the use of small case size, low cost, high valueceramic capacitors on the input and output of the supply.
Figure 126 shows a ±15V supply with 1.5kV of isolation.Output regulation remains within ±3% over the entire 5Vto 15V input voltage and ±60mA output current range,
Figure 123. 5V to –9V/250mA Isolated LAN Supply
Application Note 84
AN84-66
Figure 125. Switch Voltage and Current forFigure 123’s Circuit with Outputs of–9V/250mA and –9V/30mA
D: ISW = 0.2A/DIV
C: VSW = 20V/DIV
B: ISW = 1A/DIV
A: VSW = 20V/DIV
1µs/DIV
even with one output fully loaded and the other unloaded(±1.5% with input voltages of 10V–15V). The isolationvoltage is ultimately limited only by bobbin selection andtransformer construction.
Figure 127 implements a 12V to 5V/1A step-down regula-tor with off-the-shelf magnetics. The circuit uses an exter-nal, cascoded 100V MOSFET to extend the LT1425’s 35Vmaximum switch voltage limit. D1 and Q1 ensure theLT1425 does not start until almost 9V, guaranteeingadequate gate voltage for the MOSFET. The MUR120prevents the source from rising above the gate at turn-off.
The circuit in Figure 128 achieves even higher inputvoltages, this time in the form of a –48V to 5V/2A isolatedtelecom supply. The input voltage is too high to directlyrun Q1 or the LT1425, so a bootstrap winding is used toprovide feedback and power for the IC after start-up. Thevoltage to the VIN pin is controlled by D1, D2, Q2, Q3 andassociated components, which form the necessary start-up circuitry with hysteresis. Nothing happens until C1charges through R1 to 15V. At that point, Q2 turns on Q3,pulling the shutdown pin high. Q3, in turn, latches Q2 on,setting the turn-off voltage to approximately 11V. Switch-ing begins and, before C1 has a chance to discharge to11V, the bootstrap winding begins to supply power. If theoutput is shorted, R2 prevents C1 from being charged bythe transformer’s leakage energy, causing the supply tocontinually attempt to restart. This limits input and outputcurrent during a short circuit. Feedback voltage is feddirectly through a resistor divider to the RREF pin. Thesampling error amplifier still works, but the load compen-sation circuitry is bypassed. This results in a ±5% loadregulation over line and load. A dedicated feedback wind-ing referencing the feedback voltage to the VIN pin couldbe used to include the load compensation function andimprove regulation.
++
+
1425_06.eps
GND
N/C
LT1425
MBRS1100T3
MBRS1100T3
1 9
4
7
T1
12
11
2 6
RFB
VC
RREF
SYNC
SGND
GND
GND
SD
ROCMP
RCMPC
VIN
VSW
PGND
GND
1
2
3
4
5
6
7
8
16
15
14
13
12
11
103.01k
1%
1N759
18.4k0.1%
3k
+15V
–15V
OUTCOM
7.32k1%
75
5V TO15V
INPUTCOM
0.1µF
220pF
1µF22µF35V
15µF35V
3k15µF35V
1000pF
0.1µF
130330pF
9
MBR0540LT1
T1: COILTRONICS CTX02-13498 (516) 241-7876
Figure 126. Fully Isolated ±15V, ±60mA Supply
Application Note 84
AN84-67
++
+220µF10V
1425_07.eps
GND
N/C
LT1425
MBRS340T3
2
5
1
4
6
3
10
7
11
8
12
9
RFB
VC
RREF
SYNC
SGND
GND
GND
SD
ROCMP
RCMPC
VIN
VSW
PGND
GND
1
2
3
4
5
6
7
8
16
15
14
13
12
11
103.01k
1%
25.5k1%
9.3k1%
MMFT1N10E
2.4k
12V
INPUTCOM
0.1µF
22µF35V 220µF
10V 200Ω
5V
OUTCOM
COILTRONIXVP1-0190TURNS RATIO 1 : 1 : 1 : 1 : 1 : 112µH PER WINDING(561) 241-7876
1000pF
1000pF
MUR120
Q12N3906
0.1µF
100Ω
10
1.8k
330pF
9
D11N7557.5V
+
+ +
1425_08.eps
GND
N/C
BAV21
BAV21
MUR120
LT1425
5k
18
MBR745
10
4 11, 12
9, 10
T1
2
RFB
VC
RREF
SYNC
SGND
GND
GND
SD
ROCMP
RCMPC
VIN
VSW
PGND
GND
1
2
3
4
5
6
7
8
16
T1
1
6
15
14
13
12
11
103.16k1%
Q22N3906
Q32N3904
Q1IRF610
D17.5V
1N755
D27.5V1N755
30.1k1%
R218
R124k
50Ω1W510
10k
2.4k
100k
INPUTCOM
–36V TO–72V
3.3µF
150pF
0.1µF
0.1µF C127µF35V
150µF6.3V
150µF6.3V
5V
OUTCOM
1000pF
470pF
9
T1: COILTRONICS CTX 02-14143-X3(561) 241-7876
Figure 127. 5V/1A Step-Down, Isolated Supply
Figure 128. 5V/2A Telecommunications Supply
Application Note 84
AN84-68
HIGH ISOLATION CONVERTER USESOFF-THE-SHELF MAGNETICSby Mitchell Lee
Isolated flyback converters usually evoke thoughts (orbitter memories) of custom transformers, slipped deliveryschedules and agency approval problems. Off-the-shelfflyback transformers are available from several vendors,but these carry isolation ratings of only 300V–500V, and,rarely, of up to 1kV. Flyback transformers with isolationratings of 3750VRMS are impossible to find, and if anapplication requires this level of isolation, an expensive,custom design is likely the only solution.
Gate-drive transformers, designed to couple switchingregulator controllers to MOSFET gates, are readily avail-able from stock with high isolation ratings and low cost.These are wound on ungapped cores and have very highinductance (500µH to 2mH), and will quickly saturate in anormal flyback converter circuit.
The transformer used in Figure 129’s circuit handlessignificant current without saturating. The converteroperates from a 12V battery-backed input supply andoutputs 24V at 200mA. The key feature is that the secondcoil is not a coil at all, but rather an off-the-shelf gate drive
VIN10V TO 15V
VOUT24V/200mA
L1*100µH
C1100µF50V
D1P6KE36A
D2MUR120 C3
100µF50V
D31N914
R1200Ω
R218k
R31k
C410nF
T1**
PE63387
D4MUR120
C5100µF50V
R43.6k(6.7mA MIN LOAD)
LT1172
VIN SW
FB
GND VC
C2100nF
L1 = PE53829T1 = PE63387
PULSE ENGINEERING (619) 674-8100
***
+ +
+transformer. This component offers 3750VRMS isolationand full VDE approval at a lower cost than a comparablecustom design.
Feedback is derived from the primary winding, throughD3. R1 acts to filter the leakage-inductance spike at switchturn-off, and C4 smooths the recovered feedback voltage.Note that the transformer is wound 1:1; C4 peak detects avoltage roughly equal to the output. Sizing R1 and C4 is atrade-off between minimum load and load regulation. Asshown, a minimum load of 3600Ω is recommended.Output regulation is shown in Figure 130. Line regulationfrom 10V to 20V input at full load is 0.13%/V.
OUTP
UT V
OLTA
GE (V
)
DI_1068_04. EPS
0 50 100 150 200
30
28
26
24
22
20
TOTAL LOAD CURRENT (mA)
Figure 129. 24V/200mA Bulk Supply with 3750VRMS Isolation
Figure 130. Output Regulation for Figure 129’s Circuit
Application Note 84
AN84-69
by Kurk Mathews
Many new switching regulators are designed with a spe-cific application or topology in mind. If your requirementshappen to fall within these parameters, all is well. Unfor-tunately, when faced with unusual requirements, thedesigner is often forced to choose bare-bones, universalregulators. The LTC1624 overcomes these issues byproviding a full featured regulator that can operate in thestep-down (buck), step-up (boost), buck-boost or flybackmode.
This constant-frequency current mode controller includesa high-side differential current sense amplifier and afloating high current N-Channel MOSFET driver. In thebuck mode, an external bootstrap capacitor between theBOOST and SW pins works in conjunction with the internal5.6V regulator and diode to provide a regulated supply fora high-side driver. In the boost, buck-boost or flybackmode, the SW pin is grounded, providing drive for a low-side MOSFET.
+
VIN4.75V–24V
330µF35V
SANYO MV-GX
1µF
0.02Ω
R3220Ω
R147Ω
Q2MPS2222A
Q1IRL540N
C1220pF
0.01µF 1µF
1µF
1k15k
LTC1624
ITH/RUN GND VFB
TG
BOOST
SW
VIN SENSE–
R243k
10Ω
8 1
7
6
5
342
220pF
MURS120T3
MURS120T3
620k
1µF
18k
1µF
18k
T1
3
12
111
28
5
7
6
10
4
9
50V/75mA
–50V/75mA
T1: COILTRONICS VP3-0138, 1:1:1:1:1:1(SIX WINDINGS, EACH 11.2µH)(561) 241-7876
47Ω
An example of a wide-input-range flyback is shown inFigure 131. The circuit provides ±50V at 75mA from a 4.75to 24V source. The sum of line-, load- and cross-regula-tion is better than ±5%. The TG pin voltage is controlled bythe internal 5.6V regulator, allowing the input voltage to beabove Q1’s 16V maximum gate-to-source voltage rating.200kHz fixed frequency operation minimizes the size ofT1. The R-C snubber formed by C1 and R1 in combinationwith T1’s low leakage inductance keeps Q1’s drain voltagewell below its 100V rating. To improve cross-regulation,Q2, R2 and R3 were included to disable Burst Mode™operation (a feature that improves efficiency at light loadconditions by skipping switching cycles). The LTC1624’s95% maximum duty cycle accommodates the 5-to-1 inputvoltage range. Finally, by reconfiguring T1’s secondaries,a variety of output configurations, such as 24V out (fourwindings in parallel), single 50V/150mA or a single 100Voutput, are possible with this same basic circuit.
Figure 131. Wide-Input-Range Flyback Regulator Provides ±50V at 75mA
WIDE-INPUT-RANGE, LOW VOLTAGE FLYBACK REGULATOR
Application Note 84
AN84-70
Regultors—Switching (Low Noise)
THE LT1533 HERALDS A NEW CLASS OFLOW NOISE SWITCHING REGULATORSby Jeff Witt
Introducing the LT1533 Low Noise Switcher
The LT1533 is a switching regulator that provides asolution to EMI problems through two flexible approaches.First, the slew rates of both the current through the powerswitch and the voltage on it are easily programmed withexternal resistors. Limiting these slew rates will removethe highest harmonics from the switching waveforms.Second, the LT1533, with two 1A power switches, isdesigned to operate in push-pull circuits. Such circuits,with their low input and output current ripple, are inher-ently quiet. The result is an integrated switching regulatorthat provides very quiet output power and very low emis-sions. Figure 132 illustrates what can be achieved. The toptrace shows the output of a push-pull boost regulatorgenerating 120mA at 12V from an input of 5V. This tracewas measured using a 10MΩ oscilloscope probe with asix-inch ground lead, demonstrating that there is nosignificant inductively or capacitively coupled noise. Prob-ing the output of the LT1533 circuit with a 50Ω low noise
Figure 132. Output Ripple of an LT1533 Switching RegulatorProducing 120mA at 12V from a 5V Input
5µs/DIV 1533_01.eps
200µV/DIV
20mV/DIV
18k 15k 2.49k
21.5k
0.015µF
LT1533
COL A
PGND
COL B
RCSL
RVSL
FBGND RT CT VC
5V
47µF6.3V
25nH*
T1
4k–68k
4k–68k
1.2k
220pF
D1
D2
L1300µH
L210µH
C122µF20V
C222µF20V
12V/200mAVIN
NC
DUTY
SHDN
SYNC
1500pF
1000pF
1
1
4
4
1533_03.EPS
2
16
15
12
13
7
8
10569
4
11
3
1
14
NFB
*BEAD OR PCB TRACET1 = COILTRONICS CTX02 13666-X1
(561) 241-7876L1 = COILTRONICS CTX300-2L2 = COILTRONICS DT1608C-103D1, D2 = MOTOROLA MBRS1100T3
(800) 441-2447
++
+
amplifier reveals the real performance (second trace):peak-to-peak output ripple of the low noise switcher isonly 150µV in a 10kHz to 100MHz bandwidth.
A Closer Look at the LT1533
The LT1533 is a fixed frequency current mode PWMswitching regulator. The output voltage is regulated bycontrolling the peak switch current on each cycle of theoscillator, resulting in good transient performance andrapid current limiting. The oscillator drives a toggle flip-flop, alternately enabling one of two 0.5Ω NPN powerswitches, QA and QB. The switch current is monitored bya sense resistor at the emitter of the switch. The output
Figure 133. 5V to 12V Push-Pull PWM Converter
Application Note 84
AN84-71
Figure 134. Lowering the Slew Rates of the Power Switches (Trace A) Eliminates High Frequency Ripple at the Output (Traces B and C)
0.2µs/DIV0.2µs/DIV 1533_04.eps
TRACE B20mV/DIV
TRACE A0.5A/DIV
TRACE C500µV/DIV
TRACE B20mV/DIV
TRACE A0.5A/DIV
TRACE C500µV/DIV
voltage (either positive or negative) is compared with anaccurate internal 1.25V reference voltage by an erroramplifier whose current output, along with loop compen-sation components tied to the VC pin, determine the peakswitch current required for regulation; a comparator turnsoff the switch when this current level is reached.
The slew-control circuitry monitors the collector voltagesand emitter currents of the power switches and adjustsbase drive to control both the voltage and current slewrates. The desired rates are programmed by tying the RVSLand RCSL pins to ground with resistors between 4k and68k, corresponding to slew rates from ~80V/µs to 5V/µsand 7A/µs to 0.4 A/µs. This allows the circuit designer to
directly trade off quiet, low EMI operation with highefficiency: low slew rates result in slowly changing strayfields, which generate less interference, but increase theconduction losses in the switches.
The LT1533 oscillator presents additional opportunitiesfor managing EMI. Its wide frequency range (20kHz to250kHz) allows the designer to avoid sensitive frequen-cies. Operating frequency is set with a capacitor on the CTpin and a resistor of nominally 17k on the RT pin. TheLT1533 can also be synchronized to an external clock,allowing accurate placement of both switching frequencyand phase.
18k
LT1533
COL A
PGND
COL BRCSL
RVSL
FB
5V22µF10V
25nH*
T1
68k
4k–68k
VINNC
DUTY
SYNC
1
1
3.3
3.3
1533_05.EPS
3300pF
12V80mA
–12V80mA
4 × 1N5819
LT1121-CS8
LT1175-CS8
22µF35V
22µF35V
150k
150k
2.2µF25V
2.2µF25V
332k
332k
8 1
3 2
1, 2, 7, 8 3
45
14
2
16
15
12
13
7
8
4
11
3
1
2 × BAT85
GND RT CT VC
NFB
SHDN
9 6 5 10
10k
47k
L1100µH
L2100µH
*BEAD OR PCB TRACET1 = COILTRONICS CTX02 13716-X1
(561) 241-7876L1, L2 = COILCRAFT DT1608C-104
(847) 639-6400
+
+
+
+
+
Figure 135. 5V to ±12V DC/DC Transformer
Application Note 84
AN84-72
Push-Pull PWM Makes a Quiet Boost Converter
The push-pull converter in Figure 133 produces 200mA at12V from an input of 5V. The oscillator is set to 80kHz(note that the circuit operates at half this frequency) andthe LT1533 applies a pulse-width modulated 5V to theprimary side of the transformer. The rectified secondaryvoltage is filtered by L1 to generate 12V on C1. In thiscircuit, L1 is the primary energy storage device, so thetransformer can be made fairly small. Additional outputfiltering is provided by L2 and C2.
This topology is inherently quiet. Current through L1 intothe primary output capacitor C1 is a continuous trianglewave with little high frequency content, resulting in lowconducted output noise. With an appropriate transformerturns ratio, RMS input current is kept low, reducing thepotential for conducted noise on the input.
It is advantageous to start with a good topology, but highfrequency noise will still get around via stray capacitanceand mutual inductance; the best way to deal with this is toeliminate fast edges. Figure 134 shows several waveformsfrom the circuit as it delivers 120mA of output current. Theupper trace in each photo is the current in switch QA as itturns off. Trace B is the output voltage probed with a
2200pF 0.01µF
7.50k
18k 10k 2.49k
1000pF
10569
LT1533
COL A
COL B
PGND
RCSL
RVSL
FB
C233µF10V
L1100µH
VINNC
DUTY
SHDN
SYNC
1533_06.EPS
VOUT5V/350mA
D1
C1100µF
10V
VIN3.3V
4k–68k
4k–68k
10Ω
*50nH
14
2
15
16
12
13
1
3
11
4
GND RT CT VC
7
NFB8
*BEAD OR PCB TRACEL1 = COILTRONICS CTX100-4 (561) 241-7876D1 = MOTOROLA MBRS120T3 (800) 441-2447C1 = AVX TPSD107M010R0100 (207) 222-5111C2 = AVX TPSC336M010R0375
++
10MΩ scope probe with a six-inch ground lead. The lowertrace is the output measured with a low noise amplifier. Inthe left photo the switch slew rates are programmed totheir highest values with 3.9k resistors on the RCSL andRVSL pins. The fast switch transients induce high fre-quency ripple on the output (the higher level of noise onthe middle trace is due to the inductance of the scopeprobe’s ground lead). By lowering the slew rates (RCSL =24k and RVSL = 8.2k) this potentially troublesome outputripple is eliminated, as shown in the right photo. Theefficiency penalty is minor; the slower slew rates reduceefficiency from 73% to 70%.
This combination of appropriate circuit topology andcontrolled slew rates produces the exceptionally cleanoutput shown in Figure 132. This circuit is simply imple-mented with ordinary PCB construction, and can be placedin close proximity to sensitive circuits without the need forexpensive electrostatic or magnetic shielding.
DC Transformer with Civilized Edges
Grounding the Duty pin of the LT1533 disables the feed-back loop and runs each switch at 50% duty cycle,allowing the LT1533’s use in DC transformer circuits.Such circuits are useful for generating bipolar or isolatedsupplies; Figure 135 shows an example. The LT1533switches 5V across a 3.3:1 transformer and a diode bridgerectifies the secondary side voltages to produce nominally16V bipolar outputs that are regulated to ±12V. Short-circuit current limit at the output is provided by theLT1533’s switch current limit; the 1A switch limit istransformed to 0.3A on the secondary.
A common problem with isolated-output switchers is thatfast edges couple through stray capacitance between theprimary and secondary windings of the transformer tocreate common mode noise on the outputs. Also, linearregulators are incapable of rejecting high frequency noiseat their inputs. Both problems are greatly reduced bylimiting the switch slew rates. Shielding between thewindings can be eliminated, reducing transformer sizeand cost. LC filters on the isolated side are unnecessarywith the linear regulators rejecting ripple at the operatingfrequency and the controlled slew rates eliminating highfrequency ripple.
Figure 136. 3.3V to 5V Boost Converter
Application Note 84
AN84-73
3.3V to 5V Boost Converter
Simple switching topologies can also benefit from theLT1533’s low noise features. In a boost regulator, forexample, the current into the output capacitor is a squarewave, which contains the high frequency harmonics gen-erated by a fast power switch. Even when the rectifyingdiode is off, fast voltage waveforms at the switch couplethrough the Schottky diode’s capacitance. Fast switchingcan also excite high frequency resonant circuits formed bythe diode’s capacitance and parasitic inductance due toboard traces. All of these effects can be reduced bycontrolling the slew rate of the switch. Figure 136 showsthe LT1533 in a simple boost circuit generating 5.0V froma 3.3V input, a typical requirement when interfacing 3.3Vlogic systems to 5V high performance ADCs. The collec-tors of the two power switches are tied together andalternately energize the boost inductor. Figure 137 showsseveral waveforms at two different slew rate settings withthe circuit delivering 200mA of output current. Trace A isthe switch voltage, trace B is the current through theoutput capacitor and trace C is the AC-coupled outputvoltage in a 100MHz bandwidth. In the left photo, the slewrates are set to their maximum values (RCSL = RVSL =3.9kΩ). The rapidly switched current combined with thefinite series inductance of the output capacitor result inlarge voltage spikes on the output. The right photo shows
the same waveforms with the slew rates lowered (RCSL =RVSL = 22k), eliminating the troublesome transients. Thepenalty is a drop in efficiency from 85% to 80%.
Conclusion: a Switcher for Sensitive Systems
With two 1A power switches, the ability to control positiveor negative outputs, and a wide input operating range (2.7to 30V), the LT1533 is a highly flexible switching regula-tor. Thermal shutdown, in addition to switch-current limit,provides circuit protection. The LT1533 is packaged in thenarrow 16-lead SO, and is available in commercial andindustrial grades.
The LT1533 allows the circuit designer to add a switchingregulator to sensitive analog systems without fear ofintroducing uncontrollable noise and interference. Theprogrammable operating frequency and switch slew ratesallow final tuning to occur in the circuit, when the systemis running and interference problems may first becomeapparent. In addition to providing a way to deal withunforeseen problems, this flexibility means that sacrificesin efficiency will be limited to those needed for propersystem performance. The LT1533 is the switching regula-tor of choice for high performance analog systems.
Figure 137. Limiting Switch Slew Rates (Traces A and B) Lowers the HighFrequency Content of the Boost Regulator’s Output Ripple (Trace C)
5µs/DIV5µs/DIV 1533_07.eps
TRACE B0.5A/DIV
TRACE A5V/DIV
TRACE C20mV/DIV
TRACE B0.5A/DIV
TRACE A5V/DIV
TRACE C20mV/DIV
Application Note 84
AN84-74
LT1533 ULTRALOW NOISE SWITCHING REGULATORFOR HIGH VOLTAGE ORHIGH CURRENT APPLICATIONSby Jim Williams
The LT1533 switching regulator1, 2 achieves 100µV out-put noise by using closed-loop control around its outputswitches to tightly control switching transition time. Slow-ing down switch transitions eliminates high frequencyharmonics, greatly reducing conducted and radiated noise.
The part’s 30V, 1A output transistors limit available power.It is possible to exceed these limits while maintaining lownoise performance by using suitably designed outputstages.
High Voltage Input Regulator
The LT1533’s IC process limits collector breakdown to30V. A complicating factor is that the transformer causesthe collectors to swing to twice the supply voltage. Thus,15V represents the maximum allowable input supply.Many applications require higher voltage inputs; the cir-cuit in Figure 138 uses a cascoded3 output stage toachieve such high voltage capability. This 24V to 5V (VIN= 20V–50V) converter is reminiscent of previous LT1533circuits, except for the presence of Q1 and Q2.4 Thesedevices, interposed between the IC and the transformer,constitute a cascoded high voltage stage. They providevoltage gain while isolating the IC from their large drainvoltage swings.
SYNC
DUTY
SHDN
PGND
NFB
FBRVSLRCSL
LT1533
VINCOL A COL B14
2 15
16
8
7
4
3
11
5
6
10
1500pF
0.002µF
18k
0.01µF
0.002µF
L2
10µF
10k
1k
+
L1100µH
L3100µH
5VOUT
1312
7.5k1%
2.49k1%
220µF
MBRS140
L1, L3: COILTRONICS CTX100-3(561) 241-7876
L2: 22nH TRACE INDUCTANCE, FERRITE BEAD OR INDUCTOR COILCRAFT B-07T TYPICAL(847) 639-6400
Q1, Q2: ON SEMI MTD6N15(800) 282-9855
T1: COILTRONICS VP4-0860 AN70 F40
100µF
CT
RT
VC
OPTIONALSEE TEXT( )+10k
12k
GND
9
24VIN(20V TO 50V)
Q3MPSA42
Q42N2222
4.7µF
10k
9
4
3
101
12
2
11
7
6 T1
5
8
Q21k
220Ω10k220Ω
MBRS140
+
+Q1
Figure 138. A Low Noise 24V to 5V Converter (VIN = 20V–50V): Cascoded MOSFETs Withstand 100V Transformer Swings,Permitting the LT1533 to Control 5V/2A
Application Note 84
AN84-75
A = 5mV/DIV
B = 100µV/DIV
2µs/DIVFigure 139. MOSFET-Based Cascode Permits the Regulator toControl 100V Transformer Swings while Maintaining a LowNoise 5V output. Trace A is Q1’s Source, Trace B is Q1’s Gateand Trace C is the Drain. Waveform Fidelity through CascodePermits Proper Slew-Control Operation
Figure 141. Waveforms for Figure 139 at 10W Output: Trace AShows Fundamental Ripple with Higher Frequency ResidueJust Discernable. The Optional LC Section Results in Trace B’s180µVP-P Wideband Noise Performance
Normally, high voltage cascodes are designed simply forsupply isolation. Cascoding the LT1533 presents specialconsiderations because the transformer’s instantaneousvoltage and current information must be accurately trans-mitted, albeit at lower amplitude, to the LT1533. If this is notdone, the regulator’s slew-control loops will not function,causing a dramatic output noise increase. The AC-compen-sated resistor dividers associated with the Q1–Q2 gate-drain biasing serve this purpose, preventing transformerswings coupled via gate-channel capacitance from corruptingthe cascode’s waveform-transfer fidelity. Q3 and associatedcomponents provide a stable DC termination for the dividerswhile protecting the LT1533 from the high voltage input.
Figure 139 shows that the resultant cascode response isfaithful, even with 100V swings. Trace A is Q1’s source;traces B and C are its gate and drain, respectively. Underthese conditions, at 2A output, noise is inside 400µV peak.
Current BoostingFigure 140 boosts the regulator’s 1A output capability toover 5A. It does this with simple emitter followers (Q1–Q2). Theoretically, the followers preserve T1’s voltage andcurrent waveform information, permitting the LT1533’sslew-control circuitry to function. In practice, the transis-tors must be relatively low beta types. At 3A collectorcurrent, their beta of 20 sources ≈150mA via the Q1–Q2base paths, adequate for proper slew-loop operation.5 Thefollower loss limits efficiency to about 68%. Higher inputvoltages minimize follower-induced loss, permitting effi-ciencies in the low 70% range.
Figure 141 shows noise performance. Ripple measures4mV (Trace A) using a single LC section, with highfrequency content just discernable. Adding the optionalsecond LC section reduces ripple to below 100µV (traceB), and high frequency content is seen to be inside 180µV(note ×50 vertical scale-factor change).
SHDN
DUTY
SYNC
COL A
COL B
PGND
RVSL
RCSL
FBNFB
LT1533
GND
VIN
5V
14
2
15
16
13
12
7
11
3
4
5
6
10
1500pF
18k
0.01µF
10k
R22.49k1%
L2
4.7µF
10k
+T1 L1
300µH12V L3
33µH
89
R121.5k1%
+ +100µF 100µF
1N5817
1N5817
1N4148
1N4148
AN70 F42
CT
RT
VC
OPTIONAL FORLOWEST RIPPLE( )
330Ω
330Ω
0.05Ω
0.05Ω4.7µF+
Q2
Q10.003µF
680Ω
L1: COILTRONICS CTX300-4(561) 241-8786
L2: 22nH TRACE INDUCTANCE, FERRITE BEAD ORINDUCTOR. COILCRAFT B-07T TYPICAL(847) 639-6400
L3: COILTRONICS CTX33-4Q1, Q2: MOTOROLA D45C1
(800) 441-2447T1: COILTRONICS CTX-02-13949-X1
: FERRONICS FERRITE BEAD 21-110J
Figure 140. A 10W, Low Noise, 5V to 12V Converter: Q1–Q2 Provide 5A Output Capacity while Preserving the LT1533’s Voltage/CurrentSlew Control. Efficiency is 68%. Higher Input Voltages Minimize Follower Loss, Boosting Efficiency Above 71%
A = 20V/DIV
B = 5V/DIV(AC COUPLED)
C = 100V/DIV
10µs/DIV
Application Note 84
AN84-76
Notes:
1 Witt, Jeff. The LT1533 Heralds a New Class of Low Noise SwitchingRegulators. Linear Technology VII:3 (August 1997).
2 Williams, Jim. LTC Application Note 70: A Monolithic Switching Regulatorwith 100µV Output Noise. October 1997.
3 The term “cascode,” derived from “cascade to cathode,” is applied to aconfiguration that places active devices in series. The benefit may behigher breakdown voltage, decreased input capacitance, bandwidth
improvement or the like. Cascoding has been employed in op amps,power supplies, oscilloscopes and other areas to obtain performanceenhancement.
4 This circuit derives from a design by Jeff Witt of Linear Technology Corp.
5 Operating the slew loops from follower base current was suggested byBob Dobkin of Linear Technology Corp.
1000pF
1000pF
3pF
22µF35V ×2
0.1µF
0.1µF
220pF
1000pF
56pF
1000pF
1000pF
470pF
100pF
0.1µF
MBRS1100T3
390k220k
10kΩ
10Ω
0.033Ω
100Ω
10Ω10Ω
100Ω
1MΩ
100k
47k
Q12N2907A
0.033Ω
T115µH
ITH1
COSC
SGND
SFB1
ITH2
VOSENS2
SENSE–2
SENSE+2
PGND
KEYBOARDCONTROLLER
SIGNAL
VIN 5.2-28V; SWITCHING FREQUENCY = 200kHz5V-3A / 3.3V-3.5A / 12V-120mAM1-M4 = Si4412DYT1 = DALE LPE-6562-A092; 15µH; 1:2.2
BG2
EXT VCC
SW2
TGL2
BOOST2
AUXON
AUXFB
23
22
21
20
19
18
LTC1538CG-AUX
BOOST1
RUN/SS1
SENSE+1
SENSE–1
M1
VPRGM1
TGL1
SW1
VIN
BG1
INT VCC
28
27
26
25
24
17
16
6
7
8
9
10
11
12
13
1
2
3
4
5
14RUN/SS2 AUXDR
15
0.1µF
CMDSH-3
4.7µF, 16V
VIN5.2-28V
VOUT15V/3A
GND
VOUT23.3V/3.5A
5V STANDBYAUX ON/OFF
AUX 12V OUT
L110µH
CMDSH-3
MBRS-140T3
MBRS140T3
+
100µF10V ×2
+
3.3µF35V
+
4.7µF25V
+
100µF10V×2
+
22µF35V×2
+
+
M2
M4
M3
24V
KEYBOARDCONTROLLER
SIGNAL
220pF
0.1µF
10kΩ
L1 = SUMIDA CDRH125-100MC 10µHINPUT AND OUTPUT CAPACITORS ARE AVX-TPS SERIES HAVING A MAXIMUM ESR SPECIFICATION
CSS1
CSS2
Regulators—Switching (Multioutput)
Figure 142. LTC1538-AUX Provides 3.3V/3.5A, 5V/3A, 12V/120mA and 5V/20mA Standby Power
Application Note 84
AN84-77
LTC1538-AUX: A NEW ADDITION TO LTC’SADAPTIVE POWER CONTROLLER FAMILYby Steve Hobrecht
Notebook Computer Power Solution
The circuit shown in Figure 142 is a power solution for aportable notebook computer. The switching controllersprovide 5V at 3A, 3.3V at 3.5A and a regulated 12V/120mAoutput using the auxiliary regulator. See the LTC1538-AUX/ LTC1539 data sheet for techniques illustrating howto generate other voltage and current combinations usingthe auxiliary regulator. The circuit provides a standby 5Voutput to power a keyboard controller. The keyboardcontroller has the ability to control the run/soft-start
(RUN/SS1 and RUN/SS2) pins of the LTC1538-AUX usingsimple logic gates. The turn-on sequence is determined bythe ratio of Css1 to Css2. The secondary winding oftransformer T1 develops a somewhat unregulated voltagedue to the loading on VOUT1. The SFB1 control pin will keepthe minimum voltage of the secondary output at approxi-mately 13V, but the peak voltage is affected by the loadingand leakage inductance of the transformer. The auxiliaryregulator will keep the 12V supply well within its normal±5% specified tolerance. Short-circuit protection can beadded to this circuit if required, but it is assumed here thatthe protection will only be required at the user PCMCIAinterface and will therefore be taken care of as part of theinterface and not duplicated here.
HIGH EFFICIENCY, LOW POWER,3-OUTPUT DC/DC CONVERTERby John Seago
The recent proliferation of battery powered products hascreated a lot of interest in low power, high efficiency DC/DC converter designs. These products are small, light-weight and portable, so space for bulky batteries is limited.Often, operating time between charges is a major sellingfeature, making the efficient use of battery power veryimportant. Since many products cannot function with asingle regulated voltage, multiple-output DC/DC convert-ers are required.
Although developed for somewhat higher power levels,the single output LTC1435 can be used in applicationsrequiring a very efficient, very small, low power, multiple-output DC/DC converter (see Figure 143). This isaccomplished through the use of an overwound buckinductor. With additional windings, the inductor can pro-vide additional outputs, requiring only a diode and filtercapacitor for each output. As with the less efficient flybacktopology, the additional outputs are not as well regulatedas the primary output, but the regulation is suitable formost applications.
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
TG
BOOST
SW
VIN
INT VCC
BG
PGND
EXT VCC
U1LTC1435CS
C60.001µF
C3330pFC4, 47pF
C2, 0.1µF
C5100pF
C168pF
6V TO 20V
R1, 10k
R3, 100Ω
C80.1µF
C722µF35V
Q11/2 Si9936
C11220pF
C12100µF10V
C13100µF10V
C14100µF10V
–5V/0.05A
3.3V/0.5A
5V/0.1A
GND
D4MBR0540
D3MBR0540
T1
T1
R50.1Ω
R635.7k
R720k
Q21/2 Si9936
D1, MBR0530
D2MBRS130L
R410Ω
C9, 0.1µF
C10, 4.7µF
R2, 100Ω
COSC
RUN/SS
ITH
SFB
SGND
VOSENSE
SENSE–
SENSE+
DI1435_01.eps
R5 = IRC, LR2010-01-R100-JC7 = AVX, TPSE226M035R0300
C12, 13, 14 = AVX, TPSD107M010R0100
T1 = COILTRINICS, CTX02-13299Q1/Q2 = SILICONIX, Si9936DY
30µH
+
+
+
+
+
Figure 143. High Efficiency, 3-Output DC/DC Converter
Application Note 84
AN84-78
DUAL-OUTPUT VOLTAGE REGULATORby Peter Guan
The LTC1266-3.3 and LTC1263, as shown in the sche-matic of Figure 144, are perfect complements for oneanother. The combination of these two parts provides tworegulated outputs of 3.3V/5A and 12V/60mA from an inputrange of 4.75V to 5.5V. These two outputs are perfect fornotebook and palmtop computers with microprocessorsthat burn several amps of current from a regulated 3.3Vsupply, flash memories that consume milliamps of currentfrom a regulated 12V supply and interface and logiccomponents that still run off the 5V supply. In fact, thisquick and easy combination may well be the aspirin formany of the headaches caused by the rigorous powersupply demands in today’s electronics.
The LTC1263, using only four external components (two0.47µF charge capacitors, one 10µF bypass capacitor anda 10µF output capacitor), generates the regulated 12V/60mA output from a 5V input using a charge pump tripler.During every period of the 300kHz oscillator, the twocharge capacitors are first charged to VCC and then stackedin series, with the bottom plate of the bottom capacitorshorted to VCC and the top plate of the top capacitorconnected to the output capacitor. As a result, the outputcapacitor is slowly charged up from 5V to 12V. The 12Voutput is regulated by a gated oscillator scheme that turns
The circuit of Figure 143 provides 3.3V at 0.5A, 5V at 0.1Aand –5V at 0.05A, and has greater than 93% efficiency fortest loads between 1.25W and 2.4W with a 6V input. Loadand line regulation of the positive outputs are quite good.Each output voltage was measured with all output currentsvaried independently between 20% and 100% of their fullload range, while the input voltage was varied from 6V to20V. Table 3 shows the worst-case output voltagesmeasured.
Table 3. Worst-Case Output VoltagesOutput Minimum Maximum
3.3V 3.307V 3.315V
5V 5.03V 5.24V
–5V –4.98V –5.51V
The buck regulator with an overwound inductor is a goodsolution for those applications that do not have large loadcurrent or line voltage variations. The smaller the load andline variations, the smaller the voltage variations on theoverwound outputs. As a general rule, output voltageregulation is suitable for most applications if the switchduty cycle is kept between 15% and 50% and minimumload current is kept above 20% of maximum. Since loadvariation and line variation have an additive effect onoutput voltage, applications with relatively constant loadcurrent requirements can have a larger input voltage rangeand vice versa. For zero output current requirements, asmall preload resistor can be used.
the charge pump on when VOUT is below 12V and turns itoff when it exceeds 12V.
The LTC1266-3.3 then uses the 5V input along with the12V output from the LTC1263 and various external com-ponents, including bypass capacitors, sense resistors andSchottky diodes, to switch two external N-channel MOS-FETs and a 5µH inductor to charge and regulate the 3.3V/5A output. The charging scheme for this part, however, isvery different from that of the LTC1263. The LTC1266-3.3first charges the output capacitor by turning on the top N-channel MOSFET, allowing current to flow from the 5Vinput supply and through the inductor. By monitoring theamount of current flow in the inductor with a senseresistor, the 3.3V output is regulated by turning on and offthe top and bottom N-channel MOSFETs to charge anddischarge the output capacitor.
If we replaced the top external N-channel MOSFET with aP-channel, the LTC1266-3.3 could generate the same3.3V/5A output without the help of the LTC1263. But, sinceN-channel MOSFETs have lower gate capacitance andlower RDS(ON), their higher efficiency at high currentsmore than compensates for the extra complexity in bring-ing in another higher input voltage, especially if thatsecond input voltage is readily available.
Application Note 84
AN84-79
DI1263_01.eps
TDRIVE
PWR VIN
PINV
BINH
*COILTRONICS CTX0212801
LTC1266-3.3
1000pF
VIN
CT
ITH
SENSE–
BDRIVE
PGND
LBOUT
LBIN
SGND
SHDN
NC
SENSE+
1
2
3
4
5
6
7
8
CT180pF
CC3300pF
1µF
C2 = 0.47µF
C3 = 10µF
C4 = 10µF
C1 = 0.47µF
RSENSE0.02Ω
RC470Ω
VOUT = 3.3V/5A
VOUT = 12V/60mA
VCC
16Si9410DY
D1MBRS140T3
CIN100µF20V× 2
COUT220µF10V× 2
L*5µH
Si9410DY
15
14
13
12
11
10
9
C1–
C1+
C2–
C2+
LTC1263
VCC5V
FROM µP
GND
VOUT
VCC
1
2
3
4
8
7
6
5
SHDN
+
+
Since both of these devices are very stingy on quiescentcurrent, their combination is also very gentle to the mainpower supply, especially if that power supply is a battery.In standby mode, the LTC1263 and the LTC1266-3.3 havea total quiescent current of about 500µA. To conserve evenmore current, both of these parts can be put into shutdownmode by floating their shutdown pins or pulling them high.The total shutdown current is less than 40µA. Whenloaded, the LTC1263 has a 76% efficiency, whereas the
LTC1266-3.3 can squeeze out more than 90%. Together,with a 60mA load at the 12V output and a 5A load at the3.3V output, the overall efficiency is 87%.
The LTC1266-3.3 is available in the 16-pin SO package andthe LTC1263 is available in the 8-pin SO package. Together,these two parts provide an easy and efficient solution formultiple power supply demands.
Figure 144. 5V to 3.3V/5A and 12V/60mA Supply
Application Note 84
AN84-80
SWITCHER GENERATES TWO BIAS VOLTAGESWITHOUT TRANSFORMERby Jeff Witt
LCD displays and CCD imaging circuits in today’s portableproducts require several bias voltages of 10V to 20V at afew mA. When symmetric bipolar bias supplies are needed,the negative supply can be generated with a discretecharge pump operating from the power switch of theboost regulator that generates the positive supply. How-ever, an asymmetric bipolar supply is typically required:for example 20V and –10V for LCD displays or 15V and–7.5V for CCDs. One possible solution is to add a linearregulator to the negative output; this adds cost and greatlyreduces the efficiency of the switcher. Another possibilityis a 2-output flyback circuit, but the added cost and bulkof a transformer make this solution unappealing. Thecircuit in Figure 145 avoids these penalties, producing 20Vat 5mA and –10V at 5mA from 3.3V with 73% efficiency.The circuit uses standard surface mount parts.
The LT1316, a micropower Burst Mode switching regula-tor with an integrated 0.6A power switch, operates in anordinary boost circuit to generate the 20V (VOUT1) set byresistor divider R1 and R2. An internal comparator at the
Figure 146. Voltage Waveforms ofFigure 145’s circuit
VIN3.3V
C133µF10V
R510k
L147µF
C21µF35V
C43.3µF 35V
C33.3µF 35V
R4590k
Q12N7002
82k
R31M
150k
R264.9k
150pF
R11M
BAT54
BAT54
BAT54
BAT54
LT1316FB
LBO
LBI
8
1
2
RSET GND
6 5
43
7SHDN
VIN
VOUT120V/5mA
VOUT2–10V/5mA
L1 = COILCRAFT DO1608C-473C1 = AVX TAJB336M035RC2 = AVX TAJA105M035RC3, C4 = AVX TAJB335M035R
SW
+
+
+
+
FB pin regulates the output by gating the LT1316’s oscil-lator. A charge pump (C2 and associated diodes) coupledto the LT1316’s switch pin generates the negative outputvoltage. This negative output (VOUT2) is monitored by theLT1316’s low-battery detector through the resistor dividerR3 and R4, using the positive 20V output as a reference.When the negative output falls below 10V, the low-batterydetector output (LBO pin and lowest trace in Figure 146)turns Q1 on, enabling the charge pump and chargingoutput capacitor C4. Note that the switch pin jumpsbetween ground and ~10V during this period. Once thenegative output has been charged enough to overcome thelow-battery detector’s hysteresis, Q1 turns off and theswitch pin is free to fly to 20V, charging the positiveoutput.
This circuit can also operate directly from two alkaline orNiCd cells. Slightly higher peak currents are necessary;change R5, which determines the peak switch current ofthe LT1316, to 6.8kW and change L1 to 15mH.
Figure 145. By Gating the Charge Pump, this CircuitGenerates a Regulated Negative Output with a MagnitudeDifferent from that of the Positive Output
VOUT1 200mV/DIV(AC COUPLED)
SW PIN 20V/DIV
VOUT2 1V/DIV(AC COUPLED)
LBO PIN 5V/DIV
0.1ms/DIV
Application Note 84
AN84-81
NEW IC FEATURES REDUCE EMIFROM SWITCHING REGULATOR CIRCUITSby John Seago
One disadvantage of using a switching regulator is that itgenerates electronic noise, known as EMI (electromag-netic interference). This noise can be conducted or radi-ated, and it can affect other circuits in your product orinterfere with the operation of nearby products. TheLTC1436-PLL, LTC1437, LTC1439 and LTC1539 havefeatures that can be used to suppress this interference.
Frequently, EMI problems don’t show up until the integra-tion phase of product development. By using this EMIsuppression capability, a resistor or capacitor value changemay be all that is required to solve an interference prob-lem. The LTC1436-PLL shown in the circuit of Figure 147produces a switched 5V, 3A output and a 3.3V, 0.1A linearoutput. The circuit is configured to provide either switch-frequency synchronization or switch-frequency modula-
tion. Also, transistor Q3 ensures constant frequency atvery low output current levels, thus eliminating audiofrequencies and maintaining high efficiency using theinternal Adaptive Power™ circuitry.
Switch-Frequency Synchronization
Switching regulator noise results from switching highcurrents on and off. This creates high energy levels at theswitching frequency and all of its harmonics. A commonEMI-control technique is to synchronize the switchingfrequency to an external clock so that all harmonic fre-quencies can be controlled. The LTC1436-PLL uses aphase-locked loop for synchronization to avoid the loss ofslope compensation common to other synchronizing tech-niques. In addition, the input to the VCO in the phase-locked loop is available at the PLL LPF (phase-locked looplowpass filter) pin, so that a lowpass filter can be used tocontrol how fast the loop acquires lock.
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
PLL LPF
COSC
RUN/SS
ITH
SFB
SGND
VPROG
VOSENSE
SENSE–
SENSE+
AUXON
AUXFB
PLLIN
POR
BOOST
TGL
SW
TGS
VIN
INTVCC
BG
PGND
EXTVCC
AUXDR
DI1436_01.eps
5V3A
POR
3.3V0.1A
GND
Q3
LTC1436-PLL
R3, 10k
R647k
5.5V TO24V R7
10Ω
R4, 100Ω
R2, 10k
MOD
MOD
PLL
PLL0.01µF
C3, 0.1µF
C80.1µF
R1147k
R1035.7k
C153.3µF
C11, C12: KEMET T495X226M035ASC13, C14: AVX TPSD107M010R0065L1: SUMIDA CDRH125-10Q1 + Q2: SILICONIX Si4936DY (DUAL FET)Q3: INTERNATIONAL RECTIFIER IRLML2803R8: IRC LR2010-01-R033-J
* SEE FIGURE 151
C94.7µF
C100.1µF
Q2
Q1
D1MBRS0530 D2
MBRS130L
C4330pF
C2, 47pF
C5, 47pF
C7, 0.001µF
R5100Ω
INTVCCC6
100pFC13, C14100µF10V×2
C1222µF35V
L110µH
R920k
R80.033Ω
C1122µF35V
Q4MMBT2907ALT1
C1
SWITCH-FREQUENCY
MODULATOR*
+ +
+
+
+
Figure 147. 2-Output LTC1436-PLL Test Circuit
Application Note 84
AN84-82
–120dBm
–100dBm
–80dBm
–60dBm
–40dBm
–20dBm
1MHzDI1436_02.eps
1kHz 500kHz
VIN = 10VVO = 5V AT 3A
BW = 100Hz
–105dBm
–95dBm
–85dBm
–75dBm
–65dBm
–55dBm
30MHzDI1436_03.eps
1MHz 15MHz
VIN = 10VVO = 5V AT 3A
BW = 300Hz
Switch-Frequency Modulation
Access to the VCO input also makes it possible to modu-late the regulator’s switching frequency. Through fre-quency modulation, the peak energy of the fundamental isspread over the frequency range of modulation, thusdecreasing the peak energy level at any one frequency.This frequency spreading action increases with each har-
monic, so that the second harmonic has twice the band-width and the third harmonic has three times the band-width until all the harmonics blend together, decreasingthe signal strength at all frequencies. This can be seen inthe spectrum analyzer plots shown in Figures 148–150.
Figure 148. Output Noise Before and After Switch-Frequency Modulation
Figure 149. Output High Frequency Noise Before Switch-Frequency Modulation
Application Note 84
AN84-83
Figure 148 shows the full load output noise level from thecircuit of Figure 147, before and after switch-frequencymodulation. The black trace shows the normal outputnoise from 1kHz to 1MHz with the VCO at minimumfrequency, whereas the colored trace shows output noiseafter modulation around the center frequency. The 228kHzunmodulated switch-frequency output noise decreasedmore than 30dB through modulation between 270kHz and370kHz. Figures 149 and 150 show a 10dB to 15dBattenuation in full-load output voltage noise from 1MHz to30MHz after modulation.
–105dBm
–95dBm
–85dBm
–75dBm
–65dBm
–55dBm
30MHzDI1436_04.eps
1MHz 15MHz
VIN = 10VVO = 5V AT 3A
BW = 300Hz
The VCO in the LTC1436-PLL has an input range from 0Vto 2.4V. As shown in Figure 151, the switch frequency canbe modulated at least ±30% around the center frequencyfO. The ideal modulating signal varies an equal amountabove and below the center frequency voltage of 1.2V,with a constant slope. The reference circuit of Figure 152develops a 100Hz sawtooth voltage from 0.9V to 1.5V thatmodulates the LTC1436-PLL in Figure 147 to generate theplots shown in Figures 148–150. Modulator circuit com-plexity is largely determined by functional requirements.For most applications, a precision modulating signal is notrequired, because high order harmonics blend together.Consequently, modulating frequency, slope and peak-to-peak voltage are not critical.
VPLL LPF (V)
0.7f0
f0
1.3f0
FREQ
UENC
Y
2.5
DI1436_05.eps
0 2.01.51.00.5
COSC = 47pF
COSC = 100pF
–
+
DI1436_06.eps
+VCC
DISCH
THRESH
CONT
GND
TRIG
OUT
RESET
1
2
3
4
0.1µF
TLC555
LTC1436-PLLPIN 17
(5V)
LTC1436-PLLPIN 6
(GND)
1.5V
0.9V~10ms
1.2M
220Ω
0.1µF
2N3904
100k
150k
510k
LT1077
LTC1436-PLLPIN 1(MOD)
8
7
6
5
0V
Figure 150. Output High Frequency Noise After Switch-Frequency Modulation
Figure 151. Operating Frequency vs VPLL LPF Figure 152. Switch-Frequency Modulator
Application Note 84
AN84-84
Audio Frequency Suppression
The Adaptive Power feature of the LTC1436-PLLsignificantly reduces audio frequency generation, whilemaintaining good efficiency under very light load condi-tions. Figure 153 shows the audio frequencies generatedby the highly efficient cycle skipping mode of the LTC1436-PLL. Figure 154 shows the decrease in audio frequenciesresulting from Adaptive Power operation. Figure 155shows efficiency curves of both the cycle skipping andAdaptive Power modes along with the traditional, forcedcontinuous mode of operation.
Cycle skipping is the most efficient mode during light-loadoperation, where the output capacitor supplies load cur-rent most of the time and is replenished by bursts ofenergy at a rate determined by the load. When load currentis low enough, the burst rate falls into the audio-frequencyrange, which can cause problems. With the addition of Q3,an inexpensive SOT-23 size MOSFET, the Adaptive Powercircuitry inside the LTC1436-PLL takes control duringlight load conditions, turning off high current MOSFETsQ1 and Q2. Q3 and D2 are then used in a conventionalconstant frequency buck mode, eliminating the powerloss caused by charging and discharging the large inputcapacitance of both power MOSFETs.
The conventional way of avoiding audio-frequency inter-ference is the forced current mode, where both highcurrent MOSFETs continue to operate at full frequency andnormal duty cycle under all load conditions. This causes
10Hz 10kHz
–20dBm
–40dBm
–60dBm
–80dBm
–100dBm
DI1436_07.eps
–120dBm20kHz
10VIN5VOUT AT 3mABW = 100Hz
10Hz 10kHz
–20dBm
DI1436_08.eps
–120dBm
–100dBm
–80dBm
–60dBm
–40dBm
20kHz
10VIN5VOUT AT 3mABW = 100Hz
OUTPUT CURRENT
50
100
90
80
70
60
EFFI
CIEN
CY (%
)
10A1mA 10mA 1A100mA
(3)
(2)
(1)
10V IN5V OUT
CYCLE SKIPPING OPERATION: VARIABLE FREQUENCY COMPONENTS AT LOWER OUTPUT CURRENTS
Adaptive Power OPERATION: CONSTANT FREQUENCY WITH AUTOMATIC SWITCHOVER TO SMALL MOSFET Q3
FORCED CONTINUOUS OPERATION: CONSTANT FREQUENCY USINGLARGE MOSFETS Q1 AND Q2
1.
2.
3.
the peak-to-peak inductor current to flow, even under noload conditions. The synchronous buck topology allowsthe top switch, Q1, to put current into the output capacitor,followed by the bottom switch, Q2, taking current out ofthe output capacitor while regulating the output voltageunder no-load conditions. Although constant frequency ismaintained, high current I2R losses and high gate chargelosses continue under light load conditions. Forced-cur-rent operation is useful for fast transient response requiredfor high di/dt loads like the Intel Pentium® processor.
Cycle skipping, Adaptive Power and forced current opera-tion are all available on the LTC1436-PLL, so that the bestoperating mode can be selected for each application.
Figure 153. Audio Frequencies in Output Noise duringCycle-Skipping Operation
Figure 154. Output Noise with Adaptive Power Operation
Figure 155. Efficiency Curves for Light Load Currents
Application Note 84
AN84-85
Regulators—Switching (Micropower)
POWER MANAGEMENT AND HIGH EFFICIENCYSWITCHER MAXIMIZE NINE-VOLT BATTERY LIFEby LTC Applications Staff
The LTC1174 (3.3V, 5V and adjustable versions) canconvert a 9V battery source to system power with veryhigh efficiency. Efficiency is over 90% at load currentsfrom 20mA to 425mA and over 85% at a load current of4mA. For a given load, maximum battery life can beobtained by minimizing shutdown current during systemshutdown and maximizing converter efficiency duringoperation. A single control line to the LTC1174 can be usedto select shutdown mode or operational mode, as required.
100k
D S Q
R
9V
9V
9V100k
R1200k
C10.1µF
100k
TOCONTROLLER
9V
ANY FRONT PANEL SWITCH
1/2 CD4012
1/2 CD4013
0.22µF22µF*9V+
100k
9V
100k
2N2222 FROM OPENCOLLECTOR OUTPUTOF CONTROLLER1 = ON, 0 = OFF
5V
100k
9V
TO PIN 1OF LTC1174
0.0068µF
5V
93.1k
30.9k
FOR MINIMUM RF NOISEUSE LTC1174 - ADJUSTABLEWITH ABOVE NETWORK
VINLBIN3 6 7
5
1
84
IPGM
GND SDVOUT
SW
D11N5818
L1**50µH
LTC1174-5 0.1µF 100µF*
5V TO CONTROLS,ETC.
* AVX TPS** COILTRONICS CTX50-4
(561) 241-7876
+
For this circuit (Figure 156), power-up is initiated by a lowlevel signal on the NAND gate. This signal could come fromany front-panel switch or from an external interrupt signal.The system power is turned off by means of a low levelsignal from a controller/logic device. In either case, thecontrol signal to the LTC1174 must be latched. (A latchedturn-off signal ensures a known state on the LTC1174shutdown pin during the collapse of the 5V supply.)
The CD4012 and CD4013 are powered from the battery;the 2N2222 provides simple level shifting to the batteryrail. R1 and C1 ensure that the circuit remains in power-down mode during battery replacement. The circuit shownhere provides approximately 90% efficiency at 250mAload current, and consumes less than 1µA shutdowncurrent. Turn-on and turn-off transitions are very clean.
RUN RUN
5 SEC
TIME
STANDBY* STANDBY0.250A
0
*STANDBY TIME IS LONGIAVG < 5mA
Figure 156. Schematic Diagram of High Efficiency DC/DC Converter
Figure 157. Load Profile
Application Note 84
AN84-86
LT1307 MICROPOWER DC/DC CONVERTERELIMINATES ELECTROLYTIC CAPACITORSby Steve Pietkiewicz
The relentless push towards increasing miniaturization inportable electronic products has created the need forsmall, high speed, low voltage DC/DC converter ICs. TheLT1307 combines a current-mode, fixed frequency PWMarchitecture with Burst Mode™ micropower operation tomaintain high efficiency at light loads. It uses small, lowcost ceramic capacitors for both input and output, mini-mizing board area. By employing fixed frequency 575kHzswitching the LT1307 keeps spectral energy out of the455kHz band. Dense, high speed bipolar process technol-ogy enables the LT1307 to fit in the MSOP package, andmicropower circuitry results in just 60µA quiescent cur-rent at no load. Conversion efficiency exceeds 80%, andthe device also includes a low battery detector.
Single-Cell Boost Converter
A complete single-cell to 3.3V converter is shown in Figure158. The circuit generates 3.3V at up to 75mA from a 1.0Vinput. The 10µF ceramic output capacitor can be obtainedfrom several vendors. Efficiency, detailed in Figure 159,peaks at 80% and exceeds 70% over the 1:500 load rangeof 200µA to 100mA at a 1.25V input. Changing the valueof R1 to 1.87MΩ moves the output to 5V. Efficiency of the5V output converter is depicted in Figure 160. Figure 161’s
oscillograph shows output voltage and inductor current asthe load current is stepped from 5mA to 55mA, revealingsubstantial detail about the operation of the LT1307. Witha 5mA load, VOUT (top trace) exhibits a ripple voltage of60mV at 4kHz. The device is in Burst Mode at this outputcurrent level. Burst Mode operation enables the converterto maintain high efficiency at light loads by turning off allcircuitry inside the LT1307 except the reference and erroramplifier. When the LT1307 is not switching, quiescentcurrent decreases to 60µA. When switching, inductorcurrent (middle trace) is limited to approximately 100mA.Switching frequency inside the “bursts” is 575kHz. As theload is stepped to 55mA, the device shifts from BurstMode to constant switching mode. Inductor current in-creases to about 300mA peak and the low frequency BurstMode ripple goes away. R3 and C3 stabilize the loop.
VIN SWFB
LT1307
L110µH D1
LBO
LBI
SHDNSHUTDOWN
R3100k
R2604k1%
3.3V75mA
R11.02M
1%
C3680pF
C1 = MURATA-ERIE GRM235Y5V105Z01MARCON THCS50E1E105ZTOKIN 1E105ZY5U-C103-F
C2 = MURATA-ERIE GRM235Y5V106Z01MARCON THCS50E1E105ZTOKIN 1E106ZY5U-C304-F
C11µF
C210µF
1.5VCELL VC GND
D1 = MOTOROLA MBR0520LL1 = SUMIDA CD43-100
LOAD CURRENT (mA)
50
60
70
80
90
EFFI
CIEN
CY (%
)
3000.1 1 10 100
VIN = 1.00V
VIN = 1.25V
VIN = 1.5V
LOAD CURRENT (mA)
50
60
70
80
90
EFFI
CIEN
CY (%
)
2000.1 1 10 100
VIN = 1.00V
VIN = 1.25VVIN = 1.5V
Figure 158. Single Cell to 3.3V Boost Converter Delivers 75mAat 1.0V Input. Changing R1 to 1.87M Moves the Output to 5V
Figure 159. 3.3V Efficiency
Figure 160. Efficiency at 5V Output
Application Note 84
AN84-87
DC/DC Converter Noise Considerations
Switching regulator noise is a significant concern in manycommunications systems. The LT1307 is designed tokeep noise energy out of the 455kHz band at all load levelswhile consuming only 60µW–100µW at no load. At lightload levels, the device is in Burst Mode, causing lowfrequency ripple to appear at the output. Figure 162 detailsspectral noise directly at the output of Figure 158’s circuitin a 1kHz to 1MHz bandwidth. The converter supplies a5mA load from a 1.25V input. The Burst Mode fundamen-tal at 5.1kHz and its harmonics are quite evident, as is the575kHz switching frequency. Note, however, the absenceof significant energy at 455kHz. Figure 163’s plot reducesthe frequency span from 255kHz to 655kHz with a 455kHzcenter. Burst Mode low frequency ripple creates side-bands around the 575kHz switching fundamental. Thesesidebands have low signal amplitude at 455kHz, measur-ing –55dBmVRMS. As load current is further reduced, theBurst Mode frequency decreases. This spaces the side-bands around the switching frequency closer together,moving spectral energy further away from 455kHz. Figure164 shows the noise spectrum of the converter with the
load increased to 20mA. The LT1307 shifts out of BurstMode, eliminating low frequency ripple. Spectral energy ispresent only at the switching fundamental and its harmon-ics. Noise voltage measures –5dBmVRMS or 560µVRMS atthe 575kHz switching frequency, and is below –60dBmVRMSfor all other frequencies in the range. By combining BurstMode with fixed frequency operation, the LT1307 keepsnoise away from 455kHz, making the device ideal for RFapplications where the absence of noise in the this band iscritical.
Output filtering can reduce output conducted noise. Figure158’s circuit, supplying a 50mA load at 3.3V from a 1.3Vsource, is shown with an output filter (R4 and C4) in Figure165. The lowpass filter created by R4 and C4 places a poleat 34kHz, reducing high frequency spikes considerably.
FREQUENCY (kHz)1
OUTP
UT N
OISE
VOL
TAGE
(dBm
V RM
S)
40
30
20
10
0
–10
–20
–30
–40
–50
–6010 100 1000
RBW = 100Hz
FREQUENCY (kHz)255
OUTP
UT N
OISE
VOL
TAGE
(dBm
V RM
S)
–20
–25
–30
–35
–40
–45
–50
–55
–60
–65
–70455 655
RBW = 100Hz
FREQUENCY (kHz)255
OUTP
UT N
OISE
VOL
TAGE
(dBm
V RM
S)
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100455 655
RBW = 100Hz
Figure 161. Transient Response with 5mA to 55mA Load Step
Figure 162. Spectral Noise Plot of 3.3V Converter Delivering5mA Load; Burst Mode Fundamental at 5.1kHz is 23dBmVRMSor 14mVRMS
Figure 163. Span Centered at 455kHz Shows –55dBmVRMS(1.8µVRMS) at 455kHz. Burst Mode Creates Sidebands 5.1kHzApart around the Switching Frequency Fundamental of 575kHz
Figure 164. With the Converter Delivering 20mA,Low Frequency Sidebands Disappear. Noise isPresent Only at the 575kHz Switching Frequency
VOUT200mV/DIV
AC COUPLED
IL200mA/DIV
ILOAD
500µs/DIVVIN = 1.25V
55mA5mA
Application Note 84
AN84-88
Viewed in a 50MHz bandwidth, the filter reduces switchingspikes from about 10mVP–P to about 1mVP–P, as detailedin Figure 166. Beware, though; the oscilloscope used inFigure 166’s oscillograph (a Tektronix Type 547) is help-ing with the filtering by attenuating frequencies above50MHz. Figure 167 shows the same circuit viewed on a400MHz oscilloscope. The filter still attenuates but themagnitude of switching noise is far higher (140mVP–Punattenuated). A small amount of copper trace can be usedin place of the resistor if the attendant voltage drop isunacceptable. A surprisingly small amount of trace isneeded to create an effective filter; a PC trace of 1 oz.copper, 1 inch long by 10 mils wide, has an inductance of29nH. Inductive reactance at 50MHz (2πfL) is 9.1Ω. Acombination of copper trace and 0.1µF ceramic capacitorswill reduce high frequency spikes to acceptable levels inmost systems.
LED Driver
LEDs require current source drive. Typically, a 5V supplywith a series resistor to limit current is used to power theLED. Although simple, this approach has poor efficiencyand requires a voltage source higher than the 2V–3Vforward drop of most LEDs. Additionally, each LED re-quires its own ballast resistor. Figure 168’s circuit uses theLT1307 configured as a current source to drive a series-connected pair of LEDs from a single-cell input. The IC’slow battery detector monitors the voltage across senseresistor R1. LBO drives Q1; this provides correct phasingto the VC pin. Q1 and R2 drive the VC pin, overriding theinternal error amplifier. With 200mV across R1, 25mAflows through the LED pair. C3 provides frequency com-pensation. For proper operation, the circuit must alwayssupply enough power so as to not enter Burst Modeoperation. This precludes driving most single LEDs (highbrightness blue LEDs have a forward drop of 3.4V and canbe driven singly). In shutdown mode, the circuit draws aonly few microamperes. Start-up sequencing is detailed inFigure 169. The voltage at LBI stabilizes in about 200µswith minimal overshoot and ringing. The Lumex “Mega-Brite” red LEDs specified in Figure 168 provide enoughlight to act as a flashlight, providing young children with ahigh technology toy. Mounted on a small PC board with apush-button switch, the circuit entertained my two chil-dren for hours. They are both satisfied LT1307 customers.
Figure 166. VOUT1 is Output Voltage at 10µF CapacitorC2; VOUT2 is After 4.7Ω/1µF Output Filter. CircuitSupplies 50mA; Oscilloscope Bandwidth is 50MHz.
Figure 167. A Faster Oscilloscope Shows More High FrequencyContent at Both Outputs. Scope Bandwidth is 400MHz.
VIN SW
VC GND
FB
VOUT1
VOUT2LT1307
C41µF
L110µH
R44.7Ω
R21.02M
1%
D1
R3604k1%R1
100k
C11µF
1.5VCELL
C210µF
C3680pF
Figure 165. Figure 158’s Circuit with Output Filter R4/C4
VOUT150mV/DIV
AC COUPLED
VOUT25mV/DIV
AC COUPLED
200ns/DIV
VIN = 1.3V
VOUT15mV/DIV
AC COUPLED
VOUT25mV/DIV
AC COUPLED
200ns/DIV
VIN = 1.3V
Application Note 84
AN84-89
Figure 169. Start-Up Response of LEDCircuit. Many Switching Cycles Elapsebefore Current Flows in LEDs Becauseof C1 Charging
VIN SW
SHDN GND
FB
LBI
LB0
VC
LT1307
C11µF
L110µH D1
VIN
NC
D2
D3
R18Ω
100k
ON/OFF
VIN
R222k
100k
C21µF
Q12N3906
AACELL
C322µF
L1 = MURATA-ERIE LQH3C100K04D1 = MOTOROLA MBR0520L
C1, C2 = CERAMIC
D2, D3 = LUMEX SSL-X100133SRC/4 "MEGA-BRITE" RED LEDOR PANASONIC LNG992CF9 HIGH BRIGHTNESS BLUE LED
+
AN ULTRALOW QUIESCENT CURRENT,5V BOOST REGULATORby Sam Nork
Many battery-powered applications require an auxiliary5V supply to power infrequently used circuitry, such assmart card readers, wireless i.d. tags, or the like. Keepingthe 5V supply permanently active is desirable, since thiseliminates timing delays and inrush currents due to supplystart-up. The downside is that most 5V boost convertersconsume an unacceptable amount of quiescent currentunder no-load conditions. This problem is addressed bythe SHDN features of the LTC1516 micropower, charge-pump DC/DC converter. Toggling the SHDN pin of theLTC1516 allows the 5V supply to remain in regulation witha typical no-load input current of less than 5µA. When the5V output load is enabled, the part can supply up to 50mAof load current.
The LTC1516 produces a regulated 5V output from a 2V to5V input In shutdown mode, the output load is discon-nected from VIN and the quiescent current drops below1µA.
When the output is in regulation, the internal sense resistordraws only 1.5µA (typical) from VOUT. During no-loadconditions, this internal load causes a droop rate of only150mV per second on VOUT with COUT = 10µF. Applying a5Hz–100Hz, 95%–98% duty-cycle signal to the SHDN pinensures that the circuit in Figure 170 comes out of shut-down frequently enough to maintain regulation during no-load (or low-load) conditions. Since the part is kept inshutdown mode for the majority of the time, the no-loadquiescent current (see Figure 171) is approximately equalto (VOUT × (1.5µA + ILOAD))/(VIN × efficiency).
1
2
3
4
8
7
6
5
C1–
SHDN
GND
C2–
C1+
VIN
VOUT
C2+
LTC1516
0.22µF
FROM MPU
10µF
10µF
VOUT = 5V ±4%
SHDN PIN WAVEFORMS:
LOW IQ MODE (5Hz TO 100Hz, 95% TO 98% DUTY CYCLE)IOUT ≤ 100µA
VOUT LOAD ENABLE MODE(IOUT = 100µA TO 50mA)
0.22µF
VIN = 2V TO 5V+
+
Figure 168. Single-Cell LED Driver Supplies 25mA to LED String.Two Red LEDs Can Be Replaced by One Blue LED
Figure 170. Ultralow Quiescent Current (<5µA) Regulated Supply
VLBI100mV/DIV
ISW100mA/DIV
ONOFF
Application Note 84
AN84-90
The LTC1516 must be taken out of shutdown mode for aminimum of 200µs to allow the internal sense circuitry tostart up and keep the output in regulation. As the VOUT loadcurrent increases, the frequency with which the part istaken out of shutdown must also be increased to preventVOUT from drooping below 4.8V during the OFF phase (see
VIN (V)
0.0
2.0
4.0
6.0
I CC
(µA)
5.02.0 3.0 4.0IOUT (µA)
1
10
100
1000
MAX
SHD
N OF
F TI
ME
(ms)
10001 10 100
SHDN ON PULSE WIDTH = 200µsCOUT = 10µF
Figure 172). A 100Hz, 98% duty cycle signal on the SHDNpin ensures proper regulation with load currents as highas 100µA. When load current greater than 100µA isneeded, the SHDN pin must be forced low, as in normaloperation. The typical no-load supply current for thiscircuit with VIN = 3V is only 3.2µA.
CAPACITIVE CHARGE PUMPPOWERS 12V VPP FROM 5V SOURCEby Mitchell Lee
The LTC1263, a regulating charge pump tripler, convertsa 5V input to a regulated 12V, 60mA output. No inductorsare required; charge pumps operate with capacitors only.Figure 173 shows the LTC1263 configured to provide VPP
for two flash memory chips. The “flying” capacitors in thecharge pump, C1 and C2, are sized well within the surfacemount ceramic range. CIN and COUT, as shown, are surfacemount tantalum capacitors, such as Sprague 595D series.In the 10µF capacitance range, tantalum capacitors costless than ceramic units. The chip operates by charging C1and C2 in parallel across 5V and ground and then discharg-ing them in series across 5V and the output. In theory, theoutput could reach 15V, but an internal regulation loopmaintains the output at a constant 12V.
SHUTDOWN reduces the quiescent current of the LTC1263to less than 1µA under logic control. In shutdown mode,the output is held at 5V by an internal 500Ω, VCC-to-VOUTswitch. Output-voltage fall time is guaranteed to be lessthan 15ms for the component values shown. Output risetime coming out of shutdown is guaranteed to be less than800µs.
Designing a circuit to generate a split supply from a single5V source is usually an unpleasant chore; one to beavoided at all costs. If load current requirements aremodest, the LTC1263 can generate both 12V and –7V forop amps and biasing needs. Figure 174 shows how. The
SHDN
VOUT
VCC
VPP
FLASHMEMORY
C1+
C1–
C2+
C2–
LTC1263
CIN10µF
COUT10µF
4.75V TO 5.5V
OFF ONC1470nF
12V AT 60mAC2470nF
+
+
Figure 171. No-Load ICC vs Input Voltage for Figure 170’s Circuit Figure 172. Maximum SHDN OFF-Time vs Output Load Currentfor Ultalow IQ Operation
Figure 173. Programming Two Flash Chips with the LTC1263Charge Pump: In Shutdown Mode, the Output is Held at 5V
Application Note 84
AN84-91
LTC1263 is connected in the usual way to produce aregulated, 12V output, but a 2 diode, 2-capacitor chargepump is added to the C2+ pin. This pin switches betweenVCC and VOUT, swinging approximately 7VP–P. The resultis an outboard charge pump inverter with a –7V output.
SHDN
VOUT
C1+
C1–
C2+
C2–
LTC1263
10µF
10µF
4.75V TO 5.5V
470nF
12V OUTPUT
MBR0520L
–7V OUTPUT
470nF
1µF
10µF
VCC
MBR0520L
+
+
+
GND
–7V LOAD (mA)1
6
–7V
OUTP
UT (V
)
7
8
10 100
VCC = 5V12V LOAD = 3mA
COMMON LOAD CURRENT (mA)0
0
4OU
TPUT
(V)
8
16
12
4020 60 80
VCC = 5V
+12V
–7V
Schemes like this one often suffer from poor cross regu-lation. Although the inverting output is not directly regu-lated, the –7V load does affect the 12V output, therebyimproving cross regulation (see Figure 175). The regula-tion with a common load (such as op amps) is shown inFigure 176.
LTC1474 AND LTC1475 HIGH EFFICIENCY SWITCHINGREGULATORS DRAW ONLY 10µA SUPPLY CURRENTby Greg Dittmer
Introduction
Maximizing battery life, one of the key design require-ments for all battery-powered products, is now easier withLinear Technology’s new family of ultralow quiescentcurrent, high efficiency step-down regulator ICs, theLTC1474 and LTC1475. The LTC1474/LTC1475 are step-
down regulators with on-chip P-channel MOSFET powerswitches. These regulators draw only 10µA supply currentat no load while maintaining the output voltage. Widesupply voltage range (3V–18V) and 100% duty cyclecapability for low dropout allow maximum energy to beextracted from the battery, making the LTC1474/LTC1475ideal for moderate current (up to 300mA) battery-poweredapplications.
Figure 174. Split-Supply Generator: Cross Regulation isImproved by Driving the Inverting Charge Pump from C2+.
Figure 175. Cross Regulation with a Constant 12V Load
Figure 176. Output Regulation with a Common Load
Application Note 84
AN84-92
Other features include Burst Mode™ operation to maintainhigh efficiency over almost four decades of load current,an on-chip low-battery comparator and a shutdown modeto further reduce supply current to 6µA. The LTC1475provides on/off control with push-button switches for usein handheld products.
The LTC1474/LTC1475 are available in adjustable andfixed 3.3V/5V output voltage versions, in 8-pin MSOP andSO packages.
3.3V/200mA Step-Down Regulator
A typical application circuit using the LTC1474 is shown inFigure 177. This circuit supplies a 200mA load at 3.3V withan input supply range of 4V–18V (3.3V at no load). The0.1Ω sense resistor reduces the peak current to about285mA, which is the minimum level necessary to meet the200mA load current requirement with a 100µH inductor.The peak current can be reduced further if a higher valueinductor is used. Since the output capacitor dominates theoutput voltage ripple, an AVX TPS series low ESR (150mΩ)output capacitor is used to provide a good compromisebetween size and low ESR. With this capacitor the outputripple is less than 50mV.
Efficiency Considerations
The efficiency curves for the 3.3V/200mA regulator atvarious supply voltages are shown in Figure 178. Note theflatness of the curves over the upper three decades of loadcurrent and that the efficiency remains high down to
extremely light loads. Efficiency at light loads depends onlow quiescent current. The curves are flat because allsignificant sources of loss except for the 10µA standbycurrent—I2R losses in the switch, catch diode losses, gatecharge losses to turn on the switch and burst cycle DCsupply current losses—are identical during each burstcycle. The only variable is the rate at which the burst cyclesoccur. Since burst frequency is proportional to load, theloss as a percentage of load remains relatively constant.The efficiency drops off as the load decreases below about1mA because the non-load-dependent 10µA standby cur-rent loss then constitutes a more significant percentage ofthe output power. This loss is proportional to VIN and thusits effect is more pronounced at higher VIN.
LTC1475 Push-Button On/Off Operation
The LTC1475 provides the option of push-button controlof run and shutdown modes for handheld products. Incontrast to the LTC1474’s run/shutdown mode, which iscontrolled by a voltage level at the RUN pin (ground =shutdown, open/high = run), the LTC1475 run/shutdownmode is controlled by an internal S/R flip-flop that is set(run mode) by momentarily shorting the ON pin to groundand reset (shutdown mode) by a momentary ground at theLBI pin (see Figure 179). This provides simple on/offcontrol with two push-button switches. The simplestimplementation of this function is shown in Figure 180,with normally open push-button switches connected tothe ON and LBI pins. Note that because the switch on LBIis normally open, it doesn’t affect the normal operation ofthis input to the low-battery comparator. With a resistor
+10pF1.69M
1M
100kRUN
VIN
SENSE
SW
VFB
LBO0.1µF
10µF25V
1000pFLBI
LTC1474
LBI
LBO
MBR0530
GND
8
7
6
5
1
VOUT3.3V/200mA
VIN4V TO 18V
RUN
D1:L1:
COUT:CIN:
MBR0530SUMIDA CDRH74TPSC107006R0150THC50EIE106Z
2
3
4
0.1Ω
COUT100µF
6.3V
L1 100µH
LOAD CURRENT (mA)
50
60
70
80
90
100
EFFI
CIEN
CY (%
)
200
1474_04.eps
0.02 0.2 2 20
L = 100µHVOUT = 3.3V
RSENSE = 0.1Ω
VIN = 5VVIN = 10V
VIN = 15V
Figure 177. LTC1474 3.3V/200mA Step-Down RegulatorFigure 178. Efficiency vs LoadCurrent for Figure 177’s Circuit
Application Note 84
AN84-93
divider network connected to the LBI to monitor the inputsupply voltage level, the voltage at this pin will normally beabove the low-battery trip threshold of 1.23V. When thispin is pulled below 0.7V by depressing the switch, theinternal flip-flop is reset to invoke shutdown.
Figure 181 shows an example of push-button on/offcontrol of a LTC1475 microcontroller application with asingle push button. The push button is connected to themicrocontroller as a discrete input so that themicrocontroller can monitor the state of the push button.The LTC1475 LBI pin is connected to one of themicrocontroller’s open-drain discrete outputs so that it
LBI
ON
MODE RUN SHUTDOWN RUN
MODE RUN SHUTDOWN RUN
RUN
LTC1474
LTC1475
ON OVERRIDES SHUTDOWN WHILE RUN IS LOW
100k
10µF
1474_05.eps
VIN
SENSE
SW
OFF
VFB
VFB
VOUT
VFB
LBO
U1LTC1475
VBATT
VBATT
GND
8
7
6
5
1
2
3
4
ON
ON
LBI/OFF
1M 2.2M
100µH
100µF+
can force the LTC1475 off when it detects a depressedpush button. Because the LTC1475 supplies power to themicrocontroller, once the microcontroller is off, it can nolonger turn the LTC1475 back on. However, since the pushbutton is also connected directly to the ON pin, theLTC1475 can be turned back on directly from the pushbutton without the microcontroller. The LTC1475 thenpowers up the microcontroller. The discrete inputs ofmost microcontrollers have a reverse biased diode, D2,between the input and supply; thus a blocking diode withless than 1µA leakage is necessary to prevent the pow-ered down microcontroller from pulling down on the ONpin.
100k
100µF
100µH
10µF
1M 2.2M
1474_06.eps
VIN
SENSE
SW
VFB
VOUT
VFB
VCC
LBO
LTC1475
VBATT
VBATT
GND
8
7
6
5
1
2
3
4
ON/OFF
MMBD914LT1
µC
ON
LBI/OFF
VFB
+
Figure 179. Comparison of RUN/SHUTDOWNOperation for the LTC1474 and LTC1475
Figure 180. LTC1475 Step-Down Regulatorwith Push-Button On/Off Control
Figure 181. A Single Push-Button Controls On/Off for the LTC1475 Regulator and Microcontroller
Application Note 84
AN84-94
FREE DIGITAL PANEL METERS FROM THEOPPRESSIVE YOKE OF BATTERIESby Mitchell Lee
Digital panel meters (DPMs) have dropped in price to wellunder $10 for 3-1/2 digit models, even in single-piecequantities. These make excellent displays for many instru-ments, but suffer from one major flaw: they require afloating power supply, usually in the form of a 9V battery.This renders inexpensive meters useless for most applica-tions because no one wants multiple 9V batteries in theirproduct.
The circuit shown in Figure 182 powers up to five metersfrom a single 1.8V to 6V source. The source need not befloating, yet all five outputs are fully floating, isolated andindependent in every respect. The circuit consists of anLT1303 micropower, high efficiency DC/DC converterdriving a 5-output flyback converter. An off-the-shelfsurface mount coil, Coiltronics’ VERSA-PAC™ VP1-0190,is used as the transformer. This device is hipot tested to500VRMS—more than adequate for most applications.
Feedback is extracted from the primary by Q1, whichsamples the flyback pedestal during the switch off time.Typical DPMs draw approximately 1mA supply current.The primary is also loaded with 1mA for optimum regula-tion and ripple. Primary snubbing components, a neces-sity in most flyback circuits, are obviated by the primaryfeedback rectifier and smoothing capacitor. Although thiscircuit has been set up for 9V output (9.3V, to be exact),some DPMs need 5V or 7V. Use a 4.3kΩ or 6.2kΩ resistorin place of R1 for these voltages. The output voltage is setby
R1 = (VOUT – 0.7)/1mA.
Do not attempt to regulate the output beyond 10V or youwill exceed the maximum switch rating of the LT1303. TheLT1111 is better suited for higher voltage applications.
Output ripple measures 200mVP–P and can be proportion-ately reduced by increasing the output capacitance. Ifmore ripple is acceptable, the output capacitors can bereduced in value. A shutdown feature is available on theLT1303, useful where a “sleep” function is included tosave power.
With each output loaded at 1mA, the input current is16.5mA on a 5V supply. This figure rises to about 45mA ona 1.8V (2-cell) input. If the system is battery operated andif the battery voltage does not exceed 7V, operate thecircuit directly from the battery for best efficiency. In line-operated equipment, use a regulated 5VDC or 3.3VDCsupply.
DIDPM_01.eps
VIN
LT1303SHDNONOFF
10µF25V
1.8VDC–6VDC
10µF25V
10µF25V
FB
PGNDGND
SW
R21.2k
R18.2k
MBR0520L
MBR-0520L
Q12N3906
10µF25V
10µF25V
10.7µHCOILTRONICS
VP1-0190
DIGITALPANEL
METERS
10µF25V
5 × MBR0520L
10µF25V
+
+
+
+
+
+
+
Figure 182. LT1303 Flyback Regulator Provides Fully Floatingand Isolated 9V Supplies to Five Independent Digital PanelMeters. Substitute 4.3k for R1 if 5V Meters are Used.
Application Note 84
AN84-95
THE LTC1514/LTC1515 PROVIDE LOW POWERSTEP-UP/STEP-DOWN DC/DC CONVERSIONWITHOUT INDUCTORSby Sam Nork
Introduction
Many applications must generate a regulated supply froman input source that may be above or below the desiredregulated output voltage. Such applications place uniqueconstraints on the DC/DC converter and, as a general rule,add complexity (and cost) to the power supply. A typicalexample is generating 5V from a 4-cell NiCd battery. Whenthe batteries are fully charged, the input voltage is around6V; when the batteries are near end of life, the input voltagemay be as low as 3.6V. Maintaining a regulated 5V outputfor the life of the batteries typically requires an inductor-based DC/DC converter (for example, a SEPIC converter)or a complex hybrid step-up/step-down solution. TheLTC1514/LTC1515 family of switched capacitor DC/DCconverters handles this task with only three externalcapacitors (Figure 183).
A unique architecture allows the parts to accommodate awide input voltage range (2.0V to 10V) and adjust theoperating mode as needed to maintain regulation. Hence,the parts can be used with a wide variety of battery and/or adapter voltages. Low power consumption (IQ = 60µAtyp) and low parts count make the parts well suited forspace-conscious low power applications, such as cellularphones, PDAs and portable instruments. The parts comein adjustable and fixed output-voltage versions and in-clude additional features such as power-on reset capabil-ity (LTC1515 family) and an uncommitted comparator thatis kept alive in shutdown (LTC1514 family).
Dual Output Supply from a 2.7V to 10V Input
The circuit shown in Figure 185 uses the low-batterycomparator as a feedback comparator to produce anauxiliary 3.3V regulated output from the VOUT of theLTC1514-5. A feedback voltage divider formed by R2 andR3 connected to the comparator input (LBI) establishesthe output voltage. The output of the comparator (LBO)enables the current source formed by Q1, Q2, R1 and R4.When the LBO pin is low, Q1 is turned on, allowing currentto charge output capacitor C4. Local feedback formed byR4, Q1 and Q2 creates a constant current source from the5V output to C4. Peak charging current is set by R4 and theVBE of Q2, which also provides current limiting in the caseof an output short to ground. R5 pulls the gate of Q1 highwhen the auxiliary output is in regulation. C5 is used toreduce output ripple. The combined output current fromthe 5V and 3.3V supplies is limited to 50mA. Since theregulator implements a hysteretic feedback loop in placeof the traditional linear feedback loop, no compensation isneeded for loop stability. Furthermore, the high gain of thecomparator provides excellent load regulation and tran-sient response.
Conclusion
With low operating current, low external parts count androbust protection features, the LTC1514 and LTC1515 arewell-suited to low power step-up/step-down DC/DC con-version. The shutdown, POR and low-battery detect fea-tures provide additional value and functionality. The sim-plicity and versatility of these parts make them ideal forlow power DC/DC conversion applications.
1514_01.eps
LTC1515-3.5
100k
SHDN1ON OFF
5V
RESET
3.3V
POR2
5/33
GND
VOUT
VIN
VIN = 4 CELLS
VOUT = 5V ± 4% OR 3.3V ± 4%IOUT = 0 TO 50mA
C1+
C1–4
8
7
6
5 0.22µF 10µF 10µF+ +
VIN (VOLTS)
4.8
4.9
5.0
5.1
5.2
V OUT
(VOL
TS)
1514_XX.eps
3 4 65
Figure 183. Programmable 5V/3V Power-Supply with Power-On Reset Figure 184. VOUT vs VIN for Figure 183’s Circuit
Application Note 84
AN84-96
1514_04.eps
LTC1514-5
SHDN1
ON OFF
LBO2
LBI3
GND
VOUT
VIN
VOUT = 3.3V
VOUT = 5V
VIN = 2.7V TO 10V
C1+
C1–4
8
7
R147k
R410Ω
R3750k,1%
R2402k,1%
Q2
Q1
6
5C10.22µF
C410µF
C52.2nF
C322µF
C210µF
R5220k
Q1 = TP0610TQ2 = MMBT3906LT1
+ ++
LTC1626 LOW VOLTAGE MONOLITHIC STEP-DOWNCONVERTER OPERATES FROM A SINGLE Li-Ion CELLby Tim Skovmand
Introduction
The LTC1626 is a monolithic, low voltage, step-downcurrent mode DC/DC converter with an input supply volt-age range of 2.5V to 6V, making it ideal for single-cell Li-Ion or 3- to 4-cell NiCd/NiMH applications. A built-in0.32Ω P-channel switch (VIN = 4.5V) allows up to 0.6A of
output current. The maximum peak inductor current isexternally programmable to minimize component size inlower current applications.
The LTC1626 incorporates automatic power saving BurstMode operation to reduce gate-charge losses when theload current drops below the level required for continuousoperation. With no load, the converter draws only 160µA;in shutdown it draws a mere 1µA—making it ideal forcurrent-sensitive applications.
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
0 1 2 3 4 5 6 7
DISCHARGE TIME (HOURS)
Li-Io
n CE
LL V
OLTA
GE (V
)
SGND
SHUTDOWN
1k
3900pF
1000pF
0.1µF
(VIN = 2.7V TO 4.5V)
VOUT
COUT†
100µF10V
CIN††
47µF16VPWR VIN VIN
SW
CT
ITH
SENSE+
SENSE–
PGND
LTC1626
D1MBR0520LT1
L1*22µH
RSENSE**
VFB
10k1%
10k1%
100pF
(2.5V/0.25A)0.1
+
+
Ω
CT270pF
LBIN
LBOUT
SHDN
SINGLELi-IonCELL
+
* SUMIDA CDRH62-220** IRC 1206-R100F
† AVX TPSD107K010†† AVX TPSD476K016
Figure 185. Using the Low-Battery Comparator as a Feedback Comparator to Producean Auxiliary 3.3V Regulated Output from the VOUT of the LTC1514-5
Figure 186. Typical Single-CellLi-Ion Discharge Curve Figure 187. Single-Cell Li-Ion Battery to 2.5V Converter
Application Note 84
AN84-97
Single-Cell Li-Ion Operation
As shown in Figure 186, a fully charged single-cell Li-Ionbattery begins the discharge cycle at either 4.1V or 4.2V(depending upon the manufacturer’s charge voltage speci-fications). During the bulk of the discharge time, the cellproduces between 3.5V and 4.0V. Finally, toward the endof discharge, the cell voltage drops fairly quickly below 3V.It is recommended that the discharge be terminated some-where between 2.2V and 2.8V (again, depending upon themanufacturer’s specifications).
The LTC1626 is specifically designed to accommodate asingle-cell Li-Ion discharge curve. For example, using thecircuit shown in Figure 187, it is possible to produce astable 2.5V/0.25A regulated output voltage with as little asa 2.7V from the battery—thus obtaining the maximumpossible run time.
High Efficiency Operation
Using the circuit shown in Figure 188, efficiencies ofgreater than 90% are maintained from 20mA to 250mA ofload current with a 3.5V input supply voltage, as shown inFigure 189.
SGND
SHUTDOWN
470Ω
3900pF270pF
1000pF
0.1µF
VIN
33µH VOUT
(2.7V TO 6V)
COUT††
100µF6.3V
CIN†††
47µF 16V
PWR VIN VIN
SW
CT
ITH
SENSE+
SENSE–
P GND
LTC1626 D1†
L* RSENSE**
VFB
10k1%
10k1%
100pF
2.5V/0.25A
0.1
+
+Ω
CT
* COILTRONICS CTX33-4** IRC 1206-R100F† MBRS130LT
†† AVX TPSC107M006R0150††† AVX TPSD476K016
0.01 0.10 1.00
OUTPUT CURRENT (A)
EFFI
CIEN
CY (%
)
70
75
80
85
90
95
100
L1 = 33µHVOUT =2.5VRSENSE =0.1ΩCT = 270pF
VIN = 3.5V
SGND
SHUTDOWN
1k
3900pF 270pF
1000pF
0.1µF
L1*22µH VOUT
COUT†
100µF10V
CIN††
47µF16VPWR VIN VIN
SW
CT
ITH
SENSE+
SENSE–
PGND
LTC1626
D1MBR0520LT1
VFB
R110k1%
R210k1%
100pF
2.5V/0.25A
RSENSE**0.1Ω
+
+
CT
LBIN
LBOUT
SHDN
(VIN = 2.7V TO 6V)
3 OR 4 CELL
NiCD ORNiMH
+
FOR 3.3V:R1 = 15k 1%R2 = 9.09k 1%
* SUMIDA CDRH62-220** IRC 1206-R100F† AVX TPSD107K010
†† AVX TPSD476K016
Figure 188. High Efficiency 2.5V Step-Down Converter
Figure 189. Efficiency vs Output Load Current
Figure 190. 3- or 4-Cell NiCd/NiMH to 2.5V Converter
Application Note 84
AN84-98
Typical Applications
3- or 4-Cell NiCd/NiMH DC/DC Converter
Figure 190 is a schematic diagram that shows the LTC1626being powered from a 3- or 4-cell NiCd or NiMH batterypack. (This circuit is also suitable for operation from threeor four alkaline cells.) All the components shown in thisschematic are surface mount and have been selected tominimize the board space and height. The output voltageis set at 2.5V, but is easily programmed to 3.3V for 4-cellapplications. Simply modify the two output ladder resis-tors, R1 and R2, from 10k each to 15k and 9.09k, respec-tively, as shown in Figure 190.
Single Li-Ion 3.3V Buck/Boost Converter
The circuit shown in Figure 191 produces 3.3V from aninput voltage ranging from 2.5V to 4.5V. The two windingsof a common inductor core are used to implement thiscircuit. Note that the current sense resistor is connected toground. The table in Figure 191 shows the output currentcapability as a function of battery voltage.
Conclusion
The LTC1626 is specifically designed to operate from asingle-cell Li-Ion battery. With its low dropout, high effi-ciency and micropower operating modes, it is ideal forbattery operated products and efficiency-sensitive de-vices such as cellular phones and handheld industrial andmedical instruments.
SGND
SHUTDOWN
1k
3300pF 75pF1000pF
0.1µF
(2.5V TO 4.2V)
VOUT
COUT †
100µF10V
CIN††
100µF16VPWR VIN VIN
SW
CT
ITH
SENSE+SENSE–
PGND
LTC1626D1
MBRS130LT1
L1B33µH
RSENSE**0.1Ω
VFB
15k1%
9.09k1%
100pF
3.3V
+
+
CT
LBIN
LBOUT
SHDN
+
L1A33µH
1 2
4
3
33µF10V*
1
23
4TOP VIEW
L1B
L1B
L1A
L1A
VIN (V)
2.53.03.54.04.2
IOUT (mA)
200350
0500*0500*0500*
MANUFACTURER
COILTRONICSDALE
PART NO.
CTX33-4LPT4545-330LA
+Li-Ion
SINGLECELL
* DESIGN LIMIT
* SANYO OS-CON CAPACITOR** IRC 1206-R100F† AVX TPSD107M010R0100
††AVX TPSE107M016R0100
Figure 191. Single-Cell Li-Ion to 3.3V Buck/Boost Converter
Application Note 84
AN84-99
12V WALL CUBE TO 5V/400mA DC/DC CONVERTERIS 85% EFFICIENTby Steve Pietkiewicz
The ubiquitous 12V wall cube, power source of countlesselectronic products, generates an unregulated DC voltagebetween 8V and 18V, depending on line voltage and load.If you use a linear regulator to drop the voltage to 5V, a400mA load means the linear regulator must dissipate 5Wunder worst-case conditions. To deal with this heat, youmust provide adequate heat sinking, increasing yourproduct’s size and weight. Additionally, the heat is some-times objectionable to customers. These factors can ne-gate the cost advantage of a linear regulator. Figure 192’scircuit, a negative buck converter, delivers 5V at loads upto 400mA from a 7V–25V input with peak efficiency of85%, eliminating the need for a heat sink. Since theLT1307B (U1) is intended for use with a low input voltage,Q1 and Q2 are used to make a simple preregulator,providing 1.9V for U1’s VIN pin. The IC switches at 600kHz,allowing a low cost 22µH inductor and 10µF ceramicoutput capacitor to be used. Q3 is needed to level shift theoutput voltage because U1’s feedback pin is referenced tothe negative input. Output ripple measures 10mVP-P at aload of 400mA. The circuit’s efficiency is detailed in Figure193, and response to a load step from 150mA to 300mAis shown in Figure 194. Input bypass capacitor C1 seesworst-case RMS ripple current equal to one-half theoutput current and should have an ESR of less than 0.5Ω.Take care during construction to keep R1–R3 and Q3 closeto U1’s FB pin and away from the SW pin to preventunwanted coupling. Use a ground plane and keep tracesfor the power components short and direct.
SHDN VIN
SW
VC
FB
GND
U1LT1307B
Q22N3904
10k
10k
30k
30k
Q12N3904 1µF
CERAMIC
C210µF
CERAMIC1N5818
L122µH
R3100k
Q32N3906
C133µF25V
5V400mA
L1 = SUMIDA CD54-220
12VUNREGULATED
SUPPLY
+
–
R242.2k, 1%
R112.1k1%
DI_ADAP_01.EPS
1000pF
3 6
5
2
41
+
Although it might seem unsettling that the negative side ofthe wall cube is not grounded, remember that the 9V wallcube floats. The circuit merely regulates the negative side,rather than the more conventional positive side.
5070
72
74
76
78
80
82
84
86
88
90
100 150 200LOAD CURRENT (mA)
EFFI
CIEN
CY (%
)
250 300 350 400
DI-ADAP_02.EPS
VIN = 8V
VIN = 12V
VIN = 18V
Figure 192. This Negative Buck Converter Delivers 5V at 400mA from a 7V–25V Input
Figure 193. Efficiency Peaks at 85%; It Is Above80% Over an Input Range of 8V–18V
Figure 194. Load-Step Response; the LoadChanges from 150mA to 300mA
VOUT0.5V/DIV
50µs/DIVDI_ADAP_03.eps
IL10.5A/DIV
ILOAD0.3A/0.15A
Application Note 84
AN84-100
MICROPOWER 600kHz FIXED-FREQUENCYDC/DC CONVERTERS STEP UP FROM A 1-CELLOR 2-CELL BATTERYby Steve Pietkiewicz
Linear Technology introduces two new micropower DC/DC converters designed to provide power from a single-cell or higher input voltage. The LT1308 features anonboard switch capable of handling 2A with a voltage dropof 300mV and operates from an input voltage as low as 1V.The LT1317, intended for lower power requirements,operates from an input voltage as low as 1.5V. Its internalswitch handles 600mA with a drop of 360mV. Both devicesfeature Burst Mode operation at light load; efficiencies areabove 70% for load currents of 1mA. Both devices switchat 600kHz; this high frequency keeps associated powercomponents small and flat; additionally, troublesome in-terference problems in the sensitive 455kHz IF band are
avoided. The LT1308 is intended for generating power onthe order of 2W–5W. This is sufficient for RF poweramplifiers in GSM or DECT terminals or for digital-camerapower supplies. The LT1317, with its smaller switch, cangenerate 100mW to 2W of power. The LT1317 is availablein LTC’s smallest 8-lead package, the MSOP. This packageis approximately one-half the size of a standard 8-lead SOpackage. The LT1308 is available in the 8-lead SO package.
VIN
SW
FB
LT1308
L14.7µH
3V TO 4.2V
D1
LBO
LBI
RC47k
R2100k
5V1A
R1301k
CC22nF
1308_01,eps
C1100µF
C2100µF
Li-IonCELL
VC GND
SHDN
AVX TPS SERIESINTERNATIONAL RECTIFIER 10BQ015
+2200µF
C1,C2:D1:
COILTRONICS CTX5-1COILCRAFT DO3316-472
L1:
+
LOAD CURRENT (mA)1
EFFI
CIEN
CY (%
)
95
90
85
80
75
70
6510 100 1000
1308 F01a
VIN = 4.2V
VIN = 3.6V
VIN = 3V
VIN
SW
FB
LT1308
L14.7µH
D1
LBO
LBI
RC47k
R2100k
3.3V400mA
R1169k
CC22nF
1308_04.eps
C110µF
C2100µF
NiCDCELL
VC GND
SHDN
C1: CERAMICC2: AVX TPS SERIESD1: IR 10BQ015
+
L1: COILTRONICS CTX5-1COILCRAFT DO3316-472
LOAD CURRENT (mA)
90
85
80
75
70
65
60
55
501 100 1000
1308 G01
10
EFFI
CIEN
CY (%
)
VIN = 1.2VVOUT = 3.3VR1 = 169k
VOUT200mV/DIV
AC COUPLED
INDUCTORCURRENT
1A/DIV1ms/DIV
Figure 197. Transient response of DC/DCconverter: VIN = 3V, 0A–1A load step
Figure 195. Single Li-Ion Cell to 5V/1A DC/DC Converter
Figure 196. Efficiency of Figure 195’s Circuit Figure 199. Efficiency of Figure 198’s Circuit Reaches 81%
Figure 198. Single NiCd Cell to 3.3V 400mA DC/DC Converter
Application Note 84
AN84-101
Single Li-Ion Cell to 5V/1A DC/DC Converter for GSM
GSM terminals have emerged as a worldwide standard. Acommon requirement for these products is an efficient,compact, step-up converter to develop 5V from a single Li-Ion cell to power the RF amplifier. The LT1308 performsthis function with a minimum of external components. Thecircuit is detailed in Figure 195. Many designs use a largealuminum electrolytic capacitor (1000µF to 3300µF) at theDC/DC converter output to hold up the output voltageduring the transmit time slice, since the amplifier canrequire more than 1A. The output capacitor, along with theLT1308 compensation network, serves to smooth out theinput current demanded from the Li-Ion cell. Efficiency,which reaches 90%, is shown in Figure 196. Transientresponse of a 0A to 1A load step with typical GSM profiling(1:8 duty cycle, 577µs pulse duration) is depicted in Figure197. Voltage droop (top trace) is 200mV. Inductor current
(bottom trace) increases to 1.7A peak; the input capacitorsupplies some of this current, with the remainder drawnfrom the Li-Ion cell.
Single NiCd Cell to 3.3V/400mA Supply for DECT
Only minor changes are required in Figure 195’s circuit toconstruct a single-cell NiCd to 3.3V converter. The largeoutput capacitor is no longer required as the outputcurrent can be handled directly by the LT1308. Figure 198shows the DECT DC/DC converter circuit. Efficiency, reach-ing 81% from a 1.2V input, is pictured in Figure 199.Transient response of a typical DECT load of 50mA to400mA is detailed in Figure 200. Output voltage droop (toptrace) is under 200mV. Figure 201 zooms in on a singlepulse to show the output voltage and inductor currentresponses more clearly.
VOUT200mV/DIV
AC COUPLED
400mAILOAD
50mA
IL11A/DIV
100µs/DIV
VOUT200mV/DIV
AC COUPLED
400mA
2ms/DIV
50mA
Figure 200. DECT Load Transient Response:with a Single NiCd Cell, the LT1308 Provides 3.3Vwith a 400mA Pulsed Load. The Pulse Width = 416µs
Figure 201. DECT Load Transient Response:Faster Sweep Speed (100µs/DIV) Details VOUT andInductor Current of a Single DECT Transmit Pulse
VIN SW
FBLT1308
L1AN = 110µH
L1CN = 0.3
R447k
R1100k
5V200mA3.3V200mA
CCD BIAS–10V10mA
CCD BIAS18V10mA
VIN1.6V
TO 6V
R3340k
R22.01M
D1
C722nF
C81nF
1308_08.eps
C610µF
VC
GND
SHDN
C1, C2, C3 = AVX TPSC4, C5 = AVX TAJC6 = CERAMIC
18 2 3
L1BN = 0.7
3
4
C2100µF
C1100µF
+D2
D3
D4
+C3100µF
+ C410µF
+
C510µF
+
L1DN = 3.5
7
6
L1EN = 2
6
5
D1, D2 = IR 10BQ015D3, D4 = BAT-85L1 = COILTRONICS CTX02-13973
Figure 202. This Digital Camera Power Supply Delivers 5V/200mA, 3.3V/200mA, 18V/10mA and –10V/10mA from 2 AA Cells
ILOAD
Application Note 84
AN84-102
2-Cell Digital Camera SupplyProduces 3.3V, 5V, 18V and –10V
Power supplies for digital cameras must be small andefficient while generating several voltages. The DSP andlogic need 3.3V, the ADC and LCD display need 5V andbiasing for the CCD element requires 18V and –10V. Thepower supplies must also be free of low frequency noise,so that postfiltering can be done easily. The obviousapproach, to use a separate DC/DC converter IC for eachoutput voltage, is not cost-effective. A single LT1308,along with an inexpensive transformer, generates 3.3V/200mA, 5V/200mA, 18V/10mA and –10V/10mA from apair of AA or AAA cells. Figure 202 shows the circuit. Acoupled-flyback scheme is used, actually an extension ofthe SEPIC (single ended primary inductance converter)topology. The addition of capacitor C6 clamps the SW pin,eliminating a snubber network. Both the 3.3V and 5Voutputs are fed back to the LT1308 FB pin, a techniqueknown as split feedback. This compromise results inbetter overall line and load regulation. The 5V output hasmore influence than the 3.3V output, as can be seen fromthe relative values of R2 and R3. Transformer T1 isavailable from Coiltronics, Inc. (561-241-7876). Efficiencyvs input voltage for several load currents on both 3.3V and5V outputs is pictured in Figure 203. The CCD biasvoltages are loaded with 10mA in all cases.
LT1317 2-Cell to 5V DC/DC Converter
Figure 204 shows a simple 2-cell to 5V DC/DC converterusing the LT1317. This device generates a clean, lowripple output from an input voltage as low as 1.5V.Designed for 2-cell applications, it offers better perfor-
mance than its 1-cell predecessor, the LT1307. More gainin the error amplifier results in lower Burst Mode ripple,and an internal preregulator eliminates oscillator variationwith input voltage. For comparison, Figure 205 detailstransient responses of both the LT1307 and the LT1317generating 5V from a 3V input. The load step is 5mA to200mA. Output capacitance in both cases is 33µF. TheLT1307 has low frequency ripple of 100mV, whereas theLT1317 Burst Mode ripple of 20mV is the same as the600kHz ripple resulting from the output capacitor’s ESRwith a 200mA load.
Single Li-Ion Cell to ±4V DC/DC Converter
By again employing the SEPIC topology, a ±4V supply canbe designed with one IC. Figure 206’s circuit generates 4Vat 70mA and –4V at 10mA from an input voltage rangingfrom 2.5V to over 5V. Maximum component height is2mm. This converter uses two separate inductors (L1 andL2), so it is an uncoupled SEPIC converter. This reducesthe overall cost, but requires that all output current pass
INPUT VOLTAGE (V)1
EFFI
CIEN
CY (%
)
70
80
85
75
65
55
5
1308_09.EPS
60
5021.5 2.5 3.5 4.53 4
90
100mA LOADS
150mA LOADS 200mA LOADS
VIN
SW
FB
LT1317
L122µH
D1
LBO
LBI
RC100k
R2324k1%
5V200mA
R11M
CC680pF
1308_10.eps
C110µF10V
C233µF
2 CELLS
VC GND
SHDN
C1: CERAMICD1: MOTOROLA MBRO520LL1: 22µH SUMIDA CD43-220
SHUTDOWN
+
VOUTLT1307
100mV/DIV5V OFFSET
VOUTLT1317
100mV/DIV5V OFFSET
500µs/DIV
200mA5mA
Figure 205. The LT1317 Has ReducedBurst Mode Ripple Compared to the LT1307Figure 203. Camera Power Supply Efficiency Reaches 78%
Figure 204. 2-Cell to 5V Boost Converter Using the LT1317
ILOAD
Application Note 84
AN84-103
LT1610 MICROPOWER STEP-UP DC/DC CONVERTERRUNS AT 1.7MHZby Steve Pietkiewicz
Introduction
The LT1610, a micropower DC/DC converter IC, addressesthe issue of footprint in several ways. First, the switchingfrequency is 1.7MHz, allowing the use of small, inexpen-sive, minimal-height inductors and capacitors. Second,the frequency-compensation components have been inte-grated, eliminating the requirement for an external RCnetwork in most applications. Finally, the device comes inLTC’s 8-lead MSOP package, one-half the size of the 8-lead SO package.
The LT1610’s input voltage ranges from 1V to 8V, and the30V, 300mA switch allows several different configura-tions, such as boost, SEPIC and flyback, to be successfully
through C1. Since C1 is ceramic, its ESR is low and thereis no appreciable efficiency loss. C5 is charged to –VOUTwhen the switch is off, then its bottom plate is groundedwhen the switch turns on. The negative output is fairly wellregulated, since the diode drops tend to cancel. The circuitis switching continuously at rated load, where efficiency is75%. Output ripple is under 40mV and can be reducedfurther with conventional postfiltering techniques.
Conclusion
The LT1308 and LT1317 provide low noise compactsolutions for contemporary portable-product powersupplies.
R347k
C6680pF
L122µH
L222µH
C233µF
C110µF
C51µF
C41µF
C315µF
D2A D2B
D1
–VOUT–4V/10mA
+VOUT4V/70mA
VIN2.5V–5V
SHUTDOWN
R1 1M
R2442k
VIN SW
VC GND
LB0
SHDN
FB
LB1
LT1317
L1, L2 = MURATA LQH3C220C1 = MURATA GRM235Y5V106Z01D1 = MBR0520D2 = BAT54S (DUAL DIODE)C2 = AVX TAJB33M6010C3 = AVX TAJA156MO1O
C4, C5 = CERAMIC
+
+
VIN SW
VC PGND
FBSHDN
6 5
2
7
41
8
3
GNDCOMP
LT1610 C222µF
C122µF
1 CELL
L14.7µH D1 VOUT
3V30mA
C1, C2: AVX TAJA226M010RD1: MOTOROLA MBR0520L1: MURATA LQH3C4R7M24
1610 TA01
++
R11M
R2681k
Figure 206. This SIngle Li-Ion Cell to ±4V DC/DC Converter Has a Maximum Height of 2mm
Figure 207. This Single Cell to 3V Converter Delivers 30mA
Application Note 84
AN84-104
implemented. Output voltage can be up to 28V in boostmode. Operating quiescent current is 50µA unloaded;grounding the shutdown pin reduces the current to 0.5µA.The device can generate 3V at 30mA from a single (1V)cell, or 5V at 100mA from two cells (2V). Configured as aLi-Ion cell to 3.3V SEPIC converter, the LT1610 can deliver100mA. In boost mode, efficiency ranges from 60% at a100µA load to 83% at full load.
Single-Cell to 3V DC/DC Converter
A 1V to 3V boost converter is shown in Figure 207. Thespecified components take up very little board space. The4.7µH Murata inductor specified measures 2.5mm by3.2mm and is only 2mm high. The 22µF AVX “A” casetantalum capacitors measure 1.6mm by 3.2mm and are1.6mm tall. Circuit efficiency, which reaches 77%, isdetailed in Figure 208. Transient response to a 1mA to31mA load step is pictured in Figure 209. The devicefeatures Burst Mode operation at light loads. This can beseen at a load of 1mA. When the load is increased to 31mA,the device shifts to constant-frequency switching and peakswitch current is controlled to achieve output regulation.
2-Cell to 5V DC/DC Converter
By simply changing the feedback resistor values, theLT1610 can generate 5V. Figure 210’s circuit generates 5Vat a load of up to 100mA from a 2-cell input. Figure 211’sgraph shows efficiency the of the circuit, which reaches83%. This circuit is also suitable for 3.3V to 5V conversion,supplying over 200mA.
Li-Ion to 3.3V SEPIC Converter
Figure 212 employs the SEPIC (single ended primaryinductance converter) topology to provide a regulated3.3V output from an input that can range above or belowthe output voltage. Although the circuit requires twoinductors and a ceramic coupling capacitor, the totalfootprint of this solution is still attractive compared withalternative methods of generating 3.3V, such as a boostconverter followed by a linear regulator. The circuit can
LOAD CURRENT (mA)0.1
EFFI
CIEN
CY (%
)85
80
75
70
65
60
55
501 10 100
1610 TA02
VOUT = 3V
VIN = 1.5V
VIN = 1V
VIN = 1.25V
VIN SW
VC PGND
FBSHDN
6 5
2
7
41
8
3
GNDCOMP
LT1610 C215µF
C115µF
2 CELLS
L14.7µH D1
VOUT5V/100mA
1610 TA04
++
1M
332k
C1, C2: AVX TAJA156M010RD1: MOTOROLA MBR0520L1: SUMIDA CD43-4R7 MURATA LQH3C4R7M24Figure 208. Single-Cell Converter Efficiency Reaches 77%
Figure 209. Transient Load Response of Single-CellConverter, Load Stepped from 1mA to 31mA
Figure 210. 2 Cell to 5V Converter Delivers 100mA at 2V Input
LOAD CURRENT (mA)0.1
EFFI
CIEN
CY (%
)
90
80
70
60
501 10
1610 TA05
100 1000
VIN = 1.5V
VIN = 3V
VIN = 2V
Figure 211. 2-Cell Converter Efficiency Reaches 83%
VOUT50mV/DIV
AC COUPLED
31mA1mA
VIN = 1.25VVOUT = 3V
500µs/DIV
ILOAD
IL1100mA/DIV
Application Note 84
AN84-105
VIN SW
VC
PGND
FB
SHDN
SHUTDOWN
6 5
2
7 4
1
8 3
GND
COMP
LT1610 C222µF6.3V
C122µF6.3V
L14.7µH
L24.7µH
C34.7µF
CERAMIC D1INPUTLi-ION
3V to 4.2V
VOUT3.3V100mA
1610 TA06
++
1M
604k
C1, C2: AVX TAJA226M010RC3: AVX 1206YG475D1: MOTOROLA MBR0520L1, L2: MURATA LQH3C4R7M24
supply up to 100mA. Efficiency, while lower than that of astandard boost converter, reaches approximately 73%.Unlike a boost converter, this topology provides input-to-output isolation. The output is completely disconnected
Figure 212. Li-Ion to 3.3V SEPIC Converter Delivers 100mA
LOW NOISE 33V VARACTOR BIAS SUPPLYby Jeff Witt
Wideband tuning circuits, such as those used in cabletelevision systems, require a power supply for driving avaractor. This bias supply is usually at a voltage higherthan the system supply voltage, allowing a large tuningrange. The supply must have very little noise; voltage
680Ω150pF
L1 22µHC1
15µF10V
D3
D2
D1
C3 0.1µF47Ω
0.1µF
0.1µF0.1µF
C210µF35V150k
5.90k3300pF
33k
GND
VC
VIN SW
FB
LT1317B
VOUT33V0mA TO 10mA
VIN
D1 TO D3:C1:C2:L1:
MOTOROLA MMBD914LT1AVX TAJ156M010SANYO 35CV33GXMURATA LQH3C220
A B
SHDN +
+
ripple, for example, can appear as sidebands on a localoscillator. This circuit takes advantage of the fixed operat-ing frequency of the LT1317B boost regulator to generatea low noise 33V bias voltage.
from the battery in shutdown mode, preventing inadvert-ent battery discharge through the load. The LT1610’s sub-µA shutdown current reduces standby losses, increasingbattery life.
Figure 213. This Circuit Generates a Low Noise Bais Supply for Varactor-Based Tuning Circuits
Application Note 84
AN84-106
THE LTC1516 CONVERTS TWO CELLS TO 5V WITHHIGH EFFICIENCY AT EXTREMELY LIGHT LOADSby Sam Nork
Many battery-powered applications require very smallamounts of load current from the regulated supply overlong periods of time, followed by moderate load currentsfor short periods of time. In these types of applications (forexample, remote data-acquisition systems, hand-held re-mote controls, and the like), the discharge rate of thebattery is dominated by the overall current demands underlow load conditions. In such low load systems, a primarysource of battery drain is the DC/DC converter that con-verts the battery voltage to a regulated supply.
The circuit shown in Figure 215 converts an input voltagefrom two cells to 5V using a switched-capacitor charge-pump technique. An integral comparator on the LT1516senses the output voltage and enables the charge pump as
1
8
7
2
C1+
C1–
SHDN
VIN2 CELLS
10µF
C1 = C2 = 0.22µF
10µF
C2C1
LTC1516
ON/OFF
C2+
C2–
GND
VOUT
4
5
6
3VOUT = 5V ±4%IOUT = 0mA TO 20mA+ +
Figure 214. The Output Ripple of Figure 213’s Supply as itDelivers 5mA at 33V from a 5V Input; Traces A and B ShowRipple Before and After the RC Output Filter, Respectively
The circuit (Figure 213) is a simple boost regulator with itsoutput voltage doubled by diodes D2 and D3 and capacitorC3. With this doubler, the circuit can generate an outputvoltage greater than the voltage rating of the LT1317B’sinternal power switch. This supply can deliver 10mA at
NODE AOUTPUT RIPPLE
20mV/DIV
NODE BOUTPUT RIPPLE
20mV/DIV
1µs/DIV
33V from a 3V to 6V input, allowing operation from either3.3V or 5V logic rails. The high operating frequency(600kHz) results in low, easily filtered output ripple, asshown in Figure 214. The high frequency also allows theuse of small, low cost external components.
Figure 215. 2 Cell to 5V Converter
Application Note 84
AN84-107
the output begins to droop. The charge pump’s 2-phaseclock controls the internal switching of flying caps C1 andC2. (See Figure 216.) On phase one of the clock, the flyingcaps are connected between VIN and GND. On phase two,the negative plate of C1 is connected to VIN, the negativeplate of C2 is connected to the positive plate of C1, and thepositive plate of C2 is connected to the output. During thisphase of the clock, the potential on the top plate of C2 isapproximately 3 • VIN and the charge is dumped from C2onto the output cap to raise the output voltage. Therepeated charging and discharging of C1 and C2 continuesat a nominal frequency of 600kHz until the output voltagehas risen above the internal comparator’s trip point.
S2A
S2B
S2C
S1A
S1B
S1C
S1D
0.22µF
10µF
0.22µF
C1+
C1–
C2+
C2–
10µF
VOUT
VIN
CHARGE PUMP IOUT (mA)
50
80
70
60
90
100
EFFI
CIEN
CY (%
)
1000.01 1.00.1 10
VIN = 2V
VIN = 3VVIN = 2.75V
VIN = 2.5V
VIN = 2.25V
When the battery cells are fully charged (approximately1.5V per cell, for a nominal 3V VIN), the circuit operates asa voltage doubler to maintain regulation. In doubler mode,only C2 is charged to VIN and discharged onto VOUT whenthe charge pump is enabled. As the batteries dischargeand/or the load increases, the circuit will change fromdoubler mode to tripler mode. Under light load conditions,the part will remain in doubler mode until VIN has droppedbelow 2.55V. Under heavier loads, the part will go intotripler mode at a higher VIN to maintain regulation. Byswitching operating modes as the VIN and the load condi-tions change, the LTC1516 optimizes overall efficiency forthe life of the batteries. As shown in Figure 217, Figure215’s circuit achieves better than 70% efficiency with loadcurrents from 50µA to 20mA for almost the entire life of thebatteries.
Regulators—Linear
LOW DROPOUT REGULATOR DRIVER HANDLESFAST LOAD TRANSIENTS AND OPERATES ONA SINGLE 3V–10V INPUTby Lenny Hsiu
Introduction
The LT1573 is designed to provide a low cost solution toapplications requiring high current, low dropout and fasttransient response. When combined with an external PNPpower transistor, this device provides up to 5A of loadcurrent with dropout voltages as low as 0.35V. The LT1573’s
circuitry is designed for extremely fast transient response.This greatly reduces the bulk storage capacitance requiredwhen the regulator is used in applications with fast, highcurrent load transients.
Base-drive current to the external PNP is limited forinstantaneous protection and a time-delayed latch pro-tects the regulator from continuous short circuits. Thelatch time-out period can be varied by an external capaci-tor. Guaranteed minimum available base-drive current tothe external PNP is 250mA. The LT1573 is equipped withan active-high shutdown and a thermal shutdown func-tion. The shutdown function can be used to reset the
Figure 216. LT1516 Charge Pumpin Trippler Mode, Discharge Cycle
Figure 217. Efficiency vs VOUT for Figure 215’s Circuit
Application Note 84
AN84-108
overcurrent latch. The thermal shutdown function can beused to protect the PNP power transistor if it is thermallycoupled to the LT1573.
Basic Regulator Circuit
The adjustable-output LT1573 circuit shown in Figure 218senses the regulator output voltage from its feedback pinvia the output voltage divider and drives the base of theexternal PNP transistor to maintain the regulator output atthe specified value. For fixed-output versions of the LT1573,
LOAD
R11.6k
R21k
COUT1*
QOUTMOTOROLAD45H11
RB50ΩRD
24Ω
CC 100pF
RC 1k
CTIME***
VIN 4.5V–5.5V
VOUT = 1.265 • (1 + R1/R2)
+
+
CIN100µFTANT
COUT2**+
+
VIN
GND
LT1573
DRIVE
VOUT
COMP
FB
SHDN†
LATCH
*
*****
†
FOR T <45˚C, COUT1 = 24 × 1µF Y5V CERAMIC SURFACE MOUNT CAPACITORSFOR T >45˚C, COUT1 = 24 × 1µF X7R CERAMIC SURFACE MOUNT CAPACITORSPLACE COUT1 IN THE MICROPROCESSOR SOCKET CAVITYCOUT2 = 220µF CHIP TANTALUMCTIME = 0.5µF FOR 100ms LATCH-OFF TIME AT ROOM TEMPERATURESHDN (ACTIVE HIGH) SHOULD BE TIED TO GROUND IF NOT USED
the regulator output voltage is sensed from the feedbackpin via an internal voltage divider. In this case, the FB pinis left unconnected. The resistor RD is required for theovercurrent latch-off function. RD is also used to limit thedrive current available to the external PNP transistor andto limit the power dissipation in the LT1573. Limiting thedrive current to the external PNP transistor will limit theoutput current of the regulator, thereby minimizing thestress on the regulator circuit under overload conditions.
See the LT1573 Data Sheet for additional design details.
THE LT1575/LT1577 UltraFast LINEAR REGULATORCONTROLLERS ELIMINATE BULK TANTALUM/ELECTROLYTIC OUTPUT CAPACITORSby Anthony Bonte
Introduction
The LT1575/LT1577 family of single/dual controller ICsare new, easy-to-use devices that drive discrete N-channelMOSFETs as source followers to produce extremely lowdropout, UltraFast™ transient response regulators. Thesecircuits achieve superior regulator bandwidth and tran-sient load performance, and completely eliminate expen-sive tantalum or bulk electrolytic capacitors in the mostdemanding microprocessor applications. For example, a200MHz Pentium® processor can operate with only thetwenty-four 1µF ceramic capacitors that Intel already
requires for the microprocessor. Users realize significantsavings because all additional bulk capacitance is removed.The additional savings of insertion cost, inventory costand board space are readily apparent.
Precision-trimmed adjustable and fixed-output voltageversions accommodate any required microprocessorpower supply voltage. Dropout voltage can be user definedvia selection of the N-channel MOSFET RDS(ON). The onlyoutput capacitors required are the high frequency ceramicdecoupling capacitors. The regulator responds to tran-sient load changes in a few hundred nanoseconds—agreat improvement over regulators that respond in manymicroseconds. The ceramic capacitor network generallyconsists of ten to twenty-four 1µF capacitors, dependingon individual microprocessor requirements. The LT1575/LT1577 family also incorporates current limiting at no
Figure 218. 3.3V, 5A Microprocessor Supply
Application Note 84
AN84-109
additional system cost, provides on/off control and canprovide overvoltage protection or thermal shutdown withthe addition of a few simple external components. TheLT1575 is available in 8-pin SO or PDIP and the LT1577 isavailable in 16-pin narrow-body SO.
UltraFast 5V to 3.3V Low Dropout Regulator
Figure 219 shows the basic regulator control circuit. Theinput voltage is a standard 5V “silver box” and the outputvoltage is set to 3.5V, the Pentium P54 VRE microproces-sor supply voltage. The typical maximum output current isabout 5A in most Pentium microprocessor applications.The output capacitor network consists of only twenty-fourinexpensive 1µF ceramic, surface mount capacitors. Properlayout of this decoupling network is critical to properoperation of this circuit. Consult Linear Technology Appli-cation Note 69: LT1575 UltraFast Linear Controller MakesFast Transient Response Power Supplies, for details onboard layout.
The photo in Figure 220 shows the transient responseperformance for an output load current step of 0.2A to 5A.The main loop compensation in Figure 219’s regulatorcircuit is provided by R1 and C4 at the COMP pin. Capaci-tor C3 introduces a high frequency pole and providesadequate gain margin beyond the unity-gain crossoverfrequency of 1MHz. This compensation network limitsovershoot/undershoot to 50mV under worst-case loadtransient conditions. With a 1% specified worst-caseoutput voltage tolerance, the 100mV output voltage error
budget for a P54 VRE microprocessor is easily met withproduction margin to spare. All bulk tantalum/electrolyticcapacitors are completely eliminated.
The discrete N-channel MOSFET chosen is a low costInternational Rectifier IRFZ24 or equivalent. The inputcapacitance is approximately 1000pF with VDS = 1V. Thespecified on-resistance is 0.1Ω at room temperature andabout 0.15Ω at 125°C. At 7A output current, the dropoutvoltage is only 1.05V. This eases the restriction on localinput decoupling capacitor requirements because signifi-cant droop in the typical 5V input supply voltage ispermitted before dropout voltage operation is reached.(Note that 5V supply tolerance restrictions are typicallylimited by a ±5% tolerance so that 5V logic systems willoperate correctly.) However, a simple LC input filter caneliminate the need for large input bulk capacitance at theregulator 5V supply for additional system cost savings.
Figure 221 shows a more complete system configurationthat incorporates current limiting and current limit time-out with latch-off. Current limit is incorporated for noadditional system cost by manufacturing the current limitresistor from a Kelvin-sensed section of PC board trace. Inthis example, current limit is set to 7A. A capacitor from theSHDN pin to ground sets a fault condition time-out periodthat latches off the drive to the external MOSFET if thetime-out period is exceeded. The regulator is reset bypulling the SHDN pin low. The output voltage in thisapplication is set to 3.3V. The ±5% tolerance permitted in3.3V systems translates to a ±165mV output-voltage
Figure 220. Transient Response for0.2A–5A Output Load Step
1
2
3
4
8
7
6
5
SHDN
VIN
GND
OUT
IPOS
INEG
GATE
COMP
C21µF
C5330µF
VIN5V
GND1575/77 TA01
VOUT3.5V5A
R25Ω
R17.5k
VIN 12V LT1575-3.5
C41000pF
FOR T > 45°C:C6 = 24 × 1µF X7RCERAMIC SURFACE MOUNT CAPACITORS.
PLACE C6 IN THE MICROPROCESSOR SOCKET CAVITY
FOR T ≤ 45°C:C6 = 24 × 1µF Y5VCERAMIC SURFACE MOUNT CAPACITORS.
*Q1IRFZ24
+
C310pF
C6*24µF
2A/DIV
50mV/DIV
0
200µs/DIVI = 0.2A to 5A
Figure 219. UltraFast Transient Response 5V to 3.3V,Low Dropout Regulator
Application Note 84
AN84-110
1
2
3
4
8
7
6
5
SHDN
VIN
GND
OUT
IPOS
INEG
GATE
COMP
C21µF
C5330µF
5V
GND
1575/77 TA12
VOUT3.3V5A
R25Ω
R3*0.007Ω
R13.9k
12V LT1575-3.3
C41500pF
C11µFRESET
R3 IS MADE FROM “FREE” PC BOARD TRACE
C6 = 12 × 1µF X7R CERAMIC SURFACE MOUNT CAPACITORS.
PLACE C6 IN THE MICROPROCESSOR SOCKET CAVITY
*
**
Q2VN2222L
Q1IRFZ24
+
C310pF
C6**12µF
+ C2330µF6.3V
C30.33µF
C61500pF
R23.9k
R31.21k
R41.21k
R72.1k
R13.9Ω
R53.9Ω
R81.6k
C9 TOC20*1µF
Q1IRFZ24
VI/O3.3V
C510pF C8
1000pF
R67.5kC7
10pF
AN69 F06
FAULT RESET
C40.1µF
12V
C1330µF
6.3V
INPUT5V
+
Q2IRFZ24
*X7R CERAMIC 0805 CASE
C21 TOC44*1µF
VCORE2.8V
1
2
3
4
16
15
14
13
IPOS
INEG
GATE
COMP
SHDN
VIN
GND
FB
1/2 LT15775
6
7
8
12
11
10
9
IPOS
INEG
GATE
COMP
SHDN
VIN
GND
FB
1/2 LT1577
tolerance. This permits a 50% reduction in the number ofceramic capacitors required from twenty-four to twelve.Loop compensation is adjusted accordingly.
Figure 222 shows an application circuit using the LT1577,a dual regulator. All functions for each regulator areidentical to those of the LT1575. One section is configuredfor a 3.3V output and the other section is configured for a
2.8V output. This circuit provides all the power require-ments for a split-plane system: 3.3V for the logic supplyand 2.8V for the processor-core supply. Note that bothSHDN pins are tied to a common time-out capacitor. Ifeither or both regulators encounter a fault condition, bothregulator sections are latched off after the time-out periodis exceeded.
Figure 221. 5V to 3.3V Regulator
Figure 222. LT1577 Dual Regulator for Split-Plane Systems
Application Note 84
AN84-111
LT1579 BATTERY-BACKUP REGULATOR PROVIDESUNINTERRUPTIBLE POWERby Todd Owen
Introduction
Designed for a multitude of applications, the LT1579 is adual input, single output, low dropout regulator thatprovides an uninterruptible output voltage from two inde-pendent input voltage sources on a priority basis. Allpower supplied to the load is drawn from the primary input(VIN1) until the device senses that the primary source isfailing. At this point, the LT1579 smoothly switches fromthe primary input to the secondary input (VIN2) to maintainoutput regulation. The LT1579 is capable of providing300mA from either input at a dropout voltage of 0.4V. Totalquiescent current is 50µA: 45µA from the primary inputsource, 2µA from the secondary input source, and anadditional 3µA from the higher voltage of the two.
Circuit Examples
The basic application of the LT1579 is shown in Figure223. It uses two independent voltage sources for theinputs. These voltage sources may be batteries, walladapters or any other DC source. The low-battery com-parators are configured to give a low output if either inputvoltage drops below 5.5V. The trip points can be adjustedby changing the values of the divider resistors (R1 and R2for LB1, R3 and R4 for LB2). All logic outputs (LBO1,
1579_03.eps
5V300mAR1
2.7M 100k 100k 100k 100k1µF4.7µF
0.01µF
6V
R21M
R32.7M1µF6V
R41M
TO POWER MANAGEMENT
++
+
LT1579-5
GND
VIN1
LBI1
VIN2
LBI2
OUT
LBO1
LBO2
BACKUP
DROPOUT
SS
SHDNBIASCOMP
A6V
6V
5V
5V
5V4.8V
100mA
0
VIN1
VIN2
IIN1
100mA
0
0
1
1
01
0
0
1
IIN2
LB01
BACKUP
LB02
DROPOUT
VOUT
B C D E
1579_04.eps
Figure 223. LT1579 Basic Application
Figure 224. Basic Application Timing Diagram
Application Note 84
AN84-112
LBO2, BACKUP and DROPOUT) are open-collector out-puts that require an external pull-up resistor. They arecapable of sinking 20µA at a maximum output voltage of0.32V, which is useful for driving both CMOS and TTLlogic families. For driving LED’s, all logic outputs can sink5mA at a maximum output voltage of 1.2V.
Figure 224 is the timing diagram for the basic circuit. Notime scale is shown for the timing diagram because actualdischarge rates are a function of the load current and thetype of batteries used. The timing diagram is meant as atool to help in understanding the LT1579’s basic operation.
Five milestones are noted on the timing diagram. Time Ais where the primary input voltage drops enough to trip thelow-battery detector, LB1. The trip threshold for LB1 is setat 5.5V, slightly above the dropout voltage of the primaryinput. At time B, the BACKUP flag goes low, signaling thebeginning of the transition from the primary source to thesecondary source. Between times B and C, the inputcurrent makes a smooth transition from VIN1 to VIN2. By
time C, the primary battery has exhausted most of itsuseful charge. The primary input will still deliver a smallamount of current to the load, diminishing as the primaryinput voltage drops. By time D, the secondary battery hasdropped to a low enough voltage to trip the second low-battery detector, LB2. The trip threshold for LB2 is also setat 5.5V, slightly above where the secondary input reachesdropout. At time E, both inputs are low enough to cause theLT1579 to enter dropout, with the DROPOUT flag signal-ing the impending loss of output regulation.
Some interesting things can be noted on the timingdiagram. The amount of current available from a giveninput is determined by the input/output voltage differen-tial. As the primary voltage drops, the amount of currentdrawn from the input also drops, slowing discharge of thebattery. Dropout-detection circuitry will maintain the maxi-mum current draw from the input for the given input/output voltage differential, based on the impedance of thepass transistor. In the case shown, this causes the currentdrawn from the primary to approach zero, although it
BIASCOMP
1579_05.eps
LT1579-5
GND
VIN1 VOUT5V/300mA
LBI1
SS
LBO2
SHDN
LBO1
R22.7M
R91.5M
RESET
C11µF
C21µF
C34.7µF
C40.01µF
R11M
R31M
R410M
D1D2
R101M
MAIN GOOD
IN1
VIN2
LBI2
R62.7M
D45.1V1N751A
D1 TO D3 = 1N4148
C50.1µF VCC
GND
R51M
R8330k
R71M
D3IN2
OUT
BACKUP
1/474C02 1/4
74C02
1/474C02
DROPOUT NC
Figure 225. Added SR Latch Shuts the LT1579 Off when Both Low-Battery Detectors are Tripped
Application Note 84
AN84-113
never reaches that point. Note that the primary begins tosupply significant current again when the secondary inputdrops low enough to cause a loss in output regulation. Thisoccurs because the input/output voltage differential of theprimary input increases as the output voltage drops. TheLT1579 will automatically maximize the power drawnfrom the inputs to maintain the highest possible outputvoltage.
A final circuit example is shown in Figure 225. This circuithas a few notable changes from the basic application.First, the Secondary Select pin is connected directly toLBO1. When the primary input voltage drops below thethreshold level for LB1, the comparator output will pull theSecondary Select pin low. This forces the device to switchcompletely over to the secondary input, limiting the dis-charge voltage of the cells. Second, the logic gates usedform an SR latch. When both batteries are below thethreshold level for their respective comparators, the latchwill be set, forcing the part into shutdown. The latch isreset by pulling up on the RESET node, allowing the partto come out of shutdown.
The series resistance of a battery can cause its terminalvoltage to rise as its current decreases. This effect canreset the low-battery detector and cause the LT1579 tooscillate between the primary and secondary inputs. Tocombat this, the low-battery comparators have up to18mV of built-in hysteresis at the input to the comparator(LBI1, LBI2). The hysteresis is determined by the amountof load current on the comparator output. At no load, thecomparator hysteresis is zero, increasing to a maximumof 18mV for load currents above 20µA. For the pull-upresistor shown, load current on the output of the compara-tor is 5µA, so hysteresis will be 5mV. With the valuesshown for resistor divider R2/R3, this translates to 19mV
of hysteresis at the primary input of the LT1579. Additionalhysteresis can be added by connecting D1 and R4. Thevalues shown will give an additional 200mV of hysteresis.
When LBO1 and LBO2 are high impedance and either inputis greater than 6.5V, the logic-flag voltages can be abovethe maximum voltage rating. Internal clamps on the logicflags limit the output voltage to approximately 6.5V andthe pull-up resistor values shown will limit the current intothe logic flags to less than the maximum current rating.
Conclusion
The LT1579 can provide a continuous regulated outputvoltage to critical circuits from any of a number of differentinput sources. It will provide up to 300mA of outputcurrent at a dropout voltage of 0.4V. Should the primaryinput fail, the device switches seamlessly to the secondaryinput, maintaining output regulation. A single error amplifiercontrols both output stages so regulation remains tightregardless of which input is providing power. The LT1579can handle instantaneous removal of either one of itsinputs without losing regulation. System power manage-ment is aided by two status flags, which provide informa-tion about which input is providing power and signal theloss of output regulation. Two independent low-batterycomparators can be used to monitor input voltages. Also,an external pin can be used to force the switch to thesecondary input. Total quiescent current of the LT1579 is50µA, dropping to a mere 7µA in its low power shutdownstate. Internal circuitry guards against a number of faultconditions, including current limit, thermal limit and re-verse voltages, protecting sensitive circuitry and inputs.Whether the application is simple or complex, the LT1579is truly a “smart” regulator.
Application Note 84
AN84-114
Battery Chargers
THE LT1511 3A BATTERY CHARGER CHARGESALL BATTERY TYPES, INCLUDING LITHIUM-IONby Chiawei Liao
The LT1511 current mode PWM battery charger is thesimplest, most efficient solution for fast charging modernrechargeable batteries, including lithium-ion (Li-Ion),nickel-metal-hydride (NiMH) and nickel-cadmium (NiCd)that require constant-current and/or constant-voltagecharging. The internal switch is capable of delivering 3ADC current (4A peak current). Full charging current can beprogrammed by resistors or by a DAC to within 5%, andthe trickle charge current can be programmed to 10%accuracy. With 0.5% reference voltage accuracy, theLT1511 meets the critical constant-voltage chargingrequirement for lithium cells.
The LT1511 is equipped with a voltage-control loop tocontrol charging voltage and a current-control loop tocontrol charging current. A third control loop is providedto regulate the current drawn from the AC adapter. Thisallows simultaneous equipment operation and batterycharging without overloading the adapter. Charging current
is reduced to keep the adapter current within specifiedlevels.
The LT1511 can charge batteries ranging from 1V to 20V.Ground sensing of current is not required and the battery’snegative terminal can be tied directly to ground.
LT1511 Applications
Lithium-Ion Charging
The 3A lithium battery charger (Figure 226) chargeslithium-ion batteries at a constant 3A until the batteryvoltage reaches a limit set by R3 and R4. The charger willthen automatically go into a constant-voltage mode, withthe current decreasing to zero over time as the batteryreaches full charge. This is the normal regimen for lithium-ion charging, with the charger holding the battery at “float”voltage indefinitely. In this case no external sensing of fullcharge is needed.
Current though the R3/R4 divider is set at 15µA to mini-mize battery drain when the charger is off. The inputcurrent to the OVP pin is 3nA and this error can beneglected.
SW
BOOST
COMP1
CLN
UV
PROG
VCOVP SENSE BAT
C11µF
RS4 ADAPTER CURRENT SENSE
R7500Ω
R5† UNDERVOLTAGE LOCKOUT
R65k
DINVIN (ADAPTER INPUT)11V TO 25V
VBAT
10µF
CPROG1µF
CIN*10µF
300Ω RPROG4.93k1%
0.33µF
1k
0.47µF
RS3200Ω1%
RS2200Ω1%
L1**10µH
D21N4148 200pF
RS10.033Ω
BATTERY CURRENTSENSE
R3390k0.25%BATTERY VOLTAGE SENSE
R4162k0.25%
50pF
COUT22µFTANT
4.2V
4.2V
++
LT1511
NOTE: COMPLETE LITHIUM-ION CHARGER,NO TERMINATION REQUIRED. RS4, R7 AND C1 ARE OPTIONAL FOR IIN LIMITING *TOKIN 25V CERAMIC SURFACE MOUNT **10µH COILTRONICS CTX10-4 †CONSULT LT1511 DATA SHEET FOR R5 VALUE
VCC TO MAIN SYSTEM POWER
SPIN
D1MBR340
GND CLP
2 Li-Ion+
++
Figure 226. 3 Amp Lithium-Ion Battery Charger
Application Note 84
AN84-115
With divider current set at 15µA, R4 = 2.465/15mA = 162kand
R1 =VOUT – 1.245
1.245R2
+ (3 × 10–7)
where VOUT = battery float voltage
Lithium-ion batteries typically require float-voltage accu-racy of 1% to 2%. The accuracy of the LT1511 OVP voltageis ±0.5% at 25°C and ±1% over full temperature. Thisleads to the possibility that very accurate (0.1%) resistorsmight be needed for R3 and R4. Actually, the temperatureof the LT1511 will rarely exceed 50°C in float modebecause charging currents have tapered off to a low level,so 0.25% accuracy resistors will normally provide therequired level of overall accuracy.
Nickel-Cadmium and Nickel-Metal-Hydride Charging
The circuit in the 3A lithium battery charger (Figure 226)can be modified as shown in Figure 227 to charge NiCd orNiMH batteries. Two-level charging is needed; 2A when Q1is on and 200mA when Q1 is off. For 2A full current, thecurrent sense resistor (RS1) should be increased to 0.05Ω,so that enough signal (10mV) will be across RS1at 0.2Atrickle charge to keep charging current accurate.
For a two-level charger, R1 and R2 are found from
R1 = R2 =(2.465)(4000) (2.465)(4000)ILOW IHI − ILOW
PWM
RPROG4.7k
300Ω
PROG
CPROG1µF
Q1VN2222
5V0V
LT1511
IBAT = (DC)(3A)
All battery chargers with fast charge rates require somemeans to detect the full-charge state in the battery in orderto terminate the high charging current. NiCd batteries aretypically charged at high current until temperature rise orbattery voltage decrease is detected as an indication ofnearly full charge. The charging current is then reduced toa much lower value and maintained as a constant tricklecharge. An intermediate “top off” current may be used fora fixed time period to reduce 100% charge time.
NiMH batteries are similar in chemistry to NiCd but havetwo differences related to charging. First, the inflectioncharacteristic in battery voltage as full charge is ap-proached is not nearly as pronounced. This makes it moredifficult to use dV/dt as an indicator of full charge, andtemperature change is more often used, with a tempera-ture sensor in the battery pack. Second, constant tricklecharge may not be recommended. Instead, a moderatelevel of current is used on a pulse basis (1% to 5% dutycycle) with the time-averaged value substituting for aconstant low trickle.
If overvoltage protection is needed, R3 and R4 should becalculated according to the procedure described in lithium-ion charging section. The OVP pin should be grounded ifnot used. When a microprocessor DAC output is used tocontrol charging current, it must be capable of sinkingcurrent at a compliance up to 2.5V if connected directly tothe PROG pin.
Figure 227. 2-Step Charging
Application Note 84
AN84-116
LT1512/LT1513 BATTERY CHARGERS OPERATEWITH INPUT VOLTAGES ABOVE OR BELOWTHE BATTERY VOLTAGEby Bob Essaff
Introduction
The LT1512 and LT1513 form a unique family of constant-current, constant-voltage battery chargers that can chargebatteries from input voltages above or below the batteryvoltage. This feature can help simplify system design andadd product flexibility by allowing battery charging frommultiple sources, such as a wall adapter, a 12V automotivesystem or a 5V power supply, all with the same circuit. Theconstant-current, constant-voltage architecture makes theLT1512 and LT1513 well suited for charging NiCd, NiMH,lead-acid or lithium-ion batteries.
Both devices are current mode switching regulators thatoperate at a fixed frequency of 500kHz. Product featuresinclude a ±1% reference-voltage tolerance, 2.7V minimuminput voltage, easy external synchronization and 12µAsupply current in shutdown mode. The LT1512 and LT1513also include low loss on-chip power switches rated for 1.5amps and 3 amps respectively. High frequency switchingallows the use of small surface mount inductors andcapacitors, and the battery can be directly grounded.
Applications
The LT1512 and LT1513 are specifically optimized to usethe SEPIC converter topology, which is shown in Figure228’s typical application. The SEPIC (single-ended pri-mary inductance converter) topology has several advan-tages for battery-charging applications. It will operate withinput voltages above or below the battery voltage, has nopath for battery discharge when turned off, and eliminatesthe snubber losses of flyback designs. It also has a currentsense point that is ground referred and need not beconnected directly to the battery. The two inductors shownare actually two identical windings on one inductor core,although two separate inductors can be used.
The topology is essentially identical to a 1:1 transformer-flyback circuit except for the addition of capacitor C2,which forces identical AC voltages across both windings.This capacitor performs three tasks: it eliminates thepower loss and voltage spikes usually caused by a flyback-converter’s leakage inductance; it forces the input currentand the current in resistor R3 to be a triangle wave ridingon top of a DC component instead of forming a largeamplitude square wave; and it eliminates the voltagespikes across the output diode when the switch turns on.
When the battery is below its float voltage, set by R1 andR2, the charger is in the constant-current mode. Thesuggested value for R2 is 12.4k. R1 is calculated from:
LT1512CHARGE
SHUTDOWNIFBVC
VIN
L1A*
L1B*
0.5A
1 3
2
8
5•
•4
76
GNDVFB
VSW
WALLADAPTER
INPUT
S/S
C322µF25V
C2**1µF× 2
C50.1µF
*
**†
L1A, L1B ARE TWO 33µH WINDINGS ON ACOMMON CORE: COILTRONICS CTX33-3AVX1206Y2105KAT1ATO CALCULATE R1, R2 VALUES, SEE TEXT
C40.1µF
R424Ω
R1†
R2†
R30.2Ω
C122µF25V
D1MBRS130LT3+
+
INDUCTOR = 33µH
LT1513
LT1512
INPUT VOLTAGE (V)
0
1.21.0
0.80.6
0.40.2
1.4
1.6
2.0
1.8
2.2
2.4
CURR
ENT
(A)
300 5 10 20 2515
SINGLE LITHIUM CELL (4.1V)
SINGLE LITHIUM CELL (4.1V)
DOUBLE LITHIUM CELL (8.2V)
DOUBLE LITHIUM CELL (8.2V)
Figure 228. Battery Charger with 0.5A Output Current Figure 229. Maximum Charging Current
Application Note 84
AN84-117
Charging current in the battery, which also flows throughR3, develops a voltage on the IFB pin. The IFB pin’s 100mVsense voltage sets the programmed charging current toICHG = 100mV/R3. The RC filter formed by R4 and C4smoothes the signal presented to the IFB pin.
Charging current remains constant until the battery reachesits float voltage, at which point the LT1512/LT1513 changesto the constant-voltage mode. In this mode, the chargingcurrent will taper off as required to keep the battery at itsfloat voltage. The circuit’s maximum input voltage ispartly determined by the battery voltage. When the switchis off, the voltage on the VSW pin is equal to the inputvoltage, which is stored across C2, plus the battery volt-age. Both the LT1512 and LT1513 have a maximum inputvoltage rating of 30V and a maximum rated switch voltageof 35V, thereby limiting input voltage to 30V or 35V minusthe battery voltage, whichever is less.
Figure 229 shows the maximum available charging cur-rent for a single-cell or double-cell lithium battery pack.Note that the actual programmed charging current will beindependent of the input voltage if it does not exceed thevalues shown.
Programming the Charge Current
As mentioned earlier, charging current is set by R3, whereICHG = 100mV/R3. The charge current is programmed bychanging the effective value of R3, as shown in Figure 230.In the low charge mode, Q1 is off, setting charge currentto ICHG LOW = 100mV/R3A, or 100mV/2Ω = 50mA. In thehigh-charge mode, Q1 is on, and charge current is ICHG HI= 100mV/R3A + 100mV/(R3B + Q1’s RDS(ON)), or 100mV/2Ω + 100mV/(0.24Ω + 0.04Ω)) = 50mA + 357mA =407mA. Note that Q1’s RDS(ON) is a factor in the high-charge mode, requiring the use of a low RDS(ON) FET.
Off-State Leakage
Charging can be terminated by placing the LT1512/LT1513into shutdown mode. If the battery remains connected tothe charger when in the off state, two leakage paths thatload the battery must be considered.
The first is the 100µA resistor-divider feedback currentthat flows through R1 and R2. This current can be elimi-nated with the addition of a FET, Q1, between R1 and theR2/VFB junction, as shown in Figure 231. In this example,pulling the charge/shutdown input above 3.75V will acti-vate charging and turn on Q1, whereas driving the charge/shutdown input below 0.6V will shut down the LT1512/LT1513 and turn off Q1.
LT1512
IFBVC
VIN
L1A
L1B
1 3
2
8
5•
•4
76
GNDVFB
VSW
VIN
S/S
22µF25V
C21µF× 2
0.47µF
Q1 = SILICONIX Si9410DYC2 = AVX1206Y2105KAT1A
0.1µF
24ΩR3B
0.24Ω
R1
R2
Q1R3A2Ω
C122µF25V
MBRS130LT3
HI CHARGE
LOW CHARGE
CHARGE
SHUTDOWN
+
+
LT1512/LT1513
CHARGE
SHUTDOWN
GNDVFB
S/S
R1
Q1VN2222
BATTERY
R2
R1 =VOUT – 1.245
1.245R2
+ (3 × 10–7)
where VOUT = battery float voltage
Figure 230. 50mA/400mA Programmable Battery ChargerFigure 231. Shutdown-Controlled Disconnect
Application Note 84
AN84-118
Li-Ion BATTERY CHARGER DOES NOT REQUIREPRECISION RESISTORSby LTC Applications Staff
In constant-voltage mode charging, a Li-Ion cell requires4.1V ±50mV. This 1.2% tolerance is tight. In a regulationloop where a voltage divider is compared against a refer-ence, the accuracy is achieved by selecting a 0.7% refer-ence and a voltage divider with 0.25% tolerance resistors.Unfortunately, 0.25% precision resistors cost three timesas much as 1% resistors and have very long lead times.
One solution for moderate volume production involvesadding two 1% resistors and two jumpers to the chargercircuit, as shown in Figure 232. The jumpers are removedas necessary to bring the constant voltage to the requiredaccuracy of 1.2%.
The charger selected for this example is the LT1510 andthe number of Li-Ion cells in the battery is three. Select avalue for R4 (20k) and calculate the values for resistorsR1, R2 and R3 using the equations in Figure 232. K is therelative change required for a circuit with all its tolerancesin one direction. For example, in the case of a 0.5%reference and two 1% resistors, the total tolerance is2.5%. In order to bring it back to 1.2%, the percentagechange required is 2.5% – 1.2% = 1.3% and K = 0.013.
The jumpers J1 or J2 need to be opened based on thefollowing:
If VOUT is K/2 below nominal, remove J1.
If VOUT is K/2 above nominal, remove J2.
The second leakage path to consider is in the output diode,D1 (Figure 228). When the charger is in the off state, theoutput diode sees a reverse voltage equal to the batteryvoltage. Though the Schottky diode reverse leakage maytypically be only 10µA, its guaranteed specifications are
much worse, up to 1mA. One solution is to change theoutput diode to an ultra-fast silicon diode, such as anMUR-110. The higher forward voltage of the silicon diodewill decrease the circuit’s efficiency, but these diodes havereverse leakage specifications below 5µA.
BAT
–
+
+–
–
+J1R2
R1
VOUT = 12.3V
R3 =
OVP3-CELLLi-IONBATTERY
VREF2.465V
VIN
LT1510CONSTANT VOLTAGE/CONSTANT CURRENTBATTERY CHARGER
J2R3
R4
R1 = R4 × VOUT – VREFVREF
R2 = (R1+R2) × K
1 – (1 – K) VREFVOUT
R4 × K
TO CALCULATE K, SEE TEXT
SW
BOOST
GND
SENSE
VCC
PROG
VC
BAT
3.83k
16V TO 28V
R5100k
R1
R2
Q3VN2222
*TOKIN OR MARCON CERAMIC SURFACE MOUNT** COILTRONICS CTX33-2
C10.22µF
CIN*10µF
L1**33µH LT1510
D11N5819
D31N5819
D21N914
OVP
COUT22µFTANT
J1
J2
+–
0.1µF1k
1µF300Ω
R3
R4
+
+ 3-CELLLi-IONBATTERY
Figure 232. R2, R3, J1 and J2 Eliminatethe Need for Precision Resistors Figure 233. 3-Cell Li-Ion Charger without Precision Resistors
Application Note 84
AN84-119
LT1510 CHARGER WITH –∆V TERMINATIONby LTC Applications Staff
Any portable equipment that requires fast charge needsproper charge termination. Commonly, a LT1510 con-stant-voltage, constant-current type charger controlled bya microcontroller is used. Sometimes, however, a micro-controller is not available or is not suitable for fast-chargetermination.
When fast charging NiCd batteries with constant current,the internal battery temperature rises toward the end of thecharge. Since the temperature coefficient of NiCd is nega-tive, the temperature rise causes the battery voltage todrop. The drop can be detected and used for termination(called –∆V termination). The circuit in Figure 234 is asolution for a 3-cell (Panasonic P140-SCR) NiCd batterycharger with –∆V termination.
U1 in Figure 234 is programmed by resistor R2 for aconservative charge current of 0.8A, which is 0.57C.Typical fast-charge current is 1C. (The boldfaced C repre-sents a normalization concept used in the battery industry.A C rate of 1 is equal to the capacity of the cell in ampere-hours, divided by 1 hour. Since the capacity of the P140-SCR is 1.4 ampere-hours, C is 1.4 amperes.)
To determine the voltage droop rate, the battery wasconnected to an LT1510 charger circuit programmed fora 0.8A constant-current. The data was plotted as voltageversus time and the results are shown in Figure 235. Thevoltage slope is calculated to be –0.6mV/s. After thebattery voltage dropped 300mV from the peak of 4.93V(100mV per cell), the charger was disabled.
At the heart of the circuit in Figure 234 is U3, a sample-and-hold IC (LF398). For every clock pulse at pin 8, the outputof U3 (pin 5) updates to the input level on pin 3. When thebattery voltage drops, the input to U3 also drops. If theupdate step at the output of U3 is sufficiently negative, U2Blatches in the high state and Q1 turns on. Q1 terminates thecharge by pulling down the LT1510’s VC pin, and therebydisabling it.
U2A and the associated passive components smooth,amplify and level shift the battery voltage. The timer (U4)updates the hold capacitor (C8) every fifteen seconds. Thetimer signal stays high for 7ms, sufficient time for the holdcapacitor to be charged to the input level. U2B and theassociated parts form a latch that requires a momentarynegative voltage at pin 6 to change state. R15 supplies thenegative feedback and Q2, R16, R17 and C10 reset thelatch on turn-on.
U3’s output voltage droops at a rate proportional to thehold capacitor’s internal leakage and the leakage current atpin 6 (10pA typical). This droop is very low and does notaffect the operation of the circuit.
The minimum negative battery voltage slope required totrigger termination (–dV/dT) is 0.3mV/s. It can be calcu-lated from:
–dV/dT = VTRIG/(TCLK × GU2A) where:VTRIG is the trigger voltage of U2B,VTRIG = VREF × R12/(R11 + R12) = 5 × 1/101 = 49.5mVVREF = 5VTCLK is the clock period, 15 seconds,GU2A is the gain of the first stage, = R8/(R4 || R5) = 11
The following values were calculated: R1 = 20k, R2 =324Ω, R3 = 80.6Ω and R4 = 4.99k.
The voltage below which J1 should be opened is 12.34V– 1.3%/2 = 12.22V.
The voltage above which J2 should be opened is 12.34V+ 1.3%/2 = 12.42V.
The complete schematic can be seen in Figure 234. Q3 isoff when the charger is not powered, preventing currentdrain from the battery through the voltage divider. R5, a100k resistor, isolates the OVP pin from any high fre-quency noise on VIN. The charger in Figure 233 is pro-grammed for 1.3A constant current.
Application Note 84
AN84-120
+
–+
–+
–
+
–
12V
L1*30µH
C10.22µF
C110.22µF
CR21N5819
CR31N5819
LT1029CZ
C2†10µF
C1022µF
C70.1µF
C522µF25V
C60.1µF
C81µFECQV1HIOSJLPANASONIC
34
U2ALT1013 INPUT
OUTPUT
LOGICREFERENCE
LOGICCLK
2 85
7
U3LF398
1 3
6
5
8
7
64
HOLDCAPACITOR
2 1OFFSET
B1***
C31µF
C12
0.01µF
R19100k
R18, 100k
CLK
R20100k
C90.1µF
SW VCC1VCC2
PROGBOOST
GND
GND
U1LT1510
5VREFN/C
R1300
R8100k
30k
C40.1µF
VC
R31k
BAT
R6100k
R2110k
R5100k
R410k
5VREF
R11100k
R1410k
R710
150Ω
R15100k
R16100k
R17100k
Q22N3904
Q12N3904
U4CD45368
R930.1k
R1030.1k
NOTES: * L1 IS COILTRONICS CTX 33-2** SOLDER TO GROUND PLANE FOR HEAT DISSIPATION
*** B1 IS A NiCd 3 CELL PANASONIC P140-SCR† C2 IS A TOKIN OR MARKON CERAMIC SURFACE MOUNT
R121k
R1310k
R26.19k
SENSE
CR11N914
**1, 7, 8, 9, 10, 16
U2BLT1013
4
6
7
8
3
5
2
1
13
11
10
9
14
12
15
16
2
3
4
6
1514
13
12
11
+
+
+
The circuit in Figure 234 was built and connected to asystem that discharges the battery to 3V after termination,at constant current of 0.8A. Once the battery drops to 3V,the system reenables charging, and thus the completesystem repeats charge/discharge cycles indefinitely. Theduration of 70 charge discharge cycles was recorded. Thefollowing is condensed data from the test:
1. Average Charge Time: 2:00:55 Hours
2. Standard Deviation of Charge Time: 5:37 Minutes
3. Average Discharge Time: 1:59:14 Hours
4. Standard Deviation of Discharge Time: 48 Seconds.
The ratio of standard deviation of charge time to averagecharge time proves that the charger has good repeatability.However, the ratio of standard deviation of discharge timeto average discharge time shows that the charge level at
the time of termination is very consistent because thedischarge time at constant current is a better measure ofcharge level than charge time. A secondary terminationmethod, such as time, battery temperature, or the like, isalso recommended.
TIME (MIN)
4.2
4.4
4.3
4.5
4.7
4.6
4.8
4.9
5.0
BATT
ERY
VOLT
AGE
(V)
7:30 15:00 30:0022:30
END OF0.8A CHARGE
NEGATIVEVOLTAGE SLOPE
Figure 234. Schematic Diagram: 3-Cell NiCd Charger with –∆V Termination
Figure 235. Voltage-Droop Rate, 3-Cell NiCd Battery
Application Note 84
AN84-121
CONSTANT-VOLTAGE LOAD BOXFOR BATTERY SIMULATIONby Jon Dutra
Linear Technology has developed many new switcher-based battery charger ICs. Testing accuracy, regulationand efficiency in the lab with a battery load is inconvenientbecause the terminal voltage of a battery constantly changesas it is being charged. If much testing is to be done, a largesupply of dead batteries will be needed, since one set ofcells can quickly become overcharged. This articledescribes an active load circuit that can be used to simu-late a battery in any state of charge. The battery simulatorprovides a constant-voltage load for a battery-chargingcircuit, independent of applied charging current. Thesimulator’s impedance is less than 500mΩ at all reason-able input frequencies. Best of all, the simulator can neverbe overcharged, allowing long-term testing and debug-ging of a charger system without the possibility of batterydamage.
Circuit Operation
The simulator (Figure 236) uses an LT1211 high speed,single-supply op amp to drive the base of a high gain PNPtransistor-stage active load. Power for the LT1211—aportion of the charging current—is supplied through a
diode so the op amp and reference can survive briefperiods of zero charging current. The op amp is configuredfor a DC gain of four, so the voltage on its noninvertinginput is one fourth of the voltage that the load box is set to.With S1 open, the load-voltage adjust range will be from10V to 20V, and with S1 closed it will be approximately3.5V–10V. Low voltage operation could be improved byreplacing the top LT1004-2.5 with an LT1004-1.2 andreducing R1, the reference bias resistor, to 1k. The 510Ωand 1.1k resistors are required for high frequency stability;they suppress a 1MHz oscillation. The 1N5400 diode and4-amp fuse protect the circuit from reverse voltages.
Results
The battery simulator circuit has been tested “swallowing”currents from 30mA to 3A with the output voltage essen-tially unchanged. When simulating a battery, the voltageadjust can be increased until the charger thinks the batteryis fully charged and reduces the current into the simulator.Conversely, as the voltage is adjusted down, the batterycharger may think the battery is becoming discharged andincrease the current into the simulator.
Figure 237 shows the circuit’s capacity for current absorp-tion at two voltages, 5V and 15V, from 50mA to 3 amps.
–
+
+
+LT1004-2.5
2.5V OR 0V
5.0V OR 2.5V 100kΩ10 TURNPOT
0.033µF
0.033µF10k, 1%
0.5Ω5W 4A FUSE30k, 1%
510Ω
R1 10k
1.1kΩ1N5817 ORBAT-85
100µF25V
270µF25V
IN+
IN–
Q1*2N6667
1N5400
1/2LT1211
LT1004-2.5
S1 CLOSED ≥ 0 TO 10V RANGES1 OPEN ≥ 10V TO 20V RANGEALL RESISTORS 5% UNLESS NOTED
S1
Q1 DISSIPATES MOST OF THE POWER,MOUNT ON AN ADEQUATE HEAT SINK
*CURRENT (A)
14.7
15.1
15
14.9
14.8
15.2
15.3
15.4
15.5
4.7
5.1
4.9
5
4.8
5.2
5.3
VOLT
AGE
(V)
30 1.50.5 2 2.51
5V
15V
Figure 236. Schematic Diagram of the Battery SimulatorFigure 237. Current Absorption Capacityof the Battery Simulator at 5V and 15V
Application Note 84
AN84-122
HIGH EFFICIENCY, LOW DROPOUTLITHIUM-ION BATTERY CHARGER CHARGESUP TO FIVE CELLS AT 4 AMPS OR MOREby Fran Hoffart
Introduction
Rechargeable lithium batteries feature higher energy den-sity per volume, higher energy density per weight andhigher voltage per cell than any of the competing batterychemistries. For these reasons, manufacturers of portableequipment are adopting the lithium-ion rechargeable bat-tery as the battery of choice for high performance portableequipment. Lighter weight and increased operating timebetween charges are important features that customerswant and need from portable products.
Increased demands from laptop computers have forcedmanufacturers to use multiple cells in a combination ofseries and parallel configurations. Paralleling cells in-creases the amount of current that can be drawn from thebattery and/or increases the operating time betweencharges, but it also increases the current requirements ofthe charger.
Higher Charge Currents
Paralleling cells, regardless of cell chemistry, requiresrelatively high charge currents to bring the battery up tofull charge in a short period of time. When charging needsexceed the 3A maximum rating of the LT1511 or LT1513,the circuit shown in Figure 238 can provide much highercurrent solutions, and very high efficiency. This circuituses the LTC1435 and LT1620 in a charger that delivers 4Aor more with exceptional efficiency and low dropoutvoltage (Figures 238 and 239).
+
AVG8
R3(SEE TEXT)
1620_01.eps
SGND
SENSE+
VOSENS
PGND
BG
BOOST
INT VCC
SW
TG
5 4
6
10
VCC
IN–LT1620CMS8
6
4
IN+ 5
SENSE
PROG
RPROG = 21kFOR 4A
GND
IOUT1
7
IPROG
3
2 11
15
12D2 D1
C6, 0.33µF
C7, 4.7µF
C5, 0.1µF
Q2
R2, 0.1%(SEE TEXT)
RSENSE0.02Ω
IBATT
C322µF
C8100pF
Q1
L127µH
+VIN
C1, C222µF ×235V TANT
C40.1µF
C10100pF
C120.1µF
C141000pF
R51k
C130.033
C180.1µF
C150.1µF
R64k
C160.33µF
R71.5M
SHUTDOWN INPUT(SD = 0V)
14
16
13VIN
LTC1435CG
SFB
SENSE–
ITH
COSC
RUN/SS
8
7
3C1156pF
1
2
L1 = CTX27-4, COILTRONICSQ1, Q2 = Si4412DY, SILICONIXD1, D2 = CMDSH-3, CENTRAL
C1, C2 = 22µF,35V, AVX TPS SERIESC3 = 22µF, 25V, AVX TPS SERIES
C17, 0.01µF
C9, 100pF
+
CHARGE CURRENT (A)
80
90
85
95
100
EFFI
CIEN
CY (%
)
5
1620_02.eps
0 1 2 3 4
VBATT = 16.8V
VBATT = 12.8V
VIN = 24V
Figure 238. Complete Schematic of the High Efficiency, 4A, Constant-Voltage/Constant-CurrentCharger Using All Surface Mount Components, with a Circuit Board Area of 1.5in2
Figure 239. Charger Efficiency for 3- and 4-Cell Applications
Application Note 84
AN84-123
The LT1435 Switching Regulator Controller
The LTC1435 is a step-down current mode switchingregulator controller designed to drive two external N-channel power MOSFETs. Operating from input voltagesbetween 3.5V and 36V, this device includes a program-mable switching frequency, synchronous rectification,Burst Mode™ operation and a 99% maximum duty cyclefor low dropout voltage. Additional features include a 1%tolerance output voltage (adjustable between 1.2V and
9V), programmable soft start, logic-controlled micro-power shutdown and a secondary feedback control pin.Because external MOSFET switches are used, the maxi-mum output load current is determined by the currentcapabilities of the selected FETs.
The LTC1435 as a Battery Charger
The low dropout voltage, high current capability and highefficiency of the LTC1435 switching regulator would seemto make it an appropriate choice for high current batterychargers, but it has several limitations. The absolutemaximum output voltage of 10 volts allows only twoseries-connected lithium cells to be charged and theoutput current is not readily programmable.
Introducing the LT1620
The LT1620 is an IC designed to be used with a currentmode PWM controller (such as the LTC1435 and similarproducts) to increase the output voltage range and opti-mize the circuit for battery charging applications. Usedtogether, these two products overcome the voltage and
CHARGE CURRENT (A)
0
1.0
0.5
1.5
2.0DR
OPOU
T VO
LTAG
E (V
)
5
1620_03.eps
0 1 2 3 4
VBATT = 16.8VIPROG = 200µA
CONSTANT CURRENT PROGRAMMED FOR 4A
+
1620_04.eps
INT VCC
SENSEITH
INVCC
VOSENS
BG
TG
LT1620PROG
IOUT
SENSE
AVG
Q2
IBATT
Q1
L1
R2
R3
R6
PROGRAMCONSTANTCURRENT
RPROG
RSENSE
+VIN
CSS
SHUTDOWN INPUTVCC
LTC1435
GND
C15
COSC
RUN/SS
PROGRAMCONSTANTVOLTAGE
+
Figure 240. Charger Dropout Voltage vs Charge Current
Figure 241. Simplified Diagram of the Constant-Voltage/Constant Current Charger
Application Note 84
AN84-124
current programming limitations previously mentioned,to produce a high current, high performance constant-voltage/constant-current battery charger for lithium-ionand other battery types.
How They Work Together
To understand how the two parts work together, a briefreview of the LTC1435 operation is necessary. See Figure241. During each cycle of operation, the series MOSFETswitch Q1 is turned on by the LTC1435 oscillator (Q2 isoff). This causes a current to begin ramping up in inductorL1. When the current in L1 reaches a peak level determinedby the voltage at the ITH pin, Q1 is turned off and thesynchronous MOSFET Q2 is turned on, causing the cur-rent in L1 to ramp down to the level at which it started.Thus, a sawtooth of inductor ripple current is generated,with a peak level set by the voltage on the ITH pin. Thisinductor current is sensed via an external, low value senseresistor in series with the inductor and is used to drive theLTC1435 internal current sense amplifier as the currentmode feedback signal. This current sense amplifier has amaximum common mode voltage limit of 10V, whichlimits the maximum output voltage to 10V.
Enter the LT1620. The LT1620 also contains a currentsense amplifier, which has a common mode range that
extends up to 28V. This amplifier is used to level shift thedifferential sense voltage, which is riding on the batteryvoltage, and reference it to the internal 5V VCC voltagegenerated by the LTC1435. This level-shifted signal isused to drive the LTC1435 current sense pins, thusproviding current mode feedback for the constant-voltagefeedback loop. This signal is also used to control theconstant output current feedback loop, as explained be-low.
Constant Charge Current
The LT1620 also provides a simple method of accuratelyprogramming the constant-current output. Sinking anadjustable current from the PROG pin to ground controlsthe charge current from zero current to maximum current.This program current can be derived from a variety ofsources, such as a single resistor to ground or the outputof a DAC.
The constant-current feedback loop operates as follows.With a discharged battery connected to the charger, andassuming that the battery voltage is less than the floatvoltage programmed by R2 and R3, the error amplifier inthe LTC1435 begins pulling up on the ITH pin. This in-creases the peak inductor current in an effort to force thebattery voltage to be equal to the programmed voltage. By
RPROG
PROG
800mV
2.5kX10
AVG PIN
CAVG
40mV OFFSETIOUT VCC
+5V
+5V
gm
SENSE
LT1620
IN–
IBATT
CURRENTSENSEAMPLIFIER
IN+
RSENSE
80mV CHARGECURRENT
INDUCTORCURRENT
EXTERNAL AVERAGINGCAPACITOR, C15
X1
ITH INT VCC SENSE+
LTC1435
SENSE–
4.2V
R6
IPROG
≈800mV
1620_05.eps
Figure 242. Simplified Digram of Constant-Current Control Loop
Application Note 84
AN84-125
limiting the voltage on the ITH pin, the peak inductorcurrent and the average output current can be controlled.The ITH pin has an internal 2.4V clamp that sets the peakinductor to its maximum level. This 2.4V clamp providessome degree of current regulation, but the average batterycurrent will vary considerably as a result of dependence oninductor ripple current and LTC1435 parameter varia-tions. By adding the LT1620 to the circuit, the constantcharging current control performance is considerablyimproved. As shown in Figure 242, the signal from thecurrent sense amplifier in the LT1620 is amplified by 10,averaged by CAVG (C15 in Figure 238) and compared to thevoltage drop across R6. This voltage is developed by acurrent, IPROG, flowing through R6. When the voltage atthe LT1620 AVG pin approaches the voltage on the PROGpin, IOUT begins to pull the ITH pin of the LTC1435 down,limiting the peak inductor current and completing theconstant-current feedback loop.
Complete Charger Circuit
The circuit shown in Figure 238 can charge up to fiveseries-connected lithium-ion cells at currents up to 4A.Using low RDS(ON) MOSFET switches for the switch andsynchronous rectifier results in efficiency exceeding 95%and allows all surface mount components to be used,resulting in a design that occupies less than 1.5 in2 ofboard space. This circuit operates at a switching fre-quency of 200kHz and is capable of up to 99% duty cycle;it can operate over a very wide input voltage range, froma minimum input of only 600mV greater than the battery
charging voltage to a maximum of 28V (limited by theMOSFETs).
Constant-voltage charging with better than 1.2% accuracyand constant-current charging with 7.5% accuracy pro-vides almost ideal l ithium-ion battery chargingconditions.
In battery charger designs, an important issue is reversebattery drain current caused by the charger when the inputpower is removed or the charger is shut down, or both. Ifthe battery will remain connected to the charger forextended periods of time, it is important to minimize thisreverse drain current to prevent discharging the battery.The charger can be shut down by using the RUN/SS pin onthe LTC1435. This stops the charging current and resultsin a reverse battery drain current in the tens of microamps.
The LTC1435 and LT1620 have been configured so thatthe battery can remain connected to the charger when theinput power is removed, but because of the inherent bodydiode in the Q1 MOSFET, current can flow from the battery,through the Q1 body diode, to the LTC1435’s VIN pin,keeping it powered up. In this situation, because thecharger is effectively powered by the battery, the reversebattery drain can be several mA, which could discharge thebattery over an extended period. Figure 243 containscircuitry that automatically shuts down the LTC1435 whenthe input power is removed and puts it into a low quiescentcurrent condition. Because the LT1620 is powered fromthe LTC1435 INT VCC pin, it is also turned off.
+
1620_06.eps
TG Q1L1
D3
1M
D4SHUTDOWN
47k CSS
+VIN
Q32N3906
VCC
LTC1435
RUN/SS
D3, D4 = 1N4148
Figure 243. Circuitry that Shuts Down the Charger when Input Power is Removed, Minimizing Reverse Battery Current Drain
Application Note 84
AN84-126
When input power is applied, the charger can still be shutdown with an external signal to the RUN/SS pin. Shutdownoccurs by pulling this pin low; releasing it allows thecapacitor to charge up via the internal 3µA current source,producing a soft start.
By substituting higher current MOSFETs and changingsome component values, much higher charging currentscan be obtained.
Selecting Battery Voltage Programming Resistors
The charging voltage of lithium-ion cells is either 4.1 or 4.2volts per cell, depending on the battery chemistry. Contactthe battery manufacturer for the recommended chargevoltage. To program battery charging voltage (float volt-age) use the following equation (for best accuracy andstability, use 0.1% resistors).
VBATT = VREF
VREF = 1.19V; USE APPROXIMATELY100kΩ FOR R3
R2R3
1 +( )
R2 = R3VBATT VREF )) – 1
Selecting RSENSE
RSENSE is an external, low value resistor that is placed inthe inductor current path to develop a signal representa-tive of the inductor or charge current (IBATT). This signalis used as feedback to control the switching regulatorconstant-voltage and constant-current loops. To mini-mize overall dropout voltage and power dissipation in thesense resistor, a sense voltage of 80mV was chosen torepresent maximum charging current. Use the followingequation to select current sense resistor RSENSE. Themaximum battery charge current (MAX IBATT) must beknown.
RSENSE = 0.08VMAX IBATT
Selecting IPROG
IPROG is a current from the PROG pin to ground that is usedto program the maximum charging current. IPROG can bederived from a resistor to ground, from the output of a DACor by other methods. This program current is generatedusing resistors and the 5V VCC available from the LTC1435.
Refer to the simplified diagram of the constant-currentcontrol loop shown in Figure 242. The DC voltage acrossCAVG is proportional to the average charge current. Thisvoltage drives one input of a transconductance (gm)amplifier. A program voltage (relative to the 5V VCC line)proportional to the desired, or programmed charge cur-rent is applied to the other input of the transconductanceamplifier. This voltage should be selected to be ten timesthe average voltage dropped across RSENSE when thecharger is in a constant-current mode.
If the voltage across CAVG increases to a level equal to thevoltage at the PROG pin, the transconductance amplifierbegins pulling down on the ITH pin of the LTC1435, therebylimiting the peak inductor current, and thus the averagecharge current.
The program voltage needed on the program pin can easilybe generated by two resistors, as shown in Figure 242. Acurrent (IPROG) is generated by these resistors and the 5VVCC voltage. This IPROG develops a voltage across R6,which is used to set the maximum constant charge currentlevel. The circuit is designed for an approximate PROGvoltage of 800mV (don’t exceed the maximum spec of1.25V), referenced to the LT1620 VCC pin. Because of thegain-of-10 amplifier, this corresponds to a typical voltageacross RSENSE of 80mV (with a maximum of 125mV).
The recommended range of resistor values for R6 isapproximately 2kΩ to 10kΩ. With 0.8V across R6, this willresult in program currents (IPROG) between 400µA and80µA.
Application Note 84
AN84-127
The LT1620 was designed to reduce the charging currentto zero under all conditions when the IPROG is set to zero.To ensure that the charging current will always go to zero,an offset was designed into the transconductanceamplifier. In the equations for R6 and RPROGRAM, thisoffset is represented by using 840mV rather than 800mV.
Example:
GIVEN: MAXIMUM IBATT = 4AIPROG = 200µA (FOR MAXIMUM IBATT)
RSENSE = 0.08VMAX IBATT
0.08V4A
= 0.02Ω=
R6 = 0.84VIPROG
0.84V200µA
= 4.2kΩ=
5V – 0.84VIPROG
5V – 0.84V200µA
= 20.8kΩ=RPROG =
Once RPROG and R6 are known, the following equationscan be used to determine RPROG and IPROG for lower IBATTcurrents:
RPROG =
IPROG =
R6 [5 – 10(IBATT)(RPROG)]0.04 + 10(IBATT)(RPROG)
10(IBATT)(RPROG) + 0.04R6
PC Board Layout
As with any high frequency switching regulator, layout isimportant. Switching current paths and heat producingthermal paths should be identified and the printed circuitboard designed using good layout practices.
Even with efficiency numbers in the mid 90s, under somecharging conditions power losses can be as high as 4watts. These losses are primarily in the two MOSFETs, theinductor and the current sensing resistor. Since these aresurface mount components, the major thermal paths arethrough the pc board copper to the surrounding air.Maximizing copper area around the heat producing com-ponents, increasing board area and using double-sidedboard with feedthrough vias all contribute to heat dissipa-tion. Remember, the pc board is the heat sink.
One exception to the maximum copper area rule is theswitch node consisting of Q1’s source, Q2’s drain and theleft side of L1. This node switches between ground and VINat a 200kHz rate. To minimize radiation from this node, itshould be short and direct. Other copper traces related toinput and output capacitors and MOSFET connectionsshould also be as short as practical. See the LTC1435 datasheet for information on good layout practices and addi-tional applications information.
BATTERY CHARGER IC CAN ALSO SERVEAS MAIN STEP-DOWN CONVERTERby LTC Applications Staff
Using a power adapter with the highest feasible outputvoltage is attractive to portable system designers for acouple of reasons. Lower current is required to maintainthe same system power, which translates into a smallercable and input connector. If the adapter output voltage isconsiderably higher than the battery voltage, the adapteroutput voltage does not need to be regulated or wellfiltered, resulting in lower adapter cost.
A portable system with a high output-voltage adapter,however, requires that the system’s DC-to-DC converterfunctions over a very wide range of input voltage: from fullydischarged battery voltage to the highest adapter outputvoltage.
This problem can be resolved by using the LT1510 as boththe battery charger and the main step-down converter, asshown in Figure 244. An important feature of the circuit inFigure 244 is the glitch-free transfer from AC operation tobattery operation and back.
Application Note 84
AN84-128
The LT1510 battery charger IC is capable of chargingcurrents up to 1.5A and output (battery) voltages up to20V. High efficiency and small inductor size are achievedby a saturating switch running at 200kHz. The LT1510 iscapable of charging lithium-ion and sealed-lead-acidbatteries in the constant-voltage/constant-current con-figuration, and nickel-cadmium and nickel-metal-hyd-ride batteries in the constant-current configuration. TheLT1510 contains an internal switch and current senseresistor. All the designer needs to do in order to programthe current and voltage is select the current-programmingresistor and the voltage-divider resistors.
In the circuit shown in Figure 244, the system’s DC-to-DCconverter is connected to the SENSE pin. This way, theinternal sense resistor is bypassed for the system load butis active in regulating the charging current. The sum of thecharging current and system current should not exceedthe maximum output current allowed (limited by thermalconsiderations or peak switch current). Since the DC-to-DC converter circuit has a large input capacitor, it cannot
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
GND
VCC1
VCC2
PROG
VC
BAT
GND
GND
GND
SW
BOOST
GND
OVP
SENSE
GND
GND
U1LTC1510CS16
C9*0.1µF
CR7* 1N914
CR6* 1N914
CR11N5819
VIN
VIN
VIN
R9100k
Q4VN2222
CHARGE/TRICKLE
R812.4k
R7*100k
R6300Ω
R51k
R44.99k
BAT1
C51µF
C40.1µF
Q3VN2222
C322µF25V
CR3 1N5819
C2† 10µF
Q1*MPS3906
R31.5k
CR5 1N5819
Q2*Si9433
R2*1M
R1100k
SYSTEMON/OFFSWITCH
C1 0.22µF
L1**33µH
L22.2mH
L310µH
CR41N5817
SYSTEMLOAD
C7100µF
C6100µF
U2LT1300CS8
C8*0.1µF
VIN SW
SELECT SENSE
PGND GND
SHDN ILIM
CR2 1N914
6 7
2 4
8 1
3 5NCNC
* SEE TEXT** COILTRONICS CTX33-2† TOKIN OR MARCOM CERAMIC SURFACE MOUNT
+
+
+
be connected directly to the SENSE pin. This is because theinternal sense resistor between SENSE and BAT pins willsee a large capacitance across it, which will cause instabil-ity. A 2.2mH inductor, such as the DT1608C-222 byCoilcraft (L2), is used to isolate the input capacitance ofthe DC-to-DC converter. CR5 limits the transient currentthrough the LT1510’s internal sense resistor when the
INPUT VOLTAGE (V)
67.5
68.0
68.5
69.0
69.5
70.0
70.5
71.0
71.5
72.0
72.5
EFFI
CIEN
CY (%
)
28
DI1510_02.eps
8 2313 18
Figure 244. LT1510 Battery Charger/Main Step-Down Converter Provides Glitch-Free Transfer between AC and Battery Operation
Figure 245. System Efficiency vs Input Voltage
Application Note 84
AN84-129
LT1635 1A SHUNT CHARGERby Mitchell Lee
Most battery chargers comprise nothing more than aseries-pass regulator with current limit. In solar-poweredsystems, you can’t count on sufficient headroom to keepa series regulator alive, so a shunt method is preferred. Asimple shunt battery charger is shown in Figure 246. Itconsists of an op amp driving a shunt transistor andballast resistor, and is built around an LT1635. This devicecontains both an op amp and a reference, making itperfectly suited for regulator and charger applications.
Operation is straightforward: the battery voltage is sensedby a feedback divider composed of two 1M resistors. The
system is operating from the battery and turned on. Q2(Si9433) is required if the series resistance of 0.2Ωbetween the BAT pin and SENSE pin is too high. TheSi9433’s on resistance is 0.075Ω. The charge pumpcomprising C8, C9, CR6, CR7 and R2 biases the gate ofQ2. Q1 and R1 turn Q2 off on AC operation (VIN active). R7programs the trickle-charge current (maximum value isabout 100k) and the equivalent value of R7 and R8programs the charge current. The Charge input must bepulled low at the end of the charge.
The charger in Figure 244 is connected to a 2-cell NiCdbattery, BAT1. The system switching regulator is LT1300(U2) based and powers a 5V/250mA load. The efficiency,η, of the complete system is defined as:
LT1300 Output Power + Battery Charger PowerLT1510 Input Power
η =
The efficiency plot is shown in Figure 245. For the purposeof measurement, the battery voltage was 3.2V, the charg-ing current was 0.4A and the trickle charge was 40mA.
internal 200mV reference is amplified to 7.05V and com-pared against the feedback. RT1 introduces a TC thataccurately tracks the battery’s correct charging voltageover a wide temperature range. Because RT1 is designedto compensate for changes in battery temperature, itshould be located close to the battery and as far aspossible from the shunt elements. When the batterycharges to 14.1V, the op amp output voltage begins to rise,turning on the Darlington shunt and resisting furtherincreases in voltage. Full panel power is divided equallybetween the transistor and 7.5Ω resistor when the batteryis completely charged. Don’t forget to provide adequateheat sinking and air flow for up to 15W dissipation.
–
+–
+
2A
12V, 5AhGelcell
7.5Ω/10WDALE HLM-10
TIP121
220Ω
100nF
OA1/2 LT1635
REF1/2 LT1635
1M
1M
64.5k
105Ω
2
3 7
6
48
1 7.05V
200mV
1A SOLAR ARRAY
2.43k RT17.5k
RT1 = THERM-O-DISC 1K752J
14.1V
Figure 246. 1A Shunt Battery Charger (IDARK = 230µA; VFLOAT = 14.1V)
Application Note 84
AN84-130
800mA Li-Ion BATTERY CHARGER OCCUPIESLESS VOLUME THAN TWO STACKED QUARTERSby Fran Hoffart
Each new generation of cell phones, PDAs, portableinstruments and other handheld devices is invariably morepowerful, smaller and, most likely, thinner than the last.The circuit shown in Figure 247 is designed to charge oneor two Lithium-Ion cells at currents up to 800mA, with allcomponents equal to or less than 2.2mm (0.086 inches)tall. Using 0.031 inch PC board material, the total circuitthickness for this charger is 3.4mm (0.136in) or thethickness of two quarters. The complete 800mA constant-current/constant-voltage charger, including the PC board,occupies less volume than two quarters. This compact,low profile construction is ideal for cell phones or otherapplications where circuit height is restricted.
LT1510-5CGN High Efficiency 500kHzSwitch Mode Battery Charger IC
The charger consists of an LT1510 constant-voltage/constant-current PWM IC, which includes an onboard1.5A switch. The LT1510 is available in either 200kHz or500kHz versions; the higher frequency version allowslower value, smaller-sized inductors to be used. An inter-nal 0.5% reference allows precision battery-voltage pro-gramming and a current programming pin allows a singleresistor, PWM signal or a programming current from aDAC to control the charging current. Also included areundervoltage lockout and a low quiescent current sleepmode that is activated when input power is removed.
The charger is designed to handle 1A continuous, whichis compatible with a “20W” panel. There is no need todisconnect or diode isolate the charger during periods ofdarkness, because the standby current is only 230µA—less than 10% of the self discharge of even a small battery.
If a different or adjustable output is desired, the feedbackratio can be easily modified at the 1M divider. 14.1V is acompromise between an aggressive charge voltage and aconservative float voltage. Given the cyclic nature of
insolation, allowing periodic charging at 14.1V is notdetrimental to Gelcell™ batteries. The circuit in Figure 246will work with larger or smaller batteries than that shown.As a rule of thumb, the panel should be sized from 1W per10Ah battery capacity (a float charge under good condi-tions with a good battery) to 5W per 1Ah battery capacity(1 day recharge of a completely discharged battery underfavorable conditions of insolation).Gelcell is a trademark of Johnson Controls, Inc.
R5
11.0k0.5%
R6
1.02k0.5%
R4
4.99k0.5%
TO VIN
Q12N7002
VBAT = 2.465V 1 +R5 + R6
R4( )
IBAT = 20002.465V
R1( )L1TP3-10010µH
D1MBRM140T3
D2MBRM140T3
C20.22µF
C1
10µF
D3 MMBD914LT1
GND
SW
BOOST
GND
OVP
NC
SENSE
GND
GND
VCC
VCC
PROG
VC
NC
BAT
GND
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
LT1510-5
VIN = 12V–20V
C40.1µF
R2300
C31µF
C522µF
R16.19k1%
+VBAT = 8.4VLi-IONBATTERY(2 CELLS)
IBAT = 800mA
IBAT
R31k
DI 1510 01.eps
+
Figure 247. Compact, Low Profile, Constant-Current/Constant-Voltage Charger for Li-Ion Batteries
Application Note 84
AN84-131
Fused-Lead Package OffersLower Thermal Resistance
The LT1510-5 is available in a specially constructed 16-lead plastic SSOP package that has the die-attach paddleconnected (fused) directly to the four corner leads and fitsin the same area as an SO-8 package. This low profilefused-lead package provides a lower thermal resistanceby conducting much of the heat generated by the diethrough the copper leads to the PC board copper.
To take advantage of the improved thermal properties ofthis fused-lead package, it is important to provide as muchPC board copper around the package leads as practical.Back-side copper and internal copper layers intercon-nected by feed-through vias all contribute to the overalleffectiveness of the PC board as a heat sink.
Charger Operation
A typical charge profile for a discharged Li-Ion battery isan initial constant-current charge at 800mA until thebattery voltage rises to the programmed voltage. It then
changes to a constant-voltage charge, with the chargingcurrent gradually decreasing to near 0mA as the batteryapproaches full charge. If complete charge termination isrequired, pulling the VC pin low or sinking zero currentfrom the program pin stops the charge current. Thesesignals could be supplied by an external timer ormicroprocessor.
When the input power is removed, the LT1510-5 goes intoa low quiescent current (3µA) sleep mode, with thiscurrent coming from the battery. This low battery draincurrent allows the battery to remain connected to thecharger for an extended period of time without appreciablydischarging the battery. Additional battery-drain currentcan result from reverse leakage current in the Schottkycatch diode D1. Many Schottky diodes have relatively highleakage currents, so care must be exercised in theirselection.
Refer to the LT1510 data sheet for complete productspecifications and to design notes DN111 and DN124 andapplication note AN68 for additional applicationinformation.
ecnerefeRrotangiseD ytitnauQ rebmuNtraP noitpircseD rodneV enohP
1C 1 TZ601E1E05RCHT cimareCU5Y%02,V52,Fµ01 nocraM 0002-696)748(2C 1 A1TAM422C36021 cimareCR7X%02,V52,Fµ22.0 XVA 1115-282)702(3C 1 TAM501CZ5080 cimareCR7X%02,V01,Fµ1 XVA 1115-282)702(4C 1 A1TAM401G55080 cimareCR7X%02,V05,Fµ1.0 XVA 1115-282)702(5C 1 R022B1DCFEE citylortcelEmunimulAremyloP%02,V5.21,Fµ22 cinosanaP 0665-549)804(
2D,1D 2 3T041MRBM ykttohcSV04,A1 alorotoM 7442-144)008(3D 1 1TL419DBMM nociliSV001A2.0 alorotoM 7442-144)008(1L 1 001-3PT caP-nihTHµ01 scinortlioC 6787-142)165(1Q 1 2007N2 TEFSOMlennahC-N32-TOS xeteZ 0017-345)615(1R 1 rotsiseRpihC%1.k91.6 CRI 0097-299)215(2R 1 rotsiseRpihC%5,,Ω003 CRI 0097-299)215(3R 1 rotsiseRpihC%5,k1 CRI 0097-299)215(4R 1 rotsiseRpihC%5.0,k99.4 CRI 0097-299)215(5R 1 rotsiseRpihC%5.0,k0.11 CRI 0097-299)215(6R 1 rotsiseRpihC%5.0,k20.1 CRI 0097-299)215(1U 1 NGC5-0151TL CIregrahCyrettaB CTL 0091-234)804(
Table 4. Low Profile Components Used in Figure 247's Circuit
Application Note 84
AN84-132
SINGLE-CELL Li-Ion BATTERY SUPERVISORby Albert Lee
Recently introduced precision products from Linear Tech-nology allow designers to implement high precision appli-cations at supermicropower levels. Among these devicesare the LT1496 quad precision input/rail-to-rail output opamp and the LT1634 precision shunt voltage reference,which operate at only 1.5µA and 10µA, respectively. Evenat such low power levels, precision performance is notcompromised. The LT1496 features 475µV maximuminput offset voltage and 1nA maximum input bias current.The LT1634 achieves 0.05% initial accuracy and 25ppm/°Cmaximum temperature drift.
Figure 248 shows a single-cell Li-Ion battery supervisorycircuit. The building blocks of this circuit are the LT1496precision op amp and LT1634 voltage reference. Theuseful region of operation of a single-cell Li-Ion battery isbetween 4.2V and 3V. The cell voltage drops fairly quicklybelow 3V. System operation below this voltage can be
erratic. Although Li-Ion battery use is becoming wide-spread, it is costly to damage the battery. The supervisorycircuit protects the battery from overcharging and/oroverdraining and prevents the battery voltage from fallingout of its operating region. The LT1496 operates down to2.2V, ensuring that circuit operation is maintained whenthe battery voltage falls below 3V.
The Li-Ion battery is monitored via a voltage divider off thebattery voltage (node A). The divided voltage is fed into thepositive inputs of comparators A2 and A3 and comparedto the threshold voltages of 1.75V and 1.25V, respectively.These voltages are selected so that the minimum batterycharge voltage is 3V and the maximum is 4.2V. TheLT1634 1.25V reference is buffered by op amp A1. Theconstant 1.25V across R2 creates a 1µA constant current,so that the output of A1 is amplified to 1.75V. This outputdrives RS to provide constant bias current for the LT1634.
Depending on the battery voltage, the circuit is in one ofthe three states, as shown in Table 5.
–
+
–
+
–
+–
+
A11/4 LT1496
A21/4 LT1496
A31/4 LT1496
A41/4 LT1496
RS175k5%
LT16341.250V
R1500k0.1%
R21.25M0.1%
VBAT
VBAT
RH110M5%
RH210M5%
D1
D2
BATTERY
R31.75M0.1%
R41.25M0.1%
CHARGER
TO LOAD
RSW1M5%
SW*
10µA
1µA
1.75V
1.25V
A
OFF
D1, D2 = 1N458R1–R4 = CAR6 SERIES IRC (512) 992-7900TP0610L for 50mA LOAD*
Figure 248. Single-Cell Li-Ion Battery Supervisor Circuit
Application Note 84
AN84-133
The voltage at node A is compared to the two thresholdvoltages to determine the state of the circuit. For instance,when node A reaches or exceeds 1.75V (battery voltagereaches 4.2V), the outputs of A2 and A3 will swing to thepositive rail, terminating the charger and connecting theload to the battery. When node A falls between 1.25V and1.75V (battery voltage between 3V and 4.2V), the outputof A2 swings low, turning the charger on, while the outputof A3 stays high, leaving the load connected. When nodeA falls below 1.25V (battery voltage less than 3.0V), theoutput of A2 stays low, keeping the charger on. The outputof A3 will also swing low, which, in turn, will cause theoutput of A4 to go high, turning off the FET SW thatdisconnects the load from the battery.
If node A were to bounce around at either thresholdvoltage, the circuit would bounce between states. To avoidthis problem, hysteresis is added via the resistor anddiode networks connected between the outputs of A2 andA3 and their positive inputs. Figure 249 shows the behav-ior of VBAT vs node A entering the trip points with hyster-esis. When VBAT rises to 4.2V (node A increases to 1.75V),op amp A2’s output will switch from low to high, causingcurrent to flow through RH1. The additional current willraise node A by an amount ∆VAHYS1, which will clearly putthe circuit in state 3. The circuit will not exit state 3 untilVBAT falls to ∆VHYS1 (310mV for the circuit shown) below4.2V, which will cause node A to fall back to the upper trippoint of 1.75V (point 1 of Figure 2). Similarly, when VBATdrops below 3V (node A falls below 1.25V), op amp A3’soutput will switch low, causing current to conduct throughRH2. This will drag node A an amount ∆VAHYS2 below1.25V, which will put the circuit in state 1. The circuit willnot exit state 1 until the battery voltage is charged to anamount ∆VHYS2 (149mV for circuit shown) above 3V(point 2). This will bring node A back up to the lower trippoint, 1.25V, bringing the circuit out of state 1. The
amount of hysteresis desired can be calculated using thefollowing formulas:
High Trip Point:
VBAT =
R3RH1
• (VOHMIN + VBE + 1.75V) + (1.75V • ) + 1.75 R3R4( (
R3RH1
1 +
∆VHYS1 = 4.2V – VBAT
Low Trip Point:IRH2 = (1.25V – VOLMAX – VBE)/RH2∆VHYS2 = IRH2 • R3
where:VOHMIN = output voltage swing high (LT1496)VOLMAX = output voltage swing low (LT1496)VBE = diode voltage of 1N458
Using an automobile analogy, if the LT1496 op amp is thetransmission of the circuit (switching from one state to thenext), the LT1634 voltage reference is the engine. It notonly generates the threshold voltages, but also the amountof error that the circuit will have. How much accuracy anderror you get depends on the car you drive. Maximuminput offset voltage and input bias current for the LT1496are 475µV and 1nA, respectively. The LT1634 is a 0.05%initial accuracy, 25ppm/°C tempco, 10µA precision shuntreference. Its 1.250V output voltage will appear at theinput of A3 with an accuracy of 0.088% (initial accuracy +input offset voltage). R1 and R2 being 0.1% resistors, theworst-case ratio error will be 0.2%. The worst-case volt-
VBAT (V)
1.25
1.75
3.0 4.2
∆VAHYS1
∆VHY
S1
∆VHY
S2
∆VAHYS2
V A (V
)
1
2
etatS V TAB AedoNtuptuO
2AtuptuO
3AtuptuO
4A sutatS
1 V3< V52.1< woL woL hgiH ,ffOdaoLetatSegrahC
2 V<V3V2.4<
V<V52.1V57.1< woL hgiH woL ,nOdaoL
etatSegrahC
3 V2.4> V57.1> hgiH hgiH woL,nOdaoL
egrahCdetanimreT
Table 5. Circuit States
Figure 249. VBAT vs VA with Hysteresis
Application Note 84
AN84-134
age error across R1 will then be 0.2% or 1mV. This errorcompared to the 1.75V threshold voltage is 0.057%.Similarly, error at 1.75V due to worst-case 2nA input biascurrent is 0.057%. Total worst-case error at 1.75V will be0.202%.
VBAT error contributed by the voltage divider branch willconsist of three terms: resistor matching, op amp inputbias current and input offset voltage. The amount of erroris different at the two trip points when VBAT is 3V or 4.2V.Similar calculations as above result in 0.328% when VBAT= 3V and 0.268% when VBAT = 4.2V. Therefore, total
battery voltage error at either trip points is better than0.47%. Since only the ratios of R1 to R2 and R3 to R4 arecritical, precision matched resistors with ten times betterperformance can be used to reduce the overall error by33%.
This supervisory circuit demonstrates unparalleled per-formance achievable only with Linear Technology’ssupermicropower precision devices. The supervisory cir-cuit consumes only 20mA. Battery voltage monitoring andcontrol accuracy is better than 0.5%.
Power Management
LTC1479 PowerPath CONTROLLER SIMPLIFIESPORTABLE POWER MANAGEMENT DESIGNby Tim Skovmand
Introduction
The LTC1479 PowerPath™ controller drives low loss N-channel MOSFET switches to direct power in the mainpower path of a dual rechargeable battery system, the typefound in most notebook computers and other portableequipment.
Figure 250 is a conceptual block diagram that illustratesthe main features of an LTC1479 dual-battery powermanagement system, starting with the three main powersources and ending at the input of the DC/DC switchingregulator.
Switches SWA/B, SWC/D and SWE/F direct power fromeither the AC adapter (DCIN) or one of the two batterypacks (BAT1 and BAT2) to the input of the DC/DC switch-ing regulator. Switches SWG and SWH connect the de-sired battery pack to the battery charger. These fiveswitches are intelligently controlled by the LTC1479,which interfaces directly with the power managementmicroprocessor.
BAT1
BAT2
BATTERYCHARGER(LT1510)
SWA/B
SWC/D
SWE/F
SWG SWH
+
HIGH EFFICIENCY DC/DC SWITCHING
REGULATOR(LTC1435, ETC.)CIN
DCIN
LTC1479PowerPath™ CONTROLLER POWER
MANAGEMENTµP
LTC1479 - BLK1
BACK-UPREGULATOR
ACADAPTER
5VINRUSHCURRENTLIMITING
Figure 250. Dual PowerPath Controller Conceptual Block Diagram
Application Note 84
AN84-135
DCIN
DCIN
BAT1
BAT2
+ +
V+ SW VGG
1µF50V
1mH * 1µF50V
VCC
+
RSENSE0.033Ω
SWA SWB
SWC SWD
SWE SWF
GA GB GC GD GE GFSAB SCD SEF SENSE+ SENSE-
+
VCCP
2.2µF16V 0.1µF
VBATPOWER
MANAGEMENTµP
BDIV
VBKUP BACK-UPREGULATOR
LTC1538-AUX TRIPLE, HIGH EFFICIENCY,SWITCHING REGULATOR
DCDIV
LTC1479PowerPath™ CONTROLLER
LT1510Li-Ion BATTERY
CHARGER
GG SG GH SH
CHGMON
3.3V
5.0V
SWG SWH
DCIN
Li-IonBATTERYPACK #1
Li-IonBATTERYPACK #2
LTC1479 - FIG03
BACKUPBATTERY
* 1812LS-105 XKBC, COILCRAFT (708) 639-1469
12V AUX
RB2
RB1
RDC2
RDC1
MBRS140T3330Ω
0.1µF
Typical Application Circuit
A typical dual Li-Ion battery power management system isillustrated in Figure 251. If “good” power is available at theDCIN input (from the AC adapter), both MOSFETs in switchpair SWA/B are on—providing a low loss path for currentflow to the input of the LTC1538-AUX DC/DC converter.Switch pairs SWC/D and SWE/F are turned off to blockcurrent from flowing back into the two battery packs fromthe DC input.
Battery Charging
The LTC1479 works equally well with both Li-Ion andNiMH batteries and chargers. In this application, an LT1510constant-voltage, constant-current (CC/CV) battery chargercircuit is used to alternately charge two Li-Ion batterypacks.
The power management microprocessor decides whichbattery is in need of recharging by either querying a smartbattery pack directly or by more indirect means. After the
determination is made, switch pair SWG or SWH is turnedon by the LTC1479 to pass charger current to one of thebatteries. Simultaneously, the selected battery voltage isreturned to the voltage feedback input of the LT1510 CV/CC battery charger via a built-in switch in the LTC1479.
After the first battery is charged, it is disconnected fromthe charger circuit. The second battery is then connectedthrough the other switch pair and the second batterycharged. (The LTC1479 works equally well with the LT15113A CC/CV Battery Charger and LTC1435/LT1620 4A CC/CV Battery Charger.)
Running on Batteries
When the AC adapter is removed, the LTC1479 instantlyinforms the power management microprocessor that theDC input is no longer “good” and the desired battery packis connected to the input of the LTC1538-AUX highefficiency switching regulator through either switch pairSWC/D or SWE/F.
Figure 251. Dual Li-Ion Battery Power-Management System (Simplified Schematic)
Application Note 84
AN84-136
Back-Up Power and System Recovery
Backup power is provided by a standby switching regula-tor, which is typically powered from a small rechargeablebattery and ensures that the DC/DC input voltage does notdrop below a predetermined level (for example, 6V).
The “3-Diode Mode”
When the system is powered by the backup regulator, theLTC1479 enters a unique operating state called the “3-diode mode,” as illustrated in Figure 252. Under normaloperating conditions, both halves of each switch pair areturned on and off simultaneously. For example, when theinput power source is switched from a good DC input (ACadapter) to a good battery pack, BAT1, both gates ofswitch pair SWA/B are turned off and both gates of switchpair SWC/D are turned on. The back-to-back body diodesin switch pair SWA/B block current flow in or out of the DCinput connector.
In the 3-diode mode, only the first half of each power pathswitch pair, that is, SWA, SWC and SWE, is turned on; andthe second half , that is, SWB, SWD and SWF, is turned off.These three switch pairs now act as 3-diodes connected to
the three main input power sources. The power path diodewith the highest input voltage passes current through tothe input of the DC/DC converter to ensure that the systemcannot lock up regardless of how power is initially applied.
After “good” power is reconnected to one of the threemain inputs, the LTC1479 drives the appropriate switchpair on fully as the other two are turned off, restoringnormal operation.
Interfacing to the Power Management Microprocessor
The LTC1479 takes logic level commands directly fromthe microprocessor and makes changes at high currentand high voltage levels in the power path. Further, itprovides information directly to the microprocessor onthe status of the AC adapter, the batteries and the chargingsystem.
The LTC1479 logic inputs and outputs are TTL levelcompatible and therefore interface directly with standardpower management microprocessor. Because of the di-rect interface via five logic inputs and two logic outputs,there is virtually no latency (time delay) between themicroprocessor and the LTC1479. In this way, time-
BAT1
BAT2
SWA
SWC
SWE+
HIGHEFFICIENCY
DC/DCSWITCHINGREGULATOR
5V
3.3V
12V
CIN
DCIN
LTC1479POWER
MANAGEMENTµP
LTC1479 - FIG04
SWB
SWF
SWDON OFF
ON OFF
ON OFF
RSENSE
Figure 252. LTC1479 PowerPath Controller in “3-Diode Mode”
Application Note 84
AN84-137
critical decisions can be made by the microprocessorwithout the inherent delays associated with bus protocolsand the like. These delays are acceptable in certain por-tions of the power management system, but it is vital thatthe power path switching control be made through a directconnection to the power management microprocessor.The remainder of the power management system can beeasily interfaced to the microprocessor through eitherparallel or serial interfaces.
The Power Management Microprocessor
The power management microprocessor provides intelli-gence for the overall power system, and is easily pro-grammed to accommodate the custom requirements ofeach system and to allow performance updates withoutresorting to costly hardware changes. Many inexpensivemicroprocessors are available that can easily fulfill theserequirements.
THE LTC1473 DUAL PowerPath SWITCH DRIVERSIMPLIFIES PORTABLE POWERMANAGEMENT DESIGNby Jaime Tseng
Introduction
The LTC1473 is the latest addition to Linear Technology’snew family of power management controllers, whichsimplify the design of circuitry for switching between twobatteries or a battery and an AC adapter.The LTC1473 dualPowerPath™ switch driver drives low loss N-channelMOSFET switches that direct power in the main powerpath of a single or dual rechargeable battery system, the
type found in most notebook computers and other por-table equipment.
Overview
The power management system in Figure 253 shows theLTC1473 driving two sets of back-to-back N-channelMOSFET switches connecting the two batteries to thesystem DC/DC regulator. Each of the switches is con-trolled by a TTL/CMOS compatible input that interfacesdirectly with a power management system microproces-sor. An internal boost regulator provides the voltage tofully enhance the logic-level N-channel MOSFET switches.
BAT1
BAT2
SWA1
SWA2+
HIGHEFFICIENCY
DC/DCSWITCHINGREGULATOR
5V
3.3V
12V
CIN
DCIN
LTC1473
POWERMANAGEMENT
µP
SWB2
SWB1 RSENSE
STEP-UPSWITCHINGREGULATOR
1µF50V
C1
GATEDRIVER
GATEDRIVER
1µF50V
C2
VGG
V+
INRUSHCURRENTSENSING
AND LIMITING
C1
4700pFCTIMER
TIMER1473_01.eps
IN1
IN2
DIODE
SW
L11mH
Si9926
Si9926
MB914LT1
MBRD340
+
+
Figure 253. Dual-Battery PowerPath Switch Driver: VGG Regulator, Inrush Limiting and Switch-Gate Drivers
Application Note 84
AN84-138
SAB1
GB1
SENSE+
SENSE-
GA2
SAB2
GB2
IN1
IN2
DIODE
TIMER
V+
VGG
SW
GND
LTC1473GA1
RSENSE
0.04Ω+
POWERMANAGEMENT
µPSUPPLYMONITOR
BAT1
DCIN
BAT2
COUT
INPUT OF SYSTEMHIGH EFFICIENCY DC/DCSWITCHING REGULATOR
(LTC1435,ETC)
CTIMER4700PF
1µF1µF
1mH
Si9926
Si9926
1473_03.eps
MBRD340
MMBD2823LT1MMBD2823LT1
MMBD-914LT1
++
The LTC1473 uses a current sense loop to limit currentrushing in and out of the batteries and the system supplycapacitor during switch-over transitions or during a faultcondition. A user programmable timer monitors the timeduring which the MOSFET switches are in current limit andlatches them off if the programmed time is exceeded. Aunique “2-diode logic mode” ensures system start-up,regardless of which input receives power first.
Typical Application
A typical dual-battery system is shown in Figure 254. TheLTC1473 accepts commands from a power management
microprocessor to select the appropriate battery. Themicroprocessor monitors the presence of batteries andthe AC adapter through a supply monitor block, or, in thecase of some battery packs, through a thermistor sensor.This block comprises a resistor divider and a comparatorfor each supply. If the AC adapter is present, the twoswitches are turned off by the microprocessor and thepower is delivered to the input of the system DC/DCswitching regulator via a Schottky diode.
Figure 254. Dual-Battery Power-Management System
Application Note 84
AN84-139
SHORT-CIRCUIT-PROOF ISOLATEDHIGH-SIDE SWITCHby Mitchell Lee
Figure 255 shows a MOSFET switch, driven by theLTC1177–5 2.5kVRMS isolator. This device allows a logicsignal to control a power MOSFET and provides completegalvanic isolation. The device includes an internal currentlimiting circuit, but at higher voltages limiting the currentis just not enough for effective protection of the MOSFET.Foldback (shown on the LTC1177 data sheet) helps, butthe part has trouble starting certain types of loads whenfoldback current limiting is used. The circuit shown herelatches off in an overcurrent condition and is restarted bycycling the logic input.
Q1 and Q2 form an SCR with a holding current of less than100nA. If the load current exceeds approximately 1A, the
2.5kV ISOLATION BARRIER5V
LTC1177
VIN OUT
G1 SENSE G2
C110nF
Q22N3904
10MQ12N3906
10M
100Ω
20M
MTD3055EL
LOAD
0.5Ω1W
D11N914
24V
OFF
ON
SCR fires, shorting the MOSFET gate to source. TheLTC1177 output current (about 7µA) is more than ad-equate to hold the SCR on indefinitely. The circuit resetswhen the logic input briefly cycles off.
Inductive loads present a special problem. If the loadcreeps up on the overcurrent threshold and fires the SCR,the load’s inductance will carry the MOSFET source farbelow ground, which could destroy the MOSFET. Diode D1clamps the gate at ground, turning the MOSFET back on,and safely dissipates the stored magnetic energy in theMOSFET.
As shown the output rise time is about 2ms, allowing thecircuit to successfully charge capacitors of up to 100µF.Increase C1 proportionately to handle higher value loadcapacitors.
Figure 255. Short-Circuit Protected, Isolated High-Side Switch
Application Note 84
AN84-140
TINY MSOP DUAL SWITCH DRIVERIS SMBus CONTROLLEDby Peter Guan
Introduction
The LTC1623 SMBus switch controller offers an inexpen-sive, space-saving alternative for controlling peripheralsin today’s complex portable computer systems. Pin-to-pinconnections between the system controller and eachperipheral device not only result in complicated wiring, butalso limit the number and type of peripheral devicesconnected to the system controller. Using the SMBusarchitecture, the LTC1623 eliminates these problems byrequiring only two bus wires and allowing easy upgradesand additions of new peripherals.
VCC2.7V TO 5.5V
10µF
(PROGRAMMABLE)
LOAD 2LOAD 1
1623 F02
(FROMSMBus)
LTC1623
GND
VCC
AD0
AD1
CLK
DATA
GA
GB0.1µF
Q1
Q20.1µF
1k
1k
Q1, Q2: Si3442DV
10µF5V
1623 TA02
LTC1623
GND
VCC
AD0
AD1
CLK
DATA
GA
GB
0.1µF
0.1µF
1µF
Q1Si3442DV
Q2*
Q3*
1k
1k
3.3V
10k
TO PC CARD VCC0V/3.3V/5V
*1/2 Si6926DQ
VCC2.7V TO 5.5V
10µF
(PROGRAMMABLE)
LOAD 2LOAD 1
1623 F02
(FROMSMBus)
LTC1623
GND
VCC
AD0
AD1
CLK
DATA
GA
GB0.1µF
Q1
Q20.1µF
1k
1k
Q1: Si3442DVQ2: Si6433DQ
Applications
The main application of the LTC1623 is to control twoexternal high-side N-channel switches (Figure 256). Asseen in the figure, a 0.1µF capacitor and a 1k resistor areplaced on each gate-drive output to respectively slowdown the turn-on time of the external switch and toeliminate any oscillations caused by the parasitic capaci-tance of the external switch and the parasitic inductanceof the connecting wires.
Tracking the growing popularity of portable communica-tion systems, the LTC1623 makes a very handy single-slot3.3V/5V PC Card switch matrix. As shown in Figure 257,this circuit enables a system controller to switch either a3.3V or a 5V supply to any of its SMBus-addressedperipherals. Besides N-channel switches, the LTC1623can also be used to control a P-channel switch, as shownin Figure 258. As a result, the load connected to the P-channel switch will be turned on upon power-up of theLTC1623, whereas the other load must wait for a validaddress and command to be powered.
Figure 256. LTC1623 Controlling Two High-Side Switches
Figure 257. PC Card 3.3V/5V Switch Matrix
Figure 258. LTC1623 Controlling a P-Channel Switch (Q2)
Application Note 84
AN84-141
LTC1710: TWO 0.4Ω SWITCHES WITH SMBusCONTROL FIT INTO TINY MSOP-8 PACKAGE
by Peter Guan
Introduction
The LTC1710 SMBus dual switch (Figure 259) is a com-plete solution for supplying power to portable-equipmentperipherals without the need for external switches. Two
CHARGEPUMPS
LOAD 1
LOAD 2
LTC1710
AD1
DATA
CLK
SW1
SW05
6
3
4
7
2
18
VCC2.7V TO 5V
SW0DGND TO VCC
10µF 10µF
FROM SMBus 2.7V LOAD
5V LOAD
5
6
3
4
7
2
18
VCC 5V
SW0D2.7V
10µF 10µF
FROM SMBus
LTC1710
AD1
DATA
CLKOUT0
OUT1
internal high-side N-channel switches, each capable ofdelivering 300mA at an RDS(ON) of 0.4Ω, are available inthe tiny MSOP-8 package. With a low standby current of14µA, the LTC1710 operates on an input voltage of 2.7Vto 5.5V while maintaining the SMBus-specified 0.6V VILand 1.4V VIH input thresholds.
Figure 260 shows a circuit using SMBus peripheralsrequiring different input voltages can be simultaneouslyswitched by the LTC1710.
Miscellaneous
VID VOLTAGE PROGRAMMERFOR INTEL MOBILE PROCESSORSby Peter Guan
Microprocessor manufacturers’ relentless push for higherspeed and lower power dissipation, especially in areas ofmobile laptop computer processors, is forcing supplyvoltages to these processors to a level previously thoughtimpossible or impractical. In fact, the supply voltage hasbecome so critical that different microprocessors demanddifferent yet precise supply voltage levels in order tofunction optimally.
To accommodate this new generation of microproces-sors, LTC introduces the LTC1706-19 VID (voltageidentification) voltage programmer. This device is a preci-sion, digitally programmable resistive divider designed foruse with an entire family of LTC’s DC/DC converters withonboard 1.19V references. These converters include theLTC1433, LTC1434, LTC1435, LTC1435A, LTC1436,LTC1438, LTC1439, LTC1538-AUX, LTC1539 andLTC1624. (Consult the factory for future compatible DC/DC converter products.) The LTC1706-19 is fully compli-ant with the Intel mobile VID specifications and comes ina tiny SO-8 package. Four digital pins are provided toprogram output voltages from 1.3V to 2.0V in 50mV stepswith an accuracy of ±0.25%.
Figure 259. Typical Application: TheLTC1710 Switches Two SMBus Peripherals
Figure 260. LTC1710 Switches Two SMBusPeripherals with Different Input Voltages
Application Note 84
AN84-142
Applications
Figure 261 shows a VID-programmed DC/DC converterfor an Intel mobile processor that uses the LTC1435A andLTC1706-19 to deliver 7A of output current with a pro-grammable VOUT of 1.3V to 2.0V from a VIN of 4.5V to 22V.Simply connecting the LTC1706-19’s FB and SENSE pinsto the LTC1435A’s VOSENSE and SENSE– pins, respec-tively, closes the loop between the output voltage senseand the feedback inputs of the LTC1435A regulator withthe appropriate resistive divider network, which is con-trolled by the LTC1706-19’s four VID input pins.
Table 6 shows the VID inputs and their correspondingoutput voltages. VID3 is the most significant bit (MSB) andVID0 is the least significant bit (LSB). When all four inputsare low, the LTC1706-19 sets the regulator output voltageto 2.00V. Each increasing binary count is equivalent todecreasing the output voltage by 50mV. Therefore, toobtain a 1.30V output, the three MSBs are left floatingwhile only VID0 is grounded. In cases where all four VIDinputs are tied high or left floating, such as when noprocessor is present in the system, a regulated 1.25Voutput is generated at VSENSE.
+
+ +
M1Si4410DY
M2 Si4410DY
CIN10µF, 30V×2
L1 3.3µHDB*
RSENSE0.015Ω
COUT820µF4V×2
D1MBRS
-140T3
4.7µF
1000pF
0.22µF
COSC 43pF
CSS0.1µF
CC1000pF
CC2220pF
51pF
VIN
TG
SW
INTVCC
BOOST
BG
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
13
16
14
12
15
11
10
1
2
3
5
6
VOUT1.30V TO2.00V/7A
VIN4.5V TO 22V
SENSE– SENSE+
7 8
LTC1435A
*DB = CMDSH-3
RC10k
RF4.7Ω
CF0.1µF
VCC
FB
VID0
VID1
VID2
VID3 GND
SENSE
LTC1706-19
FROM µP
Figure 261. Intel Mobil Pentium II Processor VID Power Converter
Table 6. VID Inputs and Corresponding Output Voltages
Code VID3 VID2 VID1 VID0 Output
0000 GND GND GND GND 2.00V
0001 GND GND GND Float 1.95V
0010 GND GND Float GND 1.90V
0011 GND GND Float Float 1.85V
0100 GND Float GND GND 1.80V
0101 GND Float GND Float 1.75V
0110 GND Float Float GND 1.70V
0111 GND Float Float Float 1.65V
1000 Float GND GND GND 1.60V
1001 Float GND GND Float 1.55V
1010 Float GND Float GND 1.50V
1011 Float GND Float Float 1.45V
1100 Float Float GND GND 1.40V
1101 Float Float GND Float 1.35V
1110 Float Float Float GND 1.30V
Application Note 84
AN84-143
+
+ +
M1Si4412DY
M2 Si4412DY
CIN22µF, 35V×2
L13.3µH
DB*
RSENSE0.02Ω
COUT100µF10V×2
D1MBRS
-140T3
4.7µF
1000pF
0.22µF
COSC 39pF
CSS0.1µF
CC510pF
100pF
VIN
TGL
TGS
SW
INTVCC
BOOST
BGL
PGND
COSC
RUN/SS
ITH
SGND
VOSENSE
18
21
19
20
17
22
16
15
2
3
4
6
8
VOUT1.30V TO2.00V/5A
VIN4.5V TO 22V
SENSE– SENSE+
9 10
LTC1436A-PLL
*DB = CMDSH-3
RC10k
VCC
FB
VID0
VID1
VID2
VID3 GND
SENSE
LTC1706-19
FROM µP
M3IRLML2803
PLL LPF PLLIN
10k 0.1µF
EXTERNALFREQUENCY
SYNCHRONIZATION
1 24
7
8
1
2
VID0
VID1
VID2
VID3
6
5
SENSE
FBGND
VCC
4
3LTC1706-19
10µF
VCC 2.7V TO 5.5V
+
+
1
2
3
4
8
7
6
5
SENSE–
ITH/RUN
VFB
GND
VIN
BOOST
TG
SW
LTC1624
470pF
6.8k
100pF
1000pF
0.1µF
RSENSE0.05Ω
Si4412DY
10µH
MBRS340T3
CIN22µF35V×2
VOUT1.3V TO 2.0V
COUT100µF10V×2
VIN4.8V TO 20V
Figure 262 shows a combination of the LTC1624 and theLTC1706-19 configured as a high efficiency step-downswitching regulator with a programmable output of 1.3Vto 2.0V from an input of 4.8V to 20V. Using only oneN-channel power MOSFET, the two SO-8 packaged LTCparts offer an extremely versatile, efficient, compact regu-lated power supply.
Figure 263 shows the LTC1436A-PLL and the LTC1706-19, a combination that yields a high efficiency low noisesynchronous step-down switching regulator with pro-grammable 1.3V to 2V outputs and external frequencysynchronization capability.
Figure 262. High Efficiency SO-8, N-Channel Switching Regulator with Programmable Output
Figure 263. High Efficiency, Low Noise, Synchronous Step-Down SwitchingRegulator with Programmable Output and External Synchronization
Application Note 84
AN84-144
Besides the LTC family of 1.19V-referenced DC/DC con-verters, the LTC1706-19 can also be used to program theoutput voltages of regulators with different onboard refer-ences. Figure 264 shows the LTC1706-19 programmingthe output of the LT1575, an UltraFast™ transient response,
+
7
8
1
2
VID0
VID1
VID2
VID3
1
2
3
4
8
7
6
5
SHDN
VIN
GND
FB
IPOS
INEG
GATE
COMP
6
5
SENSE
FBKGND
VCC
4
3LTC1706-19
1µF
VCC 3.3V
VIN12V
10pF7.5k
1000pF
24µF
5.1Ω
220µF
IRFZ24
3.3V
VOUT1.27V–2.03IN 50.8mV STEPS
LT1575
low dropout regulator that is ideal for today’s power-hungry desktop microprocessors. However, since theLT1575 has a 1.21V reference instead of a 1.19V refer-ence, the output will range from 1.27V to 2.03V in steps of50.8mV.
Figure 264. UltraFast Transient Response, Low Dropout Regulator with Adjustable Output Voltage
Application Note 84
AN84-145
BATTERY CHARGER ICDOUBLES AS CURRENT SENSORby Craig Varga
It’s always fun to find applications for an IC that itsdesigner never intended. The circuit shown in Figure 264is such a design. In many cases, a circuit is required toprovide a ground-referenced output voltage that is propor-tional to a measured current. Frequently, the current mustbe measured with a shunt in the positive rail that may bewell above ground and, worse yet, may vary considerablywith time. The LT1620 was originally intended as a con-troller for a synchronous buck regulator in battery-chargerapplications. The normal operating mode for this IC is tomirror a current signal down to a 5V reference supply. Byadding a single small-signal MOSFET and a few resistors,it is possible to again mirror this signal to provide a groundreferenced output.
Circuit operation is as follows: The LT1620 operates byproducing a voltage between the VCC pin and the AVG pinthat is 10× the voltage across sense resistor R5. C2 filters
8
6
4
5
7
1
2
3
AVG
VCC
IN–
IN+
PROG
SENSE
IOUT
GND
LT1620
C20.33µF
C30.33µF
R410k1%
R1100k
C11000pF
R50.02Ω
Q1TP0610T
R331.6k
5V
LOAD
INPUT
OUTPUT
R21.2M
IL
IL
OUTP
UT V
OLTA
GE (V
)
0 6LOAD CURRENT (A)
31 2 4 5
4.0
3.5
3.0
2.5
2.0
1.5
1.0
0.5
0.0
this voltage. An internal op amp has its noninverting inputat the AVG pin (pin 8), its inverting input at the PROG pin(pin 7) and its output at the IOUT pin (pin 2). With the circuitconnected as shown in Figure 265, this amplifier will forceenough current through R4 to make the voltage drop on R4equal to the voltage across C2. This current is mirroredthrough R3 and is filtered by C3, producing a clean,ground-referenced, DC output voltage. Resistor R2 can-cels a small built-in offset in the LT1620’s amplifiers. Theoutput voltage obeys the following relationship: VO = IL(R5 • R3 • 10)/R4. Changing the value of R3 selectsdifferent scale factors.
The circuit yields excellent linearity over a wide range ofloads and input voltages. The curve shown in Figure 266was measured with the sense resistor referenced to a 5Vinput source. The curve looks the same even at inputs over25V, so only one curve is presented. Maximum inputvoltage is 36V. There is a small offset at no load, but in atypical microprocessor-based data acquisition system,only a simple 2-point calibration is needed to obtainabsolute accuracy.
Figure 265. Current Sensor Schematic Figure 266. Transfer Function
Application Note 84
AN84-146
100V, 2A, CONSTANT-VOLTAGE/CONSTANT-CURRENT BENCH SUPPLYby Mitchell Lee and Jesus Rosales
Most engineering labs are well stocked with low voltage,moderate current power supplies, but higher voltage sup-plies capable of several amperes of output current are hardto find. We solved this problem in our lab by building thesupply shown in Figure 267.
The circuit is based on U1, an LT1270 high efficiencyswitching regulator configured in a SEPIC topology, whichallows the output to be adjusted higher or lower than theinput voltage. Operation is similar to that of a flybackconverter, but the primary and secondary windings arecoupled together by capacitor C1. This allows the primaryand secondary windings to share current, reducing copperloss; it also eliminates the snubbing circuitry and lossesfound in flyback converters.
+
–
+
–
+
–
+
+
+
+ +
+
–
+
3.3k2W
2N6387IRF450
1N5817
1N5817
10Ω
10Ω
100Ω
U1LT1270
4
5
3 1
2
680µF100V
×2
15V 56µF35V
1µF
22V
2N2907
2.2k
100Ω
0.33µF
1/2LT1215
1/2LT1215
100Ω
0.01µF
10k
15k
20k
10k
0.1µF
R2010k
1µF
LT1034CZ-2.5
R13.9k
R23.9k
1/2LT1413
1/2LT1413
7
6
5
1
8 2
3
4
1N4148 13
2
0.1µF1k
3.9k
3.9k3.9k
R21, 1k
R34.5k
R43.9k
R52.7M
0.03Ω, 2W
4.7k0.1µF
1k
7
8 6
541N4148
2.2k
10µF, 200VFILM CAP
10µF, 200VFILM CAP
L120µHMUR1560
2.2k 1000pF
MUR120
MBR745
C110µF, 100VFILM CAP
T1
1k
VOUT0V TO100V
100k
VIN40V TO
60V
15k
T1 : PRIMARY: 57 TURNS 20AWGSECONDARY: 57 TURNS 20AWGMPP 55076 MAG INC CORE
L1: 18 TURNS 18AWG55380-A2 MAG INC CORE
3.9V
U2B
U2A
U3BU3A
The converter is designed to operate from an input of 40Vto 60V, supplied by a line transformer, diode bridge andfilter capacitor (not shown). Output voltage is linearlyadjustable from zero to 100V via potentiometer R20.
The current is limited by two independent loops. The firstcurrent limit loop is user controlled over a range of zero to8A by setting potentiometer R21. This setting does notinteract with changes in output voltage. A second currentlimit loop limits the maximum available current as afunction of voltage (components R1–R5 and U2), mini-mizing component stress. Under any given operatingcondition, the lower of the two loops takes control. Maxi-mum available output current is highest at low outputvoltage settings (about 8A), and decreases to 2A at 100Voutput.
Figure 267. Constant-Voltage/Constant-Current Bench Supply
Application Note 84
AN84-147
A COMPLETE BATTERY BACKUP SOLUTIONUSING A RECHARGEABLE NiCd CELLby L.Y. Lin and S.H. Lim
Battery-powered systems, including notebook comput-ers, personal digital assistants (PDAs) and portable in-struments, require backup systems to keep the memoryalive while the main battery is being replaced. The mostcommon solution is to use an expensive, nonrechargeable
lithium battery. This solution requires low-battery detec-tion, necessitates battery access and invites inadvertentbattery removal. The LTC1558 battery backup controllereliminates these problems by permitting the use of asingle, low cost 1.2V rechargeable Nickel-Cadmium (NiCd)cell. The LTC1558 has a built-in fast-/trickle-mode chargerthat charges the NiCd cell when main power is present.
C1147µF6.3V
L11†22µH
SW
SW11
1
3
7
LTC1558-3.3
8
5
6
2
4
CTL
GND
FB
VCC
VBAK
RESET
BKUP
+
C121µF
FROM µPOPEN DRAIN SOFT RESET
TOµP
CIN100µF16V×2
+
+C20.1µF
CC330pF
CC251pF
C34.7µF16V
Q2Si4412DY
Q1Si4412DY
+
BACKUPBATTERY NiCd††
1.2V
R1410k
PUSH-BUTTONRESET
R13100k
R1512k
R1151k1%
Q11Si4431DY
MAIN BATTERY4.5V TO 10V
R1221.2k
(20.0k 1% +1.21k 1%)
VIN TG
SGND PGND
LTC1435
EXTVCC
SFB
VOSENSE
ITH
RUN/SS
COSC
SW
BOOST
INTVCC
SENSE+
SENSE–
BG
9
4
6
3
2
1
5 10
14
13 16
15
12
8
7
11
COSC68pF
C51000pF
CSS0.1µF
C40.1µF
C1100pF
RC10k
D1*** L1*10µH
RSENSE**0.033Ω
D2MBRS140T3
COUT100µF10V×2
+
R520k1%
R135.7k 1%
C6100pF
VOUT3.3V
LOAD CURRENT 3A IN NORMAL MODE30mA IN BACKUP MODE
SUMIDA CDRH125-100IRC LR2010-01-R033-FCENTRAL CMDSH-3SUMIDA CDRH73-220SANYO CADNICA N-110AA
***
***†
††1558 01.eps
Figure 268. LTC1558 Backup System with an LTC1435 as the Main System Regulator
Application Note 84
AN84-148
Figure 268 shows a typical application circuit with anLTC1558-3.3 providing backup power to an LTC1435synchronous step-down switching regulator. The backupcircuit components consist of the NiCd cell, R11–R14,C11–C12, L11 and Q11. SW11 and R15 provide a soft orhard reset function.
Normal Mode (Operation from the Main Battery)
During normal operation, the LTC1435 is powered fromthe main battery, which can range from 4.5V to 10V (forexample, a 2-series or 2-series × 2-parallel Li-Ion batterypack, or the like) and generates the 3.3V system output.The LTC1558 operates in standby mode. In standby mode,the LTC1558 BKUP (backup) pin is pulled low and P-channel MOSFET Q11 is on. The NiCd cell is fast chargedby a 15mA current source connected between theLTC1558’s VCC and SW pins. Once the NiCd cell is fullycharged (according to the LTC1558’s gas-gauge counter),the LTC1558 trickle charges the NiCd cell. R14 sets thetrickle-charge current according to the formula I(TRICKLE)= 10 • (VNiCd – 0.5)/R14. The trickle-charge current is setto overcome the NiCd cell’s self-discharge current, therebymaintaining the cell’s full charge.
Backup Mode (Operation from the Backup Battery)
The main battery voltage is scaled down through resistordivider R11–R12 and monitored by the LTC1558 via the FBpin. If the voltage on the FB pin drops 7.5% below theinternal 1.272V reference voltage (due to discharging orexchanging the main battery), the system enters backup
mode. In backup mode, the LTC1558’s internal switchesand L11 form a synchronous boost converter that gener-ates a regulated 4V at VBAK. The LTC1435 operates fromthis supply voltage to generate the 3.3V output voltage.The BKUP pin is pulled high by R13 and Q11 turns off ,leaving its body diode reverse biased. The BKUP pin alsoalerts the system microprocessor. C11, a 47µF capacitor,provides a low impedance bypass to handle the boostconverter’s transient load current; otherwise, the voltagedrop across the NiCd cell’s internal resistance wouldactivate the LTC1558’s undervoltage-lockout function.Table 7 shows several values of VFB vs the VBAK voltage.Figure 269 shows the maximum output power available atthe 3.3V output vs the NiCd cell voltage. Over 100mW ofoutput power is achieved for a NiCd cell voltage greaterthan 1V. Figure 270 shows the backup time vs the 3.3Vload current using a Sanyo Cadnica N-110AA cell (stan-dard series with a capacity of 110mAhrs). Over one hourof backup time is realized for less than 80mW of 3.3Voutput power.
Table 7. VFB and VBAK Voltages
Relative %Below VREF % of VREF VFB VBAK
–0% 100% 1.272V 4.325V
–6% 94% 1.196V 4.065V
–7.5% 92.5% 1.177V 4.000V
BACKUP CELL VOLTAGE (V)1.00
OUTP
UT P
OWER
(mW
)
80
100
120
1.20 1.25 1.30 1.35 1.40
1558_02
60
40
01.05 1.10 1.15
20
180
160
140
VBAK = 4VVOUT = 3.3V
LOAD CURRENT (mA)0
0
BACK
UP T
IME
(MIN
S)
50
100
150
200
300
5 10 15 20
1558_03
25 30
250
VBAK = 4VVOUT = 3.3V
Figure 269. 3.3V Output Power vs Backup Cell Voltage Figure 270. Backup Time vs 3.3V Output Load Current
Application Note 84
AN84-149
DI_EFF_01c.eps
SHDN
VOUT
SW
8
1
5
LTC1174-3.3
GND
VIN
4
6
D1MBR0520LT1
C1100µF10V
C30.1µF
C222µF50V
VIN
VOUT
3
2
7
LBIN
LBOUT
IPGM
L1
68µH
L1 = SUMIDA CDRH74-680
+
+
WHAT EFFICIENCY CURVES DON’T TELLby San-Hwa Chee
Introduction
In switching regulators’ data sheets, there are alwaysefficiency curves that show how efficient the regulatorsare in transforming one voltage to another. Although thesecurves are useful in comparing one regulator to another,they don’t allow a system designer to determine accuratelyhow long batteries will last before they need to be replacedor recharged when they are used as the power source. Thiscomplication arises because the type of batteries used topower the system and the regulator load characteristicstrongly affect the lifetime of the batteries.
In this article, battery lifetime curves are obtained for theLTC1174 and the LTC1433. A Short Introduction to the LTC1174 and LTC1433
The LTC1174 uses a constant off-time architecture toswitch its internal P-channel power MOSFET. The input-to-output voltage ratio sets the on time and requires theinductor current to reach a preset limit. Even at low loadcurrent, the LTC1174 still requires the inductor current toreach the preset limit before it initiates the off-time cycle.Burst Mode operation of the LTC1174 enhances efficiency
DI_EFF_01a.eps
PWRVINPGND
SVIN
COSC
POR
ITH
VOSENSE
VPROG
16
15
14
13
12
11
10
9
LTC1433
D1MBRM5819
C6100µF10V
VOUT3.3V
VINL122µH
C70.1µF
1
2
3
4
5
6
7
8
SSW
NC
BSW
NC
SGND
RUNSS
LB0
LB1
C333µF, 20V
C40.1µF
R1 5.1k
C547pF
C16800pF
C2680pF
L1 = SUMIDA CD54-220
+
+
DI_EFF_01b.eps
PWRVINPGND
SVIN
COSC
POR
ITH
VOSENSE
VPROG
16
15
14
13
12
11
10
9
LTC1433
MBRM-520LT1
MBRM-5819
C6100µF
10V
VOUT3.3V
VIN
L1
L2
C70.1µF
1
2
3
4
5
6
7
8
SSW
NC
BSW
NC
SGND
RUNSS
LB0
LB1
C322µF, 20V
C40.1µF
R1 5.1k
C547pF
C16800pF
C2680pF
100µH
22µH
L1 = SUMIDA CD54-101L2 = SUMIDA CD54-220
+
+
Recovery from Backup Mode to Normal Mode
When a new main battery pack is inserted into the system,Q11’s body diode forward biases. Once the voltage at theFB pin increases to more than 6% below VREF, the boostconverter is disabled and the system returns to normal
Figure 271a. LTC1433 Single-Inductor Configuration
Figure 271b. LTC1433 Dual-Inductor Configuration
Figure 271c. LTC1174 Test Circuit
mode. The BKUP pin pulls low and turns Q11 back on. Thisallows the new battery pack to supply input power to theLTC1435. The LTC1558 now accurately replenishes theamount of charge removed from the NiCd cell through theinternal charger and gas-gauge counter.
Application Note 84
AN84-150
throughout the load-current range by switching only therequired number of cycles to bring the output into regula-tion and then stopping switching (going into sleep mode).When the output voltage has dropped slightly, the switch-ing sequence resumes. By doing this, switching losses are
reduced and are minimized when the load current is low,because the sleep duration is long.
The LTC1433 is a constant-frequency, current mode,monolithic switching regulator in which the inductor peak
INPUT VOLTAGE (V)
75
85
80
100
95
90
EFFI
CIEN
CY (%
)
9
DI_EFF_02a.eps
4 8765
FIGURE 270aFIGURE 270bFIGURE 271
INPUT VOLTAGE (V)
75
85
80
100
95
90
EFFI
CIEN
CY (%
)
9
DI_EFF_02b.eps
4 8765
FIGURE 271aFIGURE 271bFIGURE 271c
TIME (HOURS)
0
1
2
3
4
5
6
BATT
ERY
AND
OUTP
UT V
OLTA
GE (V
)
2.5
DI_EFF_03.eps
0 0.5 1.0 1.5 2.0
LTC1433 WITH DUAL INDUCTORSLTC1433 WITH SINGLE INDUCTORLTC1174HV
OUTPUT VOLTAGE
BATTERYVOLTAGE
BATTERYVOLTAGE
OUTPUT VOLTAGE
TIME (HOURS)
0
1
2
3
4
5
6
BATT
ERY
AND
OUTP
UT V
OLTA
GE (V
)
2.5
DI_EFF_04.eps
0 0.5 1.0 1.5 2.0
LTC1433 WITH DUAL INDUCTORSLTC1433 WITH SINGLE INDUCTORLTC1174HV
OUTPUT VOLTAGE
BATTERYVOLTAGE
OUTPUT VOLTAGE
BATTERYVOLTAGE
TIME (HOURS)
0
1
2
3
4
5
6
BATT
ERY
AND
OUTP
UT V
OLTA
GE (V
)
60
DI_EFF_05.eps
0 10 20 30 40 50
LTC1433 WITH DUAL INDUCTORSLTC1433 WITH SINGLE INDUCTORLTC1174HV
OUTPUT VOLTAGE
BATTERYVOLTAGE
BATTERYVOLTAGE
OUTPUT VOLTAGE
TIME (HOURS)
0
1
2
3
4
5
6
BATT
ERY
AND
OUTP
UT V
OLTA
GE (V
)
20
DI_EFF_06.eps
0 2 4 6 8 12 14 16 1810
LTC1433 WITH DUAL INDUCTORSLTC1433 WITH SINGLE INDUCTORLTC1174HV
OUTPUT VOLTAGE
BATTERYVOLTAGE
OUTPUT VOLTAGE
BATTERYVOLTAGE
Figure 272a. Efficiency Curves forFigure 271’s Circuits, ILOAD = 400mA
Figure 272b. Efficiency Curves forFigure 271’s Circuits, ILOAD = 10mA
Figure 273. Lifetime at ILOAD =400mA—Four AA Alkaline Batteries
Figure 274. Lifetime at ILOAD =400mA—Four AA NiCd Batteries
Figure 275. Lifetime with Load Step from 10mA to 410mA,10% Duty Cycle, TPERIOD = 20s—Four AA Alkaline Batteries
Figure 276. Lifetime with Load Step from 10mA to 410mA,10% Duty Cycle, TPERIOD = 20s—Four AA NiCd Batteries
Application Note 84
AN84-151
current varies according to the load current. In place ofBurst Mode operation, the LTC1433 has an AdaptivePower output stage to enhance its efficiency at low loadcurrent. Under low load conditions, the LTC1433 usesonly a fraction of its power MOSFET, effectively reducingswitching losses without introducing low frequency noisecomponents.
For more information on both parts, consult the datasheets.
The Setup
The circuits in Figures 271a, b and c were used to obtainthe lifetime data. All outputs were set at 3.3V and the powerwas supplied by either four AA alkaline (Eveready No.EN91) or four AA NiCd (Eveready No. CH15) cells or asingle 9V alkaline (Eveready No. EN22) battery. A current-sink load was set up to either draw a constant 400mA orprovide a load-step characteristic. The load stepping op-erated at 0.05Hz, going from 10mA to 410mA with a dutycycle of 10%, providing an average load current of 50mA.
In Figure 271b, the LTC1433 was set up to optimize lowload current efficiency by configuring the Adaptive Poweroutput stage with separate inductors for low and highcurrent operation.
Efficiency curves for each circuit are shown in Figure 272aand 272b. Figures 273 through 278 show the batteryvoltage and regulator output voltage versus time forvarious battery and load combinations.
4-Cell to 3.3V Configuration
Figures 273 and 274 were obtained with a load current of400mA. For Figure 273, the input power to the regulatorwas provided by four AA alkaline batteries, whereas fourAA NiCds were used in Figure 274. The alkaline batterieslasted longer than the NiCds, due to their higher energycapacity. From Figure 274, it is apparent when the NiCdgives up, from the cliff-like shape of the output voltage.
For Figures 275 and 276, a step load was applied to theregulators instead of a DC load. Figure 275 and 276 are thedata obtained for alkaline and NiCd AA cells, respectively.With the average load one-eighth of the previous experi-ment, it would be expected that the lifetime of the alkalinebatteries would be eight times longer or approximately 18hours, but Figure 275 shows a significantly better result.The main reason for this improvement has to do with theinternal resistance of the alkaline cell. At high constant DCload current, heat is dissipated by the internal resistanceof the alkaline batteries. The internal resistance increasesas the batteries voltage decreases, and hence causes moreheat to be dissipated, thus lowering the lifetime.
For the NiCd battery, internal resistance is low and remainsrelatively constant over its life span. Therefore, the lifetimeof the NiCd batteries for the load step case comes out tobe approximately the expected eight times that of a con-stant DC load current.
TIME (HOURS)
0
1
3
2
4
6
5
7
9
8
BATT
ERY
AND
OUTP
UT V
OLTA
GE (V
)
2.5
DI_EFF_07.eps
0 0.5 1.0 1.5 2.0
LTC1433 WITH DUAL INDUCTORSLTC1433 WITH SINGLE INDUCTORLTC1174HV
OUTPUT VOLTAGE
OUTPUT VOLTAGE
BATTERYVOLTAGE
BATTERYVOLTAGE
TIME (HOURS)
0
1
2
3
4
5
6
7
8
9
BATT
ERY
AND
OUTP
UT V
OLTA
GE (V
)
20
DI_EFF_08.eps
0 2 4 6 8 12 14 16 1810
LTC1433 WITH DUAL INDUCTORSLTC1433 WITH SINGLE INDUCTOR
LTC1174HV
OUTPUT VOLTAGE
BATTERYVOLTAGE
OUTPUT VOLTAGE
BATTERYVOLTAGE
Figure 277. Lifetime at ILOAD = 400mA—on a 9V Alkaline Battery Figure 278. Lifetime with Load Step from 10mA to 410mA—one 9V Alkaline Battery (10% Duty Cycle, TPERIOD = 20s)
Application Note 84
AN84-152
The above result indicates that if the load is intermittent innature, the user can operate the device much longer if thepower is provided by alkaline batteries. Again, the NiCdexhibits a sudden “death” at the end of its life, whereas thealkaline shows a much gentler decay. The gentle slopingof the output voltage of Figure 275 towards the end of thebattery life can be attributed to the on-resistance of theswitch when the regulator is in dropout.
For the above load characteristic, where the load is lightmost of the time, making full use of the Adaptive Powermode of the LTC1433 by means of the dual inductorconfiguration helps to squeeze an additional 1.5 hours oflife compared to the single inductor LTC1433 configuration.
Another important point to note is that although theefficiency for the LTC1174 is better than that of the singleinductor configuration of the LTC1433 at 10mA loadcurrent, the LTC1433 lasted 2.9 hours longer than theLTC1174 in Figure 275. The reason for this is that theLTC1174 inductor’s current always ramps up to the preset
value of 600mA whether the load current is at 10mA or at410mA. This high peak inductor current, combined withthe high internal resistance of the alkaline AA cells, short-ens the lifetime. Figure 276 shows that the use time isabout the same for the LTC1174 and the LTC1433 becauseof the low, constant internal resistance of the NiCd batteries.
9V-to-3.3V
The lifetime graphs are shown in Figures 277 and 278.Comparing the data between the 9V and the AA alkalinecells, the lifetime of the AA cells is about 2.5 times longer.This is because the energy capacity of the 9V alkaline ismuch smaller than that of the AA cells. In addition, theinternal resistance of the 9V alkaline is much higher thanthe AA cells, causing more energy to be dissipated as heat.For the load step case, the battery lasted 13.8 times longerthan a constant 400mA load. The dual inductorconfiguration of the LTC1433 lasted about an hour longerthan the single inductor one.
Application Note 84
AN84-153
sroticapaC
rodneV tcudorP enohP LRU
XVA sroticapaCpihC 2630-649)348(sroticapac/stcudorp/moc.procxva.www
XVA sroticapaCmulatnaT 1115-282)702(
stpecnoCcinortcelE sroticapaCmliFV004 0887-245)809( moc.sroticapac-ice.wwwtemeK sroticapaCmulatnaT 4240-689)804( moc.temek.wwwnocraM sroticapaCV/ChgiH 0002-696)748( lmth.nocram/ynapmoc/niam/moc.noc-imehc.www
scinortcelEataruM sroticapaC 1341-732)418( hsilgne/stcudorp/atarum/pj.ro.tenjii.www
nocihciN sroticapaCcitylortcelE 0057-348)748( moc.su-nocihcin.www
cinosanaP sroticapaCyloP 4337-373)417( mth.emoh_sroticapac_stnenopmoc_cinortceleoynaS sroticapaCnocsO 5386-166)916( moc.oedivoynas.www
eugarpS sroticapaC 0414-423)702( moc.eugarpsmoc.www
neduYoyiaT sroticapaCpihC 0514-375)804( moc.neduy-t.www//:ptth
nikoT sroticapaC 0208-234)804( moc.nikot.www
nocimehCdetinU roticapaCcitylortcelE 0002-696)748( niam/moc.noc-imehc.www
nomartiV roticapaCpihCcimareC 1626-862)302( moc.yahsiv.www
amiW sroticapaCmliF/repaP 4742-743)419( moc.asuamiw.www
www.panasonic.com/industrial_oem/electronic_components/
sedoiD
rodneV tcudorP rebmuNenohP LRUttelweHylremrof(tneligA
)drakcaP sDELRI 2130-532)008( ri/moc.tneliga.rotcudnocimes.www
rotcudnocimeSlartneC setercsiDlangiSllamS 0111-534)615( moc.imeslartnec.www
pmaLerutainiMogacihC sDEL 9898-984)102( lmc/moc.gnithgil-ils.www
stcudorPyalpsiDataD sDEL 5186-124)008( moc.sdel-pdd.www
ijuF sedoiDykttohcS 5550-217)102( lmth.e-xedni/gne/pj/oc/cirtceleijuf.www
rotcudnocimeSlareneG sedoiD 0003-748)615( moc.imesneg.www
*alorotoM setercsiD 7442-144)008( lmth.xedni/stcudorp/moc.sps-tom.www
*rotcudnocimeSNO setercsiD 0150-947)804( emoh/moc.imesno.www
cinosanaP sDEL 7125-843)102( /srotcudnocimes/meo_lairtsudni/moc.cinosanap.wwwmth.emoh_rotcudnocimes
cimeT sedoiDotohPRI 0075-079)804( moc.cimet.www
yahsiV langiSllamS/reneZsedoiD moc.yahsiv.www
xeteZ setercsiDlangiSllamS 0017-345)615( moc.xetez.www
.rotcudnocimeSNOybderutcafunamwoneraalorotoMybderutcafunamylremrofsetercsiD*0002yraunaJfosadeganahcneebtonevahsrebmuntraP
(650) 665-9301
APPENDIX A: COMPONENT VENDOR CONTACTS
The tables on this and the following pages list contactinformation for vendors of non-LTC parts used in theapplication circuits in this publication. In some cases,
components from other vendors may also be suitable. Forinformation on component selection, consult the text ofthe respective articles and the appropriate LTC data sheets.
Application Note 84
AN84-154
sremrofsnarTdnasrotcudnI
rodneV tcudorP rebmuNenohP LRU
naveleDIPA srotcudnI 0063-256)617( moc.naveled.www
scinortcelEHB srotcudnI 0959-498)216( moc.scinortcelehb.www
seigolonhceTIB sremrofsnarT 6562-744)417( moc.seigolonhcetib.www
tfarclioC srotcudnI 0046-936)748( moc.tfarclioc.www
scinortlioC sremrofsnarT/srotcudnI 6787-142)165( moc.scinortlioc.www
elaD sremrofsnarT/srotcudnI 7261-566)506( srotcudni#lmth.pf/pf/moc.yahsiv.www
adnawoG srotcudnI 4322-235)617( moc.adnawog.www
mocdiM sremrofsnarT/srotcudnI 1662-346)008( moc.cni-mocdim.www
scinortcelEataruM ,srotcudnI 1341-732)418( moc.atarum.www
cinosanaP sremrofsnarT/srotcudnI 4337-373)417( /stnenopmoc_cinortcele/meo_lairtsudni/moc.cinosanap.wwwmth.sremrofsnart_dna_slioc_srotcudni_stnenopmoc_cinortcele
spilihP srotcudnI 1182-642)419( moc.spilihp.stnenopmoc.mca.www
spilihP srotcudnIranalP 6302-742)419( moc.spilihp.stnenopmoc.mca.www
esluP srotcudnI 0018-476)916( moc.gneeslup.www
adimuS srotcudnI 7660-659)748( adimus/moc.knilnapaj.www
nikoT srotcudnI 0208-234)804( moc.nikot.www
cigoL
rodneV tcudorP rebmuNenohP LRU
dlihcriaF cigoL 2054-577)702( moc.imesdlihcriaf.www
)sirraHylremrof(lisretnI cigoL 7477-244)008( moc.lisretni.www
alorotoM* cigoL 7442-144)008( lmth.xedni/stcudorp/moc.sps-tom.www
rotcudnocimeSNO* cigoL 0150-947)804( emoh/moc.imesno.www
abihsoT cigoLcigoLetaGelgniS
/0002-554)949(0002-554)417( ceat/moc.abihsot.www
0002yraunaJfosasrebmuntrapnisegnahconneebevahereht;rotcudnocimeSNOybderutcafunamwoneraalorotoMybybderutcafunamylremrofseciveDcigoL*
srotsiseR
rodneV tcudorP rebmuNenohP LRUyeldarBnellA srotsiseRnobraC 8884-295)008( moc.ba.www
XVA srotsiseRpihC 4250-649)348( mth.rtsrpihc/srotsiser/stcudorp/moc.procxva.www
seigolonhceTIB rotsiseR/srotsiseRskrowteN 5432-744)417( moc.seigolonhcetib.www
snruoB sPIS,sretemoitnetoP 3527-057)108( moc.snruob.www
elaD srotsiseResneS 1039-566)506( moc.liofyahsiv.wwwmoc.yahsiv.wwwro
CRI srotsiseResneS 0097-299)163( moc.ttcri.www
nellAGR srotsiseRedixOlateM 0038-567)818( moc.ocagr.www
DAT srotsiseRpihC 1251-805)008( moc.mocdat.www
neduYoyiaT srotsiseRpihC 0514-375)804( moc.neduy-t.www
ygolonhceTmliFnihT srotsiseRpihCmliFnihT 5448-526)705( moc.mlif-niht.www
socoT sretemoitnetoPDMS 4666-488)748( moc.socot.www
Application Note 84
AN84-155
srotsisnarT
rodneV tcudorP rebmuNenohP LRU
rotcudnocimeSlartneC setercsiDlangiSllamS 0111-534)615( moc.imeslartnec.www
dlihcriaF sTEFSOM 6212-228)804( moc.imesdlihcriaf.www
RI sTEFSOM 1333-223)013( moc.fri.www
*alorotoM setercsiD 7442-144)008( lmth.xedni/stcudorp/moc.sps-tom.www
*rotcudnocimeSNO setercsiD 0150-947)804( emoh/moc.imesno.www
spilihP setercsiD 7244-767)104( moc.spilihp.srotcudnocimes.su-www
xinociliS sTEFSOM 5655-455)008( moc.xinocilis.www
xeteZ setercsiDlangiSllamS 0017-345)615( moc.xetez.www.0002yraunaJfosasrebmuntrapnisegnahconeraerehT;rotcudnocimeSNOybderutcafunamwoneraalorotoMybderucafunamylremrofsetercsiD* t
suoenallecsiM
rodneV tcudorP rebmuNenohP LRU
divaA skniStaeH 5662-655)417( moc.divaa.www
nospE slatsyrC 0036-787)013( moc.nospe.aee.wwwnoenifnI
snemeiSylremrof()rotcudnocimeS
scinortceleotpO 0197-752)801( mth.tnetnoc/otpo/su/moc.noenifni.www
.cnI,scitengaM .cte,seroCdioroT 4893-542)008( moc.cni-gam.www
scinortcelEFM srotallicsOlatsyrC 0756-675)419( moc.celefm.www
scinortcelEataruM seciveDFR 9875-334)077( moc.atarum.www
scinortceleotpOTQ sehctiwSFR 0441-027)804( moc.otpotq.www
mehcyaR sesuF 6584-722)008( moc.mehcyar.www
seciveDorciMFR srotcudnocimeSFR 3321-466)633( moc.dmfr.www
ameteK/ITR srosserppuSegruS 1800-036)417( moc.proc-itr.eitr.www
retruhcS sredloHdnasesuF 1136-877)707( moc.cniretruhcs.www
yollamrehT skniStaeH 1234-342)279( moc.yollamreht.www
okoT stcudorPFR 0343-996)748( moc.maokot.www
Application Note 84
AN84-156
B
Battery BackupLTC1558 System with LTC1435 Main System Regulator 147
Battery Chargers 114–115Additional Feature Circuits
LT1512/LT1513, Shutdown-Controlled Disconnect 117LTC1435/LT1620, Shutdown when Input Power is
Removed 125LTC1510, Doubles as Main System Regulator 128
GeneralLT1511, Mod for NiCd and NiMH Charging 115LT1512, 0.5 Amp 116LT1635, 1A Shunt 129
Lead-AcidLT1776/LT1620 , Wide VIN Range, High Efficiency 49
Lithium-IonLT1510, 1–2 Cell 130LT1510, 3-Cell, without Precision Resistors 118LT1511, 3 Amp 114LT1512, 50mA/400mA Programmable 117LTC1435/LT1620, 3–5 Cell 122
NiCdLT1510, 3-Cell with –∆V Termination 120
TestingConstant-Voltage Battery Simulator 121
Battery Simulators 121Battery Supervisor
Single Cell Li-Ion 132Bench Supply
100V/2A Constant Voltage, Constant Current 146
C
Component VendorsCapacitors 153Diodes 153Inductors 154Logic 154Miscellaneous 155Resistors 154Transformers 154
Current Sensor 145
L
Linear Regulators. See Regulators—Linear
M
Micropower Switching Regulators. See Regulators—Switching(Micropower)
Miscellaneous 141–149Modulator
Switch-Frequencyfor LTC1436-PLL 83
P
Power Magagement 134–141LTC1479 PowerPath Controller
3-Diode Mode 136Block Digram 134
PowerPath Switch DriverLTC1473, Dual-Battery 137, 138
SMBusLTC1623, Controls P-Channel Switch 140LTC1623, Controls Two High-Side Switches 140LTC1710, Switches Two Peripherals 141LTC1710, Switches Two Peripherals with Different
Voltages 141System
Dual Li-Ion Battery 134VID Controlled
LT1575/LTC1706, LDO with Adjustable Output Voltage 144Power Supply. See Regulators—Linear; Regulators—
Switching; Regulators—Switching (Micropower)
R
Regulators—Linear 107–111Adjustable
LT1575/LTC1706, LDO with Adjustable Output Voltage 144Battery Backup
LT1579, 6V to 5V/300mA 111LT1579, with Added Latch for Shutdown 112
Low DropoutLT1573, 3.3V/5A Microprocessor Supply 108LT1575, 1.27V–2.03V VID Controlled 34
Index
Application Note 84
AN84-157
LT1575, 5V to 3.3V with Current Limit 110LT1575, 5V to 3.3V/5A 109
Microprocessor SupplyLT1573, 3.3V/5A 108LT1577, Dual Regulator for Split-Plane Systems 110
MultioutputLT1577, Dual Regulator for Split-Plane Systems 110
Regulators—Switching. See also Regulators—Switching(Micropower)
Boost 51–53LT1339, 5V In, 28V/6A Out Synchronous 28LT1370, 5V In, 12V/2A Out 53LT1377, 4V–10V In, ±12V/100mA Out 51, 52LT1533, 3.3V to 5V/350mA Boost Converter 72LTC1266, 2.5V–4.2V In, 5V/2A Out 55LTC1624, 5V In, 12V/1A Out 20
Buck 4–5012V to 3.3V/9A Hybrid 17LT1339, 10V–18V In, 5V/50A Out 26LT1339, 48V In, 5V/50A Out 27LT1374, 6V–25V In, 5V/4.25A Out 23LT1425, 12V to 5V/1A Isolated Supply 67LT1506, 5V In, 3.3V/4A Out 29LT1676, 12V-48V In, 5V Out 48LT1676, 12V-48V In, 5V/0.5A Out 46LT1676, Minimum Component-Count 46LT1676/LTC1440, Burst Mode Configuration 48LT1776, 10V–30V In, 5V/0.4A Out 47LT1776, Minimum PC Board Area 47LTC1266, 12V In, 3.3V/12A Out 7LTC1266, 24V In, 3.3V/12A Out 8LTC1430, 3.3V In, 1.9V/6A Out 22LTC1430, Dual, Synchronized 17LTC1430A, 2.5V/30A, 2-Phase Synchronous 40LTC1433, 3.6V–12V In, 3.3V/600mA Out 11LTC1435, 18V–28V In, 14V/15A Out 12LTC1435, 5.5V–28V In, 2.9V/2.65V Out 9LTC1435A 4.5V–22V In, 1.3V–2V/7A Out 36LTC1435A, 4.5V–22V In, 1.3V–2V/7A Out 32LTC1435A, 4.5V–22V In, 1.6V/3A Out 35LTC1436A-PLL, 4.5V–22V In, 1.3V–2V/5A Out 33LTC1436A-PLL/LTC1706, 4.5V–22V to 1.3V–2V/5A 143LTC1439, 5.2V–25V In, 5V/3A, 3.3V/3A, 2.9V/2.5A Out 5LTC1473, 28V In, 5V/3A and 12V/250mA Out 4LTC1504, 5V In, 3.3V/0.5A Out 24LTC1504, Improved Transient Response 25
LTC1504, SCSI-2 Terminator 25LTC1504, Supply Splitter 25LTC1553, 5V In, 1.8V–3.5V/14A Out 14LTC1558, Battery Backup with LTC1435 Main System
Regulator 147LTC1622, 2.5V–8.5V In, 2.5V/1.5A Out 43LTC1622, Improved Transient Response 44LTC1624, 4.5V–25V In, 3.3V/2A Out 20LTC1624, 4.8V–20V In, 1.3V–3.0V Out 32LTC1624/LTC1706, 4.8V–20V to 1.3V–2.0V 143LTC1625, 12V–28V In, 12V/2.2A Out 39LTC1625, 5V–28V In, 2.5V/5A Out 38LTC1627, 1.8V/0.3A/3.3V/100mA 37LTC1627, 2 Li-Ion to 3.3V/0.5A 37LTC1627, Single Li-Ion to 2.5V/0.5A 37
Buck-Boost 56–57LT1371, 2.7V–20V In, 5V Out 56LTC1515, 3- or 4-Cell to 3.3V or 5V/50mA 57
Charge PumpLT1054, Generates –5V for LTC1419 ADC 59LTC1430, Assisted by LTC1517 22
Current-SharingLT1506, 6V–15V In, 5V/12A Out 30
Efficiency 149–152Flyback 65–69
LT1172, 10V–15V In, 24V/200mA Out Isolated Flyback 68LT1316, –48V to 5V Flyback 63LT1425, 5V to –9V/250mA Isolated LAN Supply 65LT1425, Fully Isolated ±15V, ±600mA Supply 66LTC1624, 4.75V–24V In, ±50V/75mA Out 69
ForwardLT1339, 15V–25V In, 5V/6A Out 27
Hybrid12V to 3.3V/9A Switcher plus Linear 17
Inverting 57–64Inverting, Negative-to-Positive
LT1316, –48V to 5V Flyback 63LT1425, –36V to –72V In, 5V/2A Out Telecom Supply 67LT1680, –48V to 5V/6A Telecom Supply 60
Inverting, Positive-to-NegativeLT1172, 12V to –48V/120mA Telecom Supply 61LT1370, 2.7V–13V In, –5V/3A Out 54LT1614, 5V In, –5V/200mA Out 62LT1776, 10V-28V In, –5V/300mA Out 50LTC1373, 5V to –5V for LTC1419 ADC 58LTC1433, 3V–7.5V In, –5.0V Out 11
Application Note 84
AN84-158
Regulators—Switching (continued)Isolated
LT1172, 10V–15V In, 24V/200mA Out Isolated Flyback 68LT1339, 15V–25V In, 5V/6A Out 27LT1425, 12V to 5V/1A Isolated Supply 67LT1425, 5V to –9V/250mA Isolated LAN Supply 65LT1425, Fully Isolated ±15V, ±600mA Supply 66
LCD BiasLT1316, 20V/5mA/–10V/5mA LCD 80
Low Noise 70–76LT1533, 24V to 5V/2A Converter 74LT1533, 3.3V to 5V/350mA Boost Converter 72LT1533, 5V to ±12V/80mA DC/DC Converter 71LT1533, 5V to 12V/200mA Push-Pull Converter 70LT1533, 5V to 12V/5A Converter 75LTC1436-PLL, 5V/3A/3.3V/0.1A Supply 81
Microprocessor Supply12V to 3.3V/9A Hybrid 172.9V Regulator for Portable Pentium Processor 9LTC1435/LTC1706, Pentium II Processor Supply 142Mobil Pentium II VID Power Converter 32
Multioutput 76–84LT1316, 20V/5mA/–10V/5mA LCD Bias Supply 80LT1377, 4V–10V In, ±12V/100mA Out 51, 52LT1425, Fully Isolated ±15V, ±600mA Supply 66LT1533, 5V to ±12V/80mA DC/DC Converter 71LT1776, Dual-Output SEPIC (5V/–5V) 50LTC1263/LTC1266, 3.3V/5A/12V/60mA Supply 79LTC1435, 5V/0.1A, 3.3V/0.5A, –5V/0.5A Supply 77LTC1436-PLL, 5V/3A/3.3V/0.1A Supply 81LTC1439, 5.2V–25V In, 5V/3A, 3.3V/3A, 2.9V/2.5A Out 5LTC1473, 28V In, 5V/3A and 12V/250mA Out 4LTC1538-AUX, 3.3V/3.5A, 5V/3A, 12V/120mA, 5V/20mA 76LTC1624, 4.75V–24V In, ±50V/75mA Out 69LTC1627, 1.8V/0.3A/3.3V/100mA 37
No RSENSELTC1625, 12V–28V In, 12V/2.2A Out 39LTC1625, 5V–28V In, 2.5V/5A Out 38
PolyPhaseLTC1430A, 2.5V/30A, 2-Phase Synchronous 40
SEPIC100V/2A Bench Supply 146LT1370, 2 Li-Ion Cells to 5V/2.9A 54LT1776, Dual Output (5V/–5V) 50LTC1624, 5V–15V In, 12V/0.5A Out 21
Step-Down. See Regulators—Switching: BuckStep-Up. See Regulators—Switching: Boost; Regulators—
Switching: Flyback
Supply SplitterLTC1504, 5V to 2.5V/±500mA 25
Switched CapacitorLTC1515, 3- or 4-Cell to 3.3V or 5V/50mA 57
SynchronizedLTC1430, Dual Buck 17LTC1436A-PLL/LTC1706, 4.5V–22V to 1.3V–2V/5A 143
TelecomLT1172, 12V to –48V/120mA Telecom Supply 61LT1425, –36V to –72V In, 5V/2A Out Telecom Supply 67LT1680, –48V to 5V/6A Telecom Supply 60LTC1504, SCSI-2 Terminator 25
VID Voltage ControlledLTC1435A, 4.5V–22V In, 1.3V–2V/7A Out 32LTC1436A-PLL, 4.5V–22V In, 1.3V–2V/5A Out 33LTC1436A-PLL/LTC1706, 4.5V–22V to 1.3V–2V/5A 143LTC1553, 5V In, 1.8V–3.5V/14A Out 14LTC1624, 4.8V–20V In, 1.3V–3.0V Out 32LTC1624/LTC1706 4.8V–20V to 1.3V–2.0V 143
Regulators—Switching (Micropower) 852-Cell Digital Camera Supply 101Boost
LT1307, Single-Cell to 3.3V/75mA Converter 86LT1307, Single-Cell to 3.3V/75mA Converter with Output RC
Filter 86LT1308, Single-Cell Li-Ion to 5V/1A 100LT1317, 2-Cell to 5V/200mA 102LT1317B, 33V/10mA Varactor Bias Suppy 105LT1610, 2-Cell to 5V/100mA 104LT1610, Single Cell to 3V/30mA 103Single-Cell NiCd to 3.3V/400mA 100
BuckLTC1174, 9V to 5V Converter 85LTC1474, 4V–18V In, 3.3V/200mA Out 92LTC1475, with Push-Button On/Off Control 93LTC1626, 2.7V–6V In, 2.5V/0.25A Out 97LTC1626, 3- or 4-Cell NiCd/NiMH to 2.5V/0.25A 97LTC1626, Single Li-Ion Cell to 2.5V/0.25A 96
Buck-BoostLTC1626, Single Li-Ion Cell to 3.3V/500mA 98
Charge PumpLTC1263, Flash Memory VPP Generator 90LTC1263, Split-Supply Generator (12V/–7V) 91LTC1516, 2-Cell to 5V/20mA 106LTC1516, Ultralow Quiescent Current 5V Supply 89
Flyback1.8V-6V to 9V, for Digital Panel Meters 94
Application Note 84
AN84-159Information furnished by Linear Technology Corporation is believed to be accurate and reliable.However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
Isolated1.8V-6V to 9V, for Digital Panel Meters 94
LED DriverLT1307, 25mA LED Driver 89
MultioutputLT1317, Single-Cell Li-Ion to ±4V 103
Negative BuckLT1307B, 7V–25V In, 5V/400mA Out 99
SEPICLT1317, Single-Cell Li-Ion to ±4V 103LT1610, Sigle-Cell Li-Ion to 3.3V/100mA 105
Switched CapacitorLTC1514, 2.7V–10V In, 3.3V and 5V Out 96LTC1515, 4-Cells to 5V/50mA or 3.3V/50mA 95LTC1516, 2-Cell to 5V/20mA 106
VPP GeneratorLTC1263, for 2 Flash Memory Chips 90
S
SwitchesHigh-Side
LTC1177, Short-Circuit Protected 139LTC1623, SMBus Controlled 140
P-ChannelLTC1623, SMBus Controlled 140
PC CardLTC1623, 3.3V/5V Switch Matrix 140
Switching Regulators. See Regulators—Switching
T
TransformerDetails
of LT1339 5V/6A Forward Converter 28
Application Note 84
AN84-160Linear Technology Corporation1630 McCarthy Blvd., Milpitas, CA 95035-7417(408) 432-1900 FAX: (408) 434-0507 www.linear-tech.com
an84f LT/TP 0400 4K • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 2000