A good agreement between measured and simulated results isobserved over a wide frequency range. The minimum measuredinsertion loss is 0.8 dB, including connector and feed line losses.The transmission zero at 3.7 GHz can be depicted from Figure 3 asexpected. The transmission zero at the upper stopband is shiftedslightly towards higher frequency region. Though the filter isequivalent to a parallel-coupled bandpass filter, there is no re-peated passband at twice the center frequency. The proposedstructure exhibits a 3 dB fractional bandwidth of 60%. The frac-tional bandwidth is defined as

�f ��f

f0� 2 �

� fh � f1�

� fh � f1�, (7)

where fl and fh are the low and high end of the frequency band.The developed filter is intended to be used in a microwave

camera for medical applications.

4. HIGHER ORDER FILTERS

For some applications, higher stopband attenuation may be re-quired. This can be achieved by increasing the order of the filter asshown in Figure 4.

The design of higher order filters is based on the same proce-dure. As shown in Figure 4, two identical microstrip structures areconnected using coupled transmission lines. The coupling levelbetween them is chosen to achieve characteristics presented inFigure 5. The gap for the central coupled line is equal to 0.25 mm;the parameters of all other components remain the same as for thestructure shown in Figure 1.

As expected, a filter using two cascaded structures provides animproved stopband response in comparison to filter in Figure 1.Insertion losses in high and low stopband regions are better than 40and 35 dB, respectively. The lower- and upper-band selectivity ofthis bandpass filter is about 143 and 77 dB/GHz, respectively, anddepends on transmission zeros locations.

The second harmonic suppression is better than 26 dB. Thefilter exhibits a relative 3 dB bandwidth of 59%. The centerfrequency is f0 � 5.6 GHz.

Further improvements of the filter presented in Figure 4 wouldstill be possible with regards to return loss by additional tuninglevels.

5. CONCLUSION

A new structure for a planar microwave filter is proposed, based oncoupled-line sections and quasi-lumped element resonator. Thisbandpass filter has a compact footprint, and exhibits good stopbandrejection with no repeated passband at twice the center frequencyin comparison with the traditional coupled-line filter. By introduc-ing the quasi-lumped element resonator, two transmission zeros atupper and lower stopbands are created, with adjustable locations oftransmission zeros. This resonator consists of quasi-lumped ele-ments, taking advantage of their flexibility in microwave band.The fabrication technology for the filter is simple, inexpensive, anddoes not require via-holes. An analytical representation of the filtercharacteristic has been derived and used in the analysis.

A filter has been designed and fabricated to demonstrate thecompactness and the improved performance using this technique.The passband filter centered at 5.58 GHz with a 3 dB bandwidth of60% is realized with a total area of �/6 � �/4. Measured andsimulated results exhibit good agreement.

Proposed method is suitable for a wideband as well as fornarrowband applications. Further, stopband characteristics im-provement is possible by cascading the proposed structure.

REFERENCES

1. J.-S. Hong and M.J. Lancaster, Microstrip filters for RF/microwaveapplications, Wiley, New York, 2001.

2. C.-L. Hsu, F.-C. Hsu, and J.-T. Kuo, Microstrip bandpass filters forultra-wideband (UWB) wireless communications, IEEE MTT-S IntMicrowave Symp Digest, Long Beach, CA (2005), 679-682.

3. J. Gao, et al., Short-circuited CPW multiple-mode resonator for ultra-wideband (UWB) bandpass filter, IEEE Microwave Wireless ComponLett 16 (2006), 104-106.

4. J.-S. Hong and H. Shaman, An optimum ultra-wideband microstripfilter, Microwave Opt Technol Lett 47 (2005), 230-233.

5. R. Mongia, I. Bahl, and P. Bhartia, RF and microwave coupled linecircuits, Norwood, MA, Artech House Microwave Library, 1999.

6. V. Zhurbenko, V. Krozer, and P. Meincke, Miniature microwave band-pass filter based on EBG structures, Proceedings of the 36th EuropeanMicrowave Conference, 2006, pp. 792-794.

7. G. I. Zysman and A. K. Johnson, Coupled transmission line networks ininhomogeneous dielectric medium, Trans Microwave Theory Tech 17(1969), 753-759.

8. I. Bahl, Lumped elements for RF and microwave circuits, Artech HouseMicrowave Library, Norwood, MA, 2003.

© 2007 Wiley Periodicals, Inc.

INTERNAL ANTENNA DESIGN OF 900MHZ-BAND MOBILE RADIOFREQUENCY IDENTIFICATION SYSTEM

Yongjin Kim,1 Ick-Jae Yoon,2 and Youngeil Kim2

1 Department of Electrical Information and Science, Inha TechnicalCollege, 253 Yonghyun-dong, Nam-gu, Inchon 402–752, Korea2 Embedded Systems Solution Lab., Samsung Advanced Institute ofTechnology, MT 14–1, Nongseo-dong, Giheung-Gu, Yongin-Si,Gyeonggi-do 446–712, Korea

Received 23 February 2007

ABSTRACT: The development of Radio Frequency Identification(RFID) system for tracking and controlling goods and products, andobtaining information through transponders (tags) from people and ob-jects are growing very rapidly in modern communication area. Cur-rently, researchers and engineers have taken a growing interest in put-ting a RFID reader system into a mobile cellular phone. In this study,

Figure 5 Transmission and reflection coefficient characteristics for thehigher order bandpass filter illustrated in Figure 4. [Color figure can beviewed in the online issue, which is available at www.interscience.wiley.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 9, September 2007 2079

we propose a wide band antenna for mRFID reader application. Theproposed antenna is the wideband PIFA antenna that can cover from800 MHz cellular frequency band to 900 MHz-band RFID frequencyband. © 2007 Wiley Periodicals, Inc. Microwave Opt Technol Lett49: 2079 –2082, 2007; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.22710

Key words: RFID reader antenna; wide band antenna; hand-set antenna

1. INTRODUCTION

The development of Radio Frequency Identification (RFID) sys-tem for tracking and controlling goods and products, and obtaininginformation through transponders (tags) from people and objectsare growing very rapidly in modern communication area [1]. Inantenna technology, antennas for RFID tag are mainly studied anddeveloped to reduce the size of antenna and to match input im-pedance between chip and tag antenna [2–4]. Currently, research-ers and engineers have taken a growing interest in putting a RFIDsystem into a mobile cellular phone. It makes people obtain usefulinformation from the RFID tag easily in ubiquitous society. In thatcase, the RFID reader antenna, that may be inserted in or printedon the circuit board of the mobile hand-set should be very small.On the contrary, the smaller antenna can cause a decrease ofantenna gain and bandwidth.

In this letter, we propose two bands, the wideband PIFAantenna that can cover from 800 MHz cellular frequency band to900 MHz Korea RFID frequency band. This antenna can be usedeffectively, when only single antenna can be installed inside mo-bile phone due to the physical limitation of the cellular phone bodydimension. This antenna can be mounted conformally or inserted

easily inside the mobile phone due to small size. The RFID readrange is also measured using the proposed antenna to verify theantenna performance.

2. ANTENNA STRUCTURE AND DESIGN

When the mRFID reader system is inserted into the cellular phone,reader antenna design is a very challenging issue. To provideapplicable read range, the reader antenna gain would be as high aspossible. However, the cellular phone size is very limited and isgetting smaller and smaller. Therefore, inserting an additionalantenna for 900 MHz-band mRFID reader seems inapplicable.

In this letter, we propose a wide band internal antenna that cancover frequency range of 800 MHz cellular band and 900 MHzmRFID band. When using the wide band antenna, only one antennais needed for the both services (cellular communication and mRFID).The size of the proposed antenna is very similar with the conventionalcommunication internal antenna. Therefore, no additional antennaspace is needed. It is true that modification of the system of cellularcommunication is essential to control communication and mRFIDsystem. However, physical size of the cellular phone is also a criticalissue in the aesthetic view. Using the wide band antenna for bothsystems can be one of the effective solutions.

The antenna geometry of the proposed antenna is shown inFigure 1. The detailed dimensions are shown in Table 1. One of themain goals of this antenna is having a wide band characteristic(target band width is mode than 100 MHz at 900 MHz band) tocover from 824 to 914 MHz.

As shown in Figure 1, the antenna lies in the x-z plane and thenormal direction is the y axis. The antenna carrier size is 40 �10 � 10 mm3 and the ground plane size of 40 � 65 � 1 mm3 isselected that is representative of a typical cellular phone circuit

Figure 1 Proposed wide band internal antenna

TABLE 1 Detailed Dimensions of the Proposed Antenna

Length (mm) Length (mm) Length (mm)

GW 40 Aw1 6 A12 12.16GL 65 Aw2 8 A13 5Acw 56 Aw3 13 Awt 7Aw 34 Aw4 4Ach 10 Al1 9

Figure 2 VSWR data of the proposed antenna (Awt is the varyingparameter). : Awt � 7 mm – – – – – : Awt � 10 mm ..............:Awt � 13 mm

2080 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 9, September 2007 DOI 10.1002/mop

board. The PIFA type radiating element is on the top layer of thesubstrate (FR4-Epoxy board, �r � 4.4). The geometry of theantenna is mainly dependent on the total length of the antenna (Aw,Al2, Aw2, Al4, Al1, etc.). The effective antenna length is �88 mmwhich corresponds to 0.26 � at 900 MHz. The simulation result ofinput impedance is shown in Figure 2.

Input impedance is shown as a function of frequency withAwt as the varying parameter. The detailed bandwidth compar-ison is shown in Table 2. As shown in Figure 2 and Table 2, theproposed antenna has a wide bandwidth characteristic and canbe applied for cellular communication and mRFID applicationat the same time.

3. MEASURED RESULTS

The prototype of the proposed antenna is in real hand-setdevice, and the photo of the antenna with the device is shownin Figure 3. There are some geometrical variations in theantenna geometrical parameters due to applying it in an actualhand set environment. When we put the proto-type antenna inan actual hand set, the resonance frequency is used to movedown to the low frequency range (frequency shift range de-pends on the model characteristics). The measured input im-pedance is shown in Figure 4. The impedance bandwidth forVSWR less than 3 of the proposed antenna with real hand-set is275 MHz (817.5–1090 MHz). This is �30% bandwidth at 900MHz. It seems that this bandwidth is wide enough to use bothcellular communication and mRFID application. Radiation pat-tern of the antenna at 912 MHz is shown in Figure 5. In the x-yplane, the radiation pattern is fairly omni-directional and max-imum gain is �0.86 dBi at �98°. The gain of the RFID servicedirection, at 90° is �1.4 dBi. The relation between the readerantenna gain and read range is presented in the next section.

Read range is the maximum distance that REID reader canread/write useful information from/to the tag [1]. In modulated

backscattering RFID system, transmit power from the readershould also be high enough to activate tag chip. Tag antennashould also match well with tag chip [5, 6]. The measured readrange using the proposed antenna is 85 cm with 25 dBmtransmit power. The RFID system used for this measurement isa commercial RFID system made by Alien cooperation.

4. CONCLUSION

In this study, a wide band antenna for mRFID reader application isintroduced. The proposed antenna is the wide band PIFA antenna thatcan cover from 800 MHz cellular frequency band to 900 MHz KoreaRFID frequency band. The radiation pattern is fairly omni-directional

TABLE 2 Antenna Bandwidth as a Function of Awt

Awt (mm) fr (MHz)Bandwidth (MHz)

(VSWR � 2)

7 924 10310 900 12813 882 83

Figure 3 Photo of the proposed antenna with real hand-set

Figure 4 Measured VSWR data of the proposed antenna. [Color figurecan be viewed in the online issue, which is available at www.interscience.wiley.com]

Figure 5 Measured radiation pattern of the proposed antenna at 912MHz (x-y plane). [Color figure can be viewed in the online issue, which isavailable at www.interscience.wiley.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 9, September 2007 2081

and maximum gain is �0.86 dBi. The gain of the RFID servicedirection, at 90° is �1.4 dBi. The measured read range of 85 cm isobtained.

REFERENCES

1. K. Finkenzeller, RFID handbook, 2nd ed., Wiley, West Sussex, En-gland, 2003.

2. J. Siden, P. Jonsson, T. Olsson, and G. Wang, Performance degradationof RFID system due to distortion in RFID tag antenna, In: Internationalconference on microwave and telecommunication technology, 2001, pp.371–373.

3. L. Ukkonen, D. Engles, L. Sydanheimo, and M. Kivikoski, Planarwire-type inverted-F RFID tag antenna mountable on metallic objects,In: International symposium on antennas and propagation and USNC/USRI national radio science meeting, June 2004, Monterey, CA, Vol. 1,pp. 101–104.

4. A. Delichatsios, et al., Folded microstrip patch-type RFID tag antennamounted on a box corner, In: International symposium on antennas andpropagation and USNC/USRI national radio science meeting, July2006, Albuquerque, NM, Vol. 1, pp. 3213–3216.

5. P.V. Nikitin and K.V.S. Rao, Measurement of backscattering fromRFID tags, In: Proceedings of antenna measurement techniques asso-ciation, October 2005, New port, RI, Vol. 1, pp. 300–303.

6. P.V. Nikitin and K.V.S. Rao, Performance limitations of passive UHFRFID systems, In: International symposium on antennas and propaga-tion and USNC/USRI national radio science meeting, June 2006, Vol. 1,pp. 1011–1014

© 2007 Wiley Periodicals, Inc.

ANALYTICAL ATTENUATIONCALCULATION OF ASYMMETRICALCOPLANAR WAVEGUIDE WITH FINITE-EXTENT GROUND PLANES FORCOPLANAR WAVEGUIDE MODE

Mehmet Duyar,1 Volkan Akan,2 Erdem Yazgan,3 andMehmet Bayrak4

1 Nigde Industry and Trade Governer Department, TR-51100,Nigde, Turkey2 TUBITAK SPACE, Satellite Technologies Group, Inonu Bulvari,ODTU Campus, TR-06531 Ankara Turkey3 Hacettepe Universitesi Elektrik ve, Elektronik Muh. Bolumu TR-06800, Beytepe, Ankara/Turkiye4 Department of Electrical and Electronics Engineering, Faculty ofEngineering, Selcuk University, Konya, Turkey

Received 23 February 2007

ABSTRACT: In this study, fast and new analytical formulations havebeen presented for correct loss calculations in asymmetrical coplanarwaveguides (CPW) with finite-extent ground planes for CPW mode. Togeneralize loss analysis for CPW mode, not only gaps but also finiteground planes are assumed as asymmetric. Moreover, dielectric thick-ness and ground plane widths are considered as finite for this transmis-sion line. Another important concept presented in this article is that theproposed expressions are easily applicable for CAD platforms. Throughthe analyses, conformal mapping techniques (CMT) have been em-ployed. The obtained results in this work have been compared with theresults of other similar studies (spectral domain method, mode match-ing, etc.) available in the literature. These comparisons indicate that theexpressions developed in this study are found to be more suitable foraccurate calculations of loss as well as the characterization of asymmet-rical CPW with finite-extent ground planes. © 2007 Wiley Periodicals,Inc. Microwave Opt Technol Lett 49: 2082–2087, 2007; Published on-line in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.22709

Key words: asymmetrical CPW; attenuation constant; conformal map-ping; loss

1. INTRODUCTION

Recently, the usage of coplanar transmission lines in practice hasbecome very popular in designing and manufacturing the micro-wave integrated circuits and the passive microwave components.The reason for this may be attributed to some important advan-tages of the coplanar waveguides (CPW), such as low radiationloss, no need for any ground hole applications, easiness in fabri-cations etc., over the other transmission lines types. So far, differ-ent types of CPW have been analyzed in the literature to calculatethe electrical parameters [1-4]. However, the work given in thisarticle takes the importance of the finite conductor thickness intoaccount at the beginning of the design procedure. Ghione hassummarized in length the conductor loss contributions earlier inthe CPW in his study [5]. As indicated in [5], various analysismethods to predict the CPW conductor losses has been reported inthe literature [6–9]. In addition, conductor loss calculations forsymmetric CPW with finite ground planes were also presented in[10–13] using different analytical and numerical methods.

Holloway and Kuester [14] gave closed form expressions forthe conductor loss of CPW lines. By means of defining modifiedstopping distance term, they also verified increasing the conductorloss calculation accuracy of the symmetric CPW significantly atlower frequencies. This modified approach overcomes the lowfrequency limitations of the skin effect formulation.

In this article, it is aimed to develop a formulation for theCAD-oriented expressions related with the conductor loss mech-anism of the analyzed CPWs in the literature [5, 13, 15] for CPWmode. In the reports (both the gaps and ground planes are asym-metric), CPW with finite ground planes were assumed to beasymmetric. In addition, the dielectric layer of the proposed struc-ture of CPW studied in this article were considered to have a finitecross sectional dimensions. Another crucial point in this work isthat the modified stopping distance formulation has been directlyobtained without using the data table given in [14].

The following sections of this article can be summarized as fol-lows. In Section 2, conformal mapping applications used in [5, 15]have been extended for proposed generalized asymmetrical CPW(GA CPW) and Section 3 is devoted to the variation of attenuationconstants according to the geometrical parameters of GA CPW.Detailed comparisons between the results of this study and thoseavailable in the literature have been undertaken to prove the highaccuracy achievement of the analytical expressions results relatedwith the loss and electrical parameters involved in this work.

2. ANALYSIS

The cross-sectional view of GA CPW for two cases is given inFigure 1. In Figure 1(a), the central strip width of the GA CPW isW (�2a) and the widths of ground planes are D1(� c1 � b1) andD2(� c2 � b2) for the left and the right sides, respectively. Asindicated in Figure 1(a), the gaps between central strip and groundplanes are taken as S1(� b1 � a) and S2 (� b2 � a) for the left andthe right sides, respectively. h refers to the dielectric thickness andt stands for the conductor thickness. Relative dielectric constant ofthe substrate is taken to be �r. The geometrical properties of GACPW shown in Figure 1(b) differs from that of Figure 1(a) onlythat the finite dielectric width in Figure 1(b) case is equal to 2d asshown in the diagram.

The capacitance, the effective dielectric constant and the char-acteristic impedance of GA CPW shown in Figure 1(a) can beexpressed as follows after application of successive conformal

2082 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 9, September 2007 DOI 10.1002/mop