A High Power Factor Rectifier Associated With a ZCZVS PWM Full-Bridge Inverter in

a Rectifier/Inverter System

Carlos A. Gallo(1), Marcella C. Chagas(1), Fernando L. Tofoli(2), Roberto M. Finzi Neto(3)

Universidade Federal de Uberlândia1

Faculdade de Engenharia Mecânica

Campus Santa Mônica –Av. João Naves de Ávila, 2121

CEP 38400-902 Uberlândia-MG – Brasil

Fone/Fax: 55 34 3239 4035

gallo@mecanica.ufu.br Universidade Federal de São João Del-Rei2

Departamento de Engenharia Elétrica Campus Santo Antônio – Praça Frei

Orlando, 170 – Centro São João del-Rei-MG – Brasil

CEP 36307-352 Fone: 55 32 33792368

fernandolessa@yahoo.com.br

Universidade Federal de Goiás3

UFG, Campus Catalão. Av. Dr. Lamartine P. Avelar N:1120, Setor

Universitário. Catalão – GO.

finzi@eee.ufg.br

Abstract – This work presents the operation and design of a high power factor AC-DC-AC power supply rated at 2 kW operating at high switching frequency. Good power factor is obtained using an AC-DC interleaved boost-flyback converter as a pre-regulator circuit associated with a nondissipative snubber without commutation losses. Commutation losses are practically reduced to zero and EMI emission can also be minimized. The operation of the ZCZVS PWM full-bridge inverter operating at 30kHz in the high frequency leg and at 60Hz in low frequency leg is also investigated.

1 Keywords – AC SMPS, ZCZVS, PWM Full-Bridge

Inverter.

INTRODUCTION

The power supply unit is a very important circuit for all electronic equipment, because it provides the necessary voltages for correct work of the electronic circuits, these electronic circuits are used, usually, to feed professional or domestic equipments, for instance, computer, telecommunication equipments, aviation and military, games, etc. These equipments are more sophisticated, as it can be seen in [1] and [2], and reduced size and weight are mandatory requirements, as well as increased efficiency. Generally, these equipments use AC as a primary power energy supply. Therefore, the AC energy must be converted to DC energy, because the majority of these systems require high quality DC power.

Linear power supplies are good for low power applications, but are uneconomical and inefficient when more power is required. The alternative is to use

switched mode power supplies (SMPS). This kind of power supply presents multi output DC voltages, constant switching frequency, and reduced size and weight when compared with linear power supplies, but the input stages of the switching mode power supplies are well known to be harmonic generators. Recently, there has been great interest about the reduction of input current harmonics and power factor correction (PFC). Moreover, in many single-phase applications, mainly in power supplies, the power level can reach several kilowatts, and in some situations, the input voltage can be quite high too. For these types of application the conventional Boost PFC converter has been more used due to its characteristics of dc-voltage gain, lower inductor volume and weight, and losses on the power devices, which will affect converter cost, efficiency, and power density [3]-[5]. It is perfect for pre-regulator applications, but this converter presents appreciable commutation and conduction losses, bringing reduction in the efficiency.

Conventional Resonant and quasi-resonant converters [6]-[10] provide ZCS (Zero-Current Switching) and/or ZVS (Zero-Voltage Switching) [11] [12], and these converters can operate with high-frequency, but these techniques have load limitation, because there are current and/or voltage peaks over the switches and range of frequency control, making difficult the filter components design.

The interleaved power conversion technique is a strategy of interconnection of multiple switching cells for which the operation frequency is identical,

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978-1-4244-3384-1/10/$25.00 ©2010 IEEE

but for which the internal switching instants are sequentially phased over equal a switching period fractions. This technique can be used with typical PFC IC’s [13], but in this case the switching cells operation is 40 kHz. However with this frequency the volume of filter capacitors is supposed to be greater than that at 100 kHz, because the filter must be designed for 200 kHz. A good way to reach high frequency and high-power operation is to use the non-dissipative snubber presented in references [11], [12], and [18].

An AC-DC-AC switching mode power supply provides the conversion of a given DC voltage level to AC voltage. There are several inverters that can be used as in this case, but it is necessary those converters operate in high-frequency and presenting reduction of switching losses, regulated multiple output and isolation.

A DC-AC ZCZVS PWM full-bridge inverter with a nondissipative snubber that can reach high frequency and high-power operation is used in this SMPS with AC output. This converter is well suitable for this power level and it provides:

- Soft switching for full load range;

- Conduction losses are almost the same as those observed in the hard PWM converter.

The main switches turning-on at zero current can reduce significantly the undesirable effects of the parasitic inductances related to the circuit layout. The commutation losses are practically reduced to zero and the EMI emission can also be minimized. The operation of the converter is analyzed, and design guidelines for the auxiliary commutation cell are recommended based on this analysis. Experimental results are presented to demonstrate the feasibility of the proposed 2kW inverter.

THE PROPOSED SMPS

The AC-DC and DC-AC non-dissipative converters are shown in Figure 1 and Figure 2, respectively. The converters can operate with reduced commutation losses. High power factor is obtained using the average current mode control technique with IC UC3854.

To simplify the analysis the converters will be studied separated, but it does not compromise its understanding.

Figure 1 – AC-DC interleaved boost-flyback converter associated with the non-dissipative snubber.

Figure 2 - AC-DC Full Bridge inverter using a soft commutation

cell.

Figure 3 – High power factor power supply using an interleaved boost-flyback converter and a full-bridge inverter with soft-switching.

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PRINCIPLE OF OPERATION OF THE AC-DC INTERLEAVED BOOST-FLYBACK CONVERTER

Figure 3 shows the interleaved boost converter associated with a nondissipative snubber used as a pre-regulator.

From [11] and [12], one can obtain the waveforms shown in Figure 3.

Figure 4 - Theoretical waveforms for the interleaved boost-flyback

converter associated with a nondissipative snubber.

From [11] and [12], the transfer function between Vo

and Vin is given by:

( )01 0101

13 31

22 2

S

S

DTGKD T π π

αω ωω

= +⎡ ⎤

− − + +⎢ ⎥⎣ ⎦

(1)

Where: 1:n = turns ratio.

aux

O

C

VKV

= (2)

011 1

1

r rC Lω = (3)

s

TDTΔ= (4)

j iT t tΔ = − (5)

Sf = switching frequency;

ST = switching period.

PRINCIPLE OF OPERATION OF THE DC-AC PWM FULL-BRIDGE INVERTER

The purpose of zero-current-transition (ZCT) is to force the current flowing through a device to decrease to zero before this device is turned on or off. This is accomplished by the addition of an auxiliary circuit that provides a resonant current to take away the current flowing through the main device before the switching transition. With the ZCT techniques, inverters are

expected to achieve a higher switching frequency with reduced switching losses, attenuated acoustic noise, and reduced electromagnetic interference (EMI).

Figure 2 shows a typical three-phase ZCT inverter circuit. Historically, the research with this circuit configuration goes back to the McMurray inverters for the SCR forced current commutation [19]-[20]. For the soft transition research with modern gate-turn-off devices, such as IGBTs, several ZCT control schemes were developed to achieve the zero-current turn-off [21] [22]. Recently, still with the same circuit, a new ZCT scheme was proposed, which improves the performance by enabling both the main switches and auxiliary ones to be turned on and off under zero-current conditions, and achieving a near-zero-voltage turn-on for the main switches [23]. As a result, besides the elimination of switching turn-off loss, the diode reverse recovery current and the switching turn-on loss are also substantially reduced, the distribution of current and thermal stresses in the auxiliary switches is evened out, and the resonant capacitor voltage stress is reduced by 30%. In order to provide guidelines for the scheme proposed in [23], which realizes the zero-current and near-zero-voltage switching in high power applications, this paper presents the design considerations through a study example of a 2 kW three-phase prototype inverter. For consistency, the inverter designed is still called a ZCT inverter.

DESIGN EXAMPLE

This section presents a design procedure and an example to determine the values regarding the resonant tank elements of the proposed ZCZVT commutation for the proposed inverter. The given specifications are presented in Table I.

Table I – Design specifications.

Parameter Value Output Power Po=2000W Input Voltage E=400V

Output Voltage Vo=220Vrms Output frequency fo=60Hz

The design procedure consists of four steps, as

follows [23].

a) Calculation of the output current peak, given by (6).

00

0

2 (1 )PI IV

= + Δ (6)

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(b) Calculation of the characteristic impedance. To assure the main switches turning-off under ZCS and ZVS conditions, during stages 9 and 10, the peak current taken away from the main switch to the auxiliary circuit must be larger than the output current peak. From these stages, the current peak through capacitor Cr2 is given by:

02pk

EIZ

= (7)

From (6) and (7), a new parameter k can be defined as follows:

0

pkIk

I=

(8) where k≥1. If k=1 is adopted, which is a practical design value to

compensate the parasitic losses, the characteristic impedance Z0 can be given by:

(9) c) Calculation of the resonant frequency. To minimize

the reverse recovery of the main diodes, the resonant frequency in stage 2 can be chosen to control the di/dt rate during turn-off, which is then given by (10).

0

1 12 sin2

Ididt

k

ω−

≅⎛ ⎞⎜ ⎟⎝ ⎠

(10)

The resonant frequency can be obtained from (10) as follows:

1

0

12 sin2

didt k

Iω

− ⎛ ⎞⎜ ⎟⎝ ⎠= (11)

d) Calculation of the resonant components. If the characteristic impedance and the resonant frequency are given, expressions (12) and (13) can be used to calculate the resonant inductor and capacitor.

01 2 ZL Lω

= = (12)

0

11 2C CZ ω

= = (13)

If a di/dt rate equal to 80A/µs [4] is adopted, the resonant components values L1, C1, L2 and C2 can be calculated for both legs of the converter, where the first one operates at 30kHz and the second one operates at 60Hz, as shown in Table II.

Table II – Design example. Parameter Value

I0 15.4A Z0 15.3Ohms L1 5.2μH C1 17.4nF L2 2.6μH C2 7.2nF

EXPERIMENTAL RESULTS

A prototype of the proposed switching mode power supply was built using the following parameter set:

Switches = IRFP460; CR = 4.2nF; Diodes = MUR1560; CR1 = 15.6nF; Aux. diodes=APT30D60B; CR2 = 7.8nF;

Full-bridge diodes=HFA15TB60; fS = 30kHz; Vi =110/220 V; Cf = 1mF; LR = 4μH; CR2 = 27nF; Lb = 1.2mH; P0 = 2000 W; Lf = 550μH; Vo = 220V; Lt = 600μH; Io = 9.0A; C0 = 680μF;. Lf = 100μH;

Figures 6, 7, 8 and 9 show the simulation and experimental results.

As it can be seen the commutation of the switches occurs without losses and power factor is almost unity. Figure 5 shows the power factor correction at rated load, as displacement power factor is 0.998.

Figure 5 – Input voltage and input current at nominal load.

Figure 6 and Figure 7 show the commutation in the active switches, it can be seen that the main switches do not present stresses of current and/or voltage, as well as the commutation is nondissipative. The auxiliary switches commutate with zero current.

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(a)

(b)

Figure 6 – (a) Switch M1 waveforms and (b) voltage and current waveforms regarding the resonant tank in the interleaved boost-flyback

converter.

Figure 7 – Switch S1 waveforms. Scales: Vs (100V/div.); Is (5A/div.);

tempo (1μs/div.)

Figure 8 – Output voltage and output current. Scales: Vo (200V/div.); Io (10A/div.);

tempo (5ms/div.)

Figure 9 shows efficiency as a function of the output power.

88

90

92

94

96

98

100

600 700 800 900 1000 1100 1200 1300 1400 1500 1600 1700 1800 1900 2000

Power (W)

Effic

ienc

y (%

)

A

Figure 9 – Efficiency as a function of the output power.

CONCLUSIONS

This paper has reported analytical, simulation and experimental developments of a SMPS using the PFC AC-DC Boost associated with a non-dissipative snubber. It has been demonstrated that the current waveshaping control technique in combination with the non-dissipative snubber permits highly efficient power factor correction pre-regulator without commutation losses. The proposed approach allows a good performance in high frequency of operation.

The second part of this work has presented an active auxiliary commutation topology for a three-level PWM ZCZVS full-bridge inverter based on the resonance principle. The auxiliary circuit is bi-directional, operating at ZVS and ZCS conditions.

It is important to mention that the proposed inverter consists in a three-level topology without the need of auxiliary supplies, differently of some counterparts. The proposed circuit assures the soft switching of the main devices to any type of load i.e. inductive or capacitive loads, with low current stress and high efficiency.

The inverter operation and performance are evaluated by Experimental tests, which validate the soft switching commutation in ZVS and ZCS ways for all the switches.

The objective initially proposed was reached, a switched mode power supply with power factor correction (0.998), high efficiency (95%) and low harmonic distortions (2.84% for current and 2.83% for voltage) and good regulation was analyzed theoretically, designed, and implemented.

ACKNOWLEDGEMENT

The authors gratefully acknowledge CNPq and FAPEMIG for the financial support to this work (Proc. Nb. 574001/2008-5 - INCT-EIE) and also Texas Instruments and ON Semiconductor for sending us free samples.

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