Fully Electronically Programmable Complex Filter for Multistandard Applications

Hussain Alzaher, and Noman Tasadduq Electrical Engineering Department

King Fahd University of Petroleum and Minerals Dhahran, Saudi Arabia alzaherh@kfupm.edu.sa

Abstract— This work presents an improved design of the current amplifier based optimal complex filter providing fully electronically programmable characteristics. It extends the tuning features to include the pole frequency and the gain in addition to the center frequency. The new solution avoids the employment of capacitor and/or resistor banks leading to a more compact design solution while maintaining comparable power consumption. Simulation results obtained from a 0.18 m standard CMOS process show that the proposed design can be used to accommodate several short range communication applications such as Bluetooth and ZigBee, as well as several mobile TV standards such as Integrated Services Digital Broadcasting-Terrestrial (ISDB-T) and Terrestrial Digital Multimedia Broadcasting (T-DMB). Performance characteristics of proposed design in terms of dynamic range and image rejection ratios are also provided.

Keywords—CMOS analog integrated circuits; current-mode circuits; programmable complex filter.

I. INTRODUCTION Recently the design of complex (polyphase) filters is

renewed by using them in low intermediate frequency (low-IF) wireless applications such as Bluetooth (BT) and ZigBee receivers, mobile TV, and wireless sensor networks applications [1]-[14] because they provide the solution for image rejection. The main drawback of the low-IF receivers is that they are more sensitive to mirror signal suppression. In contrast to the zero-IF receiver, the mirror signal in low-IF processing can be larger than the wanted signal. Active complex (polyphase) filters have a linear frequency transformation and a high image rejection. Therefore, they are suitable for low-IF architectures. To support various applications, more and more multistandard devices are developed as a cost-effective and size effective solution.

Mobile digital television is an emerging wireless system and its potential financial income has encouraged leading technology manufacturers and universities to target this application. In recent years there has been considerable progress in the digitization of traditional TV and the system for fixed and portable reception of digital terrestrial television, known as Digital Video Broadcasting – Terrestrial/Handheld (DVB-T/H). In addition to analog TV [11], two popular digital TV standards that are suitable for low-IF multistandard

operation are Integrated Services Digital Broadcasting-Terrestrial (ISDB-T), Terrestrial Digital Multimedia Broadcasting (T-DMB) and analog TV [12]-[13]. ISDB-T (supporting VHF and UHF Band) uses a new transmission technology “segmented OFDM transmission system” which allows fixed, mobile and portable applications in the same channel. There are 13 segments in the 6-MHz with each segment occupying 428kHz while 1 and 3 segment modes are the most commonly used. Hence, the bandwidth of the one-segment is 0.43MHz while that for three-segment is 1.29MHz. Whereas, T-DMB standard (defined in the UHF and VHF bands) has channel bandwidth of 1.5 MHz. Since both ISDB-T 1-segment and T-DMB are narrow band system therefore a low IF mode is more power efficient and more robust in 1/f-noise and DC offset. The work in [12]-[13] presented a mobile TV tuner that implements DVB-H/T, ISDB-T and T-DMB standards and covers VHF, UHF and L bands. The filter is configured for dual mode operation, lowpass for DVB-H/T direct conversion standard and low-IF for ISDB-T and T-DMB standard, to cover different standards. The complex bandpass filter center frequency is approximately 400kHz and can be varied by 12/16, 15/16 and 18/16 times channel bandwidth. Programmability is achieved using 5 bit capacitor arrays.

In addition to the mobile TV, Bluetooth (BT)/ZigBee receiver is another application that takes advantage of multistandard operation [8]-[10]. The typical bandwidth of Bluetooth signal is 1MHz with 1MHz, 2MHz or 3MHz center frequency, whereas ZigBee works at twice the bandwidth of Bluetooth and typically centered at 2MHz. Bluetooth applications are well known and some of the examples include wireless headsets, file sharing, and printing. Whereas, ZigBee is used for simple wireless connectivity and as an example could be used for control of lights, electronic devices using the mobile handset.

It can be seen from the above discussion of multistandard applications that the low-IF complex filter requires tunable bandwidth as well as center frequency. Typical receiver topologies are shown in Fig. 1. The first is atypical dual mode BT/ZigBee receiver whereas the second is suitable for ISDB-T/T-DMB reception. Clearly, the channel select filter must be programmable to accommodate different standards.

The author would like to acknowledge the support of KFUPM and King Abdulaziz City for Science and Technology (KACST) for the financial support Project No: AT-30-212.

Proceedings of 2013 IFIP/IEEE 21st International Conference on Very Large Scale Integration (VLSI-SoC)

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Fig. 1. Ttypical architecture for a multistandard rreceivers (a) Short range communications (b) Mobile TV

Most of the complex filters are based on transconductance amplifier-C (gm-C) or active-RC techniques. More recently, however, an optimal complex filter realization is identified among several voltage-mode and current-mode complex filters based on voltage, transconductance, transresistance, and current amplifiers [14]. Systematic comparison of the developed filters show that the most power efficient complex filter is the one obtained from current-mode structure based on the current amplifier (CA). The experimental results demonstrated that the optimal designs can operate with lower power consumption than active-RC filters and provide better linearity than gm-C techniques. There are three main drawbacks of that filter (1) does not promote independent programmability of center frequency and pole frequency (2) utilizes capacitor matrices for tuning the center frequency but it also changes the bandwidth (3) does not provide gain. Independent tuning of the center frequency is as important as the bandwidth as both are function of RC products. It would also make the design flexible to accommodate multiple low-IF applications. It is well known that capacitor matrices occupy relatively large silicon area and the switching transistors (quasi-static switches) within the banks are associated with finite non-linear resistances, which degrades the linearity [4]. Providing gain is not only advantageous because it relaxes the gain requirements of system’s amplifiers but also allows optimization of the linearity-noise trade-off of the filter design itself. This paper proposes the adoption of electronically tunable CA [15] in the design of CA based complex filter to provide area efficient solution to the tuning problem while maintaining low power property.

II. PROPOSED APPROACH An arbitrary lowpass filter can be converted to a bandpass

complex filter centered at c when every frequency dependent element in the LPF is modified to be a function of s-j c instead of s [3]. In the case of transforming a normal integrator to its polyphase counterpart, a complex feedback loop is realized by cross coupling between the in-phase (I) and

quadrature (Q) paths. The complex filters obtained from their biquad counterparts inherently exhibit better IRR than two cascaded stages of first-order. Two complex integrators can be used in cascade to develop two integrator loop complex filters as shown in Fig. 2. The corresponding transfer functions (TFs) are given by,

2

2 2( )( ) ( )( / )

oQoI oC

I Q c c o o

XX KH sX X s j s j Q

ωω ω ω ω

= = =− + − +

(1)

The complex bandpass filter given by (1) exhibits a center frequency of c, pole frequency of o, pole quality factor of Q and bandwidth of 2 o/Q (i.e. twice the bandwidth of the original lowpass filter).

Fig. 2. Two-integrator-loop complex filter block diagram

The corresponding optimal CA based complex filter [14] exhibits the following transfer function:

2 2

2 2 2

/( )( )

( ) ( ) /( ) 1/( )g

pc c q

K C RH s

s j s j K CR C Rω ω=

− + − + (2)

Thus, the filter exhibits c=Kc/CR, o=1/CR, Q0=1/Kq and

K=Kg. But KC, Kq and Kg are fixed gain factors obtained from presetted current-mirrors ratios. In this case, o can be tuned only by using capacitor/resistor arrays. Employing RC banks is one of the simplest approaches for tuning analog filters. However, they are associated with relatively large area. Also, they are either associated with limited tuning range and/or low tuning resolution. The simplest method to provide electronic tuning is to replace the passive resistors by triode MOS transistors but they suffer from poor linearity, and hence limited dynamic range [16]. In addition, their tuning ranges are limited particularly for low power supply voltages. Another option is to use current controlled current amplifier, which utilizes the programmable resistance in the X terminal but it is dependent on temperature and also suffers from the nonlinearity. The proposed approach is to utilize multioutput independently programmable gains of CA.

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III. PROPOSED SOLUTION Electronic tuning can be added to the filter by having CA

with programmable gains. This is can be achieved by replacing the fixed gain CA with a multi-output programmable gain current amplifier (CA). A power efficient realization for such design is proposed in [17], shown in Fig. 3. Transistor M1 and M2 set the voltage at X terminal to zero and with the help of negative feedback, formed basically by M3, an input resistance in the range of few tens of ohms is achieved [17]. A current folding output stage is used to replace the normal current mirroring output stages and hence achieve robust linearity [17]. The input current ix, which is forced by the constant currents of M2 and M6 to flow into M3, is conveyed to the output port Z by source-coupling M3 with M4 instead of using current mirroring. Since this coupled pair is biased with a constant tail current, the drain current changes in M3-M4 will be equal but with opposite sign, regardless of their matching resulting in negative type CA with unity gain (iz=ix).

Fig. 3. The proposed CA with the first output having unity gain and the others with electronically controlled gains

A second differential pair M12-M13 is connected in parallel with M3-M4 to provide two additional current outputs. When the two pairs are biased with different tail currents, it can be shown that the large signal current relationship will be given by:

222

1 1 21 1

1 ( / 4 )

1 ( / 4 )d TT

zP zN xT d T

KV III I II KV I

−= =

− (3)

where Vd=Vc-Vg3 (i.e. the differential voltage of the two source coupled pairs), K=0.5 CoxW/L with as the surface carrier mobility, Cox is the gate oxide capacitance per unit area, and W and L are the width and length of the channel. Thus, for small signals Vd << 2[min.(IT1,IT2)/K]1/2, the relationship simplifies to:

1/ 21 1 2 1( / )zP zN T T xi i I I i= = (4)

Thus, the new outputs would exhibit electronically controlled gains which can be programmed by adjusting the tail currents. Extra output currents with different gains can be obtained by adding more output stages with each stage providing both positive and negative signals.

The filter suggested in [14] is modified to promote electronically programmable characteristics of all parameters

as shown in Fig. 4. The fixed gain Kc has to be replaced by programmable gain αc. In order to make pole frequency ωo electronically programmable, current gain αo are added in appropriate signal paths. Also, the fixed gains Kq in the filter of [14] are replaced by programmable gains αq. In addition, electronically programmable gains αg provide independent control of the gain. Employing these modifications it can be shown that the filter now exhibits c=αc/CR, o=αo/CR, Q0=αo /αq and K=αg/αo. Therefore all parameters are electronically programmable.

Fig. 4. Fully programmable CA based complex filter

IV. SIMULATION RESULTS To demonstrate the versatility of the proposed filter, two

applications examples are considered. A 4th order filter is designed for BT/ZigBee dual-mode applications whereas an 8th-order filter is designed for ISDB-T and T-DMB standards having different center frequencies. The 4th-order and the 8th-order filters are developed by cascading two sections and four sections of filter of Fig. 4, respectively. Simulation is carried out using the CMOS realization of the CA. The passive components of R=40k and C=8pF are selected. The design starts with BT requirements of Butterworth response having bandwidth of 1MHz, center frequency of 1MHz and unity gain. The bandwidth and center frequency requirements are fulfilled through setting the amplifier gains such as αo=1 and αc=2, respectively. Butterworth response is obtained with αq1=1.85 (first stage) and αq2=0.77 (second stage) while unity differential gain is achieved with αg=1. On the other hand, ZigBee standard operates with a bandwidth of 2MHz and center frequency of 2MHz while Butterworth response has to be maintained. This means that both the bandwidth and center frequency must be adjusted. These requirements are achieved as follows. First the bandwidth is adjusted through changing αo to 2 while the center frequency is fixed via setting αc to 4. The Butterworth response is re-maintained by adjusting αq1 and αq2 to 3.70 and 1.54, respectively, whereas the gain is adjusted by setting αg=2.

Simulation results obtained from these designs are shown in Fig. 5. The supply voltages of the CA were ±0.75V and the total nominal current of each amplifier was approximately

15

140μA. In order to further demonstrate the flexible programmable characteristics of the proposed design, simulation results showing gain tuning for BT and center frequency tuning for ZigBee are shown in Fig. 6 and Fig. 7, respectively. The required gain settings of the various obtained responses are given in Table I. Fig. 6 shows BT gain tuning of 0dB, 6dB, and 12dB and 18dB whereas Fig. 7 shows ZigBee responses with center frequency at 2MHz, 3MHz and 4MHz.

0 0.5 1 1.5 2 2.5 3 3.5 4

x 106

-30

-25

-20

-15

-10

-5

0

5

Frequency (Hz)

Gai

n (d

B)

ZigBeeBT

Fig. 5. Simulation results showing BT and ZigBee nominal response

0 0.5 1 1.5 2

x 106

-25

-20

-15

-10

-5

0

5

10

15

20

Frequency (Hz)

Gai

n (d

B)

Gain=0dBGain=6dBGain=12dBGain=18dB

BTGain

Tuning

Fig. 6. Simulation results showing gain tuning for BT

0 1 2 3 4 5 6 7

x 106

-35

-30

-25

-20

-15

-10

-5

0

5

Frequency (Hz)

Gai

n (d

B)

ZigBee fctuning

Fig. 7. Simulation results showing center frequency tuning for ZigBee Table I. Example of gain settings for BT and ZigBee

Application Distinct Feature αo αc αq1 αq2 αg

BT

K=0dB 1 2 1.85 0.77 1 K=6dB 1 2 1.85 0.77 1.41 K=12dB 1 2 1.85 0.77 2 K=18dB 1 2 1.85 0.77 2.82

ZigBee fc=2MHz 2 4 3.70 1.54 2 fc=3MHz 2 6 3.70 1.54 2 fc=4MHz 2 8 3.70 1.54 2

The 8th-order filter prototype designed for ISDB-T and T-DMB is realized with same values of passive components and hence allowing its implementation utilizing two 4th-order sections in cascade. The ISDB-T and T-DMB standards, however, operate with one-segment bandwidth of 0.43MHz and channel bandwidth of 1.5 MHz, respectively. Also both standards require filter center frequency 12/16, 15/16 and 18/16 times channel bandwidth. The required gain settings to achieve these desired responses are given in Table II. Table II. Example of gain settings for ISDB-T and T-DMB

Appli- cation fc αo αc αq1 αq2 αq3 αq4 αg

ISDB-T

BW*12/16 0.43 0.65 0.84 0.72 0.48 0.17 0.43

BW*15/16 0.43 0.81 0.84 0.72 0.48 0.17 0.43

BW*18/16 0.43 0.98 0.84 0.72 0.48 0.17 0.43

T-DMB

BW*12/16 1.50 2.25 2.94 2.50 1.67 0.60 1.50

BW*15/16 1.50 2.81 2.94 2.50 1.67 0.60 1.50

BW*18/16 1.50 4.22 2.94 2.50 1.67 0.60 1.50

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Simulation results showing required center frequency tuning without disturbing the bandwidth for ISDB-T and T-DMB are shown in Fig. 8 and Fig. 9, respectively. In addition, Fig. 10 shows gain tuning of 0dB, 6dB, and 12dB and 18dB for T-DMB standard for the case of center frequency= BW*18/16. Theses gain settings are achieved through setting αg to 1.50, 1.80, 2.16, and 2.60, respectively.

0 2 4 6 8 10

x 105

-60

-50

-40

-30

-20

-10

0

Frequency (Hz)

Gai

n (d

B)

fc=BW*12/16fc=BW*15/16fc=BW*18/16

ISDB-T

Fig. 8. Simulation results showing center frequency tuning for ISDB-T

0 0.5 1 1.5 2 2.5 3

x 106

-50

-45

-40

-35

-30

-25

-20

-15

-10

-5

0

5

Frequency (Hz)

Gai

n (d

B)

fc=BW*12/16fc=BW*15/16fc=BW*18/16

T-DMB

Fig. 9. Simulation results showing center frequency tuning for T-DMB

0 0.5 1 1.5 2 2.5 3

x 106

-60

-50

-40

-30

-20

-10

0

10

20

Frequency (Hz)

Gai

n (d

B)

Gain=0dB

Gain=6dBGain=12dB

Gain=18dB

T-DMBGain

Tuning

Fig. 10. Simulation results showing gain tuning for T-DMB

Summary of the performance results are shown in Table III. The input third-order intercept point (IIP3) for in-band are found using two testing tones at 1.1MHz and 1.2MHz for BT, 2.2MHz and 2.4MHz for ZigBee, 0.4MHz and 0.5MHz for ISDB-T, and 1.2MHz and 1.3MHz for T-DMB. The total in-band output noise of the filter is found by integrating over the given bandwidths. The in-band spurious free dynamic ranges (defined as SFDR=2(IIP3-input noise)/3) are accordingly calculated. The signal and image frequency responses at the nominal center frequency are measured and the corresponding image rejection ratios (IRR) are determined. Table III. Performance results of proposed complex filter

Application BT ZigBee ISDB-T T-DMB Order 4 8

fc (MHz) 1 2 0.322 1.125 Power (mW) 1.68 3.36

Linearity (dBA) -27.5 -22.5 -15 -10 Noise (dBA) -137.2 -133 -141 -134 SFDR (dB) 73.1 73.6 84.0 82.7 IRR (dB) -34 -46 -43 -51

In comparison with the performance of BT complex filters

presented in [8] and [14] it can be seen that the power consumption of the proposed multistandard complex filter is more, whereas it achieves a significant improvement of 3.1dB and 7.3dB for SFDR as compared to [8] and [14], respectively. This is attributed to the better linearity of the CA utilized in this work compared to its counterparts in [8] and [14]. Note that although the SFDR reported in [8] is for far blocker, which is normally better than the in-band, the proposed filter still shows better performance. Also, it achieves much better power consumption and SFDR as compared to the BT complex filter presented in [9] (3.2mW and 45.4dB). The IRR of the BT mode can be improved if center frequency is increased to 3MHz as in [14] which can be achieved without the need for more power consumption.

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Comparison with the Zigbee complex filters presented in [7], [8] and [9] shows that again the proposed filter consumes much less power as compared to [7] and [9] (3.6mW and 3.9mW) while it has higher power consumption as compared to [8] (1mW). Whereas the improvement in SFDR as compared to [7], [8] and [9] is 18.1dB, 5.2dB and 29.5B, respectively. Also, the proposed filter shows much better power consumption as compared to the complex filer used for analog TV in [11] (14mW).

V. CONCLUSION This paper presents the design of a versatile complex filter

with fully programmable characteristics. The filter can be electronically configured to realize responses with different center frequency, bandwidth and gain. Therefore, it supports multistandard applications such as BT/ZigBee and ISDB-T/T-DMB. It also avoids the use of capacitor and/or resistor banks leading to a more area efficient design solution while maintaining comparable power consumption. Simulation results showing the flexibility of obtaining the required responses for these applications are given. A 4th-order is designed for BT and ZigBee whereas an 8th-order filter is employed for ISDB-T and T-DMB. The proposed design offers SFDR of 73.1dB for Bluetooth and 73.6dB for ZigBee while it consumes 1.68mW. Also, it achieves SFDR of 84.0dB for ISDB-T and 82.7dB for T-DMB with twice the power consumption due to the increase of the filter order. There is inter-dependent relation between noise, SFDR and power and the performance of the proposed filter can be further optimized by considering different values of passive components. The decrease in resistor values will result in improved noise performance but at the expense of lower SFDR because of increase in signal currents within the stages which decrease the signal swing. The SFDR can be improved by increasing the supply voltages and consequently, a desired compromise between power consumption and SFDR can be achieved.

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