A Novel High Step-up Ratio Bi-directional DC-DC Converter

Yu Du Alex Q. Huang Mengqi Wang Srdjan M. LukicFREEDM Systems Center, FREEDM Systems Center, FREEDM Systems Center, FREEDM Systems Center,

North Carolina State University, North Carolina State University, North Carolina State University, North Carolina State University, Raleigh, NC 27606, U.S.A. Raleigh, NC 27606, U.S.A. Raleigh, NC 27606, U.S.A. Raleigh, NC 27606, U.S.A.

ydu4@ncsu.edu aqhuang@ncsu.edu mwang7@ncsu.edu smlukic@ncsu.edu

Abstract—High step-up ratio bi-directional DC-DC converter is attractive in energy storage systems for renewable energy generation and in electric vehicle applications. A novel high step-up ratio bi-directional DC-DC converter and its alternative topology which can achieve soft switching in full load range and wide voltage range are proposed. The operation principle of the converter is analyzed in this paper. The modulation strategy and the optimum operation region are presented. A 400kHz 14.4V to 360V prototype was built to verify the analysis and the operation of the proposed converter. Zero voltage switching of all the switches was verified by experimental waveforms. The tested converter efficiency at 450W is 94.1%.

I. INTRODUCTION Electric power and transportation industries are two major

sources for primary energy consumption on Earth. According to the United States Energy Information Administration, in 2009 about 40% of U.S. primary energy was consumed in electric power sector and about 30% in transportation sector. The majority of the energy sources such as petroleum, natural gas, coal are nonrenewable or not environment-friendly. The associated energy shortage, green house gas emission and energy security issues are well known. It is promising that greater utilization of renewable energy resources can help mitigate the issues with fossil fuels. However, due to the intermittent nature of the renewable energy resources such as wind and solar energy, there is difficulty to achieve high penetration of renewable energy generation in today's grid. According to [1], one major challenge to the wide spread adoption of renewable energy is the ability to store and control the wide variety of different energy resources. The future electric distribution grid must address the issues of storage and complex control. To achieve the goal of greater utilization of renewable energy sources, both large-scale centralized installations and wide-scale distributed renewable energy generation are helpful. The energy storage systems can be used to complement renewable energy resources to optimize the energy use. Transportation electrification is another

important approach to reduce the green house gas emission and mitigate the heavy dependency on petroleum.

Batteries and ultracapacitors are key energy storage devices used for grid and transportation applications [2]. In a microgrid, the combined distributed renewable energy resources (DRERs) and distributed energy storage devices (DESDs), provide ancillary services such as peak shaving, emergency power supply, frequency and voltage control based on the regulated real and reactive power from the hybrid sources. A hybrid generator with a photovoltaic energy conversion system is proposed with supercapacitors and lead-acid batteries coupled to a DC structure through bi-directional DC-DC converters in [3]. The supercapacitors avoid frequent

This work was supported by ERC Program of the National Science Foundation under Award Number EEC-08212121.

Fig. 1: The conventional dual-active-bridge (DAB) bi-directional DC-DC

converter.

Fig. 2: The proposed basic high step-up ratio bi-directional DC-DC

converter

978-1-4577-1216-6/12/$26.00 ©2012 IEEE 524

charging and discharging of batteries and then extend the battery life. Suppercapacitors also lead to fast dynamic regulation of power whereas batteries are used for long-term supply of energy. In order to improve the rate of self-consumption for a PV generator, a 5kW photovoltaic system with Li-ion battery energy storage system was proposed in [4], which allows a temporary decoupling of the generation and injection. A bi-directional DC-DC converter connected a 5-7kWh Li-ion battery pack with the voltage range from 168V to 336V to the 600V- 680V DC link of the system.

With the increased use of energy storage devices in a variety of applications, there is a growing need of bi-directional power conversion. The applications of particular interest are electric vehicles [5], electric vehicle charging infrastructure [6]-[8], renewable power generation systems [9], hybrid power sources [10]. Particularly challenging are the applications where the energy storage system voltage varies substantially during normal operation as it does for batteries, supercapacitors or fuel cells. In most applications, high power conversion efficiency is required for all operating conditions.

The dual active bridges (DAB) DC-DC converter, shown in Fig. 1, is a widely used isolated bi-directional DC-DC converter [11]-[14]. Conventional modulation strategy uses the phase shift between the two full bridges to control the direction and amplitude of the power flow. However, the optimal operating range is limited, making it unsuitable for wide voltage range applications. Advanced modulation strategies improve the DAB performance [15]-[19].

In this paper, a novel high step-up ratio bi-directional DC-DC converter and its alternatvie topology are proposed for distributed energy storage applications to integrate small capacity battery packs, as shown in Fig. 2. Vb represents the low voltage battery pack and Vd represents the system DC grid voltage (high voltage side). The high step-up ratio is obtained by two transformers and a voltage doubler. For this converter, based on the proposed modulation strategies zero voltage switching (ZVS) can be obtained in the wide voltage range and full load range. The operation principle of the converter is analyzed in this paper. The modulation strategy and the optimum operation region are addressed. A 400kHz 14.4V to 360V/450W prototype was built to verify the analysis and the operation of the proposed converter.

II. THE FORWARD MODE OPERATION PRINCIPLES The forward mode operation is defined here as the power

transferred from the low voltage side Vb to the high voltage DC-link side Vd, or the discharging operation when a battery pack is connected to the low voltage side.

The proposed converter operation waveforms for the forward mode operation are illustrated in Fig. 3. V1 is the amplitude of the square-wave voltage between a and c in the high voltage side, as shown in Fig. 2. In steady state, V1 is equal to the half of the DC-link voltage Vd, or the capacitor voltage Vc2. V2 is the amplitude of the three-level voltage between b and c in the high voltage side. It is the sum of the voltage across the high voltage winding of two transformers. Vdo, Veo are the square-wave voltage across the low voltage winding of two transformers. The amplitude of Vdo and Veo is

Fig. 3: The forward mode operation waveforms of the proposed converter

Fig. 4 (a): The t0-t1 interval of forward mode operation

Fig. 4 (b): The t1-t2 interval of forward mode operation

Fig. 4 (c): The t2-t3 interval of forward mode operation

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equal to the half of the battery pack voltage Vb/2, since the filter consists of two equal capacitors C3 and C4 in series. V2 should be designed to be higher than V1 for the proposed converter to operate correctly.

The voltage reference polarity and current reference direction are labeled in Fig. 2. In forward mode operation, Vbc should lead Vac by a phase-shift angle φ. The phase-shift angle φ is defined in Fig. 3, where the center points of the positive part of the waveforms of Vbc and Vac are used as the reference time instant to calculate the phase-shift angle φ. For the low voltage side, Vdo and Veo are modulated in the phase-shift manner as well, where Vdo leads Veo by an angle of (1-m)·π such that a three-level voltage waveform can be generated for Vbc, and we call m modulation index.

From Fig. 4(a), at the time instant t0, the current in the equivalent series inductor L1 is negative and S1 is turned on with zero voltage switching (ZVS). In the time interval t0 – t1, Vac is equal to Vd/2, while Vdo is Vb/2 and Veo is –Vb/2. Vbc is zero, since it is the sum of these two voltages multiplied by the transformer turns ratio N. Therefore, the energy stored in L1 is discharged to C1 and the inductor current decreases. The energy stored in the filter capacitor C3 is discharged into C4.

In the time interval t1 – t2, Vac is still equal to Vd/2 and Vbc is zero. The current in L1 becomes positive and continues increasing, as shown in Fig. 4(b). The inductor L1 is charged such that S5 will be turned on with ZVS. The energy stored in the capacitor C4 is discharged back to C3.

In Fig. 4(c), at the time instant t2, S5 is turned on with ZVS. Then Veo becomes positive (Vb/2). Therefore, Vbc is equal to Vb·N (=V2) and is higher than Vd/2 (=V1). The energy stored in the inductor L1 is discharged to the output capacitor C3.

At the time instant t3, the current in the inductor L1 becomes negative. In the time interval t3 – t4 (Fig. 4(d)), the inductor L1 is charged and the energy is transferred from the output capacitor C3 to the DC-link side C1 such that S4 will be turned on with ZVS. The time interval t3 – t4 is a longer interval such that it is the major period in forward mode operation for the energy being transferred from the low voltage battery side to the high voltage DC-link side.

At the time instant t4, S4 is turned on with ZVS. Then Vdo is equal to –Vb/2 and Vbc becomes zero again. The energy stored in the inductor L1 is discharged into C1 and the current decreases. In the meanwhile the energy is also transferred from the battery to the DC-link side. The energy stored in the capacitor C3 is discharged into the capacitor C4, as shown in Fig. 4(e).

In the time interval t5 – t6, the inductor L1 is charged again by the DC-link capacitor C1 and the current in L1 becomes positive (Fig. 4(f)). The energy stored in L1 will be used to turn on the high voltage side switch S2 with ZVS. In this interval, the energy in C4 is discharged into back into C3.

In Fig. 4(g), at the time instant t6, S2 is turned on with ZVS and Vac becomes negative (-Vd/2). The other half cycle of one switching period begins. After S2 is on, the energy stored in the L1 is discharged into C2 and the current in the inductor

L1 decreases. Since Vbc is still zero, the energy in C4 continues discharging into C3 in the time interval t6 – t7.

Fig. 4 (d): The t3-t4 interval of forward mode operation

Fig. 4 (e): The t4-t5 interval of forward mode operation

Fig. 4 (f): The t5-t6 interval of forward mode operation

Fig. 4 (g): The t6-t7 interval of forward mode operation

Fig. 4 (h): The t7-t8 interval of forward mode operation

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In the time interval t7 – t8 as shown in Fig. 4(h), the high voltage side capacitor C2 charges L1 and the current in L1 becomes negative such that it can generate the ZVS condition for S6. Vbc is still zero and the energy stored in C3 is discharged back to the output capacitor C4.

At the time instant t8, S6 is turned on with ZVS, and both Vdo and Veo are equal to –Vb/2. Since S4 and S6 are simultaneously turned on in the next two intervals, the energy stored in the input side capacitor C2 is transferred into the output capacitor C4 for a short interval and then the energy is transferred back from C4 to C2 for a long interval, and net energy is injected to the high voltage side.

In Fig. 3, it can be concluded from the switch current waveforms of S1, S3 and S5 that ZVS is obtained for all the switches in the proposed high step-up ratio bi-directional DC-DC converter in forward mode operation.

III. THE REVERSE MODE OPERATION PRINCIPLES The reverse mode operation is defined as the power

transferred from the high voltage DC-link side Vd to the low voltage battery side Vb, or the charging operation if the converter is used for energy storage applications.

The proposed converter operation waveforms for the reverse mode operation are illustrated in Fig. 5. Again, V1 is the amplitude of the voltage between a and c in the high voltage side, as shown in Fig. 2. In steady state, V1 is equal to the half of the DC-link voltage Vd, or the capacitor voltage Vc2. V2 is the amplitude of the voltage between b and c in the high voltage side. It is the sum of the voltage across the high voltage winding of two transformers. Vdo, Veo are the voltage across the low voltage winding of two transformers. The amplitude of Vdo and Veo is equal to the half of the battery pack voltage Vb.

The voltage reference polarity and current reference direction are labeled in Fig. 2. In reverse mode operation, Vac should lead Vbc by a phase-shift angle φ. The phase-shift angle φ is defined in Fig. 5, where the center points of the positive part of the waveforms of Vac and Vbc are used as the reference time instant to calculate the phase-shift angle. For the low voltage side, Vdo and Veo are also modulated in the phase-shift manner, where Vdo leads Veo by an angle of (1-m)·π such that a three-level voltage waveform can be generated for Vbc.

From Fig. 6(a), at the time instant t0, the current in the equivalent series inductor L1 is negative and S1 is turned on with zero voltage switching (ZVS). In the time interval t0 – t1, Vac is equal to Vd/2, while Vdo is Vb/2 and Veo is –Vb/2. Then Vbc is zero, since it is the sum of these two voltages multiplied by the transformer turns ratio N. Therefore, the energy stored in L1 is discharged to C1 and the inductor current decreases. The energy stored in the filter capacitor C3 is discharged into C4.

In the time interval t1 – t2, Vac is still equal to Vd/2 and Vbc is zero. The current in L1 becomes positive and continues increasing, as shown in Fig. 6(b). The inductor L1 is charged such that S5 will be turned on with ZVS. The energy stored in the capacitor C4 is discharged back to C3.

Fig. 5: The reverse mode operation waveforms of the proposed converter

Fig. 6 (a): The t0-t1 interval of reverse mode operation

Fig. 6 (b): The t1-t2 interval of reverse mode operation

Fig. 6 (c): The t2-t3 interval of reverse mode operation

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In Fig. 6(c), at the time instant t2, S5 is turned on with ZVS. Then Veo becomes positive (Vb/2). Therefore, Vbc is equal to Vb·N (=V2) and is higher than Vd/2 (=V1). The energy stored in the inductor L1 is discharged to the output capacitor C3. In the meanwhile the energy is transferred from the DC-link to the high voltage side capacitor C2 and the output capacitor C3. The interval t2 –t3 is long and a major period to charge the battery pack.

At the time instant t3, the current in the inductor L1 becomes negative. In the time interval t3 – t4 (Fig. 6(d)), the inductor L1 is charged and the energy is transferred from the output capacitor C3 back to the DC-link side C1 such that S4 will be turned on with ZVS.

At the time instant t4, S4 is turned on with ZVS. Then Vdo is equal to –Vb/2 and Vbc becomes zero again. The energy stored in the inductor L1 is discharged into C1 and the current decreases. The energy stored in the capacitor C3 is discharged into the capacitor C4, as shown in Fig. 6(e).

In the time interval t5 – t6, the inductor L1 is charged again by the DC-link capacitor C1 and the current in L1 becomes positive (Fig. 6(f)). The energy stored in L1 will be used to turn on the high voltage side switch S2 with ZVS. In this interval, the energy in C4 is discharged into back into C3.

In Fig. 6(g), at the time instant t6, S2 is turned on with ZVS and Vac becomes negative (-Vd/2). The other half cycle of one switching period begins. After S2 is on, the energy stored in the L1 is discharged into C2 and the current in the inductor L1 decreases. Since Vbc is still zero, the energy in C4 continues discharging into C3 in the time interval t6 – t7.

In the time interval t7 – t8 as shown in Fig. 6(h), the high voltage side capacitor C2 charges L1 and the current in L1 becomes negative such that it can generate the ZVS condition for S6. Vbc is still zero and the energy stored in C3 is discharged back to the output capacitor C4.

At the time instant t8, S6 is turned on with ZVS, and both Vdo and Veo are equal to –Vb/2. Since S4 and S6 are simultaneously turned on in the next the interval, the energy stored in the input side capacitor C2 is transferred into the output capacitor C4.

In Fig. 5, it can be concluded from the switch current waveforms of S1, S3 and S5 that ZVS is obtained for all the switches in the proposed high step-up ratio bi-directional DC-DC converter in reverse mode operation.

IV. STEADY STATE ANALYSIS OF THE CONVERTER In the time interval t0 – t2, Vac=V1=Vd/2 and Vbc=0, so the

inductor current is,

)(1)( 01

101 ttL

VItiL −⋅⋅+= , (1)

In time interval t2 – t4, Vbc=V2=N·Vb, let V1<V2, the inductor current decreases,

)(1)()( 21

1211 ttL

VVItiL −⋅⋅−−= , (2) In time interval t4 – t6, Vbc=0 and the inductor current

increases again,

Fig. 6 (d): The t3-t4 interval of reverse mode operation

Fig. 6 (e): The t4-t5 interval of reverse mode operation

Fig. 6 (f): The t5-t6 interval of reverse mode operation

Fig. 6 (g): The t6-t7 interval of reverse mode operation

Fig. 6 (h): The t7-t8 interval of reverse mode operation

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)(1)( 41

121 ttL

VItiL −⋅⋅+= , (3)

In steady state, I3=-I0, and also substitute the high voltage bridge to the low voltage bridge phase-shift angle φ and the phase-shift angle between the two low voltage side bridges (1-m)·π into (1)-(3), one obtains,

sfLVmVI

⋅⋅⋅+−

=1

210 4

, (4)

sfLVVmVmI

⋅⋅⋅⋅⋅−⋅⋅−⋅⋅

=1

1121 4

2π

ϕππ , (5)

sfLVVmVmI

⋅⋅⋅⋅⋅−⋅⋅+⋅⋅−

=1

1122 4

2π

ϕππ , (6)

where fs is the switching frequency, m is the modulation index of Vbc. Define the voltage ratio,

1

2

VVd = , (7)

and design d>1. The power transferred by the converter is,

ϕπ

⋅⋅⋅⋅⋅⋅

=⋅⋅

= ∫6

01

211

2)(2 t

ts

Ls fL

dmVdttiT

VP , (8)

where Ts is the switching cycle period and the unit of φ is rad. The rms current of the inductor is,

s

t

t Ls

rms

fLmddmdmdmmdV

dttiT

i

⋅⋅⋅++−++−

=

= ∫

1

222232222321

6

0

2

12)12233(3

)(2

πϕπππππ

, (9)

Define the reactive power of the converter as,

12

12 2 LfiXiQ srmsLrms ⋅⋅⋅⋅=⋅= π , (10)

Substitute (9) into (10) and the ratio of the output power to the reactive power is,

22223222232 1223312

ϕπππππϕ

mddmdmdmmdmd

QPRpq

++−++−

== , (11)

For the high voltage side switch S1 and S2 to obtain zero voltage switching,

04 1

210 <

⋅⋅⋅+−

=sfLVmVI , or

dVVm 1

2

1 =< , (12)

It can be found that the ZVS of high voltage side switches is only controlled by the phase-shift angle of the two low voltage full bridges, or the modulation index m.

In order to let the low voltage side switches obtain ZVS,

<⋅⋅⋅

⋅⋅−⋅⋅+⋅⋅−=

>⋅⋅⋅

⋅⋅−⋅⋅−⋅⋅=

04

2

04

2

1122

1121

s

s

fLVVmVmI

fLVVmVmI

πϕππ

πϕππ

, (13)

By simplifying (13), one obtains

−⋅

>

−⋅

<

)1(2

)1(2

dm

dm

πϕ

πϕ , or )1(

2−

⋅< dmπϕ , (14)

The proposed operation principles described in part II and III requires (t2-t0)>0, so the phase-shift angle φ should meet,

2)1( πϕ m−< , and

2)1(maxπ

ϕ m−= , (15)

In order to find the optimum modulation index m, for given V1, V2 or d, the maximum ratio of the output power to reactive power can be obtained by solving

Fig. 7: The optimum modulation index m to minimize Rpq

Fig. 8: The ratio of the output power to reactive power and ZVS region

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=

=

0

0

pq

pq

Rddd

Rdmd

, (16)

and then selecting the result which simultaneously meets (12), (14) and (15). The optimum m is shown in Fig. 7 with different ratio of V1 to V2.

To find the optimum operation region of the proposed converter, the ratio of the output power to the reactive power, defined by (11), is plotted in Fig. 8. The ZVS boundary for switches defined by (14) and maximal phase-shift angle defined by (15) are also shown in Fig. 8. By designing the converter in the region with optimum modulation index and small φ, ZVS can be obtained for full load range and wide voltage range, and high efficiency can be potentially obtained.

Fig. 9 shows an alternative topology for the proposed high step-up ratio bi-directional DC-DC converter with additional two-time gain, where four half-bridge legs are used in the low voltage side compared with two half-bridge legs in Fig. 2. In this configuration, both Vde and Vfg are square waveforms. Vde is equivalent to Vdo and Vfg is equivalent to Veo in above analysis. Therefore, the above analysis also applies to the converter shown in Fig. 9, except that V2=2·N·Vb.

V. PROTOTYPE AND EXPERIMENTAL RESULTS A 14.4V input, 360V/450W output and 400kHz prototype

was build to verify the operation of proposed novel high step-up ratio bi-directional DC-DC converter based on the topology shown in Fig. 9. The specification of this prototype is listed in Table I. Parallel of the low voltage side sections were employed in the prototype.

Fig. 10: Test waveforms of the prototype with d=1.28, m=0.665, φ=-15°

Fig. 11: Test waveforms with d=1.28, m=0.665, φ=0° (no load)

Fig. 12: Test waveforms of ZVS for HV side MOSFET S1 at no load

Fig. 13: Test waveforms of ZVS for LV side MOSFETs S4 and S8 at no load

Fig. 9: The proposed alternative topology for high step-up ratio bi-directional DC-DC converter

TABLE I: THE PROTOTYPE SPECIFICATIONS

Low voltage side voltage Vb (V): 14.4

High voltage side voltage Vd (V): 360

Switching frequency fs (kHz): 400

High voltage side MOSFETs: STW45NM50FD

Low voltage side MOSFETs: BSC014N03LS G

Transformer turns ratio N: 8

Equivalent inductance L1 (uH): 12

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The test waveforms of the prototype in charging operation with d=1.28, m=0.665, φ=-15° are shown in Fig. 10. The inductor L1 current waveform is measured in the high voltage side of the transformer. In this case, the high voltage side output Vac leads the amplified low voltage side output Vbc by φ. The test waveforms at no load condition are shown in Fig. 11 with the same d and m but φ is zero. The ZVS is obtained for all the switches even at no load condition. Fig. 12 shows the ZVS of high voltage side switch and Fig. 13 shows the ZVS of low voltage side MOSFETs from the two phase-shifted low voltage full bridges.

The efficiency of the proposed novel high step-up ratio bi-directional DC-DC converter is shown in Fig. 14 for both forward and reverse mode operation. At 450W, the measured efficiency of the prototype is 94.1%.

VI. CONCLUSIONS A novel high step-up ratio bi-directional DC-DC converter

is proposed. By shifting the square-wave output voltage of low voltage side bridges and the coupling of two transformers, ZVS can be achieved for all the switches in the converter. The direction and magnitude of the power flow is controlled by the phase shift angle between the high voltage half bridge and low voltage bridges. An experiment prototype was built to verify the analysis as well as the operation of proposed converter. High voltage gain and ZVS are proved. 94.1% efficiency is achieved at 450W.

ACKNOWLEDGMENT This work was supported by ERC Program of the National

Science Foundation under Award Number EEC-08212121.

REFERENCES [1] A. Q. Huang, M. L. Crow, G. T. Heydt, J. P. Zheng, and S. J. Dale,

“The future renewable electric energy delivery and management (FREEDM) system: the energy internet,” in Proceedings of the IEEE, vol.99, no.1, pp:133 - 148, Jan.2011.

[2] S. Vazquez, S. M. Lukic, E. Galvan, L. G. Franquelo and J. M. Carrasco, “Energy Storage Systems for Transport and Grid Applications,” in IEEE Transactions on Industrial Electronics, vol.57, no.12, pp.3881-3895, Dec. 2010.

[3] H. Fakham, D. Lu, and B. Francois, “Power control design of a battery charger in a hybrid active PV generator for load-following applications,” in IEEE Transactions on Industrial Electronics, no. 58, vol.1, pp:85-94, Jan. 2011.

[4] M. Bragard, N. Soltau, R. W. De Doncker, and A. Schmiegel, “Design and implementation of a 5 kw photovoltaic system with li-ion battery and additional dc-dc converter,” In IEEE 2010 Energy Conversion Congress and Exposition (ECCE), pp. 2944-2949, Sept. 2010.

[5] A. Emadi, Y. J. Lee and K. Rajashekara, “Power electronics and motor drives in electric, hybrid electric, and plug-in hybrid electric vehicles,” in IEEE Transactions on Industrial Electronics, vol. 55, no. 6, pp. 2237-2245, Jun. 2008.

[6] Y. Du, X. Zhou, S. Bai, S. M. Lukic, S and A. Q. Huang, "Review of non-isolated bi-directional DC-DC converters for plug-in hybrid electric vehicle charge station application at municipal parking decks," in IEEE 2010 Applied Power Electronics Conference and Exposition, pp.1145-1151, Feb. 2010.

[7] W. Su, and M.-Y. Chow, "Performance Evaluation of An EDA-based Large-scale Plug-in Hybrid Electric Vehicle Charging Algorithm", IEEE Trans. Smart Grid, Special Issue on Transportation Electrification and Vehicle-to-Grid Application, June 2011.

[8] Y. Du, S. M. Lukic, B. S. Jacobson and Alex Q. Huang, "Review of high power isolated bi-directional DC-DC converters for PHEV/EV DC charging infrastructure," in IEEE 2011 Energy Conversion Congress and Exposition (ECCE), pp.553-560, 17-22 Sept. 2011.

[9] L. Qu and W. Qiao, “Constant Power Control of DFIG Wind Turbines With Supercapacitor Energy Storage,” in IEEE Transactions on Industry Applications, vol.47, no.1, pp.359-367, Jan./Feb.2011.

[10] W. S. Liu, J. F. Chen, T. J. Liang, R. L. Lin and C. H. Liu, “Analysis, Design, and Control of Bidirectional Cascoded Configuration for a Fuel Cell Hybrid Power System,” in IEEE Transactions on Power Electronics, vol.25, no.6, pp.1565-1575, Jun.2010.

[11] R. W. De Doncker, D. M. Divan and M. H. Kheraluwala, “A three phase soft-switched high-power-density dc/dc converter for high-power applications,” in IEEE Transactions on Industry Applications, vol.27, no.1, pp.63-73, Jan./Feb.1991.

[12] S. Inoue and H. Akagi, “A bidirectional DC–DC converter for an energy storage system with galvanic isolation,” in IEEE Transactions on Power Electronics, vol.22, no.6, pp.2299-2306, Dec.2007.

[13] S. Han and D. Divan, “Bi-directional DC/DC converters for plug-in hybrid electric vehicle (PHEV) applications,” in IEEE 2008 Applied Power Electronics Conference and Exposition, pp.:784-789, Feb.2008.

[14] F. Krismer and J. W. Kolar, “Accurate Power Loss Model Derivation of a High-Current Dual Active Bridge Converter for an Automotive Application ,” in IEEE Transactions on Industrial Electronics, vol.57, no.3, pp.881-891, Mar. 2010.

[15] H. Bai and C. Mi, “Eliminate reactive power and increase system efficiency of isolated bidirectional dual-active-bridge DC-DC converters using novel dual-phase-shift control,” in IEEE Transactions on Power Electronics, vol.23, no.6, pp.2905-2914, Nov.2008.

[16] H. Xiao, L. Guo and S. Xie, “A ZVS Bidirectional DC–DC Converter With Phase-Shift Plus PWM Control Scheme,” in IEEE Transactions on Power Electronics, vol.23, no.2, pp.813-823, Apr.2008.

[17] H. Zhou and A. M. Khambadkone, “Hybrid Modulation for Dual-Active-Bridge Bidirectional Converter With Extended Power Range for Ultracapacitor Application,” in IEEE Transactions on Industry Applications, vol.45, no.4, pp.1434-1442, Jul./Aug. 2009.

[18] G. G. Oggier, G. O. Garcia and A. R. Oliva, “Switching Control Strategy to Minimize Dual Active Bridge Converter Losses,” in IEEE Transactions on Power Electronics, vol.24, no.7, pp.1826-1838, Apr.2009.

[19] A. K. Jain and R. Ayyanar, “PWM control of dual active bridge-comprehensive analysis and experimental verification,” in IEEE Transactions on Power Electronics, vol.26, no.4, pp.1215-1227, Apr. 2011.

Fig. 14: The prototype efficiency test result

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