__________________________

The work described in this paper was fully supported by a grant from the Research Grants Council of the Hong Kong Special Administrative Region, China (Project No.: CityU 112708).

A Passive Lossless Snubber Cell with Minimum Stress and Wide Soft-Switching Range

River T. H. Li Student Member, IEEE

City University of Hong Kong Kowloon Tong, Hong Kong

50253423@student.cityu.edu.hk

Henry S.H. Chung Senior Member, IEEE

City University of Hong Kong Kowloon Tong, Hong Kong

eeshc@cityu.edu.hk

Abstract – A passive lossless snubber cell and its dual structure for reducing the switching loss of a range of switching converters are presented. The proposed snubber cell has several advantages over existing snubbering techniques. First, it provides zero-current-switching (ZCS) and zero-voltage-switching (ZVS) conditions for turning on and off, respectively, the switch over a wide load range. Second, it does not introduce extra voltage stress on the switch. Third, by taking the ripple current through the switch into account, the peak switch current during the snubber resonance period is designed to be less than the designed switch current without the snubber. Hence, the proposed snubber does not introduce extra current stress on the switch. The operating principle, procedure of designing the values of the components and soft-switching range of the snubber will be given. Connections of the snubber cells to different switching converters will be depicted. A performance comparison among the proposed snubber and two previously studied snubber cells will be addressed. The proposed snubber has been successfully applied to an example of a 200W, 380V/24V, 100kHz two-switch flyback converter operating in continuous conduction modes. Experimental results are in good agreement with the theoretical predictions.

Index terms – Snubbers, passive lossless snubber, soft-switching

I. INTRODUCTION Power switches in traditional pulsewidth modulated

(PWM) converters are operated in hard switching conditions. During the turn-on and turn-off processes, the devices have to withstand high voltage and current simultaneously, resulting in high switching losses and stresses. The classical method of reducing switching losses, dv/dt, di/dt, and stresses is to use dissipative snubbers [1]-[2]. However, dissipative snubbers produce undesirable power losses, thus limiting their application to small power or low frequency converters.

In order to overcome switching loss and enable high-frequency operation, many active snubbers that utilize soft-switching techniques have been proposed [3]-[16]. They are operated during the short switching time to perform zero-voltage-switching (ZVS) or zero-current-switching (ZCS). The main goal of the active snubbers is to obtain the advantages of the PWM and resonant converters. The

former ones have fixed-frequency operation with square current and voltage while the latter ones have low switching losses [9]. However, such merits are often offset by requiring additional switch and control circuitry, limited operating range, and high voltage/current stresses on the switches. Due to the presence of an additional switch, the switching losses will also be increased [17].

Passive snubbers are still attractive alternatives as they are generally easy to design and require fewer components. A typical passive snubber consists of two parts: turn-on snubber and turn-off snubber. The turn-on snubber limits the rate of rise of the current through the switch and allows the voltage across it to drop before its current starts increasing. The turn-off snubber limits the rate of rise of the voltage across the switch after it is turned off. The switch is made to turn on with near ZCS and turn off with near ZVS, resulting in reduced switching losses [1]-[2].

A snubber has to go through two processes, namely energy absorbing and resetting processes. The total duration taken for these two processes determine the minimum and maximum duty time of the switches. The simplest form of the energy absorbing circuit for the turn-on snubber is an inductor in series with the switch while the one for the turn-off snubber is a capacitor in parallel with the switch. Most snubber structures distinguish themselves from others by the difference of their resetting circuits.

The simplest form of the energy resetting circuit is based on using a resistor. To alleviate energy-inefficient problem associated with the dissipative snubbers, various passive lossless snubbers have been proposed [18]-[38]. The major concept is to reset the energy absorbing circuits by releasing or re-circulating the energy stored to an energy tank, such as an inductor, a capacitor, supply and/or load.

A straightforward approach to resetting the snubber is to use a switching converter, such as the forward or flyback converter in [18]-[28], to re-circulate the energy stored in the snubber. The switching action of the main switch is made common to both the main power conversion and snubber energy conversion. However, the transformer coupling effect introduces additional voltage stress across the switch and the leakage inductance of the transformer or coupled inductors also generate undesirable voltage spikes.

Another attractive approach is based on using resonant circuits with passive reactive elements and diodes only [29]-[43]. The structures of those circuits are simple and can be incorporated readily into existing converters. The energy is

685978-1-4244-2893-9/09/$25.00 ©2009 IEEE

transferred to the source or load through several LC resonant paths created by the main switch and the diodes.

One of the representative works is the structure proposed in [29]. Many counterparts for specific applications like power factor correctors, local power supplies and inverters have also been proposed [29]-[36]. The concept of operation is illustrated with the snubber proposed in [29] for the boost converter. The energy stored in the snubber inductor is firstly released to the snubber capacitor after the switch is off. When the switch is on, the stored energy will be released to a storage capacitor and to the load through the resonant paths formed by the diodes, snubber inductor, snubber capacitor, and the storage capacitor. Some improved circuits with saturable inductors added for reducing reverse recovery current of the main diode have been proposed in [37]-[39]. However, the voltage generated across the saturable inductor causes extra voltage stress on the switch and thus voltage clamping devices, like lossy zener diode, have to be added.

A systematic approach to investigating the properties and synthesis of the generalized form of that category of snubbers has been given in [40]-[43], in which different sets of minimum voltage stress (MVS) and non-minimum voltage stress (NMVS) circuit cells have been derived. Many previously-proposed structures can be derived from the two cells. The snubbers with MVS have found to have narrower soft-switching range than the ones with NMVS.

Although all snubbers discussed can help reduce switching losses, they typically exhibit at least two of the following limitations: 1) The voltage stress across the switch, particularly in

snubbers with NMVS, is high because the variation of the voltage across the snubber capacitor during resonance will generate additional off-state stress on the switch.

2) Soft-switching cannot be ensured at heavy load because the snubber cannot be completely reset before the next energy absorbing process. For example, when the load current is high, the snubber inductor has to take a long time to completely discharge. The storage capacitor will be discharging faster than that of snubber inductor in the resetting process. Then, the switch will not be turned on with ZCS.

3) The current stress on the switch is high because the switch current includes the discharging current of the snubber capacitor and main current flow for energy conversion.

Fig. 1 shows the proposed snubber cells, namely Cell A and Cell B, for dealing with the above constraints. The proposed snubber does not introduce extra voltage stress on the switch over the operating range and the energy stored in the snubber inductor Ls is ensured to be completely discharged before the storage capacitor Cst starts to be discharged. By taking the ripple current through the switch into account, the peak switch current during the snubber resonance period is designed to be less than the designed switch current without the snubber. Thus, no extra current stress is introduced on the switch.

(a) Cell A. (b) Cell B.

Fig. 1. Circuit schematic of the proposed snubber.

The operating principle of the snubber is described in Sec. II and design procedure is given in Sec. III. A performance comparison of the proposed snubber and the NMVS snubber [43] operating at different power levels will be given in Sec. IV. The proposed snubber has been successfully applied to a 200W, 380V/24V, 100kHz two-switch flyback converter. Experimental results will be given in Sec. V.

II. OPERATING PRINCIPLE Each cell shown in Fig. 1 has three nodes, namely nodes

a, b, and c. To achieve the snubbering function, a switched current source of magnitude Ix should be connected across nodes a and b, and a switched voltage source of magnitude Vx should be connected across nodes b and c. The switch that requires snubbering action is connected to node c. Fig. 2 illustrates the connection. The major difference between cells A and B is that the direction of current flowing through node a. Cell A has the current going into node a while cell B has the current coming out of node a. Table I shows the values of Ix and Vx of different converters. Fig. 3 depicts the integration of the proposed snubber cells into different converters.

The concept of operation is to firstly transfer the snubber capacitor energy to a storage capacitor Cst through a resonance process and then deliver it to the output and/or input in the successive switching action. As the snubber capacitor, Cs, and snubber inductor, Ls, are fully discharged within the switching period, the main switch can be turned on with zero current and turned off with zero voltage.

TABLE I

Value of Ix and Vx for different converters. Converter type Ix Vx

Buck IL Vin

Boost IL Vout

Buck-boost IL Vin + Vout

Ćuk IL1 + IL2 VC1

SEPIC IL1 + IL2 VC1 + Vout

2-switch forward Ipri,Tx Vin

2-switch flyback Ipri,Tx Vin

(a) Cell A (b) Cell B

Fig. 2 Required connection of the proposed snubber.

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Cell A Cell B

(a) Buck converter.

Cell A Cell B

(b) Boost converter.

Cell A Cell B

(c) Buck-boost converter.

Cell A Cell B

(d) Ćuk converter.

Cell A Cell B

(e) SEPIC.

(f) Two-switch forward converter. (g) Two-switch flyback converter. Fig. 3. Connections of the snubber cells to different switching converters.

Fig. 4 shows the key voltage and current waveforms

together with the state of diodes of Cell B. Fig. 5 shows the modes of operation. For the sake of simplicity, several assumptions have been made in the analysis as follows:

1) The switch and diodes are ideal. They have zero on-state resistance, infinite off-state resistance and zero junction capacitance.

2) All energy storage components are free of loss without parasitic element.

Before the start of a switching cycle (i.e., t0), the main diode of the converter is in freewheeling stage as shown in Fig. 5(i). The voltages of Cs and Cst are zero, and the currents of Ls and Lst are all zero. The cyclical operation is described as follows.

Mode 1a [Fig. 5(a)] ( '10 ttt <≤ ): The main switch is turned on with zero current, where Ls and the leakage inductance of the transformer, Lk, in the isolated converter, limits the rate of rise of the switch current. This mode ends when the current at node c equals Ix. The value of Ix of each converter is tabulated in Table I. Thus,

xLs

s Vdt

diL = (1)

where iLs is the current through Ls. This mode ends when the switch current reaches Ix. Hence,

x

xs

VILtt =− 01 ' (2)

Mode 1b [Fig. 5(a)] ( 11 ' ttt <≤ ): The output diode enters into the reverse recovery process. The duration of the process is approximated by the following equation

)1(2'

)'(21

11

11

SIQtt

QItt

rr

rr

rrrr

+=−

=−

Δ

Δ (3)

Fig. 4. Key voltage and current waveforms and state of diodes of Cell B.

687

where ΔQrr is the reverse recovery charge of the diode, Irr is the peak reverse-recovery current, and S is snappiness factor [1]. Since

xrr

s VStt

IL =+−

)1('11

(4)

Equation (3) can be expressed as

)1(

2SLVQI

s

xrrrr +

= Δ (5)

The peak current SI flowing through the switch is

rrinS III +=ˆ (6) By using (3) and (5), the diode is completely off when

)1(

2'11 SV

LQtt

x

srr

+=−

Δ (7)

Mode 2 [Fig. 5(b)] (t1 < t < t2): The energy stored in Cs is transferred to Cst through the resonance path Cs - Ls - Cst - D3 - Lst.

)(sin)( 122

ttZVIti a

a

xxLs −+= ω (8)

)(sin)()( 122

ttZVItiti a

a

xxLsLst −=−= ω (9)

)](cos1[1

1)( 12 ttx

Vtv axCs −−+

= ω (10)

)](cos1[1

)( 12 ttx

xVtv axCst −−+

= ω (11)

where, eq

eqa C

LZ =2 ,

eqeqa CL

12 =ω , stseq LLL += ,

sts

stseq CC

CCC

+= , and

st

s

CCx = is the ratio between Cs and

Cst. This mode ends when )( 2tvCs = xV . Thus, by using (10), xtta −=− )(cos 122ω (12) By substituting (12) into (11), xCst Vxtv =)( 2 (13)

In order to ensure the existence of 2t for charging Cs completely, the value of x in (12) should be less than or equal to one. Therefore, sst CC ≥ (14)

)sin2

(1 1

212 xtt

a

−+=− πω

(15)

( ) 2122 1sin xtta −=−ω (16)

By substituting (16) into (8) and (9),

xeq

sxLs Vx

LCIti −+= 1)( 2 (17)

xeq

sLst Vx

LCti −= 1)( 2 (18)

Mode 3 [Fig. 5(c)] (t2 < t < t3): D1 conducts. The energy stored in Lst is transferred to Cst through the resonance path, D1 - Ls - stC - D3 - stL .

])([cos)( 222 bbeq

sxxLs tt

LC

VIti φω +−+= (19)

])([cos)( 222 bbeq

sxLst tt

LC

Vti φω +−= (20)

xCs Vtv =)( (21)

])([sin)( 222 bbxCst ttVxtv φω +−= (22)

where st

eqb C

LZ =2 ,

steqb

CL1

2 =ω , and

xx

b −= −

1tan 1

2φ .

This mode ends when Lst is fully discharged, i.e., 0)( 3 =tiLst . Thus, by using (20),

x

xttb

−=− − 1tan1 1

223 ω

(23)

By substituting (23) into (19)-(22), xLs Iti =)( 3 (24) 0)( 3 =tiLst (25) xCs Vtv =)( 3 (26)

xCst Vxtv =)( 3 (27) Mode 4 [Fig. 5(d)] (t3 < t < t4): The switch current equals

Ix. This mode defines the duty cycle of the main switch. Equations (24)-(27) still hold in this mode. This mode ends when the main switch is switched off with zero voltage at t4.

Mode 5 [Fig. 5(e)] (t4 < t < t5): The main switch is switched off with zero voltage, Cs is discharged by Ix. Thus, xLs Iti =)( (28) 0)( =tiLst (29)

)()( 4ttCIVtv

s

xxCs −−= (30)

xCst Vxtv =)( (31) This mode ends at t5 when

x

CstCs

Vx

tvtv

=

= )()( 45 (32)

By using (30),

xx

s VICxtt )1(45 −=− (33)

Mode 6 [Fig. 5(f)] (t5 < t < t6): Cs continues to be discharged by Ix. Cst starts discharging by Ix through D4.

)](cos[1

)( 55 ttxx

Iti xLs −+

+= ω (34)

0)( =tiLst (35)

688

( )

( ) )(sin1

)()1(

555

5

ttxCx

I

ttxC

IVxtv

st

x

st

xxCs

−+

−−+

−=

ωω

(36)

)(sin

)1(

)()1(

)(

555

5

ttxC

I

ttxC

IVxtv

st

x

st

xxCst

−+

+−+

−=

ωω

(37)

where eqs CL

15 =ω .

This mode ends at t6 when iLs (t6) = 0. By using (34),

)sin2

(1 1

556 xtt −+≅− π

ω (38)

( ) 0)( 56 == titi LstLst (39)

)1

sin2

(

11)(

21

6

xx

x

xx

CL

xIVxtv

st

sxxCs

−++

++−≅

−π (40)

)1sin2

(

11)(

21

6

xx

xx

CL

xI

Vxtvst

sxxCst

−−+

++−=

−π (41)

Mode 7 [Fig. 5(g)] (t6 < t < t7): Cst is discharged to the load by Ix. 0)( =tiLs (42) 0)( =tiLst (43) =)(tvCs )( 6tvCs (44)

)()()( 66 ttCItvtv

st

xCstCst −−= (45)

This mode ends at t7 when )()( 67 tvtv CsCst = (46) By using (44) and (45),

x

xCLtt sts−=− 1

67 (47)

)1sin2

(

111)(

21

7

xxx

xx

xCL

IVxtvst

sxxCst

−++

++−=

−π (48)

Mode 8 [Fig. 5(h)] (t7 < t < t8): Cs is discharged by Ix again through D2. Cst continues to be discharged by Ix through D4. This mode ends when both Cs and Cst are fully discharged. 0)( =tiLs (49) 0)( =tiLst (50)

)()1(

)()( 77 ttxC

Itvtvst

xCsCs −

+−= (51)

)()1(

)()( 77 ttxC

Itvtvst

xCstCst −

+−= (52)

This mode ends at t8 when vCs(t8) and vCst(t8). By using (48) and (52),

]]1

sin2

[11

)1([

21

78

xx

xx

xLx

CIV

CLtts

s

x

xss

−++

+

−+=−

−π (53)

Mode 9 [Fig. 5(i)] (t8 < t < t9) : D conducts and Ix will supply to the load. This mode ends when the main switch is turned on again. This completes one switching cycle.

Based on Fig. 4, the minimum on time ton,min and minimum off time toff,min of the main switch are calculated as follows. ton,min is the minimum time required to transfer the energy stored in the snubber capacitor Cs to the reset circuit while toff,min is the minimum time required to transfer the energy stored in the reset circuit to the other part of the converter

(a) Mode 1(a) and mode 1(b) [t0 ~ t1]. (b) Mode 2 [t1 ~ t2].

(c) Mode 3 [t2 ~ t3]. (d) Mode 4 [t3 ~ t4].

(e) Mode 5 [t4 ~ t5]. (f) Modes 6 [t5 ~ t6].

(g) Mode 7 [t6 ~ t7]. (h) Mode 8 [t7 ~ t8].

(i) Mode 9 [t8 ~ t9].

Fig. 5. Modes of operation.

689

circuit. ton,min is the total time taken from Mode 1 to Mode 3 (i.e., from t0 to t3) while the one of toff,min is the total time taken from Mode 5 to Mode 8 (i.e., from t4 to t8). ton,min is obtained by adding up the time durations given in (2), (15), and (23), and toff,min is obtained by adding up the time durations given in (33), (38), (47), and (53). Thus,

11tan1sin2

1 1

2

1

2min, −+⎟

⎠⎞

⎜⎝⎛ ++= −−

xx

VILt

bax

xson ω

πω

(54)

and )11(min, xIVCtx

xsoff += (55)

III. DESIGN PROCEDURE The values of the passive components are designed as follows: 1. Design the value of Ls

The value of Ls is determined by considering the rate of rise of the switch current in Mode 1b when Ix is maximum, Ix,max. Thus,

max,

,1

x

xdbs I

VtL ≤ (56)

where t1b,d is the designed duration of Mode 1b and Ix,max is the peak value of Ix. It should be noted that Ls can be realized by the leakage inductance of the transformer in the isolated converters, such as flyback and forward converter.

2. Design the values of Cs and Cst Cs and Cst are designed by the following iterative procedure. Firstly, x ∈ [0, 1] is randomly selected. Secondly, in order to ensure that the switch voltage stress in Mode 5 is less than Vx. By substituting x into (40) and make vCs less than Vx, it can be shown that

22

1

2

1sin2

)1(

ˆ

1

⎟⎟

⎠

⎞

⎜⎜

⎝

⎛ −++

⎟⎟⎠

⎞⎜⎜⎝

⎛

++≥

−

xxx

xVI

xxLC

x

xss

π (57)

The minimum value of Ix, Ix,min, that can ensure soft-switching, is calculated by substituting x into (55) with the designed minimum off time toff,min,d. Thus,

)11(min,,

min,xt

VCI

doff

xsx += (58)

The value of x is accepted if Ix,min is below the designed minimum input current, for example, 5% of the peak input current with the converter powering rated load at minimum input voltage. Otherwise, another value of x is chosen and the above iterative is process is repeated. With the chosen value of x, Cst calculated from the value of Cs because

xC

C sst = .

3. Design of Lst The value of Lst is determined by considering that the current stress on the main switch is not higher than the designed current ripple ΔI on the main switch. Thus, based on (9),

ssx

st Lx

CI

VL −

+⎟⎠

⎞⎜⎝

⎛=1

2

Δ (59)

4. The value of ton,minis calculated by using (54) and the values of Cs, Cst, Ls, and Lst designed above and compared with the designed minimum duty time of the main switch dminTs. If the condition of

min,min ons tTd ≥ is satisfied, the above set of component values will be taken. Otherwise, new set of values will be chosen again.

IV. COMPARISON WITH PRIOR-ART SNUBBER The performance of the proposed snubber and NMVS snubber in [43] is compared. The two snubbers are designed for a 750W, 250V/48V buck converter studied in [43]. The proposed snubber is designed with the procedure given in Sec. III. Table II shows the component values of the proposed snubber. According to Table I, the values of Ix and Vx are 15.5A and 250V, respectively, at the rated load. Fig. 6 shows the operating region of the proposed snubber cell that can provide soft-switching conditions to the main switch. The region is formed by considering several boundaries. The first two boundaries relating to the minimum on and off times of the switch. They are ton,boundary and toff,boundary. ton,boundary is the relationship between the values of Vx and Ix that can just satisfy the minimum on time in (54). Similarly, toff,boundary is the relationship between the values of Vx and Ix that can just satisfy the minimum off time in (55). The next two boundaries are the maximum and m i n i m u m v a l u e s o f V x . The minimum value of Vx is 48V, the rated output voltage. The next boundary is the maximum value of Ix. The final boundary is the vertical line intercepting toff,boundary and maximum Vx. The snubber cell can provide soft-switching conditions for Ix varying between 0.28A to 15.5A. Fig. 7 shows the extra voltage stress on the main switch versus the inductor current. It can be seen that the extra voltage stress on the switch is kept zero until the inductor current IL is higher than 15.5A. Conversely, the NMVS snubber gives an extra voltage stress from 25V to 55V. At the rated condition,

690

Fig. 6. Operating region of the proposed snubber cell in a buck converter with 250V Vin and 48V Vout.

Fig. 7. Extra voltage stress on the main switch versus the inductor current.

(a) Without any snubber (b) With the proposed snubber (vds: 100V/div. ids: 1A/div). (vds: 100V/div. ids: 1A/div).

Fig. 8. Switching waveforms of the main switch. (Timebase: 1μs/div).

(a) Without any snubber (b) With the proposed snubber

Fig. 9. Switching trajectories of the main switch (x-axis: id 1A/div, y-axis: vds100V/div).

the current stress on the main switch with the proposed snubber is 18.11A while the one with NMVS snubber is 21.3A. It can be seen that the proposed snubber gives lower voltage and current stress than NMVS snubber on the main switch.

V. EXPERIMENTAL VERIFICATIONS A continuous conduction mode, 200W, 380V/24V two-

switch flyback converter has been built and evaluated. The switching frequency is 100kHz. Table III shows the component values of the snubber cell that designed by following the design procedure given in Sec. III. The leakage inductance of the flyback transformer is utilized as the snubber inductor and the value of 15μH. The snubber cell is designed such that the current peak in mode 2 is equal to the current ripple, 2.5A, of the converter at rated load. No extra current stress is put on the main switch.

Fig. 8(a) and (b) show the switching waveforms of the main switch with and without the proposed snubber cell respectively. Fig. 9(a) and (b) show the x-y plots corresponding to Fig. 8(a) and (b). With the proposed snubber cell, the trajectory loop area, and thus the switching loss, is significantly reduced. Fig. 10 shows the converter

efficiency against load power. By the proposed snubber cell, the efficiency is higher by about 2% than hard switching.

TABLE II Values of the components in the snubber for comparison.

Item Value Item ValueCs 2.1nF Cst 2.47nFLs 400nH Lst 10μH

TABLE III

Values of the components in the snubber for the two-switch flyback Item Value Item ValueCs 1nF Cst 2.2nFLst 22μH

Fig. 10. Converter efficiency against load power.

Fig. 11. Picture of the proposed snubber switch module for 2-switch

flyback converter.

Finally, Fig. 11 shows the picture of the prototype of the proposed snubber cell integrated into the main switches to form a snubber switch module for the 2-switch flyback converter.

VI. CONCLUSION Two passive lossless snubber cell structures with

minimum voltage and current stress on the main switch have been proposed. They can be applied to different converters, so that the main switch can be turned on with zero current and turned off with zero voltage. The operating principles and design procedure of the snubber cell have been described. The distinct feature of this snubber is that it does not introduce extra voltage stress on the switch and has wide operating range. The current stress is designed to be the same as the hard-switched converter. Thus, no extra current stress is added. A 200W, 380V/24V two-switch flyback converter prototype has been evaluated and provide good agreement with theoretical prediction.

References

The proposed snubber cells

Mosfet driver

vds

id

vin vds

id

80V voltage overshoot

vds vds

id id

2.82A3.0A

384V

460V

691

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