High-voltage high-frequency powerswitching transistor module with

switching-aid-circuit energyrecovery

B.W. Williams, Dip.Eng., B.Sc. B.E.. Ph.D.. M.Eng.Sc.

Indexing terms: Transistors, Switches and switching theory, Switching-aid-circuit energy recovery

Abstract: Optimal base-drive requirements for the high-voltage power switching transistor are given. Theserequirements are used to minimise switching stresses and the amount of switching-aid circuit energy to berecovered. A passive resonant recovery technique is analysed and employed to transfer stored switching-aidcircuit energy into the load. Experimental results for a 600 V, 100 A, 100 kHz Darlington switching moduleincorporating 95% efficient switching-aid circuit energy recovery are presented.

List of principal symbols

A = areaBmax — maximum magnetic flux densityC = capacitanceD — diodeF — FET switching elementHm — maximum magnetic flux intensity before satura-

tioni = instantaneous collector currentIc = collector currentIm = maximum collector currentI mag = magnetising current/ = aircore inductance/s = saturable inductancelT = total inductance (/s + l0)L = mean magnetic-flux path lengthn = CJCO

N = number of turnsR = resistanceSOA = safe operating areat = timetfi(tfv) = collector-current fall time (voltage)tOn(min) — minimum transistor on timeTj = junction temperaturev = instantaneous collector voltageKeo(sus) = sustaining collector-emitter breakdown voltageVm = supply voltageco0 = natural frequency = ^Jl/lT Co

Zo = characteristic impedance = y/lT/C0

1 Introduction

High-voltage power-switching transistors with collectorratings in excess of 1000 V and 200 A can be foundincreasingly in chopper and inverter drive systems. Driveratings of up to 1 MVA have been attained by using anumber of devices in parallel and series so as to achievehigh-voltage and current requirements. A Darlington tran-sistor drive is used in order to reduce base-drive ratings. Inorder to fully utilise the device's electrical capabilitiesrequires proper base-drive conditions and the use ofswitching-aid circuits, called snubbers. Optimal base-driveelectrical conditions minimise the collector switching time,and hence losses, thereby reducing the amount of snubber-

Paper 2863B (P6), received 6th September 1983

The author is with the Department of Electrical Engineering, Imperial College ofScience and Technology, Exhibition Road, London SW7 2BT, England

IEE PROCEEDINGS, Vol. 131, Pt. B, No. I, JANUARY 1984

ing required. Snubbers are used to further minimise tran-sistor switching losses. Under such conditions thetransistor can operate at its full ratings to frequencies inexcess of 100 kHz.

Two switching-aid circuits can be employed on power-switching transistors, one for transistor turn-on, the otherfor turn-off. Transistor turn-off losses are usually greaterthan the turn-on losses, and many transistor circuits onlyemploy the turn-off snubber. In the case of the turn-offsnubber, energy is diverted from the switching-off tran-sistor into a parallel capacitor. The turn-on switching-aidcircuit employs an inductor in series with the collector inorder to control the rate of rise of the collector currentduring the collector-voltage fall time. Generally this induc-tor energy jllj,, and also the capacitor snubber energyjCVm, are losses, ultimately being dissipated as heat inresistors. Nevertheless the total energy losses of theswitching-aid circuits plus transistor can be less than thosefor an unaided transistor.

At just a few kilohertz, 5% of the total power input maybe snubber losses. A number of methods have been pro-posed which reclaim some or all of this energy [1-6].Recovery schemes employing passive techniques arefavoured, but previous circuits have introduced transistorvoltages in excess of the supply voltage, thereby preventingthe transistor's full potential from being exploited [1-3, 5,6].

A simple integral turn-on and turn-off switching-aid andrecovery scheme is proposed within this paper, which isefficient, requires only eight passive components and doesnot impose any transistor-collector voltages in excess ofthe supply voltage.

2 Base drive requirements

One of the objectives of a high-performance base-drivecircuit is to reduce switching time, and hence minimisestresses and losses, particularly at high frequencies. Properbase-drive techniques greatly enhance the switchingrobustness of a power transistor and any resultant appar-ent base-circuitry complexity is therefore usually justifi-able, particularly in high-voltage applications. Thefollowing features are considered as being desirable tran-sistor base-circuit properties:

(i) floating base with fast optocoupler signals(ii) controllable current source base on-drive(iii) high-current pulse at turn-on(iv) low base-to-emitter impedance

(v) antisaturation circuit(vi) high reverse base current at turn-off(vii) base-to-emitter reverse voltage bias during the off

period(viii) overcurrent detection and linear-region operation

protection(ix) Darlington pair for high gain

3 Practical base-drive circuit

Fig. 1 shows a high-performance base-drive circuit whichincorporates all the desirable base-drive features listed inSection 2.

• 12V

nential current impulse into each base for fast turn-on. Theelements Lx and Cx form a resonant circuit with F3 , thesupply, F4 and both transistor base-emitter junctions, andthe resultant sinusoidal-current impulse rapidly cuts off thebase-emitter junction of Tv. The resonant circuit formedby L2, C2, F3 , the supply, F4 and the base emitter of T2produces a current impulse which rapidly cuts off the baseemitter of the Darlington output transistor T2. The diodesD3, D4, D5 and D6 prevent undesired capacitor currentreversal. Thus both Darlington transistors turn on and offsimultaneously, thereby significantly reducing turn-on andturn-off delay times.

Reverse base-emitter junction bias of 4-2 V for both(<20V)

Fig. 1 Practical high-performance transistor drive circuit

A single floating-voltage rail is used, which may beobtained from an auxiliary 50 Hz transformer or winding,or from a DC/DC convertor. A fully controlled HEXFETbridge controls the direction of Darlington base current.

The Zener-diode/Schottky-diode series pairs Dzl DR1and DZ2 DR2 limit the base-emitter junction reverse bias.The stabilising base-to-emitter 22 Q carbon low-induct-ance resistors Rt and R2 improve the voltage-supportingcapabilities of both Darlington transistors. Schottkydiodes Dbl and Db2 reduce reverse base-drive currentrequirements and losses in RY and R2.

The transistor Tc performs a 2 A controlled-current-source function while the diodes Das and D±, along withthe base emitter of Tc, form an antisaturation circuit forthe Darlington driver T\.

The capacitors C, and C2 are the key components inproviding base-current impulses at turn-on and turn-off toboth Darlington transistors. Fig. 2 shows the basic tech-niques employed [5] for achieving a boost forward basecurrent at turn-on due to an RC circuit decay, Fig. 2a, anda reverse base-emitter-junction sinusoidal-current impulseat turn-off due to an LC circuit oscillation in Fig. 2b. Thecapacitor performs a dual function, being utilised in boththe turn-on and turn-off circuit. As illustrated by thecapacitor initial charges in Fig. 2, the stored charge fromthe previous operation significantly reinforces the nextcycle.

At turn-on, R2 Cx and R4rC2 provide a decaying expo-

Darlington transistors is provided during the off state viathe path F4 DZ2 Dr2 DZi Drl DZ3 D2 D3 F3 and the supply.

Fig. 3 shows practical base voltage and current wave-12V

-V/R

12V

-v/z

Fig. 2 Base current boost at

a switch-on: forward base currentb switch-off: reverse base current

1EE PROCEEDINGS, Vol. 131, Pt. B, No. 1, JANUARY 1984

forms for the Darlington transistors Tx and T2. A high-current pulse at turn-on, Figs. 3a and 3b, ensures rapid

Fig. 3 Practical base-drive waveforms

a Turn-on base voltage and current for transistor T2

b Turn-on base voltage for transistor T, and total collector current for transistors7", and T2

c Turn-off base voltage for transistor T2

d Turn-off base voltage for transistor Tj, total collector current for transistors T,and T2, and current measurement noise (lowest trace)Current measurement = 5 A/div. Horizontal scale = 100 ns/div.

conduction, and the pulse magnitude and duration arevaried by adjusting the R4C2 values for T2 and R^Civalues for Tx. A large reverse base-emitter bias voltage andcurrent are used for fast turn-off. The magnitude andperiod of the reverse base current are specified by the L2

C2 and LlCl values employed. The R, C and / values usedare shown in Fig. 1. Reverse base currents are not shownin Figs. 3c and 3d because of current probe insertionimpedance effects. The current-probe ground noise duringthe collector-current fall time is also shown in Fig. 3d.

It can be seen in Figs. 3a and 3b that the initial turn-onbase-current pulse extends to incorporate the freewheeling-diode reverse recovery period. At turn-off both transistorbase-emitter junctions are reverse biased by 4.2 V, thuseliminating any collector-current focusing [5].

Fig. 4 shows the overvoltage or linear operation detec-12V

Fig. 4 Collector overvoltage detection and protection circuit

tion and protection circuit, which is basically a collector-voltage comparator. Normally the base-emitter junction ofthe pnp transistor To is cut off. The base-emitter junctionbecomes forward biased when the Darlington enters thelinear region, exceeding a voltage

Vref vTeb - Dov (1)

Transistor To turns on, and the monostable and then opto-coupler feedback elements are activated. The optocouplerfeeds back an isolated signal to the main control circuitry,which then initiates turn-off. A long delay time may occurbetween the detection of an overvoltage and the beginningof the actual transistor Darlington turn-off process asmany active elements are involved. Rapid initiation of theturn-off process is achieved by using the monostable 2 JISpulse to override the base control signal which allows thebase drive to be reversed less than 100 ns after linear oper-ation occurs. The monostable can be replaced by a flip-flop and other minor circuitry if the optocoupler feedbacksignal is not used.

The elements RDCD form a delay circuit to preventovervoltage detection while the collector voltage is fallingduring turn-on. The capacitor CD is discharged at turn-offwhen an overvoltage is detected because of the rising col-lector voltage. This generated signal is redundant.

Although the base-drive circuitry may appear quitecomplex, all components are low powered and a minimumof magnetic components is involved. The required andachieved high performance of the Darlington transistor

IEE PROCEEDINGS, Vol. 131, Pt. B, No. 1, JANUARY 1984

switching module justifies the techniques and circuitryimplemented. The maximum base switching rate is inexcess of 400 kHz.

4 Switching-aid networks

During both the switch-on and switch-off intervals, for aninductive load, a period exists when the transistor simulta-neously supports the supply voltage Vm and conducts thefull load current Im. The base-drive circuit cannot alter thepeak power loss, but a good base drive can minimise theswitching duration, and hence energy losses, by minimisingrise and fall times.

Power switching transistors usually have switchingtimes of the order of microseconds. Thus the whole safeoperating area (SOA) bounded by Vceo(sus) and ICmax can beexploited during switching without second breakdown orthe need for power derating. Therefore, in switching appli-cations, the overriding operation constraint is to limit thejunction operating temperature to below the maximumallowable junction temperature TJmax [7]. A switching-aidnetwork modifies the switching trajectory across the SOAand avoids the condition of simultaneous maximumvoltage and current. Transistor losses are reduced and thetransistor's full electrical current and voltage ratings canbe exploited provided Tjmax is not exceeded.

The typical turn-on and turn-off unified switching-aidcircuit is shown in Fig. 5. The inductor /s controls the rate

4-

Dsy

c

ov TFig. 5 Unified switching-aid circuit for power transistors

of rise of collector current at turn-on, while the capacitorCs controls the rate of rise of collector voltage at turn-off.In each case, the snubber prevents a switching conditionwhere Im and Vm occur simultaneously. The capacitor turn-off snubber function is well known and treated extensivelyin numerous relevant texts [5]. The inductor turn-onsnubber function can be designed using a similar approachoutlined in Reference 5 for the turn-off snubber.

5 Switching-aid circuit with totalenergy recovery

Various techniques for switching-aid-circuit energy re-covery or partial recovery have been considered [1-6]. Anelegant solution is deemed to be any recovery techniquewhich involves:

(a) only passive, lossless elements (/, C, D)(b) no resistors or Zener diodes(c) no transistor current stresses and no collector

voltage in excess of Vm

10

(d) recovered energy transferred to the load(e) high efficiency and high-frequency operation.

A circuit for the proposed technique is shown in Fig. 6 andincorporates, with eight passive components, all the desir-

Fig. 6 Proposed unified passive switching-aid-circuit energy recoveryscheme

able features listed. The configuration includes bothturn-on and turn-off switching-aid circuits with unifiedtotal energy recovery from both switching-aid circuits.

5.1 Switching-aid circuit and energy-recovery oper-ation

With reference to Fig. 6, consider the transistor T2 in theoff-state supporting the voltage Vm. The load current Im isconducted by the load freewheeling diode Df. The snubbercapacitor Cs is charged to the supply voltage Vm while Co istotally discharged. Various circuit voltage and currentwaveforms are shown in Fig. 7.

5.1.1 Transistor turn-on: The saturable inductor /s isdesigned to have a low magnetising current (Imag =HmL/N) and to saturate after the transistor voltage hasfallen to near zero at turn-on (Vm = 2NBmaxA/tfv). Thefreewheeling diode recovers at a di/dt controlled by thesaturated inductance of /s. The diode recovery current isconducted by the transistor, with the load current beingtransferred from the freewheeling diode Df to the tran-sistor T2.

Simultaneously at transistor turn-on, the snubbercapacitor Cs resonates its stored energy to capacitor Co viathe path Cs Do l0 Co ls T2. The diode Do prevents any currentreversal after energy transfer from Cs to Co is complete. Ifwe neglect circuit losses, let

Cs = nC0 where n > 0

and

then the snubber-capacitor voltage changes according to

(V) (2)

while the capacitor Co charges according to

VCo = Vm — — (1 - cos cot) (V) (3)

The resonant-circuit current is

, = — sin a>t 0 ^ cot ^ n (A) (4)

IEE PROCEEDINGS, Vol. 131, Pt. B, No. 1, JANUARY 1984

Fig. 7 Collector circuit voltage and current waveforms

Turn-ona Collector voltage and current (7", + T2), 10 A/div.b Snubber capacitor resonant voltage and current Ilo,2 A/div.Turn-ofT:c Collector voltage and current (T, + T2), 5 A/div.: and ground current measure-ment noised Discharge voltage for Co and inductor lscurrent, 5 A/div.

where

CO =1 (\ + n

lrCn= COn n

(rad/sec)

lT 1 + n7 — -L. ' — 7 (fi)

The transistor current is Im + Ir.As it is required that the snubber capacitor must fully

discharge, from eqn. 2 we therefore require

When n < 1, eqn. 2 predicts that Cs will charge negatively,but within the practical circuit Cs is clamped to 0 Vbecause Ds will conduct, allowing the transfer of storedenergy in lT to Co.

That is, at cot = cos" l(n),

VCs = 0 (V)

Vc =nVm (V)Co m v /

1/

(A)

and the transistor current falls to lm.The resonant current through lT Co Ds is

1, = *—^ sin (coot + ct>) (A)

where

cj) = t a n J

and the transfer capacitor Co voltage is given by

(5)

VCo = ^ i Vm cos (coot (V) (6)

The final capacitor Co voltage is x / n Vm, while the peakresonant current is VJZ which is designed to be smallcompared with 7W, by making the total resonant timeequal to the minimum transistor on time ton(min). Thisproduces a high Q for the charge resonant circuit, which isa key factor for high-efficiency energy transfer.

5.7.2 Transistor turn-off: At transistor turn-off, thecapacitor Cs initially retains no charge, Co is charged tosfn Vm volts and the collector current is Im. A number ofphases and parallel processes occur as the transistor collec-tor current falls to zero.

Phase IThe collector current begins to fall and the excess loadcurrent is diverted to the snubber capacitor Cs via thediode Ds. The saturable inductor ls conducts the full loadcurrent Im. The transistor collector is clamped to thevoltage of the snubber capacitor which increases fromzero, according to

(7)

Phase I is that of conventional turn-off snubber action.

Phase IIWhen the snubber capacitor charges to Vm — VCg, that isKnO — \A)> t n e capacitor Co begins to discharge into theload via diode Dc. Because a portion of the load current Im

is provided from CQ, the current in ls is correspondinglyreduced to less than Im; hence the snubber capacitor Cs

charge rate is decreased.

IEE PROCEEDINGS, Vol. 131, Pt. B, No. 1, JANUARY 1984

If n « 1, that is CSK CO, then the inductor /s currentcan be approximated by

(8)

which indicates that the inductor current decreases at halfthe rate of the collector fall current. For small n, both cur-rents fall at the same rate.

The snubber capacitor voltage can be approximated by

4t/»C.n* 1 (9)

which is half the rate of that of the conventional snubber(cf. eqn. 7).

Phase IIIIt is assumed that the collector current falls to zero beforethe snubber capacitor Cs charges to the supply voltage Vm.Snubber operation is inconsequentional to transistor lossesafter the collector current reaches zero. When the collectorcurrent reaches zero, the snubber capacitor Cs is chargedto Vm via the inductor /s. The snubber capacitor Cs isclamped to Vm by the diode Dr, which in conjunction withdiode Ds prevents the transistor collector voltage fromexceeding the supply voltage Vm.

Any energy stored in /s is transferred via the path /s Ds Dr

load. Any load-current deficit has been supplied by thecapacitor Co, which, when discharged, is clamped to 0 V asthe freewheeling diode Df provides a path for load current.

All components have now attained the original requiredinitial conditions ready for transistor turn-on.

6 Inverter bridge with switching-aidcircuit and energy recovery

High-voltage power switching transistors are findingincreasing use in bridge-type applications, for high-powerstepper motor drives, AC machine control, RF inductionheating etc. If the transistors in the same bridge leg alter-nate from on to off with minimal dead time and the load isalways inductive, then simple lossless turn-off snubberingis possible [8, 9]. However, generally these two operationalrequirements preclude the use of a simple lossless turn-offsnubbers on most bridge circuits where a leading powerfactor may occur and versatile control, such as pulse widthmodulation, is required. Also, severe transistor stressingmay occur during snubber discharge at turn-on.

• Vm

load

Fig. 8 Proposed snubber energy recovery circuit adapted for one leg ofa bridge

The snubber-energy-recovery scheme shown in Fig. 6for a DC chopper is adaptable to the leg of a bridge asshown in Fig. 8. Basically two recovery schemes areemployed and each works independently of the other. Norestrictions are placed on the load power factor or theform of control employed. The diodes DbXa and Dblb

prevent undesired charge transfer from Cs to Co when theassociated freewheeling diode conducts [1, 2]. The diodesalso prevent reverse transistor current flow [5]. Ultrafastrecovery diodes for Dbl are not necessary, and voltage andcurrent ratings are Vm and lm, respectively.

The turn-on switching-aid inductance /s not only con-trols di/dt at turn-on, but also reduces current surges dueto bridge dv/dt effects [5].

7 ConclusionThis paper has considered the features and use of the high-voltage power-switching transistor in chopper and inverterapplications. One transistor application area not con-sidered is transformer load switching. Such applicationsusually involve collector voltages in excess of the supplyvoltage.

Transistor electrical switching characteristics have beenconsidered and base-drive requirements outlined. A high-performance practical base-drive circuit has been given,which incorporates all the features deemed desirable tomake the power transistor reliable and robust. This base-drive circuit has also been used in a 50 kW, 600 V tran-sistor inverter.

Turn-on and turn-off switching protection have beentreated and a simple, efficient and passive technique forrecovering protection circuit energy has been outlined.Over 95% energy-recovery efficiency is achievable. Thisrecovery technique is applicable to both transistorchoppers and inverters and may be used with thyristors orany other controlled switching element where switching-aid circuits are normally used.

The proposed energy-recovery scheme is directly applic-able to gate turn-off (GTO) thyristor applications. TheGTO usable frequency and power range can be signifi-cantly extended as it is currently limited by impracticallylarge snubber losses.

The proposed base and collector circuit techniques andcircuitry have been employed on a Darlington transistorswitching module and operated at rated Vceo{sus) = 600 Vand ICmax = 100 A at 100 kHz into an inductive load.

8 AcknowledgmentThe authors wish to thank both MEDL and Westcode formaking available experimental devices and data.

9 References

1 BOEHRINGER, A., and KNOLL, H.: Transistorschalter im Bereichhoher Leistungen und Frequenzen', ETZ Arch. 100, 1979, pp. 664-670

2 BOEHRINGER, A., el al.\ Transformatorlose Transistor-Pulsumrichter mit Augangsleistungen bis 50 kVA\ E. und M., 96, 1979,pp. 538-545

3 HSU, P.S.: 'A simple "lossless" turn-on plus turn-off snubber'. PCI '82Motorcon. '82, San Francisco, 29 March 1982

4 SLICKER, J.M.: 'A PWM inverter for an AC electric vehicle drive'.IEEE IAS Conference Record, 1981, pp. 292-299

5 PETER, J.M. el al.\ 'The power transistor in its environment'.Thomson-CSF, Semiconductor division, SESCOSEM, 1978

6 KNOLL, H.: 'High current transistor choppers'. Proc. IFAC, 1977, pp.307-315

7 'Power transistors'. Thomson-CSF, Semiconductor division, SESCO-SEM, 1981

8 EVANS, P.D.: 'Some aspects of power transistor inverter design', Elec-tron. Power Appl., 1979, 2, pp. 73-80

9 TRW Power Semiconductors, Data Book 08B, 1980

12 IEE PROCEEDINGS, Vol. 131, Pt. B, No. 1, JANUARY 1984