antenna at 2.4 GHz and this frequency the loop is caused an

omni directional pattern at antenna. But at higher frequency, the

inductance effect of the CRLH cells are reduced so the maxi-

mum current distribution are concentrated CRLH capacitances.

Therefore, in this condition we cannot see that effective current

loop, so directive pattern is predictable and according to Figure

6(g) the antenna pattern shows directive characteristic.

4. CONCLUSION

This article shows the effect of CRLH cells on designing dual

band antenna base on Zeroth-order resonance. Here, two differ-

ent form of CRLH antenna with corner ground is presented with

two different radiation pattern the first antenna has omnidirec-

tional radiation and second antenna shows omnidirectional pat-

tern at lower frequency and semidirectional pattern for higher

frequency with more bandwidth. First antenna, has two resonan-

ces at 1.85 and 5. 5 GHz for GSM 1800 and 5.2 GHz WLAN

indoor application. In the second antenna, first band support

2.38–2.48 GHz that covers the WLAN. The second operation

band is between 4.18 and 9.85 GHz for covering WLAN. The

current distribution shows that how these CRLH cells are helped

for change in pattern of antenna and frequencies effects on

inductance and capacitance of the CRLH shows in current dis-

pense. As shows in this article, the CRLH cells are shaped the

current as a loop in antenna surface and this current loop is

affected on antenna radiation pattern and making a omnidirec-

tional pattern. Most of the simulation result are compared with

measured experimental result for check the truth of result.

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VC 2015 Wiley Periodicals, Inc.

ULTRACOMPACT Si SLOT WAVEGUIDE-BASED POLARIZATION ROTATORS

Thang Q. Tran and Sangin KimDepartment of Electrical and Computer Engineering, Ajou University,Suwon 443-749, Gyeonggi Province, Korea; Corresponding author:sangin@ajou.ac.kr

Received 23 August 2014

ABSTRACT: A novel silicon asymmetric tilted slot waveguide-basedpolarization rotator is presented. A large birefringence in the silicon

slot waveguide facilitated a compact polarization rotator of

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 57, No. 4, April 2015 779

subwavelength scale. A 1287 nm long TM to TE and TE to TM polariza-

tion rotator was achieved with an extinction ratio of 10 dB and aninsertion loss of less than 5 dB at an operating wavelength of 1500 nm.Further performance improvement is possible for polarization rotator

designed to work on TM to TE (or TE to TM) conversion: a 711 nmlong TM to TE polarization rotator with 9.25 dB extinction ratio and2.2dB insertion loss was also numerically demonstrated. VC 2015 Wiley

Periodicals, Inc. Microwave Opt Technol Lett 57:779–785, 2015; View

this article online at wileyonlinelibrary.com. DOI 10.1002/mop.28953

Key words: polarization rotator; waveguide optics; silicon on insulator;slot waveguide

1. INTRODUCTION

The design of ultracompact polarization rotator is necessary for

highly integrated optical design. Nano-sized optical device

designs often require a unique polarization to function correctly.

The solution for such problem is polarization diversity, which

requires the design of an efficient and compact polarization rota-

tor for future integrated photonics. The design of an efficient

polarization rotator with low insertion loss, high extinction ratio,

and short operation length is still a difficult task.

It is acknowledged that the most difficult issue in

birefringence-based directly coupled polarization rotator is the

matching of the coupling coefficients [1]. In direct coupling Si-

based polarization rotating device, the device length is so small

that all propagation losses are negligible, and the single most

important loss is the coupling loss, resulting from the mode mis-

match between the input and output waveguide and the device.

These coupling losses are often high, and tuning such losses to

achieve good performance is often difficult. For such reason, the

majority of recent advances in polarization rotator are modal

evolution-based devices [1–3]. However, directly coupled devi-

ces, due to the lack of taper, can be made much shorter.

Besides, in the directly coupled devices, the propagating modes

can be made to be drastically different, so that a high birefrin-

gence, and thus, a further reduced device length are obtained.

There are recent advances in hybrid plasmon-based polarization

rotators [4–8], however, the device that offers an ultrashort

device length at a reasonably low insertion loss is still out of

reach. The best device so far achieved rotation at a device

length of 3.2 lm and an insertion loss of �1.38 dB [8]. In [5], a

TM to TE polarization rotator was designed at 3 lm long and

11 dB extinction ratio, but with very high insertion loss of 10

dB. In [6], loss compensator is required, in [9], nonlinear mate-

rial is needed, and the device length is relatively long (�6 lm).

In [10], very short conversion length was achieved. However,

due to very large coupling waveguide width of approximately 5

lm, single-mode operation of the output waveguide cannot be

guaranteed, which can be an obstacle to integration with other

devices and require an additional taper structure. In [11], a 3

lm device and a 4.8 lm tapered design were realized, however,

subwavelength conversion length was not obtained. Similarly, a

hybrid plasmonic waveguides design was presented in [8],

achieved a device length of 3.2 lm and an insertion loss of

�1.38 dB. However, as far as we know, no work in literature

has achieved yet a sub wavelength scale device length.

In this work, we propose a polarization rotator based on a

silicon-air slot waveguide with single-mode input/output wave-

guides. The silicon-air slot waveguide shows a giant optical

birefringence between the slot mode and the dielectric mode

mainly due to the large different of permittivity between Si and

air. A directly coupled polarization rotator design based on a

slanted dielectric slot waveguide, to our best knowledge, has not

been carried out. The design based on traditional silicon tech-

nology further adds to the attractiveness of the device. We have

shown that a 1287 nm long TM to TE and TE to TM simultane-

ous polarization rotator and a 711 nm long polarization rotator

optimized for TM to TE operation can be realized at

k 5 1500 nm.

2. OPERATION PRINCIPLE AND DESIGN CONSTRAINTS

For a birefringence-based polarization rotator with index differ-

ence of Dn, the optimal device length will be L5k=ð2DnÞ,where k is an operating wavelength. Generally, the larger the

index difference Dn is, the shorter the device length, thus more

desirable. Slot waveguide-based polarization rotators are espe-

cially promising as a slot mode exists only for a certain polar-

ization with electric field normal to the Si-air interface, which is

usually referred to as TM polarization, and has much lower

effective index than dielectric modes.

Figure 1 shows the proposed structure, where the polarization

rotating part consists of a slant slot waveguide on SiO2 and two

identical single-mode waveguides are adopted as input and out-

put waveguides. In this work, the single-mode waveguide is

designed to support only the lowest order mode for each polar-

ization, so that no additional taper structure is needed for inte-

gration with other waveguide-based devices. In the slant slot

Figure 1 (a) 3D structure of the polarization rotating device. The device consists of a inclined Si slot waveguide, represented in yellow, with an input

and an output Si waveguide, represented in green color. The substrate, represented in dark green, is SiO2. (b) Five design parameters of the polarization

rotating device: the height of the device (H), width of the foundation of the device (D), the slot width (DS), the rotating angle (h), and the width of the

input and output rib waveguide (not shown, identical by design). The height of the input and output waveguide was selected to be identical to the device

height H for ease of fabrication purposes. Both figures are to scale to the optimized version of the device, with H 5 360 nm, D 5 865 nm, DS 5 180 nm,

h 5 56.24�, and the input waveguide width is 400 nm. The device’s length is L 5 1287 nm. [Color figure can be viewed in the online issue, which is

available at wileyonlinelibrary.com]

780 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 57, No. 4, April 2015 DOI 10.1002/mop

waveguide, it is desired that only two guided modes exist: one

is a slot mode of a low effective index with electric field normal

to the Si-air interface and the other is a dielectric mode of a

high effective index with electric field parallel to the Si-air

interface.

When the slant slot waveguide supports only two modes

referred to as low- and high-index modes (LIM and HIM), an

operation principle of the polarization rotator is described in

Figure 2. At the interface between the input/output waveguide

and the slant slot waveguide, light from input waveguide of TE

or TM polarization is coupled to both HIM and LIM of the slant

slot waveguide. Those two modes experience different phase

retardations and each of those two modes is coupled to both TE

and TM polarization modes of the output waveguide.

Before discussing the mode coupling conditions for the

polarization rotation operation, let us define the nomenclature

of the coupling coefficients. We call Cjk the input coupling

coefficients in which j can be either E or M which corresponds

to a coupling with TE and TM mode, respectively, and k can

be either H or L which corresponds to a coupling to the HIM

or the LIM, respectively; and we call Ckj the output coupling

coefficients with k and j holding the same meaning. The cou-

pling coefficient can be calculated based on the overlap

integral:

Figure 2 Mechanism of an ideal direct coupling-based polarization rotator. Each arrow corresponds to a coupling direction. Light from input wave-

guide of TE or TM polarization is coupled to both the HIM and the LIM of the polarization rotator. Each of those two modes is coupled to both the TE

and the TM modes of the output waveguide. If complete destructive interference occurs at one of the polarization mode of the output waveguide, the

polarization rotator can be realized

Figure 3 Four propagating modes of the polarization rotator. (a) LIM or slot mode, (b) Dielectric or HIM, (c) and (d) antisymmetric modes. Due to

symmetric device design, antisymmetric modes will have very small coupling coefficients to the waveguide modes, and thus, negligible interference to

the working of the device. Depends on the design parameters, however, other undesired asymmetrical modes may also be supported by the device which

may have significant negative influences on its performance. For the designed waveguide, however, these are the only modes supported. The optimal

device parameters are H 5 360 nm, D 5 865 nm, DS 5 180 nm, h 5 56.24�, and the input waveguide width is 400 nm. [Color figure can be viewed in

the online issue, which is available at wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 57, No. 4, April 2015 781

C5jÐ Ð

E1�E2dAj2Ð Ð

jE1j2dAÐ ÐjE2j2dA

; (1)

where E1 and E2 are vector fields of the corresponding modes

in the input waveguide and the slant slot waveguide, respec-

tively. As the input and output waveguides are identical, which

has an added benefit of simplifying design, Cjk 5 Ckj is

satisfied.

If TE polarized light is assumed to be launched through the

input waveguide, there are four optical paths which can be

taken by light: TE-HIM-TE, TE-LIM-TE, TE-HIM-TM, and

TE-LIM-TM. Due to the different in effective indices between

the HIM and the LIM, there will be phase differences among

the optical paths. If the slant slot waveguide length is deter-

mined such that the phase retardation difference between the

HIM and the LIM is p, the TE-HIM-TE and the TE-LIM-TE

paths will be completely out of phase, and the TE-HIM-TM

and the TE-LIM-TM will be completely in-phase. To have a

complete destructive interference between TE-HIM-TE and

TE-LIM-TE at the output waveguide, they need to have

the same amplitude, that is, CEH*CHE 5 CEL*CLE. As

CEH 5 CHE and CEL 5 CLE, the additional condition for TE-to-

TM polarization rotation becomes CEH 5 CEL. The output

amplitude of TM polarization in such case would be

(CEH*CHM 1 CEL*CLM). Similar reasoning can be said for a

TM polarized input, which results in an amplitude matching

condition of CMH 5 CML and an rotated output amplitude of

(CMH*CHE 1 CML*CLE).

An important consequence of the foregoing is that the TE to

TM conversion condition (CMH 5 CML) and the TM to TE con-

version condition (CEH 5 CEL) can be achieved independently.

This implies that the interference-based device does not neces-

sary offer simultaneous TE to TM and TM to TE rotations. By

satisfying only one of the polarization rotation conditions, one

can potentially realize a polarization rotator with better perform-

ance in some specific applications where a simultaneous polar-

ization rotator device is not required.

3. DEVICE DESIGN AND ANALYSIS

Designing the device involves optimizing five parameters

including those four geometrical parameters presented in Figure

1(b) and the width of the input/output waveguide. The guided

modes in the slant slot waveguide and the input/output wave-

guide are calculated using the finite element method, and the

corresponding coupling coefficients are calculated based on the

field profiles of the modes. From the calculated effective index

difference between the HIM and the LIM, the device length can

be estimated. By repeating this process with those five geometri-

cal parameters varied, we should find an optimum set of the

parameters satisfying the polarization conversion condition. This

optimization process is extremely time-consuming. In this work,

we focused on minimization of the device length with an

acceptable insertion loss in the parameter optimization process.

Figure 4 FEM calculated coupling coefficients CEL, CEH, CML, and CMH, versus different input waveguide width, with different slant angle h. The

rest of the parameters, which are the same for the four cases, are H 5 865 nm, D 5 360 nm, DS 5 180 nm. [Color figure can be viewed in the online

issue, which is available at wileyonlinelibrary.com]

782 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 57, No. 4, April 2015 DOI 10.1002/mop

There appeared to be a tradeoff between the insertion loss of

the device and the device length. A larger slot size will increase

its modal area, thus increase its coupling coefficient and reduce

the insertion loss. However, the larger slot size will reduce the

effective index of the dielectric mode and thus, Dn, which con-

sequently results in an increase of the device length. In the

design of the slant slot waveguide, it should also be considered

to prevent the appearance of higher-order asymmetrical modes.

Due to the sharp edges and intrinsic asymmetrical properties of

the polarization rotator, multiple nondesired propagating modes

may exist within the polarization rotator. An unintended cou-

pling of the input light to the undesired higher-order modes will

have a detrimental effect to the performance reducing the

extinction rate of the polarization rotator. To suppress those

higher-order modes, the slant slot waveguide should be made

small enough. However, if the device is small enough to prevent

the unwanted modes, the existence of the HIM and the LIM is

not guaranteed. Moreover, reducing the transverse dimension of

the device will reduce the coupling efficiency to the input/output

waveguides. Therefore, the transverse dimension of the device is

designed so as to allow up to the second order antisymmetric

modes. If the single-mode input/output waveguide is adopted,

the excitation of the antisymmetric modes will be negligible

bringing about an order of magnitude lower coupling coeffi-

cients compared to the HIM and the LIM, thus, will not seri-

ously interfere with the operation of the polarization rotator.

From our investigation, it has been found that H> 200 nm is

required to support the LIM and to keep the excitation of the

antisymmetric modes low enough H< 420 nm is needed. Figure

3 represent all the guided modes of the slant slot waveguide of

H 5 360 nm, D 5 865 nm, Ds 5 180 nm, and h 5 56.24�, which

is close to the optimal structure. The HIM have the effective

index nHi 5 2.4797, and the LIM have the effective index

nLo 5 1.8966, thus, the index different Dn is 0.5831, which cor-

responds to a device length of � 1287 nm at k 5 1500 nm. Fig-

ures 3(a) and 3(b) show the HIM and the LIM, respectively.

Their related antisymmetric modes are represented in Figures

3(c) and 3(d), respectively.

From our extensive investigation, it has been found that the

slant angle, h does not seriously affect the existence of the

guided modes and Dn of the slant slot waveguide and instead,

dramatically change the coupling coefficients to the input/output

waveguide. So, we could achieve the satisfactory optimization

run by scanning of the input waveguide width with different

angle h while keeping the rest of the geometrical parameters

constant, which is represented in Figure 4. This is just one of

many scans performed; extensive runs at different H, D, and Ds

were also performed, but due to the limitation of the pages they

are not presented. To maximize the coupling efficiency and

make the device fabrication simple, the height of the input/out-

put waveguide was chosen to be the same as D. For

D 5 360 nm, single-mode operation of the input/output

Figure 6 FDTD simulation of the simultaneous polarization rotator design for (a) TE to TM conversion and (b) TM to TE conversion. Simulation was

done in complex, and color represent the field modulus to filter out the effect of phase. The device parameters are H 5 360 nm, D 5 865 nm,

DS 5 180 nm, h 5 56.24�, and the input waveguide width is 400 nm. The device’s length is L 5 1287 nm. The extinction ratio for TE to TM is 10.48 dB

with an insertion loss of 4.59 dB. For TM to TE rotation, the extinction ratio is 5.82 dB and the insertion loss is 4.41 dB. [Color figure can be viewed

in the online issue, which is available at wileyonlinelibrary.com]

Figure 5 Two propagating modes of the input and output waveguides, (a) TE mode and (b) TM mode. Input waveguide width is 400 nm. [Color fig-

ure can be viewed in the online issue, which is available at wileyonlinelibrary.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 57, No. 4, April 2015 783

waveguide was achieved at the width of less than � 680 nm at

k 5 1500 nm. As such, the width of the input/output waveguide

was scanned in the range from 300 to 600 nm. As one can see

in Figure 4, the TE-to-TM and TM-to-TE polarization conver-

sion condition (CMH 5 CML and CEH 5 CEL) can be achieved

for h 5 58.5� over a wide range of the input/output waveguide

width.

After rough geometrical parameter optimization based on the

mode analysis and the coupling coefficient calculation, 3D finite

different time domain (FDTD) simulation was done on the

designed device, and some fine-tuning of the parameters was per-

formed. The fine-tuning is necessary due to internal reflection

which was not taken into account in the foregoing optimization

and the existence of multiple undesired leaky modes in the oper-

ating devices which may interfere with the device operation due

to its short length of subwavelength scale. For the FDTD simula-

tion, the MIT FDTD package MEEP [11] was used. The FDTD

was done with complex field to control phase information, and

the field modulus was plotted. The resulting optimized polariza-

tion rotator consists of the slant slot waveguide of H 5 360 nm,

D 5 865 nm, Ds 5 180 nm, h 5 56.24�. As for the input/output

waveguide width, the larger, the better performance is achieved

as far as the excitation of only the fundamental TE or TM mode

is provided. From our calculation, the optimum width appeared

to be 1200 nm. However, due to the need to ensure single-mode

propagation of the input/output coupling waveguide, a 400 nm

wide waveguide was considered at a significant performance pen-

alty. Figure 5 shows the TE and the TM mode field profiles in

the input/output waveguide width of 400 nm. The length of the

device was optimized for maximum polarization conversion,

which is L 5 1287 nm. Figure 6 shows the FDTD simulation

result for the designed polarization rotator. In Figure 6(a), TE-to-

TM polarization conversion is clearly demonstrated. The polariza-

tion extinction ratio at the output is 10.48 dB and the insertion

loss is 4.59 dB. Whereas, for TM to TE rotation in Figure 6(b),

the performance is relatively poor, and the extinction ratio of

5.82 dB and the insertion loss of 4.41 dB were obtained. This

rather poor performance stems from the lower coupling coeffi-

cient of CML and CMH than CEL and CEH as seen in Figure 4(c).

As previously mentioned, if one is interested in only a TM

polarization rotator, much better performance can be obtained.

In this work, the optimized TE to TM polarization rotator was

also designed by the same design process as described earlier.

The designed device has geometrical parameters of H 5 400 nm,

D 5 500 nm, Ds 5 50 nm, h 5 36�, and L 5 711 nm, and the

input/output waveguide width is 500 nm. Figure 7 shows the

FDTD simulation for the designed TM to TE polarization rota-

tor, where the polarization extinction ratio was of 9.25 dB and

the insertion of 2.19 dB are achieved.

4. CONCLUSION

The design of polarization rotator is critical for optical commu-

nication applications. For ultrahigh density integrated optical

devices, minimizing the size of the optical components and the

polarization rotator in particular are very important. We have

demonstrated an ultracompact TM to TE polarization rotator of

a submicron based on direct coupling of slot waveguide with

low insertion loss. It is clear that these devices can be further

optimized by performing an algorithm such as particle swarm

optimization in conjunction with FDTD simulation given enough

computational resources. For critical applications where space is

a concern, and polarization rotators capable of working with

both polarization is not required [1, 5], the proposed TM to TE

polarization rotator device is well suited.

The fabrication of the designed devices would not be a prob-

lem as a Si reactive ion etching process with an arbitrary slant

angle has been well established [12].

ACKNOWLEDGMENT

This work was supported by National Research Foundation of

Korea Grant (NRF-2014R1A2A2A01006720, NRF-2008-

0061906 and NRF-2009-0094046).

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VC 2015 Wiley Periodicals, Inc.

A 60 GHz LOW POWER DIRECT-CONVERSION QUADRATURE-PHASETRANSMITTER IN 130 nm CMOS WITHLOW PROFILE CAVITY BACKEDANTENNA

Bo Chen,1,2 Guodong Su,3 Li Shen,1 and Jianjun Gao1

1 School of Information Science and Technology, East China NormalUniversity, Shanghai 200062, People’s Republic of China;Corresponding author: jjgao@ee.ecnu.edu.cn2 School of Electrical and Electronic Engineering, NanYangTechnological University, Singapore 6397983 School of Electrical Engineering, Zhejiang University,Zhejiang 310007, People’s Republic of China

Received 24 August 2014

ABSTRACT: This article presents the analysis, design, and implemen-tation of a 60 GHz transmitter in 130 nm bulk CMOS, focusing on lowpower and high performance applications. The low power transmitter

lineup consists of two double-balanced Gilbert-cells with dynamic cur-rent injection, on chip FGCPW meandering ring-hybrid balun, Lange

coupler, combiner, preamplifier, and low power transformer coupledpower amplifier. A planar antenna with above 6 dB gain is also imple-mented. The measured output 21 dB compression point of the power

amplifier is 9.6 dBm with 8.6 dB gain, and the 3 dB bandwidth is from47 to 67 GHz (power amplifier) while drawing 90 mA from a 1.2 V sup-ply. The mixer, preamplifer, and drive amplifier consume 12, 26, and 68

mA, respectively. The transmitter up-converting an IF of 200 MHz to RFat 60.2 GHz achieve a total gain of 17.17 dB (simulated 21.09 dB) and

saturation power higher than 9.53 dBm. VC 2015 Wiley Periodicals, Inc.

Microwave Opt Technol Lett 57:785–789, 2015; View this article online

at wileyonlinelibrary.com. DOI 10.1002/mop.28962

Key words: CMOS; transmitter; power amplifier; millimeter wave;antenna

1. INTRODUCTION

Scaling in CMOS technology has permitted the integration of

mm-wave systems with high-speed data communications of gig-

abit per second (Gb/s) opening in the 57–66 GHz band [1]. In

the past few years, implementations of CMOS building blocks

such as low-noise amplifiers, mixers, and power amplifiers have

been designed to operate in the spectrum of 60-GHz band [2].

The Federal Communication Commission has allocated several

frequency bands at millimeter waves for high data rate wireless

communication. In 60 GHz band, IEEE 802.15.3c-2009 standard

are given approval recently. The European Computer Manufac-

turers Association International (ECMA) outlines similar

requirements in the ECMA-387 standard [3]. Also, the Wire-

lessHD consortium specification (version 1.0a [4]) for the trans-

mission of high-definition video resides in this unlicensed band.

The IEEE 802.11ad standard defines four 2.16-GHz-bandwidth

channels around the 60-GHz frequency [5].

This article describes the design and implementation of direct

conversion millimeter-wave CMOS transmitter in 130 nm

CMOS. In this article, Section 2 details the circuits’ implemen-

tation of various blocks. The measurement results are summar-

ized in Section 3. Finally, conclusions are drawn in Section 4.

2. CIRCUITS DESIGN AND IMPLEMENTATION

This section describes the circuits used to implement the direct

up-conversion transmitter in 130-nm bulk CMOS. All building

blocks use differential signal except the preamplifier. The

direct-conversion zero-IF architecture of this work is shown in

Figure 1. Figure 2(a) illustrates the schematic of the dynamic

current injection Gilbert up conversion mixer. The channel

length of M3 and M4 are chosen to be 0.12 lm to increase the

transconductance of the IF. The switching transistors are opti-

mized to have a small overdrive voltage thus a relatively low

LO swing is required. TL1 to TL4 are four transmission lines

which form the output matching network. Cross-coupled PMOS

pair (M9�M10) forms the current bleeding in the Gilbert mixer

core. The transistors of M9 and M10 must be large enough to

inject a sufficient current and the PMOS cross-coupled transis-

tors M9/M10 can provide an equivalent negative input transcon-

ductance 2gm8,9 at the source terminals of the LO switching

transistors M5�M8. This in turn results in a conversion gain

enhancement due to negative resistance compensation [6].

The preamp is implemented as a five-stage cascode distribute

amplifier. The preamplifier has the same structure which is

shown in Figure 2(b). The interstage transmission line helps dis-

tribute the parasitic capacitance and increasing the bandwidth of

the amplifier [7]. The input and cascode transistors are 30-lm

wide and are laid out as 12-finger devices. A three-stage trans-

former biased drive amplifier is placed before the mixer to

improve the power of LO. The transistor widths of drive ampli-

fier are 40 lm while the length is the 0.12 lm to get enough

gain.

The power amplifier (PA) is implemented as a three-stage dif-

ferential common source transformer-coupled amplifier as shown

in Figure 2(c). Differential amplifier reduces the amplifier’s sensi-

tivity to the surrounding ground plane and reduces the need for

by decoupling capacitors. Single turn overlay transformers are

used in this design for input, inter- and output stage matching

networks as well as for power combining in output stage. The

pseudodifferential neutralized technique is adopted here to offer

more stability than normal common source amplifier [8].

Figure 3(a) shows the 60 GHz Lange coupler which is used

to generate I, Q signal, miniaturization is always a desirable

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 57, No. 4, April 2015 785