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Design With PIN DiodesThe PIN diode finds wide usage in RF, UHF and microwave circuits. It is fundamentally adevice whose impedance, at these frequencies, is controlled by its DC excitation. A unique feature of the PIN diode is its ability to control large amounts of RF power with much lowerlevels of DC.

By Gerald HillerSenior Scientist

Alpha Industries

PIN Diode FundamentalsThe PIN diode is a current controlled resistor at radio

and microwave frequencies. It is a silicon semiconductordiode in which a high resistivity intrinsic I-region issandwiched between a P-type and N-type region. Whenthe PIN diode is forward biased, holes and electrons areinjected into the I-region. These charges do no immedi-ately annihilate each other; Instead they stay alive for anaverage time called the carrier lifetime, t. This results inan average stored charge, Q, which lowers the effectiveresistance of the I-region to a value RS.

When the PIN diode is at zero or reverse bias there isno stored charge in the I-region and the diode appears asa capacitor, CP shunted by a parallel resistance RP.

PIN diodes are specified for the following parameters:

RS series resistance under forward bias

CT total capacitance at zero or reverse bias

RD parallel resistance at zero or reverse bias

VR maximum allowable DC reverse bias voltage

t carrier lifetime

q AVE average thermal resistance or

PD maximum average power dissipation

q pulse pulse thermal impedance or

PP maximum peak power dissipation

By varying the I-region width and diode area it is possible to construct PIN diodes of different geometrics to result in the same RS and CT characteristic. Thesedevices may have similar small signal characteristics.However, the thicker I-region diode would have a higherbulk or RF breakdown voltage and better distortionproperties. On the other hand the thinner device wouldhave faster switching speed.

There is a common misconception that carrier lifetime, t, is the only parameter that determines the lowestfrequency of operation and the distortion produced. This is indeed a factor, but equally important is the thickness of the I-region, W, which relates to the transittime frequency of the PIN diode.

Area A

Figure 1.

2 Alpha Industries, Inc. [781] 935-5150 • Fax [617] 824-4579 • Email [email protected] • www.alphaind.com

Design With PIN DiodesDesign With PIN Diodes

1ƒ >2π r (where r is the resistivity of the I-region).

At lower frequencies the PIN diode acts like a varactor.

Low Frequency ModelAt low frequencies (below the transit time frequency of

the I-region) and DC the PIN diode behaves like a siliconPN junction semiconductor diode. Its I-V characteristicdetermines the DC voltage at the forward bias currentlevel. PIN diodes often are rated for the forward voltage,VF, at a fixed DC bias.

The reverse voltage ratings on a PIN diode, VR, are aguarantee from the manufacturer that no more than aspecified amount, generally 10µA, of reverse current willflow when VR is applied. It is not necessarily theavalanche or bulk breakdown voltage, VB, which is deter-mined by the I-region width (approximately 10V/ µm.)PIN diodes of the same bulk break down voltage mayhave different voltage ratings. Generally, the lower thevoltage rating the less expensive the PIN diode.

Large Signal ModelWhen the PIN diode is forward biased the stored

charge, Q, must be much greater than the incrementalstored charge added or removed by the RF current, IRF. Toinsure this the following inequality must hold:

Q > > IRF

2 πf

RF Electrical Modeling of PIN Diode

Figure 2


Design With PIN Diodes

Under reverse bias the diode should not be biasedbeyond its DC voltage rating, VR. The avalanche or bulkbreakdown voltage, VB, of a PIN diode is proportional tothe I-region wide, W, and is always higher than VR. In atypical application maximum negative voltage swingshould never exceed VB. An instantaneous excursion of theRF signal into the positive bias direction generally does notcause the diode to go into conduction because of slowreverse to forward switching speed. The DC reverse biasneeded to maintain low PIN diode conductance has beenanalysed6 and is related to the magnitude of the RF signaland I-region width.

Switching Speed ModelThe switching speed in any application depends on

the driver circuit as well as the PIN diode. The primaryPIN properties that influence switching speed may beexplained as follows:

A PIN diode has two switching speeds from forwardbias to reverse bias TFR, and from reverse bias to forwardbias TRF. The diode characteristic that affects TFR is t, carrier lifetime. The value of TFR may be computed fromthe forward current, IF, and the initial reverse current IR,as follows:

IF = loge (1 + IF ) (Secs)

IR

TRF depends primarily on I-region width, W, as indi-cated in the following chart which shows typical data:

Thermal ModelThe maximum allowable power dissipation, PD, is

determined by the following equation:

TJ - TAPD = Watts

qwhere TJ is the maximum allowable junction tempera-ture (usually 175ºC) and TA is the ambient or heat sinktemperature. Power dissipation may be computed as theproduct of the RF current squared multiplied by thediode resistance, RS.

For CW applications the value of thermal resistance,q, used is the average thermal resistance, qAV.

In most pulsed RF and microwave applications wherethe duty factor, DF, is less than 10 percent and the pulsewidth, TP, is less than the thermal time constant of thediode, good approximation of the effective value of q inthe above equation may be computed as follows:

q = DF x qAV + qTP °C/W

Where qTP is the thermal impedance of the diode forthe time interval corresponding to TP.

The following diagram indicates how junction temperature is affected during a pulsed RF application.

Figure 3

Figure 4

I-Width To 10mA from To 50mA from To 100mA fromµm 10V 100V 10V 100V 10V 100V

175 7.0µS 5.0µS 3.0µS 2.5µS 2.0µS 1.5µS100 2.5µs 2.0µS 1.0µS 0.8µS 0.6µS 0.6µS50 0.5µS 0.4µS 0.3µS 0.2µS 0.2µS 0.1µS



PIN Diode ApplicationsSwitches

PIN diodes are commonly used as a switching ele-ment to control RF signals. In these applications, the PINdiode can be biased to either a high or low impedancedevice state, depending on the level of stored charge inthe I-region.

A simple untuned single-pole, single-throw (SPST)switch may be designed using either a single series orshunt connected PIN diode as shown in Figure 5. Theseries connected diode switch is commonly used whenminimum insertion loss is required over a broad fre-quency range. This design is also easier to physicallyrealize using printed circuit techniques, since no throughholes are required in the circuit board.

A single shunt mounted diode will, on the other handproduce higher isolation values across a wider frequencyrange and will result in a design capable of handlingmore power since it is easier to heat sink the diode.

Multi-throw switches are more frequently used thansingle-throw switches. A simple multi-throw switch maybe designed employing a series PIN diode in each armadjacent to the common port. Improved performance isobtained by using “compound switches,” which arecombinations of series and shunt connected PIN diodes,in each arm.

For narrow-band applications, quarter-wave spacedmultiple diodes may also be used in various switchdesigns to obtain improved operation in the followingsection, we shall discuss each of these types of switchesin detail and present design information for selectingPIN diodes and predicting circuit performance.

Figure 5 Figure 6



Series Connected SwitchFigure 6 shows two basic types of PIN diode series

switches, (SPST and SPDT), commonly used in broad-band designs. In both cases, the diode is in a “passpower” condition when it is forward biased and presentsa low forward resistance, RS, between the RF generatorand load. For the “stop power” condition, the diode is atzero or reverse bias so that it presents a high impedancebetween the source and load. In series connectedswitches, the maximum isolation obtainable dependsprimarily on the capacitance of the PIN diode, while theinsertion loss and power dissipation are functions of thediode resistance. The principal operating parameters of aseries switch may be obtained using the following equa-tions:

A. Insertion Loss (Series Switch)

IL = 20 log10 [1 + RS/2ZO] dB (1)

This equation applies for a SPST switch and is graph-ically presented in Figure 7 for a 50 ohm impedancedesign. For multi-throw switches, the insertion loss isslightly higher due to any mismatch caused by thecapacitance of the PIN diodes in the “off” arms. Thisadditional insertion loss can be determined from Figure10 after first computing the total shunt capacitance of all“off” arms of the multi-throw switch.

B. Isolation (Series Switch)

I = 10 log10 [1 + (4πfCZ0) -2] dB (2)

This equation applies for a SPST diode switch. Add 6dB for a SPNT switch to account for the 50 percent volt-age reduction across the “off” diode due to the termina-tion of the generator in its characteristic impedance.Figure 8 graphically presents isolation as a function ofcapacitance for simple series switches. These curves areplotted for circuits terminated in 50 ohm loads.

C. Power Dissipation (Series Switch in Forward Bias)

Where the maximum available power is given by:

It should be noted that Equations 3 and 4 apply onlyfor perfectly matched switches. For SWR (o) values otherthan unity, multiply these equations by [20/(o + 1)]2 toobtain the maximum required diode power dissipationrating.

D. Peak Current (Series Switch)

Figure 7 Insertion loss for PIN Diode series

switch in 50Ω system.

Figure 8 Isolation for SPST Diode series switch in

50Ω system. Add 6 dB to isolation for multi throw switches (SPNT).

2



1.0

Figure 9 Shunt Connected Switches 2-5

Figure 10 Insertion loss for shunt PIN switch in 50Ω system.

Shunt Connected SwitchFigure 9 shows two typical shunt connected PIN

diode switches. These shunt diode switches offer highisolation for many applications and since the diode maybe heat sinked at one electrode, it is capable of handlingmore RF power than a diode in a series type switch.

In shunt switch designs, the isolation and power dis-sipation are functions of the diode’s forward resistance,whereas the insertion loss is primarily dependent on thecapacitance of the PIN diode. The principal equationsdescribing the operating parameters of shunt switchesare given by:

A. Insertion Loss (Shunt Switch)

IL = 10 1og10 [1 + (πf CTZ0) 2] [dB] (10)

This equation applies for both SPST and SPNT shuntswitches and is graphically presented in Figure 10 for a50 ohm load impedance design.

B. Isolation (Shunt Switch)

This equation, which is illustrated in Figure 11,applies for a SPST shunt switch. Add 6 dB to these val-ues to obtain the correct isolation for a multi-throwswitch.



Figure 11 Isolation for SPST shunt PIN switches in

50Ω system. Add 6 dB to isolation for multi-throw switches (SPNT).

Figure 12 Compound Switches

Compound and Tuned SwitchesIn practice, it is usually difficult to achieve more than

40 dB isolation using a single PIN diode, either in shuntor series, at RF and higher frequencies. The causes of thislimitation are generally radiation effects in the transmis-sion medium and inadequate shielding. To overcomethis there are switch designs that employ combinationsof series and shunt diodes (compound switches) andswitches that employ resonant structures (tunedswitches) affecting improved isolation performance.

The two most common compound switch configura-tions are PIN diodes mounted in either ELL (series-shunt) or TEE designs as shown in Figure 12. In theinsertion loss state for a compound switch the seriesdiode is forward biased and the shunt diode is at zero orreverse bias. The reverse is true for the isolation state.Th/s adds some complexity to the bias circuitry in com-parison to simple switches. A summary of formulas usedfor calculating insertion loss and isolation for compoundand simple switches is given in Figure 13.

Figure 13 Summary of Formulas for SPST Switches.

(Add 6 dB to Isolation to obtain value for single-pole multiple throw switch.)



Figure 14 shows the performance of an ELL typeswitch a diode rated at 3.3pF, maximum capacitance,and 0.25W, RS maximum at 100 mA. In comparison, asimple series connected using the same diode switchwould have similar insertion loss to the 100 MHz contour and the isolation would be 15 dB maximum at100 MHz, falling off at the rate of 6 dB per octave.

A tuned switch may be constructed by spacing twoseries diodes or two shunt diodes a wavelength apart asshown in Figure 15. The resulting value of isolation in thetuned switch is twice that obtainable in a single diodeswitch. The insertion loss of the tuned series switch is high-er than that of the simple series switch and may be com-

Figure 14 Series Shunt Switch

Figure 15

Figure 16 Quarter Wave Line Equivalent



l /4

l /4

¥

ZO

2πof

12πfOZO

puted using the sum of the diode resistance as the RSvalue in equation 1. In the tuned shunt switch the inser-tion loss may even be lower than in a simple shunt switchbecause of a resonant effect of the spaced diode capaci-tances.

Quarter-wave spacing need not be limited to frequen-cies where the wavelength is short enough to install a dis-crete length of line. There is a lumped circuit equivalentwhich simulates the quarter-wave section and may beused in RF band. This is shown in Figure 16. These tunedcircuit techniques are effective in applications havingbandwidths on the order of 10 percent of the center fre-quency.

Transmit-Receive SwitchesThere is a class of switches used in transceiver appli-

cations whose function is to connect the antenna to thetransmitter (exciter) in the transmit state and to thereceiver during the receiver state. When PIN diodes areused as elements in these switches they offer higher reli-ability, better mechanical ruggedness and faster switch-ing speed than electro-mechanical designs.

The basics circuit for an electronic switch consists of a

PIN diode connected in series with the transmitter, and ashunt diode connected a quarter wavelength (l /4 sec-tion (Figure 16) and, of course, are preferable for trans-ceivers that operate at long wavelengths.

When switched into the transmit state each diodebecomes forward biased. The series diode appears as alow impedance to the signal heading toward the antennaand the shunt diode effectively shorts the receiver’santenna terminals to prevent overloading. Transmitterinsertion loss and receiver isolation depend on the dioderesistance. If RS is 1Ω greater than 30 dB isolation andless than 0.2 dB insertion, loss can be expected. This per-formance is achievable over a 10 percent bandwidth.

In the receive condition the diodes are at zero orreverse bias and present essentially a low capacitance, CT,which creates a direct low-insertion-loss path betweenthe antenna and receiver. The off-transmitter is isolatedform this path by the high impedance series diode.

The amount of power, PA, this switch can handledepends on the power rating of the PIN diode, PD, andthe diode resistance. The equation showing this relation-

Figure 18 Broadband Antenna Switch

Figure 17 Quarter Wave Antenna Switches



The Alpha SMC1842-201 is a surface mount PIN dioderated at 2.5W dissipation to a 2.5 contact. The resistanceof this diode is a .75W (max) at 50mA. A quarter-waveswitch using SMC1842-201 may then be computed tohandle 40 watts with a totally mismatched antenna.

It should be pointed out that the shunt diode of thequarter-wave antenna switch dissipates about as muchpower as the series diode. This may not be apparent fromFigure 17; however, it may be shown that the RF currentin both the series and shunt diode is practically identical.

Broadband antenna switches using PIN diodes may bedesigned using the series connected diode circuit shownin Figure 18. The frequency limitation of this switchresults primarily from the capacitance of D2.

In this case forward bias is applied either to D1 duringthe transmit of D2 during receive. In high power application(>50W) it is often necessary to apply reverse voltage on D2during transmit. This may be accomplished either by a

negative polarity power supply at Bias 2 or by having theforward bias current of D1 flow through resistor R toapply the required negative voltage.

The selection of diode D1 is based primarily on its powerhandling capability. It need not have a high voltage ratingsince it is always forward biased in its low resistance statewhen high RF power is applied Diode D2 does not pass high RF current but must be able to hold off the RF voltagegenerated by the transmitter. It is primarily selected onthe basis of its capacitance which determines the upper frequency limit and its ability to operate at low distortion.

Using Alpha SMC1842-201 as D1, and a SMP1302-001or SOT-23 PIN diode which are rated at 0.3pf max as D2,greater than 25 dB receiver isolation may be achieved upto 400 MHz. The expected transmit and receive insertionloss with the PIN diodes biased at 50 mA are 0.1 dB and 0.3dB respectively. This switch can handle RF power levelsup to 40 watts.

Practical Design HintsPIN diode circuit performance at RF frequencies is

predictable and should conform closely to the designequations. When a switch is not performing satisfacto-rily, the fault is often not due to the PIN diode but toother circuit limitations such as circuit loss, bias circuitinteraction or lead length problems (primarily whenshunt PIN diodes are employed).

It is good practice in a new design to first evaluate thecircuit loss by substituting alternatively a wire short oropen in place of the PIN diode. This will simulate the cir-cuit performance with “ideal PIN diodes.” Any defi-ciency in the external circuit may then be correctedbefore inserting the PIN diodes.

Figure 2 RF AGC/Leveler Circuit

Figure 1 Typical Diode Resistance vs

Foward Current



PIN Diode AttenuatorsIn an attenuator application the resistance characteris-

tic of the PIN diode is exploited not only at its extremehigh and low values as in switches but at the finite valuesin between.

The resistance characteristic of a PIN diode when forward biased to IF1depends on the l-region width (W)carrier lifetime (T), and the hole and electron mobilities µp µn as follows:

RS = W 2

/ (µp + µn) IFt. ohms (1)

For a PIN diode with an I-region width of typically250µm, carrier lifetime of 4µS, and µn of .13, µp of .05m2/v•s, Figure 1 shows the RS vs current characteristic.

In the selection of a PIN diode for an attenuator appli-cation the designer must often be concerned about therange of diode resistance which will define the dynamicrange of the attenuator. PIN diode attenuators tend to bemore distortion sensitive than switches since their operat-ing bias point often occurs at a low value of quiescentstored charge. A thin I-region PIN will operate at lower for-ward bias currents than thick PIN diodes but the thickerone will generate less distortion.

PIN diode attenuator circuits are used extensively inautomatic gain control (AGC) and RF leveling applica-tions as well as in electronically controlled attenuatorsand modulators. A typical configuration of an AGCapplication is shown in Figure 2. The PIN diode attenu-ator may take many forms ranging from a simple seriesor shunt mounted diode acting as a lossy reflectiveswitch or a more complex structure that maintains aconstant matched input impedance across the fulldynamic range of the attenuator

Although there are other methods for providingAGC functions such as varying the gain of the RF tran-sistor amplifier, the PIN diode approach generallyresults in lower power drain, less frequency pulling,and lower RF signal distortion The latter results areespecially true, when diodes with thick I-regions andlong carrier lifetimes are used in the attenuator circuits.Using these PIN diodes, one can achieve wide dynamicrange attenuation with low signal distortion at frequencies ranging from below 1 MHz up to well over1 GHz.

Reflective AttenuatorsAn attenuator may be designed using single series or

shunt connected PIN diode switch configurations asshown in Figure 3. These attenuator circuits utilize thecurrent controlled resistance characteristic of the PINdiode not only in its low loss states (very high or lowresistance) but also at in-between, finite resistance values.

The attenuation value obtained using these circuitsmay be computed from the following equations:

These equations assume the PIN diode to be purelyresistive. The reactance of the PIN diode capacitance,however, must also be taken into account at frequencieswhere its value begins to approach the PIN diode resis-tance value.

Figure 4 Quadrature Matched Hybrid Attenuator

(Series Connected Diodes)

Figure 3 SPST PIN Diode Switches



Matched AttenuatorsAttenuators built from switch design are basically

reflective devices which attenuate the signal by produc-ing a mismatch between the source and the load.Matched PIN diode attenuator designs, which exhibitconstant input impedance across the entire attenuationrange, are also available which use either multiple PINdiodes biased at different resistance points or bandwidth-limited circuits utilizing tuned elements. They aredescribed as follows:

Quadrature Hybrid AttenuatorsAlthough a matched PIN attenuator may be achieved

by combining a ferrite circulator with one of the previoussimple reflective devices, the more common approachmakes use of quadrature hybrid circuits. Quadraturehybrids are commonly available at frequencies frombelow 10 MHz to above 1 GHz, with bandwidth cover-age often exceeding a decade. Figures 4 and 5 show typi-

cal quadrature hybrid circuits employing series andshunt connected PIN diodes. The following equationssummarize this performance:

The quadrature hybrid design approach is superior tothe circulator coupled attenuator from the standpoint oflower cost and the achievement of lower frequency oper-ation. Because the incident power is divided into twopaths, the quadrature hybrid configuration is also capa-ble of handling twice the power and this occurs at the 6dB attenuation point. Each load resistor, however, mustbe capable of dissipating one half the total input power atthe time of maximum attenuation.

Both the above types of hybrid attenuators offer gooddynamic range. The series connected diode configurationis, however, recommended for attenuators used primar-ily at high attenuation levels (greater than 6 dB) while theshunt mounted diode configuration is better suited forlow attenuation ranges.

Figure 5 Quadrature Hybrid Matched Attenuator

(Shunt Connected PIN Diodes) Constant impedance attenuator circuit. The power incident on port A divides equally between ports B and C, port D is isolated. The mismatch produced by the PIN Diode resistance in parallel with the load resistance at ports B & C reflects part of the power. The reflected power

exits port D isolating port A. Therefore, A, appears matched to the input signal.

Figure 6 Quarter Wave Matched Attenuator

(Series connected Diodes)

Figure 8 Lumped Circuit Equivalent

Of Quarter Wave Line

Figure 7 Quarter Wave Matched Attenuator

(Shunt Connected Diodes)



l /4

Quadrature hybrid attenuators may also be con-structed without the load resistor attached in series orparallel to the PIN diode as shown. In these circuits theforward current is increased from the 50%, maximumattenuation/Rs value to lower resistance values. Thisresults in increased stored charge as the attenuation islowered which is desirable for lower distortion. The pur-pose of the load resistor is both to make the attenuatorless sensitive to individual diode differences and increasethe power handling capacity by a factor of two.

Quarter-Wave AttenuatorsAn attenuator matched at the input may also be built

using quarter-wave techniques. Figures 6 and 7 showexamples of these circuits. For the quarter-wave section alumped equivalent may be employed at frequencies toolow for practical use of line lengths. This equivalent isshown in Figure 8.

The performance equations for these circuits are givenbelow:

Figure 11 Attenuatin of PI Attenuators

Figure 9 Bridged Tee Attenuator

Figure 10 PI Attenuator

(The π and Tee are broadband matched attenuator circuits.)



A matched condition is achieved in these circuitswhen both diodes are at the same resistance. This condi-tion should normally occur when using similar diodessince they are DC series connected, with the same for-ward bias current flowing through each diode. The seriescircuit of Figure 6 is recommended for use at high atten-uation levels while the shunt diode circuit of Figure 7 isbetter suited for low attenuation circuits.

Bridged TEE and PI AttenuatorsFor matched broadband applications, especially

those cover the low RD (1 MHz) through UHF, attenua-tor designs using multiple PIN diodes are employed.Commonly used for this application are the bridged TEEand PI circuits shown in Figures 9 and 10.

The attenuation obtained using abridged TEE circuitmay be calculated from the following:

The relationship between the forward resistance ofthe two diodes insures maintenance of a matched circuitat all attenuation values.

The expressions for attenuation and matching condi-tions for the PI attenuator are given as follows:

Using these expressions, Figure 11 gives a graphicaldisplay of diode resistance values for a 5Off PI attenua-tor. Note that the minimum value for Rs1 and Rs is 50%.In both the bridged TEE and PI attenuators, the PINdiodes are biased at two different resistance pointssimultaneously which must track in order to achieveproper attenuator performance.

PIN Diode ModulatorsPIN diode switches and attenuators may be used as

RF amplitude modulators. Square wave or pulse modu-lation use PIN diode switch designs whereas linear modulators use attenuator designs.

The design of high power or distortion sensitive mod-ulator applications follows the same guidelines as theirswitch and attenuator counterparts. the PIN diodes they employ should have thick I-regions. Seriesconnected or preferably back-to-back configurationsalways reduce distortion. The sacrifice in using thesedevices will be lower maximum frequencies and highermodulation current requirements.

The quadrature hybrid design is recommended as abuilding block for PIN diode modulators. Its inherentbuilt-in isolation minimizes pulling and undesired phase modulation on the driving source.

Figure 12 Switched Line Phase Shifter

Figure 13 Loaded Line Phase Shifter



PIN Diode Phase ShiftersPIN diodes are utilized as series or shunt connected

switches in phase shifter circuit designs. In such cases,the elements switched are either lengths of transmissionline or reactive elements. The criteria for choosing PINdiodes for use in phase shifters is similar to those used inselecting diodes for other switching applications. Oneadditional factor, however, that must often be considered,is the possibility of introducing phase distortion particu-larly at high RF power levels or low reverse bias voltages.Of significant note is the fact that the properties inherentin PIN diodes which yield low distortion, i.e., a long car-rier lifetime and thick I-regions, also result in low phasedistortion of the RF signal. Three of the most commontypes of semiconductor phase shifter circuits, namely: theswitched line, loaded line and hybrid coupled designsare described as follows:

A. Switched Line Phase ShifterA basic example of a switched line phase shifter circuit

is shown in Figure 12. In this design, two SPDT switchesemploying PIN diodes are used to change the electricallength of transmission line by some length. . The phaseshift obtained from this circuit varies with frequency andis a direct function of this differential line length as shownbelow:

The switched line phase shifter is inherently a broadband circuit producing true time delay, with the actualphase shift dependent only on A,g. Because of PIN diodecapacitance limitations this design is most frequentlyused at frequencies below 1 GHz.

The power capabilities and loss characteristics of theswitched line phase shifter are the same as those of aseries connected SPDT switch. A unique characteristic ofthis circuit is that the power and voltage stress on eachdiode is independent of the amount of differential phase

shift produced by each phase shifter. Thus, four diodesare required for each bit with all diodes having the samepower and voltage ratings.

B. Loaded Line Phase ShifterThe loaded line shifter design shown in Figure 13

operates on a different principle than the switched linephase shifter. In this design the desired maximum phaseskirt is divided into several smaller phase shift sections,each containing a pair of PIN diodes which do not com-pletely pertubate the main transmission line. A majoradvantage of this phase shifter is its extremely highpower capability due partly to the use of shunt mounteddiodes plus the fact that the PIN diodes are never in thedirect path of the full RF power.

In loaded line phase shifters, a normalized suscep-tance, Bn, is switched in and out of the transmission pathby the PIN diodes. Typical circuits use values of Bn,much less than unity, thus resulting in considerabledecoupling of the transmitted RF power from the PINdiode. The phase shift for a single section is given as fol-lows:

The maximum phase shift obtainable from a loadedline section is limited by both bandwidth and diodepower handling considerations. The power constraint onobtainable phase shift is shown as follows:



Figure 14 Hybrid Coupler Reflective Phase Shifter

The above factors limit the maximum phase shiftangle in practical circuits to about 45°. Thus, a 180° phaseshift would require the use of four 45° phase shift sec-tions in its design.

• Reflective Phase ShifterA circuit design which handles both high RF power

and large incremental phase shifts with the fewest numberof diodes is the hybrid coupled phase shifter shown inFigure 14. The phase shift for this circuit is given below:

The voltage stress on the shunt PIN diode in this cir-cuit also depends on the amount of desired phase shift or“bit” size. The greatest voltage stress is associated withthe 180° bit and is reduced by the factor (sinø/2) 1⁄2 forother bit sizes. The relationship between maximumphase shift, transmitted power, and PIN diode ratings isas follows:

In comparison to the loaded line phase shifter, thehybrid design can handle up to twice the peak powerwhen using the same diodes. In both hybrid and loadedline designs, the power dependency of the maximum bitsize relates to the product of the maximum RF currentand peak RF voltage the PIN diodes can handle. By judi-cious choice of the nominal impedance in the plane ofthe PIN diode, the current and voltage stress can usuallybe adjusted to be within the device ratings. In general,this implies lowering the nominal impedance to reducethe voltage stress in favor of higher RF currents. For PINdiodes, the maximum current rating should be specifiedor is dependent upon the diode power dissipation ratingwhile the maximum voltage stress at RF frequencies isdependent on I-region thickness.

PIN Diode Distortion Model

The beginning sections of this article concerned withlarge signal operation and thermal considerations allowsthe circuit designer to avoid conditions that would lead tosignificant changes in PIN diode performance or exces-sive power dissipation. A subtle but often significantoperating characteristic is the distortion or change in sig-nal shape which is always produced by a PIN diode inthe signal it controls.

The primary cause of distortion is any variation ornon-linearity of the PIN diode impedance during theperiod of the applied RF signal. These variations couldbe in the diode’s forward bias resistance, Rs, parallelresistance, Rp, capacitance, CT, or the effect of the lowfrequency I-V characteristic. The level of distortion canrange from better than 100 dB below, to levels approach-ing the desired signal. The distortion could be analyzedin a fourier series and takes the traditional form of har-monic distortion of all orders, when applied to a singleinput signal, and harmonic intermodulation distortionwhen applied to multiple input signals.

Non-linear, distortion generating behavior is oftendesired in PIN and other RF oriented semiconductordiodes. Self-biasing limiter diodes are often designed asthin I-region PIN diodes operating near or below theirtransit time frequency. In a detector or mixer diode thedistortion that results from the ability of the diode to fol-low its I-V characteristic at high frequencies is exploited.In this regard the term “square law detector” applied to adetector diode implies a second order distortion genera-tor. In the PIN switch circuits discussed at the beginningof this article, and the attenuator and other applicationsdiscussed here, methods of selecting and operating PINdiodes to obtain low distortion are described.

There is a common misconception that minoritycartier Lifetime is the only significant PIN diode para-meter that affects distortion. This is indeed a major fac-tor, but another important parameter is the width of theI-region, which determines the transit time of the PINdiode. A diode with a long transit time will have more ofa tendency to retain its quiescent level of stored charge.The longer transit time of a thick PIN diode reflects itsability to follow the stored charge model for PIN dioderesistance according to:

The effect of carrier lifetime on distortion relates to thequiescent level of stored charge induced by the DC forward

AP5508-98 Alpha Industries, Inc. [781] 935-5150 • Fax [617] 824-4579 • Email [email protected] • www.alphaind.com 17


bias current and the ratio of this stored charge to theincremental stored charge added or removed by the RFsignal.

The distortion generated by a forward biased PINdiode switch has been analyzed and has been shown tobe related to the ratio of stored charge to diode resistanceand the operating frequency. Prediction equations for thesecond order intermodulation intercept point (IP2) andthe third order intermodulation intercept point (IP3)have been developed from PIN semiconductor analysisand are presented as follows:

In most applications, the distortion generated by areversed biased diode is smaller than forward biasedgenerated distortion for small or moderate signal size.This is particularly the case when the reverse bias appliedto the PIN diode is larger than the peak RF voltage pre-venting any instantaneous swing into the forward biasdirection.

Distortion produced in a PIN diode circuit may bereduced by connecting an additional diode in a back toback orientation, (cathode to cathode or anode to anode).This results in a cancellation of distortion currents. Thecancellation should be total, but the distortion producedby each PIN diode is not exactly equal in magnitude andopposite in phase. Approximately 20 dB distortionimprovement may be expected by this back to back con-figuration.

Distortion in Attenuator CircuitsIn attenuator applications, distortion is directly relat-

able to the ratio of RF to DC stored charge. In such appli-cations, PIN diodes operate only in the forward bias stateand often at high resistance values where the storedcharge may be very low. Under these operating condi-tions, distortion will vary with charges in the attenuationlevel. Thus, PIN diodes selected for use in attenuator cir-cuits need only be chosen for their thick I-region width,since the stored charge at any fixed diode resistance, Rs,is only dependent on this dimension.

Consider an SMP1304-001 PIN diode used in an appli-cation where a resistance of 50W is desired. The data sheetindicated the 1 mA is the typical diode current at whichthis occurs. Since the typical carrier lifetime for this diodeis 1 mS, the stored charge for the diode at 50W is 1.0 nC. Iftwo PIN diodes, however, are inserted in series, toachieve the same 50W resistance level, each diode mustbe biased at 2 mA. This results in a stored charge of 2 nC

log

log

per diode or a net stored charge of 4 nC. Thus, adding asecond diode in series multiples the effective storedcharge by a factor of 4. This would have a significant pos-itive impact on reducing the distortion produced byattenuator circuits.

Measuring DistortionBecause distortion levels are often 50 dB or more

below the desired signal, special precautions are requiredin order to make accurate second and third order distor-tion measurements. One must first ensure that the signalsources used are free of distortion and that the dynamicrange of the spectrum analyzer employed is adequate tomeasure the specified level of distortion. These require-ments often lead to the use of fundamental frequencyband stop filters at the device output as well as pre-selec-tors to clean up the signal sources employed. In order toestablish the adequacy of the test equipment and signalsources for making the desired distortion measurements,the test circuit should be initially evaluated by removingthe diodes and replacing them with passive elements.This approach permits one to optimize the test set-up andestablish basic measurement limitations.

Since harmonic distortion appears only at multiples ofthe signal frequency these signals may be filtered out innarrow band systems. Second order distortion, caused bythe mixing of two input signals, will appear at the sumand difference of these frequencies and may also be fil-tered. As an aid to identifying the various distortion sig-nals seen on a spectrum analyzer, it should be noted thatthe level of a second distortion signal will vary directly atthe same rate as any change of input signal level. Thus, a10 dB signal increase will cause a corresponding 10 dBincrease in second order distortion.

Third order intermodulation .distortion of two inputsignals at frequencies FA and FB often produce in-band,nonfilterable distortion components at frequencies of 2FA- FB and 2FB - FA. This type of distortion is particularlytroublesome in receivers located nearby transmittersoperating on equally spaced channels: In identifying andmeasuring such signals, it should be noted that thirdorder distortion signal levels vary at twice the rate ofchange of the fundamental signal frequency. Thus a 10 dBchange in input signal will result in a 20 dB change of thethird order signal distortion power observed on a spec-trum analyzer.Bibliography1. Garver, Robert V., “Microwave Control Devices,” Artech House, Inc.,Dedham, MA., 1976.2. Mortenson, K.E., and Borrego, J.M., “Design Performance andApplication of Microwave Semiconductor Control Components,”Artech House, Inc., Dedham, MA., 19723. Watson, H.A., “Microwave Semiconductor Devices and Their CircuitApplication.” McGraw Hill Book Co., New York, NY., 1969.4. White, Joseph F., “Semiconductor Control,” Artech House, Inc.,Dedham, MA., 1977.5. G. Hiller and R. Caverly, Distortion in PIN Diode Control CircuitsIEEE MTT Transactions, May, 1987.6. R. Caverly and G. Hiller, Establishing the Reverse Bias to a PIN Diodein a High Power Switch. IEEE MTT Transactions, Dec., 1990.

Alpha Industries, Inc. [781] 935-5150 • Fax [617] 824-4579 • Email [email protected] • www.alphaind.com 4-29-98

Singapore/Thailand/MalaysiaPangaea Corporation20 MacTaggart Rd. #07-02Khong Guan Industrial Bldg.368079 SingaporeTel: (65) 282-1870Fax: (65) 282-9879USA Office for PangaeaPangaea Associates3 Uplands Dr.Littleton, MA 01460 USATel: (978) 486-0942Fax: (978) 486-4647

AlphaAMERICASHeadquartersAlpha Industries20 Sylvan Rd.Woburn, MA 01801Tel: (781) 935-5150Fax: (617) 824-4579Alabama/Mississippi/Western TennesseeBeacon Electronics7501 Memorial Pkwy. S., #105Huntsville, AL 35802Tel: (205) 881-5031Fax: (205) 883-9516Arizona/New MexicoYoungwirth & Olenick Associates4855 E. Warner Rd. Suite 24-138Phoenix, AZ 85044Tel: (800) 515-1554Fax: (800) 515-1574Colorado/UtahCain-Pollock, Inc.21874 Unbridled Ave.Parker, CO 80134Tel: (303) 805-2515Fax: (303) 805-2514New York/Northern New JerseyTrionic Associates320 Northern Blvd., Suite 23Great Neck, Long Island, NY 11021Tel: (516) 466-2300Fax: (516) 466-2319FloridaBeacon Electronics2170 W. State Rd. 434, Suite 116Longwood, FL 32779Tel: (407) 788-1155Fax: (407) 788-1176

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QuebecCain-Sweet, Co.65 Brunswick Blvd., Suite 107Dollards Des OrmeauxQuebec, H9B 2N4 CanadaTel: (514) 421-0848Fax: (514) 421-1324USA Headquarters/Cain-Sweet, Co.Cain-Sweet, Co.1409 140th Place NE Suite 105Bellevue, WA 98007-3963Tel: (425) 462-2118Fax: (425) 451-9418Ontario/Manitoba/Atlantic ProvincesCain-Sweet, Co.1171-2720 Queensview Dr.Ottawa, Ontario, K2B 1A5 CanadaTel: (613) 820-2606Fax: (613) 820-2709TorontoCain-Sweet, Co.5555 Elgington, Suite 202Etobicoke, Ontario M9C 5M1 CanadaTel: (416) 695-1444Fax: (416) 695-2999BrazilVermont RepresentativesCommercial S/C Ltda.Rua Dom Paulo Pedrosa 400 Real Parque, 05687-000Sao Paulo, SP BrasilTel: (55) 11-846-7372Fax: (55) 11-846-0308MexicoMexitek S.A.Porfirino Diaz 53Col. Del ValleMexico 03100, D.F. Tel: (52) 5-575-0269Fax: (52) 5-575-9981

EUROPE • MIDDLE EAST • AFRICAAustraliaElectronic Development Sales Pty. Ltd.11-13 Orion Rd., Unit 2ALane Cove NSW 2066 AustraliaTel: (61) 2-9418-6999Fax: (61) 2-9418-6550DenmarkTech Partner A/STomsagervej 18 8230 AabyhojDenmarkTel: (45) 874-61600Fax: (45) 874-61616England/Ireland/Scotland/WalesAlpha Industries, Ltd.19/21 Chapel St.Marlow Bucks, SL73HNEnglandTel: (44) 1628-475562Fax: (44) 1628-474078Finland/RussiaIntegrated Electronics Oy AbLaurinmaenkuja 3AP.O. Box 31, FIN-00441Helsinki, FinlandTel: (358) 9-586-1770Fax: (358) 9-586-1771

France/BelgiumMillimondes S.A.18/22 Avenue Edouard Herriot92350 Pessis RobinsonFranceTel: (33) 1-45-37-12-30Fax: (33) 1-46-32-81-06Germany/Austria/Switzerland/Yugoslavia/Eastern EuropeMunicom GmbHFuchsgrube 4Traunstein 83278GermanyTel: (49) 861-16677-0Fax: (49) 861-16677-88GreeceKostas Karayannis S.A.Kapodistriou Avenue, No. 58GR 142 35 Nea IoniaAthens, GreeceTel: (30) 1-68-00-4601Fax: (30) 1-68-53-522IndiaAarjay International Pvt. Ltd.167, I Main, II CrossDomuir - II StageBangalore, 560 071 IndiaTel: (91) 80-5262874Fax: (91) 80-5274616

USA Office for IndiaElkay International Inc.15 Commerce BoulevardSuccasunna, NJ 07876 USATel: (201) 927-8647Fax: (201) 927-5370IsraelOrmic Components1 Atididim Technological ParkP.O. Box 58029Tel Aviv, 61580 IsraelTel: (972) 3-6491118Fax: (972) 3-6471942ItalyMicroelit S.P.A.Via Sardegna, 120146 Milan, ItalyTel: (39) 2-481-7900Fax: (39) 2-481-3594

Microelit S.P.A.Via Nicola Marchese, 10 00141 Rome, ItalyTel: (39) 6-86894323Fax: (39) 6-8275270

Netherlands/Belgium/LuxembourgSemi Dice International BVMiddel 6, 1551SP WestzaanP.O. Box 303 1520 AH WormerveerThe NetherlandsTel: (31) 756476969Fax: (31) 756408795NorwayDatamatik asJerikoveien 16, P.O. Box 12Lindeberg Gaard1007 Oslo NorwayTel: (47) 22-301730Fax: (47) 22-300273South AfricaSimm Electronics1279 Mike Crawford Ave.Suite 201Die AnkerCenturion, South AfricaTel: (27) 12-663-8578Fax: (27) 12-663-8753

SpainIberica de Componentes, S.A.Avda Somosierra, 112, 1 of A28709 San Sebastian de los ReyesMadrid, SpainTel: (34) 91-6541856Fax: (34) 91-6531019SwedenNordimarBox 49S-182 05 Djursholm SwedenTel: (46) 8-753-3760Fax: (46) 8-622-6304TurkeyEltronik Elektronik Sanayli

Ve Ticaret Ltd. StiFarabi Sckak No. 44/406690 CankayaAnkara, TurkeyTel: (90) 312-467-9952Fax: (90) 312-467-9828USA Office for TurkeyEltronik Inc.8427 Kennedy Blvd.North Bergen, NJ 07047 USATel: (201) 453-0401Fax: (201) 453-0959

ASIA • PACIFICChina/Hong KongPangaea (Hong Kong) Ltd.Room 1810 Tai Yau Building181 Johnston RoadWanchai, Hong KongTel: (852) 2836-3301Fax: (852) 2834-7340KoreaS-TEC Int’l. Co., Ltd.P.O. Box 577 Yoido-DongYeung-Deung-PoSeoul, KoreaTel: (82) 2-784-9898Fax: (82) 2-784-8600

JapanM-RF Co., Ltd.1-7-9 Izumi-choKanda, Chiyoda-kuTokyo 101 JapanTel: (81) 3-5821-3623Fax: (81) 3-5821-3625Osaka BranchM-RF K.K.2-4-5 Minami-MorimachiKita-KuOsaka 530 JapanTel: (81) 6-313-7180Fax: (81) 6-313-7182

PhilippinesPangaea InternationalTrading CorporationUnit 703 Alabang Business TowerBlk. 1, Lot 6, Acacia Ave.Madrigal Business ParkAyala Alabang1780 Muntinlupa City, PhilippinesTel: (632) 807-8430Fax: (632) 809-1355

TaiwanAmeleks Corp.6F, 48 Keelung Rd., Sec. 2Taipei, Taiwan R.O.C.Tel: (886) 2-758-6958Fax: (886) 2-758-6987

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