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1. I.J. Bahl and P. Bhartia, Microstrip antennas, Artech House, NewDelhi, 1980, pp. 1, 46–47, 57, 230.

2. Z. Qi and B. Lieng, Design of microstrip antenna with broader band-width and beam, Antennas Propag Soc Int Symp 3A (2005), 617–620.

3. S. Maci and G. Biffi Gentili, Dual frequency patch antennas, IEEETrans Antennas Propag 39 (1997), 13–18.

4. S.O. Kundukulam, M. Paulsou, C.K. Aanandan, and P. Mohanan,Slot-loaded compact microstrip antenna for dual-frequency operation,Microwave Opt Technol Lett 31 (2001), 379–381.

5. W.-C. Liu and H.-J. Liu, Compact triple band slotted monopoleantenna with asymmetrical CPW grounds, Electron Lett 37 (2006),840–842.

6. V. Hamsakutty and K.T. Mathew, Dual frequency hexagonal dielectricresonator antenna for DCT and WLAN applications, Proc Natl SympAntennas Propag 54 (2006), 199–200.

7. L. Liu, S. Zhu, and R. Langley, Dual band triangular patch antennawith modified ground plane, Electron Lett 43 (2007), 140–141.

8. X. Wang, W. Chin, and Z. Feng, Multiband antenna with parasiticbranches for laptop applications, Electron Lett 43 (2007), 1012–1013.

9. S.-J. Lin and J.-S. Row, Bandwidth enhancement for dual frequencymicrostrip antenna with conical radiation, Electron Lett 44 (2008),2–3.

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11. R. Chatterjee, Antennas theory and practice, Wiley Eastern Ltd, 1998,pp. 156–158.

12. D.M. Pozar, Microwave engineering, Addison Wesley PublishingCompany Inc., Reading, MA, 1990, pp. 185–186, 302.

13. K.F. Lee and W. Chen, Advances in microstrip and printed antennas,Wiley, New York, 1997, pp. 145.

14. J.-S. Row, Dual-frequency triangular planar inverted-F antenna, IEEETrans Antennas Propag 53 (2005), 874–876.

© 2009 Wiley Periodicals, Inc.

DESIGN OF MULTILAYER TRIANGULARSUBSTRATE INTEGRATED WAVEGUIDEFILTER IN LTCC

Wei Shen,1,2 Xiao-Wei Sun,2 Wen-Yan Yin,1 Jun-Fa Mao,1 andQi-Fu Wei11 Center for Microwave and RF Technologies, School of ElectronicInformation and Electrical Engineering, Shanghai Jiao Tong University,Shanghai 200240, People’s Republic of China; Correspondingauthor: weishen@sjtu.edu.cn2 Institute of Microsystem and Information Technology of ChineseAcademy of Sciences, Shanghai 200050, People’s Republic of China

Received 22 February 2009

ABSTRACT: This article proposes a four-pole quasi-elliptic functiontriangular substrate integrated waveguide bandpass filter, which wasfabricated using multilayer low-temperature cofired ceramic technology.In our design, neural network inverse model is adopted to fast capturethe sizes of TSIW cavities, with the coupling matrix transformation im-plemented to achieve cross coupling on the basis of multilayer TSIW. Aslot on the top metal layer is introduced so as to suppress the twohigher spurious modes. The filter has 65% size reduction in comparisonwith its planar counterpart, with good agreement obtained between thesimulated and measured S-parameters. © 2009 Wiley Periodicals, Inc.Microwave Opt Technol Lett 51: 2582–2585, 2009; Published online inWiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.24680

Key words: index terms-bandpass filter (BPF); coupling matrix; neuralnetwork (NN) inverse model; triangular substrate integrated waveguide(TSIW); S-parameters

1. INTRODUCTION

It is understood that both positive and negative coupling areneeded to generate transmission zeros at finite frequencies so as toachieve a high selectivity in a cross-coupled bandpass filter (BPF)[1]. However, it is not easy to get a negative coupling in manycases in the design of substrate integrated waveguides (SIW) filters[2].

In the past several years, SIWs have attracted special attention,which are normally realized by introducing metallic via-holes indielectric substrates [2, 3]. Evidently, standard low-temperaturecofired ceramic (LTCC) technology seems to be the most efficientmethod in the realization of SIW-based components, because of itsvery flexible three-dimensional integration capabilities, low toler-ance in processing and low dielectric loss of LTCC materials [4].On the other hand, we know that normal SIW filters are built basedon rectangular and circular cavities. However, by taking area anddesign flexibility into account, triangular cavities, such as isoscelesright one, etc., are still useful in the design of SIW-based filters.

In this article, a neural network (NN) inverse model is adoptedto determine the sizes of a triangular substrate integratedwaveguide (TSIW) cavity. A novel four-pole multilayer TSIWfilter is proposed and fabricated using a standard LTCC technol-ogy. One slot is introduced on the top metal surface to suppress thetwo higher spurious modes in the filter, with its good performancedemonstrated numerically and experimentally.

2. FILTER ANALYSIS AND DESIGN

2.1. TSIW ResonatorsThe electric field distributions of the fundamental and the second-order modes in a triangular cavity are shown in Figure 1. Here, thelength of the isosceles right side of the triangular is given by a,with the diameter of metallized via-holes and center-to-centerpitch between two adjacent via-holes defined by d and p, respec-tively.

To the best of our knowledge, there is no closed-form equationof the fundamental resonant frequencies as a function of thegeometrical parameters of a TSIW cavity. But in our design, a NNinverse model [5] will be utilized to build up the nonlinear rela-tionship among them. We can get the side length a from theinverse model for its uniqueness of the input and output relation-ship.

Figure 1 The electric field distribution in the TSIW cavity correspond-ing to the (a) fundamental and (b) second-order modes, respectively. [Colorfigure can be viewed in the online issue, which is available at www.inter-science.wiley.com]

2582 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 11, November 2009 DOI 10.1002/mop

However, direct training the NN inverse model may becomedifficult due to the nonuniqueness between the input and outputvariables. Here, d is chosen to be 0.18 mm, and 0.26 mm astraining samples, with the curve of the fundamental resonantfrequencies plotted in Figure 2. Inverse model exchanges a andresonant frequency f1 from the direct model. As shown in Figure2, once d is determined, uniqueness between f1 and a will beobtained.

2.1. Coupling MatrixTwo cross-coupled SIW filters are presented in [4]. However, thecoupling matrix is not suitable for constructing multilayer filtersbecause two ports are not located in the same plane [4], it isinconvenient to connect bottom microstrip line to the top layer bya buried via hole, and further insertion loss may be introduced.

The fabricated filter has the central frequency of 15 GHz andfractional bandwidth of 5%.We can obtain the coupling matrix asfollows:

�M� � �0 0.0439 0 � 0.0087

0.0439 0 0.0386 00 0.0386 0 0.0439

� 0.0087 0 0.0439 0� (1)

and the external quality factor is qe � 18.76 obtained by theclassical method [6, 7].

The relationship between the S-parameters and the couplingmatrix is described by [6]:

S11 � 1 � j2qe�1�A�11

�1, S21 � � j2qe�1�A�n1

�1 (2)

where A � pU � jqe � m, p is the low pass frequency, U is theidentity matrix of n � n, qe represents external quality factor, andm is the coupling matrix.

Here, for design of multilayer structure, we suppose J� diag� � 1,1,1,1�4�4, m in (2) is replaced by J � m � J, thenS-parameters keep unchanged. The coupling coefficients can beextracted from matrix J � m � J, M14 � 0.0087, and M12 �� 0.0439, with other elements unchanged.

2.1. Structure of Filter and Spurious SuppressionThe structure of the cross-coupled TSIW filter is shown in Figure3(a). We can get the initial sizes of cavity from the NN model,electric coupling M12 is realized by etching a circular aperture onthe middle metal layer in Figure 3(a). Magnetic coupling can beobtained by a postwall iris in common post wall and rectangle slotson middle metal layer. The external quality factor qe can beextracted from the phase response of S11 of a singly fed cavity.

Then, the initial model of the filter can be constructed in HighFrequency Structure Simulator (HFSS). The model is tuned byadjusting the geometries of the coupling mechanisms until idealresponses are obtained [8].

A slot on the top metal layer is introduced to suppress the twohigher spurious modes. The surface current distribution, corre-sponding to the second-order mode in TSIW, is plotted in Figure3(b). If another slot is etched on resonator 4, poor performance ofsuppression is achieved due to the same spurious mode in resona-tor 1 and 4. Only the first higher mode in circular and rectangularcavity is suppressed and the second higher mode is not impacted.Then wider stopband can be obtained in triangular than other twostructures.

3. RESULTS AND DISCUSSION

Two filter samples were fabricated using eight-layer LTCC sub-strate of Ferro-A6, with a relative permittivity of � � 5.9, and aloss tangent of tan� � 0.0015. The thickness of each layer is0.096 mm. All metallic via holes used to build TSIW have thesame diameter of 0.18 mm, with the geometrical sizes and itsphoto shown in Figure 4. Their measured and simulated S-param-eters are plotted in Figure 5, and the measured in-band return and

Figure 2 Nonlinear relation between the fundamental resonant frequen-cies and the side length of TSIW

Figure 3 (a) Structure of the filter and (b) its surface current of thesecond-order mode. [Color figure can be viewed in the online issue, whichis available at www.interscience.wiley.com]

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 11, November 2009 2583

insertion losses are below �16dB and about 1.6 dB, respectively.There are two spurious passband at 22 and 29 GHz in Figure 5(b).The spurious suppression responses are shown in Figure 6 forcomparison.

In Figure 6, excellent spurious suppression response isachieved by introducing a slot on the top metal plane, and thespurious response is below �20 dB from 15 to 34 GHz. On theother hand, we would like to say that good agreement is obtainedbetween the simulated and measured S-parameters here, except themeasured central frequency up shift 0.5% as compared with thesimulation result. This may be contributed to the bigger LTCCshrinkage. Therefore, TSIW cavities are smaller than expected andcause central frequency shift. The insert loss of the filter withsuppressed slot is 1.6 dB which is nearly impacted by the slot.

Figure 4 Geometry and photo of the fabricated filter (unit: mm). (a) Topview, (b) bottom view, and (c) its photo. [Color figure can be viewed in theonline issue, which is available at www.interscience.wiley.com]

Figure 5 S-parameters of the filter as a function of frequency. (a) narrowband and (b) wideband cases, respectively

Figure 6 The S-parameters of the filter with spurious suppression

2584 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 11, November 2009 DOI 10.1002/mop

4. CONCLUSION

In this article, a novel multilayer TSIW filter is proposed andrealized using a standard LTCC technology. The size of the TSIWis fast captured by the NN inverse model. It is demonstrated thatour proposed filter has very good selectivity, spurious suppression,and in particular it is compact in geometry. It can be directlyintegrated with other planar circuits at a low cost for microwaveand millimeter-wave applications.

ACKNOWLEDGMENTS

This work was supported by the National Basic Research Programunder Grant 2009CB320204, and by the NSF under Grant90607011 of China.

REFERENCES

1. J.S. Hong and M.J. Lancaster, Couplings of microstrip square open-loopresonators for cross-coupled planar microwave filters, IEEE Trans Mi-crowave Theory Tech 24 (1996), 2099–2109.

2. X.P. Chen and K. Wu, Substrate integrated waveguide cross-coupledfilter with negative coupling structure, IEEE Trans Microwave TheoryTech 56 (2008), 142–149.

3. H.J. Tang, W. Hong, Z.C. Hao, J.X. Chen, and K. Wu, Optimal designof compact millimeter-wave SIW circular cavity filters, Electronic Lett41 (2005), 1068–1069.

4. T.M. Shen, C.F. Chen, T.Y. Huang, and R.B. Wu, Design of verticallystacked waveguide filters in LTCC, IEEE Trans Microwave TheroyTech 55 (2007), 1771–1779.

5. H. Kabir, Y. Wang, M. Yu, and Q.J. Zhang, Neural network inversemodeling and applications to microwave filter design, IEEE TransMicrowave Theory Tech 56 (2008), 867–879.

6. J.S. Hong and M.J. Lancaster, Microstrip filters for RF/microwaveapplications, Wiley, New York, 2001.

7. R.S. Amari, Synthesis of cross-coupled resonator filters using an ana-lytical gradient-based optimization technique, IEEE Trans MicrowaveTheory Tech 48 (2000), 1559–1564.

8. T. Shen, H.-T. Hsu, K.A. Zaki, A.E. Atia, and T.G. Dolan, Full-wavedesign of canonical waveguide filters by optimization, IEEE TransMicrowave Theory Tech 51 (2003), 504–511.

© 2009 Wiley Periodicals, Inc.

KU-BAND SUBSTRATE INTEGRATEDWAVEGUIDE TRANSITIONS BETWEENLAYERS

Kaijun Song1,2 and Yong Fan1

1 School of Electronic Engineering, University of Electronic Scienceand Technology of China, Chengdu, 610054, China; Correspondingauthor: kjsong@ee.uestc.edu.cn, kaijsong@ee.cityu.hk2 State Key Lab of Millimeter Waves, Department of ElectronicEngineering, City University of Hong Kong, Hong Kong

Received 25 February 2009

ABSTRACT: Two types of the substrate integrated waveguide (SIW)transitions between layers are presented in this letter. The SIW circuitshave been used to achieve the transition from one layer to other layer.These two types of SIW transitions between layers operating in Ku-bandhave been designed and fabricated. The measured 10-dB return lossbandwidth of the transition between adjacent layers is about 7 GHz, andthe measured 1-dB insertion loss bandwidth is about 5 GHz; the mea-sured 13-dB return loss bandwidth of the unadjacent-layers transition isdemonstrated to be about 3.7 GHz, and the measured insertion losses

are less than 2 dB from 11.2 GHz to 15 GHz. The simulated and mea-sured results indicate that these two SIW transitions take the advantagesof broadband, low insertion loss, and low profile. © 2009 Wiley Period-icals, Inc. Microwave Opt Technol Lett 51: 2585–2588, 2009;Published online in Wiley InterScience (www.interscience.wiley.com).DOI 10.1002/mop.24685

Key words: transition; Ku-band; substrate integrated waveguide (SIW);multilayered PCB

1. INTRODUCTION

Multilayered printed circuit boards (PCB) and low-temperaturecofired ceramics (LTCC) circuits are the two low-cost fundamentalbuilding blocks, and can be widely used in high-density integrationof RF and microwave systems. Obviously, the transitions betweenthe layers are the key circuit elements in order to ensure circuit andsystem integration in multilayered LTCC or PCB circuits. A layer-to-layer transition is required to achieve the low insertion loss andbroadband for these high-density integration systems. The tradi-tional transitions between layers in multilayered PCB or LTCCcircuits have the disadvantage of high insertion loss and aredifficult to integrate. Hence, to solve these problems, it is impor-tant to look into alternatives with low insertion loss, broadband,and ease of integration in the multilayered PCB or LTCC circuits.

New technologies of passive components always stimulate thedesign of microwave devices including transitions between layers.Recently, a convenient and interesting planar integration schemecalled substrate integrated waveguide (SIW) has already attractedmuch interest [1, 2]. The SIW is synthesized in a planar substratewith arrays of metallic via, and can be fabricated by standard PCBor the LTCC process. Compared with the traditional waveguide ormicrostrip-like transmission-line structures, SIW takes the advan-tage of both waveguide and microstrip structures, like the high-Qfactor, high power capacity, small size, low loss, and the possibil-ity of integration. In view of the above advantages, it is anappropriate choice for realizing components in microwave andmillimeter wave high-density integrated circuits [2–7]. A numberof integrated components have been developed including numer-ous power dividers, filters, antennas, mixer, and oscillators [2–7].Therefore, it has the potential to design a layer-to-layer transitionwith perfect performance in multilayered PCB or LTCC circuitsusing the SIW circuits.

In this article, we have presented two SIW layer-to-layer tran-sitions: transitions between adjacent layers and transitions betweenunadjacent layers. These SIW transitions between layers take theadvantages of low cost, low loss, broadband, low profile, etc., andcan easily be integrated into RF/microwave multilayered PCB orLTCC circuits.

2. STRUCTURE AND DESIGN OF THE SIW TRANSITIONSBETWEEN LAYERS

The proposed SIW transition between adjacent layers is integratedin PCB circuits, as shown in Figure 1. There is a microstrip-to-SIW transition at the top layer, while there is a SIW-to-microstriptransition at the bottom payer. A taper transition between themicrostrip line and the SIW is used in this SIW layer-to-layertransition, which can provide broadband characteristics. The SIWis a type of dielectric-filled waveguide that is synthesized in aplanar substrate with arrays of metallic via to realize circumfer-ential walls. The metallic vias inserted to suppress the parallel-plate leaky modes create electrical walls on the SIW walls, andthen the SIW can be viewed as a conventional rectangularwaveguide.

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 51, No. 11, November 2009 2585