design of an ultra-high efficiency gan high-power amplifier
TRANSCRIPT
Design of an Ultra-High Efficiency GaN High-PowerAmplifier for SAR Remote SensingTushar Thrivikraman
Radar Science and EngineeringJet Propulsion Laboratory
California Institute of Technology4800 Oak Grove Drive, Pasadena, CA 91109
James HoffmanRadar Science and Engineering
Jet Propulsion LaboratoryCalifornia Institute of Technology
4800 Oak Grove Drive, Pasadena, CA 91109818-354-4384
Abstract—This work describes the development of a high-poweramplifier for use with a remote sensing SAR system. The am-plifier is intended to meet the requirements for the SweepSARtechnique for use in the proposed DESDynI SAR instrument. Inorder to optimize the amplifier design, active load-pull techniqueis employed to provide harmonic tuning to provide efficiencyimprovements. In addition, some of the techniques to overcomethe challenges of load-pulling high power devices are presented.The design amplifier was measured to have 49 dBm of outputpower with 75% PAE, which is suitable to meet the proposedsystem requirements.
TABLE OF CONTENTS
1 INTRODUCTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 MIXED-SIGNAL ACTIVE LOAD-PULL MEA-
SUREMENT . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23 LOAD-PULLING HIGH POWER TRANSISTORS . . 34 HIGH POWER AMPLIFIER DESIGN . . . . . . . . . . . . . . . 45 SUMMARY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
REFERENCES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6BIOGRAPHY . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1. INTRODUCTIONRequirements for next generation SAR remote sensing sys-tems demand new technology to allow these systems tobe feasible. Increased swath size, high resolution, rapidglobal coverage, as well as sub-cm interferometry and po-larimetry require advanced techniques such as SweepSAR,which would be employed by the proposed Earth Radar Mis-sion’s (ERM) DESDynI (Deformation, Ecosystem Structure,and Dynamics of Ice) SAR Instrument (DSI). SweepSARwould use multiple transmit/receive (T/R) channels and digi-tal beamforming to achieving simultaneously high resolutionand large swath [1].
The SweepSAR technique (Fig. 1) would use a large aper-ture reflector with a linear patch feed array, with each setof patches fed by a single T/R module. On transmit, allT/R modules would be used in unison, sub-illuminatingthe reflector creating a large swath on the ground. Whileon receive, individual beams would be formed by stitchingmultiple receivers together using digital beamforming [2].This technique would produce, for transmit, an electricallysmall antenna, illuminating a large area on the ground, whileon receive, smaller beams would be formed, yielding higherresolution. Due to the large swath, a receiver would have
978-1-4673-1813-6/13/$31.00 c©2013 IEEE.
Figure 1. SweepSAR technique highlighting transmit andreceive operation. Beamforming on transmit would producea single large beam covering a wide-swath. Digital beam-forming on receive would allow for multiple high resolutionbeams.
valid data across many transmit events, therefore, any trans-mit event would cause a loss of science data (gaps in theswath). Therefore, the transmit pulse width should be narrowas possible, limiting the total amount of power available toilluminated the ground. However, due to the size of the swath,the transmit energy would be spread over a large area, whichwould demand a longer pulse width and higher peak transmitpower. A longer pulse width is not an option, therefore,multiple high-power T/R modules would be required.
Previous generations of high-power amplifiers for use inremote sensing applications utilized GaAs and Si Bipolartransistors and are not suitable for large arrays containingmultiple high-power amplifiers. However, Gallium Nitride(GaN) High Electron mobility transistors (HEMTs) are anemerging technology that offers high-power density as wellas high efficiency, making them an effective solution forSweepSAR applications. The high breakdown voltage ofGaN as well as its excellent thermal properties make it a per-fect candidate for high-power amplifiers [3]. For commercialapplications, GaN has begun to become the technology ofchoice for RF transmitters in a variety of market segments.
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DUT
a1 b2
a2b1
S11*
S21*
S22*
S12*
Systemload
reflection𝛤L
Figure 2. Signal flow diagram for a large-signal load-pullmeasurement system.
This work explores the design of a high-power GaN amplifierthat is designed using a mixed-signal active load-pull system(MSALPS). The MSALPS allows for an optimized designthrough the use of harmonic tuning to determine optimalimpedance points. Section 2 provides a detailed discussion ofthe mixed-signal active load-pull system, section 3 discussesthe details of performing a load-pull on a high power device,and section 4 highlights the final amplifier design.
2. MIXED-SIGNAL ACTIVE LOAD-PULLMEASUREMENT
A “load-pull” is performed to characterize devices underlarge signal conditions, where simple linear two-port networktheory is not valid. Load-pulling presents the device withdesired impedances and measures the large signal response(output power, efficiency, etc) as shown in Fig. 2. Theseresponses can be plotted as contours on a smith chart, yieldinga simple graphical method for determining optimal designimpedances.
Typical load-pull systems use mechanical tuners to presenta mismatch to the device output, which can be calibrated tocreate a desired impedance at the device. Mechanical tunersare limited to tuning a narrow band signal as well as in theamount of reflection and therefore impedances that can beachieved at the device.
A solution to the problems with mechanical tuners is the useof active load-pull. This technique uses amplifiers in tuningloops to inject signals back into the DUT, allowing for higherreflection coefficients. Active load-pull systems can either beclosed- or open-loop as shown in Fig. 3 [4]. The closed-loopmethod, Fig. 3a, injects a sampled output signal back intothe DUT with appropriate phase and amplitude to producethe desired reflection. This method is optimal for fast devicecharacterization, but can be prone to oscillations and requiresfiltering. The MSALPS that is discussed in this work is anopen-loop active system (Fig. 3b), that utilizes independentarbitrary waveform generators (AWGs) to create independentchannels for source and load fundamentals and harmonics.The added complexity of this system allows for improvedperformance and greater ability to optimize the performanceof the device under test.
Anteverta-mw, in partnership with Maury Microwave, havedeveloped an open-loop active load-pull system that allowsfor wide-band modulated signals to be characterized. Thesystem contains a National Instrument PXI chassis, whichallows rapid and synchronous sampling of the incident andreflected waves as shown in Fig. 4. Each tuning looprequires a separate digital AWG that generates the appropriatewaveforms and is unconverted to RF for injection into thedevice. This procedure is handled automatically through thesoftware and algorithms developed by Anteverta-mw. The
ɸDUT
injection amplifierbias tee amplitude and phase
controlcouplerRF source
DUT
injection amplifierbias teeAWG source AWG source
(a)
(b)
Figure 3. Closed (a) and open (b) active load pull systemconfigurations. In (a) for the closed loop case, the injectedsignal is a sampled version of the output signal while in theopen loop case (b) the injected signal is generated by anindependent source.
DigitalAWG
x4DigitalA/D
PA @ fo
Output section
b2,f0 b2,2f0 a2,f0
LO LO
Input section
To DC
Bias Tee
High power fixture with bias decoupling
I 1
On-wafer configurationTo DC
To DC
BaseBand I V
To DC
Digital AWG Digital A/D Digital AWG
PA @ 2f0
PA @ f0RF @ f0
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a1,f0 a1,2f0 b1,f0 b1,2f0
LO LO LO
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LO
a1,f0 a1,2f0 a1,BB b1,f0 b1,2f0 b1,BB b2,f0 b2,2f0 b2,BB a2,f0 a2,2f0 a2,BB
RF Source
LO Source
To LO
X2
To RF@ f0
To RF@ 2f0
HPR
aREF
BaseBand I V
Bias Tee
I V BaseBandReference
Planes
I V BaseBand
Reference Planes
Load Reference
Plane
DUT Output section
b2,f0 b2,2f0 a2,f0
LO LO
Input section
To DC
Bias Tee
High power fixture with bias decoupling
I 1
On-wafer configurationTo DC
To DC
BaseBand I V
To DC
Digital AWG Digital A/D Digital AWG
PA @ 2f0
PA @ f0RF @ f0
RF @ 2f0
I/Q f0 signal
I/Q 2f0 signal
PA @ 2f0
PA @ f0 RF @ f0
RF @ 2f0
I/Q f0 signal
I/Q 2f0 signal
DUT
DUT
a1,f0 a1,2f0 b1,f0 b1,2f0
LO LO LO
a2,2f0
LO
a1,f0 a1,2f0 a1,BB b1,f0 b1,2f0 b1,BB b2,f0 b2,2f0 b2,BB a2,f0 a2,2f0 a2,BB
RF Source
LO Source
To LO
X2
To RF@ f0
To RF@ 2f0
HPR
aREF
BaseBand I V
Bias Tee
I V BaseBandReference
Planes
I V BaseBand
Reference Planes
Load Reference
Plane
a1 b1
NI PXI Chassis
b2 a2
PA @ fo
PA @ 2fo,3fo
source fo
load foload 2fo
load 3fo
receiver receiver receiver receiver
Figure 4. Block diagram of mixed-signal active load-pullsystem (MSALPS). The system is configured for four tuningloops, the fundamental source, and load fundamental, second,and third harmonics. Each of the loops require an individualarbitrary waveform generator that produces the appropriatewaveform to inject into the device to produce the desired Γ.
directional couplers are used to measure the a and b wavesfrom the input and output and digitized for software analysis.A diplexer is used to power combine the fundamental andharmonic signals for injection into the device. Tuning loop Ais used for the fundamental source, while tuning loops B, C,and D are used for the load fundamental, second, and thirdharmonic respectively.
The measurement is performed in two steps. The first stepis to determine the proper injection signals for all powersand impedance conditions. As shown in Fig. 5, the injectedsignals, a1,2 are calculated from the reflections to convergeto the correct impedance values. Once these waveforms havebeen determined, a detailed measurement is performed foreach load and power condition. The ‘real-time’ measure-ment mode allows for rapid characterization by measuringdifferent impedances conditions sequentially during a signalacquisition. Fig. 6 depicts typical waveforms for a real-timemeasurement using the MSALPS. This technique greatlyreduces time needed for load-pull characterization, a typicalmeasurement that could take hours on a traditional passivetuner can take a few minutes on the active load-pull system.Detailed measurement theory of the MSALPS can be foundin [5], [6], [7].
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SPECIFICATIONS SUBJECT TO CHANGE WITHOUT NOTICE
t e c h n i c a l d a t a 4T- 095
General Description
Series MT2000
Figure 1. Principle of the Mixed-Signal Load Pull setup as a Signal-Flow diagram
Wideband Generation/Detection
Figure 2. Output spectrum of the DUT at f0 and 2f0 tested with multi-carrier WCDMA(with and without delay compensation).
"Real-Time" Load Pull
Figure 3. Injection signals as used in the "real-time" multi-dimensional parameter sweeps.
Figure 5. Signal flow graph highlighting software developedinjected waveforms to produced desired impedance at device.As is the source signal, while a1,2 are the injected signals toproduce the desired impedance. [Courtesy of Anteverta-mw]
SPECIFICATIONS SUBJECT TO CHANGE WITHOUT NOTICE
t e c h n i c a l d a t a 4T- 095
General Description
Series MT2000
Figure 1. Principle of the Mixed-Signal Load Pull setup as a Signal-Flow diagram
Wideband Generation/Detection
Figure 2. Output spectrum of the DUT at f0 and 2f0 tested with multi-carrier WCDMA(with and without delay compensation).
"Real-Time" Load Pull
Figure 3. Injection signals as used in the "real-time" multi-dimensional parameter sweeps.Figure 6. Time series of ‘real-time’ load-pull waveforms,which allows for rapid characterization of many impedancespoints. [Courtesy of Anteverta-mw]
3. LOAD-PULLING HIGH POWERTRANSISTORS
Even though the basic concepts of load-pulling are wellunderstood, it can be challenging to implement these mea-surements in practice. For high-power devices (over 50 W),it is especially difficult to load-pull such devices due to thevery low matching impedances, the high thermal stresses,high dissipated power in the device, and the instabilities thatcould arise at some impedances during the load-pull.
For active load-pull systems, the injected power required isa function of the DUT’s output power and impedance. Forvery high power devices with low output impedance, it canbe challenging to inject enough power into the device from a50 Ω source to achieve the necessary gamma. The low deviceimpedance causes much of the device power to be reflectedand secondly any ohmic losses reduce the power available fortuning. In order to overcome this obstacle, a low-loss, high-power test fixture was developed that transforms the 50 Ωinjection amplifier impedance to 10 Ω.
The test fixture is required to be broad band to be able to tunethe fundamental, second, and third harmonics to optimize thedesign. This design uses a Klopfenstein taper that incorporatebias feed network for the gate and drain of the device.Incorporating the bias network allowed for improving theoverall loss, maintaining wide bandwidths, and handling thehigh bias currents and voltages. The feed network consistedof an RF choke and DC blocking cap. In addition, test pointswere added on the board to monitor the dc bias closer tothe device to reduce the effects of losses. The test fixtureis fully characterized over all desired tuning frequencies, andis deembeded from the measure device performance. Fig. 7depicts a picture of the fabricated test fixture with a tefloninsert that secures the device leads to the board without theneed for solder. Fig. 8 shows the calibrated response ofthe 10 Ω test fixture from 500 MHz to 5 GHz. For thisapplication, the low pulsed duty cycle allowed for simpleforced air cooling, however higher average power devices
could require inserts that provide for liquid cooling.
Figure 7. High-power transforming test fixture forMSALPS. Fixture transforms from 50 Ω at the injectionamplifiers to 10 Ω at the device reference plane
Figure 8. Reflection from DUT reference plane of 10 Ω ofinput and output transforming test fixture boards. Test fixtureexhibits broad band performance from 500 MHz to 5 GHzand less than 1 dB of insertion loss (not shown).
One drawback of the broadband test fixture is that it mightincrease the chance for oscillations outside the desired injec-tion bandwidth. These devices are typically unstable at lowfrequency due to their high gain, and therefore care should betaken to avoid causing these low-frequency oscillations. Dueto the large currents and voltages, a large amount of powerdissipation in the device could occur or large gate voltageswings could damage the gate junction and cause deviceburnout. Most of these failures cause permanent devicedamage, and can be observed by high gate leakage currents.In addition, care should to be taken to protect measurementequipment from high-power oscillations that could damagesensitive receive components.
If oscillations with high power devices are observed, a simplereactive feedback network can be added to the DUT to reducelow-frequency gain and improve stability. For the 100 Wdevice measured in this work, a 30 pF capacitor with a 220 Ωseries resistor connected with a short wire, yielding a few nHof inductance, improves the low frequency stability, withoutimpacting performance in the desired frequency band. Fig. 9shows an image of this feedback network across the device.As shown in Fig. 10, S-parameter response at L-band is un-changed, while low-frequency stability is greatly improved.
Using the 10 Ω transforming fixture with integrated bias teesand this feedback network, a 100 W GaN HEMT device ischaracterized on the MSALPS. These measured results arepresented in the next section.
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Figure 9. Image of load-pull test fixture with feedback net-work to improve low-frequency stability. Network includes a220Ω resistor and a 30 pF capacitor in series.
Figure 10. Return loss of simulated amplifier feedback net-work to improve stability of high power device. Pre-feedbackis shown in the shaded lines, amplifier with feedback is shownin the bold lines. Feedback network only had slight in-bandchanges.
4. HIGH POWER AMPLIFIER DESIGNThe first step in performing the load-pull characterization isto determine the appropriate input matching condition. Fig.11 shows the source pull results of the the max transducergain (GT ) at 1.25 GHz referenced to 10 Ω. The very lowinput impedance can be difficult to realize and achieve a wide-bandwidth and maintain device stability, a trade-off is madeto use a slightly higher impedance, reducing the overall gain.However, the small-signal gain is still high enough for thisgiven application. The source impedance was set to 2.2 - 1.4j.
The next step was to determine suitable load impedances.Care should be taken to avoid unstable regions of the deviceduring the load-pull. Typically this is achieved by lookingat stability criteria using measured or simulated s-parameterdata. Even though these metrics are better suited for linear,small-signal, they will indicate areas or conditions that couldpose potential instabilities and should be avoided, but they donot guarantee stability in large-signal operation.
The output power and power-added efficiency (PAE) contoursare shown in Fig. 12. Load-pull contours are circles ofconstant performance for that plotted metric. In these plots,
−1 −0.8 −0.6 −0.4 −0.2 0 0.2 0.4−1
−0.8
−0.6
−0.4
−0.2
0
0.2
0.4
16.5
1717.5
18
18.519
GainT
Figure 11. Transducer gain source pull of 100 W GaNHEMT with a 10 Ω reference impedance. Over 19 dB ofsmall-signal gain can be achieved by this device.
to avoid potentially unstable regions, fully closed contoursare not depicted, however, the trends can still be understood.For example, the 49.5 dBm output power line is the left mostcolored line in red, any impedance points inside this contourwill have higher power, while points outside will have lowerpower. The peak power is achieved to the left of the sweptimpedance values while the peak PAE is achieved to theupper-left.
The peak output power was measured to be just below 50dBm, while the efficiency was over 60%. These contoursshow the tradeoff between output power and PAE. Plottingthe power contours for the second and third harmonic (Fig.13), it is clear that higher efficiencies can be achieved byharmonic tuning. The optimal impedance point for highestefficiency is a trade-off between maintaing output power atthe fundamental, while controlling harmonic power levels.By controlling the harmonic impedances independently of theload impedance, these contours can be adjusted to achievehigh efficiencies while maintaining output power.
Now that source and load impedances are determined, har-monic impedances can be controlled independently to im-prove efficiencies while maintaining power at the fundamen-tal. Typically, the optimal impedances for the second andthird harmonic will be occur at high Γ, so the load-pullcan be restricted to a phase sweep around the smith chart.Fig. 14 plots the PAE vs the second harmonic power at 3dB gain compression for a series of second harmonic loadimpedances (shown in the inset). Over 70% PAE is achievedwith the second harmonic less than 30 dBc with phase anglesnear π/2. The same impedance sweep can be performed atthe third harmonic, shown in Fig. 15, which increase theefficiency to 72 % and reduce the third harmonic output toless than 36 dBc.
The load-pull amplifier results exhibits a total efficiency over70% with an output power of approximately 49 dBm asshown in Fig. 16, which plots the PAE versus the load outputpower at the fundamental. The output power is slightly lowerthan expected, which is most probably due to slight losses
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−0.6 −0.5 −0.4 −0.3 −0.2 −0.1 0 0.1 0.2 0.3
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0
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0.2
PLPAE
40
45
50
55
60
48.0
48.549.0
49.5
Higher PAE
Higher Pout
Figure 12. Output power and PAE contours for load-pullresult at 1.25 GHz. Output power approaches 50 dBm whilepeak efficiency is over 60 %.
−0.6 −0.5 −0.4 −0.3 −0.2 −0.1 0 0.1 0.2 0.3 0.4
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3335
29
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PL (2f0)PL (3f0)
Figure 13. Power contours for second and third harmonic atload. Peak efficiencies correspond to areas of low harmoniccontent.
in the feedback network. Further design optimization can beperformed to reduce these losses and enhance performance.
The final step in the amplifier design is the fabrication ofthe appropriate matching networks. In order to maintainbandwidth, as well as terminate harmonics at the correctimpedance, a stepped impedance line with shunt capacitivetuning is used. A depiction of the matching network synthesisis shown in Fig. 17a. and a 3D representation of thematching network board is shown in Fig 17b. The advantageof the stepped impedance line is easily ability to tune theperformance to account for parasitics or device variations tooptimize performance.
5. SUMMARYThis work presents the characterization and design of a 100W GaN power amplifier for advanced SAR T/R modules.Such advanced SAR systems utilize new techniques such asSweepSAR that demand high performance transmitters. Suchtechniques would use sophisticated beamforming to achievewide swaths and high resolution imagery, but would place
30 30.5 31 31.5 32 32.5 33 33.5 34 34.5 3568.5
69
69.5
70
70.5
71
PL_2f0 (dBc)
PAE
(%)
Figure 14. Efficiency as a function of second harmonicpower in dBc for swept second harmonic terminations at 3dB gain compression.
22 24 26 28 30 32 34 36 3870.6
70.8
71
71.2
71.4
71.6
71.8
72
72.2
PL_2f0 (dBc)
PAE
(%)
Figure 15. Efficiency as a function of third harmonic powerin dBc for swept third harmonic terminations at 3 dB gaincompression.
strict requirements on the RF hardware. Previous poweramplifier technologies would not suit the needs for high peakpower and efficiencies. GaN HEMT’s for space-based radarsallow for improve performance by offering high breakdown,better thermal performance, as well as high power densityperformance.
In order to optimize the design of these GaN power ampli-fiers, a mixed-signal active load-pull system is employed toperform large-signal harmonic tuning. Techniques to per-form such high-power device characterization is discussed,highlighting the challenging of high power device load-pull.This system allows for sophisticated device characterizationand is able to achieve 49 dBm output power with almost 75% efficiency. Future work aims to characterize instantaneousbandwidth performance of the typical RF chirp signal as wellas further characterization of dynamic voltage and currentwaveforms to better understand device reliability.
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39 40 41 42 43 44 45 46 47 48 4925
30
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45
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75
PL_f0_dBm
PAE
Figure 16. Measured PAE as a function of fundamental loadoutput power for high power amplifier design. PAE is over 70% with an output power approaching 49 dBm.
(a) (b)
Figure 17. (a) Matching network synthesis of steppedimpedance transformer and (b) model of matching networkboard
ACKNOWLEDGMENT
This work is supported by the NASA Earth Radar Missiontask at the Jet Propulsion Laboratory, California Institute ofTechnology, under a contract with the National Aeronauticsand Space Administration.
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BIOGRAPHY[
Tushar Thrivikraman is an RF engi-neer at the Jet Propulsion Lab at theCalifornia Institute of Technology in theRadar Science and Engineering sectionwhere he has focused on the develop-ment of RF hardware for air- and space-borne SAR systems. He received his PhDin Electrical and Computer Engineeringfrom the Georgia Institute of Technologyin 2010. His research under Dr. John
Cressler in the SiGe Devices and Circuits Research Group atGeorgia Tech focused on SiGe BiCMOS radar front-ends forextreme environment applications.
James Hoffman is a Senior Engineerin the Radar Technology DevelopmentGroup at JPL. He received his BSEEfrom the University of Buffalo, followedby MSEE and PhD from Georgia Techin planetary remote sensing. He hasworked in the design of instruments forremote sensing applications for morethan 10 years. In previous technologydevelopment tasks, he successfully de-
veloped a new low power digital chirp generator, which hasbeen integrated into several radar flight instruments. Hehas experience designing radar systems for both technologydevelopment and space flight hardware development, andis currently the RF lead for the proposed DESDynI radarinstrument.
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