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LiU-ITN-TEK-A--12/085--SE

Design and PerformanceAnalysis of Low-Noise

Amplifier with Band-PassFilter for 2.4-2.5 GHz

Muneeb Mehmood Abbasi

Mohammad Abdul Jabbar

2012-12-12

LiU-ITN-TEK-A--12/085--SE

Design and PerformanceAnalysis of Low-Noise

Amplifier with Band-PassFilter for 2.4-2.5 GHz

Examensarbete utfört i Elektroteknikvid Tekniska högskolan vid

Linköpings universitet

Muneeb Mehmood AbbasiMohammad Abdul Jabbar

Handledare Adriana SerbanExaminator Magnus Karlsson

Norrköping 2012-12-12

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© Muneeb Mehmood Abbasi, Mohammad Abdul Jabbar

Design and Performance Analysis of

Low-Noise Amplifier with Band-Pass

Filter for 2.4-2.5 GHz

Mohammad Abdul Jabbar

Muneeb Mehmood Abbasi

Supervisor: Dr. Adriana Serban

Examiner: Dr. Magnus Karlsson

Department of Science and Technology

Linköping University, SE-601 74 Norrköping, Sweden

Norrköping 2012

i

Abstract

Low power wireless electronics is becoming more popular due to durability, portability and

small dimension. Especially, electronic devices in instruments, scientific and medical (ISM)

band is convenient from the spectrum regulations and technology availability point of view. In

the communication engineering society, to make a robust transceiver is always a matter of

challenges for the better performance.

However, in this thesis work, a new approach of design and performance analysis of Low-Noise

Amplifier with Band-Pass filter is performed at 2.45 GHz under the communication electronics

research group of Institute of Science and Technology (ITN). Band-Pass Filtered Low-Noise

Amplifier is designed with lumped components and transmission lines. Performances of

different designs are compared with respect to noise figure, gain, input and output reflection

coefficient. In the design process, a single stage LNA is designed with amplifier, ATF-58143.

Maximally flat band-pass (BPF) filters were designed with lumped components and distributed

elements. Afterwards, BPF is integrated with the LNA at the front side of LNA to get a compact

Band-Pass Filtered Low-Noise Amplifier with good performance.

Advanced Design System (ADS) tool was used for design and simulation, and each design was

tuned to get the optimum value for noise figure, gain and input reflection coefficient. LNA

stand-alone gives acceptable value of noise figure and gain but the bandwidth was too wide

compared to specification. Band-Pass Filtered Low-Noise Amplifier with lumped components

gives also considerable values of noise and gain. But the gain was not so flat and the bandwidth

was also wide. Then, Band-Pass Filtered Low-Noise Amplifier was designed with transmission

lines where the optimum value of noise figure and gain was found. The gain was almost flat

over the whole band, i.e., 2.4-2.5 GHz compared to LNA stand-alone and Band-Pass Filtered

Low-Noise Amplifier designed with lumped components. It is observed that deviations of

results from schematic to layout level are considerable, i.e., electromagnetic simulation is

needed to predict the Band-Pass Filtered Low-Noise Amplifier performance.

Prototype of LNA, Band-Pass Filtered Low-Noise Amplifier with lumped and transmission

lines are made at ITN’s PCB laboratory. Due to unavailability of exact values of Murata

components and for some other technical reasons, the measured values of Band-Pass Filtered

Low-Noise Amplifier with lumped components and transmission lines are deviated compared to

predicted values from simulation.

ii

Acknowledgement

With all praises to the almighty and by His blessings we have finally completed this thesis.

We would like to express our gratitude to Dr. Magnus Karlsson who has graciously provided us

his valuable time whenever we required his assistance. His counseling, supervision and

suggestions were always encouraging and it motivated us to complete the job at hand. He will

always be regarded as a great mentor for us.

We would also like to thank Dr. Adriana Serban for her valuable comments and suggestions.

Finally the unwavering support from our loving families was an inspiration for us and we are

extremely grateful to them.

iii

Dedicated…

To parents, sisters and brothers

iv

Table of Contents

1 Introduction..................................................................................... 1

1.1 Background and Motivation ....................................................... 1

1.2 Objectives .................................................................................. 2

1.3 Outline of the Thesis .................................................................. 2

2 Theoretical Background ................................................................. 4

2.1 ISM Band ................................................................................... 4

2.1.1 ISM Band Operation ............................................................... 4

2.1.2 Application ............................................................................. 5

2.2 Radio Receiver Basics ................................................................ 5

2.3 Network Analysis ....................................................................... 6

2.3.1 Two-Port Network .................................................................. 6

2.3.2 S-Parameter ............................................................................ 6

2.4 Types of Noises .......................................................................... 7

2.4.1 Thermal Noise ........................................................................ 7

2.4.2 Shot Noise .............................................................................. 8

2.4.3 Flicker Noise ........................................................................... 8

2.5 Noise Figure ............................................................................... 8

2.6 Active Device: FET .................................................................... 9

2.7 Design Process of BFP-LNA ...................................................... 9

2.7.1 Band-Pass Filter .................................................................... 10

2.7.2 Low-Noise-Amplifier (LNA) ................................................ 14

2.7.3 Matching Network between BPF and LN A .......................... 16

3 Design of LNA ............................................................................... 18

3.1 Design Specification ................................................................. 18

3.2 Transistor Selection .................................................................. 18

3.2.1 Features ................................................................................ 18

3.2.2 Applications .......................................................................... 18

3.3 Q-Point Determination ............................................................. 19

3.4 DC Biasing Network ................................................................ 20

3.5 Design of LNA with S2P File ................................................... 20

3.5.1 Stability ................................................................................ 21

3.5.2 Using Ideal Components without Biasing Network ............... 22

3.5.3 Using non-Ideal Components without Biasing Network ........ 24

3.5.4 Using Ideal Components with Biasing Network .................... 26

3.5.5 Using non-Ideal Components with Biasing Network ............. 29

v

3.6 Design of LNA with Electrical Model ...................................... 31

3.6.1 Design with Ideal Components.............................................. 31

3.6.2 Design with non-Ideal Components ...................................... 34

3.7 Layout Design of LNA ............................................................. 36

3.7.1 Design with non-Ideal Components ...................................... 37

4 Design of BPF-LNA ...................................................................... 40

4.1 Design Specifications of BPF ................................................... 40

4.2 Design of Maximally Flat BPF ................................................. 40

4.2.1 Design with Lumped Components ........................................ 40

4.2.2 Design with Distributed Elements ......................................... 42

4.3 Design of BPF-LNA with Lumped Components ...................... 47

4.3.1 Schematic Design with Ideal Components ............................ 47

4.3.1 Layout Design ....................................................................... 49

4.4 Design of BPF-LNA with Distributed Elements ....................... 52

4.4.1 Design of Schematic ............................................................. 52

4.4.2 Design of Layout .................................................................. 54

5 Prototypes & Measurements ........................................................ 58

5.1 Prototype of LNA ..................................................................... 58

5.1.1 Measurement Results ............................................................ 59

5.2 Prototype of BPF-LNA with Lumped Elements........................ 60


5.3 Prototype of BPF-LNA with Distributed Elements ................... 63


5.4 Comparison of Layouts and Measured Results ......................... 65

6 Conclusion and Future Works ..................................................... 66

6.1 Conclusion ............................................................................... 66

6.2 Future Works............................................................................ 66

7 References ...................................................................................... 68

vi

List of Abbreviations

ISM Instruments Scientific and Medical

WLAN Wireless Local Area Network

LNA Low-Noise-Amplifier

BPF Band-Pass Filter

IMN Input Matching Network

OMN Output Matching Network

LPD433 Low Power Device, 433 MHz

PMR446 Private Mobile Radio, 446 MHz

CEPT European Conference of Postal and Telecommunications Administrations

ETSI European Telecommunications Standards Institute

ITU International Telecommunication Union

ITU-R The ITU Radio-communication Sector

FCC Federal Communications Commission

WDCT Digital Cordless Telecommunications

RFID Radio Frequency Identification

HiperLAN High Performance Radio LAN

Wi-Fi Wireless Fidelity

PCB Printed Circuit Board

PCS Personal Communications Service

WCDMA Wideband CDMA

ADS Advanced Design System

WLL Wireless Local Loop

ITN Department of Science and Technology

SMD Surface Mounted Device

CHAPTER 1

1

1 Introduction

Demand of wireless communication systems with robust transmitting and receiving

performance is growing tremendously due to the modern technology intense society. Frequency

spectrum is a natural resource as well as limited and need to be used very keenly with high

attention of distribution. Instruments Scientific and Medical (ISM) band is unlicensed and

becomes most popular because of its free uses. The engineering community is giving high

attention as well to design devices which is compatible with this band. Cordless phone, Wireless

LAN, Bluetooth, Wi-Fi all are operated in the 2.4 to 2.5 GHz.

In wireless communications, receivers need to be able to detect and amplify incoming low-

power signals without adding much noise. Therefore, to filter out the unwanted signal, a Band-

Pass filter (BPF) is placed before low noise amplifier (LNA)’s placement. A low noise amplifier

(LNA) is often used as the first stage of these receivers. To design an LNA integrated with

Band-Pass Filter (BPF), with trade-off or suitable compromise between gain and noise is always

a matter of challenge.

1.1 Background and Motivation

Thesis work is a partial requirement of Master of Science in Wireless Networks and Electronics

at Department of Science and Technology (ITN), Linköping University. In this thesis work,

integration of Band-Pass filter with LNA will be performed where; BPF will be designed by

both lumped and distributed elements. While BPF is designed with lumped components, no

need to design an input matching network (IMN) in the front side of LNA, matching network

between BPF and LNA will be fixed as IMN in front of LNA. In figure 1-1 a typical receiver

block diagram is shown where, the BPF and LNA are put in the same block i.e. BPF will be

integrated with LNA and this integrated block will be acting as a single block.

Electronic devices such as microwave oven to Bluetooth all are operated in ISM band. To keep

in mind the scarcity of electromagnetic spectrum, design of equipments in ISM band is

convenient for the engineering and technological entrepreneur as it is free of cost. However, in

CHAPTER 1

2

this thesis work, it is supposed to design and analysis the performance of band-pass filtered low-

noise amplifier (BPF-LNA) at 2.45 GHz with lumped and distributed elements.

Figure. 1-1 Block diagram of super-heterodyne receiver with combined BPF-LNA [1]

In general, BPF and LNA are different blocks in a receiver. Here it is tried to compact BPF with

LNA in a single block which would be cost effective and have less circuit complexity and the

dimension of the receiver will be reduced as well. Making a larger antenna is not cost effective

rather putting an LNA to boost up the antenna signal to compensate for the feedline losses going

from the antenna (outdoor) to the receiver (indoor). To design BPF-LNA, at first, it was needed

to choose such a transistor which gives maximum gain and minimum noise figure (NF). ATF-

58143 is selected for the whole design process.

It is highly expected that the outcome of the thesis would be highly appreciated by the industry

people due to its robustness and cost effectiveness. LNA is being used in many applications

such as ISM radio, cellular handset, GPS receiver, cordless phone, satellite communication and

wireless LAN etc.

1.2 Objectives

The main objectives of this thesis work are following:

• Literature review on BPF and LNA

• Selection of suitable substrate for BPF-LNA

• Design and simulation of all the design in Advanced Design Tools (ADS)

• Optimization of LNA and BPF-LNA

• Fabrication of prototype of LNA and BPF-LNAs and performance analysis

• Evaluation of noise figure, gain, input and output reflection coefficient

1.3 Outline of the Thesis

Chapter 1 Describes a brief idea about the thesis background and motivation

Chapter 2 Theoretical background consists of literature review

Chapter 3 Design of LNA with ATF-58143is described in details

Chapter 4 Design of BPF-LNA with the maximally flat BPF is depicted elaborately.

CHAPTER 1

3

Chapter 5 Fabrication process and comparison of results of BPF-LNAs are shown

Chapter 6 Concludes the thesis works and expectation of future works within this topic

CHAPTER 2

4

2 Theoretical Background

To have a better understanding and supporting of the thesis work, a theoretical background

literature is included in this part. Relevant theories are described briefly.

2.1 ISM Band

The ISM radio band is radio band (a small portion of radio spectrum) which is reserved

internationally for the use of radio frequency (RF) energy for the purpose of industrial, scientific

and medical equipments other than communications [2]. In general, communications equipment

operating in these bands must have to tolerate any interference generated by the ISM

equipments and for the case of ISM device operation, users have no regulatory protection. In

spite of the intention of the original allocation, the uses of these bands become very popular for

short-range communication and low power communication electronics systems.

2.1.1 ISM Band Operation

ITU-R has defined the ISM bands in 5.138, 5.150, and 5.280 of the radio regulations [3]. Due to

the national radio regulations of spectrum management, individual countries' use of the bands

designated in these sections may differ. Some communication devices which are using the ISM

bands, it must tolerate any interference from ISM equipments. Normally unlicensed operations

are allowed to use these bands, because the unlicensed operations are supposed to tolerate any

external or internal interference from other devices. However, the ISM bands do have the

licensed operations. Because of high possibilities of harmful interferences, licensed use of the

ISM bands is not high. By the part 18 of the Federal Communications Commission (FCC), uses

of ISM bands are being governed in USA, at the same time, part 15 contains the rules and

regulations for unlicensed communication devices even though those use the ISM frequency

bands [4].

According to European commission’s short range device regulations, the use of the ISM band is

being governed in Europe [5]. In most of the European zones, for license-free voice

communication, LPD433 band is allowed using analog frequency modulation [6].

CHAPTER 2

5

2.1.2 Application

Microwave oven is one of most common examples of ISM device which operates at 2.45 GHz.

Lately ISM bands have been shared with license-free communications applications for example

915 MHz and 2.450 GHz are for wireless sensor networks. 915 MHz, 2.450 GHz and 5.800

GHz are for wireless LNA and cordless phones respectively [3]. In radio frequency

identification (RFID) applications such as biometric and contactless smart cards, ISM bands are

being used widely [3].

Some low power remote control toys, gas powered cars and miniature aircraft use 2.4 GHz band

range. Worldwide Digital Cordless Telecommunications (WDCT) is an ISM band technology

which uses the 2.4 GHz radio spectrum. Wireless LAN devices use the following bands [3]:

• Bluetooth 2450 MHz band

• HIPERLAN 5800 MHz band

• IEEE 802.11/Wi-Fi 2450 MHz and 5800 MHz bands

2.2 Radio Receiver Basics

The super-heterodyne receiver is one of the most popular forms of receiver which is widely

used today in a variety of applications from broadcast receivers to two way radio

communications links as well as many mobile radio communications systems [1]. At the early

stage of radio communication technology development, the super-heterodyne receiver offers

many advantages in many applications.

Figure. 2-1 Block diagram of super-heterodyne receiver [1]

In this section, a typical block diagram (figure 2-1) of wireless receiver is drawn. According to

this figure, the typical functionalities will be described shortly. The basic function of receiver is

to recover the transmitted baseband signal by the reversing the functions of transmitter. An

important component of receiver is antenna which receives the radiated electromagnetic waves

from some other sources of broad frequency ranges [1]. Then the signal passes through a band-

pass filter which provides some selectivity by filtering out received signals with unwanted

CHAPTER 2

6

frequencies and passing some signals of desired frequency band. The desired signal from BPF

will pass through a low-noise-amplifier (LNA). The basic function of LNA is to amplify the

very weak received signal at the same time to minimize the noise power which is added to the

received signals [1]. By putting a BPF in before LNA reduces the possibilities to add other

interfering signals to the desired signal, this is how, the amplifier cannot be overloaded with

other high power signals. The output from LNA is feed to a mixer which is used to down-

convert the received radio signal to a lower frequency signal. A local oscillator (LO) is set at the

level of the frequency which is near to the RF input and the output of the mixer will be

relatively low and it could be filtered out by the IF band-pass filter [1]. The high gain IF

amplifier raises the power level of the filtered signal thus the baseband information can be

recovered without distortion [1].

2.3 Network Analysis

In this section, two-port network and S-parameter will be discussed briefly.

2.3.1 Two-Port Network

A two-port network is an electrical circuit which consists of four terminals to be connected with

other external network or circuit [7]. It is represented by four variables such as at the input port

voltage, �� current, �� and at the output port voltage, �� and current, �� [8]. Figure 2-3 shows a

two-port network which has four terminals.

Figure. 2- 2 Two-port scattering network with source and load [9]

2.3.2 S-Parameter

Scattering parameters or S-parameters have significant role in RF system design. RF engineers

use S-parameter to define the relationship between input-output of an electrical network in

terms of incident and reflected power waves [10]. According to figure 2-2, an incident

normalized power wave, �� and a reflected normalized power wave, ��

The mathematical expression for incident and reflected normalized power wave can be written

as:

�� = ��

�� + �� (1)

CHAPTER 2

7

�� = ��

�� − �� (2)

Where,

� = Port 1or 2

�� = Characteristics impedance of the connectinglines [10]

Four waves such as ��, �� ,�� and �� are related through following equations (3) and (4) where

�� , �� , �� and �� are the S-parameters of the above network [10]

�� = �� + �� (3)

�� = �� + �� (4)

Combining equation (3) and (4), the matrix form is as follows:

��

� = �� (5)

Where,

�� = Input reflection coefficient

�� = Input reflection coefficient

�� = Forward voltage gain

�� = Reversed voltage gain

2.4 Types of Noises

Noise is an undesired random disturbance in the communication systems which can degrade the

useful signal [f]. It comes from natural or man-made sources. For wireless system performance

evaluation, noise is an important factor to be taken into account. Normally, noise exists in all

radio frequency (RF) and microwave systems. Receiver performances can be limited by the

noises effect [1]. There are several parameters such as signal-to-noise ratio; dynamic range, bit

error rates and minimum detectable signal level all are directly dependent on the noise effect

[1]. In the following sections, some noises of electronics devices are discussed briefly:

2.4.1 Thermal Noise

Due to random thermal motions of electrons inside electronics devices generate some noises

which are called thermal noise. Thermal noise is also called as Johnson–Nyquist noise [11].

Throughout the whole spectrum, the power spectral density is almost equal. The amplitude of

the signal is very close to the Gaussian probability density function [11].

The electrons in a resistor are in a random motion, with a kinetic energy which is proportional

to the temperature; T. Due to these random motions of these electrons, small random voltage

fluctuations is produced across the terminal of the resistor. Calculations shows, the mean value

of this produced voltage is zero but r.m.s. value is not zero, which can be calculated using the

following equation through a narrow frequency bandwidth, B [1].

CHAPTER 2

8

V� = �4kTBR (6)

Where,

k = 1.380x10)�* J/K (Boltzmann’s constant)

T = Temperature, degree Kelvin (°K)

B = Bandwidth, Hz

R = Resistance, �

2.4.2 Shot Noise

Due to thermal fluctuations of stationary charge carriers, a different type of noise is generated

which is called shot noise. In case of higher frequencies and low level temperature, shot noise

behaves as the dominant source of electronic noise [12]. Shot noise follows Poisson distribution,

and the r.m.s. value of current fluctuations can be modeled by the following equation [13]:

+, = 2.�∆0 (7)

Where,

. = Charge of an electron

� = DC current flowing

∆0 = Bandwidth

2.4.3 Flicker Noise

Flicker noise or pink noise is inversely proportional to the frequency. At higher frequencies the

noise is not considerable but at low frequency, it is troublesome [14]. Because of the imperfect

contacts between conductors and semiconductor, this type of noise is generated inside

electronics devices [15, 16].

This noise can be expressed by the following mathematical equation: [17]

�1�� = 23456�7

(8)

Where,

8�9 = Oxide capacitance per unit length

: = Process dependent constant

; = Channel width

< = Channel length

0 = Frequency

2.5 Noise Figure

Noise figure (NF) is one of the most important parameters to evaluate the radio performance of

communication system. It is a measurement of degradation of signal-to-noise ratio (SNR)

between the input and output of the component [1].

CHAPTER 2

9

When the network is noisy, the output noise power is greater than the output signal power; this

is how, output SNR will be decreased because of high output noise power. Once the noise and

desired signal are applied to the input of a noiseless network, may be both the noise and signal

will be amplified or attenuated by the same degree, that’s why, SNR will not be changed [1].

The noise figure (NF) can be calculated using the following mathematical equation:

=> = ?@/B@?�/B�

= ?BC@?BC�

≥ 1 (9)

=>�EF� = 10 log => (10)

Where,

�, = Input signal power

=, = Input noise power

�� = Output signal power

=� = Output noise power

Using the following Friis equation, noise figure (NF) of LNA of a receiver can be obtained:

=>J�J = 1 + �=>� − 1� + �BKL)��MN

+ �BKO)��MNML

+ ⋯ + �BKQ)��MNML…M@SN

(11)

Where,

T, = Gain of each stage

=>U = Noise figure of each stage

From Friis equation (equation no. 11), it is understandable that the total noise figure �=>J�J� is

dominated by the noise figure of first stage, =>� which is the noise figure of the low-noise-

amplifier (LNA). Simultaneously the gain of the first stage T� reduces the noise in the

consecutive stages [18].

2.6 Active Device: FET

Amplification is one of the most critical functions in all the wireless receivers and transmitters.

Engineers pay high attention for designing the semiconductor transistor to get the acceptable

value of amplification. Today, microwave and RF amplifiers commonly use three-terminal

solid–state devices such as silicon or silicon germanium (SiGe) bipolar transistors, gallium

arsenide (GaAs) field effect transistors (FETs) and high electron mobility transistors (HEMTs)

etc [1]. RF and Microwave transistors are used as amplifiers which are low-cost, reliable and

can be easily integrated due to high gain and low noise figure in the millimeter wave range [1].

2.7 Design Process of BFP-LNA

Band-pass filter and low-noise-amplifier have to be designed individually. Once these two

blocks are designed, integration of these two blocks make a single module named BPF-LNA.

CHAPTER 2

10

The following figure 2-4 shows the complete block of BPF-LNA where two blocks (BPF and

LNA) are connected through a matching network of lumped or transmission lines.

��

��

��

��

��

Figure. 2-3 Complete BPF-LNA block diagram

2.7.1 Band-Pass Filter

In RF transmitter and receiver, filters are key components which is used to selectivity pass or

reject signals based on frequency. Generally, there are four types of filters such as low-pass,

high-pass, band-pass and band-stop filter. Combination of high-pass filter and low-pass filter

make a band-pass filter (BPF) which is used to reject unwanted frequency bands and pass a

narrow pass-band [1].

Normally, a pre-select BPF is setup in front of the first RF amplifier to the RF tuning range of

the receiver (see figure 2-1). To make noise figure as less as possible, the filter should have low

insertion loss (IL) as a result the cut-off characteristics of the filter will not be very sharp [1].

There are several classes of band-pass filter such as Butterworth or maximally flat, Chebyshev

and elliptical BPF. BPF can be designed in some ways like using lumped components and

distributed components. In this thesis work, maximally flat BPF is considered to design with

lumped and distributed components. More details can be found in the chapter-4. Some

parameters need to keep in mind during design of filters such as:

• Insertion Loss: An ideal filter has zero insertion loss (IL) when it is integrated in to the

RF circuitry as it does not introduce any power loss in the pass-band. But in practical a

filter has some power loss in the pass-band. 0 dB line shows how much power is

deviated which is quantified as insertion loss .It can be stated as the following

mathematical equation: [10]

�< = 10 VWX YZ@[Z\

] = −10 VWX�1 − ^_,�^�� (12)

Where,

Pa = Power delivered to the load

Pb� = Input power from the source

^�b�^ = Reflection coefficient looking towards the filter [10]

CHAPTER 2

11

• Ripple: In a band-pass filter, flatness is highly desired and it can be achieved by

controlling the ripple. The less difference between maximum and minimum of the

amplitude of the pass-band will provide more flat band filter. Design of Chebyshev is a

better way to control the magnitude of the ripple in the pass band [10].

• Bandwidth: In case of a band-pass filter, the difference between upper and lower

frequencies is defined as the bandwidth which is measured at the 3 dB attenuation. The

value of the bandwidth can be written by the following expression [10]:

F;* cd = 0e * cd − 05 * cd (13)

Where,

F;* cd = Bandwidth

0e * cd = Upper Frequency

05 * cd = Lower Frequency

• Shape Factor: Sharpness is a highly expected factor in the filter design. The following

factor depicts the sharpness of the band-pass filter which is calculated using the ratio of

bandwidths at 60 dB and 3 dB [10].

�> = d4 fg hid4 O hi = 3j fg hi)3\ fg hi

3j O hi)3\ O hi (14)

Where,

�> = Shape factor

F; kl cd = Bandwidth at 60 dB attenuation and

F; * cd = Bandwidth at 3 dB attenuation

• Rejection: Infinite number of components makes filter ideal, but its circuit becomes

more complex which is not practically convenient. That is why, in practical, finite

number of components are used to design filters which is mostly specified 60 dB as the

rejection rate [10].

However, it is not practically possible to make high performance band-pass filter in the

integrated circuit form. Due to inherent losses of RF and microwave integrated circuits, filter

experiences high insertion losses and low attenuation rates in out-band. Now-a-days, in most of

the devices, off-chip filter is being used which is optimized for better performance but at the

same time it is costly [1].

2.7.1.1 Lumped-Components Filter

Generally, the filters which are designed by lumped components (inductor, capacitor) are called

lumped components filters. Lumped components are considered to design the filters when it is

needed to reduce the dimension of the filter and if the assigned frequency band is low [10]. At

high frequency, filter design with lumped components become less ideal [19]. There are some

problems to design filter at higher frequencies, for example, the wavelengths become equal to

the dimensions of the lumped components which causes of different types of losses and

degradation of performances [10]. The terms "tee" and "pi" are used to describe lumped element

CHAPTER 2

12

filters, and other networks. A tee element starts with a series element, while a pi network starts

with a shunt element as shown below [19].

�

�

�

�

�

�

�

�

�

�

�

��

�

(a) (b)

Figure. 2-4 Network topology a. Pi Network low-pass filter b. Tee network high-pass filter [19]�

�

The following figures represent band-pass filters of Tee and Pi Networks of order 3

�

Figure. 2-3 Band-pass filter with Tee networks with order 3 [19]

�

Figure 2-5 Band-pass filter with Pi networks with order 3 [19]

CHAPTER 2

13

2.7.1.2 Distributed-Elements Filter

A distributed element filter is an electronic filter which contains capacitance, inductance and

resistance interns of transmission lines instead of conventional discrete circuit elements. The

functionalities of this distributed element filters are same as conventional one [20]. To design

RF and microwave circuit at higher frequency using distributed elements is convenient rather

lumped elements. At high frequency, to design with lumped components have some losses

because of deviation in behaviour [21].

There are two ways to convert lumped components to distributed components such as Richard

transform and Kuroda’s identity [22]. These two methods consider mn transmission lines. To

form a lumped component from a transmission line, the width of microstrip line ( �l ) is used

[23]. There are several ways to design distributed elements filters such stub filters and coupled

lines.

In this thesis work, stub filter is designed and implemented. Stub filter is implemented by using

quarter wave ( mo ) transmission lines which is connected to the quarter wave (

mo ) stubs [24]. A

stub behaves like a capacitor or an inductor over a narrow band and in case of wide range of

frequencies it shows resonance properties. The impedance of the stub can be found by its length

[25].

(a)

(b)

Figure. 2-6 Quarter wave stub resonator [22] (a) equivalent circuit of short-circuit (b) equivalent

circuit of open-circuit

In figure 2-6 (a), quarter wave stub resonator of equivalent short-circuit and in figure 2-6 (b)

quarter wave stub resonator of open-circuit are designed respectively. According to RF

CHAPTER 2

14

principles, short-circuit quarter-wavelength stub works as shunt LC anti-resonators and open-

circuit quarter-wavelength stub works as series LC resonator. To build complex filters, stubs

can be used in combination with impedance transformers which could be most useful in case of

band-pass applications [25]

Figure. 2-7 Band-pass filter using quarter wave transmission lines and short-circuit stubs [24]

In figure 2-7, a band-pass filter is shown using ( mo ) transmission lines and quarter wave (

mo )

short-circuit stubs. The short circuit stubs are used to pass the required frequency signal through

the transmission lines by behaving as an open circuit at the joint of transmission line and stub.

And short-circuit stubs behave as a short circuit at the joint of the transmission line and stubs for

all other out of band frequencies

2.7.2 Low-Noise-Amplifier (LNA)

Low-noise-amplifier (LNA) is one of the most important key components of the communication

system. It is used in the input stage of the receiver. It deals with two important parameters such

as gain (in dB) and the noise figure [1]. In a few words, the purpose of the LNA is to amplify

the received signal to acceptable levels while minimizing the noise which is added from the

channel.

According to Friss equation (equation no. 11), it is very important for RF and microwave

engineers to design RF receiver with low noise at the input stage. Once the signal is received by

the antenna, passing through the BPF and LNA, it is not possible to get the high gain and low

noise at the same time. That’s why, it is important to consider a trade-off between gain and

noise figure [26].

2.7.2.1 Design Specification

Before going to design BPF-LNA, design specification should be made properly. The following

things should be given attention such as bandwidth and central frequency for measurements,

noise figure (NF), gain, transistor model, Q-point, source impedance, load impedance, matching

network.

CHAPTER 2

15

2.7.2.2 Transistor

In order to make an LNA, the choice of transistor is critical. This is one of the most important

steps in designing a low-noise-amplifier (LNA). Different types of transistors are available for

LNA applications. According to specifications, appropriate transistor should be selected for

low-noise-amplifier due to its low noise figure and high gain [27]. The numbers of transistors

are limited at the interested frequency. In this thesis work, ATF58413 is chosen.

2.7.2.3 Stability Analysis

Stability test is one of the most important tasks to verify whether the amplifier is stabled or not.

Due to improper stability, an RF circuit approaches to be oscillated. To verify the stability of a

transistor, Rollet’s conditions are used such as [10]:

: = �)^?NN^L)^?LL^Lp^∆^L� ^?NL^^?LN^ > 1 (15)

∆ = ^��^^��^ − ^��^^��^ (16)

If : > 1 and ^∆^ < 1 then the amplifier is stabled throughout the selected frequency band and

bias conditions.

By putting a shunt conductance or a series resistance either at input port or output port, an

amplifier can be stabilized. It is recommended not to put a resistive element at the input side as

it causes additional noises to be amplified. After stabilization through adding resistors, may be

gain can be low or noise figure increases so it’s a trade off [10].

2.7.2.4 Q-Point Selection

The operating point of a device is known as Q-point, which is the steady-state operating

condition of an active device without applying any input signal. Here, at first a suitable Q-point

needs to be found for correct biasing of the transistor throughout the entire bandwidth.

2.7.2.5 DC Biasing Network

Biasing is a process of setting up the bias point at the middle of the DC load line applying drain

voltage and current [27]. In a field-effect transistor (FET), bias is the DC voltage supplied from

a battery which is applied at the drain. According to the selected Q-point, the biasing circuit is

designed to operate the transistor at that Q-point.

2.7.2.6 Input and Output Matching Networks

Matching networks is one of the important steps to design LNA. Impedance matching is used to

minimize the reflections and obtain an acceptable amount of noise figure and maximum gain by

making the load impedance equal to the source impedance [22].To get an optimal value of input

reflection coefficient, gain and noise figure (NF); input matching network is tuned and for

output reflection coefficient; output matching network (OMN) is tuned. The following figure (2-

7) shows a general transistor amplifier circuit where IMN and OMN are designed with the

transistor.

CHAPTER 2

16

Figure. 2-8 A general transistor amplifier circuit [1]

Generally it is not possible to obtain both minimum noise figure and maximum gain for an

amplifier. So, some sort of compromise must be made. This can be done by using constant gain

circles and circle of constant noise figure to select a usable trade-off (check it from book, trade

off: up-down or compromise: linear) between noise figure and gain [1]. IMN and OMN can be

designed by lumped and distributed components. More details will be discussed in Chapter: 3.

Four parameters are considered to check the design of LNA such as gain (��), noise figure

(NF) and input reflection coefficient (��)

2.7.3 Matching Network between BPF and LNA

Integration of band-pass filter (BPF) and low-noise-amplifier (LNA) can be performed using

matching network which is shown in the figure 2-4. This matching network can be designed

using lumped elements or quarter-wave transmission lines.

2.7.3.1 Matching Network with Lumped Components

There are different topologies of matching networks which can be designed by lumped elements

such as T-networks, Pi-network and L-network. In this thesis work, T-network is used as

connector between BPF and LNA.

(a) (b)

Figure. 2-9 Topology a. Pi network low-pass filter b. Tee network high-pass filter [19]�

2.7.3.2 Matching Network with Distributed Elements

The connection between BPF and LNA can be made using quarter transmission line as well. In

this case, IMN of LNA is removed and matching network by quarter-wave transmission line is

placed. If the impedance of BPF, �,� and impedance of LNA, �5 are known, these two values

can be used to calculate the characteristics impedance, �l of the of quarter-wave transmission

line.

CHAPTER 2

17

Figure.2-10.Input and load impedance matched through mo line [10]

�l can be determined using the following equation [10]:

�l = �5�,� (17)

Where,

�l = Characteristic impedance of the line

�,� = Impedance from BPF

�5 = Impedance from LNA

Once, �l is calculated, afterwards using Agilent ADS’s line calculation option, corresponding

height and width of the transmission line can be found as well.

CHAPTER 3

18

3 Design of LNA

In this chapter, the design procedure of LNA is described step by step in the following sub-

sections. The operating frequency of the design is 2.45 GHz. The design is simulated and

optimized in Advanced Design System (ADS)

3.1 Design Specification

The design specifications for the low noise amplifier are as follows:

• Gain > 15.5 dB

• Noise Figure < 0.55 dB

• Used lumped components for – matching networks

• Bandwidth: 100 MHz from 2.4 GHz to 2.5 GHz

3.2 Transistor Selection

The AVAGO Technologies’ ATF58143 is chosen for designing BPF-LNA due to its following

features.

3.2.1 Features

There are some mentionable features of ATF58143 such as [28]

• Low noise and high linearity performance

• Enhancement Mode Technology

• Excellent uniformity in product specifications

• Low cost surface mount small plastic package SOT-343 in Tape-and-Reel packing

option available

• Lead-free option available

3.2.2 Applications

Applications of AVAGO Technologies’ ATF-58143 are following [28]:

CHAPTER 3

19

• Cellular /PCS/WCDMA base stations

• Pre-driver amplifier for 3-4 GHz WLL

• Low noise and high linearity application at 450 MHz to 6 GHz.

3.3 Q-Point Determination

The following circuit (figure 3-1) is setup for I-V characteristics simulation in Advanced Design

System (ADS). Figure 3-2 shows different I-V curves with respect to different Vst

Figure. 3-1 I-V characteristics simulation setup in ADS

Figure. 3-2 I-V curves of ATF-58143

For this thesis work, such a Q point is chosen according to data sheet in which it is possible to

get the minimum noise figure at 2.45 GHz which is the central frequency. In the data sheet, it is

S1

G

D

S2

ATF58143_ADS_model

IDS

SRC3

X2

SRC4

Vdc=VDS

Vdc=VGS

1 2 3 40 5

20

40

0

60

VGS=0.450

VGS=0.472

VGS=0.494

VGS=0.516

VGS=0.538

VGS=0.550

Vds (V)

Ids (

mA

)

CHAPTER 3

20

seen that, at this Q point (Vut = 3 V, Iut = 30 mA), the minimum Noise Figure �NF{b� � and

the Gain are 0.55 dB and 16.5 dB respectively.

3.4 DC Biasing Network

Figure 3-3 shows the setup for the desired Q point at Vst = 0.516 V.

Figure. 3-3 DC biasing network setup in ADS

In order to get the Q point, the above circuit is designed. At drain, it is needed to have Iut = 30

mA. In the above circuit setup, it is 30.3 mA which is very close to desired one. The drain

voltage, according to Q point, is, Vut = 3 V, so here, by the above setup, exactly it is found 3 V.

In order to achieve these specification (i.e. drain current, Iut = 30 mA, and Vut = 3 V at Vst =

0.5 V). There are three resistors used such as R1, R2 and R3. R1 and R2 are used for voltage

divider and by changing their values, getting the gate voltage, Vst = 0.51 V

3.5 Design of LNA with S2P File

In this section, LNA is designed using S2P file with ideal and no-ideal components with and

without biasing networks.

515 mV

3.30 V3.30 V

3.00 V

Gate

Source

Drain

95.4 uA

DC

S1

G

D

S2

-15.2 mA

0 A

30.3 mA

-15.2 mA

VarEqn

30.4 mA

-30.4 mA

-30.4 mA95.4 uA

VAR

DC

R

I_Probe

ATF58143_ADS_model

R

V_DC

R

VAR1

DC1

R2

IDS

X1

R1

R3

SRC1

R=26 kOhm

Vdc=VDD

R=10 Ohm

VDD =3.3 V

R=5.4 kOhm

CHAPTER 3

21

3.5.1 Stability

S2P file is used to check the stability of the transistor. First the S2P file is run alone and it is

found that in the whole bandwidth (BW) i.e. from 2.4 GHz to 2.5 GHz, the transistor is

unstable, because the value of stability factor is : < 1

Figure. 3-4 Schematic for stability test

In order to make it stabled, a series resistor is connected in front of the drain and by changing its

different values it was found that stability factor becomes, : > 1only in the central frequency

(at 2.45 GHz). To get stability factor : > 1 in the whole bandwidth, a shunt resistor (R1) is

connected to the drain as shown in the figure 3-4.

Figure. 3-5 Transistor stability test

When a 100 � shunt resistor is connected to the drain, the value of stability factor, : > 1

through the whole bandwidth as shown in figure 3-5. Along to x-axis frequency and along to y-

axis stability factor are plotted.

Drain

Source

Gate

2

1

111 1

1

2

1

2

Ref

1 21 2

3 S2P

SNP1

Term2Term1

R1

R=100 Ohm

Z=Z0 OhmZ=Z0 Ohm

2.2 2.4 2.6 2.82.0 3.0

1.0

1.1

0.9

1.2

Frequency (GHz)

Sta

bili

ty F

acto

r (K

)

CHAPTER 3

22

3.5.2 Using Ideal Components without Biasing Network

In the following circuit, input matching network (IMN) and output matching network (OMN)

are designed by using Smith chart tool in ADS.

Figure. 3-6 Schematic with ideal components without biasing network

According to this figure, several parameters will be discussed such as forward voltage gain

��, noise figure (NF), input reflection coefficient, ��. The circuit is optimized in order to

get required noise figure (NF) and power gain.

Figure. 3-7 Simulation result of noise figure (NF)

In figure 3-7, along to x-axis frequency and along to y-axis noise figure are plotted. Noise figure

is found as 0.57 dB and minimum noise figure (NF) can be achieved 0.55 dB at the central

frequency 2.45 GHz. But if this amount of noise figure is achieved, by changing input matching

Gate

Source

Drain

2

1

11 1 11 1

1

2

1

2

Ref

1 2

1 2

3

2

1

2

1

1 21 2

S2PL1 L2

C2

C1

SNP1

Term1 Term2

R1

C=1.216 pF

R=

L=2.1 nH L=1 nH

C=1.08 pF

Z=Z0 Ohm Z=Z0 Ohm

R=100 Ohm

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

1

2

3

4

5

0

6

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

CHAPTER 3

23

network, the value of input reflection coefficient, �� goes higher than -6 dB which is

undesirable.

Figure. 3-8 Simulation result of forward voltage gain

In figures3-8 and 3-9 along to x-axis frequency and along to y-axis input reflection coefficient

and forward voltage gain are plotted respectively. Here, forward voltage gain is 14.6 dB and it

can be achieved 17 dB at the central frequency 2.45 GHz but if forward voltage gain is

increased by changing the IMN and OMN, noise figure also increases. To design LNA, main

concern is to get the acceptable value of noise figure and forward voltage gain, ��

Figure. 3-9 Simulation result of input reflection coefficient

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

0

5

10

15

-5

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-8

-6

-4

-2

-10

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 3

24

The value of input reflection coefficient can be changed by changing the value of IMN’s

components. The value of input reflection coefficient of figure 3-9 can be decreased but if the

value decreases, then noise figure increases. So, there is see a trade-off between noise figure and

input reflection coefficient. Practically, its value should be less than -6 dB, so here, the value is

achieved which is less than -6 dB.

3.5.3 Using non-Ideal Components without Biasing Network

The following circuit is designed with non-ideal components without biasing network. The

schematic was simulated and found the following responses of input reflection coefficient, noise

figure and forward voltage gain. Table 1 shows the list of components used in figure 3-10.

Figure. 3-10 Schematic with non-ideal components without biasing network

Table 1 List of components

Resistor (�) Capacitor (pF) Inductor (nH)

R1 = 100 C1 = 1.2 L1 = 2.2

-- C2 = 1.1 L2 = 1.2

Gate

Source

Drain

2

1

11 1 11 1

1

2

1

2

Ref

1 2

1 2

3

1 21 2

2

1

2

1

S2P

C2C1

L1 L2

SNP1Term1Term2

R1Z=Z0 Ohm

Z=Z0 Ohm

CHAPTER 3

25


Non-ideal components deviates the result from the ideal components because there are some

parasitic effects involved in non-ideal components. In figure 3-11, the noise figure is found as

(NF) 0.578 dB and it can be achieved as 0.56 dB which is the minimum noise figure at central

frequency 2.45 GHz. If the desired amount of noise figure is achieved by changing the input

matching network, the input reflection coefficient can be high.


1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

2

4

6

8

10

0

12

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-10

-5

0

5

10

15

-15

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

CHAPTER 3

26

In figure 3-12 it is found that forward voltage gain is 14.58 dB and it can be achieved 16.70 dB

at the central frequency 2.45 GHz but if this forward voltage gain is increased by changing the

IMN and OMN, noise figure (NF) will be increased also. The non-ideal components gain

slightly decreases as compared to ideal components and the reason is parasitic effects in non-

ideal components.

Figure. 3-13. Simulation result of input reflection coefficient

Input reflection coefficient can be varied with the change of the value of IMN’s components.

There is a trade-off between noise figure (NF) and input reflection coefficient. In practical,

value of �� should be less than -6 dB. Here, it is found that the input reflection coefficient is -

9.27 dB which is acceptable.

3.5.4 Using Ideal Components with Biasing Network

The following circuit of LNA is designed with ideal components and biasing network is added

as well. After simulation of the schematic, the following responses of input reflection

coefficient, noise figure and forward voltage gain are found which are described briefly.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-8

-6

-4

-2

-10

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 3

27

Figure. 3-14 Schematic with ideal components with biasing network

Figure. 3-15 Simulation results of noise figure (NF)

It is found that there is a very small difference in results between with and without biasing

design. Because S2P file has already saved data for AC signal and applying DC voltage cannot

change its results. In figure 3-15, the noise figure is 0.62 dB and it can be achieved 0.59 dB

which is the minimum noise figure (NF) at the central frequency 2.45 GHz.

Drain

Source

Gate 2

1

11 1

1

2

1

2

Ref

1 21 2

3

11 2

1

1 2

2

1

1

1

1 2 1

1

2

1

2

1

2

2 1

2

1

21 2 11 2

C

L S2P C4

C2

C1

R2

L4

R3

R1

L3

L1

SRC1

L2

SNP1

Term2Term1 R4 C3

C=8.2 pFC=8.06 pFL=1 nH

R=26 kOhm

L=5.6 nH

R=

R=10 Ohm

R=5.4 kOhm

L=2.525 nH

R=

R=

C=1.244 pF

Vdc=3.3 V

R=

Z=Z0 OhmZ=Z0 Ohm R=100 Ohm C=1.08 pF

L=2.1 nH

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

1

2

3

4

5

0

6

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

CHAPTER 3

28


In figure 3-16 the forward voltage gain, �� is 14.99 dB and it can be achieved 16.72 dB at

central frequency 2.45 GHz but once the forward voltage gain is increased, the noise figure

(NF) will also be increased. Gain has also no effect of DC biasing.


1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

0

5

10

15

-5

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-8

-6

-4

-2

-10

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 3

29

There is a trade-off between noise figure (NF) and input reflection coefficient. In practical,

value of S (11) should be less than -6 dB. In figure 3-17, the value of input reflection coefficient

is -9.687 dB

3.5.5 Using non-Ideal Components with Biasing Network

The following circuit is designed with non-ideal components with biasing network. After

simulation the schematic, the following responses of input reflection coefficient, noise figure

and forward voltage gain are found.

Figure. 3-18 Schematic with non-ideal components with biasing network


Resistor Capacitor (pF) Inductor (nH)

R1 = 5.4 k� C1 = 8.0 L1 = 2.2

R2 = 26 k� C2 = 1.2 L2 = 2.7

R3 = 10 � C3 = 1.1 L3 = 5.6

R4 = 100 � C4 = 8.2 L4 = 1.2

Gate

Source

Drain

2

1

2 12 1

2

1

11 1

1

1

11 1

1

2

1

2

Ref

1 21 2

3

1

2

1

2

1

2

1 221 1 2

2

1

2 1

2

1S2P

C3

C4

C2

L1C1

L4

SRC1

L3L2

SNP1

Term1Term2

R3

R1 R2

R4

Vdc=3.3 V

Z=Z0 OhmZ=Z0 Ohm

CHAPTER 3

30


In figure 3-19, along to x-axis frequency and along to y-axis noise figures are plotted. Noise

figure (NF) is 0.618 dB and minimum noise figure is 0.602 at the central frequency 2.45 GHz.


In figure 3-20 along to x-axis frequency and along to y-axis forward voltage gain are plotted.

Here, forward voltage gain, �� is 14.813 dB and it can be achieved up-to 16.42 dB at central

frequency 2.45 GHz but when the forward voltage gain increases, noise figure (NF) also

increases.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

2

4

6

8

0

10

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-10

-5

0

5

10

15

-15

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

CHAPTER 3

31


When noise figure (NF) increases, input reflection coefficient decreases, so there is a trade-off

between noise figure (NF) and input reflection coefficient. In the case of LNA design, value of

�� should be less than -6 dB. In figure, 3-21, the value of input reflection coefficient is -8.706

dB.

3.6 Design of LNA with Electrical Model

In this section, the LNA is designed with electrical model; but Electrical model does not explain

the results in all the frequencies as compared to S2P file. So for layout design S2P file was used

to design LNA. Biasing network is designed by using electrical model.

3.6.1 Design with Ideal Components

The following circuit is designed with ideal components. After simulation the schematic,

following responses of input reflection coefficient (��), noise figure (NF) and forward voltage

gain (��) are observed.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-8

-6

-4

-2

-10

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 3

32

Figure. 3-22 Schematic with ideal components


In figure 3-23, along to x-axis, frequency and along to y-axis noise figure are plotted. The noise

figure (NF) is 0.567 dB and it can be obtained up-to 0.545 dB at central frequency 2.45 GHz. In

order to achieve the desired amount of NF by changing input matching network, the value of

input reflection coefficient will go high

S1

G

D

S2

2

1

3

41

2

1

2

1

1

1 1

1

21

2 12 11

1

2

1

2

2

1

1

1 2

2

1

1

2

1 1 2

2

1

21

2 1

1

21

ATF58143_ADS_modelC3

C1

R4

C2C5L4

R3

L2

L3

R1 R2

SRC1

Term2Term1

X1C4

L1

C=3 pF

C=1.2 pF

C=8.2 pF

R=100 Ohm

C=8.2 pFL=1.136 nH

R=

L=1.552 nHR=

R=10 Ohm

R=

L=2.775 nH

R=L=5.6 nH {t}

R=5.4 kOhm R=26 kOhm

Vdc=3.3 V

Z=50 OhmZ=50 Ohm C=1.71 pF

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

2

4

6

0

8

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

CHAPTER 3

33


In figure 3-24, forward voltage gain is found as 13.50 dB and it can be achieved 14.57 dB at

central frequency 2.45 GHz but if this forward voltage gain increases by changing the IMN and

OMN, noise figure also increases. As the aim is to design LNA, the main target is to get

minimum noise figure and required forward voltage gain, ��


In figure 2-25, along to x-axis frequency and along to y-axis input reflection coefficient are

plotted. If the value of input reflection coefficient decreases, then noise figure (NF) increases.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

0

5

10

15

-5

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-12

-10

-8

-6

-4

-2

-14

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 3

34

So, we see a trade-off between noise figure and input reflection coefficient. Expected value of

input reflection coefficient is less than -6 dB and here, it is achieved less than -6 dB.

3.6.2 Design with non-Ideal Components

The following circuit is designed with non-ideal components. After simulation of the schematic,

the following responses of input reflection coefficient, noise figure and forward voltage gain are

found which are described briefly.

Figure. 3-26 Schematic with non-ideal components



R1 = 5.4 k� C1 = 8.2 L1 = 1.5

R2 = 26 k� C2 = 1.6 L2 = 2.7

R3 = 10 � C3 = 3.0 L3 = 5.6

R4 = 100 � C4 = 1.2 L4 = 1.2

-- C5 = 8.2 --

S1

G

D

S2

2

1

3

41

2

1

2

1

1

1 1

1

21

2 12 11

21

2

1

1

1 2

1

1 2

2

1

1

2

1

2

1

2

1

2

1

21

2 1

ATF58143_ADS_model C5

C1

C4

C2

L3L2

R4

L4

L1

R3

C3

R1 R2

SRC1

Term2

Term1

X1

Vdc=3.3 V

Z=50 Ohm

Z=50 Ohm

CHAPTER 3

35


Non-ideal components deviates the result from the ideal components because there are some

parasitic effects involved in non-ideal components. In figure 3-27, along to x-axis, frequency

and along to y-axis noise figure are plotted. From the figure, the noise figure is found as (NF)

0.573 dB and the minimum noise figure is 0.556 dB at central frequency 2.45 GHz


1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

2

4

6

8

10

12

0

14

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-10

-5

0

5

10

15

-15

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

CHAPTER 3

36

In figure 3-28, along to x-axis, frequency and along to y-axis, forward voltage gain are plotted.

Here, the forward voltage gain is 13.32 dB and it can be obtained up-to 14.34 dB at the central

frequency 2.45 GHz but if gain is increased, by changing the IMN and OMN, noise figure also

increases.


The value of input reflection coefficient can be changed by changing the value of input

matching network (IMN) components. The expected value of input reflection coefficient is less

than -6 dB and in figure 3-29, the value is achieved less than -6 dB

3.7 Layout Design of LNA

In the following sections, layout of LNA is designed with Roger’s substrate (Rogers 4350B)

which specifications are [29]:

• Substratethickness, | = 0.254 ~~

• Relative dielectric constant, �� = 3.48

• Conductorthickness, � = 35 µ~

• Dielectric loss tangent, �� = 0.0004

• Conductivity of conductor is 5.8 ∗ 10ll��/~

• Conductorsurfaceroughness is 0.001~~

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-12

-10

-8

-6

-4

-2

-14

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 3

37

3.7.1 Design with non-Ideal Components

The following layout is designed with non-ideal components using S2P file. Figure 3-30 and 3-

31 represent the layout and the layout symbol of LNA respectively with components which

dimension is 36.7 mm x 14 mm. A number of vias are created for better grounding. After

generation the symbol, it was simulated and the following responses of input reflection

coefficient, forward voltage gain and noise figure are seen which are described briefly.

Figure. 3-30 Layout of LNA

Figure. 3-31 Layout symbol of LNA with lumped components

1

1

1

2

1

2

1

2

1

2 1 2 1

2

1

2

1

1

11

1

1 2

3

2 121212 1

1

2

1

2

1

2

3 7

3 6

3 5

3 43 33 23 13 02 9

2 8

2 7

2 6 2 5 2 4

2 3

2 22 1

2 0

1 9

1 8

1 7

1 6

1 51 4

1 3

1 2 1 1

1 0

9

87

6

5

4

321 21

1

2

c e ll_ 2

S 2 P

I _ _ 4 8

L 1

C 1

L 4

L 3

C 2 C 3 C 4 C 5

L 2

R 1 0

R 3

R 2R 1

T e r m 2

S R C 1

T e r m 1

S N P 1

P a r t N u m b e r = L Q G 1 8 H N 1 N 8 S 0 0

P a r t N u m b e r = G Q M 1 8 7 5 C 2 E 8 R 2 C B 1 2



P a r t N u m b e r = G Q M 1 8 7 5 C 2 E 1 R 3 B B 1 2P a r t N u m b e r = G Q M 1 8 7 5 C 2 E 3 R 0 B B 1 2P a r t N u m b e r = G Q M 1 8 7 5 C 2 E 1 2 0 G B 1 2 P a r t N u m b e r = G Q M 1 8 7 5 C 2 E 8 R 2 C B 1 2


R = 1 0 0 O h m

R = 1 0 O h m

R = 2 6 k O h mR = 5 . 4 k O h m

Z = 5 0 O h m

V d c = 3 . 3 V

Z = 5 0 O h m

CHAPTER 3

38

Figure. 3-32 Layout result of noise figure (NF)

In figure 3-32, along to x-axis, frequency and along to y-axis noise figure are plotted. Here the

noise figure (NF) is 0.92 dB and it can be achieved up-to 0.69 dB which is minimum noise

figure at the central frequency 2.45 GHz. These results are deviated from schematic results

(figure 3-19) because now transmission lines are used to connect the non-ideal components. All

the parasitic effects are also considered. That is why, noise figure deviates from 0.618 dB to

0.92 dB.

Figure. 3-33 Layout result of forward voltage gain

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

5

10

15

0

20

Frequency (GHz)

No

ise

Fig

ure

(N

B)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-10

-5

0

5

10

-15

15

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

CHAPTER 3

39

In figure 3-33, the forward voltage gain is 14.73 dB at the central frequency 2.45 GHz but if this

forward voltage gain is increased by changing the IMN and OMN, noise figure also increases.

Gain is very close to the schematic results which is 14.813 dB.

Figure. 3-34 Layout result of input reflection coefficient

When noise figure (NF) increases, input reflection coefficient decreases, so there is a trade-off

between noise figure (NF) and input reflection coefficient. As the value of �� is acceptable up-

to -6 dB, in figure, 3-34, the value is -15.30 dB which is less than -6 dB. Schematic results have

low noise figure as compared to layout results but at layout better value of �� is found as

compared to schematic level (-8.70 dB), so it’s a trade-off between noise figure and input

reflection coefficient.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-14

-12

-10

-8

-6

-4

-2

-16

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 4

40

4 Design of BPF-LNA

In this chapter, the procedure of design of BPF is described step by step. Afterwards, a matching

network is designed to integrate LNA with BPF. Lastly, layout of BPF-LNA is designed.

4.1 Design Specifications of BPF

There are numbers of specifications have to consider to design maximally flat BPF such as stop-

band frequencies, pass-band frequencies, stop-band attenuation, pass-band attenuation and filter

order. In this thesis work, stop band is set-up at 0.1 GHz and 3 GHz. Pass-band is set-up from

2.3 GHz to 2.6 GHz. Stop-band attenuation is set-up at 40 dB and pass-band attenuation is set-

up at 3 dB. In addition, filter order is 4.

4.2 Design of Maximally Flat BPF

In this section, maximally flat band-pass filter is designed with lumped components and

distributed elements.

4.2.1 Design with Lumped Components

The following circuit of maximally flat band-pass filter of order 4 is designed with ideal lumped

components. The pass band is selected from 2.3-2.6 GHz and attenuation for pass-band is -3dB.

Series resonators have very low impedance for the desired bandwidth which is 2.4-2.5 GHz.

Parallel resonators have very high impedance for desired bandwidth to stop the signal from

ground. Filter order 4 is used to design band-pass filter. Higher order of filters has higher loss

due to more components but more sharp and flat response. Circuit complexity goes high as well

with physical dimension. However, the schematic was simulated and the following responses of

input reflection coefficient �� and forward transmission �� are observed. This circuit is

designed alone on the required band and then it will be connected with LNA by matching

network.

CHAPTER 4

41

Figure. 4-1 Schematic of band-pass filter using lumped components


In figure 4-2 and 4-3, along to x-axis, frequency and along to y-axis input reflection coefficient

and forward transmission are plotted respectively. The value of ��is -120 dB at the central

frequency which is 2.45 GHz. Band-Pass filter is showing very appropriate results

for��individually but when this BPF is attached with the LNA circuit using matching network

then it is needed to further optimization to get better results for whole BPF-LNA.

C

Term1

L4

L3

L2

L1

C4

C3

C2

C1Term2

Z=50 Ohm

R=1e-12 OhmL=20.289939 nH

R=1e-12 Ohm

L=216.183563 pH

R=1e-12 OhmL=48.984245 nH

R=1e-12 Ohm

L=521.913291 pH

C=208.765314 f F

C=19.593698 pF

C=86.473425 f F

C=8.115975 pFZ=50 Ohm

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-120

-100

-80

-60

-40

-20

-140

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 4

42

Figure. 4-3 Layout result of forward transmission

In figure 4-3, the forward transmission, �� is showing that the signal is passing at -3 dB

attenuation from 2.3 GHz to 2.6 GHz. But, the desired bandwidth is from 2.4 GHz to 2.5 GHz.

As a margin, from 2.3 GHz to 2.6 GHz is selected.

4.2.2 Design with Distributed Elements

In this project work, all the measurements were performed at the central frequency 2.45 GHz.

At the higher frequencies such as approximately at 1 GHz, lumped components behaves

differently and that is why, use of transmission lines theory is a best option instead [26].

In this section, order 4 stub filter with maximally flat response is designed in figure 4-4. Series

transmission lines have very low impedance for desired bandwidth which is from 2.4 GHz to

2.5 GHz. Parallel short circuit stubs have very high impedance for the desired bandwidth to stop

the signal from ground. After simulation, the following responses of input reflection coefficient,

�� and forward transmission, �� are found. This circuit is designed alone on the required

band and then it will be connected with LNA by a matching network.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-80

-60

-40

-20

-100

0

Frequency (GHz)

Fo

rwa

rd T

ran

sm

issio

n (

S2

1)

dB

CHAPTER 4

43

Figure. 4-4 Schematic of band-pass filter using distributed elements


In figure 4-5, the value of �� is -10.80 dB at the central frequency which is 2.45 GHz. Band-

Pass filter is showing very acceptable results for ��< -6 dB alone but when this BPF is attached

with LNA using matching network then it is needed further optimization to get better results for

the overall BPF-LNA. In this case, quarter wave transmission line is used for matching between

BPF and LNA.

Cros1 Cros2

TL3 TL6 TL9

TL1 TL11

TL2 TL5

TL4

TL8

TL7

TL10

Tee1 Tee2

Term1

Term2

W4=3.574 mm

W3=1.007 mm

W2=3.574 mm

W1=0.86 mm

W4=3.574 mm

W3=0.86 mmW2=3.574 mm

W1=1.007 mm

L=19.36 mm

W=0.86 mm

L=19.4678 mm

W=1.007 mm

L=19.36 mm

W=0.86 mm

L=5.1 mm

W=0.73 mmL=4.75 mmW=0.73 mm

L=16.91 mm

W=0.613 mm

L=16.91 mm

W=3.574 mm

L=16.91 mmW=3.574 mm

L=16.91 mm

W=3.574 mm

L=16.91 mmW=3.574 mm

L=16.91 mm

W=0.613 mm

W3=0.613 mmW2=0.86 mm

W1=0.65 mm

W3=0.613 mm

W2=0.577 mm

W1=0.86 mm

Z=50 Ohm

Z=50 Ohm

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-30

-25

-20

-15

-10

-5

-35

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 4

44


In figure 4-6, the forward transmission, �� is showing that the signal passing at -3 dB

attenuation is from 2.26 GHz to 3.09 GHz. But the desired bandwidth is from 2.4 GHz to 2.5

GHz. As a margin the bandwidth is selected from 2.3 GHz to 3.09 GHz because at layout it will

be left shifted.

In this part, layout of BPF is designed with the previous specifications of section 3.7. The

following layout is designed for stub filter which is shown in figure 4-4. Figure 4-7 and 4-8

represent the layout of the stub filter and layout symbol of stub-filter respectively which

dimension is 56.90 mm x 38.63 mm. A number of vias are created for better grounding. After

generation the symbol, it was then simulated in the schematic window and got the following

responses of input reflection coefficient, �� and forward transmission, ��

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-60

-50

-40

-30

-20

-10

-70

0

Frequency (GHz)

Fo

rwa

rd T

ran

sm

issio

n (

S2

1)

dB

CHAPTER 4

45

Figure. 4-7 Layout of BPF

Figure. 4-8 Layout symbol of BPF

BPF Layout TX af t er m eet ing 17oct

Ter m 2Ter m 1

I __1

Z=50 O hmZ=50 O hm

CHAPTER 4

46


In figure 4-9, the forward transmission, �� is showing that the signal passing at -3 dB

attenuation is from 2.0 GHz to 2.67 GHz. As already mentioned in the schematic of this filter

that the signal at -3 dB will be shifted towards left side, so in this case it is seen that it moves

from (2.3-3.09) GHz to (2.0-2.67) GHz. The required bandwidth which is (2.4-2.5) GHz is still

in the range of -3 dB attenuation.


1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-80

-60

-40

-20

-100

0

Frequency (GHz)

Fo

rwa

rd T

ran

sm

issio

n (

S2

1)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-40

-30

-20

-10

-50

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 4

47

In figure 4-10, the value of ��is -19.35 dB at the central frequency which is 2.45 GHz. Band-

pass filter is showing very acceptable results for ��as compared to schematic level which is -

10.80 dB

4.3 Design of BPF-LNA with Lumped Components

In this section, BPF-LNA is designed using lumped components. Schematic and layout designs

results are compared as well.

4.3.1 Schematic Design with Ideal Components

Order 4 lumped-filter with maximally flat response is connected with LNA through �-matching

network in figure 4-11. The schematic is simulated and found the following responses of noise

figure (NF), input reflection coefficient, and forward voltage gain.

Figure. 4-11 Schematic of BPF-LNA using ideal lumped components



R1 = 5.4 k� C1 = 8.11 L1 = 0.52

R2 = 26 k� C2 = 0.086 L2 = 48.98

R3 = 10 � C3 = 19.95 L3 = 0.21

R4 = 100 � C4 = 0.20 L4 = 20.21

-- C5 = 2.0 L5= 1.80

-- C6 = 1.0 L6= 2.77

-- C7= 3.0 L7= 5.60

-- C8= 1.28 L8= 1.50

-- C9= 8.20 --

Ref

1 2

S2PTerm1

L8

L7L6

L4L2

L1 L3

L5C4

C1

C2

C3

C7SNP1

SRC1

R2R1

C8

R3

Term2

C6C5

C9

R4Z=50 Ohm

Vdc=3.3 V

Z=50 Ohm

CHAPTER 4

48


In figure 4-12, the noise figure (NF) is 0.761 dB and minimum noise figure is 0.611 dB at the

central frequency 2.45 GHz


In figure 4-13, forward voltage gain, �� is 14.28 dB at the central frequency 2.45 GHz and it is

giving almost flat gain from 2.31 GHz to 2.54 GHz. If this forward voltage gain is changed by

changing the IMN and OMN, noise figure will be also increased. The main reason for

connecting the BPF with LNA is that it gives flat gain only in the desired bandwidth.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

20

40

60

80

0

100

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-80

-60

-40

-20

0

-100

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

CHAPTER 4

49


In figure 4-14, the value of �� is -35 dB at the central frequency which is 2.45 GHz. Band-pass

filter stand alone has the value of �� is -120 dB and after connecting with LNA the overall ��

increases but it is still acceptable because it is less than -6 dB.

4.3.1 Layout Design

With the previous substrate specifications of section 3.7, the following layout of BPF-LNA is

designed with non-ideal components. Figure 4-15 and 4-16 represent the layout and layout

symbol of BPF-LNA respectively which dimension is 51.3 mm x 14 mm. A number of vias are

created for better grounding. After generation the symbol, it was then simulated in the

schematic window and got the following responses of noise figure, input reflection coefficient,

�� and forward voltage gain, ��

Figure. 4-15 Layout of BPF-LNA

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-30

-20

-10

-40

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 4

50

Figure. 4-16 Layout symbol of BPF-LNA with lumped components


In figure 4-17, the noise figure (NF) is 1.37 dB and the minimum noise figure 1.051 dB can be

achieved at the central frequency 2.45 GHz. In order to achieve the desired amount of NF,

input matching network’s component should be changed which causes increase the value of

input reflection coefficient.

R e f

1 2

BPFLNA lum ped new layout

S2P

I __40

L8

L5

L4

L1

L2

L3

C9

SNP1

SRC1

L7

L6

C3

C2

C1

C4 C5

C6

C8

C7

R4

R3

R1R2

Ter m 1

Ter m 2

Par t Num ber=LQ G 18HN1N2S00

Par t Num ber =LQ G 18HN1N2S00

Par t Num ber=LQ G 18HN6N8J00




Par t Num ber =G Q M 1875C2E3R0BB12

Vdc=3. 3 V



Par t Num ber=G Q M 1875C2E1R0CB12

Par t Num ber =G Q M 1885C2A1R0BB01

Par t Num ber=G Q M 1885C2A1R0BB01

Par t Num ber=G Q M 1875C2E1R0CB12 Par t Num ber =G Q M 1875C2E1R5CB12

Par t Num ber=G Q M 1875C2E1R2BB12

Par t Number =G Q M 1875C2E1R0BB12

Par t Number =G Q M 1875C2E3R0CB12

R=100 O hm

R=10 O hm

R=5. 4 kO hmR=26 kO hm

Z=50 O hm

Z=50 O hm

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

10

20

30

40

50

60

0

70

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

CHAPTER 4

51


In figure 4-18, forward voltage gain is 13.26 dB at the central frequency 2.45 GHz and it is

providing almost flat gain from 2.35 GHz to 2.60 GHz. The flatness becomes more and the

bandwidth is expanded as compared to schematic level because non-ideal components are used

with very selective values for it from Murata library


The value of ��is also changed very much from schematic level which is -10.80 dB but still

acceptable. The reason for this change is also the use of non-ideal components with very

selected components library from Murata. In this design, non-ideal components availability

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-60

-40

-20

0

-80

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-14

-12

-10

-8

-6

-4

-2

-16

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 4

52

problem was faced in band-pass filter. That is why; a new filter is designed with transmission

line in order to solve this problem.

4.4 Design of BPF-LNA with Distributed Elements

In this section, BPF-LNA is designed using distributed elements. Schematic and layout designs

results are compared as well.

4.4.1 Design of Schematic

Order 4 stub filter with maximally flat response is connected with LNA through quarter wave

matching network in figure 4-20. The schematic is simulated and found the following responses

of noise figure (NF), input reflection coefficient and forward voltage gain.

Figure. 4-20 Schematic of BPF-LNA using distributed elements



R1 = 5.4 k� C1 = 3.0 L1 = 2.7

R2 = 26 k� C2 = 1.2 L2 = 5.6

R3 = 10 � C3 = 8.2 L3 = 1.5

R4 = 100 � -- --

Ref

1 2

S2P

L1 L2

L3TL6

TL8

TL7

TL10

Tee1 Tee2

TL9

TL1 TL11

TL2

Term1TL12TL3

Cros2

C1

Cros1

SNP1

SRC1

R2R1

C2

R3

Term2C3

R4

TL4

TL5

W=1.007 mm W=0.86 mmL=19.36 mm

W=3.574 mmL=16.91 mm

W=0.613 mmL=16.91 mm

W2=0.86 mmW3=0.613 mm W3=0.613 mm

W1=0.65 mm W1=0.86 mmW2=0.577 mmW=0.73 mm

L=5.1 mmW=0.73 mmL=4.75 mm

W=0.613 mmL=16.91 mm

W=3.574 mmL=16.91 mm

W3=1.007 mm

Z=50 Ohm

L=19.4678 mmL=18.56 mmW=0.65 mmSubst="MSub1"

W2=3.574 mm

W4=3.574 mm

W=0.86 mmL=19.36 mm

W1=1.007 mmW2=3.574 mm

W1=0.86 mm

Vdc=3.3 V

W4=3.574 mmW3=0.86 mm

Z=50 Ohm

L=16.91 mmW=3.574 mm

L=16.91 mmW=3.574 mm

CHAPTER 4

53


In figure 4-21, along to x-axis frequency and along to y-axis noise figure are plotted. The noise

figure (NF) is 1.193 dB and the minimum noise figure is 1.166 dB at central the frequency 2.45

GHz


In figure 4-22, forward voltage gain is 12.93 dB at the central frequency 2.45 GHz and the gain

is almost flat from 2.24 GHz to 2.59 GHz. Once this forward voltage gain increases by changing

the IMN and OMN, noise figure increases also at the same time.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

20

40

60

80

0

100

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-80

-60

-40

-20

0

-100

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

CHAPTER 4

54


In figure 4-23, the value of �� is -8.31 dB at the central frequency which is 2.45 GHz. Band-

pass filter stand-alone has the value of �� is -10.80 dB and after connecting with LNA the

overall value of �� increases also but it is still acceptable which is less than -6 dB

4.4.2 Design of Layout

With the previous substrate specifications of section 3.7, the following layout of BPF-LNA is

designed with distributed components. Figure 4-24 and 4-25 represent the layout and layout

symbol of BPF-LNA respectively which dimension is 97.10 mm x 39.75 mm. A number of vias

are created for better grounding. There are two big vias are created to tie-up Vuu. However,

after generation the symbol, it was then simulated in the schematic window and found the

following responses of noise figure, input reflection coefficient, �� and forward voltage gain,

��.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-20

-15

-10

-5

-25

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 4

55

Figure. 4-24 Layout of BPF-LNA with distributed components

Figure. 4-25 Layout symbol of BPF-LNA

V _ D C

S 2 P

B P F L N A T X la y o u t a f t e r m e e t in g o c t 1 7

L 1

L 2

L 3

R 4

R 2

R 1

R 3

S R C 2

S N P 1

C 1

C 2

C 3

T e r m 2T e r m 1

I _ _ 1

P a r t N u m b e r = L Q G 1 8 H N 2 N 7 S 0 0P a r t N u m b e r = L Q G 1 8 H N 5 N 6 S 0 0


R = 1 0 0 O h m

R = 2 6 k O h m

R = 5 . 4 k O h m

R = 1 0 O h m

V d c = 3 . 3 V

P a r t N u m b e r = G Q M 1 8 7 5 C 2 E 1 R 8 B B 1 2



Z = 5 0 O h mZ = 5 0 O h m

CHAPTER 4

56


In figure 4-26, the noise figure (NF) is 1.05 dB and it can be achieved the minimum noise figure

0.94 dB at the central frequency 2.45 GHz. In order to achieve the desired amount of NF, input

matching network’s component should be changed which causes increase the value of input

reflection coefficient.

Figure. 4-27 Layout result of forward voltage gain

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

20

40

60

0

80

Frequency (GHz)

No

ise

Fig

ure

(N

F)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-60

-40

-20

0

-80

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

CHAPTER 4

57

In figure 4-27, the forward voltage gain 12 dB at the central frequency 2.45 GHz and the gain is

almost flat from 2.15-2.66 GHz. The flatness and the bandwidth are almost close to the

schematic level because now non-ideal components are not being used for filter design. The use

of transmission lines has solved the components unavailability problem.

Figure. 4-28 Layout result of input reflection coefficient

In figure 4-28, the value of �� is also changed from the schematic level which is -10.36 dB and

it is a better value than schematic level. Using of transmission lines to design filter is the reason

for this improvement.

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-15

-10

-5

-20

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 5

58

5 Prototypes & Measurements

In this chapter, prototype of LNA, BPF-LNA with lumped components and BPF-LNA with

transmission lines are described which are fabricated at PCB laboratory of ITN. In all the

prototypes Roger’s substrate RO4350B is used. Once the prototype is fabricated, components

are soldered and parameters are measured using vector network analyzer. All the components

are used in the prototype are of standard size 0603 inch.

5.1 Prototype of LNA

The following figure shows the complete prototype of LNA stand-alone which is tested and

measured to see the performances. After getting the result from network analyzer, the generated

S2P file is run in ADS and found the following results of input reflection coefficient and

forward voltage gain.

Figure. 5-1 Photograph of the prototype of LNA stand-alone

CHAPTER 5

59

The following tables show the values of components which were used at layout level and

prototype respectively. Due to unavailability of components, the values of table 7 were used for

the prototype of LNA.

Table 6 Used values at layout

Resistor (k�) Capacitor (pF) Inductor (nH)

26 1.3 1.8

5.4 3.0 2.7

Table 7Used values at prototype


27 1.0 2.2

5.6 1.0 2.2

5.1.1 Measurement Results

In figure 5-2, along to x-axis and along to y-axis frequency and forward voltage gain are

plotted. The forward voltage gain is 7 dB at the central frequency 2.45 GHz and the gain is

almost flat from 1.8-4.1 GHz. The flatness is almost same to the layout level.

Figure. 5-2 Measurement result of forward voltage gain

In figure 5-3, along to x-axis and along to y-axis frequency and forward voltage gain are plotted

respectively. The value of �� is -14 dB which is almost close to the layout level gain. The

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-25

-20

-15

-10

-5

0

5

-30

10

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

CHAPTER 5

60

values which are used at the layout level, in most of cases, those component values were not

found in the desired companies. That is why, the result of prototype level is deviated from the

layout level.

Figure. 5-3 Measurement result of input reflection coefficient

5.2 Prototype of BPF-LNA with Lumped Elements

The following figure shows the complete prototype of BPF-LNA with lumped components

which is tested and measured to see the performances at the prototype. After getting the result

from network analyzer, the generated S2P file is run in ADS and found the following results of

input reflection coefficient and forward voltage gain.

Figure. 5-4 Photograph of the prototype of BPF-LNA with lumped elements

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-14

-12

-10

-8

-6

-4

-2

0

-16

2

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 5

61


prototype level respectively.



26 1.0 1.2

5.4 1.2 2.7

-- 3.0 --

-- 1.5 --

-- 3.0 --

Table 9 Used values in prototype


27 2.2 2.2

5.6 1.0 2.2

-- 2.2 --

-- 2.2 --

-- 3.3 --


In figure 5-5, Forward voltage gain with respect to frequency is shown. The forward voltage

gain is 10 dB at the central frequency 2.45 GHz and the gain is not as flat as expected. But the

value of gain is satisfactory.

CHAPTER 5

62


In figure 5-6, along to x-axis and along to y-axis frequency and input reflection coefficient are

plotted respectively. The value of �� is – 5.5 dB which is almost acceptable but still it is little

higher than – 6 dB.


1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-60

-40

-20

0

-80

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

in (

S2

1)

dB

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-14

-12

-10

-8

-6

-4

-2

0

-16

2

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 5

63

5.3 Prototype of BPF-LNA with Distributed Elements

The following figure shows the complete prototype of BPF-LNA with distributed components

which is tested and measured to see the performances at the prototype. After getting the result

from network analyzer, the generated S2P file is run in ADS and found the following results of

input reflection coefficient and forward voltage gain which are shown in figures 5-8 and 5-9.

Figure. 5-7 Photograph of the prototype of BPF-LNA with distributed element


prototype level respectively.



26 1.8 1.2

5.4 3.6 2.7

Table 11 Used values at prototype


27 1.0 2.2

5.6 3.3 2.2

CHAPTER 5

64


In figure 5-8, forward voltage gain is shown where the value of forward voltage gain is 7 dB at

the central frequency 2.45 GHz and the gain is not as flat as layout level. The bandwidth is also

wide which is not expected.


In figure 5-9, along to x-axis and along to y-axis frequency and input reflection coefficient are

plotted respectively. The value of ��is – 5.0 dB which is higher than -6.


1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-60

-40

-20

0

-80

20

Frequency (GHz)

Fo

rwa

rd V

olta

ge

Ga

int (S

21

) d

B

1.5 2.0 2.5 3.0 3.5 4.0 4.51.0 5.0

-8

-6

-4

-2

-10

0

Frequency (GHz)

Inp

ut R

efle

ctio

n C

oe

ffic

ien

t (S

11

) d

B

CHAPTER 5

65

5.4 Comparison of Layouts and Measured Results

In this section, the comparison of layout and measured results are shown in tabular form. In case

of layout, the results are much better than the measurement results. The optimal values of

components were not found in the Murata and other companies, that is why, other closest

available values were used. Table-6, 8, 10 show the components values which are used in layout

design and simulation, and table-7, 9, 11 are used in the respective prototype. Since equivalent

values of layout components are not used in prototype, that is why, the results are much

deviated into the prototype.Table-12 shows the summary of layout results of LNA stand-alone,

BPF-LNA (Lumped) and BPF-LNA (T-Line). Table-13 shows the summary of measurement

results of LNA stand-alone, BPF-LNA (Lumped) and BPF-LNA (T-Line).

Other non-ideal effects also ought to be responsible for some of the unexpected deviation

between simulation and measurement results. The surface of the conductor layer is not ideally

smooth, then the signal might not propagate entirely as expected. In addition, may be, the

connectors and other SMD components are not placed properly which results a small gap

between the conductor layer and the respective component and these various situations can

cause a capacitive/inductive effect, which causes deviation as well. In the PCB, wires are used

for grounding and Vuu instead of using layer. Throughout the whole design stability was

checked and the operation point was steady. Finally, noise figure was not measured due to

unavailability of instrument.

Table 12 Comparison of layout parameters

Prototypes S 11 (dB) S21 (dB) Noise Figure (NF) (dB)

LNA-Stand alone -15.30 14.73 0.92

BPF-LNA (Lumped) -10.80 13.26 1.37

BPF-LNA (T-Line) -10.36

12 1.05

Table 13 Comparison of measured parameters

Prototypes S 11 (dB) S 21 (dB)

LNA-Stand alone -14.0 7

BPF-LNA (Lumped) -5.5 10

BPF-LNA (T-Line) -5.0 7

CHAPTER 6

66

6 Conclusion and Future Works

6.1 Conclusion

According to the design specifications, LNA stand-alone, BPF-LNA with lumped components,

and transmission lines are designed withATF-58143,and their performance are compared in the

simulation (schematic and layout) and measurement level (prototype level).Optimization was

performed according to get the desired responses in all the designs. All the PCB prototypes

were fabricated using a standard PCB (etch based) process. In case of LNA stand-alone, the

optimum value input reflection coefficient and gain are -14 dB and 7 dB respectively but the

bandwidth is too wide compared to the specification and gain is not much flat. BPF-LNA with

lumped components has a input reflection coefficient and gain of – 5.5 dB and 10 dB,

respectively and the bandwidth is narrower than LNA-stand-alone but still it is wider than the

specified 100 MHz. BPF-LNA with transmissions lines was measured and the input reflection

coefficient is – 5.0 dB and the gain is 7 dB. As the required values of components were not used

to prototype due to unavailability of components that is why, the measurement results of PCB

level is not satisfactory. Once the required values are used, the bandwidth and gain ought be

narrow and almost flat respectively, over the whole bandwidth which is expected for the desired

signal with minimum noise.

Furthermore, throughout the whole design, transistor was stable. The level of satisfaction of this

thesis work is satisfactory. However, due to the parasitic effects and unavailability of required

Murata components, there are some deviations from expectations in the measured results. This

thesis work gives a closer and wide view of all the relevant background theories and design

technologies to the designer. PCB lab works gave a manufacturing hands-on experience which

implies expanding reality of theoretical knowledge.

6.2 Future Works

Though the responses are satisfactory, but still there are scopes to improve the performances.

Some of the circuits can be improved in design and with more proper optimization to have

CHAPTER 6

67

better responses. In future, different classes of BPFs such as elliptical, Chebyshev with different

orders can be designed with LNA which will provide more options to compare for the better

one. Furthermore, exact values of components which were used in the designs can be purchased

and made new prototypes, which may produce better responses of BPF-LNA (designed with

lumped components). However, the acquired knowledge from this thesis work can help to

design the whole RF receiver system in the ISM band.

REFERENCES

68

7 References

[1] David M. Pozar, Microwave and RF Wireless System, John Willey & Sons, Inc. Third

Edition, 2000. Chapter-10

[2] International Telecommunication Union, "Industrial, Scientific and Medical (ISM)

applications of radio frequency energy in the field of telecommunications.", 19 October,

2009.

[3] International Telecommunication Union, Visited date: 14 August, 2012 http://www.itu.int

[4] Federal Communications Commission. “Authorization of Spread Spectrum Systems

Under Parts 15 and 90 of the FCC Rules and Regulations". 18 June, 1985. Retrieved

2007-08-31.

[5] European Commission, Visited date: 16 August, 2012, http://ec.europa.eu/

[6] Electronic Communications Committee, “ERC Recommendation 70-03”,

http://www.erodocdb.

[7] Ghosh, Smarajit, “Network Theory: Analysis and Synthesis”, Prentice Hall of India ISBN

81-203-2638-5 pp. 353

[8] Two-Port Networks, Visited date: 30 August, 2012 , http://fourier.eng.hmc.edu

[9] Andres Moran Valerio, Alonso Perez Garrido, Thesis title “Design and Implementation of

6-8.5 GHz LNA”, Institute of Science & Technology, Linkoping University, 2008-11-07,

pp. 13

[10] Reinhold Ludwig, Pavel Bretchko, “RF Circuit Design”, Prentice-Hall, Inc. New Jersey

07458, ISBN: 0-13-095323-7, 2000. Chapter-2, 4

[11] C. D. Motchenbacher, J.A. Connelly, “Low-Noise Electronic System Design”. Wiley

Interscience. 1993.

[12] Dennis V. Perepelitsa, “Johnson Noise and Shot Noise”, MIT, Department of Physics, 27

November , 2006. pp. 1

[13] M. Blanter, M. Büttiker, Physics Reports on “Shot Noise in Mesoscopic Conductors”,

2000. DOI:10.1016/S0370-1573(99) 00123-4.

[14] Jimmin Chang, A.A. Abidi and C.R. Viswanathan, "Flicker Noise in CMOS Transistors

from Subthreshold to Strong Inversion at Various Temperatures". IEEE Transactions on

Electron Devices, Vol. 41, No. 11 November 1994. pp. 1965

REFERENCES

69

[15] Devendra K. Misra, “Radio-Frequency and Microwave Communication Circuits;

Analysis And Design”, John Wiley and Sons, 2001. Chapter-2

[16] Henry W.Ott, “Noise Reduction Techniques in Electronic Systems”, John Wiley and Sons

1988. Chapter-8

[17] Behzad Razavi, “Design of Analog CMOS Integrated Circuits”, McGraw-Hill, 2000,

Chapter 7: Noise.

[18] Adriana Serban Craciunescu, “Low-Noise Amplifier for Ultra-Wideband System”. LiU-

TEK-LIC-2006.

[19] Lumped Element Filters, Visited date: August, 2012

http://www.microwaves101.com/encyclopedia/Lumpedfilters.cfm

[20] F.R. Connor, “Wave Transmission”, Edward Arnold Ltd., 1972 ISBN 0-7131-3278-7, pp.

13-14

[21] M. Afzal, N. Ahmad , Thesis title, “Investigation of Different Diplexer Design

Techniques for 4G Mobile Communications”, Institute of Science & Technology,

Linkoping University, 2011, LiU-ITN-TEK-A-11/068-SE, pp. 7

[22] David M. Pozar, Microwave Engineering, John Willey & Sons, Inc. Second Edition,

1998, pp. 449-450

[23] C. Zhu, L Yao, J Zhao, “Novel Microstrip Diplexer based on a Dual Band Bandpass Filter

for WLAN Systems”, State Key Laboratory of Millimeter Waves, Southeast University,

Nanjing, 210096, China, Proceedings of Aisa-Pacific Microwave Conference-2010.

[24] Zverev, Matthaei, Young, Jones Dishall, “Handbook of Filter Synthesis”, Artech House,

ISBN 0-890-06099-1-5, Nov. 1951, Chaper-8.

[25] Matthaei, L. George, Young, Leo, Jones, “Microwave Filters, Impedance-Matching

Networks and Coupling Structures”, McGraw-Hill 1980, ISBN 0-89006-099-1

[26] Robin S. Johansson, Torbjörn E. Karlsson, "Low Noise Amplier 2.45 GHz", Lund

Institute of Technology, Lund University, Sweden. May 16, 2011

[27] Venkat Ramana. Aitha, Mohammad Kawsar Imam,Master’s Thesis title “Low Noise

Amplifier for Radio Telescope at 1.42 GHz”, Computer and Electrical Engineering,

Halmstad University, Sweden, IDE0747, May 2007 pp. 29

[28] Data Sheet, AVAGO Technologies, ATF-58143, Visited date: 17 August, 2012

http://www.avagotech.com/

[29] Roger R04350B data sheet, Visited date: August, 2012 www.rogerscorp.com

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