crosstalk and emi noise new4 · 2019. 3. 5. · however, crosstalk analysis of microstrip lines...
TRANSCRIPT
1
Abstract— In this paper, a simple approach has been proposed to determine parameters
of CPMLs with BGS. A break or slot in the ground plane raises the impedance of
microstrip lines. Firstly, the S-parameters of the transmission lines are obtained using an
electromagnetic simulator. Then the L and C values are extracted using circuit modeling
approaches. Measurements were performed using a vector network analyzer for the
fabricated CPMLs. Results show that the parallel or crossed break with decoupling
capacitors or ferrite beads enhances the noise isolation and lead to a reduction of signal
reflection and the radiated emission. Frequency characteristics such as crosstalk, return
loss and EMI radiated noise in the crossed break are degraded if capacitors or beads, are
not used.
Index Terms—break in ground structure (BGS), coupled pair of microstrip lines
(CPMLs), crosstalk effect, EMI noise, slot.
I. INTRODUCTION
Faster signal transients in high-density electronic systems are the current trend in new
high-speed circuit design. Using a cheaper substrate with a smaller number of routing
Crosstalk and EMI Noise Investigation for a Coupled
Pair of Microstrip Lines (CPMLs) with a Break in Ground
Structure (BGS)
Morteza Kazerooni1, Ahmad Cheldavi1 and Mahmoud Kamarei2
1College of Electrical Engineering, Iran University of Science and Technology, Tehran, Iran
2Faculty of Electrical and Computer Engineering University of Tehran, Tehran, Iran
2
layers is an attractive proposition at high frequencies. Routing on a severely size-
constrained circuit board can inadvertently create non-ideal conditions such as a
reference plane with breaks or slots produced by various power and ground islands.
Electromagnetic problems such as crosstalk and EMI noise, which are usually ignored at
relatively low frequencies in these circuits, are critical for overall system performance [1-
3]. Crosstalk or radiation coupling is also a fundamental topic for the estimation of
susceptibility and immunity since there are various types of transmission lines such as
microstrip lines and electronic equipments. However, crosstalk analysis of microstrip
lines including complex structures and a non-uniform ground plane (e.g. BGS) is usually
difficult. The models used for these structures are usually based on the transverse
electromagnetic (TEM) approximation [4], where the per-unit-length parameters are
determined first and then the crosstalk of the propagation signals is generally estimated
by using transmission line theory. Since the medium surrounding the microstrip is
inhomogeneous, numerical techniques such as the method of moments [5], finite-
difference solution technique [6], or finite element method [7] have been used to predict
the per-unit-length parameters. Other solutions have also been developed, e.g. [8] for
time-domain crosstalk between two parallel lossless lines.
In this paper, the problems of crosstalk, return loss and EMI noise in CPMLs with a
BGS are considered and investigated. First, the scattering parameters are obtained using
the full-wave EM simulator Ansoft HFSS version 11.0. Then the L and C values can be
extracted from a circuit modeling approach. These parameters are frequency dependent.
Generally, the solution method is based on the quasi-TEM assumption although in some
3
cases, this assumption is not valid. Finally, the simulation results are compared to
measurements of the structures. The simulated results show good agreement with
measurements. In this paper three examples of CPMLs are considered; one with uniform
and others with non-uniform ground planes.
II. BREAK IN GROUND STRUCTURE (BGS)
For high-speed signal transition systems, the power and ground planes are used as
important references. Therefore, when high-speed transmission lines cross a break in the
ground plane, crosstalk and radiated emission are greatly increased [9-14]. Fig. 1(a)
shows the existence of a crossed break in the ground plane. A crossed break in the
ground plane (Fig. 1(a)) differs from a crossed slot in the ground plane (Fig. 1(b)).
Current flows in the circuit that has the minimum impedance. This means that at high
frequencies, the current in the ground plane tends to flow directly under the signal trace.
Due to the break in the ground plane most of the return current takes a circuitous route
with minimum length around the break and increases the length electrically.
a crossed break in Ground Plane
Ground Plane
Direct Signal Trace 1
Crosstalked Current Flow
Direct Signal Trace 2
Direct Signal Trace 3
Return Signal Trace 1
Return Signal Trace 2
Return Signal Trace 3
(a)
4
a crossed slot in Ground Plane
Ground Plane
Direct Signal Trace 1
Crosstalked Current Flow
Direct Signal Trace 2
Direct Signal Trace 3
Return Signal Trace 1
Return Signal Trace 2
Return Signal Trace 3
(b)
Fig.1. Current distribution in a typical high density circuit. a) a crossed break in ground plane and b) a
crossed slot in ground plane
Increased crosstalk leads to unpredictable signal quality. Radiated emission, on the
other traces, may lead to electromagnetic interference (EMI) noise. Therefore, a ground
plane with a break can interrupt the return current and adversely affect EMI, crosstalk,
signal rise/fall times and characteristic impedance.
In most cases, using breaks in the ground plane is unavoidable. Therefore, some
discrete decoupling capacitors are used to reduce the signal integrity degradation, as
show in Fig. 2. A decoupling capacitor is sometimes placed near the signal trace across
the break to maintain the high frequency return current path, but this is a poor solution.
5
a complete crossed break Ground Plane
Ground Plane
Decoupling Capacitor
Fig. 2. Using decoupling capacitors in ground plane with a complete crossed break
Also, employing decoupling capacitors leads to additional inductance in the board
parameters and therefore creates an undesirable resonance in the frequency response as
well as increased manufacturing costs.
For high frequency design, it is desirable to have a uniform ground plane with no
breaks. However, breaks in the ground or power planes are a common feature in densely
packaged and multilayered PCBs. Breaks due to vias and thru holes divert the ground
plane return current are often used to isolate unintentional signals from intended signals
in a PCB.
The current must return to its source, so the return current follows the break parasitic
inductance and potential signal integrity (SI) problems can occur. Also, the current
through the equivalent inductance results in a noise voltage between the two pieces of the
ground plane that can lead to EMI events. Therefore:
1- A break is often cut in the ground plane to isolate the ground of the noisy drivers
from the sensitive loads by increasing the impedance of the noise route. Therefore, by
6
adjusting the dimensions of the break and changing the current distribution on the ground
plane, good susceptibility can be obtained. Fig. 3(a) shows an illustration of the current
flow scenario in a BGS design.
2- One can properly select the break or slot dimensions to create an electromagnetic
band gap (EBG) structure which suppresses the noise.
3- A complete crossed break in the ground plane can be placed on between the
connector and the microstrip as illustrated in Fig. 3(b). Using a ferrite bead to bridge the
break allows the intended low frequency signals to have a return current path, while
effectively maintaining the high impedance of the break to the high frequency
unintentional signals. High-speed I/O signals such as video should never be run over the
complete crossed break in the ground plane with a ferrite bead.
Noisy Driver Sensitive LoadDirect Current
Return current
A crossed break
Ground Plane
Capacitive current or displacement current is
proportional to the width of the break and
degree of charge density
Inductive current or conduction current is
proportional to the area of the break and degree
of current density
Effective area of the break
(a)
7
(b)
Fig. 3. a) Current flow scenario in a BGS design and b) one method for isolation of low and high
frequency PCB areas with the complete crossed break
III. EQUIVALENT CIRCIT OF A BREAK AND SLOT
Due to existence of a break or slot, an additional capacitance and inductance will be
created (Fig. 4(a)). A break and slot translate into a capactance due to the edge of the
defect and an increased inductance due to the longer current paths. Hence the increase in
area of the current loop causes an increase in EMI emissions or inductive noise since it is
directly proportional to the total defected area. For high speed signals, the signal trace
looks more like a transmission line and impedance mismatches cause reflections.
Therefore quantification of the increase in inductance is desirable for EMI estimation.
The break or slot section can be modeled exactly by lumped frequency dependent L and
C elements assuming that the wavelength is larger than the defect dimensions (Fig. 4(b)).
It is important to note that the transmission line is assumed lossless. The extraction of
simulated results shows that βα << allowing for the assumption of a lossless line. This
model is a real physical model. Therefore, all changes in the defect can be clearly
8
represented by values of the lumped elements. The total ABCD matrix for the
transmission line with the break or slot becomes the product of the three sections (Fig.
4(c)). Two sections are simple transmission lines and a lumped LC resonator has been
placed between them. Therefore, in spite of the existence of the break or slot, the TEM
approximation is correct.
The following equations present the approximate extracted equivalent circuit LC
parameters for the break or slot which are constant with frequency. The parallel
capacitance value in Fig. 4(b) for the given dimensions can be extracted from the
attenuation pole location. The capacitance of the equivalent circuit, can be obtained as:
)ff(2
1Z2fC 2
c200
c
−π⋅= (1)
Once the capacitance value of the equivalent circuit is extracted, the series equivalent
inductance for the given defected section can be calculated as:
Cf41L 2
02π
= (2)
To validate the circuit model, two transmission lines with a break and slot have been
simulated using Ansoft Designer 3.0. The proposed structures are designed on substrates
with relative permittivity 2.33. Also, the values of lA,, lB,, lC are 34.8 ,34.8 and
12.875mm, respectively. The width of the microstrip trace (W) on the board and its
length are 2.25 and 71.6mm, respectively. The dielectric substrate was 0.787 mm thick.
The width of the break or slot (wb) is 2mm and the length of the break and slot are 36.25
and 22mm, respectively. The extracted L, and C values are 4 nH, and 0.376 pF for the
break and 3.792 nH, and 0.247 pF for the slot, where fc and f0 are the resonant frequency
9
and 3-dB cutoff frequency, respectively. The simulation results for the amplitude show
relatively excellent agreement as illustrated in Fig. 4(d) up to 5-6 GHz. This means that
the model is valid up to this frequency. Therefore, above 5-6 GHz non-TEM modes are
dominant. The models can be improved by adding a resistor parallel to the LC resonator,
which represents the radiation from the break or slot.
Length
wb
lA lB
lCw
wb
lA lB
Length
lC
lC
w
(a)
(b)
The break or slot circuit
Port 2Port 1
Measurement plane
Circuit plane Circuit plane
Measurement plane
lA
p0 v,Zp0 v,Z
lB
(c)
10
0 1 2 3 4 5 6 7 8 9 10-60
-50
-40
-30
-20
-10
0
freq(GHz)
Am
plitu
de(d
B)
S parameters of the crossed break using full wave analysis and circuit model simulation
S11
S21
Red and solid lines are fromcircuit model simulation
0 1 2 3 4 5 6 7 8 9 10
-80
-70
-60
-50
-40
-30
-20
-10
0
freq(GHz)
Am
plitu
de(d
B)
S parameters of the crossed slot using full wave analysis and circuit model simulation
S11 S21
Red and solid lines are fromcircuit model simulation
(d)
Fig. 4. a) Dimensions of two microstrip circuits with a crossed break and slot in the ground planes, b)
the break or slot equivalent circuit model, c) schematic of embedded the break or slot by the transmission
lines (lA=lB=(Length-ws)/2) and d) comparison of the scattering parameters using full wave analysis and
circuit simulation
IV. EXTRACTION OF LINE PARAMETERS OF TRANSMISSION LINE WITH
BGS
Based on EM simulation results, the additional inductance and capacitance of a
transmission line with a BGS is established and these parameters are extracted using
circuit theory. The S-parameters as shown in (3) are obtained from Ansoft HFSS
simulator and can be used to derive the total L and C parameters of the transmission line.
This method assumes a TEM mode approximation. The procedure is detailed as:
[ ] ⎥⎦
⎤⎢⎣
⎡=
)f(S)f(S)f(S)f(S
)f(S2221
1211 (3)
The relations between the S-parameters, Z11 and Y11 and the propagation constant for
the lossless line are given as:
11
)l).f(cot(Z.j)f(S)f(S))f(S1)).(f(S1(
))f(S).f(S))f(S1)).(f(S1.((Z)f(Z 0
21122211
21122211p11 β−=
−−−
+−+= (4)
)l).f(cot(Y.j)f(S)f(S))f(S1)).(f(S1(
))f(S).f(S))f(S1)).(f(S1.((Y)f(Y 0
21122211
21122211p11 β−=
−++
++−= (5)
where Zp is the port impedance of 50Ω with Yp being the corresponding admittance.
β , l, Z0 and Y0 are the propagation constant, length, characteristic impedance and
admittance of the interconnect, respectively. Then:
11
110 Y
Z)f(Z = (6)
and
l/)Y.Z(cot)f( 11111−=β (7)
The relation between inductance and capacitance for a transmission line is given by:
εμ= .)f(C).f(L (8)
and
υ==
)f(C1
)f(C)f(L)f(Z0 (9)
therefore, the inductance and capacitance per unit length of a microstrip line are given
by:
)f()f().f(Z
f2
f2.)f(C
1
)f(C1)f(L 0
2 ωβ
=π
υπ
υ=
υ= (10)
and
12
)f(L1)f(C 2υ
= (11)
where
εμ=υ
.1 (12)
Simulations were performed for investigation of inductance and capacitance per unit
length for the proposed transmission line with and without the break and slot (Fig.5).
At low frequencies, the current is distributed uniformly throughout the conductor.
However, as the frequency rises, there will be a tendency for the current to concentrate at
the surface of the conductor. It is assumed that the current decreases exponentially inside
the conductor (skin effect). As a result, the outer portions of the conductor contribute less
than the inner parts to the overall inductance (current has more difficulty passing through
the inner parts due to the skin effect). Thus, if current is concentrating on the surface, the
inductance will be decreased. Therefore, with an increase of frequency, the inductance
decreases.
0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.50
0.5
1
1.5
2
2.5
3
3.5x 10
-7
freq(GHz)
Indu
ctan
ce(H
/m)
Frequency dependent inductance
Line Without the crossed break and slot
Line with the crossed breakLine with the crossed slot
0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.50
1
2
3
4
5
6
7
8
9x 10
-9
freq(GHz)
Cap
acita
nce(
F/m
)
Frequency dependent capacitance
Line without the crossed break and slot
Line With the crossed breakLine with the crossed slot
Fig. 5. Comparison of the frequency dependent inductance and capacitance per unit length for the
proposed transmission line with and without the break and slot
13
As can be seen from Fig. 5, with increasing frequency, the value of inductance for the
line is decreased. The current distribution is supposed to be a vector quantity and a
function of frequency. Hence, the inductance of the transmission line decreases to a
limited extent with an increase in frequency. According to this figure, the crossed break
has caused more values of inductance with respect to the standard transmission line. This
increase is more than 0.07 m/Hμ up to 0.7 GHz. The inductive noise in this frequency is
high. This means that the break in ground plane creates additional inductance in the
transmission lines. Also the results show almost no difference in capacitance line in these
traces at low frequencies. As can be seen, that the capacitive noise increases slowly at
very high frequencies.
V. EXAMPLES OF THREE TYPES OF BREAKS AND SLOTS IN THE GROUND
PLANES OF CPMLS
Fig. 6(a) illustrates types of the breaks and slots that were cut on the ground plane of
the test boards underneath the CPMLs. The cross sections of the CPMLs with the BGS
are shown in Fig. 6(b). For these test boards S=10mm, the break length in the CPMLs
with the parallel break is 71.6mm and the slot length in the crossed and parallel slot is 20
and 28mm, respectively.
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Study board without break
Study board with parallel break
Study board with crossed break
Study board with parallel slot
Study board with crossed slot
(a) (b)
Fig. 6. a) Three basic types of the test boards for crosstalk assessment and b) the cross sections of the
CPMLs with the breaks
With a lumped representation, two coupled transmission lines are described only by L
and C matrices of size of 2 x 2. Two lossless, symmetrical reciprocal transmission lines
are described by:
LLLL
LLLL
LM
M
2221
1211 == (13)
and
CCCC
CCCC
CM
M
2221
1211
−−
=−
−=
(14)
where LM and CM are the mutual inductance and capacitance, respectively. As a second
method, this structure may be described by its eigenvalues, even-mode and odd-mode
15
characteristics. The wave can be decomposed to even and odd mode waves. Each wave
has its own characteristic impedance and propagation constant. The even-mode and odd-
mode characteristic impedances and propagation delays are:
M
Moe CC
LLZ−+
= M
Moo CC
LLZ+−
= (15)
and
)CC)(LL(t MMpde −+= )CC)(LL(t MMpdo ++= (16)
The relative capacitive mc and the magnetic ml coupling can be defined as:
CCc M
m = L
Ll mm = (17)
According to the above equations, in a homogenous medium, the relative coupling
capacitance and inductance are similar, therefore the two modes propagate with the same
speed: tpde = tpdo. If the two modes propagate with the same speed, no far-end crosstalk
will appear with matched terminations. This is the case in coupled planar striplines. In
surface microstrips, however, due to the inhomogeneity of the propagation medium, tpde
and tpdo are different. In surface microstrips, since the dielectric fills only the volume
bellow the strip, the capacitive coupling is weaker than the inductive coupling: lm > cm. In
the next section, the mutual couplings and effects of the breaks and slots on CPMLs are
investigated.
VI. INVESTIGATION OF MUTUAL COUPLINGS ON PROPOSED CPMLS
Mutual capacitance and inductance in two lossless, symmetrical reciprocal
16
transmission lines can be determined using simulation or experiment. At low frequencies
where the transmission lines are electrically short, either capacitive or inductive coupling
dominates for an open or short termination, respectively, of the driven line [15]. In either
case, the coupling is proportional to the mutual parameters, and the mutual capacitance
and inductance can be determined from S21 simulations or measurements. An equivalent
circuit model for this procedure is shown in Fig. 7.
ZS
ZLVS
Driven/Generator Line(TL1)-Z0
(1) (2)
ZFEZNE
Coupled/Receptor Line(TL2 )-Z0
(3) (4)
Near End Coupling
Far End Coupling
Length
Length
ZS
ZL
VS
ZNE ZFE
I1
jωLmI1
V1jωCmV1
Fig. 7. Schematic and equivalent circuit of symmetrical mutually CPMLs
The electrical parameters of CPMLs can be measured using a LCR meter or generally
calculated by well-known reference [16].
Fig. 7 shows the setup was used to determine S21 and derive expressions for mutual
inductance and mutual capacitance between Port 1 connected to one microstrip trace and
Port 2 attached alternately to the near-end and far-end of the other microstrip trace. To
calculate mutual capacitance, both traces were terminated with an open circuit. Also, to
calculate mutual inductance, both traces were terminated with a short circuit. The mutual
inductance and mutual capacitance were determined from S21 as follows [17]:
17
LengthZS
CPI
m ...221
ω= (18)
LengthZS
L PIm ..2
.21
ω= (19)
where ZPI is the port impedance (50 ohm). Fig. 8(a) shows the mutual coupling for
proposed structures. Whole of the cases, except for the crossed break, the CPMLs with
non uniform ground planes can be modeled using the symmetrical CPMLs with uniform
ground planes. In this case equations (13) to (19) can be used to determine the
characteristic impedances, phase constants and mutual couplings. From this figure, in the
CPMLs with a break or slot, the inductive coupling increases at low frequencies. Since
the return current is evenly distributed on the ground plane, existence of a slot causes a
disturbance. Cutting a break in the ground plane between and parallel the signal traces
can effectively isolate the driven circuit from the coupled circuit and reduce the inductive
coupling at higher frequencies e.g. above 1.05 GHz as illustrated in Fig.8(a).
At high frequencies with a parallel slot in the ground plane, the return current is
allowed to spread out across the ground plane just as it was able to do when there was a
solid ground plane. Since the width of slot is small, the return current is not adversely
disrupted and the mutual inductance is not too affected. When the return current is
restricted to the side under the source trace in the parallel break case, the inductive
coupling is decreased. However, in the parallel slot case, the isolation is generally
ineffective and can increase inductive coupling.
18
At lower frequencies, parallel breaking and slotting (bellow 0.9 GHz and 0.6 GHz,
respectively) in the ground plane can increase capacitive coupling but at higher
frequencies these defects can decrease it.
When a crossed slot is cut into the ground plane, the current distribution is abruptly
changed (Fig. 8(a)) and more of the electric and magnetic fluxes that would normally
couple to the ground plane from the source trace couple to the victim trace, thereby
increasing mutual inductance and capacitance. A similar effect occurs in the crossed
break as well. However, in this case, the mutual couplings are very much higher than the
crossed slot. Therefore, the line parameters of two mutual traces are not the same and the
CPMLs is not a symmetrical structure. Thus, one cannot use the above equations. The
proposed equivalent circuit of the crossed break is shown in Fig. 8(b). This equivalent
circuit contains two lumped inductances and two lumped capacitances and related
couplings such that LB1> LB2 and CB2 > CB1. To determine these values, the method
which is discussed in this paper can be used for each line individually. The extracted LB1,
LB2, CB1 and CB2 values are 5.16 nH, 1.89 nH, 0.292 pF and 0.881 pF, respectively.
Mutual couplings can be obtained from full wave simulation and circuit modeling by
comparison of standard CPMLs and CPMLs with the crossed break.
Fig. 8(c) illustrates how the breaks and slots change the distributions of return currents
at low or high frequencies. The magnitude of surface current distribution on the ground
of the test board with the crossed break is graphically presented in Fig. 8(d) at 1GHz. The
difference of current level due to the break is obvious.
19
20
(a)
(b)
21
(c)
(d)
Fig. 8. a) Mutual inductance and capacitance for the proposed CPMLs, b) the equivalent circuit for the
CPMLs with the crossed break, c) schematic of the current distribution in the proposed breaks and slots and
d) the magnitude of surface current at the 1 GHz for CPMLs with the crossed break in the ground plane.
22
VII. INVESTIGATION OF CROSSTALK EFFECT, RETURN LOSS AND EMI
NOISE IN PROPOSED CPMLS
In order to further investigate the conditions under which the crosstalk and return loss
is increased, a number of experiments and computer EM simulations were performed.
The far-end crosstalk cannot be neglected in the inhomogeneous medium case and can be
a dominant factor when the rise and fall times of the applied input signal are very short.
The effect of the parallel and crossed break on the near and far-end crosstalk of the
investigated test boards is shown in Figs. 9(a) and 9(b). Due to the crossed break in the
ground, the crosstalk in neighboring transmission lines is increased. Therefore, the
equivalent expression may be interpreted as referred to as inductive and capacitive
couplings. The influence of these parameters on the performance shows an impressive
destruction in signal integrity. This degradation is about 26dB for the near-end and 33dB
for the far-end. Also, the far-end crosstalk in the CPMLs with the parallel break is
improved. However, the near-end crosstalk in the board with the parallel break is slightly
increased at high frequencies.
Fig. 9(c) shows the measured and simulated results of S11 for the three proposed
CPMLs test boards. As shown in this figure, the S11 in the CPMLs is increased
significantly with the crossed break.
To survey the EMI noise in these test boards, the E-field pattern is a good gauge. As
can be seen from Fig.10, the realized pattern gain in the CPMLs with the crossed break
has been increased. This means that EMI noise in this case is higher than the other test
boards.
23
The proposed test boards in this paper were fabricated and measured from 0.5 to 1.5
GHz. Photographs of three fabricated circuits with the breaks are shown in Fig. 11.
0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.5
x 109
-70
-60
-50
-40
-30
-20
-10Comparison of measurement and simulation
freq(GHz)
Nea
r-en
d cr
osst
alk(
dB)
measured
(a)
0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.5
x 109
-55
-50
-45
-40
-35
-30
-25
-20
-15
-10Comparison of measurement and simulation
freq(GHz)
Far
-end
cro
ssta
lk(d
B)
measured
(b)
24
0.5 0.6 0.7 0.8 0.9 1 1.1 1.2 1.3 1.4 1.5
x 109
-55
-50
-45
-40
-35
-30
-25
-20
-15
-10
-5Comparison of measurement and simulation
freq(GHz)
Ret
urn
loss
(dB
)
measured
S11(
dB)
(c)
Fig. 9. Comparison of the measured and simulated of a) near-end crosstalk, b) far-end crosstalk and c)
return loss in the proposed test boards
-50.00
-40.00
-30.00
-20.00
90
60
30
0
-30
-60
-90
-120
-150
-180
150
120
Fig. 10. Comparison of the radiation pattern in E-plane between the three proposed CPMLs
25
The proposed breaks
Fig. 11. Photograph of the three fabricated CPMLs
VIII. CONCLUSIONS
A simple technique for extraction of L, C and mutual coupling parameters of CPMLs
with a break and slot in ground plane has been developed.
Breaking or slotting (defecting) the ground plane is an effective way to reduce or
suppress noise, but this approach may create some undesirable effects on the
transmission lines such as CMPLs. A narrow parallel slot cut into the ground plane may
be ineffective at reducing inductive coupling at high frequencies. However, it can reduce
the capacitive coupling at these frequencies. Adding a parallel break in the ground plane
actually decreases the mutual capacitance and inductance at high frequencies. Three
factors, namely the types, orientation and dimensions of the defect determine the effect
of the break or slot on capacitive and inductive couplings in CPMLs. Also, if the defect
crosses the traces, mutual couplings increase.
To reduce far-end crosstalk between two traces in the CPMLs, it may be advisable to
apply a break in ground plane such as using a parallel break.
26
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