carrier aggregation lte challenges(devices)

IEEE Communications Magazine • December 201376 0163-6804/13/$25.00 © 2013 IEEE

INTRODUCTION

This article addresses the radio frequency (RF)design challenges of terminals for carrier aggre-gation (CA). CA is one of the key features ofthe Third Generation Partnership Project(3GPP) Long Term Evolution (LTE), helping toachieve higher data rates by transmitting multi-ple carriers from or to the same terminal [1]. InRelease-10 of LTE, up to five carriers can beaggregated, each with a bandwidth of 1.4, 3, 5,10, 15, or 20 MHz, thereby allowing for overallbandwidth of up to 100 MHz. Wide bandwidthsare very desirable, as they enable mobile broad-band services such as high-definition videostreaming to take off. Unfortunately, in manyareas of the world, there are currently few, ifany, contiguous spectrum allocations of 20 MHzor higher. CA is the technical solution to over-come the spectrum fragmentation, and it pro-vides the flexibility needed to adapt to a widevariety of spectrum scenarios. However, withflexibility come RF design challenges of termi-nals, as will become clear in this article.

Carriers may be contiguous or non-contigu-ous (NC) in frequency within the same band(intra-band CA) or across multiple bands (inter-band CA), which allows for full utilization offragmented spectrum. From a digital basebandpoint of view, there is little difference amongthese CA types. From an RF implementationpoint of view, however, the design complexity ofthe terminal heavily depends on the CA type,

with intra-band contiguous CA being the leastcomplex. Moreover, the RF characteristics of aterminal vary with the aggregated frequencybands.

During initial access, a CA-capable terminalbehaves similarly to a terminal from earlierreleases; that is, there is a single carrier, referredto as a primary component carrier (PCC). Uponsuccessful connection to the network, dependingon its own capabilities and the network configu-ration, a terminal may be configured with addi-tional carriers in the uplink (UL) and downlink(DL), which are referred to as secondary compo-nent carriers (SCCs). Therefore, in reality, a ter-minal may be configured with multiple CCs eventhough not all of them are currently used. Inorder to save battery consumption, the networkcan activate/deactivate the configured carriersfor a particular terminal as needed.

In Release-10, the number of DL carriers isalways greater than or equal to the number ofUL carriers (i.e., only DL-heavy asymmetries aresupported). This is useful to handle differentspectrum allocations, for example, if an operatorhas more spectrum available for DL than forUL. In addition, this helps to provide a terminalwith DL-heavy traffic such as high-definitionvideo streaming. For example, in Fig. 1a, termi-nal X1 supports two aggregated carriers in DL,whereas it supports a single carrier (i.e., no CA)in UL.

To facilitate load balancing across carriersand handle different capability among terminals,the configuration of carriers is terminal-specific.In other words, the configuration of PCC andthe set of carriers in each direction may varyfrom terminal to terminal. For example, in Fig.1a, the PCC and the set of carriers supported byterminal X2 are different from those supportedby terminal X3. Therefore, it readily follows thatthe number of carriers configured in a networkmay be different from the number of carrierssupported by a terminal. Since each carrier usesthe legacy physical layer structure in order toachieve backward compatibility, it is fully acces-sible to a terminal from earlier releases (e.g.,terminal X3). The association between UL andDL carriers is cell-specific and signaled as partof the system information.

CA is supported for both frequency-division

ABSTRACT

Carrier aggregation is a key feature of 3GPPLTE that addresses the support of higher datarates and utilization of fragmented spectrumholdings. In this article, the relevant design chal-lenges of terminals are discussed. The transmit-ter architectures are reviewed, and the minimumamount of power amplifier back-offs is evaluat-ed. In addition, several receiver architectures arecompared from the perspective of design trade-off. The radio impairments affecting the receiverperformance are analyzed and the simulationresults are provided. The corresponding siliconimplementation is presented together with themeasurement results.

RADIO COMMUNICATIONS

Chester Sungchung Park, Konkuk University

Lars Sundström, Anders Wallén, and Ali Khayrallah, Ericsson

Carrier Aggregation for LTE-Advanced:Design Challenges of Terminals

PARK_LAYOUT_Layout 12/3/13 2:56 PM Page 76

IEEE Communications Magazine • December 2013 77

duplex (FDD) and time-division duplex (TDD)with all carriers using the same duplex scheme.Throughout this article, FDD operation isassumed, since it is generally more challengingfrom the perspective of RF implementations(e.g., transmitter emission).

The readers are referred to [1, 2] for moretechnical details of CA, and [3] for a generalunderstanding of radio design. In this article, thedesign challenges of terminals are described,assuming intra-band NC CA (unless otherwisementioned). The coexistence of different net-works is explained, taking into account theunwanted emission requirements of a transmit-ter and the selectivity and blocking requirementsof a receiver. Several transmitter architecturesfor CA operation are introduced. Consideringthe nonlinearity of the transmitter (both RFchain and power amplifier [PA]), the minimumPA back-off required to meet the unwantedemission requirements is evaluated. In addition,several receiver architectures for CA operationare introduced, and the radio impairmentsaffecting the receiver performance are discussedtogether with some simulation results. Specifical-ly, the radio impairments include transmitteremission, phase noise, receiver nonlinearity,local oscillator (LO) coupling, limited harmonicrejection, and limited image rejection.

COEXISTENCE ISSUESComponent carriers occupied by one operatormay be located adjacent to those occupied byanother operator. In general, networks belong-ing to different operators are deployed in anuncoordinated way. In Fig. 1b, base stations Xand Y, belonging to operators X and Y, respec-tively, are placed in different locations. TerminalX2 (connected to the network of operator X) islocated closer to base station Y than terminal Y(connected to the network of operator Y).Therefore, an interfering signal from terminalX2 seen by the receiver of base station Y may bemuch stronger than a wanted signal from termi-nal Y (since the path loss from terminal X2 tobase station Y is much smaller than the path lossfrom terminal Y to base station Y). Likewise, aninterfering signal from base station Y seen bythe receiver of terminal X2 tends to be muchstronger than a wanted signal from base stationX, as illustrated in Fig. 1b.

In order to satisfy regulatory requirementsand guarantee coexistence with other operators,the transmitter of a terminal should be designedto comply with the unwanted emission require-ments, which set limits on all emission outsidethe channel bandwidth. For example, theunwanted emissions from terminal X2 should bekept reasonably low in order to protect the want-ed signal from terminal Y at the receiver of basestation Y. The unwanted emissions from thetransmitter are divided into out-of-band (OOB)emissions and spurious emissions [4]. The OOBemissions are unwanted emissions immediatelyoutside the assigned channel bandwidth, whichare specified in terms of a spectrum emissionmask (SEM) and an adjacent channel leakagepower ratio (ACLR). The spurious emissions aredefined for all frequency ranges other than those

covered by the SEM. In the case of intra-bandNC CA, the unwanted emission requirementsfor single-carrier operation are applied to eachof the carriers so as to keep the unwanted emis-sion to a maximum of the emission level for sin-gle-carrier operation. For example, the SEM andspurious emissions are set to the maximum ofthose of all carriers [5], as illustrated in Fig. 2.The ACLR is defined on a per-carrier basis andcalculated with respect to the total power ofaggregated carriers (being used as the numera-tor) [5].

Similarly, in order to guarantee the receptionof a wanted signal in the presence of stronginterfering signals (possibly generated by anoth-er operator), the terminal receiver should bedesigned to meet the selectivity and blockingrequirements [4]. Note that an interfering signalmay be located within the gap between twoaggregated carriers. For example, in Fig. 1b, thereceiver of terminal X2 should be able to receivetwo aggregated carriers from base station X inthe presence of a strong interfering signal frombase station Y inside the gap.

TRANSMITTER

TRANSMITTER ARCHITECTURESIn order to support CA, the baseline transmitterfor single-carrier operation can be extended inseveral ways. The choice of transmitter architec-ture for CA operation depends on how to com-bine the aggregated carriers in the transmitterchain. Here we consider three options to illus-trate the design trade-offs. Assuming two con-tiguous or non-contiguous carriers, the left side

Figure 1. Examples of CA configuration: a) terminal specific configurations;b) coexistence with uncoordinated operators.

(a)Base station X

Downlink Uplink

PP

P

P

P

P

XX

P

PP

X1

X3

Y

X2

Y

Y XXBase station Y

(b)

Terminal Y XX Y

Terminal X2

Y XX


IEEE Communications Magazine • December 201378

of Fig. 3a illustrates various transmitter architec-tures depending on where the carriers are com-bined: at the digital baseband (option 1), at theRF chain before power amplification (option 2),or after power amplification (option 3). Option 1is characterized by a single transmitter chain.Conversely, option 3 includes two separate trans-mitter chains, each consisting of a basebandchain, an RF chain, and a PA. The numbers ofbaseband chains, RF chains, and PAs depend onthe transmitter architecture. Furthermore, it ispossible to extend option 3 into a multi-antennaarchitecture by removing the combiner and con-necting each PA to an antenna.

Option 1 is the least complex architecture, forexample, in terms of chip size and power con-sumption, since it has only a single transmitterchain. This transmitter architecture is generallywell suited for contiguous CA. The otherextreme, option 3, is the most complex, sinceeach carrier has its own transmitter chain. Thistransmitter architecture is well suited for inter-band CA. Option 2, a compromise betweenthese two options, is considered a reasonablecandidate for intra-band NC CA.

When choosing the transmitter architecture,one of the most important factors to consider isthe amount of PA back-off required to satisfythe unwanted emission requirements mentionedearlier. The required PA back-off largelydepends on whether the carriers are amplified ina single PA or two different PAs. For example,in options 1 and 2, two carriers are combinedfirst and then amplified with a single PA. Thesituation is similar to a two-tone test for a PA;that is, a large PA back-off may be required tosuppress intermodulation (IM) products (inorder to comply with SEM and spurious emis-

sion requirements). In contrast, in option 3, eachcarrier is amplified by a separate PA, and thetwo carriers are added up in the combiner. Thistends to reduce the IM products and thus lowerthe required PA back-off (e.g., if two PA outputsare sufficiently well isolated from each other).Note that the use of a combiner results in inser-tion loss of at least 3 dB and therefore doublesthe power consumption. For options 1 and 2,this is not an issue since the power levels ofcombined signals are even lower.

MINIMUM PA BACK-OFFIn this section, the amount of PA back-offrequired to satisfy the unwanted emission require-ments is evaluated. The minimum PA back-off ofa transmitter heavily depends on the air interfaceof LTE UL, which is characterized by discreteFourier transform spread orthogonal frequency-division multiplexing (DFTS-OFDM) [2]. There-fore, the transmitter supports flexible resourceallocation in frequency. The frequency domainresource allocation keeps the cubic metric (CM)from being used as an indicator of the requiredPA back-off, since the IM products do not alwaysfall into the OOB domain. For this reason, themaximum allowable PA back-off of an LTE ter-minal is generally specified in terms of resourceallocation as well as modulation order [4].

The CA operation tends to increase the mini-mum PA back-off, especially when the gap widthis so large that the IM products of carriers occurfar apart from the carriers (e.g., in the spuriousemission domain). In the case of intra-band NCCA, the required PA back-off is generally maxi-mized when the UL resource is allocated on theoutermost edges (since the IM products reach asfar as possible). For example, assuming two 5MHz carriers with 5 MHz gap and option 2, thethird order IM products (referred to as IM3products hereafter) fall into the spurious emis-sion domain with the outermost edge allocation(Fig. 4a), whereas they fall into the SEM domainwith the innermost edge allocation (Fig. 4b).(Note that the measurement bandwidth of powerspectral density [PSD] is either 1 MHz [redcurve] or 30 kHz [blue curve].) Hence, it is clear-ly shown that the outermost edge allocation leadsto even larger minimum PA back-off than theinnermost edge allocation for the same resourceallocation (0.9 MHz per carrier in this example).However, two aggregated carriers with sufficient-ly large gap generally require relatively largeminimum PA back-off, regardless of whether theUL resource is on the outermost or innermostedges, since (an entire portion of) the IM3 prod-uct belongs to the spurious emission domain.

In addition, the minimum PA back-off gener-ally decreases with resource allocation, since theIM products spread out more in frequency, low-ering the PSD for a given PA back-off. (Notethat both the SEM and the spurious emissionsare specified in terms of PSD, not the absolutepower level [4].) For example, the allocation of2.7 MHz per carrier (Fig. 4c) leads to smallerminimum PA back-off than the allocation of 0.9MHz per carrier (Fig. 4a).

Assuming random resource allocations, Fig.4d shows the simulation results of the minimumPA back-off for intra-band NC CA as a function

Figure 2. Unwanted emission requirements: a) single-carrier operation ; b)intra-band NC CA operation .

Frequency [MHz]

(a)

-30-40

-30

-35

PSD

[dB

m]

-25

-20

-15

-10

-5

-20 -10 0 10 20 30 40

Frequency [MHz]

(b)

-30-40

-30

-35

PSD

[dB

m]

-25

-20

-15

-10

-5

-20 -10 0 10 20 30 40



of resource allocation as specified in 3GPP [5].It is clearly shown that the envelope of minimumPA back-off values decreases linearly withresource allocation. Here the PSD of two carri-ers is assumed to be equal. In general, the enve-lope of minimum PA back-off values depends onthe PSD ratio [6]. It is also worth mentioning

that, given the resource allocation and PSDratio, the minimum PA back-off varies substan-tially depending on the resource allocation ratioacross carriers. The resource allocation ratio thatmaximizes the required PA back-off is approxi-mately inversely proportional to the PSD ratio.For example, if carrier 1 has twice as high PSD

Figure 3. Terminal implementation: a) transmitter and receiver architectures; b)implementation example of receiver architecture (option 2).

3 bits

IF filter

5-20MHz

IF mixer

4.75-34.75 MHz

IQ17 bits

3 bits

RF HB

LO HB

IF FILTERTESTMUX

DCXO

LO LB

RF LB

CSF

C BI

AS

DIG

CTR

L SP

I

SEQ

UEN

CER

IF M

IXER

LO IF

ADC

CSF ADC

CSF ADC

LO DIG

(a)

(b)

ReceiverTransmitter

LNA A/D Digitalbaseband1

Option 1Option 1

Option 2

LO1LO1

LO2

LNA

Option 2

PA1

LNA

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LO1

D/ADigitalbaseband1

A/D Digitalbaseband1LO1

A/D Digitalbaseband2LO2

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A/D Digitalbaseband2

LO1

PA1

PA1

PA2

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LO2D/ADigital

baseband2

Option 3 LO1D/ADigital

baseband1

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baseband2

IQ27 bits

IQ37 bits

CLK

LNA

LO HB

ADC

ADC

Switch unit

1.8-2.7 GHz

RFHB

LNA

LO LB0.7-1.5 GHz

RFHB

3 bits

3 bitsADC

ADC3 bits

3 bitsADC

ADC

SEQDIVLO IFDCXO

LDOs BIAS TESTMUX DIG CTRL SPI

+1,2,4,8,16195-390 MHz26 MHz

4x64bits

LO DIG

312 MHz

Formatterscrambler

LVDS drivers



as carrier 2, the required PA back-off is maxi-mized when the resource allocation of carrier 1is half that of carrier 2.

Considering the impact of PA back-off onUL coverage, it can be concluded that the ULscheduling decision of the base station has a sig-nificant impact on the UL coverage. For exam-ple, it is desirable to avoid the resourceallocation on the outermost channel edges if theterminal would be power-limited with the corre-sponding PA back-off. However, it is not alwayspossible to avoid such resource allocation, forexample, since the UL control transmission(PUCCH), if any, is typically small resource allo-cation scheduled around the edges of UL chan-nel bandwidth [2].

RECEIVER

RECEIVER ARCHITECTUREAssuming two contiguous or non-contiguous car-riers, the right side of Fig. 3a illustrates variousreceiver architectures differentiated by where

the carriers are separated from each other: at adigital baseband (option 1), at an intermediatefrequency (IF) chain (option 2), or at an RFchain (option 3). The numbers of RF chains, IFchains and baseband chains vary with choice ofreceiver architecture. For example, option 1 con-sists of a single RF chain, a single IF chain, andmultiple digital baseband chains, whereas option3 consists of two receiver chains, each consistingof an RF chain, an IF chain, and a digital base-band chain.

Options 1 and 2 are equipped with a singleRF LO whose center frequency may be set tothe center of the whole frequency range span-ning two carriers that are down-converted to IFsimultaneously. An example of option 1 is a sin-gle wideband homodyne receiver that processesthe down-converted frequency range as a singlecarrier signal. An example of option 2 is a het-erodyne receiver where an IF receiver is dedicat-ed to each carrier as shown in [8], which is moredetailed in Fig. 3b. On the other hand, option 3is equipped with two RF LOs whose center fre-

Figure 4. Minimum PA back-off for intra-band NC CA for Option 2: a) outermost edge allocation of 0.9 MHz per carrier (11.0 dB back-off); b) innermost edge allocation of 0.9 MHz per carrier (3.4 dB backoff); c) outermost edge allocation of 2.7 MHz per carrier (8.6 dBbackoff); d) PA simulation with random resource allocation with respect of inter-carrier gap.

Frequency [MHz]

(a)

-30-40

-60

-70

PSD

[dB

m /

1 M

Hz]

or

[dBm

/ 30

khz

]

-50

-40

-30

-20

-10

0

10

-20 -10 0 10 20 30 40Frequency [MHz]

(b)

-30-40

PSD

[dB

m /

1 M

Hz]

or

[dBm

/ 30

khz

] 10

20

0

-10

-20

-30

-40

-50

-20 -10 0 10 20 30 40

Frequency [MHz]

(c)

-30-40

-60

-70

PSD

[dB

m /

1 M

Hz]

or

[dBm

/ 30

khz

]

-50

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10

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00

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imum

req

uire

d PA

bac

koff

[dB

] 12

14

10

8

6

4

2

8 9

106

7654321

5 MHz gap10 MHz gap20 MHz gap


quencies may be set to those of two carriers.Thus, each receiver chain down-converts a carri-er to baseband separately. An example of option3 is a receiver where one homodyne receiver isdedicated to each carrier.

Option 1 is the least complex architecture, forexample, in terms of chip size and power con-sumption since it has only a single receiver chain.However, it may have design issues such as gaincontrol of analog circuitry, especially when thereis a significant time delay among carriers. Thisreceiver architecture is generally well suited forcontiguous CA. The other extreme, option 3, isthe most complex, since each carrier has its ownreceiver chain. This receiver architecture is wellsuited for inter-band CA. Each of the receiverarchitectures has its own pros and cons. Forinstance, option 1 tends to consume less batterypower, while it is more susceptible to the adja-cent channel interference.

In the remainder of this section, the radioimpairments affecting the receiver performancewill be discussed. Again, for simplicity, intra-band NC CA with two carriers is assumed,although most of the arguments are generallyapplicable to other CA types.

TRANSMITTER EMISSIONThe unwanted emission from the transmittermay cause non-negligible interference to thereceiver of the same terminal. Although this mayalso be true for single-carrier operation, it iseven more likely for CA operation. For instance,considering a scenario with two UL carrierslocated at a distance of integer multiples of theinter-carrier spacing from a DL carrier, the IMproducts of two UL carriers fall on top of DLcarriers. In particular, when the inter-carrierspacing is half the duplexer distance, the IM3product falls on top of the close-in DL carrier,possibly causing severe interference to thereceiver, as shown in Fig. 5a.

The presence of even a single UL carrier maycause severe interference to the receiver becauseof limited duplexer isolation. If the close-in ULcarrier is configured as the PCC by the network(as shown in Fig. 5b), the minimum distancebetween UL and DL carriers is given as theduplexer distance minus the inter-carrier spac-ing; thus, it may be much smaller than theduplexer distance. In an extreme case where twoUL carriers are located on both ends of thetransmit band, the gap between UL and DL car-riers is as small as the duplexer gap.

In general, the emission due to spectralregrowth from the transmitter increases with theresource allocation in UL. For example, the allo-cation of 5 MHz (Fig. 6a) creates more emissionthan the allocation of 1.8 MHz (Fig. 6b). Thus, itis possible to keep the interference to the receiv-er below a certain level by limiting the resourceallocation in UL. However, even small resourceallocation may create non-negligible interferenceat a small frequency offset [9]. For example,when small resource is allocated around theclose-in edge of UL channel bandwidth, the IMproducts of UL transmission, and its IQ imageappears at a small frequency offset, as illustratedin Fig. 6c. On the other hand, when smallresource is allocated around the far-off edge of

UL channel bandwidth, the UL transmission cre-ates the counter IM3 (CIM3) onto a DL carrier,which tends to be about 20 dB weaker than theaforementioned IM3 product of UL transmissionand its IQ image. The UL data transmission(PUSCH) may be such small resource allocation


Figure 5. Receiver impairments: a) transmitter emission with one UL carrier; b)transmission emission with two UL carriers; c) transmitter phase noise; d)receiver phase noise; e) receiver nonlinearity; f) LO coupling or harmonic rejec-tion; g) image rejection.

Downlink

(a)

Uplink

Downlink

(b)

Uplink

Downlink

(c)

LO

Uplink

Downlink

(d)

LO

LO

Uplink

DownlinkUplink

(e)

DownlinkUplink

(f)

(g)

DownlinkUplink



around the edges of UL channel bandwidth,depending on the network scheduling. In addi-tion, the PUCCH transmission, if any, is typicallysmall resource scheduled around the edges ofUL channel bandwidth together with frequencyhopping over two slots (with each slot spanning0.5 ms). The impact on the DL data reception(PDSCH) also depends on the network schedul-ing, since the interference hits only a small frac-tion of the channel bandwidth. However, the DLcontrol reception (PDCCH) may be most likelyinterfered, since it generally spans the wholechannel bandwidth [2].

Assuming that the close-in UL carrier is con-figured as the PCC by the network and theresource allocation starts from the nearby edgeof UL channel bandwidth, the amount of trans-mitter emission on the close-in DL carrier isevaluated in Fig. 6d with respect to UL resourceallocation. The transmitter emission is measuredat the antenna port of terminal. The simulationparameters are set as follows:• Duplexer distance : 80 MHz• Duplexer gap : 15 MHz• Receiver noise figure: 6.5 dB• Insertion loss: 5 dBIt is clearly shown that the receiver performanceis severely affected by the transmitter emissionwhen the inter-carrier gap is extremely large.

PHASE NOISEThe ideal implementation of LOs provides theterminal (both transmitter and receiver) with asinusoidal waveform (i.e., a single tone) whosefrequency is set to the center frequency of thewanted signal. However, the desired LO signal isaccompanied by a phase noise that decreaseswith the offset from the desired frequency. Inthe case of intra-band NC CA, the impact ofphase noise becomes more pronounced, sincethe minimum distance between UL and DL car-riers may be far smaller than the duplex distance(depending on the inter-carrier spacing).

The phase noise of transmitter LOs causesinterference to a DL carrier (referred to as in-band noise) after the up-conversion of a trans-mitter, as illustrated in Fig. 5c. In general, thecloser the UL carrier is to the DL carrier (i.e.,the smaller the minimum distance between ULand DL carriers), the more interference thephase noise causes to the DL carrier.

Similarly, the phase noise of receiver LOs caus-es interference from a UL carrier (referred to asreciprocal mixing) after the down-conversion ofthe received signal, as illustrated in Fig. 5d. Again,the interference due to phase noise decreases withthe distance between UL and DL carriers.

Assuming the phase noise of –143 dBc/Hz atthe offset of 20 MHz and the slope of –20 dBper decade [10], the amount of interference dueto phase noise is evaluated as a function of inter-carrier gap in Fig. 6d. It is shown that the phasenoise may dominate the receiver performanceloss (e.g., when the inter-carrier gap is between45 and 55 MHz).

RECEIVER NONLINEARITYReceiver nonlinearity may cause further interfer-ence from a UL carrier to a DL carrier. In par-ticular, a UL carrier may create an IM2 product

on the DL carriers. The interference due to non-linearity of analog baseband tends to becomestronger with a smaller minimum distancebetween UL and DL carriers, since the channelselection filter provides less attenuation of theUL carrier. However, considering the wide-bandcharacteristics of the RF front-end (e.g., mixer),the interference due to receiver nonlinearity ismostly independent of the distance between ULand DL carriers, as illustrated in Fig. 6d. Herethe receiver’s second order intercept point (IIP2)[3] is assumed to be 45 dBm. It should be notedthat the receiver nonlinearity often dominatesthe receiver performance loss (e.g., when theinter-carrier gap is smaller than 45 MHz).

LO COUPLINGIn the case of options 2 and 3, the simultaneousoperation of multiple LOs on the same die maycause severe coupling between LOs. More specif-ically, because of limited isolation between LOs,an LO signal applied to one mixer may alsoappear, at an attenuated level, at the LO input ofthe other mixer. Taking option 3 as an example,at the mixer for carrier 1, other than the desiredLO signal at the frequency of carrier 1 (LO1),several additional spurs may appear, whichinclude the LO signal at the frequency of carrier2 (LO2) and the IM products of these two LOsignals. For example, the latter spurs may occurwhen two LO signals pass through nonlineardevices such as an LO buffer. The relativestrength of these spurs is subject to the receiverimplementation. These spurs appear at the mixer,thereby resulting in unwanted frequency conver-sion. For example, if the inter-carrier spacing isequal to half the duplexer distance, the unwantedfrequency conversion causes interference from aUL carrier to a DL carrier, as illustrated in Fig.5f. In order to avoid severe desensitization, thespur at the IM3 frequency should be about 80 dBsmaller than the desired LO signal. It is worthmentioning that the LO coupling may be depen-dent on the radio frequency planning (e.g.,through the VCO coupling).

HARMONIC REJECTIONThe widely used switched mixer typically leads tonon-negligible conversion at harmonics of theLO center frequency. When it comes to an RFLO, such frequency conversion is not so prob-lematic, since most of the interfering signals atthe harmonics are easily suppressed by theduplexer (receiver) filter. However, option 2cannot rely on the selectivity of a duplexer as theharmonic frequencies of the IF LO may be with-in the DL band. The corresponding frequencyconversion is very similar to the frequency con-version due to LO coupling (Fig. 5f) in the sensethat a couple of spurs occur around the desiredLO signal. Harmonic rejection mixers can beused to mitigate such interference [11]. Themeasurement results on the left side of Fig. 6eshow that it is possible to achieve a harmonicrejection ratio of 85 dB.

IMAGE REJECTIONThe presence of gain and phase imbalance ofreceiver (quadrature) paths leads to limitedimage rejection of the receiver.

In order to avoid

severe desensitiza-

tion, the spur at the

IM3 frequency

should be about

80 dB smaller than

the desired LO

signal. It is worth

mentioning that the

LO coupling may be

dependent on the

radio frequency

planning, e.g.,

through the

VCO coupling.



Figure 6. Impact of receiver impairment: a) transmitter emission with 5 MHz allocation; b)transmitter emission with 1.8 MHz alloca-tion; c) transmitter emission with 180 kHz allocation; d) interference as a function of intercarrier gap; e) harmonic rejection (measure-ments); f) image rejection ratio (measurements).

TX leakage - 1RBTX leakage - 10RBsTX leakage - 25RBsThermal noiseTX phase noiseRX IM2RX phase noise

Distance from UL center frequency (MHz)

(a)

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2636 2641

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Carrier 1

PARK_LAYOUT_Layout 12/3/13 2:56 PM 3


In the case of options 1 and 2, one DL carrieris seen as the image of the other DL carrier withrespect to the center frequency of RF LO (e.g.,LO1 in Options 1 and 2). If two DL carriers haveunequal bandwidth, an unwanted signal adjacentto the narrowband DL carrier may interfere with(a fraction of) the wideband DL carrier, as illus-trated in Fig. 5g. Recall that the unwanted signalmay be substantially stronger than the DL carri-ers, since it may come from a different operator,as mentioned earlier. Thus, the image rejectionof a conventional receiver may not suffice to pro-tect the wanted DL carriers. Note that this is nota problem with two carriers with equal band-width, since the image always comes from theother carrier whose power level is typically simi-lar. It is possible to avoid the interference due tothe image by calibrating the gain and phase ofanalog circuitry, as depicted in [8, 12]. The mea-surement results on the right side of Fig. 6e showthat a digitally calibrated receiver provides morethan 55 dB image rejection ratio. (It is worthmentioning that further improvement of imagerejection ratio is possible, e.g., by digitally com-pensating for gain and phase imbalance [13].) Incontrast, option 3 does not have such an imageissue (regardless of whether two carriers haveequal bandwidth or not), since each carrier isdown-converted by a dedicated LO.

CONCLUSIONAn overview of design challenges of CA-capableterminals is presented in this article. Differentradio architectures are discussed from the per-spective of design trade-off. The simulationresults show that the support of CA generallyleads to significant increase of minimum PAback-off in order to meet the unwanted emissionrequirements. In addition, the radio impairmentsaffecting the receiver performance are discussed,which include transmitter emission, phase noise,receiver nonlinearity, LO coupling, limited har-monic rejection, and limited image rejection. Thesimulation results show that receiver performanceis mostly determined by either transmitter emis-sion or phase noise for a large inter-carrier gap,whereas it is dominated by receiver nonlinearityfor a small inter-carrier gap. The silicon imple-mentation shows significant enhancement of har-monic rejection and image rejection capability.

ACKNOWLEDGMENTThe authors would like to thank Peter Jakobssonand David Duperray from ST-Ericsson for theirinsightful comments.

REFERENCES[1] K. I. Pedersen et al., “Carrier Aggregation for LTE-

Advanced: Functionality and Performance Aspects,”IEEE Commun. Mag., vol. 49, no. 6, June 2011, pp.89–95.

[2] E. Dahlman, S. Parkvall, and J. Sköld, 4G LTE/LTE-Advanced for Mobile Broadband, Elsevier, 2011.

[3] B. Razavi, RF Microelectronics, Prentice-Hall, 1988.[4] 3GPP TS 36.101, “Evolved Universal Terrestrial Radio

Access (E-UTRA); User Equipment (UE) Radio Transmis-sion and Reception,” v. 11.2.0, Sept. 2012.

[5] 3GPP TR 36.823, “Evolved Universal Terrestrial Radio Access(E-UTRA); Carrier Aggregation Enhancement; UE and BSRadio Transmission and Reception,” v0.5.0, Jan. 2013.

[6] 3GPP R4-125599, “MPR Simulations for NC Intra-BandCA,” Oct. 2012.

[7] C. S. Park, “Dependence of Power Amplifier Backoff onResource Allocation for Non-Contiguous Carrier Aggre-gation,” Electronics Letters, vol. 49, no. 15, July 2013,pp. 962–64.

[8] L. Sundström et al., “A Receiver for LTE Rel-11 and BeyondSupporting Non-Contiguous Carrier Aggregation,” Proc.IEEE ISSCC ’13, San Francisco, CA, Feb. 2013.

[9] 3GPP R4-123306, “Reference Sensitivity for Non-Con-tiguous Intra-Band CA,” May 2012.

[10] D. B. Leeson, “A Simple Model of Feedback OscillatorNoise Spectrum,” Proc. IEEE, vol. 54, Feb. 1966, pp.329–30.

[11] L. Sundström et al., “Complex IF Harmonic RejectionMixer for Non-Contiguous Dual Carrier Reception in 65nm CMOS,” to be published (invited paper), IEEE J.Solid-State Circuits.

[12] L. R. Wilhelmsson et al., “Design of a ConfigurableAnalog Receiver Front-End Supporting LTE CarrierAggregation,” Proc. IEEE VTC ’13, Dresden, Germany,June 2013.

[13] C. S. Park and F. S. Park, “Digital Compensation of IQImbalance for Dual-Carrier Double Conversion Receiv-er,” IEICE Trans. Commun., vol. E95-B, no. 5, May2012, pp. 1612–19.

BIOGRAPHIESCHESTER SUNGHCUNG PARK ([email protected])received his Ph.D. degree from the Korea Advanced Insti-tute of Science and Technology (KAIST), Daejeon, in2006. After about two years with Samsung ElectronicsInc., Giheung, Korea, he joined Ericsson Research, USA,where he worked on 3GPP-LTE digital baseband/front-end and participated in 3GPP-LTE standardization of car-rier aggregation and MIMO. Since 2013, he has beenwith Konkuk University, Seoul, Korea, working on algo-rithm and system-on-a-chip (SoC) architecture for digitalsignal processing.

LARS SUNDSTRÖM received his Ph.D. in applied electronicsfrom Lund University, Sweden, in 1995. From 1995 to2000 he was an associate professor at the CompetenceCenter for Circuit Design at the same university wherehis research focused on linear radio transmitters and RFASIC des ign. In 2000, he jo ined Er icsson Researchwhere he presently holds the position of senior special-ist with interests ranging from RF, analog, and mixed-s ignal IC design to radio architectures for cel lu lartransceivers.

ANDERS WALLÉN received his Ph.D. in automatic controlfrom Lund University in 2000. Since then, he has beenwith Ericsson Research, currently holding a masterresearcher position. He has participated in 3GPP standard-ization for both HSPA and LTE, and has primarily workedwith terminal front-end requirements and physical layeralgorithms.

ALI KHAYRALLAH received M.S. and Ph.D. degrees from theUniversity of Michigan, Ann Arbor, and his B.E. degreefrom the American University of Beirut. He is currentlydirector of research at Ericsson in San Jose, California,and has held various research positions with the companysince 1995. Previously, he was on the faculty of the Elec-tr ical Engineering Department at the University ofDelaware.

The simulation

results show that

receiver performance

is mostly determined

by either transmitter

emission or phase

noise for a large

inter-carrier gap,

whereas it is

dominated by

receiver nonlinearity

for a small

inter-carrier gap.


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