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Thyristors and Triacs Power Semiconductor Applications Philips Semiconductors

CHAPTER 6

Power Control with Thyristors and Triacs

6.1 Using Thyristors and Triacs

6.2 Thyristor and Triac Applications

6.3 Hi-Com Triacs

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Using Thyristors and Triacs

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6.1.1 Introduction to Thyristors and Triacs

Brief summary of the thyristor family

The term thyristor is a generic name for a semiconductorswitch having four or more layers and is, in essence, ap-n-p-n sandwich. Thyristors form a large family and it ishelpful to consider the constituents which determine thetype of any given thyristor. If an ohmic connection is madeto the first p region and the last n region, and no otherconnection is made, the device is a diode thyristor. If anadditional ohmic connection is made to the intermediate nregion (n gate type) or the intermediate p region (p gatetype), thedevice is a triode thyristor. If an ohmic connectionismade toboth intermediateregions, thedevice isa tetrodethyristor. All such devices have a forward characteristic ofthe general form shown in Fig. 1.

There are three types of thyristor reverse characteristic:blocking (as in normal diodes), conducting (large reversecurrents at low reverse voltages) and approximate mirrorimageof theforward characteristic (bidirectional thyristors).Reverse blocking devices usually have four layers or lesswhereas reverse conducting and mirror image devicesusually have five layers.

The simplest thyristor structure, and the most common, isthe reverse blocking triode thyristor (usuallysimply referredto as the thyristor or SCR silicon controlled rectifier). Itscircuit symbol and basic structure are shown in Fig. 2.

The most complex common thyristor structure is thebidirectional triode thyristor, or triac. The triac (shown inFig. 3) isable topass current bidirectionallyand is thereforean a.c. power control device. Its performance is that of apair of thyristors in anti-parallel with a single gate terminal.The triac needs only one heatsink, but this must be largeenough to remove the heat caused by bidirectional currentflow. Triac gate triggering circuits must be designed withcare to ensurethat unwantedconduction, ie. loss ofcontrol,does not occur when triggering lasts too long.

Thyristors and triacs are both bipolar devices. They havevery lowon-statevoltages but, because theminority chargecarriers in the devices must be removed before they can

block an applied voltage, the switching times arecomparatively long. This limits thyristorswitching circuits tolow frequency applications. Triacs are used almostexclusivelyatmainssupply frequenciesof50or60Hz,whilein some applications this extends up to the 400Hz supplyfrequency as used in aircraft.

Thevoltage blocking capabilitiesof thyristors and triacs arequite high: the highest voltage rating for the Philips rangeis 800V, while the currents (IT(RMS)) range from 0.8A to 25A.

The devices are available as surface mount components,

or as non-isolated or isolated discrete devices, dependingon the device rating.

Fig. 1 Thyristor static characteristic

Fig. 2 Thyristor circuit symbol and basic structure

Fig. 3 Triac circuit symbol and basic structure

On-statecharacteristic

Off-statecharacteristic

Avalanchebreakdownregion

Reversecharacteristic

Reversecurrent

Forwardcurrent

Reversevoltage

Forwardvoltage

ILIH

V(BO)

I = 0GI > 0G

Anode Anode

Gate

Gate

Cathode Cathode

p

n

p

n

J1J2J3

MT1

MT2

Gate Gate

MT1

MT2

n

n

n

n

p

p

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Thyristor operation

Theoperationof thethyristorcan beunderstood from Fig. 2.When the thyristor cathode is more positive than the anodethen junctions J1 and J3 are reverse biased and thedevice

blocks. When the anode is more positive than the cathode,junctions J1 and J3 are forward biased. As J2 is reversebiased, then the device still blocks forward voltage. If thereverse voltage across J2 is made to reach its avalanchebreakdown level then the device conducts like a singleforward-biased junction.

The two transistor model of Fig. 4 can be used to considerthe p-n-p-n structure of a thyristor as the interconnection ofan npn transistor T1 and a pnp transistor T2. The collectorof T1 provides the base current for T2. Base current for T1is provided by the external gate current in addition to thecollector current from T2. If the gain in the base-collector

loop of T1 and T2 exceeds unity then the loop current canbe maintained regeneratively. When this condition occursthen both T1 and T2 are driven into saturation and thethyristoris said tobe latched. Theanode tocathodecurrentis then only limited by the external circuit.

Fig. 4 Two transistor model of a thyristor

There are several mechanisms by which a thyristor can belatched. The usual method is by a current applied to thegate. This gate current starts the regenerative action in thethyristor and causes the anode current to increase. Thegains of transistors T1 and T2 are current dependent andincrease as the current through T1 and T2 increases. With

increasing anode current the loop gain increasessufficiently such that the gate current can be removedwithout T1 and T2 coming out of saturation.

Thus a thyristor can be switched on by a signal at the gateterminal but, because of the way that the current thenlatches, the thyristor cannot be turned off by the gate. Thethyristor must be turned off by using the external circuit tobreak the regenerative current loop between transistors T1and T2. Reverse biasing the devicewill initiate turn-off oncethe anode current drops below a minimum specified value,called the holding current value, IH.

Thyristor turn-on methods

Turn-on by exceeding the breakovervoltage

When the breakover voltage, VBO, across a thyristor isexceeded, the thyristor turns on. The breakover voltage ofa thyristor will be greater than the rated maximum voltageof the device. At the breakover voltage the value of thethyristor anode current is called the latching current, IL.

Breakover voltage triggering is not normally used as atriggering method, andmostcircuit designs attempt toavoidits occurrence. When a thyristor is triggered by exceedingVBO the fall time of the forward voltage is quite low (about1/20th of the time taken when the thyristor isgate-triggered). As a general rule, however, although athyristor switches faster with VBO turn-on than with gateturn-on, the permitted di/dt for breakover voltage turn-on islower.

Turn-on by leakage current

As the junction temperature of a thyristor rises, the leakagecurrent also increases. Eventually, if the junctiontemperature is allowed to rise sufficiently, leakage currentwould become large enough to initiate latching of theregenerative loop of the thyristor and allow forwardconduction. At a certain critical temperature (above Tj(max))the thyristor will not support any blocking voltage at all.

Turn-on by dV/dt

Any p-n junction has capacitance - the larger the junctionarea the larger the capacitance. If a voltage ramp is appliedacross the anode-to-cathode of a p-n-p-n device, a currentwill flow in the device to charge the device capacitanceaccording to the relation:

If the charging current becomes large enough, the densityof moving current carriers in the device induces switch-on.

Turn-on by gate triggering

Gate triggering is theusual methodof turning a thyristoron.Application of current to the thyristor gate initiates thelatching mechanism discussedin theprevious section. Thecharacteristic of Fig. 1 showed that the thyristor will switchto its on-statecondition with forward bias voltages less thanVBO when the gate current is greater than zero. The gatecurrent and voltage requirements which ensure triggeringof a particular device are always quoted in the device data.As thyristor triggering characteristics are temperaturedependant, the amplitude and duration of the gate pulsemust be sufficient to ensure that the thyristor latches underall possible conditions.

T1

T2

Anode

Cathode

Gate

iA

iG

iC = C.dv

dt(1)

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During gate turn-on, the rate of rise of thyristor anodecurrent dIF/dt is determined by the external circuitconditions. However, the whole active area of the thyristor(or triac) cannot be turned on simultaneously: the areanearest to the gate turns on first, followed by the remainder

of the device. At turn-on it is important that the rate of riseof current does not exceed the specified rating. If dIF/dt isexcessive then only a limited area of the device will havebeen turned on as the anode current increases. Theresulting localised heating of the device will causedegradation and could lead to eventual device failure.

A suitably high gate current and large rate of rise of gatecurrent (dIG/dt) ensures that the thyristor turns on quickly(providing that the gate power ratings are not exceeded)thus increasing the thyristor turn-on di/dt capability. Oncethe thyristor has latched then the gate drive can bereducedor removed completely. Gate power dissipation can alsobe reduced by triggering the thyristor using a pulsed signal.

Triac operation

The triac can be considered as two thyristors connected inantiparallel as shown in Fig. 5. The single gate terminal iscommon to both thyristors. The main terminals MT1 andMT2 are connected to both p and n regions of the deviceand the current path through the layers of the devicedepends upon the polarity of the applied voltage betweenthemain terminals. Thedevice polarity is usually describedwith reference to MT1, where the term MT2+ denotes thatterminal MT2 is positive with respect to terminal MT1.

Fig. 5 Anti parallel thyristor representation of a triac

The on-state characteristic of the triac is similar to that of athyristor and is shown in Fig. 6. Table 1 and Fig. 7

summarise the different gate triggering configurations fortriacs.

Due to the physical layout of the semiconductor layers in atriac, the values of latching current (IL), holding current (IH)and gate trigger current (IGT) vary slightly between thedifferent operating quadrants. In general, for any triac, thelatching current is slightly higher in the second (MT2+, G-)quadrant than the other quadrants, whilst the gate triggercurrent is slightly higher in fourth (MT2-, G+) quadrant.

Fig. 6 Triac static characteristic

Quadrant Polarity of MT2 wrt MT1 Gate polarity

1 (1+) MT2+ G+2 (1-) MT2+ G-3 (3-) MT2- G-4 (3+) MT2- G+

Table 1. Operating quadrants for triacs

Fig. 7 Triac triggering quadrants

For applications where the gate sensitivity is critical andwherethe devicemusttrigger reliablyand evenlyfor appliedvoltages in both directions it may be preferable to use anegative current triggering circuit. If the gate drive circuit isarranged so that only quadrants 2 and 3 are used (i.e. G-operation) then thetriac is neverused in thefourth quadrantwhere IGT is highest.

On-state

Off-state

Reversecurrent

Forwardcurrent

Reversevoltage

Forwardvoltage

ILIH

V(BO)

I = 0GI > 0G

On-state

Off-state

LIHI

(BO)V

I = 0G

I > 0G

T2-

T2+

Quadrant 1Quadrant 2

Quadrant 4Quadrant 3

G+G-

MT2+

MT2-

IG

IG

IG

IG

++

- -

+

-

+

-

MT2

MT1

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For some applications it is advantageous to trigger triacswith a pulsating signal and thus reduce the gate powerdissipation. To ensure bidirectional conduction, especiallywith a very inductive load, the trigger pulses mustcontinueuntil theend ofeach mainshalf-cycle. If singletriggerpulses

are used, one-way conduction (rectification) results whenthe trigger angle is smaller than the load phase angle.

Philips produce ranges of triacs having the same currentand voltage ratings but with different gate sensitivities. Adevicewitha relatively insensitivegate will bemore immuneto false triggering due to noise on the gate signal and alsowill be more immune to commutating dv/dt turn-on.Sensitive gate triacs are used in applications where thedevice is driven from a controller IC or low power gatecircuit.

The diac

It is also worthwhile to consider the operation andcharacteristicsof thediac in thecontext ofmultilayer bipolardevices.The diac ismore strictlya transistor thana thyristor,but has an important role in many thyristor and triactriggering circuits. It is manufactured by diffusing an n-typeimpurity into both sides of a p-type slice to give a twoterminal device with symmetrical electrical characteristics.As shown in the characteristic of Fig. 8, the diac blocksapplied voltages in either direction until the breakovervoltage, VBO is reached. The diac voltage then breaksbackto a lower output voltage VO. Importantdiac parameters arebreakover voltage, breakover current and breakbackvoltage as shown in the figure.

Fig. 8 Diac static characteristic and circuit symbol

Gate requirements for triggering

To a first approximation, the gate-to-cathode junction of athyristor or triac acts as a p-n diode. The forwardcharacteristic is as shown in Fig. 9. For a given thyristortype there will bea spread in forward characteristics of gate

junctions and a spread with temperature.

Fig. 9 Thyristor gate characteristic

The gate triggering characteristic is limited by the gatepower dissipation. Figure 9 also shows the continuous

power rating curve (PG(AV)=0.5W) for a typical device andthe peak gate power curve (PGM(max)=5W). When designinga gate circuit to reliably trigger a triac or thyristor the gatesignal must lie on a locus within the area of certain devicetriggering. Continuoussteady operation woulddemand thatthe 0.5W curve be used to limit the load line of the gatedrive circuit. For pulsed operation the triggering locus canbe increased. If the 5W peak gate power curve is used, theduty cycle must not exceed

Atthe otherendof thescale,the level belowwhichtriggeringbecomes uncertain is determined by the minimum numberof carriers needed in the gate-cathode junction to bring thethyristorinto conduction by regenerative action. The triggercircuit load line must not encroach into the failure to triggerregion shown in Fig. 9 if triggering is to be guaranteed. Theminimum voltage andminimum current to trigger alldevices(VGT and IGT) decreases with increasing temperature. Datasheets for Philips thyristors and triacs show the variation ofVGT and IGT with temperature.

Thyristor commutation

A thyristor turns off by a mechanism known as naturalturn-off, that is, when the main anode-cathode currentdrops below the holding value. It is important to remember,however, that the thyristor will turn on again if the reappliedforward voltage occurs before a minimum time period haselapsed; this is because the charge carriers in the thyristorat the time of turn-off take a finite time to recombine.Thyristor turn-off is achieved by two main methods - selfcommutation or external commutation.

Gate voltage, V G (V)

Gate current, IG(A)

P = 5W

= 0.1

= 1.0

VGT

IGT

Failureto trigger

GM(max)

P = 0.5WG(AV)

Gate power

ratings

Gate-cathodecharacteristic

max

=PG(AV)

PGM=

0.5

5= 0.1 (2)

Reverse

current

Forwardcurrent

Reversevoltage

Forwardvoltage

V(BO)

(BO)V

I(BO)

I(BO)

VO

VO

Breakbackvoltage

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Self CommutationIn self-commutation circuits the thyristor will automaticallyturn off at a predetermined time after triggering. Thethyristor conduction period is determined by a property ofthe commutation circuit, such as the resonant cycle of anLC-circuit or the Volt-Second capability of a saturableinductor. The energy needed for commutation is deliveredby a capacitor included in the commutation circuit.

LC circuit in series with the thyristorWhen the thyristor is triggered, the resulting main currentexcites the resonant circuit. After half a resonant cycle, theLC circuit starts to reverse the anode current and turns thethyristor off. The thyristor conduction interval is half aresonant cycle. It is essential for proper commutation thatthe resonant circuit be less than critically damped. Fig. 10shows the circuit diagram and the relevant waveforms for

this arrangement.

LC Circuit in parallel with the thyristor

Initially the capacitor charges to the supply voltage. Whenthe thyristor is triggered the load current flows but at thesame time the capacitor discharges through the thyristor inthe forward direction. When the capacitor has discharged(i.e.after oneresonant half-cycle of theLC circuit), it beginsto charge in the opposite direction and, when this chargingcurrent is greater than the thyristor forward current, thethyristor turns off. The circuit diagram and commutationwaveforms are shown in Fig. 11.

External commutation

If the supply is an alternating voltage, the thyristor canconduct only during the positive half cycle. The thyristornaturally switches off at the end of each positive half cycle.The circuit and device waveforms for this method ofcommutation are shown in Fig. 12. It is important to ensurethat the duration of a half cycle is greater than the thyristorturn-off time.

Reverse recovery

In typical thyristors the reverse recovery time is of the orderof a few micro-seconds. This time increases with increaseof forward current andalso increasesas theforward currentdecay rate, dIT/dt, decreases.Reverse recovery time is theperiod during which reverse recovery current flows (t1 to t3in Fig. 13) and it is the period between the point at whichforward current ceases and the earliest point at which thereverse recovery current has dropped to 10% of its peakvalue.

Fig. 10 Commutation using a series LC circuit

Fig. 11 Commutation using a parallel LC circuit

RR

leakage

L

CE

+

Ithyristor

R

L

C

E

IR

+ I thyristor

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Reverse recovery current can cause high values of turn-oncurrent in full-wave rectifier circuits (where thyristors areused as rectifying elements) and in certain inverter circuits.It should also be remembered that, if thyristors areconnected in series, the reverse voltage distribution canbe

seriously affected by mismatch of reverse recovery times.

Fig. 12 Thyristor commutation in an a.c. circuit

Fig. 13 Thyristor turn-off characteristics

Turn-off time

Turn-off time is the interval between the instant whenthyristorcurrentreversesand thepointat whichthethyristorcan block reapplied forward voltage (t1 to t4 in Fig. 13). If

forward voltage is applied to a thyristor too soon after themain current has ceased to flow, the thyristor will turn on.The circuit commutated turn-off time increases with:

-junction temperature-forward current amplitude-rate of fall of forward current-rate of rise of forward blocking voltage-forward blocking voltage.

Thus the turn-off time is specified for defined operatingconditions. Circuit turn-off time is the turn-off time that thecircuit presents to thethyristor; it must,of course, be greaterthan the thyristor turn-off time.

Triac commutation

Unlike thethyristor, thetriac canconduct irrespectiveof thepolarity of the applied voltage. Thus the triac does notexperience a circuit-imposed turn-off time which allowseach anti-parallel thyristor to fully recover from itsconducting state as it is reverse biased. As the voltageacrossthe triac passes through zero andstarts to increase,then the alternate thyristor of the triac can fail to block theapplied voltage and immediately conduct in the oppositedirection. Triac-controlled circuits therefore require carefuldesign in order to ensure that the triac does not fail tocommutate (switch off) at the end of each half-cycle asexpected.

It is important to consider the commutation performance ofdevices in circuits where either dI/dt or dV/dt can be large.In resistive load applications (e.g. lamp loads) currentsurges at turn-on or during temporary over-currentconditions may introduce abnormally high rates of changeof current which may cause the triac to fail to commutate.In inductive circuits, such as motor control applications orcircuits where a dc load is controlled by a triac via a bridgerectifier, it is usually necessary to protect the triac againstunwanted commutation due to dv(com)/dt.

The commutating dv(com)

/dt limit for a triac is less than thestatic dv/dt limit because at commutation the recentlyconducting portion of the triac which is being switched offhas introduced stored charge to the triac. The amount ofstored charge depends upon the reverse recoverycharacteristics of the triac. It is significantly affected by

junction temperature and the rate of fall of anode currentprior to commutation (dI(com)/dt). Following high rates ofchange of current the capacity of the triac to withstand highreappliedrates of change of voltage is reduced. Data sheetspecifications for triacs give characteristics showing the

R

i thyristor

IT

IR

V

D

VR

dIT

dt

dVD

dt

t0

t1

t2

t3

t4

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maximum allowable rate of rise of commutating voltageagainst device temperature andrate of fall of anode currentwhich will not cause a device to trigger.

Fig. 14 Inductive load commutation with a triac

Consider the situation when a triac is conducting in onedirection and the applied ac voltage changes polarity. Forthe case of an inductive load the current in the triac doesnot fall to its holding current level until some time later. Thisis shown in Fig. 14. At the time that the triac current hasreached the holding current the mains voltage has risen tosome value and so the triac must immediately block thatvoltage. The rate of rise of blocking voltage followingcommutation (dv(com)/dt) can be quite high.

The usual method is to place a dv/dt-limiting R-C snubberinparallel with the triac.Additionally, because commutatingdv/dt turn-on is dependent upon the rate of fall of triaccurrent, then in circuits with large rates of change of anodecurrent, the ability of a triac to withstand high rates of rise

of reapplied voltage is improved by limiting the di/dt usinga series inductor. This topic is discussed more fully in thesection entitled Using thyristors and triacs.

Conclusions

This article has presented the basic parameters andcharacteristics of triacs and thyristors and shown how thestructure of the devices determines their operation.Important turn-on and turn-off conditions and limitations ofthe devices have been presented in order to demonstrate

the capabilitiesof the devices andshow the designer thoseareas which require careful consideration. The devicecharacteristics which determine gate triggeringrequirements of thyristors and triacs have been presented.

Subsequent articles in this chapter will deal with the use,operation and limitations of thyristors and triacs in practicalapplications, and will present some detailed design andoperational considerations for thyristors and triacs in phasecontrol and integral cycle control applications.

VDWM

-dI/dtdV

com/dt

Time

Time

Time

Supplyvoltage

Loadcurrent

Voltageacrosstriac

Trigger

pulses

Current

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6.1.2 Using Thyristors and Triacs

This chapter is concerned with the uses, operation and

protection of thyristors andtriacs. Twotypes of circuit coverthe vast majority of applications for thyristors and triacs:static switching circuits and phase control circuits. Thecharacteristics anduses of these two types of circuit will bediscussed. Various gate drive circuits and protectioncircuits for thyristor and triacs are also presented. The useofthesecircuitswill enabledesigners tooperate thedevicesreliably and within their specified limits.

Thyristor and triac control techniques

There are two main techniques of controlling thyristors andtriacs - on-off triggering (or static switching) and phasecontrol. In on-off triggering, the power switch is allowed to

conduct for a certain number of half-cycles and then it iskept off for a number of half-cycles. Thus, by varying theratio of "on-time" to "off-time", the average power suppliedto the load can be controlled. The switching device eithercompletely activates or deactivates the load circuit. Inphase control circuits, the thyristor or triac is triggered intoconduction at some point after the start of each half-cycle.Control is achieved on a cycle-by-cycle basis by variationof the point in the cycle at which the thyristor is triggered.

Static switching applications

Thyristors and triacs are the ideal power switching devicesfor many high power circuits such as heaters, enabling the

load to be controlled by a low power signal, in place of arelay or other electro-mechanical switch.

Ina highpower circuit where the power switch may connector disconnect the load at any point of the mains cycle thenlarge amounts of RFI (radio frequency interference) arelikely to occur at the instants of switching. The largevariations in load may also cause disruptions to the supplyvoltage. The RFI and voltage variation produced by highpower switching in a.c. mains circuits is unacceptable inmany environments and is controlled by statutory limits.The limits depend upon the type of environment (industrialor domestic) and the rating of the load being switched.

RFI occurs at any time when there is a step change incurrent caused by the closing of a switch (mechanical orsemiconductor). The energy levels of this interference canbequitehigh in circuitssuch as heating elements.However,if the switch is closed at the moment the supply voltagepasses through zero there is no step rise in current andthus no radio frequency interference. Similarly, at turn-off,a large amount of high frequency interference can becaused by di/dt imposed voltage transients in inductivecircuits.

Circuit-generated RFIcanbe almost completely eliminated

by ensuring that the turn-on switching instants correspondto the zero-crossing points of the a.c. mains supply. Thistechnique is known as synchronous (or zero voltage)switching control as opposed to the technique of allowingthe switching points to occur at any time during the a.c.cycle, which is referred to as asynchronous control.

In a.c. circuits using thyristors and triacs the devicesnaturally switch off when the current falls below the deviceholding current. Thus turn-off RFI does not occur.

Asynchronous control

In asynchronous control the thyristor or triac may betriggeredat a point in the mains voltage other than the zerovoltage crossover point. Asynchronous control circuits areusually relatively cheap but liable to produce RFI.

Synchronous control

In synchronous control systems the switching instants aresynchronised with zero crossings of the supply voltage.They also have the advantage that, as the thyristorsconduct over complete half cycles, the power factor is verygood.This methodofpowercontrol ismostlyused tocontrol

temperature.Therepetition period, T, is adjusted to suit thecontrolled process (within statutory limits). Temperaturerippleis eliminated when therepetitionperiodis made muchsmaller than the thermal time constant of the system.

Figure 1 shows the principle of time-proportional control.RFIand turn-on di/dtarereduced,andthe best power factor(sinusoidal load current) is obtained by triggeringsynchronously. The average power delivered to a resistiveload, RL, is proportional to ton/T (i.e. linear control) and isgiven by equation 1.

where: T is the controller repetition periodton is controller on timeV(RMS) is the rms a.c. input voltage.

Elsewhere in this handbook theoperationof a controller i.c.(the TDA1023) is described. This device is specificallydesigned to implement time-proportional control of heatersusing Philips triacs.

Pout =V(RMS)

2

RL.ton

T(1)

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Fig. 1 Synchronous time-proportional control

Phase control

Phase control circuits are used for low power applications

such as lamp control or universal motor speed control,where RFI emissions can be filtered relatively easily. Thepower delivered to the load is controlled by the timing ofthe thyristor (or triac) turn-on point.

The two most common phase controller configurations arehalf wave control, where the controlling device is a singlethyristorandfullwave control,where thecontrollingdeviceis a triac or a pair of anti-parallel thyristors. These twocontrol strategies are considered in more detail below:

Resistive loads

The operation of a phase controller with a resistive load is

thesimplest situation to analyse. Waveforms for a full wavecontrolled resistive load are shown in Fig. 2. The triac istriggered at angle , and applies the supply voltage to theload. The triac then conducts for the remainder of thepositivehalf-cycle, turning offwhentheanodecurrentdropsbelow the holding current, as the voltage becomes zero at=180.The triac is then re-triggered at angle (180+),andconducts for the remainder of the negative half-cycle,turning off when its anode voltage becomes zero at 360.

The sequence is repeated giving current pulses ofalternating polarity which are fed to the load. The durationof each pulse is the conduction angle , that is (180-).The output power is therefore controlled by variation of thetrigger angle .

For all values of other than =180 the load current isnon-sinusoidal. Thus, because of the generation ofharmonics, the power factor presented to the a.c. supplywill be less than unity except when =0.

For a sinusoidal current the rectified mean current, IT(AV),and the rms current, IT(RMS), are related to the peak current,IT(MAX), by equation 2.

Fig. 2 Phase controller - resistive load

where

From equation 2 the crest factor, c, (also known as the

peak factor) of the current waveform is defined as:

The current form factor, a, is defined by:

Thus, for sinusoidal currents:

For the non-sinusoidal waveforms which occur in a phasecontrolled circuit, the device currents are modified due tothedelaywhichoccurs beforethe powerdeviceis triggered.Thecrest factorof equation4andthe form factorof equation5 can be used to describe variation of the currentwaveshape from the sinusoidal case.

tON

T

Inputvoltage

Triggersignal

Outputcurrent

Trigger

Conduction

angle,

Voltage

Current

Supply

voltage

Triac

Triac

Device

triggers

Trigger

angle,

O = wt

O = wt

O = wt

IT(AV) =

2.IT(MAX)

= 0.637I

T(MAX)

IT(RMS) =

IT(MAX)

2= 0.707I

T(MAX) (2)

IT(MAX) =

VT(MAX)

RL=

2V(RMS)RL

(3)

Crest factor, c =IT(MAX)

IT(RMS)(4)

Form factor, a =IT(RMS)

IT(AV)(5)

a =I

T(RMS)IT(AV)

= 1.111; c =I

T(MAX)IT(RMS)

= 1.414 (6)

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Half wave controller

Figure 3a) shows the simplest type of thyristor half-wavephase controller for a resistive load. The load currentwaveform is given in Fig. 3b). Thevariationof average load

current, IT(AV), rms load current, IT(RMS) and load power overthe full period of the a.c mains, with trigger angle are givenin equation 7.

N.B. When using equation 7 all values of must be inradians. For each case the maximum value occurs when=180 (= radians).

At =180 the crest factor and form factor for a half wavecontroller are given by:

Full wave controller

Figure 4 shows the circuit and load current waveforms fora full-wave controller using two antiparallel thyristors, or atriac, as the controlling device. The variation of rectifiedmean current, IT(AV), rms current, IT(RMS), and load power withtrigger angle are given by equation 9.

N.B. When using equation 9 all value of must be inradians. For each case the maximum value occurs when=180 (= radians).

Fig. 3 Half wave control

Fig. 4 Full wave control

The variation of normalised average current, IT(AV)/IT(AV)max,rms current IT(RMS)/IT(RMS)max, and power, P(out)/P(out)max, for

equations 7 and 9 are plotted in Fig. 5.

Figure 6 shows the variation of current form factor withconduction angle for the half wave controller and the fullwave controller of Figs. 3 and 4.

Fig. 5 Current and power control using conductionangle

a) b)

Trigger

Voltage

Current

Supplyvoltage

Thyristor

Thyristor

IT(MAX)

IT(AV) = IT(AV)max.

(1 cos)2

IT(RMS) = IT(RMS)max.

12sin2

1

2

P(out) = P(out)max.

1

2sin2

IT(AV)max =

IT(MAX)

IT(RMS)max =

IT(MAX)

2

P(out)max =

IT(MAX)2

RL

4 (7

)

a) b)

Trigger

Voltage

Current

Supplyvoltage

Triac

Triac

IT(MAX)

IT(MAX)

a =IT(RMS)

IT(AV)= 1.571; c =

IT(MAX)

IT(RMS)= 2.0 (8)

IT(AV)

IT(AV)max

IT(RMS)

IT(RMS)max

P(OUT)

P(OUT)max

Amplitude

Conduction angle

0 30 60 90 120 150 1800

0.2

0.4

0.6

0.8

1

IT(AV) = IT(AV)max.

(1 cos)2

IT(RMS) = IT(RMS)max.

1

2

sin2

1

2

P(out) = P(out)max.

12sin2

IT(AV)max =

2IT(MAX)

IT

(RMS

)max

=IT(MAX)

2

P(out)max =IT(MAX)2

RL

2(9)

499


16/84


Fig. 6 Variation of form factor with conduction angle

Inductive loads

The circuit waveforms for a phase controller with aninductive load or an active load (for example, a motor) aremore complex than those for a purely resistive load. Thecircuit waveforms depend on the load power factor (whichmay be variable) as well as the triggering angle.

For a bidirectional controller (i.e triac or pair of anti-parallelthyristors),maximumoutput, thatis, sinusoidal loadcurrent,occurs when the trigger angle equals the phase angle.When the trigger angle, , is greater than the load phaseangle, , then the load current will become discontinuousandthe triac (orthyristor)will blocksome portion of the inputvoltage until it is retriggered.

If the trigger angle is less than the phase angle then theload current inone direction will not have fallen back to zeroat the time that the device is retriggered in the oppositedirection. This is shown in Fig. 7. The triac fails to betriggered as the gate pulse has finished and so the triacthen acts as a rectifier. In Fig. 7 the triac is only triggeredby the gate pulses when the applied supply voltage ispositive (1+ quadrant). However, the gate pulses whichoccurone half period later have no effect because the triac

is still conducting in the opposite direction. Thusunidirectional current flows in the main circuit, eventuallysaturating the load inductance.

This problem can be avoided by using a trigger pulse trainas shown in Fig. 8. The triac triggers on the first gate pulseafter the load current has reached the latching current IL inthe 3+ quadrant. The trigger pulse train must cease beforethe mains voltage passes through zero otherwise the triacwill continue to conduct in the reverse direction.

Fig. 7 Triac triggering signals - single pulse

Fig. 8 Triac triggering signals - pulse train

Gate circuits for thyristors and triacs

As discussed in the introductory article of this chapter, athyristor or triac can be triggered into conduction when avoltageofthe appropriatepolarity isapplied acrossthemainterminals and a suitable current is applied to the gate. Thiscan be achieved using a delay network of the type shownin Fig. 9a). Greater triggering stability and noise immunitycan be achieved if a diac is used (see Fig. 9b). This givesa trigger circuit which is suitable for both thyristors andtriacs.

Figure 10 shows several alternative gate drive circuitssuitable for typical triac and thyristor applications. In eachcircuit the gate-cathode resistor protects the device fromfalse triggering due to noise - this is especially importantforsensitivegate devices. Inadditionopto-isolated thyristorand triac drivers are available which are compatible withthe Philips range of devices.

Form

factor

IT(RMS)

I T(AV)

Conduction angle

0 30 60 90 120 150 1800

1

2

3

4

5

6

Half-wave rectifier

Full-wave rectifier

Trigger

Device failsto trigger

Conductionangle

Voltage

Current

Supplyvoltage

Triac

Triac Inductoriron coresaturation

Devicetriggers

Trigger

Conductionangle

Voltage

Current

Supplyvoltage

Triac

Triac

Devicetriggers

Devicetriggers

Fails totrigger

500


17/84


Fig. 10 Alternative triac triggering circuits

1k0

220R

BT145

Load

1k0BT145

Load

180R

10k

12V

BC337

1k0

Load

10k

12V

1k0

Load

10k

12V

BC337

4k7100nF

BT145

BT145BAW62

2:1

a)

b)

Fig. 9 Basic triac triggering circuits

In some applications it may be necessary to cascade asensitive gate device with a larger power device to give asensitive gate circuit with a high power handling capability.A typical solution which involves triggering the smallerdevice (BT169) from a logic-level controller to turn on thelarger device (BT151) is shown in Fig. 11.

Figure 12 shows an isolated triac triggering circuit suitablefor zero voltage switching applications. This type of circuitis also known as a solid state relay (SSR). The function of

the Q1/R2/R3 stage is that the BC547 is on at all instantsin time when the applied voltage waveform is high and thusholds the BT169 off. If the BT169 is off then no gate signalis applied to the triac and the load is switched off.

Fig. 11 Master-slave thyristor triggering circuit

Fig. 12 Opto-isolated triac triggering circuit

R

E

IR

+

R

E

IR

+ BT151BT169

R

IR

+

-

R1R3

R2 R4

BC547

BT169

100R

1K0

BT138100R

100nF

Q1

501


18/84


If the input signal is switched high then the photo-transistorturns on. If this occurs when the mains voltage is high thenQ1 remains on. When the line voltage passes through itsnext zero crossing in either direction the photo transistorensures that Q1 stays off long enough for the BT169 to

trigger. This then turns the triac on. Once the thyristor turnson, the drive circuit is deprived of its power due to the lowervoltage drop of the BT169. The triac is retriggered everyhalf cycle.

Voltage transient protection

Therearethree major sources of transientwhichmayaffectthyristor and triac circuits:

-the mains supply (e.g. lightning)-other mains and load switches (opening and closing)-the rectifying and load circuit (commutation)

In order to ensure reliable circuit operation these transientsmust be suppressed by additional components, removedat source or allowed for in component ratings.

Three types of circuit are commonly employed to suppressvoltage transients - a snubber network across the device,a choke between the power device and external circuit oran overvoltage protection such as a varistor.

Series line chokes

A series choke may be used to limit peak fault currents toassist in the fuse protection of thyristors and triacs. If thechoke is used in conjunction with fuse protection, it mustretain its inductance to very large values of current, and so

for this reason it is usually an air-cored component.Alternatively,if thechokeis only requiredto reducethe dv/dtacross non-conducting devices then the inductance needsonly to be maintained up toquite lowcurrents. Ferrite-coredchokes may be adequate provided that the windings arecapable of carrying the full-load current. Usually only a fewmicrohenries of inductance are required to limit the circuitdi/dt to an acceptable level. This protects the devices fromturning on too quickly and avoids potential devicedegradation.

For instance, a 220V a.c. supply with 20H sourceinductance gives a maximum di/dt of (2202)/20=16A/s.Chokes used to soften commutation should preferably besaturable so as to maintain regulation and avoiddeterioration of the power factor. As their impedancereduces at high current, they have very little effect on theinrush current.

The addition of di/dt limiting chokes is especially importantin triac circuits where the load is controlled via a bridgerectifier. At the voltage zero-crossing points the conductiontransfers between diodes in the bridge network, and therate of fall of triac current is limited only by the strayinductance in the a.c. circuit. The large value ofcommutating di/dt may cause the triac to retrigger due to

commutating dv(com)/dt. A small choke in the a.c circuit willlimit the di(com)/dt to an acceptable level. An alternativetopology which avoids triac commutation problems is tocontrol the load on the d.c. side.

Snubber networks

Snubber networks ensure that the device is not exposed toexcessive rates of change of voltage during transientconditions. This is particularly important when consideringthe commutation behaviour of triacs, which has beendiscussed elsewhere.

Fig. 13 Triac protection

Thefollowing equationscanbe used to calculatethe valuesof the snubber components required to keep the reapplieddv/dt for a triac within the dv(com)/dt rating for that device.The parameters which affect the choice of snubbercomponentsare the value of load inductance, frequency ofthe a.c. supply and rms load current. The value of thesnubber resistor needs to be large enough to damp thecircuitandavoidvoltageovershoots. Thesnubbercapacitorshould be rated for the full a.c. voltage of the system. Thesnubber resistor needs to be rated at 0.5W.

For circuits where the load power factor, cos, 0.7 thesnubber values are given approximately by:

where: L is the load inductancef is the supply frequencyIT(RMS) is the rms device currentdv(com)/dt is the device commutating dv/dt rating.

The presence of a snubber across the device can improvethe turn-on performance of the triac by using the snubbercapacitor discharge current in addition to the load currentto ensure that the triac latches at turn-on. The value of thesnubber resistor must be large enough to limit the peakcapacitor discharge current through the triac to within theturn-on di/dt limit of the device.

Load

SnubberVaristor

Choke

C 25L

fIT(RMS)

dV(com)/dt

2

R = 3LC (9)

502


19/84


Varistor

The use of a metal oxide varistor (MOV), as shown inFig. 13, protects the device from transient overvoltageswhich may occur due to mains disturbances.

Overcurrent protection

Like all other semiconductor devices, triacshave an infinitelife if they are used within their ratings. However, theyrapidly overheat when passing excessive current becausethe thermal capacitance of their junction is small.Overcurrent protective devices (circuit breakers, fuses)must, therefore, be fast-acting.

Inrush condition

Motors, incandescent lamp or transformer loads give riseto an inrush condition. Lamp and motor inrush currents areavoided by starting the control at a large trigger angle.

Transformer inrush currents are avoided by adjusting theinitial trigger angle toa valueroughlyequalto theloadphaseangle. No damage occurs when the amount of inrushcurrent is below the inrush current rating curve quoted inthe device data sheet (see the chapter Understandingthyristor and triac data).

Short-circuit condition

Fuses for protecting triacs should be fast acting, and theamount of fuse I2t to clear the circuit must be less than theI2t rating of the triac. Because the fuses open the circuit

rapidly, they have a current limiting action in the event of ashort-circuit. High voltage fuses exhibit low clearing I2t butthe fuse arc voltage may be dangerous unless triacs witha sufficiently high voltage rating are used.

Conclusions

This paper has outlined the most common uses andapplicationsof thyristorand triac circuits.The type of circuitused depends upon the degree of control required and thenature of the load. Several types of gate circuit and deviceprotection circuit have been presented. The amount of

device protection required will depend upon the conditionsimposed on the device by the application circuit. Theprotection circuits presented here will be suitable for themajority of applications giving a cheap, efficient overalldesign which uses the device to its full capability withcomplete protection and confidence.

503


20/84


21/84


6.1.3 The Peak Current Handling Capability of Thyristors

The ability of a thyristor to withstand peak currents many

times the size of its average rating is well known. However,there is little informationabout thefactors affectingthe peakcurrent capability. This section will investigate the effect ofpulse duration on the peak current capability of thyristors.

Data sheets for thyristors always quote a figure for themaximum surge current that the device can survive. Thisfigureassumesa halfsinepulse witha width of either10 msor 8.3 ms, which are the conditions applicable for 50/60 Hzmains operation. This limit is not absolute; narrow pulseswith much higher peaks can be handled without damagebut little information is available to enable the designer todetermine how high this current is. This section will discusssome of the factors affecting a thyristors peak currentcapability andreview the existing prediction methods. It willgo on to present the results of an evaluation of the peakcurrent handling capabilities for pulses as narrow as 10 sfor the BT151, BT152 and BT145 thyristors. It will alsopropose a method for estimating a thyristors peak currentcapability fora half sine pulsewith a durationbetween10 sand 10 ms from its quoted surge rating.

Energy Handling

Inaddition to themaximum surgecurrent, data sheetsoftenquote a figure called "I2t for fusing". This number is used toselectappopriate fuses fordevice protection. I2t represents

the energy that can be passed by the device withoutdamage. In fact it is not the passage of the energy whichcausesdamage,butthe heating of thecrystal by the energyabsorbed by the device which causes damage.

If the period over which the energy is delivered is long, theabsorbed energy has time to spread to all areas of thedevice capable of storing it - like the edges of the crystal,the plastic encapsulation, the mounting tab and for verylong times the heatsink - therefore the temperature rise inthe crystal is moderated. If, however, the delivery period isshort - say a single half sine pulse of current with a durationof


22/84


Fig. 1 Predicted ITSM multiplying factors

Circuit Description

The circuits used to perform the required measurementswere of the form shown in Fig. 2. They produce half sinepulses of current from the resonant discharge of C via L.Triggering of the device under test (DUT) itself is used toinitiate the discharge. The gate signal used for all the testswas a 100 mA / 1 s pulse fed from a pulse generator insingle-shot mode.

Themagnitudeof thecurrent pulse is adjusted by changing

the voltage to which C is initially charged by varying theoutput of the PSU. The pulse is monitored by viewing thevoltage across R3 on an digital storage oscilloscope. R1and D protect the power supply. R1 limits the current fromthe supply when DUT fails and during the recharging of C.D attempts to prevent any high voltage spikes being fedback into the PSU.

Fig. 2 Surge current test circuit

Pushbutton S1 and resistor R2 are a safety feature. R2keeps C discharged until S1 is pressed. The trigger pulseneeds a button on the pulse generator to be pressed whichmeans both hands are occupied and kept away from thetest circuit high voltages.

Choice of L & C

The width of the half sine pulse from an LC circuit is:

and the theoretical peak value of the current is:

These equations assume that the circuit has no seriesresistance to damp the resonant action which would resultin a longer but lower pulse. Minimising these effects wasconsidered to be important so care was taken during thebuilding of the circuits to keep the resistance to a minimum.To this end capacitors with low ESR were chosen, theinductorswere wound using heavy gauge wire and the loopC / L / DUT / R3 was kept as short as possible.

It was decided to test the devices at three different pulsewidths - 10 s, 100 s and 1 ms - so three sets of L and Cwere needed. The values were selected with the help of aspreadsheet program running on an PC compatiblecomputer.Thevalues which were finally chosen are shownin Table 1. Also given in Table 1 are the theoretical peakcurrents that the L / C combination would produce for ainitial voltage on C of 600 V.

Test Procedure

Asmentionedearlier, thetest methodcalled foreachdeviceto be subjected to a series of current pulses of increasingamplitude. The resolution with which the current capability

is assessed is defined by the size of each increase incurrent. It was decided that steps of approximately 5%would give reasonable resolution.

Experimentation indicated that the clearest indication ofdevice damage was obtained by looking for changes in theoff-state breakdown voltage. So after each current pulsethe DUT was removed from the test circuit and checked ona curve tracer. This procedure did slow the testing but itwas felt that it would result in greater accuracy.

Pulse Width C (F) L (H) Ipeak (A)

10 s 13.6 0.75 2564

100 s 100 10 1885

1 ms 660 154 1244

Table 1. Inductor and Capacitor Values

It was also decided that, since this work was attempting todeterminethe current that a devicecouldsurvive - notwhichkilledit, thefigure actuallyquotedin theresults fora devicescurrent capability would be the value of the pulse prior tothe one which caused damage.

15

14

13

12

11

10

9

8

7

6

5

4

3

2

110us 100us 1ms 10ms

Width of Half Sine Pulse

I t = const.

I t = const.

I t = const.

I t = const.log(1/t)

2

3

4

PeakCurrentMultiplyingFactor

tpulse = L C

Ipeak = V CL

DC PSU

0-600V

DR1 L

CR2

DUT

R3

S1 Vak

Trigger

Ia

Pulse

506


23/84


Fig. 3 Peak current capability measurements

Test ResultsFigure 3 is a graph showing the measured currentcapabilitiesof all of the testeddevices. Table2 summarisesthe measurements bygiving the mean of the results for thethree device types at each of the pulse widths. Table 3expresses the mean values as factors of the device ITSMrating. This table also gives the factors that the variousrules would have predicted for the various pulse widths.

Mean Peak Current Capability (Amps)

Pulse Width BT151 BT152 BT145

10 s 912 1092 1333100 s 595 1021 1328

1 ms 264 490 697

Table 2. Measured Current Capability

Measured Predicted FactorFactor (by Int rule)

Pulse BT BT BT n=2 n=3 n=4 n=Width 151 152 145 log(1/t)

10 s 9.1 5.5 4.4 31.6 10.0 5.6 4.0

100 s 6.0 5.1 4.4 10.0 4.6 3.2 3.2

1 ms 2.6 2.4 2.3 3.2 2.2 1.8 2.2

Table 3. Measured and Predicted ITSM MultiplicationFactors

Interpretation of Results

It had been hoped that the measurements would give clearindication of which of the rules would give the mostaccurate prediction of performance. However, aninspection of Table 3 clearly shows that there is no

correlation between any of the predicted factors and themeasured factors. In fact the variation in the factorsbetween the various device types would indicated that norule based on an Int function alone can give an accurateprediction. This implies that something else will have to be

taken into account.

FurtherstudyofFig. 3 reveals that thedifference in thepeakcurrent capability of the threedevice types is becoming lessas the pulses become shorter. This could be explained bya reduction in the active area of the larger crystals, makingthem appear to be smaller than they actually are. This isconsistent with theknownfact that notall areas ofa thyristorturn on simultaneously - the conduction region tends tospread out from the gate. If the pulse duration is less thanthe time it takes for all areas of the device to turn on, thenthe current flows through only part of the crystal, reducingthe effective size of the device. If the rate at which the

conduction area turns on is constant then the time takenfor a small device to be completely ON is shorter than fora large device. This would explain why the performanceincrease of the BT145 starts falling off before that of theBT151.

Proposed Prediction Method

The above interpretation leads one to believe that theoriginal energy handling rule, which says that I2t is aconstant, may still be correct but that the performance itpredicts will roll off if the pulse duration is less than somecritical value. The equation which was developed to havethe necessary characteristics is:

which simplifies to:-

where tcrit is proportional to - but not necessarily equal to -the time taken to turn on all the active area of the crystaland is calculated from:-

where: A = crystal areaR = constant expressing therate at which the areais turned on.

Preferably, A should be the area of the cathode but thisinformation is not always available. As an alternative thetotal crystal area can be used if the value of R is adjustedaccordingly. This will inevitably introduce an error becausecathode and crystal areas are not directly proportional, butit should be relatively small.

#####

########

#####

@@@

@

@@@@@@

@

@@@@

*** ******

*

*

**

****

10us 100us 1ms 10ms100

1000

Width of half-sine pulse

PeakCurrent(amp

s)

# BT151 @ BT152 * BT145

Ipk = ITSM

0.01

tp

1

2

tp

tp + tcrit

1

2

Ipk = ITSM 0.01tp + tcrit

tcrit = AR

507


24/84


R wasdeterminedempirically tobe approximately 0.02 m2/sUsing this value of R gives the values of tcrit shown in Table3. Using these values in the above equation predicts thatthe peak current handling capability of the BT151, BT152and BT145 would be as shown in Fig. 4.

Device tcrit

BT151 148 sBT152 410 sBT145 563 s

Table 3. Calculated Values of tcrit

Conclusions

The first conclusion that can be drawn from this work is thata thyristor, with average rating of only 7.5A, is capable ofconducting, without damage, a peak current greater than

100timesthis value ina short pulse.Furthermorethepowerrequired to trigger the device into conducting this currentcan be


25/84


6.1.4 Understanding Thyristor and Triac Data

The importance of reliable and comprehensive data for

power semiconductor devices, together with theadvantages of the absolute maximum rating system, isclear. This present article describes the data sheetdescriptions of Philips thyristors and triacs, and aims toenable the circuit designer to use our published data to thefull and to be confident that it truly describes theperformance of the devices.

A brief survey of short-form catalogues is an insufficientmethod of comparing different devices. Published ratingsand characteristics require supporting information to trulydescribe the capabilities of devices; thus comparisonsbetween devices whose performance appears to be similarshould not be made on economic grounds alone.

Manufacturers have been known to quote ratings in sucha way as to give a false impression of the capabilities oftheir devices.

Ratings and characteristics given in published data shouldalways be quoted with the conditions to which they apply,and these conditions should be those likely to occur inoperation. Furthermore, it is important to define the ratingor characteristic being quoted. Only if data is both completeandunambiguouscan a true comparison be made betweenthe capabilities of different types.

Thyristors

Thyristoris a generic term fora semiconductor devicewhichhas four semiconductor layers and operates as a switch,having stable on and off states. A thyristor can have two,three, or four terminals but common usage has confinedthe term thyristor to three terminal devices. Two-terminaldevices are known as switching diodes, and four-terminaldevices are known as silicon controlled switches. Thecommon, or three-terminal, thyristor is also known as thereverse blocking triode thyristor or the silicon controlledrectifier (SCR). Fig. 1 shows the circuit symbol and aschematic diagram of the thyristor. All Philips thyristors arep-gate types; that is, the anode is connected to the metaltab.

The thyristor will conduct a load current in one directiononly, as will a rectifier diode.However, thethyristor will onlyconduct this load current when it has been triggered; thisis the essential property of the thyristor.

Fig. 2 shows the static characteristic of the thyristor. Whena small negative voltage is applied to the device, only asmall reverse leakage current flows. As the reverse voltageis increased, the leakage current increases until avalanchebreakdown occurs. If a positive voltage is applied, thenagain a small forward leakage current flows whichincreases as the forward voltage increases. When the

forward voltage reaches the breakover voltage V(BO),

turn-on is initiatedby avalanchebreakdown andthe voltageacross the thyristor falls to the on state voltage VT.

However, turn-on can occur when the forward(anode-to-cathode) voltage is less than V(BO) if the thyristoris triggered by injecting a pulse of current into the gate. Ifthe device is to remain in the on state, this trigger pulsemust remain until the current through the thyristor exceedsthe latching current IL. Once the on state is established, theholding current IH is the minimum current that can flowthrough thethyristor andstill maintain conduction. Theloadcurrent must be reduced to below IH to turn the thyristor off;for instance, by reducing the voltage across the thyristor

and load to zero.

Fig. 1 Thyristor circuit symbol and basic structure

Fig. 2 Thyristor static characteristic

Thyristors are normally turned on by triggering with a gatesignal but they can also be turned on by exceeding eitherthe forward breakover voltage or the permitted rate of rise

Anode Anode

Gate

Gate

Cathode Cathode

p

n

p

n

On-statecharacteristic

Off-statecharacteristic

Avalanchebreakdownregion

Reversecharacteristic

Reversecurrent

Forwardcurrent

Reversevoltage

Forwardvoltage

ILIH

V

(BO)

509


26/84


of anode voltage dVD/dt. However, these alternativemethods of switching to the conducting state should beavoided by suitable circuit design.

TriacsThe triac, or bidirectional triode thyristor, is a device thatcan be used to pass or block current in either direction. Itis therefore an a.c. power control device. It is equivalent totwothyristors in anti-parallel with a commongate electrode.However, itonly requires oneheatsink compared to the twoheatsinks required for the anti-parallel thyristorconfiguration. Thus the triac saves both cost and space ina.c. applications.

Figure 3 shows the triac circuit symbol and a simplifiedcross-section of the device. The triac has two mainterminalsMT1and MT2(the load connections)anda single

gate. The main terminals are connected to both p and nregions since current can be conducted in both directions.The gate is similarly connected, since a triac can betriggered by both negative and positive pulses.

Fig. 3 Triac circuit symbol and basic structure

Fig. 4 Triac static characteristic

The on state voltage/current characteristic of a triacresembles that of a thyristor. The triac static characteristicof Fig. 4 shows that the triac is a bidirectional switch. Theconditionwhen terminal2 of thetriacis positivewith respectto terminal 1 is denoted in data by the term T2+. If the triac

is not triggered, the small leakage current increases as thevoltage increases until the breakover voltage V(BO) isreached and the triac then turns on. As with the thyristor,however, the triac can be triggered below V(BO) by a gatepulse,providedthat the current through thedeviceexceedsthe latching current IL before the trigger pulse is removed.Thetriac, like thethyristor,has holding current valuesbelowwhich conduction cannot be maintained.

When terminal 2 is negative with respect to terminal 1 (T2-)the blocking and conducting characteristics are similar tothose in the T2+ condition, but the polarities are reversed.The triac can be triggered in both directions by eithernegative (G-) or positive (G+) pulses on the gate, as shownin Table 1. Theactualvaluesof gate trigger current, holdingcurrent and latching current may be slightly different in thedifferent operating quadrants of the triac due to the internalstructure of the device.

Quadrant Polarity of T2 wrt T1 Gate polarity

1 (1+) T2+ G+2 (1-) T2+ G-3 (3-) T2- G-4 (3+) T2- G+

Table 1. Operating quadrants for triacs

Device data

Anode to cathode voltage ratings

The voltage of the a.c. mains is usually regarded as asmooth sinewave. In practice, however, there is a varietyof transients, some occurring regularly and others onlyoccasionally (Fig. 5). Although some transients may beremoved by filters, thyristors must still handle anode tocathode voltages in excess of the nominal mains value.

The following reverse off-state voltage ratings are given inour published data:

VRSM: the non-repetitive peak reverse voltage. This is theallowable peak value of non-repetitive voltage transients,and is quoted with the maximum duration of transient thatcan be handled (usually t < 10ms).

VRRM: the repetitive peak reverse voltage. This is theallowable peak value of transients occurring every cycle.

VRWM: the peak working reverse voltage. This is themaximum continuous peak voltage rating in the reversedirection, neglecting transients. It corresponds to the peaknegative value (often with a safety factor) of the sinusoidalsupply voltage.

MT1

MT2

Gate Gate

MT1

MT2

n

n

n

n

p

p

Reversecurrent

Forwardcurrent

Reversevoltage

Forwardvoltage

IHIL

V(BO)

Blocking

Blocking

ILIH

V(BO)

T2+

T2-

510


27/84


Fig. 5 Diagrammatic voltage waveform showing deviceanode voltage ratings

The forwardoff-state voltages corresponding to VRSM, VRRM

and VRWM are listed below.VDSM: the non-repetitive peak off-state voltage applied inthe forward direction.

VDRM: the repetitive peak off-state voltage applied in theforward direction.

VDWM: the peak working off-state voltage applied in theforward direction.

Both the repetitive and non-repetitive voltage ratings aredetermined partly by the voltage limit that prevents thethyristor being driven into forward or reverse breakdown,and partly by the instantaneous energy (resulting from anincrease in leakage current) that can be dissipated in the

device without exceeding the rated junction temperature.

Whenathyristor isto operate directlyfromthemains supply,it is advisable to choose a device whose repetitive peakvoltage ratings VRRM andVDRM are atleast 1.5 times the peakvalue of the sinusoidal supply voltage. This figure formspart of the device type number; for example BT151-650R,where 650 corresponds to VDRM, VRRM=650V and the finalR (for Reverse) indicates that the anode of the device isconnected to the metal tab.

Anode-to-cathode current ratings

The following current ratings, described by the waveforms

shown in Fig. 6, are given in our published data. Note thatthe suffix T implies that the thyristor is in the on state.

IT(AV): the average value of the idealised mains currentwaveform taken over one cycle, assuming conduction over180. For devices mounted on heatsinks, the IT(AV) ratingshould be quoted for a particular mounting-basetemperature Tmb; our devices are generally characterisedat a mounting-base temperature of at least 85C. A devicecan have an artificially high current rating if the

mounting-base temperature is unrealistically low; ratingswith no associated mounting-base temperature should beregarded with suspicion.

IT(RMS): the rms on-state current. This rating gives the

maximum rms current that the thyristor can handle. It isimportant for applications when the device currentwaveformis describedby a high value form factor. Forsuchconditions the rms current rather than the average currentmay be the limiting rating.

ITRM: the repetitive peak forward current. This rating is thepeak current that can be drawn each cycle providing thatthe average and rms current ratings are not exceeded.

ITSM: the non-repetitive (surge) peak forward current. Thisrating is the peak permitted value of non-repetitivetransients, and depends on the duration of the surge. Ourpublished data quotes the ITSM rating for t=10ms, the

duration of a half-cycle of 50Hz mains. However, somemanufacturers quote ITSM for t=8.3ms (half-cycle of 60Hzmains), and thus surge ratings for devices quoted att=8.3ms should be approximately downrated (multiplied by0.83) before comparing them with t=10ms surge ratings.

The surge rating also depends on the conditions underwhich it occurs. Our data sheets quote ITSM rating under theworst probable conditions, that is, Tj=Tj(max) immediatelyprior to the surge, followed by reapplied VRWM(max)immediately after the surge. An unrealistically high ITSMrating could be quoted if, for example, Tj


28/84


dI/dt: the rate of rise of on-state current permissible aftertriggering. An excessive rate of rise of current causes localheating and thus damage to the device. The rate of rise ofcurrent is determined by both the supply and loadimpedances, and can be limited by additional series

inductance in the circuit.

Fig. 7 Non-repetitive surge current as a function of time

I2t: a dimensional convenience specifying the capability ofa thyristor to absorb energy. This rating is required for theselection of fuses to protect the thyristor against excessivecurrents caused by fault conditions. It is normally only validover the range 3 to 10ms. In our published data, a value isquoted for 10ms, in which case:

The user should match the minimum I2t capability of thethyristor to the worst case I2t let-through of a range ofnominally rated fuses in order to select a fuse that willprotect the device under worst probable conditions.

Values of I2t other than those quoted for 10ms can beestimated by referring to the appropriate published curvesof non-repetitive surge current against time. For example,Fig. 7 is thenonrepetitive surge current curve fora thyristorwhose I2t at 10ms is 800A2s. From Fig. 7, ITS(RMS) at 3ms is470A and therefore I2t at 3ms is given by:

To summarise, when selecting an appropriate fuse thefollowing conditions must be taken into account.

1. The fuse must have an rms current rating equal to, orless than, that of the thyristor it is to protect.

2. The I2t at the rms working voltage must be less thanthat of the thyristor taken over the fuse operating time.

3. The arc voltage of the fuse must be less than the VRSMrating of the thyristor.

Gate-to-cathode ratings

The following gate-to-cathode ratings are given in thepublished data.

VRGM: the gate peak reverse voltage.

PG(AV): the mean gate power, averaged over a 20ms period.

PGM: the peak gate power dissipation.

Thegate-to-cathode power ratings shouldnotbe exceededifover-heatingof thegate-cathodejunctionis tobeavoided.

Temperature ratings

Two temperature ratings are given in the published data.

Tstg: thestorage temperature. Bothmaximumandminimumvalues of the temperature at which a device can be storedare given.

Tj: the junction temperature. This is one of the principalsemiconductor ratings since it limits the maximum powerthat a device can handle. The junction temperature ratingquoted inour published data is the highest value of junctiontemperature at which the device may be continuouslyoperated to ensure a long life.

Thermal characteristics

The following thermal resistances and impedances aregiven in our data.

Rth(j-a): the thermal resistance between the junction of thedevice and ambient (assumed to be the surrounding air).

Rth(j-mb): the thermal resistance between the junction andmounting base of the device.

Rth(mb-h): thethermal resistance between themountingbaseof thedevice and theheatsink (contact thermal resistance).

Zth(j-mb): the transient thermal impedance between thejunction and mounting-base of the device. The value givenin the published data is for non-repetitive conditions and aparticular pulse duration. Under pulse conditions, thermalimpedances rather than thermal resistances should beconsidered. Higher peak power dissipation is permittedunder pulse conditions since the materials in a thyristorhave a definite thermal capacity, and thus the critical

junction temperature will not be reached instantaneously,even when excessive power is being dissipated in thedevice. The published data also contains graphs of Zth(j-mb)against time (for non-repetitive conditions) such as thoseshown in Fig. 8.

0.001 0.003 0.01 0.03 0.1 0.3 1 3 100

100

200

300

400

500

600

Duration (s)

ITS(RMS)

I2t=

i2.dt (1)

=

ITSM

2

2

10.103 (A2s)

I2t(3ms) = I

TS(RMS)2 t

= 4702 3.103

= 662.7A2s

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Fig. 8 Thermal impedance between the junction andmounting-base as a function of time

The values of the various thermal resistances between thethyristor junction and the surroundingsmust be consideredto ensure that the junction temperature rating is notexceeded. The heat generated in a semiconductor chipflows by various paths to the surroundings. Fig. 9 showsthe various thermal resistances to be taken into account inthis process. With no heatsink, the thermal resistance fromthe mounting-base to the surroundings is given by Rth(mb-a).When a heatsink is used, the heat loss direct to thesurroundings from the mounting-base is negligible owingto the relatively high value of Rth(mb-a) and thus:

Fig. 9 Heat flow paths

Rth(mb-a) = Rth(mb-h) + Rth(h-a) (2)

Where appropriate, our published data contains powergraphs such as that in Fig. 10. These characteristics relatethe total power P dissipated in the thyristor, the averageforward current IT(AV), the ambient temperature Ta, and thethermal resistance Rth(mb-a), with the form factor, a, as aparameter. They enable the designer to work out therequired mounting arrangement from the conditions underwhich the thyristor is to be operated.

1E-05 0.0001 0.001 0.01 0.1 1 100.001

0.003

0.01

0.03

0.1

0.3

1

3

10

Time (s)

Z th(j-mb) (K/W) Tj

Tmb

Th

Ta

Rth(j-mb)

Rth(mb-a)

Rth(mb-h)

Rth(h-a)

Fig. 10 Derivation of the appropriate Rth(mb-a) for a given value of IT(AV), aand Tamb

0 5 10 15 200

5

10

15

20

25

30

35

40

45

50

0 20 40 60 80 100 120125

120

115

110

105

100

95

90

85

80

75

a=4

a=2.8 2.2 1.9 1.6

Rth(mb-a)

=0.5K/W1K/W2K/W

3

4

6

10

P(W) Tmb (oC)

IT(AV)

(A) Ta (

oC)

513


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Usually, the characteristics are designed for use in 50Hzsinusoidal applications, when the procedure below shouldbe followed.

1. Determine the values of IT(AV) and IT(RMS) for the relevant

application.2. Determine the form factor, which is given by:

3. Starting from the appropriate value of IT(AV) on a curvesuch as Fig. 10, move vertically upwards to intersectthe appropriate form factor curve (interpolating ifnecessary).

4. This intersection gives the power dissipated in thethyristor on the left-hand axis of the combined graphand the mounting base temperature on the right hand

axis.

5. Moving horizontally across from this intersection to theappropriate value of ambient temperature gives therequired mounting base to ambient thermal resistanceRth(mb-a).

6. The required heatsink thermal resistance Rth(h-a) cannow be calculated from Equation 2 since the mountingbase to heatsink thermal resistance Rth(mb-h) is given inthe published data.

Example

The thyristor to which Fig. 10 applies is operated at an

average forward current IT(AV) of 12A and an rms forwardcurrent IT(RMS) of 19.2A. The maximum anticipated ambienttemperature is 25C. Now, Equation 3 gives,

Figure 10 gives the power as P=20W and themounting-base temperature as Tmb=105C. Also, at thispower and ambient temperature of 25C, Fig. 10 gives thevalue of Rth(mb-a) to be 4C/W. The published data gives thevalue of Rth(mb-h) (usinga heatsink compound) to be 0.2C/Wand then Equation 2 gives

Mounting torque

Two values of mounting torque are given in the publisheddata. A minimum value is quoted below which the contactthermal resistance rises owing to poor contact, and amaximum value is given above which the contact thermalresistance again rises owing to deformation of the tab orcracking of the crystal.

Fig. 11 Forward current vs. forward voltage

The surface of a device case and heatsink cannot beperfectly flat, and thus contact will take place on severalpoints only, with a small air-gap over the rest of the contactarea. The use of a soft substance to fill this gap will lowerthe contact thermal resistance. We recommend the use ofproprietary heatsinking compounds which consist of asilicone grease loaded with an electrically insulating andgood thermal conducting powder such as alumina.

Anode-to-cathode characteristicsThe following anode-to-cathode characteristics areincluded in the published data.

IR: the reverse current. This parameter is given forthe worstprobable conditions; that is, the reverse voltageVR=VRWM(max) and a high Tj.

ID: the off-state current. This parameter is again given forthe worst probable conditions; that is, the forward voltageVD=VDWM(max) and a high Tj.

IL: the latching current (Fig. 2). This parameter is quoted ata particular value of junction temperature.

IH: the holding current (Fig. 2). This parameter is quoted ata particular value of junction temperature.

VT: the forward voltage when the thyristor is conducting.This parameter is measured at particular values of forwardcurrent and junction temperature.The junction temperatureis usually low (Tj=25C, for example) since this is the worstcase. The measurement must be performed under pulseconditions to maintain the low junction temperature. Thepublished data also contains curves of forward currentagainst forward voltage, usually for two values of the

junction temperature: 25C and Tj(max) (Fig. 11).

0 0.2 0.4 0.6 0.8 1.0

0

1.0

2.0

3.0

IF(A)

VF(V)

MAX

TYP

25 C

150 C

a =IT(RMS)

IT(AV)(3)

a =19.2

12= 1.6

Rth(h a) = 4 0.2 = 3.8 C/W

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Fig. 12 Definition of rate of rise of off-state voltagedVD/dt

dV/dt: the rate of rise of off-state voltage that will not triggerany device. This parameter is given at maximum values of

junction temperature Tj(max) and forward voltageVD=VDRM(max).

The values of dVD/dt quoted in our published data arenormally specified assuming an exponential waveform.This facilitates the design of RC snubber circuits for deviceprotection when required. Fig. 12 illustrates the definitionof dVD/dt. The final voltage applied to the device VDM ischosenasVDRM(max) andthejunction temperatureis Tj=Tj(max).

Fig. 12 shows that dVD/dt is given by the expression:

where T is the exponential time constant.

The dVD/dt capability of a thyristor increases as the junctiontemperature decreases. Thus curvessuch as those shownin Fig. 13a) are provided in the published data so thatdesigners can uprate devices operated at lower junctiontemperatures.

The dVD/dt characteristic can also be increased byoperating the device at a low supply voltage. Thus thepublished data also contains curves such as Fig. 13b)which shows how dVD/dt increases as the ratio VDM/VDRMmax decreases. Note that VDM is unlikely to be greater than2/3VDRM(max) (usually owing to the restriction of VDWM(max)) andtherefore the fact that dVD/dt approaches zero as VDMincreases above the value of 2/3VDRM(max) does not causeproblems.

dVDdt

VDM

0.63VDM

T 2T 3T 4T Time

dVD

dt=

0.63VDM

T

= 0.63 2/3VDRM(max)T

=0.42VDRM(max)

T(V/s)

a) Junction temperature, Tj b) Applied voltage, VD

Fig. 13 Derating of maximum rate of rise of off-state voltage

0 20 40 60 80 1000

200

400

600

800

1,000

1,200

1,400

Expl

dVDdt

Expl

dVDdt

Ratingpoint

Ratingpoint

Tj

( o C) VD /VDRM(max)(%)

20 40 60 80 100 120 1400

250

500

750

1,000

1,250

1,500

1,750

2,000

515


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a) Minimum VGT that will trigger all devices b) Minimum IGT that will trigger all devices

Fig. 14 Gate characteristics vs. junction temperature

-50 0 50 100 150 Tj ( C)

VGT (V)3

2

1

0 0

50

100

150

0 50 100 150 -50

IGT (mA)

Tj ( C)Tj ( C)

Gate-to-cathode characteristics

The following gate-to-cathode characteristics are given inthe published data.

VGT: thegate-to-cathodevoltage that will trigger alldevices.This characteristic should be quoted for particular valuesof applied voltage VD and low junction temperature.

IGT: the gate-to-cathode current that will trigger all devices.This characteristic should be quoted for the sameconditions given above.

A gate drive circuit must be designed which is capable ofsupplyingat least therequiredminimum voltageandcurrentwithout exceeding the maximum power rating of the gate

junction. Curves such as those shown in Fig. 14 (whichrelate the minimum values of VGT and IGT for safe triggeringto the junction temperature) are provided in data. Thefollowing design procedure is recommended to construct agate drive circuit load-line on the power curves shown inFig. 15.

1. Determine the maximum average gate powerdissipation PG(AV) from the published data (normally0.5W, 1.0W, or 2.0W) and then use the appropriatechoice of x-axis scaling in Fig. 15.

2. Estimate the minimum ambient temperature at whichthe device will operate, and then determine theminimum values of VGT and IGT from curves such asFigs. 14a) and 14b) in the published data. Note that itis assumed that at switch-on Tj=Ta.

3. Determine the minimum open-circuit voltage of thetrigger pulse drive circuit: this is the first co-ordinate onthe load line at IG=0.

4. Using the appropriate horizontal scaling for the device(PG(AV)=0.5W, 1.0W or 2.0W), plot a second point on thepower curve whose co-ordinates are given by VGT(min)and 5IGT(min). Construct a load line between these twopoints. The slope of this load gives the maximumallowable source resistance for the drive circuit.

5. Check the power dissipation by ensuring that the loadline must not intersect the curve for the maximum peakgate powerPGM(max) which is theoutermost (=0.1)curveofFig. 15.Theloadlinemust also notintersect thecurvewhich represents the maximum average gate powerPG(AV) modified by the pulse mark-space ratio, where:

For instance, in Fig. 15, for a thyristor with PG(AV)=1W,the =0.25 curve canbe used for a gate drive with a 1:3mark-space ratio giving an allowable maximum gatepower dissipation of PGM(max)=4W.

An illustration of how the above design procedure operatesto give an acceptable gate drive circuit is presented in thefollowing example.

Example

A thyristor has the VGT/Tj and IGT/Tj characteristics shownin Fig. 14 and is rated with PG(AV)=0.5W and PGM(max)=5W. Asuitable trigger circuit operating with max=0.25,VGT(min)=4.5V, IGT(max)=620mA and Ta(min)=-10C is to bedesigned. Determine its suitability for this device.

PGM(max) =

PAV

(5)

516


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Fig. 15 Gate circuit design procedure - power curves

0

2.5

5.0

7.5

10.0

12.5

15.0

17.5

20.0

22.5

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0

0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0

0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0

= 0.100

= 0.111

= 0.125

= 0.143

= 0.167

= 0.200

= 0.250

= 0.333= 0.500

= 1.000

Gate current, IG

(A)

Gate voltage, VG

(V)

(PG(AV)=0.5W)

(PG(AV)=1.0W)

(PG(AV)=2.0W)

A

B C

1. Select the top x-axis scale of Fig. 15 (PG(AV)=0.5W).2. From Fig. 14, VGT(min)=1.75V, and IGT(min)=66mA.

3. At minimum supply voltage, the open-circuit gatevoltage is 4.5V, giving point A in Fig. 15. Point B isplotted at the co-ordinates VGT(min) and 5xIGT(min), that isat 1.75V and 330mA, and load line ABC is constructedas shown. Note that point C is the maximum currentrequired at IG=570mA and is within the capabilityof thedrive circuit.

4. As required the load line does not intersect the PG(max)(=0.1). The gate drive duty cycle, , is 0.25. ThereforePGM(max) = PG(AV)/ = 0.5/0.25 = 2W. As required, the loadline ABC does not intersect the

=0.25 curve.

Switching characteristics

Two important switching characteristics are usuallyincludedin ourpublisheddata.They arethegate-controlledturn-on time tgt (divided into a turn-on delay time, td, and arise time, tr) and the circuit-commutated turn-off time, tq.

Gate-controlled turn-on time, tgt

Anode current does not commence flowing in the thyristorat the instant that the gate current is applied. There is aperiodwhich elapses between the application of the trigger

pulse and the onset of the anode current which is knownas the delay time td (Fig. 16). The time taken for the anodevoltage to fall from 90% to 10% of its initial value is knownas the rise time tr. The sum of the delay time and the risetime is known as the gate-controlled turn-on time tgt.

Thegate controlled turn-on time depends on theconditionsunder which it is measured, and thus the followingconditions should be specified in the published data.

-Off-state voltage; usually VD=VDWM(max).-On-state current.-Gate trigger current; high gate currents reduce tgt.-Rate of rise of gate current; high values reduce tgt.

-Junction temperature; high temperatures reduce tgt.

Circuit-commutated turn-off time

When a thyristor has been conducting and isreverse-biased, it does not immediately go into the forwardblocking state: minority charge carriers have to be clearedaway by recombination and diffusion processes before thedevice can block reapplied off-state voltage. The time fromthe instant that the anode current passes through zero totheinstant that the thyristor is capable of blocking reappliedoff-state voltage is the circuit-commutated turn-off time tq(Fig. 17).

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Fig. 16 Thyristor gate-controlled turn-on characteristics

Fig. 17 Thyristor turn-off characteristics

The followingconditions should be given when

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