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Chapter 1:INTRODUCTION

In 1889 Tesla and Ferraris published a paper on methods of producing polyphase currents, and

the former exhibited a crude type of three phase motor at the Frankfurt Exhibition in 1891. Since

then induction machines have made steady progress in their development and applications.

Today three phase induction motors is an important class of electric motors which finds wide

applicability in industry and farm sector. In modern day era of ergonomics energy efficient

motors have a pivotal role to offer.

In this project an effort has been made to study the effects of variation of machine efficiency

with variation in design parameters such as stator core length (L) and average value of air gap

flux density and to optimize the performance of the motor such that it is in compliance with the

IE4 standards. Studies reveal that the losses occurring in the machine may depend either directly

or indirectly on these parameters, thus directly affecting efficiency of the motor.

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Chapter 2:CONVENTIONAL APPROACH TO INDUCTION

MOTOR DESIGN

We have to relate the output of the machine with its main dimensions. Lets use the following

nomenclature:-

Eph=Induced EMF per phase in volts

Iph=Current per phase

Tph=No of turns per phase

=Flux per pole in the airgap

P=Number of poles

Kw=Winding factor

Bav=Average value of the flux density in the airgap

ac=Ampere conductor per meter on the armature periphery

D=Armature diameter or stator bore diameter(m)

L=Stator core length

ns=Synchronous speed in r.p.s.

=Full load efficiency

cos =Full load power factor

=Pole pitch= D/P

2.1 OUTPUT EQUATION

2.1.1 Specific magnetic loading(Bav)

Bav=P / DL

2.1.2 Specific electric loading(ac)

ac=3x2xIphxTph/ D

Further we have the relation, f=nsxP/2

Figure 2Error! No text of specified style in document..1 Stator and rotor slots

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2.1.3 kVA Equation

kVA = 3 x 4.44 x Kw x ns x P/2 x Bav x (DL / P) x ac x D / (3 x 2)x 10-3

=(1.11 2 Kw Bav ac ) D2 L

=CoD2L ns

where, Co=1.11 2 Kw Bav ac x 10

-3

2.1.4 Efficiency and power factor

kVA=h.p. x0.746/( x cos )

Thus to get kVA from h.p. or kW, & cos is to be highly satisfied or suitably assumed.

2.1.5 Calculation of main dimensions

kVA=(1.11 2 Kw Bav ac ) D2 L

2.1.6 Seperation of D & L

D2L= kW /Co x ns

The D2L product obtained above has to be split up into two components D & L. To

incorporate features like cost, efficiency etc.

Table 2.1: Choice of L/

Design Feature Ratio L/ 1. Minimum cost 1.5 to 2

2. Good efficiency 1.5

3. Good ovrall design 1

Assuming Ventilating ducts =nvd

length of each duct= bvd

Fig. 2.2 Main dimensions D & L

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Gross iron Length (Ls) = L - nvd x bvd Assuming Iron factor =ki

Net Iron length (Li)= Ls x ki

Core inner diameter (D) = (D2L/L)0.5

2.2 STATOR DESIGN

Stator of an induction motor consists of stator core and stator slots.

Stator slots: in general two types of stator slots are employed in induction motors viz, open clots and semi closed slots. Operating performance of the induction motors depends upon the shape of the slots and hence it is important to select suitable slot for the stator slots.

(i) Open slots: In this type of slots the slot opening will be equal to that of the width of the slots as shown in Fig 10. In such type of slots assembly and

repair of winding are easy. However such slots will lead to higher air gap

contraction factor and hence poor power factor. Hence these types of slots are

rarely used in 3 induction motors.

(ii) Semi closed slots: In such type of slots, slot opening is much smaller than the width of the slot as shown in Fig 10 and Fig 11. Hence in this type of slots assembly of windings is more difficult and takes more time compared to open slots and hence it is costlier. However the air gap characteristics are better compared to open type slots.

(iii) Tapered slots: In this type of slots also, opening will be much smaller than the slot width. However the slot width will be varying from top of the slot to bottom of the slot with minimum width at the bottom as shown in Fig. 2.

(i) Open type (ii) Semiclosed type (iii) Tapered type

Fig. 2.3 Different types type slots

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2.2.1 Stator winding design

2.2.1.1. Turns per phase

m=Bav x DL/P

Es=4.44 Kws f m Ts

Ts=Es / (4.44 Kws f m Ts)

2.2.1.2. Stator conductor section

Stator current per phase Is=kVA x 1000/(3 x Es)

Cross sectional area of each conductor as=Is/s

2.2.2. Stator slot design

2.2.2.1. Number of stator slots

Let Ss be the number of stator slots

Stator slot pitch yss=D/Ss

Number of conductors per slot Zss=3x2xTs/Ss

2.2.2.2. Size of stator slots

Approx.area per slot= Copper section per slot/Space factor

=Zss aa / Space factor

The value of space factor varies from 0.25 to 0.4. Lower value to be selected for

higher voltage machines in order to allow more space for insulation.

2.2.2.3.Stator slot insulation

The slot liner is latheroid or micanite paper having thickness from 0.75mm to 4mm

thick micanite seperators between the two layers.

2.2.3. Stator teeth design

Once the dimension of the slots is fixed, the tooth dimension is also fixed. But a

check is applied to see that the flux densityin the teeth is within the required range.

The value of flux density in the tooth should lie between 1.3 to 1.7Wb/m2.

Bts= m/((Ss/P) x Li x Wts)

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2.2.4. Depth of stator core

dcs=depth of stator core behind the slot

dss=depth of stator slot

Do=outside diameter of stator core lamination

Area of cross section of stator core = m/(2Bcs)

where Bcs= flux density in the stator core

dcs= m/(2 x Bcs x Li)

Do=D+2dss+2dcs

2.2.5. Losses in stator teeth:

The following steps explain the calculation of iron loss in the stator teeth

(i) Calculate the area of cross section of stator tooth based on the width of the tooth at

1/3rd

height and iron length of the core as A'ts= b'ts x li m2

(ii) Calculate the volume all the teeth in stator Vts = A'ts x hts x Ss m3

(iii) Compute the weight of all the teeth based on volume and density of the material as

Wts = Vts x density. ( density of the material can be found in DDH) (7.8 x 10-3

kg/m3)

(iv) Corresponding to the operating flux density in the stator teeth of the machine iron loss

per kg of the material can be found by referring to the graph on pp179 of DDH.

(v) Total iron losses in teeth= Iron loss /kg x weight of all teeth Wts.

Fig. 2.4 : Flux density vs core loss

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2.2.6. Losses in stator core

Similar to the above calculation of iron loss in teeth, iron loss in stator core can be estimated.

(i) Calculate the area of cross section of the core as Acs = dcs x li m2

(ii) Calculate the mean diameter of the stator core below the slots as Dmcs= D + 2 hts + dcs

(iii) Compute the volume of stator core as Vcs = Acs x Dmcs m3

(iv) Calculate the weight of the stator core as Wcs = Vcs x density

(v) Corresponding to the operating flux density in the stator core of the machine iron loss per kg of the material can be found by referring to the graph on pp 179 of

DDH.

(vi) Total iron losses in core = Iron loss /kg x weight of core Wcs

Total iron losses in induction motor = Iron loss in stator core + iron losses in stator teeth.

In addition friction and windage loss can be taken into account by assuming it as 1- 2%

of the output of the motor.

Hence total no load losses = Total iron losses + Friction and windage

2.3. ROTOR DESIGN

There are two types of rotor construction. One is the squirrel cage rotor and the other is the slip ring rotor. Most of the induction motor are squirrel cage type. These are having the advantage of rugged and simple in construction and comparatively cheaper. However they have the disadvantage of lower starting torque. In this type, the rotor consists of bars of copper or aluminum accommodated in rotor slots. In case slip ring induction motors the rotor complex in construction and costlier with the advantage that they have the better starting torque. This type of rotor consists of star connected distributed three phase windings.

Between stator and rotor is the air gap which is a very critical part. The performance parameters of the motor like magnetizing current, power factor, over load capacity, cooling and noise are affected by length of the air gap. Hence length of the air gap is selected considering the advantages and disadvantages of larger air gap length.

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Advantages

(i) Increased overload capacity (ii) Increased cooling (iii) Reduced unbalanced magnetic pull (iv) Reduced in tooth pulsation (v) Reduced noise

Disadvantages

(i) Increased Magnetising current (ii) Reduced power factor

Magnetising current and power factor being very important parameters in deciding the performance of induction motors, the induction motors are designed for optimum value of air gap or minimum air gap possible. Hence in designing the length of the air gap following empirical formula is employed.

Air gap length lg = 0.2 + 2DL mm

2.3.1. Number of slots

Proper numbers of rotor slots are to be selected in relation to number of stator slots otherwise undesirable effects will be found at the starting of the motor. Cogging and Crawling are the two phenomena which are observed due to wrong combination of number of rotor and stator slots. In addition, induction motor may develop unpredictable hooks and cusps in torque speed characteristics or the motor may run with lot of noise. Let us discuss Cogging and Crawling phenomena in induction motors.

Crawling: The rotating magnetic field produced in the air gap of the will be usually

nonsinusoidal and generally contains odd harmonics of the order 3rd

, 5th

and 7th

. The third

harmonic flux will produce the three times the magnetic poles compared to that of the

fundamental. Similarly the 5th

and 7th

harmonics will produce the poles five and seven times

the fundamental respectively. The presence of harmonics in the flux wave affects the torque

speed characteristics. The Fig. 16 below shows the effect of 7th

harmonics on the torque

speed characteristics of three phase induction motor. The motor with presence of 7th

harmonics is to have a tendency to run the motor at one seventh of its normal speed. The 7th

harmonics will produce a dip in torque speed characteristics at one seventh of its normal

speed as shown in torque speed characteristics. Cogging: In some cases where in the number of rotor slots are not proper in relation to

number of stator slots the machine refuses to run and remains stationary. Under such

conditions there will be a locking tendency between the rotor and stator. Such a phenomenon

is called cogging.

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Hence in order to avoid such bad effects a proper number of rotor slots are to be selected in relation to number of stator slots. In addition rotor slots will be skewed by one slot pitch to minimize the tendency of cogging, torque defects like synchronous hooks and cusps and noisy operation while running. Effect of skewing will slightly increase the rotor resistance and increases the starting torque. However this will increase the leakage reactance and hence reduces the starting current and power factor.

Selection of number of rotor slots: The number of rotor slots may be selected using the following guide lines. (i) To avoid cogging and crawling: (a)Ss Sr (b) Ss - Sr 3P (ii) To avoid synchronous hooks and cusps in torque speed characteristics P, 2P,

5P. (iii) To noisy operation Ss - Sr 1, 2, (P 1), (P 2)

2.3.2. Rotor Bar Current Bar current in the rotor of a squirrel cage induction motor may be determined by comparing the mmf developed in rotor and stator.

Hence the current per rotor bar is given by

Ib = ( Kws x Ss x Z's ) x I'r / ( Kwr x Sr x Z'r ) ;

where

Kws winding factor for the stator, Ss number of stator slots, Z's number of conductors / stator slots, Kwr winding factor for the rotor, Sr number of rotor slots, Z'r number of conductors / rotor slots and I'r equivalent rotor current in terms of stator current

I'r = 0.85 Is

where is stator current per phase.

2.3.3. Cross sectional area of Rotor bar Sectional area of the rotor conductor can be calculated by rotor bar current and assumed value of current density for rotor bars. As cooling conditions are better for the rotor than the stator higher current density can be assumed. Higher current density will lead to reduced sectional area and hence increased resistance, rotor cu losses and reduced efficiency. With increased rotor resistance starting torque will increase. As a guide line the rotor bar current density can be assumed between 4 to 7 Amp/mm

2.

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Hence sectional area of the rotor bars can be calculated as Ab = Ib /b mm2. Once the cross

sectional area is known the size of the conductor may be selected form standard table.

2.3.4. Shape and Size of the Rotor slots

Generally semiclosed slots or closed slots with very small or narrow openings are employed for the rotor slots. In case of fully closed slots the rotor bars are force fit into the slots from the sides of the rotor. The rotors with closed slots are giving better performance to the motor in the following ways: (i) As the rotor is closed the rotor surface is smooth at the air gap and hence the

motor draws lower magnetizing current. (ii) reduced noise as the air gap characteristics are better (iii) increased leakage reactance (iv) reduced starting current (v) Over load capacity is reduced (vi) Undesirable and complex air gap characteristics. From the above it can be

concluded that semiclosed slots are more suitable and hence are employed in rotors.

2.3.5. Copper loss in rotor bars

Knowing the length of the rotor bars and resistance of the rotor bars cu losses in the rotor bars can be calculated.

Length of rotor bar lb = L + allowance for skewing Rotor bar resistance = 0.021 x lb / Ab Copper loss in rotor bars = Ib

2 x rb x number of rotor bars.

2.3.6. End Ring Current

All the rotor bars are short circuited by connecting them to the end rings at both the end rings. The rotating magnetic field produced will induce an emf in the rotor bars which will be sinusoidal over one pole pitch. As the rotor is a short circuited body, there will be current flow because of this emf induced. The distribution of current and end rings are as shown in Fig. 17 below. Referring to the figure considering the bars under one pole pitch, half of the number of bars and the end ring carry the current in one direction and the other half in the opposite direction. Thus the maximum end ring current may be taken as the sum of the average current in half of the number of bars under one pole.

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Fig. 2.5 : currents in cage rotor bars and end rings

Maximum end ring current Ie(max) = ( Number rotor bars / pole) Ib(av)

= x Sr/P x Ib/1.11

Hence rms value of Ie = 1/22 x Sr/P x Ib/1.11

= 1/ x Sr/P x Ib/1.11

2.3.7. Area of end ring

Knowing the end ring current and assuming suitable value for the current density in the end rings cross section for the end ring can be calculated as

Area of each end ring Ae = Ie / e mm2,

current density in the end ring may be assume as 4.5 to 7.5 A/mm2

.

2.3.8. Copper loss in End Rings

Mean diameter of the end ring (Dme) is assumed as 4 to 6 cms less than that of the rotor.

Mean length of the current path in end ring can be calculated as lme = Dme. The

resistance of the end ring can be calculated as

re = 0.021 x lme / Ae

Total copper loss in end rings = 2 x Ie2 x re

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2.3.9. Equivalent Rotor Resistance

Knowing the total copper losses in the rotor circuit and the equivalent rotor current equivalent rotor resistance can be calculated as follows.

Equivalent rotor resistance r'r = Total rotor copper loss / 3 x (Ir

' )

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2.3.10. Flux density in rotor tooth

It is required that the dimension of the slot is alright from the flux density consideration. Flux density has to be calculated at 1/3

rd height from the root of the teeth. This flux

density has to be limited to 1.8 Tesla. If not the width of the tooth has to be increased and width of the slot has to be reduced such that the above flux density limitation is satisfied. The flux density in rotor can be calculated by as shown below. Diameter at 1/3rd height Dr' = D - 2/3 x htr x 2

Slot pitch at 1/3rd height = 'r = x Dr' /Sr

Tooth width at this section = b'tr = 'sr bsr

Area of one rotor tooth = a'tr = b'tr x li

Iron length of the rotor li = (L- wd x nd)ki, ki = iron space factor

Area of all the rotor tooth / pole A'tr = b't x li x Sr /P

Mean flux density in rotor teeth B'tr = / A'tr

Maximum flux density in the rotor teeth < 1.5 times B'tr

2.3.11. Depth of stator core below the slots

Below rotor slots there is certain solid portion which is called depth of the core below slots. This depth is calculated based on the flux density and flux in the rotor core. Flux density in the rotor core can be assumed to be between 1.2 to 1.4 Tesla. Then depth of the core can be found as follows.

Flux in the rotor core section c =

Area of stator core Acr = /2Bcr

Area of stator core Acr = Li x dcr

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Hence, depth of the core dcr = Acr / Li

Inner diameter of the rotor can be calculated as follows Inner diameter of rotor = D - 2lg - 2htr 2 dcr

2.4. ESTIMATION OF OPERATING CHAHRACTERSTICS

2.4.1. No load current

No load current consists of two components:

(i)Loss component of the current, Ie

(ii)Magnetizing component of current, Im

The loss component is in quadrature with magnetizing component of current.

Io=(Ie2+Im

2).5

2.4.2. Loss component

This component of current is responsible for supplying the iron losses in the magnetic

circuit. Hence this component can be calculated from no load losses and applied voltage.

Iron loss component of current Iw= Total no load losses / ( 3 x phase voltage)

2.4.3. Magnetising current

Magnetising current of an induction motor is responsible for producing the required

amount of flux in the different parts of the machine. Hence this current can be calculated

from all the magnetic circuit of the machine. The ampere turns for all the magnetic circuit

such as stator core, stator teeth, air gap, rotor core and rotor teeth gives the total ampere

turns required for the magnetic circuit. The details of the magnetic circuit calculations are

studied in magnetic circuit calculations. Based on the total ampere turns of the magnetic

circuit the magnetizing current can be calculated as Magnetizing current Im= p AT60 / (2.34 kw Tph ) where p no of pairs of poles,

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2.4.4. No load Power Factor:

No load power factor of an induction motor is very poor. As the load on the machine

increases the power factor improves. No load power factor can be calculated knowing the

components of no load current.

No load power factor cos0 = Iw / I0

2.4.5. Efficiency

Pin=Input power

Pout=Output power

W=Total losses=Total no load losses + Total copper losses

%efficiency=Pout*100/Pin

=Pout*100/(Pout+W)

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Chapter 3: ENERGY EFFICIENT MOTORS

Energy-efficient motors, also called premium or high- efficiency motors, are 2 to 8%

more efficient than standard motors. Motors qualify as "energy-efficient" if they meet or

exceed the efficiency levels listed in the National Electric Manufacturers Association's

(NEMA's) MG1-1993 publication.

Energy-efficient motors owe their higher performance to key design improvements and

more accurate manufacturing tolerances. Lengthening the core and using lower-electrical-

loss steel, thinner stator laminations, and more copper in the windings reduce electrical

losses. Improved bearings and a smaller, more aerodynamic cooling fan further increase

efficiency.

Advantages of energy efficient motors

Saves energy & money

Near uniform efficiency from 50% to 100% of full load ensuring

energy savings even at part load conditions also.

Short payback period

Substantial savings after payback period

Fig. 3.1 : %Efficiency vs %Loading

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Design of Premium Efficiency Motors needs special knowledge, experience and test

facilities, equipped with precision instrumentation. The task of design is, to get the

efficiency up by minimizing and balancing the single losses, especially those created in

the stator coils, the stator iron (magnetizing) and the losses within the rotor by slip. In

comparison to standard electrical motors compliant e.g. to IE1 for IE3 motor

manufacturing, more iron and copper material are used. IE3 motors are heavier and

physically bigger, than IE1 motors. Typically use of higher slot fill in the copper

winding, use of thinner laminations of improved steel properties, reducing the air gap,

better design of cooling fan, use of special and improved bearings etc. can ensure higher

efficiency in the motors.

The high electrical conductivity of copper versus other metallic conductors enhances the

electrical energy efficiency of motors. Increasing the mass and cross section of copper in

a coil increases the electrical energy efficiency of the motor. Copper motor rotors, a new

technology designed for motor applications where energy savings are prime design

objectives, are enabling general-purpose induction motors to meet and exceed National

Electrical Manufacturers Association (NEMA) premium efficiency standards.

IEC 60034-30 specifies electrical efficiency classes for single-speed, three-phase, 50 Hz

and 60 Hz, cage-induction motors that:

have 2, 4, or 6 poles (3,000; 1,500; and 1,000 RPM at 50 Hz)

have rated output between 0.75 and 375 kW

have a rated voltage up to 1000 V

are rated on the basis of either duty type S1 (continuous duty) or S3 (intermittent

duty) with a rated cyclic duration factor of 80% or higher

The standard also mentions a future level above IE3 to be called IE4 Super Premium

Efficiency although these products are not commercially available yet and might go

beyond AC induction motor technology. There are still lower efficiency motors in use

now (i.e. Eff3), but they will not be used anymore in the new classification.The following

motors are excluded from the new efficiency standard:

Motors made solely for inverter operation

Motors completely integrated into a machine (pump, fan, or compressor) that

cannot be tested separately from the machine

Rotor losses in IE3 systems are considerably reduced by using copper instead of

aluminium as the conductor material for the squirrel cage. The slip under load, which is

proportional to the rotor losses, is significantly decreased compared with aluminium

motors. Unlike aluminium motors, IE3 motors with a copper rotor do not require an

increased amount of iron or need merely a moderate increase. Other measures can also be

taken to save energy in IE3 motors.

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There is a 34% energy efficiency difference between IE1 and IE3 standard motors,

but the differences, and the absolute level of efficiency, depend on the output of the

motor relative to its rating.

Fig 3.2 : %efficiency vs output for different IE classes

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Chapter 4: SELECTION OF ELECTRIC AND MAGNETIC

MATERIALS FOR THE DESIGN OF ENERGY EFFICIENT

MOTORS

4.1. SELECTION OF CONDUCTOR MATERIAL FOR ELECTRIC CIRCUIT

The high electrical conductivity of copper is an important design factor that helps to

improve the energy efficiency of motors. This is important because motors and motor-

driven systems are very significant consumers of electricity, accounting for 43%-46% of

all global electricity consumption and 69% of all electricity used by industry.

Electric motors do not transfer 100% of the input electrical energy into kinetic

mechanical energy. A certain percentage of electrical energy is lost during the

conversion to mechanical energy. These losses, which are manifested as electrical power

losses (waste heat due to the electrical resistance of the windings, conductor bars and end

rings), magnetic core losses, stray load losses, mechanical losses, and brush contact

losses, reduce what is known as the energy efficiency of motors. The electrical power

losses account for more than half of a motors total losses.

This is a problem for several reasons. First, inefficient electric motors waste electrical

energy, thereby increasing electrical demand and associated electricity costs required to

power motors. Second, when electricity is generated by oil- or coal-fed power plants, the

burning of fossil fuels produces carbon footprints from the usage of natural resources and

the emissions of greenhouse gases. Electrical energy losses from inefficient motors,

therefore, waste precious natural resources, cause increased emissions of greenhouse

gases, and increase operating costs (i.e., increases utility bills). Third, waste heat from

inefficient motors increases maintenance and decreases the life of the motor.

A well-designed motor can convert over 90% of its input energy into useful power for

decades. When the efficiency of a motor is raised by even a few percentage points, the

savings, in kilowatt hours (and therefore in cost), are enormous.

The main parts of an AC induction motor are the fixed housing body (stator), a rotating

assembly (rotor), and electromagnets consisting of coils of copper or aluminium wire

around a core of magnetic steel.

Copper and aluminium can both be used in the stator coils, although copper coils are the

standard as they are more flexible and they enhance motor electrical efficiencies due to

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their higher electrical conductivity. In standard induction motors, instead of being wound

in coils, the rotor conductors are die-cast in the shape of a squirrel cage within a core of

magnetic steel. Aluminium die-cast rotors are the standard material but copper die-

casting of rotors is an improved new technology that is increasingly used to enhance

motor energy efficiency. Induction motors can be designed with wound-rotor motors

instead of a squirrel-cage. In a wound-rotor motor, the rotor winding is made of many

turns of insulated wire.

Other advantages to using copper rather than aluminium in AC motors include:

Lower coefficient of expansion for copper: aluminium will creep and move

approximately 33% more than copper.

Higher tensile strength for copper: copper is 300% stronger than aluminium and

thus able to withstand high centrifugal force and the repeated hammering from

currentinduced forces during each start.

Higher melting point of copper: copper can better withstand thermal cycling over

the life of the motor.

Copper has the second highest electrical conductivity of all metals (5.96 107

Siemens/meter at 20C) and is much more affordable. Copper is commonly used in

motors, including the highest quality motors because of its high electrical conductivity.

Copper is an excellent metal to use for a motor's coils because: 1) it has less electrical

resistance than almost any other non-precious metal; 2) it is easily made into wires; 3) it

is not too expensive; 4) it can perform and survive at high temperatures; and 5) it can

easily be recycled when the motor needs to be replaced.

In general, older, standard-efficiency motors have higher losses than premium motors

that meet more current energy standards. One of the design elements of premium motors

is the reduction of heat losses due to the electrical resistance of the conductors. To

improve the electrical energy efficiencies of induction-type motors, one design

consideration is to reduce load loss by increasing the cross section of the copper coils.

Increasing the mass of copper in a coil increases the electrical energy efficiency of the

motor.

The electrical efficiency of motors can be improved by replacing the standard aluminium

electrical conductor in the motor rotor with copper, which has a much higher electrical

conductivity. Until recently, die-cast motor rotors were produced only from aluminium

while researchers worked on solving technological issues with copper pressure die-

casting. Today, copper pressure die-casting is a proven technology and thousands of die-

cast copper motor rotors are produced annually for motor applications where energy

savings are prime design objectives.

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The use of copper in place of aluminium for conductor bars and end rings of induction

motor rotors results in improvements in motor energy efficiency due to a significant

reduction in I2R losses. Motor modeling by a number of manufacturers has demonstrated

that motors with copper rotors yield overall rotor loss reductions from 15 to 20%

compared to aluminium.

The advantages of motors with copper motor rotors on an equivalent basis with

aluminium include the following:

Motors have longer lives: they generate less heat and reduce thermal stresses,

including those on insulation, which enable them to operate longer.

Motors are smaller: the increased electrical conductivity of the copper rotor

material plus the need for a smaller volume of steel enables the motors to be

shorter in length.

Motors have 15% higher energy efficiency ratings, so therefore consume less

energy.

Motors have lower overall manufacturing costs.

4.2. SELECTION OF MATERIAL FOR MAGNETIC CIRCUIT

Electrical steel, also called lamination steel, silicon electrical steel, silicon steel, relay

steel or transformer steel, is specialty steel tailored to produce certain magnetic

properties, such as a small hysteresis area (small energy dissipation per cycle, or low

core loss) and high permeability.

The material is usually manufactured in the form of cold-rolled strips less than 2 mm

thick. These strips are called laminations when stacked together to form a core. Once

assembled, they form the laminated cores of transformers or the stator and rotor parts

of electric motors. Laminations may be cut to their finished shape by a punch and die, or

in smaller quantities may be cut by a laser, or by wire erosion. When low carbon steel is

alloyed with small quantities of silicon, the added volume resistivity helps to reduce

eddy current losses in the core. Silicon steels are probably of the most use to designers

of motion control products where the additional cost is justified by the increased

performance. These steels are available in an array of grades and thicknesses so that the

material may be tailored for various applications. The added silicon has a marked

impact on the life of stamping tooling, and the surface insulation selected also affects

die life. Silicon steels are generally specified and selected on the basis of allowable core

loss in watts/lb . The grades are called out, in increasing order of core loss by M

numbers, such as M19, M22, M27, M36 or M43, with each grade specifying a

maximum core loss. (Note that this means that material can be substituted up , as

M19 for M36, but not vice versa.) The higher M numbers (and thus higher core losses)

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are progressively lower cost, although only a few percent is saved with each step down

in performance. M19 is probably the most common grade for motion control products,

as it offers nearly the lowest core loss in this class of material, with only a small cost

impact, particularly in low to medium production quantities.

In addition to grade, there are a number of other decisions to make regarding silicon

steels. These are:

1. Semi vs. Fully processed material,

2. Annealing after stamping,

3. Material Thickness,

4. Surface insulation.

Fully processed material is simply material which has been annealed to optimum

properties at the steel mill. Semi processed material always requires annealing after

stamping in order to remove excess carbon as well as to stress relieve. The better

grades of silicon steel are always supplied fully processed while semi processed is

available only in grades M43 and worse. The designer considering semi processed M43

should evaluate Low Carbon Steel which may provide equivalent performance at lower

cost.

4.2.1. Interpolation of core losses for M22 steel using MATLAB

We have the following set of points for different values of specific losses

corresponding to different values of flux density.

Table 4.1: B vs P

B(Wb/m2) P(W/kg)

0.2 0

1.0 1.2

1.2 3.0

Since it can be assumed that the core losses vary quadratically with change in B,

one can apply second order curve fitting techniques to obtain a continuously

varying core loss curve between P and B.

The MATLAB code for this interpolation procedure is as under

x=[.2 1 1.5]; y=[0 1.2 3]; c=polyfit(x,y,2) B=0:.2:2; W=polyval(c,B) plot(B,W) title('INTERPOLATION OF CORE LOSSES FOR M22')

22

xlabel('B(Wb/m2)'); ylabel('P(W/kg)');

Fig. 4 : Interpolated core loss curve for M22 steel

23

Chapter 5 : DEVELOPMENT OF ALGORITHM FOR OPTIMAL

DESIGN OF THREE PHASE INDUCTION MACHINE

Fig. 5 : Flow diagram of optimal design of Induction machine

24

5.1. MATLAB CODE FOR OPTIMAL DESIGN OF INDUCTION MACHINE

%INDUCTION MACHINE DESIGN

%Output equation Vph=input('Vph='); %phase voltage P=input('P='); %Poles Bav=input('Bav='); %Average airgap flux density Kw=input('Kw='); %winding factor f=50; %frequency ns=2*f/P; %Synchronous speed in r.p.s. D=input('D='); %Armature diameter(m) Yp=(pi*D)/P; %Pole pitch L=1.6*Yp %Stator core length(m) Iph=(5*746)/(.82*.91*3*Vph) %Current per phase flux=Bav*pi*D*L/P; %Flux per pole Tph=Vph/(4.44*50*.966*flux) %Turns per phase ac=(3*2*Iph*Tph)/(pi*D); %Specific electrical loading Co=1.11*(pi*pi)*Kw*Bav*ac*10^-3; %Output coefficient kVA=Co*(D*D)*L*ns %Output equation

%STATOR DESIGN

Kws=Kw; %Stator winding factor Ts=Tph; %Stator turns Is=Iph; %Stator current per phase dels=input('dels(current density)='); %Stator current density as=Is/dels; %Cross section area of stator

conductor Ss=input('Ss'); %Stator slots yss=(pi*D/Ss); %Stator slot pitch

Lmts=2*L+2.3*Yp+.24; %Length of mean turn of stator

winding rho=2.1*10^-8; %Resistivity of stator conductor

material rs=rho*Lmts*Ts/as %Stator winding resistance per

phase Ks=input('Ks='); %Stacking factor Li=Ks*L; %Net iron length Wts=input('Wts='); %Width of stator teeth Bts=flux/((Ss/P)*Li*Wts) %Flux density in stator tooth dcs=input('dcs='); %Depth of stator core behind the

slot Bcs=(flux/2)/(dcs*Li) %Flux density in stator core dss=input('dss='); %Depth of stator slot Do=D+2*(dss+dcs); %Outside diameter of stator core

lamination Wos=input('Wos(slot opening)'); %Stator slot opening(mm) Lg=(1/1000)*(.2+2*sqrt(D*L)) %Airgap length ratios=Wos/(2*Lg); %Wos/2Lg ratio Kcs=(2/pi)*(atan(ratios)-(1/ratios)*log10(sqrt(1+ratios*ratios))); %Carter's coefficient

25

Kgs=yss/(yss-Kcs*Wos); %Gap contraction factor for

stator slots

%ROTOR DESIGN Sr=input('Sr='); %Rotor slots Dr=D-2*Lg %Rotor diameter ysr=(pi*Dr/Sr); %Rotor slot pitch at the airgap Nb=Sr; %Number of rotor bars Ib=.85*Is*Ts/Nb %Rotor current per phase delb=input('delb(current density)='); %Rotor bar current density ab=Ib/delb %Rotor conductor area Lb=L+(2*.012)+.01; %Rotor bar length rb=2.1*10^-8*Lb/ab %Resistance of each bar Prb=Sr*Ib*Ib*rb %Total copper loss in bars %End ring Ie=Sr*Ib/(pi*P) %End ring current dele=input('dele(end ring density)='); %End ring current density ae=Ie/dele %area of end ring de=input('de(depth of end ring)=') %depth of end ring dsr=input('dsr='); %Depth of rotor slot Deo=Dr-2*dsr; %Outer diameter of end ring Dei=Deo-2*de; %Inner diameter of end ring De=(Deo+Dei)/2; %Mean diameter of end ring re=(2.1*10^-8)*pi*De/ae; %Resistance of each end ring Pre=2*Ie*Ie*re %Total copper loss in end ring Wtr=input('Wtr='); %Width of rotor tooth Btr=flux/((Sr/P)*Li*Wtr); %Flux density in rotor tooth Wor=input('Wor='); %Rotor slot opening(mm) ratior=Wor/(2*Lg); %Wor/2Lg ratio Kcr=(2/pi)*(atan(ratior)-(1/ratior)*log10(sqrt(1+ratior*ratior))); %Carter's coefficient Kgr=ysr/(ysr-Kcr*Wor); %Gap contraction factor for rotor

slots dcr=input('dcr='); %Depth of rotor core behind the

slot Bcr=Bcs; %Rotor core density

%OPERATING CHARACTERSTICS %Calculation of magnetizing current % 1.For airgap Kg=Kgs*Kgr; %Gap contraction factor for slots Ag=pi*D*L/P; %Area of airgap B60=1.36*Bav; %Flux density at 6odeg angle from

interpolar axis ATg=800000*B60*Kg*Lg; %MMF for airgap % 2.For stator teeth B=Bts; if(B1.49&&B

26

% 3.For stator core lcs=(1/3)*(pi)*(D+2*dss+dcs)/P; %Length of magnetic path through

stator core B=Bcs; if(B1.49&&B

27

%Stator copper losses Wcus=3*Is*Is*rs %Stator coppper losses

%Rotor copper losses Wcur=Prb+Pre %Rotor copper losses

Wcu=Wcus+Wcur %Total copper losses

%TOTAL LOSSES Wfw=input('f/w losses='); %Frictional and windage losses W=Wi+Wcu+Wfw %Total losses

%PERFORMANCE EVALUATION Rs=7.419; %Total resistance referred to

stator side Xs=23.8678; %Total reactance referred to

stator side Zs=(Rs*Rs+Xs*Xs)^.5; %Standstill impedance Isc=Vph/Zs; %Short circuit current angsc=72.73 efficiency=(5*746)/((5*746)+W) %efficiency at full load s=(Wcur)/(5*746+Wcur+Wfw) %full load slip Tst=3*Vph*Vph*2.055/(50*pi*Zs*Zs) %Starting torque

28

5.2. DESIGN DATA SHEET

5.2.1 Rating

1. Full load output 3.73 kW

2. Line voltage 440 V

3. Supply frequency 50 Hz

4. Phase 3

5. Connection

6. Poles 4

7. Synchronous r.p.s. 25

8. Full load efficiency .91

9. Full load power factor .82

10. Full load current per phase 3.7869

11. Full load line current 6.56

12. Specific magnetic loading 0.54

13. Output coefficient 135.882

5.2.2. Stator

1. Type of lamination 0.5 mm Lohys

2. Winding type Mush

3. Connection

4. Voltage per phase 440 V

5. Flux per pole 5.9 mWb

6. Turns per phase 348

7. Number of slots 36

8. Slots per pole per phase 3

9. Winding factor 0.955

10. Slot pitch 9.2 mm

11. Conductors per slot 10

12. Conductor area 0.9467 mm2

13. Bare diameter 1.2 mm

14. Insulated diameter 1.315 mm

15. Current density 4 A/mm2

16. Length of mean turn 0.6936 m

17. Resistance of stator winding per phase 5.3639 ohm

18. Stator copper loss at full load 230.3287 W

19. Depth of slot 15 mm

29

20. Depth of core behind slot 25 mm

21. Outer diameter of laminations 185 mm

5.2.5. Rotor

1. Length of airgap .43 mm

2. Rotor diameter 104 mm

3. Number of slots 28

4. Rotor type Cage

5. Number of bars 28

6. Number of bars per slot 1

7. Slot pitch 11.7 mm

8. Rotor bar current 40.055 A

9. Rotor bar area 10 mm2

10. Current density in bar 4A/mm2

11. Length of each bar 165.9 mm

12. Resistance of each bar 0.348 miliohm

13. Copper loss in bars 15.6144 W

14. End ring current 89.1390 A

15. Section area 14.857 mm2

16. Resistance of each ring 0.261 miliohm

17. Copper loss in two end rings 4.1495 W

18. Total rotor copper loss 19.764 W

19. Rotor resistance referred to stator 2.0555 ohm

20. Depth of slot 17.6 mm

21. Depth of core behind slot 23.1 mm

22. Inner diameter of rotor 63.3 mm

5.2.4. Performance analysis

1. Total resistance referred to stator 7.419 ohm

2. Total leakage reactance referred to stator 23.8678 ohm

3. Impedance at standstill 25 ohm

4. Short circuit current 17.6 A

5. Short circuit power factor .297

6. Short circuit angle 72.73 deg

30

5.2.5. Efficiency

1. Total full load losses 365.9342 W

2. Output at full load 3730 W

3. Input at full load 4095.93 W

4. Full load efficiency 91.07%

5. Starting torque 12.1629 Nm

31

Chapter 6: CONCLUSIONS

The efficiency of a three phase induction machine can be significantly improved if the proper choices of the magnetic and electric materials are made. Materials like M22 have very low core

loss components, thus contributing to the improved performance of the machine. With the help

of an optimizing algorithm, we have been able to achieve efficiency as high as 91.05%. However

it is still not in compliance with the latest IE4 (Premium efficiency) standards, which state that

the efficiency of an energy efficient motor(4-P and 3.7 kW rating) should be more than 92%.

In order to further enhance the machine performance and increase the efficiency up to 92%,

some modifications in the conventional design procedure need to be made. Use of a better

magnetic material core can certainly be of great advantage. It helps in reducing the losses in the

core components of the motor. Another approach to improve the machine performance can be

fine tuning of the dimensions, with the help of modern state of art technologies available today.

More the precision in the design and fabrication , better will be the performance of the machine

that can be achieved.

32

REFERENCES

1. Principles of Electrical Machine DesignR. K. Agarwal

2. A Course in Electrical Machine Design A. K. Sawhney

3. Performance and Design of A C Machines M. G. Say

4. www.phasemotorparts.com (December 7,2012)

5. http://en.wikipedia.org/wiki/Copper_in_energy_efficient_motors (May 4,2013)

6. http://motorsummit.ch/data/files/MS_2010/ms_swiss_10/5_anibal.pdf (May 4, 2013)

7. http://www.motorsystems.org/files/otherfiles/0000/0038/MEPS_Guide_1st_Edition_

February_2009.pdf (May 6, 2013)

8. http://beeindia.in/energy_managers_auditors/documents/guide_books/3Ch2.pdf

(May 7, 2013)

9. http://www.energy.ca.gov/process/pubs/motors.pdf(May 7,2013)

10. http://en.wikipedia.org/wiki/Electrical_steel (May 8, 2013)