soft-switched bidirectional buck-boost converters 2017/apec 17 2... · soft-switched bidirectional...

7
Soft-Switched Bidirectional Buck-Boost Converters Yungtaek Jang and Milan M. Jovanović Delta Products Corporation Power Electronics Laboratory 5101 Davis Drive, Research Triangle Park, NC, USA Abstract— A bidirectional buck-boost converter with a new soft-switching active-snubber cell that reduces switching losses is introduced. Soft-switching cell consists of an active snubber switch, a snubber inductor, and a two-winding transformer with associated magnetizing current reset circuit. The soft- switching cells enable the buck and boost rectifier to turn off with a controlled turn-off rate of their current to minimize corresponding reverse-recovery losses. In addition, in the introduced soft-switching cell, the power-controlling buck and boost switches turn on with zero-voltage switching (ZVS) and the snubber switches turn off with zero-current switching (ZCS). The performance of the proposed bidirectional converter was evaluated on a 5-kW prototype exchanging energy between a 400-V bus and a battery with voltage range between 200 V and 300 V. The 100-kHz prototype circuit exhibits the maximum full-load efficiency of 99.1% in the boost- mode and 98.2% in the buck-mode operation. I. INTRODUCTION Bidirectional converters are increasingly being used in power systems with energy-storage capabilities, such as “smart-grid” and automotive applications where they are employed to condition charging and discharging of energy- storage devices, such as batteries and super-capacitors. For example, in automotive applications, isolated bidirectional dc- to-dc converters are used in electric vehicles (EVs) to provide bidirectional energy exchange between the high-voltage (HV) battery and the low-voltage (LV) battery, while non-isolated dc-dc bidirectional converters are typically employed to optimize the traction inverter performance by pre-regulating its input voltage and providing energy regeneration. Because a battery’s operating voltage range depends on the battery’s state of charge, achieving high efficiency across the entire operating voltage range of the battery is a major design challenge in bidirectional converter designs. Non-isolated bidirectional dc-dc converters are almost exclusively implemented by buck-boost converter topology [1]-[14]. At higher power levels, the continuous-conduction- mode (CCM) operation is preferred over discontinuous- conduction-mode (DCM) operation because of better performance. As described in [15], the major limitations of CCM operation of the high-voltage, high-power buck and boost converters at high frequencies are related to switching losses caused by the reverse recovery of rectifiers and capacitive turn-on switch loss of switches due to their “hard” switching. Generally, in unidirectional buck and boost converter, the reverse-recovery-related losses can be virtually eliminated by using SiC or GaN rectifiers instead of more cost effective fast-recovery Si rectifiers. Since in bidirectional buck-boost converter both switches consists of a combination of a controllable switch and an antiparallel rectifier, SiC or GaN rectifier can only be used if it is co-packaged with an IGBT, or by employing emerging SiC and GaN MOSFET switches [16]. The IGBT implementation is limited to a relative low frequency due to a limited switching speed of IGBTs, which increases the size of the converter; whereas, for the time being, the SiC or GaN switch implementation is not attractive primarily due to increased cost. Today’s most cost effective high-frequency implementation that employs high- voltage Si MOSFETs is only possible if the reverse-recovery- related losses of the slow parasitic antiparallel body diode are significantly reduced. In this paper, active-snubber methods that offer reduced switching losses of semiconductor switches in the bidirectional buck-boost converter are described. The major feature of these active-snubber methods that are extensions of the active-snubber method described in [17] is soft switching of all semiconductor switches. Specifically, the rectifiers are turned off with a controlled turn-off rate of their current to minimize their reverse-recovery losses, the power-controlling switch is turned on with zero-voltage switching (ZVS), and the auxiliary switch in the active snubber is turned off with zero-current switching (ZCS). Because of fully soft-switched operation, proposed circuits exhibit improved efficiency and EMI performance compared to their conventional “hard”- switched counterparts and also enable the employment of semiconductor switches with a relatively slow antiparallel diode (rectifier) at high frequencies. II. SOFT-SWITCHED BIDIRECTIONAL BUCK-BOOST CONVERTERS Figure 1 shows the proposed soft-switched bidirectional buck-boost converter. As shown in Fig. 1, the converter includes two soft-switching cells, each consists of active switch S A, inductor LS, isolation transformer TR, and reset- voltage circuit DA-CR-RR-DB. In Fig. 1, switches SA1 and SA2 are unidirectional current switches that can carry current only in one direction. If bidirectional current switches such as MOSFETs are used, than a rectifier in series with the switch needs to be employed to prevent conduction of its antiparallel diode. Sources V1 and V2 can be any kind of DC power sources or their combinations that can deliver and store (receive) electric energy. 978-1-5090-5366-7/17/$31.00 ©2017 IEEE 287

Upload: truongkhanh

Post on 14-Apr-2018

228 views

Category:

Documents


0 download

TRANSCRIPT

Soft-Switched Bidirectional Buck-Boost Converters

Yungtaek Jang and Milan M. Jovanović

Delta Products Corporation Power Electronics Laboratory

5101 Davis Drive, Research Triangle Park, NC, USA

Abstract— A bidirectional buck-boost converter with a new soft-switching active-snubber cell that reduces switching losses is introduced. Soft-switching cell consists of an active snubber switch, a snubber inductor, and a two-winding transformer with associated magnetizing current reset circuit. The soft-switching cells enable the buck and boost rectifier to turn off with a controlled turn-off rate of their current to minimize corresponding reverse-recovery losses. In addition, in the introduced soft-switching cell, the power-controlling buck and boost switches turn on with zero-voltage switching (ZVS) and the snubber switches turn off with zero-current switching (ZCS). The performance of the proposed bidirectional converter was evaluated on a 5-kW prototype exchanging energy between a 400-V bus and a battery with voltage range between 200 V and 300 V. The 100-kHz prototype circuit exhibits the maximum full-load efficiency of 99.1% in the boost-mode and 98.2% in the buck-mode operation.

I. INTRODUCTION

Bidirectional converters are increasingly being used in power systems with energy-storage capabilities, such as “smart-grid” and automotive applications where they are employed to condition charging and discharging of energy-storage devices, such as batteries and super-capacitors. For example, in automotive applications, isolated bidirectional dc-to-dc converters are used in electric vehicles (EVs) to provide bidirectional energy exchange between the high-voltage (HV) battery and the low-voltage (LV) battery, while non-isolated dc-dc bidirectional converters are typically employed to optimize the traction inverter performance by pre-regulating its input voltage and providing energy regeneration. Because a battery’s operating voltage range depends on the battery’s state of charge, achieving high efficiency across the entire operating voltage range of the battery is a major design challenge in bidirectional converter designs.

Non-isolated bidirectional dc-dc converters are almost exclusively implemented by buck-boost converter topology [1]-[14]. At higher power levels, the continuous-conduction-mode (CCM) operation is preferred over discontinuous-conduction-mode (DCM) operation because of better performance. As described in [15], the major limitations of CCM operation of the high-voltage, high-power buck and boost converters at high frequencies are related to switching losses caused by the reverse recovery of rectifiers and capacitive turn-on switch loss of switches due to their “hard” switching. Generally, in unidirectional buck and boost converter, the reverse-recovery-related losses can be virtually eliminated by using SiC or GaN rectifiers instead of more

cost effective fast-recovery Si rectifiers. Since in bidirectional buck-boost converter both switches consists of a combination of a controllable switch and an antiparallel rectifier, SiC or GaN rectifier can only be used if it is co-packaged with an IGBT, or by employing emerging SiC and GaN MOSFET switches [16]. The IGBT implementation is limited to a relative low frequency due to a limited switching speed of IGBTs, which increases the size of the converter; whereas, for the time being, the SiC or GaN switch implementation is not attractive primarily due to increased cost. Today’s most cost effective high-frequency implementation that employs high-voltage Si MOSFETs is only possible if the reverse-recovery-related losses of the slow parasitic antiparallel body diode are significantly reduced.

In this paper, active-snubber methods that offer reduced switching losses of semiconductor switches in the bidirectional buck-boost converter are described. The major feature of these active-snubber methods that are extensions of the active-snubber method described in [17] is soft switching of all semiconductor switches. Specifically, the rectifiers are turned off with a controlled turn-off rate of their current to minimize their reverse-recovery losses, the power-controlling switch is turned on with zero-voltage switching (ZVS), and the auxiliary switch in the active snubber is turned off with zero-current switching (ZCS). Because of fully soft-switched operation, proposed circuits exhibit improved efficiency and EMI performance compared to their conventional “hard”-switched counterparts and also enable the employment of semiconductor switches with a relatively slow antiparallel diode (rectifier) at high frequencies.

II. SOFT-SWITCHED BIDIRECTIONAL BUCK-BOOST CONVERTERS

Figure 1 shows the proposed soft-switched bidirectional buck-boost converter. As shown in Fig. 1, the converter includes two soft-switching cells, each consists of active switch SA, inductor LS, isolation transformer TR, and reset-voltage circuit DA-CR-RR-DB. In Fig. 1, switches SA1 and SA2 are unidirectional current switches that can carry current only in one direction. If bidirectional current switches such as MOSFETs are used, than a rectifier in series with the switch needs to be employed to prevent conduction of its antiparallel diode. Sources V1 and V2 can be any kind of DC power sources or their combinations that can deliver and store (receive) electric energy.

978-1-5090-5366-7/17/$31.00 ©2017 IEEE 287

In the bidirectional buck-boost converter in Fig. 1, switch S2 and diode D1 are boost switch and rectifier, respectively, whereas switch S1 and diode D2 are buck switch and rectifier, respectively. In this mode, boost switch S2 is controlled to provide regulation of the boost output, i.e., regulate voltage V2 and/or current I2 and/or power V2I2. Similarly, in the buck mode, buck switch S1 is controlled to provide regulation of the buck output, i.e., regulate voltage V1 and/or current I1 and/or power V1I1. In both modes, the switch in parallel with corresponding rectifier, i.e., switch S1 in the boost mode and switch S2 in the buck mode, can be either kept continuously open or preferably controlled as synchronous rectifiers to improve efficiency.

When the circuit in Fig. 1 operates in the boost mode, i.e., when power is transferred from V1 to V2, switch SA1 is turned off so that only soft-switching cell #2 is active, as illustrated in Fig. 2(a) by removing (not shown) the components of soft-switching cell #1. To facilitate the explanation of operation, Fig. 2(b) shows key waveforms of the circuit in Fig. 2(a) during a switching cycle assuming, for simplicity, that buck switch S1 is kept continuously off during the boost-mode operation.

As can be seen from Fig. 2 (b), before switch SA2 is turned on at t=T0, all switches are off. As a result, during this period the entire inductor current iL flows through antiparallel diode D1 of switch S1, because current iLS drawn by soft-switching cell #2 is zero. To reduce the reverse-recovery current of rectifier D1, switch SA2 is turned on slightly prior to turning on boost switch S2. After switch SA2 is turned on, current iLS starts flowing because voltage V2 is impressed across the series connection of inductor LS2 and primary winding NP2 of transformer TR2. Current iLS flowing through primary winding NP2 induces a current in secondary winding NS2 that flows through diode DB2 to source V2. As long as the current flows through the secondary winding, the voltage across secondary winding NS2 is clamped to V2,

inducing constant primary voltage = ( ) ∙⁄ =∙ , where n2=NP2/NS2 is the turns ratio of transformer TR2. As a result, after switch SA2 is turned on at t=T0, constant voltage = − = − ∙ is applied across inductor LS2 causing its current iLS2 =iLS to increase linearly. Because the current of a large buck-boost inductor can be considered approximately constant during a switching cycle, the current in rectifier D1 must decrease at the same rate, i.e., ⁄ = − ( − )⁄ . By controlling the

Fig. 1. Proposed soft-switched bidirectional buck-boost converter.

V1

S A2

L

C R2V R2

+

-

RR2NP2 N S2

DA2

L S2

S 2

S 1

S A1

C R1V R1

+

-

RR1

NP1 N S1

DA1

L S1

V2

TR1

TR2

DB2

DB1

SOFT-SWITCHING CELL #1

SOFT-SWITCHING CELL #2

D 2

D 1

I 1

I 2

A1

B1

A2

B2

(a)

(b)

Fig. 2. (a) Circuit diagram of active part of soft-switched buck-boost

converter in Fig. 1 when it operates in boost mode, i.e., transferpower from source V1 to source V2, where V2>V1 and (b) keywaveforms.

V1

L

C R2 RR2NP2 N S2

DA2

L S2

V2

TR2

DB2

SOFT-SWITCHING CELL #1

i L i LS

VP2

VR2

S 1D 1

i D1

VS1

S 2 D 2

VS2

iS2

VLS2

A1

B1

A2

B2

S A2 SOFT-SWITCHING CELL #2

t

t

t

t

ON OFF

S1

OFFON

i D1

dt=

V2

L S2

- VP2

i LS

dt=

LS2

VP2

i D1

t

i LS

OFF

t

i S2

t

VS1

t

ZVS

I 1

S2

SA2

VS2

TON

TS

T0 T1 T2 T3 T5 T6T4

288

rectifier current turn-off rate, the reverse-recovery losses can be minimized. The current turn-off rate in the circuit in Fig. 1(a) can be adjusted by a proper selection of turns ratio n2 of transformer TR2 and value of inductor LS2.

After rectifier current iD1 reaches zero at t=T1, i.e., after rectifier D1 is turned off, inductor current iLS starts discharging output capacitance of switch S2 and charging the output capacitance of switch S1, not shown in Fig. 1, by resonance between the output capacitance of the switches

and inductor LS2. As illustrated in Fig. 2(b), during this resonance, voltage vS2 across switch S2 decreases while voltage vS1 increases since the sum of the switch voltages is constant, i.e., vS1+vS2=V2. To achieve ZVS, switch S2 should be turned on when or shortly after its output capacitance is fully discharged. As shown in Fig. 2(b), switch S2 is turned on when switch voltage vS2 reaches zero at t=T2. When switch S2 is turned on, a negative primary-winding voltage is impressed across inductor LS2, i.e., vLS2= -VP2, causing a linear decrease of inductor current iLS with slope ⁄ =− ⁄ . After inductor current is reset to zero at t=T3, switch SA2 can be turned-off with ZCS, as shown in Fig. 2(b). When switch SA2 is turned off at t=T4, the magnetizing current of the transformer, not shown in the figures, which was flowing through closed switch SA2 is diverted through diode DA2 to the reset-voltage circuit comprising capacitor CR2 in parallel with resistor RR2. For proper operation, the value of reset voltage VR2 needs to be set by selecting the value of resistor RR2 so that the magnetizing current reaches zero before a new cycle is initiated at t=T6 by turning on switch SA2. Since voltage stress on switch SA2 is given by the sum of voltage V2 and reset voltage VR2, i.e., by V2+VR2, it is desirable to use the minimum reset voltage so that the stress of switch SA2 is also minimized.

Figures 3(a) and (b) show the active parts of the buck-boost circuit in Fig. 1 and its key waveforms, when the circuit operates in the buck mode, i.e., when power is transferred from V2 to V1. In the buck mode, switch SA2 is turned off so only soft-switching cell #1 is active, as illustrated in Fig. 3(a) by omitting the components of soft-switching cell #2. Since the operation in the buck mode is identical to that in the boost mode, the waveforms of the corresponding components in both circuits are identical, as can be seen from comparing waveforms in Figs. 2(b) and 3(b).

(a)

(b)

Fig. 3. (a) Circuit diagram of active part of soft-switched buck-boost

converter in Fig. 1 when it operates in buck mode, i.e., transferpower from source V2 to source V1, where V2>V1 and (b) keywaveforms.

V1

S A1

C R1 RR1

NP1 N S1

DA1

L S1

V2

TR1

DB1

SOFT-SWITCHING CELL #1

SOFT-SWITCHING CELL #2

VR1

L

i L i LS

VP1

S 1 D 1

i D2

VS1

S 2D 2

VS2

iS1

VLS1

A1

B1

A2

B2

t

t

S2

i D2

dt=

V2

L S1

- VP1

i LS

dt=

LS1

VP1

i D2

i LS

OFF

t

i S1

t

VS1

t

ZVS

I 1

t

t

ON OFF

OFFON

S1

SA1

VS2

TON

TS

T0 T1 T2 T3 T5 T6

t

T4

Fig. 4. Proposed soft-switched buck-boost converter with common snubber

inductor LS.

V1

S A2

L

C R2V R2

+

-

RR2NP2 N S2

DA2

S 2

S 1

S A1

C R1V R1

+

-

RR1

NP1 N S1

DA1

V2

TR1

TR2

DB2

DB1

L S

SOFT-SWITCHING CELL

D 2

D 1

I 1

I 2

289

Many variations of the circuit are possible depending how soft-switching cells are connected and how reset-voltage circuit is implemented. Because inductors LS1 and LS2 do not operate at the same time, inductors LS1 and LS2 in Fig. 1 can be implemented as a coupled inductor, i.e., by having both inductor windings on a single core, or by implementing the circuit with a common inductor LS for both soft-switching cells as shown in Fig. 4

Further reduction of components can be achieved by using a single transformer and reset-voltage circuit, i.e., by employing the same switching cell in both the boost and the buck mode as shown in Fig. 5 with two additional switches SB1 and SB2. These switches are used to provide proper polarity of the reset-voltage for the boost and buck mode. In the boost mode, switch SB1 is turned on and switch SB2 is turned off so the reset voltage VR is positive. In the buck mode, switch SB2 is turned on and switch SB1 is turned off so that the reset voltage is negative.

III. EXPERIMENTAL RESULTS

The performance of the proposed converter with the proposed soft-switching cell shown in Fig. 5 was evaluated on a 5-kW prototype circuit that was designed to exchange energy between a 400-V bus and a battery with voltage range between 200 V and 300 V. Figure 6 shows the experimental prototype circuit with the full description of the employed power components. It should be noted that the turns ratio (1:4) of transformer TR is chosen to provide full ZVS over the entire voltage and load ranges.

As shown in Fig. 6, the prototype is built using SPW47N60CFD MOSFETs (VDS = 600 V, RDS = 0.083 Ω, COSS=2.2 nF, Qrr=2 μC) from Infineon for switches S1 and S2 and STGW35HF60WD IGBTs (VDS = 600 V, VCE(SAT) = 1.65 V, COSS=235 pF, Qrr=90 nC) from ST for switches SA1 and SA2. It should be noted that two additional diodes DA1

and DA2, implemented with FR305CT (600 V, 3 A), are added in series with switches SA1 and SA2 to eliminate unnecessary reverse current caused by parasitic ringing between snubber indictor LS and output capacitances of switches SA1 and SA2, respectively. To obtain the desired inductance of filter inductor LB of approximately 330 μH, inductor LB was built by winding 37 turns of magnet wire (AWG #14) on two common toroidal cool-mu cores (77258A7) from Magnetics. Transformer TR was built using a pair of ferrite cores (PQ3230, 3C96) with 10 turns of Litz wire (Φ 0.1mm, 120 strands) for the primary winding and 40

Fig. 5. Proposed soft-switched buck-boost converter with single softswitching cell.

V1

S A2

L B

S A1

V2TR

CR V R

+

-

RR

SB2 SB1

DB2

DB1

L S

SOFT-SWITCHING CELL

S 2

S 1

I 1

I 2

NP1

N S1

D 2

D 1

DA2 DA1

A

B

CFig. 6. Experimental prototype circuit.

TR PQ3230-3C96N1=10 turns

Litz 0.1mmx120N2=40 turns

Litz 0.1mmx50L =1.9mHL =1.6uH

M_N2

LK_N1

PQ2016-3C96Litz 0.1mm x 170

6T, 1.5 uH

C2

S A2

LB

S 2

S 1S A1

CR RR

NP1 N S1

DA1

D B2

DB1

TR

SB1 SB2

DA2

L S

4x560uF/450 V

2x560uF/450 V

V 400 V

2

V 200 - 400 V

1

C1

BYM26C

77258A7two coresAWG#14

37T, 331 uH

47nF/400V

30 kΩ/2W

GW35HF60WD

47N60CFD

47N60CFD

GW35HF60WD

BYM26C

FR305CT (600 V, 3A)

FR305CT (600 V, 3A)GW35HF60WD

(a)

(b)

Fig. 7. Measured gate-voltage waveforms of switches when it operates in

boost mode. Time scales are (a) 2 μS/div. and (b) 400 nS/div.

290

turns of Litz wire (Φ 0.1mm, 50 strands) for the secondary winding. The measured magnetizing and leakage inductances at the primary winding are 119 μH and 1.6 μH, respectively. To achieve the desired turn-off rate of the rectifier current to approximately 150A/μsec, the inductance value of snubber inductor LS=VO/(iLS/dt) of 3 μH was selected. Since the leakage inductance of transformer TR is in series with snubber inductor LS, i.e., it is a part of inductor LS, the inductance of an external inductor should be approximately 1.5 μH. The inductor was built using a pair of ferrite cores (PQ-2016, 3C96) with 6 turns of Litz wire (Φ 0.1mm, 170 strands) and approximately 3.08 mm gap. Litz wire was used to reduce the fringing-effect-induced winding loss near the gap of the inductor core.

The voltage regulation for both directions was implemented by a TMS320F28027 microcontroller with 32-bit CPU from TI. A simple two-pole two-zero compensator is implemented to the output voltage control loop. The bandwidth of the voltage control loop is set to approximately 5 kHz which is about twenty times lower than that of the switching frequency.

Figures 7 and 8 show measured gate-voltage waveforms of switches when the prototype operates in boost and buck mode, respectively. As shown in Fig. 7 (a), switch SA1 is disabled for boost mode operation and S1 operates as a synchronous rectifier. Figure 7 (b) shows expanded waveforms of the gate signals. The dead time between switches S2 and S1 is set to be 900 ns to prevent any accidental overlapping of the gate signals. The delay time

between the turn-on transient of switch SA2 and that of switch S2 is set to be 500 ns. As shown in Fig. 8 (a), switch SA2 is

(a)

(b)

Fig. 8. Measured gate-voltage waveforms of switches when it operates in

buck mode. Time scales are (a) 2 μS/div. and (b) 400 nS/div.

(a)

(b)

Fig. 9. Measured waveforms of proposed converter when it operates in

boost mode. Time scales are (a) 500 nS/div. and (b) 2 μS/div.

(a)

(b)

Fig. 10. Measured waveforms of proposed converter when it operates in

buck mode. Time scales are (a) 500 nS/div. and (b) 2 μS/div.

291

disabled for buck mode operation and S2 operates as a synchronous rectifier. Figure 8 (b) shows expanded waveforms of the gate signals. The dead time and delay time are 900 ns and 500 ns, respectively.

Figure 9 shows the measured waveform of snubber inductor current iLS and the gate waveforms of switches SA2 and S2 of the experimental circuit when it operates in boost mode and delivers full power from 200-V input to 400-V output. As it can be seen from the waveforms, switch S2 turns on with ZVS and switch SA2 turns off with ZCS. Figure 10 shows the measured waveform of snubber inductor current iLS and the gate waveforms of switches SA1 and S1 of the experimental circuit when it operates in buck mode and delivers full power from 400-V input to 200-V output. The waveforms show ZVS of switch S1 and ZCS of switch SA1.

Figures 11 and 12 show measured efficiencies and losses of the prototype converter with and without the proposed soft-switching cell when it operates in boost and buck modes, respectively. It should be noted that the converter with soft switching exhibits much higher efficiency than the converter with hard switching. In fact, because of high losses, the output power of the implementation without the proposed soft-switching active-snubber cell is thermally limited to

around 2 kW. The prototype circuit exhibits the maximum full-load efficiency of 99.1% at boost mode operation and 98.2% at buck mode operation with the switching frequency of 100 kHz.

IV. SUMMARY

In this paper, a bidirectional buck-boost converter with new soft-switching active-snubber cells that reduce switching losses has been introduced. The soft-switching cells enable the buck and boost rectifier that are the body-diode of the corresponding switches to turn off with a controlled turn-off rate of their current to minimize corresponding reverse-recovery losses. In addition, in the introduced soft-switching cell, the power-controlling buck and boost switches turn on with ZVS and the snubber switches turn off with ZCS. The performance of the proposed bidirectional converter was evaluated on a 5-kW prototype exchanging energy between a 400 V bus and a battery with voltage range between 200-V and 300 V. The 100-kHz prototype circuit exhibits the maximum full-load efficiency of 99.1% in the boost-mode operation and 98.2% in the buck-mode operation.

(a)

(b)

Fig. 11. Measured (a) efficiencies and (b) losses of experimental prototype

with and without proposed soft switching circuit as functions ofoutput power when it operates in boost mode.

400 600 800 1000 1500 2000 2500 3000 3500 4000 4500 5000

Output Power [W]

95

95.5

96

96.5

97

97.5

98

98.5

99

99.5

100

Eff

icie

nc

y [

%]

Hard Switching

V = 400 VBULK

Soft Switching

200 V

V = 300 VBAT

300 V

200 V

68 deg

T = 36 degSW

66 deg

42 deg

400 600 800 1000 1500 2000 2500 3000 3500 4000 4500 5000

Output Power [W]

0

10

20

30

40

50

60

70

80

90

100

110

Lo

ss

[W

]

Hard Switching

Soft SwitchingV = 400 VBULK

V = 200 VBAT

(a)

(b)

Fig. 12. Measured (a) efficiencies and (b) losses of experimental prototype

with and without proposed soft switching circuit as functions ofoutput power when it operates in buck mode.

400 600 800 1000 1500 2000 2500 3000 3500 4000 4500 5000

Output Power [W]

92

92.5

93

93.5

94

94.5

95

95.5

96

96.5

97

97.5

98

98.5

99

Eff

icie

nc

y [

%]

V = 400 VBULK

Soft Switching

200 V

V = 300 VBAT

300 V

200 V

85 deg

T = 48 degSW

81 deg

55 deg

Hard Switching

400 600 800 1000 1500 2000 2500 3000 3500 4000 4500 5000

Output Power [W]

0

10

20

30

40

50

60

70

80

90

100

110

120

130

140

150

160

170

180

Lo

ss

[W

]

Hard Switching

Soft Switching

V = 400 VBULK

V = 200 VBAT

292

REFERENCES [1] D.M. Sable, F.C. Lee, and B.H. Cho, “A zero-voltage-switching

bidirectional battery charger/discharger for the NASA EOS satellite,” in Proc. IEEE APEC, 1992, pp. 614 – 621.

[2] K.T. Chau, T.W. Ching, and C.C. Chan, “Bidirectional soft-switching converter-fed dc motor drives,” in Proc. IEEE PESC, 1998, pp. 416 – 422.

[3] F. Caricchi, F. Crescimbini, F.G. Capponi, and L. Solero, “Study of bi-directional buck-boost converter topologies for application in electrical vehicle motor drives”, in Proc. IEEE APEC, 1998, pp. 287 – 293.

[4] Y. Zhang and P. C. Sen, “A New Soft-Switching Technique for Buck, Boost, and Buck-Boost Converters,” IEEE Trans. Ind. Appl., vol. 39, pp.1775–1781, Nov./Dec. 2003.

[5] C.G. Yoo, W.-C. Lee, K.-C. Lee, and B.H. Cho, “Transient current suppression scheme for bidirectional dc-dc converters in 42V automotive power systems,” in Proc. IEEE APEC, 2005, pp. 1600-1604.

[6] L. Solero, A. Lidozzi, and J.A. Pomilio, “Design of Multiple-Input Power Converter for Hybrid Vehicles”, IEEE Trans. Power Electron., vol. 20 no. 5, pp. 1007-1016, Sep. 2005

[7] O. Garcia, P. Zumel, A. de Castro, and A. Cobos, “Automotive dc-dc bidirectional converter made with many interleaved buck stages,” IEEE Trans. Power Electron., vol. 21, no. 3, pp. 578 – 586, May 2006.

[8] D.P. Urciuoli and C.W. Tipton, “Development of a 90 kW bi-directional dc–dc converter for power dense applications,” in Proc. IEEE APEC, 2006, pp. 1375–1378.

[9] J. Zhang, J.-S. Lai, R.-Y. Kim and W. Yu, “High-power density design of a soft switching high-power bidirectional dc–dc converter,” IEEE Trans. Power Electron., vol. 22, no. 4, pp. 1145 – 1153, Jul. 2007.

[10] P. Das, S.A. Mousavi, and G. Moschopoulos, “Analysis and design of a nonisolated bidirectional ZVS-PWM DC–DC converter with coupled inductors,” IEEE Trans. Power Electron., vol. 25, no. 10, pp. 2630–2641, Oct. 2010.

[11] L. Jiang, C.C. Mi, S. Li, M. Zhang, X. Zhang, and C. Yin, “A Novel

Soft-switching bidirectional dc-dc converter with Coupled Inductors,” IEEE Trans. Ind. Appl., vol. 49, no. 6, pp. 2730-2740, Nov. 2013.

[12] J.H. Lee, D.H. Yu, J.G. Kim, Y.H. Kim, S.C. Shim, D.Y. Jung, Y.C. Jung, and C.Y.Won, “Auxiliary switch control of a bidirectional soft-switching dc/dc converter,” IEEE Trans. Power Electron., vol. 28, no. 12, pp. 5446-5457, Dec. 2013.

[13] M. Pavlovsky, G. Guidi, and A. Kawamura, “Buck/Boost dc–dc converter topology with soft switching in the whole operating region,” IEEE Trans. Power Electron., vol. 29, no. 2, pp. 851-862, Feb. 2014.

[14] M. Mohammadi and H. Farzanehfard, “Analysis of Diode Reverse Recovery Effect on the Improvement of Soft-Switching Range in Zero-Voltage-Transition Bidirectional Converters”, IEEE Trans. Ind. Electron., vol. 62, no. 3, pp. 1471-1479, Mar. 2015.

[15] M.M. Jovanović, “A technique for reducing rectifier reverse-recovery-related losses in high-power boost converters,” IEEE Trans. Power Electron., vol. 13, no. 5, pp. 932-941, Sep. 1998.

[16] Z. Wang, J. Ouyang, J Zhang, X. Wu, and K. Sheng, “Analysis on reverse recovery characteristic of SiC MOSFET intrinsic diode,” in Proc. IEEE Energy Conversion Congress and Exposition (ECCE), 2014, pp. 2832–2837.

[17] Y. Jang, M.M. Jovanović, K.H. Fang, and Y.M. Chang, “Soft-switched high-power-factor boost converter,” IEEE Trans. Power Electron., vol. 21, no. 1, pp. 98-104, Jan. 2006.

293