resonant dc dc buck - boost converter for the battery...
TRANSCRIPT
117 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
Resonant DC – DC Buck - Boost Converter for the Battery Charger
and PV Applications
1D. Govind, 2 Nandkumar Wagh 1Assistant Professor, 2 Professor
Department of Electrical Engineering,
Vidya Pratishthan’s Kamalnayan Bajaj Institute of Engineering and Technology,
Baramati, Pune(M.S.)-India
Abstract—In power electronic switches, soft switching is
a possible way of reducing losses. Soft switching refers
to the operation of power electronic switches as zero-
voltage switches (ZVS). All the power electronic
switching devices undergoes zero-voltage switching
during turn-off. In the converter, the switches undergo
zero-capacitive turn-on losses unlike switches in other
soft-switched topologies. This soft-switching technique
can also be applied to other classical switched mode
power converters. The structure of the proposed
converter is simpler and cheaper than other resonant
power converters. In this paper, single-switch resonant
power converter offering the advantages of soft
switching, reduced switching losses, and increased
energy conversion efficiency for Photovoltaic
applications is presented. This circuit topology integrates
a single-switch resonant converter with zero-voltage-
switching. The operating principles and the steady-state
analyses of the proposed interleaved buck, boost buck-
boost converters are discussed and performance of grid
connected ZVS is verified with simulation results.
Keywords—ZVS, buck, boost, buck –boost.
This paper is divided in five sections. Section-I
presents the introduction of switching modes in
converters and the literature related to it.
Section-II presents all converter topologies.
Simulation results of the converter topologies are
depicted in Section-III.
Section-IV and V deals with the comparison of
converter topologies and the comparative analysis.
I. INTRODUCTION
DC-DC converters are finding more and more
use in portable applications such as cell phones,
laptops etc. In order to achieve higher power
density and high voltage profile, these converters
are usually require to operate at higher switching
frequencies with higher efficiencies[1]-[2].
When the switching frequencies continues to
increase , then in order to meet the future
requirements of power density and efficiency, the
Resonant DC-DC converters redraw people’s
attention. Resonant converters are good alternative
because of its soft-switching power transfer
characteristic. These converters can considerably
reduce the switching loss and obtain friendly EMI
characteristics [3]. Therefore we can operate the
converter at higher frequencies without sacrificing
the efficiency, so high efficiency and high power
density can be achieved simultaneously. In
Resonant converters because of smooth voltage and
current waveforms, noise and interference and
stress on switching devices are reduced and
parasitic circuit elements such as transformer
leakage inductance can be taken into account.
In the 1970’s, conventional PWM power
converters were operated in a switched mode
operation. Power switches have to cut off the load
current within the turn-on and turn-off times under
the hard switching conditions. Hard switching
refers to the stressful switching behavior of the
power electronic devices. The switching trajectory
118 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
of a hard-switched power device is shown in Fig.1.
During the turn-on and turn-off processes, the
power device has to withstand high voltage and
current simultaneously, resulting in high switching
losses and stress. Dissipative passive snubbers are
usually added to the power circuits so that the dv/dt
and di/dt of the power devices could be reduced,
and the switching loss and stress are diverted to the
passive snubber circuits [4].
However, the switching loss is proportional to
the switching frequency, thus it is required to limit
the maximum switching frequency of the power
converters. Typical converter switching frequency
was limited to a few tens of kilo-Hertz (typically 20
kHz to 50 kHz) in early 1980’s.
I
VOff
On
Soft-switching
Hard-switching
Safe Operating Area
snubbered
Fi
g.1 Typical switching trajectories of power
switches
The stray inductive and capacitance in the power
circuits and power devices still cause considerable
transient effects, which in turn give rise to
electromagnetic interference (EMI) problems [3].
Fig.2. shows ideal switching waveforms and typical
practical waveforms of the switch voltage. The
transient ringing effects are major causes of EMI.
In the 1980’s, lots of research efforts were diverted
towards the use of resonant converters. The concept
was to incorporate resonant tanks in the converters
to create oscillatory, usually sinusoidal voltage and
current waveforms so that zero voltage switching
(ZVS) or zero current switching (ZCS) conditions
can be created for the power switches. The
reduction of switching loss and the continual
improvement of power switches allow the
switching frequency of the resonant converters to
reach hundreds of kilo-Hertz (typically 100 kHz to
500 kHz). Consequently, sizes of elements can be
reduced and the power density of the converters is
increased. Various forms of resonant converters
have been proposed and developed.
Fig.2. Typical switching waveforms of (a) hard-
switched and (b) soft-switched devices
Resonance Technology
There are basically two types of soft-switching
techniques:
1. Zero Current Switching (ZCS)
2. Zero Voltage Switching (ZVS)
Either of this technique can greatly reduce and even
completely eliminates the switching losses in a
converter. High power level converters usually use
IGBT switches due to low conduction losses and
higher power capability, but IGBT is not as fast as
MOSFET [5]- [8] and its switching frequency
cannot be increased beyond 100 KHz even if softly
switched. On the contrary to Insulated Gate Bipolar
Junction Transistor (IGBT), Metal Oxide
Semiconductor Field Effect Transistor (MOSFET)
is resistive device. When it is turned on, the
conduction losses are higher as compared to IGBT
at higher power levels. However, MOSFET is a
faster device and is able to operate up to a few
MHz’s.
Fig.3. Shows current and voltage waveforms of
hard and resonant switching system with portion of
losses in both.
Fig.3. Current and voltage waveforms of hard and
resonant switching systems
119 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
ZVS converters have three resonant states: over
resonance (completed resonance), optimum
resonance (critical resonance) and quasi resonance
(sub resonance). Only the quasi resonance state has
two zero crossing points in a repeating period. A
resonant switch is a sub-circuit comprising a
semiconductor switch S and resonant elements, Lr
and Cr. The switch S can be implemented by a
unidirectional or bidirectional switch, which
determines the operation mode of the resonant
switch. [4].
II. ZERO VOLTAGE RESONANT SWITCH
(ZVS)
In zero voltage switching resonant converters,
the resonant capacitor provides a zero-voltage
condition for the switch to turn on and off [7]. A
quasi-resonant buck converter designed for half-
wave operation using a ZV resonant switch as
shown in Fig.4. In a ZV resonant switch, a
capacitor Cr is connected in parallel with the switch
S for achieving zero-voltage-switching (ZVS). If the
switch S is a unidirectional switch, the voltage
across the capacitor Cr can oscillate freely in both
positive and negative half-cycle. Thus, the resonant
switch can operate in full-wave mode. If a diode is
connected in anti-parallel with the unidirectional
switch, the resonant capacitor voltage is clamped by
the diode to zero during the negative half-cycle.
The resonant switch will then operate in half-wave
mode. The objective of a ZV switch is to use the
resonant circuit to shape the switch voltage
waveform during the off time in order to create a
zero-voltage condition for the switch to turn on [6].
Lr
S
(a)
Cr
Lr
CrS
(b)
Fig.4. Zero-voltage resonant switch (a) half wave
(b) full wave mode
Steady-State Analysis of Quasi-Resonant
Converters (QRC) to simplify the steady-state
analysis of the converters, some assumptions needs
to be made.
1. The filtering components Lo, Lin, Lf, and Co
are very large compared to the resonant
components Lr, Cr.
2. The output filter Lo-Co-R is treated as a
constant current source, Io.
3. The output filter Co-R is treated as constant
voltage source, Vo.
4. Switching devices and diodes are ideal.
5. Reactive circuit components are ideal.
A ZVS-QRC designed for half-wave operation is
illustrated with a buck type dc-dc converter; shown
in Fig.5.
Fig.5. Circuit diagram of a buck zero-voltage-
switching resonant converter
Fig.5, depicts the circuit structure of a buck
zero-voltage switching resonant converter. It differs
from a conventional buck PWM converter in that it
has an additional resonant tank that comprises a
resonant inductor Lr , a resonant capacitor Cr , and
a diode Dr . The inductor Lr is connected in series
to power switch Q to limit di/dt of the power
switch, and the capacitor Cr is installed as an
auxiliary energy transfer element. Lr and Cr
constitute a series resonant circuit, whose
oscillation is initiated by turning off the power. Dm
is a freewheeling diode. Capacitor Cf and inductor
Lf comprise a low pass filter, which not only filters
high-frequency ripple Signal, but also provides a
stable dc source for load. The freewheeling diode in
the ZVS converter is commutated under soft-
switching. This characteristic makes the ZVS
technique particularly appealing for high-frequency
conversion applications. Therefore, implementing
soft-switching for both the power switch and the
freewheeling diode in such a circuit is particularly
valuable. To simplify the analysis, the output filter
120 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
inductance is assumed to be sufficiently large to be
regarded as an ideal dc current source Io, during a
high-frequency resonant cycle. In one switching
cycle, the circuit operation can be divided into four
modes, whose associate equivalent circuits are
displayed in Fig.5. The parameters are defined as
follows. Characteristic impedance: ro
r
LZ
C ;
resonant angular frequency: o r rL C ; resonant
frequency: fr = ωo/2π; switching period: Ts
In a single switching cycle, the circuit operates
in the following four modes.
Mode I: Linear stage0 1( )t t t : Prior to t0, the
power switch Q is on, and conducts a drain current
that equals the output current Io, and the
freewheeling diode Dm is off. Fig. 6 depicts the
equivalent circuit. At t0, Q is turned off. The current
through the resonant inductor Lr does not change
instantaneously, and so the current is diverted
around the power switch through the resonant
capacitor Cr. The current of the resonant inductor
equals the output current Io and the capacitor
voltage vcr, which increases, as given by
0
1( )
t
ocr o
r r
IV t I dt t
C C (1)
Voltage across freewheeling diode Dm is
determined by
( ) ( ) ox in cr in
r
Iv t V v t V t
C (2)
Fig.6. Equivalent circuit of Model I
vx declines to zero at time t1 , when Dm is turned on
by soft switching. The constant output current
linearly increases the voltage across the resonant
capacitor, until the input voltage is reached
1
in r
o
V Ct
I (3)
Model I is completed when t = t1 , namely vcr (t1) =
Vin . The time interval TI in Model I is obtained
using (5). Moreover, Model II is initiated when vx
decreases to zero
in rI
o
V CT
I (4)
Mode II: Resonant stage1 2( )t t t ): After t1, the
freewheeling diode Dm becomes forward-biased,
and Cr and Lr resonate. The instantaneous voltage
across Cr and the resonant inductor current can be
evaluated, respectively as
1( ) cos ( )
rL o oi t I t t (5)
1
1
1( ) ( ) ( )
t
cr cr cr
r t
v t i t dt v tC
(6)
With initial condition vCr(t1) = Vin (7)
1
1
1( ) cos ( )
t
cr o o in
r t
v t I t t dt VC
(8)
Fig.7. Equivalent circuit of Model II
1( ) sin ( )cr o o o inv t Z I t t V (9)
The maximum ( )crv t that occurs at '
1t is
determined by
' 1
1 1
1(sin )
r
t t
(10)
The maximum value of crv is determined by
,Cr Peak in o ov V Z I (11)
The voltage across the freewheeling diode in Fig. 7
can be written as
( ) 0crv t
121 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
The freewheeling diode current wave shape follows
a cosine function during this interval, and equals Io
minus iLr (t). The resonant time is determined by
solving the resonant capacitor voltage equation
under the condition when the voltage is zero.
1
2 1
1[sin ( ) ]in
o o o
Vt t
Z I
(12)
This model is completed at t = t2 , when vCr(t2) = 0
and iLr (t2) = −Io . Moreover, the time interval
during Model II is determined using (13)
11[sin ( ) ]in
II
o o o
VT
Z I
` (13)
The above equation indicates that load current Io
is so large that Zo.Io > Vin. Otherwise, the voltage
of the power switch would not return to zero
naturally, and the power switch has to be turned on
at a nonzero voltage, causing turn-on losses. This
interval ends at t2, when vcr decreases to zero and
the anti-parallel diode Dr begins to conduct.
Mode III: Recovery Stage2 3( )t t t : After Dr is
turned on, the voltage across Cr is held at zero. The
turn-on signal of Q is applied, when the anti
parallel diode is conducting to achieve ZVS. During
this interval, the inductor current is expressed as
2
2
1( ) ( )
t
L in Lr
r t
i t V dt i tL
(14)
2 2 1( ) cos[ ( )]in
r
Vt t Io o t t
L
(15)
3 3 2 2 1( ) ( ) cos[ ( )]inLr o o
r
Vi t Io t t I t t
L
(16)
3 0 2 1 2( )[1 cos ( )]r o
in
L It t t t
V (17)
Fig.8. Equivalent circuit of Model III
The resonant inductor current iLr (t) is linearly
returned from its negative peak of minus Io to its
positive value of positive Io . Consequently, iLr (t)
increases linearly and iDm decreases linearly. This
model is completed at t = t3 when vCr(t3) = 0 and iLr
(t3) = Io. The commutation interval in this stage is
expressed by
0 2 1 2( )[1 cos ( )]r oIII
in
L IT t t t
V (18)
Notably, the voltage across the switch Q is zero,
when the power switch is turned on. It enables the
turn-on switching loss to be avoided and the total
efficiency of the converter to be increased
accordingly.
Mode IV: Freewheeling stage3 4( )t t t : When
iL (t) reaches Io at t3, the freewheeling diode Dm is
turned off, and the zero-voltage-switched converter
resembles a conventional square-wave power
processor. The charging current flows through
power switch Q and resonant inductor Lr.
Accordingly,
( )L oi t I and ( ) 0Crv t (19)
The power switch conducts Io as long as it is kept
on until t4. At t4, the power switch is turned off
again, beginning another switching cycle. The
duration of this mode is TIV expressed as
( )IV s I II IIIT T T T T (20)
Fig.9. Equivalent circuit of Model IV
The output voltage Vo is determined as
0
1Ts
BA x
s
V v dtT
1
10 3
1[ (1 )]
t Ts
in in
s t
tV dt V dt
T t
13[ ( )]
2
ins
s
V tT t
T (21)
13[1 ( )]
2BA in s
tV V f t (22)
The output voltage varies with the switching
frequency. Fig. 9 illustrates the equivalent circuit of
122 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
Model IV and Fig. 10 shows key steady-state
waveforms of the buck ZVS converter.
Voltage Gain: The condition ZoIo > Vin must hold
to ensure that the operation is under zero voltage
switching. 1o in
o r
I VC
,
or
in o
IC
V (23)
Similarly, because of the condition ZoIo >Vin must
hold such that o o r inI L V
Fig.10. Steady-state waveforms of the developed
ZVS resonant converter
inr
o o
VL
I (24)
Given Io and TS, TI, TII and TIII and the output
voltage Vo can be determined. However, the
voltage conversion ratio is normally best expressed
in terms of load resistance R and switching
frequency fs. Vo = RIo, so the energy stored in the
resonant inductor is 2 3
3 1
1 2
( )
t t
i in Lr in Lr in o s
t t
W V i dt V i dt V I T t t (25)
The energy released by the filter inductor to the
load is
o o o sW V I T (26)
2
1
t
inin
o O Ot
Io ViLrdt CrV
I Z
(27)
3 2 2
2 1
2
[1 cos ( )]
2
t
O O
int
LrI t tiLrdt
V
(28)
Let 2 1( )O t t
2 1 3 2( ) [ ( ) ( )]o s II III o sI T T T I T t t t t
(1 cos )[ ]o
o s
in o
I LrI T
V
(29)
Let the normalized load resistance
o
o i
Rr
Z W
(1 cos ) [ ]
2
r O r inin o
O in
L I C VV I Ts
V
(30)
o o o sW V I T
Wo equals Wi , when the converter power
dissipation is ignored.
1[ (1 cos ) ]
2
O O O r O r in
in o s in O
V I L C V
V I T V I
(31)
For a lossless system, in the steady state, these two
energies are equal. Hence, the voltage ratio is
expressed by (32) as
(1 cos )1 [ ]
2 2
fs X rX
fr r X
(32)
Where BA
in
VX
V and
2 1( )O t t denote the
voltage ratio and pulse width angle, respectively.
The relationship between input and output voltages
is a function of the pulse width angle, the
characteristic impedance of the resonant-switch
converter and the output load current. The variation
of modulation index (M) and switching frequency
fns as a function of duty ratio (D) is shown in fig.11.
Fig.11. Relationship between M and fns
123 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
ZVS converters can be operated in full-wave
mode. The circuit schematic is shown in Fig.12. (a).
The circuit waveforms in steady state are shown in
Fig.12. (b). The operation is similar to half-wave
mode of operation, except that VCr can swing
between positive and negative voltages. The
relationships between M and at different r are
shown in Fig.12. (c).
Cf
Cr
Lr
Lf
Df
+
Vo
-
+
voi
-
ILr
+ vc -
Dr
Io
Fig.12. (a) Schematic diagram
Fig.12. (b) Circuit waveforms
Fig.12. (c) Relationship between M and fns
Comparing Fig.11 with Fig.12 (c), it can be seen
that M is load-insensitive in full-wave mode [7]-
[10]. This is a desirable feature. However, as the
series diode limits the direction of the switch
current, energy will be stored in the output
capacitance of the switch and will dissipate in the
switch during turn-on. Hence, the full-wave mode
has the problem of capacitive turn-on loss, and is
less practical in high frequency operation. In
practice, ZVS-QRCs are usually operated in half-
wave mode rather than full-wave mode.
III. SIMULATION RESULTS
Simulation of all six types of zero voltage switching
(ZVS) converters is performed using PSIM
software.
ZVS Buck Converter:
ZVS Resonant BUCK Converter with half wave
topology
The Simulation Model of ZVS Buck Converter
is shown in Fig.14a. By adding Lr,Cr to the normal
buck converter it will become as a resonant
converter. Consider the converter specifications as:
Specifications: Vi = 24V, f = 25kHZ, D = 70%, Lr
= 316µH, Cr = 22nF, Ro = 50Ω, Vo = 14V.
The resonant capacitor is placed across the switch.
This capacitor makes the switch turn on at zero
voltage position. By proper maintenance of firing
pulses switch turn on and off takes place at zero
voltage positions.
Fig.14b. shows that, whenever voltage across
switch is zero at that position ,switching (either ON
or OFF) takes place. The switching losses and EMI
decrease because in normal converter switching
takes place at non- zero value of voltage because of
this dv/dt increases and consequently EMI
increases. Where as ZVS switching takes place at
zero voltage position, so dv/dt decreases and EMI
decreases.
Fig.14a. Simulation Model of ZVS Buck Converter
with half wave topology
124 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
0.0
0.20
0.40
0.60
0.80
1.00
Firing_pulses
0.0
-10.00
10.00
20.00
30.00
40.00
50.00
60.00
voltage_across_switch
19.90 19.92 19.94 19.96 19.98 20.00
Time (ms)
0.0
-0.10
-0.20
-0.30
-0.40
0.10
0.20
0.30
Inductor_current
Fig.14b. Firing Pulses, Current through Switch and
Voltage across Resonant Capacitor vs Time
0.0
2.50
5.00
7.50
10.00
12.50
Output_voltage
0.0 5.00 10.00 15.00 20.00 25.00 30.00
Time (ms)
0.0
0.10
0.20
0.30
output_current
Fig.15 Output Voltage and Output Current vs Time
The output voltage dependency with respect to the
switching frequency is shown in Fig.16. This gives
the output voltage for different normalised loads
and switching frequency. If the load changes for
particular switching frequency voltage gain varies.
This is the disadvantage of this topology. To
overcome this full wave mode topology is
introduced.As the switching frequency increases
output voltage decreases.
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.90.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
Normalised frequency(fs/fo)
Vo
ltag
e G
ain
(M
)
Q = 0.1
Q = 0.2
Q = 0.5
Q = 0.8
Fig.16. Relationship between M and fns
ZVS Resonant BUCK Converter with full wave
topology
Simulation Model of ZVS Buck Converter in
full wave mode is shown in Fig.17. By adding anti
parallel diode across switch current through switch
goes to negative also. This can be observed from
Fig.18.
Specifications:
Vi = 24V, f = 25KHZ, D = 60%, Lr = 316µH, Cr =
22nF, Ro = 100Ω, Vo = 14.8V.
Fig.17. Simulation Model of ZVS Buck Converter
with full wave topology
Fig.18. show that whenever voltage across
switch is zero, at position, switching (either ON or
OFF) takes place. Fig.19. shows the output voltage
and output current.
125 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
0.0
0.20
0.40
0.60
0.80
1.00
firing_pulses
0.0
20.00
40.00
60.00
Voltage_across_switch
49.90 49.92 49.94 49.96 49.98 50.00
Time (ms)
0.0
-0.20
-0.40
0.20
0.40
Inductor_current
Fig.18. Firing Pulses, Current through Switch and
Voltage across Resonant Capacitor vs Time
0.0
5.00
10.00
15.00
20.00
25.00
30.00
Output_voltage
0.0 0.02 0.04 0.06 0.08 0.10
Time (s)
0.0
0.20
0.40
output_current
Fig. 19. Output Voltage and Output Current vs
Time
0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 10
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1
Normalised frequency(fns)
Vo
lta
ge G
ain
(M
)
Q = 0.45
Q = 0.9
Q = 1.4
Q = 1.9
Fig.20. Relationship between M and fns
ZVS Boost Converter:
PSIM Simulation Model of ZVS Boost Converter in
half wave mode is shown in Fig.21.
Specifications:
Vi = 24V, fs= 25KHZ, D = 60%, Lr = 316µH, Cr =
22nF, Ro = 100Ω, Vo = 30V.
Fig.21. Simulation Model of ZVS Boost Converter
Fig.22. shows whenever voltage across switch is
zero, at position switching (either ON or OFF) take
place. Fig.23 shows the output voltage and output
current wave forms.
0.0
0.20
0.40
0.60
0.80
1.00
Firing_pulses
0.0
20.00
40.00
60.00
voltage_across_switch
49985.00 49990.00 49995.00 50000.00
Time (us)
0.0
-2.00
-4.00
2.00
4.00
Inductor_current
Fig.22. Firing Pulses, Current through Switch and
Voltage across Resonant Capacitor vs Time
0.0
5.00
10.00
15.00
20.00
25.00
30.00
35.00
Output_voltage
5.00 10.00 15.00 20.00
Time (ms)
0.0
0.25
0.50
0.75
1.00
1.25
1.50
1.75
Output_current
Fig.23. Output Voltage and Output Current vs
Time
126 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
The output voltage variation with switching
frequency is shown in Fig.24. This gives the output
voltage for different normalised loads(Q). As the
switching frequency increases output voltage
decreases. Voltage gain is greater than one because
its boost converter its output is always greater than
input so M >1 always.
0.3 0.4 0.5 0.6 0.7 0.8 0.9 10
0.5
1
1.5
2
2.5
3
3.5
4
Normalised frequency(fs/fo)
Vo
lta
ge G
ain
(M
)
Q = 0.25
Q = 0.5
Q = 1
Q = 2
Fig.24. Relationship between M and fns
ZVS Buck – Boost Converter
Buck Mode Operation: The Simulation Model of
ZVS Buck-Boost Converter in half wave mode
shown in Fig.25. Specifications: Vi = 24V, f =
25kHZ, D = 45%, Lr = 316µH, Cr = 22nF, Ro =
100Ω,
Vo=12.5V.
Fig.25. Simulation Model of ZVS Buck-Boost Converter
0.0
0.20
0.40
0.60
0.80
1.00
Firing_pulses
0.0
-20.00
20.00
40.00
60.00
80.00
voltage_across_switch
39990.00 39992.50 39995.00 39997.50 40000.00
Time (us)
0.0
-2.00
-4.00
-6.00
2.00
4.00
6.00
Inductor_current
Fig.26. Firing Pulses, Current through Switch and
Voltage across Resonant Capacitor vs Time
Fig.26. shows whenever voltage across switch is
zero, at position switching (either ON or OFF) take
place in buck mode operation. Fig.27. shows the
output voltage and output current wave forms.
Fig.28. show that whenever voltage across switch is
zero, at position switching (either ON or OFF)
takes place in buck mode operation. Fig.29. shows
the output voltage and output current.
0.0
-2.50
-5.00
-7.50
-10.00
-12.50
-15.00
Output_voltage
0.0 5.00 10.00 15.00 20.00
Time (ms)
0.0
-1.00
-2.00
-3.00
output_current
Fig.27. Output Voltage and Output Current vs
Time
Boost Mode of Operation:
In this mode of operation output is more than
input in order to get this D should be maintained
greater than 0.5. In this simulation D is selected as
70%.
0.0
0.20
0.40
0.60
0.80
1.00
Firing_pulses
0.0
25.00
50.00
75.00
100.00
125.00
voltage_across_switch
29985.00 29990.00 29995.00 30000.00
Time (us)
0.0
-5.00
-10.00
5.00
10.00
Inductor_current
Fig.28 Firing Pulses, Current through Switch and
Voltage across Resonant Capacitor vs Time.
127 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
0.0
-10.00
-20.00
-30.00
-40.00
Output_voltage
0.0 5.00 10.00 15.00 20.00
Time (ms)
0.0
-0.50
-1.00
-1.50
-2.00
-2.50
-3.00
-3.50
output_current
Fig.29. Output Voltage and Output Current vs Time
The output voltage dependency with switching
frequency for different loads are shown in Fig.30.
As the switching frequency increases output voltage
decreases in negative with negative slope . At fns
=0.45 boost operation starts for all Q > 0.05, if it is
less than 0.45 converter operates in buck converter.
0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-2.5
-2
-1.5
-1
-0.5
0
Normalised frequency(fs/fo)
Vo
ltag
e G
ain
(M
)
Q = 0.1
Q = 0.5
Q = 1
Q = 1.5
Fig.30. Relationship between M and fns
IV. COMPARISION OF ZVS CONVERTER
WITH NORMAL CONVERTER
ZVS resonant converters can considerably
reduce the switching loss by switching on the
device at zero voltage position, so the efficiency of
the converter increases compared to the normal
converter. The comparison of ZVS boost with
normal boost converter with different parameters is
shown in table-1.The power losses and efficiency of
ZVS and normal boost converters are calculated
and presented
Table – 1: Comparison of ZVS Boost Converter with Normal Boost Converter
Parameters Normal Boost Converter ZVS Boost Converter
R = 100Ω R = 150Ω R = 100Ω R = 150Ω
Input Volatge (Vin) 24 24 24 24
Input Current(Iin) 5.82 3.4 1.258 1.156
Input Power(Pin) 139.68 9.456 30.192 27.4
Output Voltage(Vo) 109 100 53.52 60
Output Current(Io) 1.09 0.65 0.535 0.4
Output Power(Po) 118.8 65 28.63 24
Efficiency 85 79.6 94.8 86.5
128 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
V. COMPARATIVE ANALYSIS OF ZVS RESONANT CONVERTERS:
The steady state performance of different converters, considering different parameters like rise time, peak
overshoot, efficiency and output voltage ripple is presented in table-2.
Table -2: Performance analysis of various ZVS converters.
Converter Type Vin Iin Vo Io Efficien
cy
Rise
Time
(msec)
Peak
Over
Shoot
Output
Voltage
Ripple(mv)
Buck 24 0.185 13.5 0.27 82 6 Zero 5
Buck Full wave
mode 24 0.202 14.1
0.28
1 81.7 2.5 25.2 2
Boost 24 1.08 34.8 0.7 92 2.3 42 10
Buck-boost 24 0.47 21.8 0.45 87 6 Zero 10
CONCLUSION
The converter topologies has been described and
the simulation results of all are presented. To
minimize the switching losses at the time of switch
on, zero voltage condition is provided by
introducing resonant circuit across switch.
Simulation of various ZVS Resonant DC-DC
converters are carried out using PSIM and output
voltage, current, voltage across resonant capacitor
are obtained. The variation of voltage gain with
switching frequency is plotted in MATLAB
environment. Simulation results and the calculated
switching loss and efficiency shows that, there is a
great improvement in efficiency of ZVS Resonant
converters compared to normal converter.
REFERENCES
[1] Kwang-Hwa Liu, Ramesh Oruganti and Fred
C.Y.Lee, “Quasi-Resonant converter topologies and
Characteristics”, IEEE Transactions on Power
Electronics, vol. PE-2, no.1, January 1987, pp .
[2] M. A. Jabbar, Ashwin M. Khambadkone and Guo
Chun, “Quasi-Resonant-Converters-Based High-
Efficiency Spindle Motor Drives for Magnetic Data
Storage” IEEE Transactions on Industrial
Electronics , vol .51, NO. 6, December 2004, pp
1338-1343.
[3] H. Chung, S. Y. R. Hui, and K. K. Tse, “|Reduction
of Power Converter EMI Emission Using Soft-
Switching Technique”, IEEE Transactions on
Electromagnetic Compatibility, vol. 40, no. 3,
August1998,pp. 282-287.
[4] Ivo Barbi, J.C. Bolacell, D.C. Martins, and F.B.
Libano,“Buck Quasi – Resonant Converter
Operating at Constant Frequency : Analysis, Design
and Experimentation”, IEEE Transactions on Power
Electronics, vol.5,no.3,1990, pp.873-880.
[5] Jung G. Cho and Gyu H. Cho, “Single-Cycle
ResonantConverters: A New Group of Quasi-
Resonant Converters Suitable for High-Performance
dc/dc and ac/ac Conversion Applications”, IEEE
Transaction on Industrial Electronics, vol. 38, no. 4,
August 1991, pp.260-267.
[6. Wojciech A. Tabisz, Pawel M. Gradzki, and
Fred.C.Y.Lee, “Zero-Voltage-S witched Quasi-
Resonant Buck and Flyback Converters-
Experimental Results at 10 MHz”, IEEE
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April 1989, pp. 194-204.
[7] Kwang-Hwa Liu, and Fred C.Y.Lee, “Zero-Voltage
Switching Technique in DC/DC Converters”, IEEE
Transactions on Power Electronics, vol. 5, no. 3,
July 1990, pp. 194-204.
[8] Guichao Hua and Fred C. Lee, “Soft-Switching
Techniques in PWM Converters”, IEEE
Transactions on Industrial Electronics, vol. 42, no.
6, December 1990, pp. 595-603
[9] B.P. Divakar and D. Sutanto, “A Novel Converter
for Fuel-Cells Applications”, IEEE International
Conference on Power Electronics and Drives
Systems, June 2005, vol.2, pp.162-165.
[10] Ying-Chun Chuang and Yu-Lung Ke, “A Novel
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129 D. Govind, Nandkumar Wagh
International Journal of Electronics, Electrical and Computational System
IJEECS
ISSN 2348-117X
Volume 6, Issue 6
June 2017
[11] Robert C. N. Pilawa-Podgurski, Anthony D.
Sagneri, Juan M. Rivas, David I. Anderson, and
David J. Perreault, “Very-High-Frequency Resonant
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1664.
Govind.D. was born in nizamabad, India, on April
18, 1988. He received the B.Tech degree in
electrical and electronics engineering from
Jawaharlal Nehru Technical University, Hyderabad,
India, in 2008 and the M.Tech degree in electric
drives and power electronics at the Indian Institute
of Technology, Roorkee. His current research
interests include power electronics applications and
drives, multilevel inverter, power-quality (PQ),
solar photovoltaics and application.
Nandkumar Wagh was born on 20th December
1962. He obtained his U.G. and P.G. degree from
Government College of Engineering Amravati
(M.S.). He obtained the Ph D in Electrical
Engineering from Maulana Azad National Institute
of Technology, Bhopal (M.P.) His research interest
is specifically focused in artificial intelligence
applications to Electrical Power System and Power
Electronics and Drives. He has more than 25
national and international publications to his credit.
He is a member of professional societies such as
IETE, IE (I), ISTE, and IENG.