ieee a 2.4 ghz radio solution

7/27/2019 IEEE a 2.4 GHz Radio Solution

1/1170 International IC Taipei Conference Proceedings

A 2.4 GHz radio solutionfor Bluetooth and wireless

home networking

AbstractA complete radio transceiver solution for the 2.4GHz ISM band

demonstrating a high level of integration along with reduced

component count and board size has been constructed. A single

conversion receiver and a direct conversion transmitter have

been fabricated on a single integrated circuit (IC). The archi-

tecture and functionality of the system blocks are described in

detail. In addition, the performance of the transceiver is evalu-

ated for several emerging wireless data and voice standards in-

cluding Bluetooth.

The first section of the paper concentrates on the system

requirements of the transceiver as used in the 2.4GHz ISM bandincluding Bluetooth, HomeRF, and upbanded DECT standards.

The subsequent sections will present details of the various cir-

cuit blocks in a complete 2.4GHz ISM transceiver solution and

the enabling BiCMOS transceiver IC. Conclusions will be

drawn about the impact on size, cost, and performance of mo-

bile terminals.

I. IntroductionEmerging applications in the unlicensed 2.4 GHz Industrial,

Scientific & Medical (ISM) band have generated a great amount

of interest in the wireless communications industry worldwide.

One of the main reasons for this is the availability of the spec-trum around the world. As shown in Table 1, spectrum within

the 2.4 GHz frequency range is common to Europe, Japan, and

the USA. This commonality allows manufacturers to develop

one product to address the global market with minimal changes.

Couple this with the need for connectivity in the exploding

Home PC market, handheld service access devices, and infor-

mation appliances and you have a very compelling segment of

the wireless communications market.

New standards have also been developed in accordance to

this available frequency range and FCC part 15 [1]. Standard

specifications are also attractive to product manufacturers. Not

only do the open standards provide an overview of protocol,

product capabilities and features, but they also help to guaran-

tee reliable high-quality radio links, and interoperability be-

tween products of any brands. This equates to customer accep-

tance, and market penetration.

There are some specific applications, however, where stan-

dard protocol may not be an immediate necessity. For example,

in wireless home networks and cordless phones, several com-

panies are providing proprietary solutions available in the mar-

ket today. Much recent development activity is occurring in

Bluetooth and HomeRF solutions [2],[3]. These two standards

have hundreds of member companies in their respective work-

ing groups, and have recently released version 1.0 specifica-

tions. Both Bluetooth and HomeRF are loosely based on the

Digital Enhanced Cordless Telephony (DECT) standard popu-

larized in Europe [4]. Both of these standards are specifically

designed with frequency hopping algorithms to work well inthe presence of microwave ovens, and also include paging

modes to maximize battery life.

II. StandardsBluetooth advertises Wireless Connections Made Easy. Ini-

tially it is a standard that will deploy radio-based wireless ports

replacing the IR-based wireless ports (IRDA) with 10m range.

Bluetooth offers freedom from IRDAs line-of-sight require-

ments and allows users point-to-multipoint connectivity

Bluetooth acts like a radio-based wireless pico-LAN. The

objective of the Bluetooth Group is to get the radio-based wire-

less port adopted on the motherboard, in peripherals, into PDA

products and new digital cell phones. Whereas Bluetooth aims

at connecting portable devices (10-100m), the Shared Wire-

less Access Protocol -Cordless Access or SWAP-CA, com-

monly referred to as HomeRF, is focused on the Home Net-

working (300 m).

William O. KeeseHead of the Applications Group

National Semiconductor

Vikas VinayakSenior Applications Engineer


*except Spain and France

Table 1: Frequency Allocation

Christopher LamApplications Engineer



2/11International IC Taipei Conference Proceedings 71

The mission of the HomeRF Working Group is to enable

the existence of a broad range of interoperable consumer de-

vices via RF digital communications, in and around the home.

HomeRFis a PC-centric wireless solution that provides a gate-

way between the PC, the PSTN and other HomeRF nodes. It

supports Isochronous (I), Asynchronous (A) and combined I/A

data transfer.

HomeRF operates at 100 mW transmit power at 50 Hops/s.

It also supports up to 6 wireline quality voice connections basedon 32 Kb/s ADPCM and DECT call processing. Some of the

system specifications are compared in Table 2.

Another option providing a quick time to market solution is

moving DECT to the 2.4 GHz band. Several manufacturers have

introduced products based on 2.4 GHz DECT available today.

Because DECT is a mature high performance cordless com-

munications standard, there exist many advantages in this tech-

nology [5]. The radio architecture and TDMA controller can

remain the same as for DECT.

Higher protocol layers can remain unchanged, and full

DECT throughput can be maintained. Interaction with WANs,

POTS, ISDN, and frame relay has already been specified and

proven in Europe.One thing worthy of noting is the typical DECT modulation

(1.152Mbits/s GFSK, modulation index = 0.5) is not 20dB down

at 1 MHz away as specified in [1]. Two options exist to achieve

compliance: 1) lower the bit rate, or 2) decrease the modulation

bandwidth. Lowering the bit rate to 1Mbits/s while keeping the

same BT product and modu-

lation index will decrease the

Tx spectrum enough to meet

the spectral requirements, and

have a minimal impact on the

link channel budget. By

changing the BT product andmodulation index to 0.35, the

spectrum can also be met at

full DECT data rates, but the

increased ISI will cause deg-

radation in the link channel

by several dB. Theoretically,

you could also keep the origi-

nal 1.152MHz channel spac-

ing, because the FCC states

only that the channels must

have at least 1MHz spacing.

However, since 2.4GHz

DECT operation exists as

largely proprietary solutions,

it is feasible to change to a

narrower 1.0MHz channel

spacing, and even change the

time slot structure to accom-

modate non zero blind slot

operation.

An overview of the dif-

ferent standards comparing

some of the key parameters

is shown in Table 2. Param-

eters from Bluetooth andHomeRF are derived directly

from [2] and [3] respectively.

As stated previously, 2.4GHz

DECT solutions today are

mostly proprietary solutions, therefore are subject only to FCC

part 15, rather than a rigorous type approval. Specifications

for upbanded DECT have been adapted from [4] where appli-

cable for comparison. The similarities and differences between

the standards are evident from examining Table 2. The

Bluetooth and HomeRF performance requirements are sig-

nificantly relaxed from the DECT standard. Notably, the mini-

mum sensitivity is reduced considerably, and the in-band

blocking levels are also not as stringent. HomeRF particu-larly has a relaxed interference specification, with no adja-

cent channel specification, only -10dB of C/I required at 3

channels away, and an intermodulation protection (IMp) re-

quirement of only -39.5dBm.

The receiver intermodulation performance is typically mea-

sured using 3 signals: a desired signal and 2 undesired block-

ing signals. All three signals are on 3 different RF channels.

One of the unwanted signals is a modulated signal and the

other is not modulated (CW). The frequencies of the unwanted

signals are chosen such that one of their third-order

intermodulation products (IMp) will appear as an unwanted

interferer on the same channel as the wanted signal, or in other

words as a co-channel interferer.For example in Bluetooth, the level of the wanted signal is

specified to be PREF = -64 dBm and the level for each of the

two unwanted signals is PINT = -39 dBm. The CCI needed is

11dB. Assuming the wanted signal is much greater than the

sensitivity limit of the device, no contribution from the noise

Table 2: Parameter Comparison for 2.4GHz Applications



floor of the wanted band is added to the CCI number.

For IMp, the following is needed:

System IIP3 = 1/2( PINT - PREF +CCI) + PINT =

+ 1/2[(-39) - -64 + 11] -39 = -21dBm

Since there is no direct co-channel interferer (CCI) specifi-

cation in HomeRF [3], it is inferred by the clear channel

assesment level of -80dBm. With an input level 3 dB above thesensitivity limit, -73 (same as IMp) this gives a CCI of -7dBc.

The phase noise numbers are a combination of the Tx and

Rx requirements and are also inferred from the specifications.

In order to relate the noise as dBc/Hz, the spectrum must betranslated to dBc/Hz, which can be done as:

10log(2*BW) = 10log(1 MHz) = 60dBc/Hz

For the transmit, the VCO phase noise is assumed as the

only contributor to the noise generated in adjacent channels.

For example, with a transmit power of 100mW, the noise in the

+/- 1MHz adjacent channels due to the VCO is determined for

Bluetooth as:

- 10log(100mW/1mW) - 60dBc/Hz =

- 80dBc/Hz @ 550 kHz

In the receive mode, reciprocal mixing can occur when an

interfering signal mixes with residual noise of the VCO creat-

ing an in-band interferer (CCI). For Bluetooth, the required C/

I is 11 since it is a Co-Channel Interferer that is created. For

example, the requirements of the +/- 1MHz adjacent channel

are calculated as below (adding 60dBc/Hz for the 1MHz mea-

surement BW):

0dB - 11dB - 60 dBc/Hz =-71dBc/Hz @ 1 MHz

Note, that if no channel filtering is provided at IF, and a

second down conversion is to be used, the sum of the two VCOsphase noise is required to meet the above requirements. Some

additional margin may be desired as the actual adjacent chan-

nel covers offsets of N*1.0MHz +/- the modulation spectrum.

III. RF TransceiverA radio transceiver was designed for Frequency Modulation

(FM) based schemes such as GFSK [7]. These modulation

methods are popular for low cost, low power applications. The

high tolerance to system non-linearity allows for decreased

operating current and lower voltage headrooms. In particular,

the transmitter side of the radio benefits enormously. There is

only a single frequency modulated carrier which is insensitive

to amplifier non-linearities[8],[9]. Non-coherent demodulationmeets the Bit Error Rate (BER) performance requirements of

these protocols and

translates to simpler

transceiver architectures

reducing the cost of the

solution[10],[11].

The block level

schematic diagram of

the transceiver is shown

in Figure 1. It consists

of a BiCMOS radio

transceiver IC, a power

amplifier, VCO, voltageregulator, Transmit/Re-

ceive (T/R) switch, ce-

ramic filter and SAW

filter.

The LMX3162

BiCMOS IC [13] con-

tains the phase lock loop

(PLL), transmit and re-

ceive functions. The

1.3GHz PLL is shared between transmit and receive sections.

The transmitter portion of the LMX3162 includes a frequency

doubler and a high frequency buffer and employs direct VCOmodulation. The receiver part consists of a 2.5GHz low noise

down converting mixer, an intermediate frequency (IF) ampli-

fier, a high gain limiting amplifier, a frequency discriminator, a

received strength signal indicator (RSSI), and an analog DC

compensation loop. The receiver section has single conversion

architecture and the received signal is demodulated by a Quadra-

ture Discriminator[12]. The IC features an on chip regulators

to allow supply voltages ranging from 3.0 to 5.5 volts, and is

specified for operation from -10 to +70 degree centigrade. Two

additional voltage regulators provide a stable supply source to

external discrete stages in the TX and RX chains.

The ceramic filter is shared between transmit and receive

sections, while the SAW filter provides selectivity at IF. The

RSSI output may be used for channel quality monitoring and

regulation of transmitted power, as required by HomeRF and

Bluetooth respectively. The external regulator supplies the VCO

and prevents frequency pulling. The T/R switch enables either

the transmit section or the receive section to connect to the an-

tenna. The power amplifier is implemented using a discrete bi-

polar transistor. An external VCO module provides the local

oscillator.

A. PLL SectionThe transceiver contains a phased locked loop (PLL). The PLL

runs at one half of the ISM band (2.4 - 2.5 GHz) and employsan integrated frequency doubler to synthesize the desired fre-

quencies as shown in Figure 1. This architecture alleviates the

disturbance to the local oscillator (LO) when the power

Figure 1: Transceiver Schematic



amplifier (PA) is switched on. The radiation is isolated by off-

setting the PA output frequency from the LO frequency.

The PLL is shared between transmit and receive sections

since HomeRF, Bluetooth and Upbanded DECT are half-du-

plex TDMA systems. The data is transmitted and received in

different time slots. The length of time is determined by the

protocol employed. The PLL must be able to hop to the desired

carrier frequency in a given amount of time before data trans-

mission and reception starts. This time is called the lock time.It is defined as the time the PLL takes to settle down to an

acceptable frequency error. Table 2 shows that the demanding

requirement for lock time comes from HomeRF. Requirements

for Bluetooth (220 ms) and Upbanded DECT (416.67 ms with 1

blind slot) are more relaxed. Most Upbanded DECT solutions

employ blind slot operation, which allows certain latency and

also provides enough time to acquire lock.

To improve performance as well as reduce the cost of imple-

mentation, the transceiver transmits and receives data in the

open loop mode [6]. The PLL is first locked at the desired car-

rier frequency and then shut down during the data transmission

or reception. In this short duration, the VCO is subsequently

modulated by the baseband signal in transmit mode or idling in

the receive mode. There are two issues of concern with open

loop mode of operation.

The first issue is the drift of the VCO as the loop capacitors

discharge or charge because of the leakage currents associated

with the charge pump of the PLL and the varactor of the VCO.

This implies the drifting of the VCO must be negligible com-

pared to the carrier frequency. In order to achieve low frequency

drift (



frequency is doubled and is also used as the PLL input. The

frequency doubled signal is then buffered and is output from the

LMX3162 at -7.5dBm. This signal needs to be high pass fil-

tered to reduce the level of what is now a sub-harmonic at half

the frequency. The level of this unfiltered harmonic is about

11dB below the desired output. The amount of amplification

required by the next stage, off-chip amplifier depends upon the

protocol being employed. Table 2 shows the required levels.

This transmit section of this radio was tested for Bluetooth op-eration, in the low power mode. The transmit power required is

0dBm. Adding to this 1dB of loss in the T/R switch and 2dB

loss in the post amplifier ceramic bandpass filter, the power

output from the PA desired is 3dBm, computing to a gain of

10.5dB. The amplifier was designed on a Siemens BFP420, a

25GHz bipolar transistor. The amplifier runs off a voltage regu-

lator internal to the LMX3162, which has a 2.7volt output rated

at 10mA. The transistor was biased at a collector-emitter volt-

age of 1.7volt with a quiescent current of 10mA. It delivers

+3dBm when its input is connected to the output of the

LMX3162.

Some Bluetooth options, and also other protocols, require

higher transmit powers. These are in the range of +20dBm andabove. This mandates the use of high efficiency power amplifi-

ers. Some suitable parts are the ITT2302 for upbanded DECT

and the ITT2304 for Bluetooth and HomeRF from ITT

GaAsTEK [16]. These amplifiers interface to the -7.5dBm out-

put of the LMX3162 directly, and also offer an on board T/R

switch, 14dB Gain LNA and 3.6volt operation. The BFP420

based PA is not required for the ITT2304.

After power amplification, the transmit signal is filtered to

remove all the harmonics and the sub-harmonics of the carrier.

This is required by regulatory organizations such as FCC and

also by the 2.4GHz protocols. The spurii levels acceptable are

dependent on the protocol employed. The filter chosen is a twostage ceramic filter from Murata, part number

DFC22R44P084LHA. It has a 3dB bandwidth of 84MHz cen-

tered at 2442MHz, and a typical insertion loss of 2dB. For higher

performance, Murata offers a 3 stage ceramic filter in the same

footprint. The wideband output and unmodulated transmit spec-

tra are shown in Figure 3 and Figure 4 respectively[v1]. The

sub-harmonic meets the Bluetooth specification (-36dBm)

without any filtering. A low pass filter can be added for addi-

tional margin. The ceramic filter is shared by the TX and the

RX modes, and a TX/RX switch is placed between the PA and

the T/R switch.

The unmodulated signal shows the phase noise of the trans-

mit signal. This signal is also used as a Local Oscillator (LO)

in the receive mode, and has to satisfy certain requirements of

phase noise in both the TX and RX modes. These requirements

are shown in Table 1.

Figure 5 shows a PRBS15 modulated Bluetooth signal,

which requires baseband Gaussian filtering with BT=0.5 andFSK modulation with typical peak-to-peak frequency devia-

tion of 320kHz. Figure 5 [v2]also shows the superimposed spec-

trum of a similar signal generated by a HP ESG4433 signal

generator. The trace with the larger first sideband [v3]is from

the signal generator. FCC requires the radiated power at off-

sets of 500kHz or more to be 20dB less than the peak power

being transmitted, all measurements being made with in a

Figure 3: Harmonics of Tranmitter Output

Figure 4: Phase Noise of Local Oscillator

Figure 5: Bluetooth Modulated Signals



100kHz bandwidth. The Bluetooth signal easily meets this

requirement.

The VCO gets its power supply from an external low noise

voltage regulator, the National Semiconductor LP2980. This

regulator isolates the supply to the VCO from the power supply

to the other parts of the radio and various transients. Most im-

portant of these transients occurs when the Power Amplifier

(PA) is switched on. The peak power required to be delivered

can be in excess of +20dBm. At typical operating voltages of2.7 to 3.6 volts, the supply currents of these PAs are greater

than 150mA. This can result in IR voltage drops from the inter-

nal resistance of the battery and traces on the PCB. The VCO

has a worst case pushing figure of approximately 5MHz/V. The

VCO used oscillates at half the desired transmit frequency so

the pushing figure also doubles to 10MHz/V. Most 2.4GHz pro-

tocols require frequency accuracies of the order of +/-50kHz

(See Table 2). This is the total frequency accuracy required of a

transmitter and includes effects due to the temperature coeffi-

cient and initial accuracy of the crystal employed for the PLL,

the frequency jump resulting from opening the PLL, and fre-

quency pushing of the VCO. A typical error budget distribution

would allocate +/-20kHz out of the total +/-50kHz to errorscaused by pushing. Dividing this by 10MHz/V, the allowed noise

on the power supply pin of the VCO is +/-2mV. Now consider

that the connection of the battery to the PA and the VCO, and

that the two traces have 2cm in common. Assuming 35 micron

thick copper traces which are 12mil wide, the series resistance

of the trace evaluates to about 22 milliohms, assuming resistiv-

ity of 20 nano-Ohm/m for copper. The internal resistance of a

typical Ni-MH battery used in handsets is about 0.25 Ohms.

The total series resistance becomes 0.272 Ohms[v4], and the

IR drop computes to 40.8mV. This is far in excess of what is

allowed. The voltage regulator shields the VCO from all such

switching events.

C. Receive SectionA ceramic filter selects the desired frequency band received by

the antenna. The LMX3162 processes the received signals at

an IF of 110.5MHz, and the LO is chosen to be 110.5MHz be-

low the RF Input. The ceramic filter rejects the image frequency

221MHz below the RF Input. It also removes or attenuates out

of band blockers.

While the transceiver meets the BER requirements of

Bluetooth and HomeRF, some applications require the use of

an LNA external to the LMX3162. A LNA built using a BFP420

can achieve 13dB gain and 2dB noise figure. It can run off the

second voltage regulator internal to the LMX3162. A typical

schematic for the LNA is shown in Figure 6.

Once the received signal enters the LMX3162, it is

downconverted by the low noise 2.5GHz mixer to 110.5 MHz.

The mixer has 17dB gain with 11dB Noise Figure. The OIP3 of

this mixer is 7.5dBm. The LO is derived from the frequency

doubler internally, and in the receive mode the TX output buffer

of the LMX3162 is shut down to conserve power.

After the RF signal gets downconverted to IF, it needs to be

bandlimited for channel selection. Filtering is very helpful in a

multi-signal environment. The IF filter provides the desired se-

lectivity and prevents generation of spurii in the Limiter and

the Quadrature Detector. Limiters are inherently very nonlin-ear devices and in a hostile environment stronger, unwanted

signals can capture the receiver if no IF filtering is employed

[18]. Bandlimiting is also required to provide an optimum BER,

the primary requirement for which is to limit the noise power

as much as possible. This filtering can be done before the

Quadrature Detection and after it. However, it is a known fact

that discriminators have non-linear performance with respect

to the input signal-to-noise ratio [19]. This is called the Thresh-

old Effect. The discriminator needs to see a signal-to-noise ra-

tio that is above a certain threshold, otherwise its performance

degrades very rapidly. Hence there is a requirement to filter the

IF signal early on in the chain.

In the radio the IF Filter is implemented by a high volume,

readily available DECT SAW Filter. The SAFU110.6MSA40T

[17] Murata SAW filter is centered at 110.6 MHz, and has a3dB bandwidth of 1.5 MHz. The minimum insertion loss is about

3dB. The SAW filter is matched to LMX3162 using two induc-

tors and two capacitors. As the market for 2.4GHz products

increases, SAW filters that have been designed specifically for

these protocols will be made available by various vendors and

improve the performance of these radios.

After the initial filtering, the IF signal is amplified by the IF

Amplifier and fed to the IF limiter. A resistor and two capaci-

tors provide some filtering.

The LMX3162 demodulates the IF signal by quadrature de-

modulation [12]. The phase shifting tank consists of a capaci-

tor, an inductor, and a varactor. The varactor is used to tune the

tank, along with its associated parasitics, at exactly 110.5MHz.

The voltage at the output of the quadrature discriminator is mea-

sured by the baseband controllers ADC and compared to a ref-

erence, and the generated error signal is amplified and con-

verted back to an analog signal by using a Digital to Analog

Converter (DAC). The scheme is shown in Figure 7.

Using elements with different Qs and values can change

the Q of the tank. Higher Qs mean a narrower bandwidth and

greater sensitivity. This translates to better SNR and also greater

ISI [20][21]. Conversely, lower Qs result in slightly lower SNR

and lesser ISI. The LMX3162 has a 1pF quadrature shift ca-

pacitor internal to it.

Once the signal has been demodulated, it is passed throughan active low pass filter. This further limits the noise band-

width of the system and provides larger peak to peak voltage

output. The quadrature tank and the active LPF are shown in

Figure 7.

Figure 6: PA and LNA Schematic



The LMX3162 provides a feature that eliminates the effect

of any initial frequency offset between the Transmitter and the

Receiver. Most RF protocols have an initial Synchronization

Field during which a string of alternating 1s and 0s is trans-

mitted. During this time, the demodulated signal is well known

and its average represents the nominal center frequency of the

transmitter. This average level can be captured by a Sample-

And-Hold (S&H) circuit and used as a reference level for a

data slicing comparator which converts the demodulated signal

into logic level 0s and 1s. The LMX3162 features an onboard

S&H. An internal resistor of 3kW and external capacitor

together provide the averaging time constant. The S&H actionis controlled by the S_Field input of the LMX3162, as shown

in Figure 7.

IV. Performance MeasurementsA. BERThe transceiver achieves a BER of 1E-3 for a Bluetooth modu-

lated signal at an input power of -83dBm, with the received

signal being offset by the +/-

115kHz frequency offset al-

lowed by Bluetooth. In this

measurement the transceiver

does not employ an LNA and

uses a low cost DECT SAW

filter. This easily meets the

Bluetooth minimum sensitiv-

ity requirement of -70dBm.With the use of a high per-

forming LNA and an opti-

mally chosen SAW filter, sen-

sitivity of -96dBm can be

achieved. The eye diagram of

the received signal for 1E-3

BER is shown in Figure 9,

and Figure 10 shows the eye

diagram for an input level of

-40dBm.

B. Synthesizer

The frequency offsetsallowed are given in Table 2[VV5]. The transmitter must be

within this limit and the receiver must be able to receive with

this frequency offset. A typical jump on opening the loop and

open loop frequency drift for a 500uS period at room tempera-

ture is shown in Figure 8. The parameters affecting these per-

formance criteria have been discussed in Section III.A. The fre-

quency jump is independent of the operating temperature, but

the frequency drift doubles every 10 degree centigrade follow-

ing the same law as the leakage currents it is caused by do. The

typical drift measured indicates a combined LMX3162 charge

pump, VCO varactor and PCB surface leakage current to be

around 55pA. The maximum operating temperature specifiedby Bluetooth is 35 Degrees ambient. Adding to this 10 degrees

due to self-heating of the product, the drift specification must

be met at 45 degrees Centigrade. This is 20 degrees above the

room temperature, and quadruples the leakage current to 220pA.

The maximum continuous Transmit time for Bluetooth is 5 slots,

or 3.125ms. Bluetooth has a channel spacing of 1 Mhz, but the

IF of the radio is at 110.5MHz. This means the LO must be

synthesized with a frequency resolution of 0.5MHz. The PLL

output is doubled in frequency, and this halves the resolution

required from the PLL to 0.25MHz. For 0.25MHz frequency

resolution, the phase comparison frequency that must be used

is 0.25MHz. The lock time requirements for Bluetooth are

220uS, and budgeting for margins and programming time re-

duces this to 180uS. To achieve a lock time of less than 180uS

with a phase comparison frequency of 0.25MHz requires a

total loop filter capacitance of about 5nF. Using this value, the

frequency drift on account of leakage is calculated as:

Df = ILeakage Dt KVCO/C

The drift in 3.125ms is about 20kHz. This meets the

Bluetooth requirement of 40kHz. This is because the National

Semiconductor proprietary charge pump provides excellent

leakage characteristics (typical 700pA at 85C).

As discussed in Section III.A, the lock and drift require-ments of HomeRF are demanding. The Alps VCO employed

had a KVCO of 140MHz/V, referred to 2.4GHz. The phase

detector comparison frequency was 250kHz. A second order

Figure 7: Quadrature Demodulation

Figure 8: Open Loop Frequency Jump and Drift



loop filter is used. The loop bandwidth is 40kHz and the phase

margin is 5.

Figure 2 shows the component values of the filter. The lock

time is measured from the falling edge of signal that shuts down

the PLL (PLL_PD) to 1225 MHz (2450 MHz at Tx Output)

within 5 kHz (10 kHz at Tx Output). The lock time is 111ms as

shown in Figure 11.

The frequency drift due to the current leakage is measured

starting from the rising edge of PLL_PD. Figure 12 demon-

strates the frequency drift of the VCO due at 85C. The very low

drift of 778Hz/ms is much better than what is required for

HomeRF.

Figure 12 shows two frequency jumps. The first is due to

the phase noise of the PLL and appears as a jump due to the

finite response time of the modulation domain analyzer. Thesecond jump is due to the common mode operating range of the

op-amp. Even though the control voltage line varies a few hun-

dred mVs, the frequency jumps tens of Kilohertz because of the

large VCO gain and the bias current coming out of the op-amp.

Placing a 10-MW resistor in parallel with the loop filter will

sink away the bias current of the opamp and prevent saturation.

The occurrence of the second jump depends on the selection of

the opamp and the CMOS switch. In the current design, the

second jump occurs only after the loop is open for a long time

(>100 ms). In case of HomeRF, the loop will not be idle or

open for such long time.

C. InterferenceAnother difference between ideal and real operating conditions

for a receiver is the presence of signals other than the wanted

ones. This is especially true for an unlicensed environment like

the 2.4GHz band. A typical transceiver must operate in a multi-

transmitter environment, with unwanted signals that are either

on the desired or on adjacent channels. These hostile signalscan capture the desired channel, cross-modulate information

on to the desired signal, or simply reduce the demodulated sig-

nal to noise ratio and lower the BER sensitivity of the radio.

Large, out of band signals can even desensitize the radio

Figure 9: Eye Diagram for BER=1E-3

Figure 10: Eye Diagram for -40dBm Input

Figure 11: PLL Lock Time

Figure 12: Open Loop Frequency Drift at 85C



completely. The performance of the radio is shown in Table 3

and is seen to meet all the requirements.The second test in the table is Co-Channel Interference

(CCI). The radio meets a BER of 1E-3 for CCI 11dB below for

a desired input down to -80dBm. At the specified desired sig-

nal level of -60dBm, the radio even meets the BER for a CCI

that is only 6db. Bluetooth requirements have relaxed for a pe-

riod of three years. During this period, the radios may meet a

relaxed CCI specification of 14dB. For this CCI level, this ra-

dio suffers no degradation in BER sensitivity and performs at -

83dBm. Figure 13 shows the eye diagram for the Bluetooth

CCI test signal.

The third test is first of the Adjacent Channel Interference

(ACI) requirements the radio has to meet. A radio must per-

form at 1E-3 BER with an undesired, Bluetooth signal interfer-

ing at 1 MHz offset. The desired signal must be at the same

level as the interferer, both of them being at -60dBm. The three

year relaxed specification for the 1MHz test is an ACI 4dB be-

low the desired signal. The radio meets the relaxed specifica-

tion. The low cost, off-the-shelf-available SAW filter selected

for this radio has a 3dB bandwidth of 1.5 MHz and is wider

Table 3: Interference Test for Bluetooth

Figure 13: Eye Diagram for Bluetooth CCI Test

Figure 14: Eye Diagram for Bluetooth ACI 1MHz Test


Figure 16: Demodulated Signal for Bluetooth ACI 2MHz Test



than what is required for a Bluetooth signal. Figure 14 shows

the eye diagram for the Bluetooth ACI 1MHz test signal.

The radio meets the 2MHz ACI specification for levels from

-81dBm to -30dBm. Figure 15 shows the eye diagram for the

Bluetooth ACI 2MHz test signal. Figure 16 shows a single trace

of the demodulated signal, and a 2 MHz signal is clearly visible.

The fifth test measures the performance of the radio for a

Bluetooth ACI that is 3MHz offset. The radio meets the speci-

fication from -72 to -35dBm. Figure 17 shows the eye diagramfor the Bluetooth ACI 3MHz test signal. Figure 18 shows a

single trace of the demodulated signal, and a 3MHz signal is

clearly visible.

The seventh test measures the intermodulation properties

of the radio. The test requires the desired signal to at -64dBm,

a -39dBm static sine wave at 3 MHz offset and another

Bluetooth signal at -39dBm at 4MHz offset. The radio meets

the requirements. Figure 19 shows the Bluetooth

intermodulation test signal.

The radio also meets the performance requirements for out

of band interferers.

D. TransmitterThe transmitter section of the radio too must meet some speci-

fications. Being unlicensed radiators, these radios have to com-

ply with the FCC requirement that the radiated power at offsets

of 500kHz or more must be 20dB less than the peak power

being transmitted, all measurements being made in 100kHz

bandwidth. In addition, Bluetooth requires the power at 2 and

3MHz offsets to be below -20dBm and -40dBm respectively,

all measurements being made in 100kHz bandwidth.

Figure 5 shows that that the radio meets these requirements

when used as a Bluetooth transmitter. Table 4 shows the out of

band spurious emission requirements of Bluetooth, and Figure

3 shows that these are met.

V. ConclusionThis paper has covered key design, specification and test is-

sues for emerging 2.4GHz wireless communication applications.

Three 2.4GHz protocols and their impact on transceiver speci-

fications were discussed. A new single chip BiCMOS radio

transceiver was presented. A radio built around this chip was

made and its performance measured for the 2.4GHz protocols.

The single chip transceiver itself contains most of the function-

ality required for 2.4GHz radios and is an enabling component

of low cost, small size wireless applications.

This paper is published with due permission from Penton Publishing for the

IIC-Taipei 2000 conference.


Figure 18: Demodulated Signal for Bluetooth ACI 3MHz Test

Figure 19: Bluetooth Intermodulation Test Signal

Table 4: Out of Band Spurious Emission for Bluetooth


11/11

AcknowledgementsThe authors would like to thank Eric Lindgren, Doug Steen,

Tai Wong, Erik Ankney, John Lund, Finn Anderson, and Jim

Stubstad whose contributions were instrumental in developing

the single chip transceiver solution.

References[1] Document CFR47, Part 15, Sections 15.205, 15.209,

15.247, Federal Communications Commission, FCC,USA. www.fcc.gov

[2] Specification of the Bluetooth System, Version 1.0 Draft

Foundation, www.Bluetooth.com

[3] HomeRF SWAP Specification, www.HomeRF.org

[4] DECT Reference Documents ETS 175 1-9, European

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signing for DECT, RF Design Magazine, April 1993.

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[7] S. Sampei, Applications of Digital Wireless Technolo-gies to Global Wireless Communications, Prentice Hall,

NJ, USA, pp. 71-72.

[8] S. Sampei, Applications of Digital Wireless Technolo-

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NJ, USA, pp. 55-57.

[9] B. Razavi, RF Microlectronics, Prentice Hall, NJ, USA,

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[10] B. Razavi, RF Microlectronics, Prentice Hall, NJ, USA,

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[11] M. S. Roden, Digital Communication System Design,

Prentice Hall, NJ, USA, pp. 376-385.

[12] E. A. Richley, Design of Quadrature Detectors, RF De-sign Magazine, Vol.14, No. 5, May 1991, pp 68-72.

[13] LMX3162 Datasheet, National Semiconducotor,

www.national.com

[14] Alps Electric (USA), Inc., www.alps.com

[15] C. A. Harper, Passive Electronic Component Handbook,

p.80, 2nd Edition, McGraw-Hill, 1997.

[16] ITT GaAsTEK, ITT2302 and ITT2304 Datasheets,

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[17] Murata, RF and Microwave Products Catalog,

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[18] V. Manassewitsch, Frequency Synthesizers, John Wiley

& Sons, pp. 79-85.

[19] H. Taub and D. H. Schilling, Principles of Communica-

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[20] K. Murota and K. Hirade, GMSK Modulation for Digi-

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munication, Vol. COM-29, No.7, pp. 1044-50, July 1981.[21] J. D. Gibson, The Mobile Communications Handbook,

IEEE Press, pp. 535-536.

Authors contact detailsVikas Vinayak


Wireless Communications Division

2900 Semiconductor Drive

M/S D3-500, Santa Clara, CA 95052-8090 USA

Phone: 1 408 721 2228

E-mail: [email protected]

William O. KeeseNational Semiconductor



M/S D3-500, Santa Clara, CA 95052-8090 USA

Phone: 1 408 721 4494


Christopher Lam




M/S D3-500, Santa Clara, CA 95052-8090 USAPhone: 1 408 721 5724


This paper is published with due permission from

Penton Publishing for the IIC-Taipei 2000 conference.