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International Journal of Electrical Engineering & Technology (IJEET)

Volume 7, Issue 3, May–June, 2016, pp.93–105, Article ID: IJEET_07_03_008

Available online at

http://www.iaeme.com/ijeet/issues.asp?JType=IJEET&VType=7&IType=3

ISSN Print: 0976-6545 and ISSN Online: 0976-6553

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HIGH EFFICIENT BRIDGELESS BOOST

RECTIFIER FOR LOW VOLTAGE ENERGY

HARVESTING APPLICATIONS

K. Ajith

Assistant Professor,

Department of Electrical and Electronics Engineering,

Kakatiya Institute of Technology & Science,

Warangal, Andhra Pradesh, India

T. Praveen Kumar

Assistant Professor




V. Ramaiah

Professor




Sai Teja

PG Scholar




ABSTRACT

A single phase ac-dc bridgeless boost rectifier for low voltage energy

harvesting applications is proposed in this paper. The conventional bridge

type boost converters for low voltage energy harvesting requires more

components hence they suffer from high power loss and require more number

of energy storage components like inductors and capacitors. Conventional

converters can be modeled for boost operation or buck-boost operation alone.

The proposed converter overcomes the above mentioned draw backs of

conventional converter. Detailed analysis of proposed convertor is also

presented under boost, buck-boost mode operations. The proposed converter

K. Ajith, T. Praveen Kumar, V. Ramaiah and Sai Teja


operation is analyzed using MATLAB/SIMULINK environment both open loop

and closed loop conditions.

Keywords: Bridge Less Boost Rectifier, Buck-Boost Converter, Electro-

Magnetic Generator, Energy Harvesters

Cite this Article: K. Ajith, T. Praveen Kumar, V. Ramaiah and Sai Teja, High

Efficient Bridgeless Boost Rectifier For Low Voltage Energy Harvesting

Applications. International Journal of Electrical Engineering & Technology,

7(3), 2016, pp. 93–105.

http://www.iaeme.com/ijeet/issues.asp?JType=IJEET&VType=7&IType=3

1. INTRODUCTION

Wireless sensor systems are receiving increasing interest since they offer flexibility,

ease of implementation and the ability to retrofit systems without the cost and

inconvenience of cabling. Furthermore, by removing wires there is the potential for

embedding sensors in previously inaccessible locations. At present, the majority of

wireless sensor nodes are simply battery-powered. Despite measures such as low

power techniques for communicating (e.g. IEEE 802.15.4 and Zigbee protocols) and

the intelligent management of the sensor node’s power consumption, batteries will

still require periodical replacement. Replacing batteries is not compatible with

embedded applications nor it is feasible for networks with large numbers of nodes.

The advances made in low power wireless systems present an opportunity for

alternative types of power source. Solutions such as micro fuel cells and micro turbine

generators are capable of high levels of energy and power density. However, they

involve the use of chemical energy and require refueling. Energy harvesting

approaches that transform light, heat and kinetic energy available in the sensor’s

environment into electrical energy offer the potential of renewable power sources

which can be used to directly replace or augment the battery. Such renewable sources

could increase the lifetime and capability of the network and mitigate the

environmental impact caused by the disposal of batteries [1-2].

Kinetic energy harvesters convert mechanical energy present in the environment

into electrical energy. The past decade has seen an increasing focus in the research

community on kinetic energy harvesting devices. Typically, Kinetic energy is

converted into mechanical energy using electromagnetic, piezoelectric, or electrostatic

transduction mechanisms. In comparison to electrostatic and piezoelectric transducers,

electromagnetic transducers outperform in terms of efficiency and power density. The

limitation of the energy harvesters is low voltage output. In order to overcome this,

we employ bridge boost rectifier to step up the voltage. However, the bridge rectifier

would incur considerable voltage drop, making the low voltage rectification

infeasible. Though we employ CMOS diodes with low-voltage drops, they require

additional bias networks or external comparators. Thus, both complexity and power

loss of the circuitry would increase. Another approach to maximize the conversion

efficiency in low-voltage rectification is to use bridgeless direct ac-dc converters,

which employ split capacitors, bidirectional switches or two parallel dc-dc converters

to condition positive and negative input voltages separately. The above topologies

employ large capacitors to suppress ripple voltage, which makes these topologies

impractical due to limited size of energy harvesters. Also, the additional switches

would incur extra switching losses and driving circuit dissipations. To overcome

above drawbacks, a new bridgeless boost rectifier is considered with unique

integration of boost and buck-boost converters. The circuit employs only two power

High Efficient Bridgeless Boost Rectifier For Low Voltage Energy Harvesting Applications


electronic switches, simplifying the circuit and thereby reducing cost and size of the

rectifier. In addition, power losses are also minimized leading to high efficiency [1-5].

ELECTRO-MAGNETIC GENERATOR

A general diagram of an electromagnetic generator is demonstrated in Fig. 1, where k

is spring stiffness constant; m is the proof-mass; DE and DP represent electrical and

parasitic dampers, respectively. Essentially, the energy harvesting system consists of a

spring, a proof mass, and an electrical damper. The extrinsic vibrations excite the

internal oscillation between the proof mass (magnet) and electrical damper (coils) [2].

Figure 1 General diagram of electromagnetic micro-generator

The internal oscillation produces a periodically variable magnetic flux in the coil,

which induces a corresponding alternating output voltage. The output voltage level of

the micro scale and meso scale energy harvesting devices is usually in the order of a

few hundred millivolts depending on the topology of device. The output ac voltage

should be rectified, boosted, and regulated by power converters to fulfill the voltage

requirement of the loads. Nonetheless, miniature energy harvesting systems have rigid

requirement on the size and weight of power electronic interfaces. In energy

harvesting systems, power electronic circuit forms the key interface between

transducer and electronic load, which might include a battery. The micro generator

output has to be processed by a suitable power converter before providing to the

electronic loads. The electrical and physical characteristics of the power conditioning

interfaces determine the functionality, efficiency, and the size of the integrated

systems. The power electronic circuits are employed to

1. Regulate the power delivered to the load, and

2. Actively manage the electrical damping of the transducers so that maximum power

could be transferred to the load.

2. MATHEMATICAL MODELING

I. Large Signal Modeling

In order to develop a large signal model and analyze the steady-state operation of the

considered bridgeless converter circuit, several assumptions are made during one

switching cycle [3-6].

1. The output filter capacitor C is large enough to keep the output voltage Vo constant.

2. The input is a sinusoidal voltage source. The switching frequency is much higher than

the input voltage frequency. During each switching cycle, the input voltage could be

treated as a constant voltage source and be expressed as (1), where Ti is the period of

input voltage.

(t) = Vmsin (2πt/Ti) (1)

Internal series resistances of passive components are not taken into account for

convenience in calculations.



According to Fig. 2, during the boost operation cycle [t0,t3), the average transistor

voltage , can be expressed as

= (1-d1-d2) (t) + d2 (t) (2)

The average switch current is2a (t) is derived by integrating the is2 (t) waveform as

depicted in Fig. 6(a) during one switching period

is2a(t)=

(3)

The average diode voltage (t), and the average diode current id2a (t) can be expressed, respectively, as

(t) = d1 (t) + (1-d1-d2) ( (t) - (t)) (4)

Figure 2 Boost Operation Waveform

id2a (t) =

(5)

According to the inductor volt-second balance, d2(t) can be related to d1(t) as

d2 = d1

(6)

From (2) - (6), by manipulation we obtain the following,

=

(7)

From (7), we can derive that the average switch voltage is proportional to the averageswitch current is2a (t).In other words the switch behaves like an equivalent

resistor with resistance Re (d1), where

Re (d1) =

(8)

Also we obtain,

(t) (t) =

(9)

Substituting (8) into (9), the equivalent power out of the diode can be presented as



p (t) = (t) (t) =

(10)

According to (10), the diode behaves like a controlled power source, the power of

which is equals to the power consumption of the equivalent resistor Re (d1).Similar

derivations can also be applied to the buck-boost operation cycle [

), and the same

expressions can be derived. So, The switch behaves like an equivalent resistor Re (d1)

and the diode behaves like a controlled power source, the power of which is equals to

the power consumption of the equivalent resistor Re (d1).Thus, the switch network

could be modeled by a loss-free resistor and a dependent power source, as illustrated

in Fig. 3.

Figure 3 Averaged switch model of general two-switch network in Discontinuous

Conduction Mode (DCM)

The equivalent circuit of the topology in the steady state can be modeled by

replacing the switch network with its averaged model. In the steady-state condition, in

which the inductor and capacitor can be, respectively, replaced with short circuit and

open circuit, the circuit can be simplified to the circuits of Fig. 4, where d1(t) = D.

Figure 4 DCM dc equivalent circuits representing

(a) Boost operation and (b) Buck-boost operation

II. Converter Analysis

In boost operation mode [0, Ti /2), the instantaneous power delivered to the equivalent

resistor in dc equivalent circuit model can be expressed as

p(t) =

=

(11)

The instantaneous current flowing through the load resistor R can be expressed as

i(t) =

(12)

Hence, the instantaneous power delivered to the load is expressed as

pL(t) = Vo i(t) =

(13)



For each boost cycle [0, Ti /2), the total energy delivered to the load is obtained as

Ein =

(14)

Now, the energy consumed by the load is determined by

E0=

(15)

According to the principle of conservation of energy, the energy delivered to the load

is equal to the energy provided by the power supply.

Eo= Ein

=

(16)

The integral part of (16) is a function of boost ratio (Vm/Vo), which would be

infinitely close to Ti/2, if the ratio is much larger than 1. For low-voltage energy

harvesting, the input voltage is usually smaller than 0.5 V. However, most of the

electronic loads should be driven by a 3.3 V voltage. With the boost ratio larger than

6.6, the integral would become equal to Ti/2. Thus, (16) becomes

(17)

Similar equations can also be driven for the buck-boost operation (Ti/2 to Ti)

according to the model provided in Fig. 4(b).Different from the boost operation, the

power consumed by the loss-free resistor is all delivered to the load. The derivations

of the buck boost dc equivalent model lead to

(18)

From equations (16) and (18), the required relations of the converter can be

established. If the gain boost ratio is much higher than unity; equation (17) can be

preferred.

III. Design Procedure

If the voltage gain is much larger than unity, according to (17) and (18), the boost and

buck-boost operations share the same duty cycle D, which is determined by the boost

ratio (Vo/Vm), the inductance of the inductor L, and the switching frequency fs.

Equation (19) defines the commutation relationship of the duty cycle of the proposed

boost/buck-boost converter [13].

D = d1 = =

(19)

The boost ratio is defined according to the specific application, while the load

resistance R is dependent on the output power level. With the specified power and

voltage demands, the inductance is designed according to the desired range of duty

cycle and switching frequency. The larger the switching frequency is, the smaller the

inductance would be. In order to design a smaller inductor with purpose of obtaining

smaller size and weight, a higher switching frequency is preferred. However, the

higher the switching frequency is, the higher the switching loss would be. A tradeoff

between the size of inductor and the switching loss should be taken into account in the

design process. Both the boost and buck-boost operations of the converter provide the

same inductor current ripple, which can be expressed as



(20)

The maximum current ripple corresponds to the peak input voltage. According to

previous analyses, the inductor, diodes, and MOSFETs share the same value of

current ripple, which is defined in the following equation given by

(21)

From (21), the current ratings of all those components could be found. Thus the

value of inductor can be designed as follows: Let us assume maximum current flows

through inductor given by ipk.. Therefore the peak or maximum value of inductor

current is given by

D2 TS = ipk (22)

We have, Condition for having discontinuous conduction is

d2< 1- d1max (23)

Let us assume d1max=0.8 and then d2<0.2 for designing for discontinuous

conduction. From (22) we have

d2 =

< 0.2 (24)

L <

(25)

The voltage ratings of the MOSFETs and diodes are normally chosen higher than

V0 with an appropriate margin for safe operation. The turn-on resistances of

MOSFETs and the forward voltage drop of diodes are the major components, which

impact the efficiency. The output capacitor C is assumed to be very high so that it

filters out the voltage ripple and also maintains the output voltage V0 constant. The

charging and discharging currents of the capacitor decides the voltage ripple. The

value of bus capacitance can be calculated from the relation given below

C =

(26)

Where, = Peak inductor current

dV = Ripple voltage (0-10% of output voltage)

The validity of the assumptions can be checked by using the two given equalities

mentioned

Ts<< RC and

3. CONTROL STRATEGY

For a dynamic Electromagnetic energy harvester system, if the external excitation

frequency is different from the intrinsic resonance frequency, the Piezo Electric

Instrument (PEI) should be able to match its input impedance with the internal

impedance of the harvester so that maximum power point (MPP) could be tracked.

This paper proposed a new topology, which has the maximum power point tracking

(MPPT) capability. However, the main objective of this paper is to introduce the

circuit topology, which is capable of satisfying the voltage requirement (3.3 V) of an

electronic load. Thus, a voltage feedback control loop is utilized to regulate the load



voltage. The simplified scheme of the controller and power stage is illustrated in Fig.

5. The converter is designed to operate in DCM [13].

Figure 5 Control Circuit for the Boost Converter

The output voltage is filtered by a passive low-pass filter and then fed to the

analog-to-digital converter (ADC) of the controller. The difference between the ADC

output and the desired voltage is calculated and compensated through the PI algorithm

to generate an adjustable duty cycle signal. The switching signals of S1 and S2 are

dependent on the polarity of the input voltage. A sign detector is used to determine the

input voltage polarity. The sign detector is composed of a voltage reference, an op-

amp, and the on-chip analog comparator. The op-amp operates as an analog adder,

where a dc bias (voltage reference) is added to the input voltage. The signal

summation is compared with the voltage reference to detect the polarity [13].

4. SIMULATION & RESULTS

The proposed direct AC to DC converter circuit with boost and buck-boost converter

is simulated using MATLAB. The converter is simulated for both open loop and

closed loop operation. A closed loop operation of the circuit is modeled and

simulated. Here, the actual voltage is compared with the reference voltage (3.3V) and

the error signal is given to a PI controller. Then, a hysteresis control is given by using

a saturation block which limits the PI controller output (control voltage) to remain

within certain limits. The control voltage is again compared with a saw tooth

waveform from the signal generator so that gate pulses are being given to the boost

and buck-boost converter switches according to the detection of positive and negative

peaks, respectively. The simulation is done for obtaining a dc output voltage, VO

=3.3V from the electromagnetic micro generator which is modeled as a sinusoidal

voltage source with the specification, Vin = 400mV and frequency 100Hz. For

converter operation the SIMULINK model of boost and buck-boost converters

created, circuit parameters used for the simulation are given in the following Table 2.

Table 2 Components and Parameters in the Simulink

PARAMETER VALUE

VIN 0.4 V ,100 HZ

VO 3.3 V

Power Inductor 4.7 µH

Switching Frequency 50 kHz

Filter Capacitor 100 µF

Load 200 ohm



Theoretical Calculations: The corresponding duty ratio and theoretical values of the

converter for R= 200Ω and R = 300Ω are calculated using mathematical modeling

and designing procedure analyzed in previous chapters and are as follows in the Table

3 given below.

Table 3 Theoretical Calculations for Different Loads

Vin(V) VO

(V) FS(kHZ) L(µH) R

L (Ω ) D iL,max(A)

0.4 3.3 50 4.7 200 .5656 0.9532

0.4 3.3 50 4.7 300 .4618 0.786

The simulation has been performed for both conventional bridge converter and

considered bridgeless boost converter. The results of the conventional and new

converter considered for boosting can be compared. The results show that the gain of

bridgeless converter is much more than that of the conventional bridge converter.

I. Conventional Bridge Boost Converter

The boost converter Simulink model and simulation results are shown in Fig 6.1 and

Fig 6.2 .The operation is performed for 4 V ac, 100 Hz and output settles at 5 V dc.

The gate signals to the transistor are given using the 555 timer at 50 KHz and 60%

duty ratio. The boost converter employs a diode bridge rectifier, which converts input

ac to dc and then presents the boost circuit which undergoes a second stage i.e.

boosting the rectified dc to the level of load requirement.

II. Open Loop Operation

The Simulink model and simulation results of open loop operation of the bridgeless

boost converter employed are shown in Fig. 8. The output is settled at 3 Volts. But the

requirement of the load is 3.3 V, so irrespective of any variations in the input the

output voltage must be constant to feed the load.

Fig. 6 Simulation Model of Conventional Bridge

Converter

Fig.7 Simulation Results of Conventional Converter



Fig. 8 Simulation model of open loop operation

of bridgeless converter

Fig. 9 Open loop simulation results

We can observe from above, open loop simulation does not yield the expected

output as shown in the results in Fig. 9. Also, as input to the vibrational source vary

depending on the environment and surroundings, the output voltage will also vary

accordingly. In order to maintain constant voltage irrespective of any variations in the

input and other circuit parameters, a controlled operation is essential. Therefore,

closed loop operation is preferred which controls the output as per the requirement of

the load.

III. Closed Loop Operation

The Simulink model of closed loop operation of the bridgeless boost converter

employed is shown in Fig. 10 and simulation results are also presented. We can now

clearly observe that after reaching steady state the output voltage is constant and as

per requirement of the load. The pulses in the simulation circuit are generated by

comparing carrier wave with a reference wave, according to the required frequency

and duty ratio. The duty ratio automatically settles to the required in order to maintain

voltage constant.

The actual voltage is compared with the reference voltage (3.3V) and the error signal

is given to a PI controller

Then, a hysteresis control is given by using a saturation block which limits the PI

controller output (control voltage) to remain within certain limits.

The control voltage is again compared with a saw tooth waveform from the signal

generator so that gate pulses generated are being given to the boost and buck-boost

converter switches according to the detection of positive and negative peaks,

respectively.

With 200 Ω resistive load the converter is capable of tightly regulating output

voltage and delivering 52 mW to the load. Fig. 11 shows the waveforms of input

voltage, gate signals of both switches, as well as the input current. During the positive

input cycle, S1 is turned ON, while S2 is driven by the boost control scheme. When

the circuit operates in the negative input cycle, S2 is turned ON, while S1 is controlled

under the buck-boost conditioning strategy. The waveforms of inductor current are

depicted in Fig. 12. We can observe that from Fig. 6.8 the conduction of diodes alter

every half cycle according to the boost and buck boost operation.



Fig. 14 and Fig.15 demonstrate the boost and buck-boost operations

correspondingly. The gate pulses of the switches can be observed in accordance with

the input current. Closed-loop voltage control successfully stabilizes the duty cycle at

0.72 at the steady state. As seen from Fig. 16, the output voltage is regulated at 3.3-V

dc with approximately 0.2 V (i.e., 12%) voltage ripple. Closed-loop voltage control

successfully stabilizes the duty cycle at 0.78 at the steady state. The output voltage for

R =300 Ω is demonstrated in Fig. 17.

Fig. 10 Closed loop Simulink model of the

converter

Fig. 11Closed Loop Simulation Results (R=200

Ω)

Fig. 12Inductor Current of Closed Loop Operation Fig. 13. Diode Voltages and Currents



Fig. 14 Input Current and Buck Boost Gate Pulse

Vg1

Fig. 15 Input Current and Boost Gate Pulse Vg2

Fig. 16 Input and Output Voltages(R=200Ω) Fig. 17 Input and Output Voltages(R=300Ω)

5. CONCLUSION

A single stage ac–dc topology for low-voltage low-power energy harvesting

applications is proposed in this project. The topology uniquely combines a boost

converter and a buck-boost converter to condition the positive input cycles and

negative input cycles, respectively. Only one inductor and one filter capacitor are

required in this topology. The closed loop simulation has been carried out in power

electronic simulation software MATLAB. Output voltage is tightly regulated at 3.3 V

with 12% ripple through closed-loop voltage control. Using a low pass filter at the

output the waveform can be smoothened by reducing the ripple content. The

measured conversion efficiency is 66% at 52.5mW. In comparison to state-of-the-art

low-voltage bridgeless rectifiers; this study employs the minimum number of passive

energy storage components, and achieves the maximum conversion efficiency. The

future research will be focused on investigating and designing integrated three-phase

power electronic interfaces for electromagnetic energy harvesting.



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