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117 D. Govind, Nandkumar Wagh

International Journal of Electronics, Electrical and Computational System

IJEECS

ISSN 2348-117X

Volume 6, Issue 6

June 2017

Resonant DC – DC Buck - Boost Converter for the Battery Charger

and PV Applications

1D. Govind, 2 Nandkumar Wagh 1Assistant Professor, 2 Professor

Department of Electrical Engineering,

Vidya Pratishthan’s Kamalnayan Bajaj Institute of Engineering and Technology,

Baramati, Pune(M.S.)-India

Abstract—In power electronic switches, soft switching is

a possible way of reducing losses. Soft switching refers

to the operation of power electronic switches as zero-

voltage switches (ZVS). All the power electronic

switching devices undergoes zero-voltage switching

during turn-off. In the converter, the switches undergo

zero-capacitive turn-on losses unlike switches in other

soft-switched topologies. This soft-switching technique

can also be applied to other classical switched mode

power converters. The structure of the proposed

converter is simpler and cheaper than other resonant

power converters. In this paper, single-switch resonant

power converter offering the advantages of soft

switching, reduced switching losses, and increased

energy conversion efficiency for Photovoltaic

applications is presented. This circuit topology integrates

a single-switch resonant converter with zero-voltage-

switching. The operating principles and the steady-state

analyses of the proposed interleaved buck, boost buck-

boost converters are discussed and performance of grid

connected ZVS is verified with simulation results.

Keywords—ZVS, buck, boost, buck –boost.

This paper is divided in five sections. Section-I

presents the introduction of switching modes in

converters and the literature related to it.

Section-II presents all converter topologies.

Simulation results of the converter topologies are

depicted in Section-III.

Section-IV and V deals with the comparison of

converter topologies and the comparative analysis.

I. INTRODUCTION

DC-DC converters are finding more and more

use in portable applications such as cell phones,

laptops etc. In order to achieve higher power

density and high voltage profile, these converters

are usually require to operate at higher switching

frequencies with higher efficiencies[1]-[2].

When the switching frequencies continues to

increase , then in order to meet the future

requirements of power density and efficiency, the

Resonant DC-DC converters redraw people’s

attention. Resonant converters are good alternative

because of its soft-switching power transfer

characteristic. These converters can considerably

reduce the switching loss and obtain friendly EMI

characteristics [3]. Therefore we can operate the

converter at higher frequencies without sacrificing

the efficiency, so high efficiency and high power

density can be achieved simultaneously. In

Resonant converters because of smooth voltage and

current waveforms, noise and interference and

stress on switching devices are reduced and

parasitic circuit elements such as transformer

leakage inductance can be taken into account.

In the 1970’s, conventional PWM power

converters were operated in a switched mode

operation. Power switches have to cut off the load

current within the turn-on and turn-off times under

the hard switching conditions. Hard switching

refers to the stressful switching behavior of the

power electronic devices. The switching trajectory



IJEECS

ISSN 2348-117X

Volume 6, Issue 6

June 2017

of a hard-switched power device is shown in Fig.1.

During the turn-on and turn-off processes, the

power device has to withstand high voltage and

current simultaneously, resulting in high switching

losses and stress. Dissipative passive snubbers are

usually added to the power circuits so that the dv/dt

and di/dt of the power devices could be reduced,

and the switching loss and stress are diverted to the

passive snubber circuits [4].

However, the switching loss is proportional to

the switching frequency, thus it is required to limit

the maximum switching frequency of the power

converters. Typical converter switching frequency

was limited to a few tens of kilo-Hertz (typically 20

kHz to 50 kHz) in early 1980’s.

I

VOff

On

Soft-switching

Hard-switching

Safe Operating Area

snubbered

Fi

g.1 Typical switching trajectories of power

switches

The stray inductive and capacitance in the power

circuits and power devices still cause considerable

transient effects, which in turn give rise to

electromagnetic interference (EMI) problems [3].

Fig.2. shows ideal switching waveforms and typical

practical waveforms of the switch voltage. The

transient ringing effects are major causes of EMI.

In the 1980’s, lots of research efforts were diverted

towards the use of resonant converters. The concept

was to incorporate resonant tanks in the converters

to create oscillatory, usually sinusoidal voltage and

current waveforms so that zero voltage switching

(ZVS) or zero current switching (ZCS) conditions

can be created for the power switches. The

reduction of switching loss and the continual

improvement of power switches allow the

switching frequency of the resonant converters to

reach hundreds of kilo-Hertz (typically 100 kHz to

500 kHz). Consequently, sizes of elements can be

reduced and the power density of the converters is

increased. Various forms of resonant converters

have been proposed and developed.

Fig.2. Typical switching waveforms of (a) hard-

switched and (b) soft-switched devices

Resonance Technology

There are basically two types of soft-switching

techniques:

1. Zero Current Switching (ZCS)

2. Zero Voltage Switching (ZVS)

Either of this technique can greatly reduce and even

completely eliminates the switching losses in a

converter. High power level converters usually use

IGBT switches due to low conduction losses and

higher power capability, but IGBT is not as fast as

MOSFET [5]- [8] and its switching frequency

cannot be increased beyond 100 KHz even if softly

switched. On the contrary to Insulated Gate Bipolar

Junction Transistor (IGBT), Metal Oxide

Semiconductor Field Effect Transistor (MOSFET)

is resistive device. When it is turned on, the

conduction losses are higher as compared to IGBT

at higher power levels. However, MOSFET is a

faster device and is able to operate up to a few

MHz’s.

Fig.3. Shows current and voltage waveforms of

hard and resonant switching system with portion of

losses in both.

Fig.3. Current and voltage waveforms of hard and

resonant switching systems



IJEECS

ISSN 2348-117X

Volume 6, Issue 6

June 2017

ZVS converters have three resonant states: over

resonance (completed resonance), optimum

resonance (critical resonance) and quasi resonance

(sub resonance). Only the quasi resonance state has

two zero crossing points in a repeating period. A

resonant switch is a sub-circuit comprising a

semiconductor switch S and resonant elements, Lr

and Cr. The switch S can be implemented by a

unidirectional or bidirectional switch, which

determines the operation mode of the resonant

switch. [4].

II. ZERO VOLTAGE RESONANT SWITCH

(ZVS)

In zero voltage switching resonant converters,

the resonant capacitor provides a zero-voltage

condition for the switch to turn on and off [7]. A

quasi-resonant buck converter designed for half-

wave operation using a ZV resonant switch as

shown in Fig.4. In a ZV resonant switch, a

capacitor Cr is connected in parallel with the switch

S for achieving zero-voltage-switching (ZVS). If the

switch S is a unidirectional switch, the voltage

across the capacitor Cr can oscillate freely in both

positive and negative half-cycle. Thus, the resonant

switch can operate in full-wave mode. If a diode is

connected in anti-parallel with the unidirectional

switch, the resonant capacitor voltage is clamped by

the diode to zero during the negative half-cycle.

The resonant switch will then operate in half-wave

mode. The objective of a ZV switch is to use the

resonant circuit to shape the switch voltage

waveform during the off time in order to create a

zero-voltage condition for the switch to turn on [6].

Lr

S

(a)

Cr

Lr

CrS

(b)

Fig.4. Zero-voltage resonant switch (a) half wave

(b) full wave mode

Steady-State Analysis of Quasi-Resonant

Converters (QRC) to simplify the steady-state

analysis of the converters, some assumptions needs

to be made.

1. The filtering components Lo, Lin, Lf, and Co

are very large compared to the resonant

components Lr, Cr.

2. The output filter Lo-Co-R is treated as a

constant current source, Io.

3. The output filter Co-R is treated as constant

voltage source, Vo.

4. Switching devices and diodes are ideal.

5. Reactive circuit components are ideal.

A ZVS-QRC designed for half-wave operation is

illustrated with a buck type dc-dc converter; shown

in Fig.5.

Fig.5. Circuit diagram of a buck zero-voltage-

switching resonant converter

Fig.5, depicts the circuit structure of a buck

zero-voltage switching resonant converter. It differs

from a conventional buck PWM converter in that it

has an additional resonant tank that comprises a

resonant inductor Lr , a resonant capacitor Cr , and

a diode Dr . The inductor Lr is connected in series

to power switch Q to limit di/dt of the power

switch, and the capacitor Cr is installed as an

auxiliary energy transfer element. Lr and Cr

constitute a series resonant circuit, whose

oscillation is initiated by turning off the power. Dm

is a freewheeling diode. Capacitor Cf and inductor

Lf comprise a low pass filter, which not only filters

high-frequency ripple Signal, but also provides a

stable dc source for load. The freewheeling diode in

the ZVS converter is commutated under soft-

switching. This characteristic makes the ZVS

technique particularly appealing for high-frequency

conversion applications. Therefore, implementing

soft-switching for both the power switch and the

freewheeling diode in such a circuit is particularly

valuable. To simplify the analysis, the output filter



IJEECS

ISSN 2348-117X

Volume 6, Issue 6

June 2017

inductance is assumed to be sufficiently large to be

regarded as an ideal dc current source Io, during a

high-frequency resonant cycle. In one switching

cycle, the circuit operation can be divided into four

modes, whose associate equivalent circuits are

displayed in Fig.5. The parameters are defined as

follows. Characteristic impedance: ro

r

LZ

C ;

resonant angular frequency: o r rL C ; resonant

frequency: fr = ωo/2π; switching period: Ts

In a single switching cycle, the circuit operates

in the following four modes.

Mode I: Linear stage0 1( )t t t : Prior to t0, the

power switch Q is on, and conducts a drain current

that equals the output current Io, and the

freewheeling diode Dm is off. Fig. 6 depicts the

equivalent circuit. At t0, Q is turned off. The current

through the resonant inductor Lr does not change

instantaneously, and so the current is diverted

around the power switch through the resonant

capacitor Cr. The current of the resonant inductor

equals the output current Io and the capacitor

voltage vcr, which increases, as given by

0

1( )

t

ocr o

r r

IV t I dt t

C C (1)

Voltage across freewheeling diode Dm is

determined by

( ) ( ) ox in cr in

r

Iv t V v t V t

C (2)

Fig.6. Equivalent circuit of Model I

vx declines to zero at time t1 , when Dm is turned on

by soft switching. The constant output current

linearly increases the voltage across the resonant

capacitor, until the input voltage is reached

1

in r

o

V Ct

I (3)

Model I is completed when t = t1 , namely vcr (t1) =

Vin . The time interval TI in Model I is obtained

using (5). Moreover, Model II is initiated when vx

decreases to zero

in rI

o

V CT

I (4)

Mode II: Resonant stage1 2( )t t t ): After t1, the

freewheeling diode Dm becomes forward-biased,

and Cr and Lr resonate. The instantaneous voltage

across Cr and the resonant inductor current can be

evaluated, respectively as

1( ) cos ( )

rL o oi t I t t (5)

1

1

1( ) ( ) ( )

t

cr cr cr

r t

v t i t dt v tC

(6)

With initial condition vCr(t1) = Vin (7)

1

1

1( ) cos ( )

t

cr o o in

r t

v t I t t dt VC

(8)

Fig.7. Equivalent circuit of Model II

1( ) sin ( )cr o o o inv t Z I t t V (9)

The maximum ( )crv t that occurs at '

1t is

determined by

' 1

1 1

1(sin )

r

t t

(10)

The maximum value of crv is determined by

,Cr Peak in o ov V Z I (11)

The voltage across the freewheeling diode in Fig. 7

can be written as

( ) 0crv t



IJEECS

ISSN 2348-117X

Volume 6, Issue 6

June 2017

The freewheeling diode current wave shape follows

a cosine function during this interval, and equals Io

minus iLr (t). The resonant time is determined by

solving the resonant capacitor voltage equation

under the condition when the voltage is zero.

1

2 1

1[sin ( ) ]in

o o o

Vt t

Z I

(12)

This model is completed at t = t2 , when vCr(t2) = 0

and iLr (t2) = −Io . Moreover, the time interval

during Model II is determined using (13)

11[sin ( ) ]in

II

o o o

VT

Z I

` (13)

The above equation indicates that load current Io

is so large that Zo.Io > Vin. Otherwise, the voltage

of the power switch would not return to zero

naturally, and the power switch has to be turned on

at a nonzero voltage, causing turn-on losses. This

interval ends at t2, when vcr decreases to zero and

the anti-parallel diode Dr begins to conduct.

Mode III: Recovery Stage2 3( )t t t : After Dr is

turned on, the voltage across Cr is held at zero. The

turn-on signal of Q is applied, when the anti

parallel diode is conducting to achieve ZVS. During

this interval, the inductor current is expressed as

2

2

1( ) ( )

t

L in Lr

r t

i t V dt i tL

(14)

2 2 1( ) cos[ ( )]in

r

Vt t Io o t t

L

(15)

3 3 2 2 1( ) ( ) cos[ ( )]inLr o o

r

Vi t Io t t I t t

L

(16)

3 0 2 1 2( )[1 cos ( )]r o

in

L It t t t

V (17)

Fig.8. Equivalent circuit of Model III

The resonant inductor current iLr (t) is linearly

returned from its negative peak of minus Io to its

positive value of positive Io . Consequently, iLr (t)

increases linearly and iDm decreases linearly. This

model is completed at t = t3 when vCr(t3) = 0 and iLr

(t3) = Io. The commutation interval in this stage is

expressed by

0 2 1 2( )[1 cos ( )]r oIII

in

L IT t t t

V (18)

Notably, the voltage across the switch Q is zero,

when the power switch is turned on. It enables the

turn-on switching loss to be avoided and the total

efficiency of the converter to be increased

accordingly.

Mode IV: Freewheeling stage3 4( )t t t : When

iL (t) reaches Io at t3, the freewheeling diode Dm is

turned off, and the zero-voltage-switched converter

resembles a conventional square-wave power

processor. The charging current flows through

power switch Q and resonant inductor Lr.

Accordingly,

( )L oi t I and ( ) 0Crv t (19)

The power switch conducts Io as long as it is kept

on until t4. At t4, the power switch is turned off

again, beginning another switching cycle. The

duration of this mode is TIV expressed as

( )IV s I II IIIT T T T T (20)

Fig.9. Equivalent circuit of Model IV

The output voltage Vo is determined as

0

1Ts

BA x

s

V v dtT

1

10 3

1[ (1 )]

t Ts

in in

s t

tV dt V dt

T t

13[ ( )]

2

ins

s

V tT t

T (21)

13[1 ( )]

2BA in s

tV V f t (22)

The output voltage varies with the switching

frequency. Fig. 9 illustrates the equivalent circuit of



IJEECS

ISSN 2348-117X

Volume 6, Issue 6

June 2017

Model IV and Fig. 10 shows key steady-state

waveforms of the buck ZVS converter.

Voltage Gain: The condition ZoIo > Vin must hold

to ensure that the operation is under zero voltage

switching. 1o in

o r

I VC

,

or

in o

IC

V (23)

Similarly, because of the condition ZoIo >Vin must

hold such that o o r inI L V

Fig.10. Steady-state waveforms of the developed

ZVS resonant converter

inr

o o

VL

I (24)

Given Io and TS, TI, TII and TIII and the output

voltage Vo can be determined. However, the

voltage conversion ratio is normally best expressed

in terms of load resistance R and switching

frequency fs. Vo = RIo, so the energy stored in the

resonant inductor is 2 3

3 1

1 2

( )

t t

i in Lr in Lr in o s

t t

W V i dt V i dt V I T t t (25)

The energy released by the filter inductor to the

load is

o o o sW V I T (26)

2

1

t

inin

o O Ot

Io ViLrdt CrV

I Z

(27)

3 2 2

2 1

2

[1 cos ( )]

2

t

O O

int

LrI t tiLrdt

V

(28)

Let 2 1( )O t t

2 1 3 2( ) [ ( ) ( )]o s II III o sI T T T I T t t t t

(1 cos )[ ]o

o s

in o

I LrI T

V

(29)

Let the normalized load resistance

o

o i

Rr

Z W

(1 cos ) [ ]

2

r O r inin o

O in

L I C VV I Ts

V

(30)

o o o sW V I T

Wo equals Wi , when the converter power

dissipation is ignored.

1[ (1 cos ) ]

2

O O O r O r in

in o s in O

V I L C V

V I T V I

(31)

For a lossless system, in the steady state, these two

energies are equal. Hence, the voltage ratio is

expressed by (32) as

(1 cos )1 [ ]

2 2

fs X rX

fr r X

(32)

Where BA

in

VX

V and

2 1( )O t t denote the

voltage ratio and pulse width angle, respectively.

The relationship between input and output voltages

is a function of the pulse width angle, the

characteristic impedance of the resonant-switch

converter and the output load current. The variation

of modulation index (M) and switching frequency

fns as a function of duty ratio (D) is shown in fig.11.

Fig.11. Relationship between M and fns



IJEECS

ISSN 2348-117X

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June 2017

ZVS converters can be operated in full-wave

mode. The circuit schematic is shown in Fig.12. (a).

The circuit waveforms in steady state are shown in

Fig.12. (b). The operation is similar to half-wave

mode of operation, except that VCr can swing

between positive and negative voltages. The

relationships between M and at different r are

shown in Fig.12. (c).

Cf

Cr

Lr

Lf

Df

+

Vo

-

+

voi

-

ILr

+ vc -

Dr

Io

Fig.12. (a) Schematic diagram

Fig.12. (b) Circuit waveforms

Fig.12. (c) Relationship between M and fns

Comparing Fig.11 with Fig.12 (c), it can be seen

that M is load-insensitive in full-wave mode [7]-

[10]. This is a desirable feature. However, as the

series diode limits the direction of the switch

current, energy will be stored in the output

capacitance of the switch and will dissipate in the

switch during turn-on. Hence, the full-wave mode

has the problem of capacitive turn-on loss, and is

less practical in high frequency operation. In

practice, ZVS-QRCs are usually operated in half-

wave mode rather than full-wave mode.

III. SIMULATION RESULTS

Simulation of all six types of zero voltage switching

(ZVS) converters is performed using PSIM

software.

ZVS Buck Converter:

ZVS Resonant BUCK Converter with half wave

topology

The Simulation Model of ZVS Buck Converter

is shown in Fig.14a. By adding Lr,Cr to the normal

buck converter it will become as a resonant

converter. Consider the converter specifications as:

Specifications: Vi = 24V, f = 25kHZ, D = 70%, Lr

= 316µH, Cr = 22nF, Ro = 50Ω, Vo = 14V.

The resonant capacitor is placed across the switch.

This capacitor makes the switch turn on at zero

voltage position. By proper maintenance of firing

pulses switch turn on and off takes place at zero

voltage positions.

Fig.14b. shows that, whenever voltage across

switch is zero at that position ,switching (either ON

or OFF) takes place. The switching losses and EMI

decrease because in normal converter switching

takes place at nonzero value of voltage because of

this dv/dt increases and consequently EMI

increases. Where as ZVS switching takes place at

zero voltage position, so dv/dt decreases and EMI

decreases.

Fig.14a. Simulation Model of ZVS Buck Converter

with half wave topology



IJEECS

ISSN 2348-117X

Volume 6, Issue 6

June 2017

0.0

0.20

0.40

0.60

0.80

1.00

Firing_pulses

0.0

-10.00

10.00

20.00

30.00

40.00

50.00

60.00

voltage_across_switch

19.90 19.92 19.94 19.96 19.98 20.00

Time (ms)

0.0

-0.10

-0.20

-0.30

-0.40

0.10

0.20

0.30

Inductor_current

Fig.14b. Firing Pulses, Current through Switch and

Voltage across Resonant Capacitor vs Time

0.0

2.50

5.00

7.50

10.00

12.50

Output_voltage

0.0 5.00 10.00 15.00 20.00 25.00 30.00

Time (ms)

0.0

0.10

0.20

0.30

output_current

Fig.15 Output Voltage and Output Current vs Time

The output voltage dependency with respect to the

switching frequency is shown in Fig.16. This gives

the output voltage for different normalised loads

and switching frequency. If the load changes for

particular switching frequency voltage gain varies.

This is the disadvantage of this topology. To

overcome this full wave mode topology is

introduced.As the switching frequency increases

output voltage decreases.

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.90.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

Normalised frequency(fs/fo)

Vo

ltag

e G

ain

(M

)

Q = 0.1

Q = 0.2

Q = 0.5

Q = 0.8


ZVS Resonant BUCK Converter with full wave

topology

Simulation Model of ZVS Buck Converter in

full wave mode is shown in Fig.17. By adding anti

parallel diode across switch current through switch

goes to negative also. This can be observed from

Fig.18.

Specifications:

Vi = 24V, f = 25KHZ, D = 60%, Lr = 316µH, Cr =

22nF, Ro = 100Ω, Vo = 14.8V.

Fig.17. Simulation Model of ZVS Buck Converter

with full wave topology

Fig.18. show that whenever voltage across

switch is zero, at position, switching (either ON or

OFF) takes place. Fig.19. shows the output voltage

and output current.



IJEECS

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June 2017

0.0

0.20

0.40

0.60

0.80

1.00

firing_pulses

0.0

20.00

40.00

60.00

Voltage_across_switch

49.90 49.92 49.94 49.96 49.98 50.00

Time (ms)

0.0

-0.20

-0.40

0.20

0.40

Inductor_current

Fig.18. Firing Pulses, Current through Switch and


0.0

5.00

10.00

15.00

20.00

25.00

30.00

Output_voltage

0.0 0.02 0.04 0.06 0.08 0.10

Time (s)

0.0

0.20

0.40

output_current

Fig. 19. Output Voltage and Output Current vs

Time

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 10

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

Normalised frequency(fns)

Vo

lta

ge G

ain

(M

)

Q = 0.45

Q = 0.9

Q = 1.4

Q = 1.9


ZVS Boost Converter:

PSIM Simulation Model of ZVS Boost Converter in

half wave mode is shown in Fig.21.

Specifications:

Vi = 24V, fs= 25KHZ, D = 60%, Lr = 316µH, Cr =

22nF, Ro = 100Ω, Vo = 30V.

Fig.21. Simulation Model of ZVS Boost Converter

Fig.22. shows whenever voltage across switch is

zero, at position switching (either ON or OFF) take

place. Fig.23 shows the output voltage and output

current wave forms.

0.0

0.20

0.40

0.60

0.80

1.00

Firing_pulses

0.0

20.00

40.00

60.00


49985.00 49990.00 49995.00 50000.00

Time (us)

0.0

-2.00

-4.00

2.00

4.00

Inductor_current



0.0

5.00

10.00

15.00

20.00

25.00

30.00

35.00

Output_voltage

5.00 10.00 15.00 20.00

Time (ms)

0.0

0.25

0.50

0.75

1.00

1.25

1.50

1.75

Output_current

Fig.23. Output Voltage and Output Current vs

Time



IJEECS

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June 2017

The output voltage variation with switching

frequency is shown in Fig.24. This gives the output

voltage for different normalised loads(Q). As the

switching frequency increases output voltage

decreases. Voltage gain is greater than one because

its boost converter its output is always greater than

input so M >1 always.

0.3 0.4 0.5 0.6 0.7 0.8 0.9 10

0.5

1

1.5

2

2.5

3

3.5

4


Vo

lta

ge G

ain

(M

)

Q = 0.25

Q = 0.5

Q = 1

Q = 2


ZVS Buck – Boost Converter

Buck Mode Operation: The Simulation Model of

ZVS Buck-Boost Converter in half wave mode

shown in Fig.25. Specifications: Vi = 24V, f =

25kHZ, D = 45%, Lr = 316µH, Cr = 22nF, Ro =

100Ω,

Vo=12.5V.

Fig.25. Simulation Model of ZVS Buck-Boost Converter

0.0

0.20

0.40

0.60

0.80

1.00

Firing_pulses

0.0

-20.00

20.00

40.00

60.00

80.00


39990.00 39992.50 39995.00 39997.50 40000.00

Time (us)

0.0

-2.00

-4.00

-6.00

2.00

4.00

6.00

Inductor_current



Fig.26. shows whenever voltage across switch is

zero, at position switching (either ON or OFF) take

place in buck mode operation. Fig.27. shows the

output voltage and output current wave forms.

Fig.28. show that whenever voltage across switch is

zero, at position switching (either ON or OFF)

takes place in buck mode operation. Fig.29. shows

the output voltage and output current.

0.0

-2.50

-5.00

-7.50

-10.00

-12.50

-15.00

Output_voltage

0.0 5.00 10.00 15.00 20.00

Time (ms)

0.0

-1.00

-2.00

-3.00

output_current

Fig.27. Output Voltage and Output Current vs

Time

Boost Mode of Operation:

In this mode of operation output is more than

input in order to get this D should be maintained

greater than 0.5. In this simulation D is selected as

70%.

0.0

0.20

0.40

0.60

0.80

1.00

Firing_pulses

0.0

25.00

50.00

75.00

100.00

125.00


29985.00 29990.00 29995.00 30000.00

Time (us)

0.0

-5.00

-10.00

5.00

10.00

Inductor_current

Fig.28 Firing Pulses, Current through Switch and

Voltage across Resonant Capacitor vs Time.



IJEECS

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Volume 6, Issue 6

June 2017

0.0

-10.00

-20.00

-30.00

-40.00

Output_voltage

0.0 5.00 10.00 15.00 20.00

Time (ms)

0.0

-0.50

-1.00

-1.50

-2.00

-2.50

-3.00

-3.50

output_current

Fig.29. Output Voltage and Output Current vs Time

The output voltage dependency with switching

frequency for different loads are shown in Fig.30.

As the switching frequency increases output voltage

decreases in negative with negative slope . At fns

=0.45 boost operation starts for all Q > 0.05, if it is

less than 0.45 converter operates in buck converter.

0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1-2.5

-2

-1.5

-1

-0.5

0


Vo

ltag

e G

ain

(M

)

Q = 0.1

Q = 0.5

Q = 1

Q = 1.5


IV. COMPARISION OF ZVS CONVERTER

WITH NORMAL CONVERTER

ZVS resonant converters can considerably

reduce the switching loss by switching on the

device at zero voltage position, so the efficiency of

the converter increases compared to the normal

converter. The comparison of ZVS boost with

normal boost converter with different parameters is

shown in table-1.The power losses and efficiency of

ZVS and normal boost converters are calculated

and presented

Table – 1: Comparison of ZVS Boost Converter with Normal Boost Converter

Parameters Normal Boost Converter ZVS Boost Converter

R = 100Ω R = 150Ω R = 100Ω R = 150Ω

Input Volatge (Vin) 24 24 24 24

Input Current(Iin) 5.82 3.4 1.258 1.156

Input Power(Pin) 139.68 9.456 30.192 27.4

Output Voltage(Vo) 109 100 53.52 60

Output Current(Io) 1.09 0.65 0.535 0.4

Output Power(Po) 118.8 65 28.63 24

Efficiency 85 79.6 94.8 86.5



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June 2017

V. COMPARATIVE ANALYSIS OF ZVS RESONANT CONVERTERS:

The steady state performance of different converters, considering different parameters like rise time, peak

overshoot, efficiency and output voltage ripple is presented in table-2.

Table -2: Performance analysis of various ZVS converters.

Converter Type Vin Iin Vo Io Efficien

cy

Rise

Time

(msec)

Peak

Over

Shoot

Output

Voltage

Ripple(mv)

Buck 24 0.185 13.5 0.27 82 6 Zero 5

Buck Full wave

mode 24 0.202 14.1

0.28

1 81.7 2.5 25.2 2

Boost 24 1.08 34.8 0.7 92 2.3 42 10

Buck-boost 24 0.47 21.8 0.45 87 6 Zero 10

CONCLUSION

The converter topologies has been described and

the simulation results of all are presented. To

minimize the switching losses at the time of switch

on, zero voltage condition is provided by

introducing resonant circuit across switch.

Simulation of various ZVS Resonant DC-DC

converters are carried out using PSIM and output

voltage, current, voltage across resonant capacitor

are obtained. The variation of voltage gain with

switching frequency is plotted in MATLAB

environment. Simulation results and the calculated

switching loss and efficiency shows that, there is a

great improvement in efficiency of ZVS Resonant

converters compared to normal converter.

REFERENCES

[1] Kwang-Hwa Liu, Ramesh Oruganti and Fred

C.Y.Lee, “Quasi-Resonant converter topologies and

Characteristics”, IEEE Transactions on Power

Electronics, vol. PE-2, no.1, January 1987, pp .

[2] M. A. Jabbar, Ashwin M. Khambadkone and Guo

Chun, “Quasi-Resonant-Converters-Based High-

Efficiency Spindle Motor Drives for Magnetic Data

Storage” IEEE Transactions on Industrial

Electronics , vol .51, NO. 6, December 2004, pp

1338-1343.

[3] H. Chung, S. Y. R. Hui, and K. K. Tse, “|Reduction

of Power Converter EMI Emission Using Soft-

Switching Technique”, IEEE Transactions on

Electromagnetic Compatibility, vol. 40, no. 3,

August1998,pp. 282-287.

[4] Ivo Barbi, J.C. Bolacell, D.C. Martins, and F.B.

Libano,“Buck Quasi – Resonant Converter

Operating at Constant Frequency : Analysis, Design

and Experimentation”, IEEE Transactions on Power

Electronics, vol.5,no.3,1990, pp.873-880.

[5] Jung G. Cho and Gyu H. Cho, “Single-Cycle

ResonantConverters: A New Group of Quasi-

Resonant Converters Suitable for High-Performance

dc/dc and ac/ac Conversion Applications”, IEEE

Transaction on Industrial Electronics, vol. 38, no. 4,

August 1991, pp.260-267.

[6. Wojciech A. Tabisz, Pawel M. Gradzki, and

Fred.C.Y.Lee, “Zero-Voltage-S witched Quasi-

Resonant Buck and Flyback Converters-

Experimental Results at 10 MHz”, IEEE

Transactions on Power Electronics, vol. 1, no. 2,

April 1989, pp. 194-204.

[7] Kwang-Hwa Liu, and Fred C.Y.Lee, “Zero-Voltage

Switching Technique in DC/DC Converters”, IEEE

Transactions on Power Electronics, vol. 5, no. 3,

July 1990, pp. 194-204.

[8] Guichao Hua and Fred C. Lee, “Soft-Switching

Techniques in PWM Converters”, IEEE

Transactions on Industrial Electronics, vol. 42, no.

6, December 1990, pp. 595-603

[9] B.P. Divakar and D. Sutanto, “A Novel Converter

for Fuel-Cells Applications”, IEEE International

Conference on Power Electronics and Drives

Systems, June 2005, vol.2, pp.162-165.

[10] Ying-Chun Chuang and Yu-Lung Ke, “A Novel

High-Efficiency Battery Charger With a Buck Zero-

Voltage-Switching Resonant Converter”, IEEE

Transactions on Energy Conversion, vol. 22, no. 4,

December 2007, pp. 848-854.



IJEECS

ISSN 2348-117X

Volume 6, Issue 6

June 2017

[11] Robert C. N. Pilawa-Podgurski, Anthony D.

Sagneri, Juan M. Rivas, David I. Anderson, and

David J. Perreault, “Very-High-Frequency Resonant

Boost Converters” IEEE Transactions on Power

Electronics , vol. 24, no. 6, June 2009 , pp. 1654-

1664.

Govind.D. was born in nizamabad, India, on April

18, 1988. He received the B.Tech degree in

electrical and electronics engineering from

Jawaharlal Nehru Technical University, Hyderabad,

India, in 2008 and the M.Tech degree in electric

drives and power electronics at the Indian Institute

of Technology, Roorkee. His current research

interests include power electronics applications and

drives, multilevel inverter, power-quality (PQ),

solar photovoltaics and application.

Nandkumar Wagh was born on 20th December

1962. He obtained his U.G. and P.G. degree from

Government College of Engineering Amravati

(M.S.). He obtained the Ph D in Electrical

Engineering from Maulana Azad National Institute

of Technology, Bhopal (M.P.) His research interest

is specifically focused in artificial intelligence

applications to Electrical Power System and Power

Electronics and Drives. He has more than 25

national and international publications to his credit.

He is a member of professional societies such as

IETE, IE (I), ISTE, and IENG.

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