dissertation torsten mack

A MEMS-Based Reconfigurable RF Receiver

Front-End Utilizing Multi-Port Technology

A Thesis Submitted to the

Technical Faculty of the

University of Erlangen-Nuremberg

In Partial Fulfillment

of the Requirements for the Degree

DOKTOR-INGENIEUR

by

Torsten Mack

Erlangen 2005

Rekonfigurierbarer Hochfrequenz-Empfanger

mit Mikro-Elektromechanischen Systemen

auf Basis von Multi-Tor-Technologie

Der Technischen Fakultat der

Universitat Erlangen-Nurnberg

zur Erlangung des Grades

DOKTOR-INGENIEUR

vorgelegt von

Torsten Mack

Erlangen 2005

Als Dissertation genehmigt von

der Technischen Fakultat der

Friedrich-Alexander Universitat Erlangen-Nurnberg

Tag der Einreichung: 15. Juni 2005

Tag der Promotion: 11. August 2005

Dekan: Prof. Dr. Albrecht Winnacker

Berichterstatter: Prof. Dr.-Ing. Dr.-Ing. habil. Robert Weigel

Prof. Dr.-Ing. Franz X. Kartner (MIT)

Acknowledgements

This work was conducted at the DaimlerChrysler Research Facility in Ulm, Germany,Department of Vehicle Sensing and Communications Microwave, and at the Massa-chusetts Institute of Technology in Cambridge, MA, USA, between November 2001and March 2005.

I would like to thank Prof. Robert Weigel from the Friedrich-Alexander-UniversityErlangen-Nuremberg, Chair of Technical Electronics, and Prof. Franz Kartner fromMassachusetts Institute of Technology, Research Laboratory of Electronics for theirinvolvement and the supervision of this work.

I would especially like to thank Prof. Franz Kartner for embracing me as a member ofhis research group during my research period in the United States.

I would like to gratefully acknowledge the enthusiastic supervision of Dr. Johann-Friedrich Luy from DaimlerChrysler Research and Technology. His knowledge andexpertise helped guide me through this work.

I want to sincerely thank Dr. Bernd Schauwecker and Dr. Karl Strohm for numerousdiscussions and countless hours of technical support during the design phase and fab-rication of the MEMS.

I want to thank Dietrich Eisbrenner for his generosity in taking the extra time in theclean-room and assisting me during the fabrication process of the MEMS.

Many thanks to Dr. Thomas Muller for his inspiration in the field of SDR and forhelping the thesis take shape and progress forward.

Many thanks also to Thomas Eireiner and Dr. Konrad Bohm for their intellectualsupport and for the various fruitful discussions that were essential for the success ofthis work.

Furthermore, I want to thank Winfried Simon from IMST GmbH and Dr. Jan Mehnerfrom FEMWARE GmbH for supporting the development process of the MEMS withtheir simulations.

Thanks also to Francois Deborgies and Laurent Marchand from ESA for their supportand suggestions in the development of the MEMS.

Many thanks to all my interns and Master Thesis students, especially to AlexanderHonold.

Last but not least, I would like to thank my parents, Inge and Eugen, and my girlfriendAnnie Seapan. Without their encouragement and support, this work would not havebeen possible. Many thanks also to my sincere friends Micky, Andrea, and Annie formaking me dinner and proofreading this thesis.

Abstract

The aim of this work is the design and evaluation of a reconfigurable, universal, multi-band, multi-standard receiver front-end. This front-end is based on software definedradio (SDR) which leads to a significant reduction of hardware circuit complexity.For down conversion, the multi-port receiver principle has been chosen as it is a verypromising candidate to cope with the large frequency ranges needed for receiving mul-tiple standards.

The original multi-port (or six-port) theory applies only to alternative network analyz-ers. Since the late 1990s, the six-port principle has also been used for radio frequency(RF) communications receivers, but the principle of the frequency conversion processwas never thoroughly described. Therefore, an accurate mathematical description ofthe frequency conversion and demodulation processes in multi-port receivers needs tobecome established. With this detailed understanding of the multi-port theory, thesix-port receiver can then be evaluated and its performance can be compared to thatof conventional architectures.

To integrate present and future frequency bands from 1 GHz through 40 GHz intoa single receiver, the hardware of the multi-port interferometer itself needs to be re-configurable. Hardware reconfigurability in the analog front-end necessitates multi-ple routing structures, i.e. receive/ transmit (RX/TX) switches or front-end selectorswitches. It is obvious that these switches must have a very low loss so as not to degradethe signal-to-noise (SNR) ratio. On the other hand, RX/TX switches must supportvery high powers. As we will see, these two requirements, namely low loss and highpower, cannot be achieved sufficiently with conventional PIN diode switches. The newand upcoming micro-electromechanical systems (MEMS) technology offers an elegantway to accomplish these specifications. As low loss, high power switches are the keyelement for future multi-band multi-standard transceivers, much effort has been putin the design, simulation, fabrication, and evaluation of new suitable radio frequency(RF) MEMS switches – in particular, a single pole double throw (SPDT) switch. RFMEMS switches designed with this new technology can cover a frequency range fromDC to 40 GHz with an insertion loss below 1 dB and can handle several watts of power.

When the performance of the new MEMS switches is known, they can be applied andevaluated in the context of the six-port receiver. For an accurate evaluation of thereconfigurable six-port interferometer, S-parameter measurements need to be analyzedin respect to signal attenuation and phase relations. A detailed analysis of these pa-rameters will further improve the understanding of the multi-port principle, especiallythe requirements for covering a large frequency range.

In the context of communications the meaningful physical entity is the symbol er-ror rate. The multi-band, multi-standard six-port receiver front-end must meet therequirements of today’s communication standards. This implies a good noise per-formance which can be characterized by the symbol error rate of the demodulatedsymbols with respect to the bit energy over noise (Eb/N0) ratio.

Contents

1 Introduction 11.1 Motivation and State of the Art Technology . . . . . . . . . . . . . . . 11.2 Contribution and Outline . . . . . . . . . . . . . . . . . . . . . . . . . 2

2 Theoretical Background of Multi-Port Receivers 52.1 The Software Defined Radio Concept . . . . . . . . . . . . . . . . . . . 5

2.1.1 Introduction to Software Defined Radio . . . . . . . . . . . . . . 52.1.2 Software Defined Radio Architectures . . . . . . . . . . . . . . . 6

2.2 The Theory of Diode Detectors . . . . . . . . . . . . . . . . . . . . . . 82.2.1 Semiconductor Diode Circuit Model . . . . . . . . . . . . . . . . 92.2.2 Diode Detectors in Multi-Port Applications . . . . . . . . . . . 10

2.3 Simple Description of Multi-Port Receivers . . . . . . . . . . . . . . . . 122.4 Mathematical Description of the Multi-Port Receiver . . . . . . . . . . 13

2.4.1 Theory of Additive Mixing . . . . . . . . . . . . . . . . . . . . . 132.4.2 The Multi-Port Theory . . . . . . . . . . . . . . . . . . . . . . . 162.4.3 Calibration Method and IQ Calculation . . . . . . . . . . . . . . 17

2.5 Frequency Conversion in Multi-Port Receivers . . . . . . . . . . . . . . 18

3 RF MEMS For Signal Routing 253.1 Motivation and Introduction to RF MEMS . . . . . . . . . . . . . . . . 25

3.1.1 Typical Applications of RF MEMS . . . . . . . . . . . . . . . . 273.2 Overview of the RF MEMS Under Investigation . . . . . . . . . . . . . 283.3 Theoretical Background of the Simulations . . . . . . . . . . . . . . . . 30

3.3.1 Mechanical Domain Simulations . . . . . . . . . . . . . . . . . . 303.3.2 Electrostatic Domain Simulations . . . . . . . . . . . . . . . . . 323.3.3 Fluid Domain and Transient Response Simulations . . . . . . . 333.3.4 Electromagnetic Domain Simulations . . . . . . . . . . . . . . . 34

3.4 Design, Layout, and Simulation Results . . . . . . . . . . . . . . . . . . 353.4.1 Shunt Airbridge Switch . . . . . . . . . . . . . . . . . . . . . . . 363.4.2 Toggle Switch . . . . . . . . . . . . . . . . . . . . . . . . . . . . 403.4.3 Single Pole Double Throw (SPDT) Switch . . . . . . . . . . . . 463.4.4 RF Cross . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47

3.5 Process Flow and Fabrication . . . . . . . . . . . . . . . . . . . . . . . 493.6 SEM Micrographs and Experimental RF Measurement Results . . . . . 52

i

Contents

3.6.1 Experimental Results of the Shunt Airbridge Switch . . . . . . . 523.6.2 Experimental Results of the Toggle Switch . . . . . . . . . . . . 543.6.3 Experimental Results of the SPDT Switch . . . . . . . . . . . . 563.6.4 Experimental Results of the RF Cross . . . . . . . . . . . . . . 59

3.7 Additional Measurements and Reliability Results . . . . . . . . . . . . 603.7.1 Switching Time Measurement Results . . . . . . . . . . . . . . . 613.7.2 RF Power Measurement . . . . . . . . . . . . . . . . . . . . . . 633.7.3 Switch Cycle Measurement Results . . . . . . . . . . . . . . . . 653.7.4 DC Contact Resistance . . . . . . . . . . . . . . . . . . . . . . . 663.7.5 Temperature Dependency and Reliability . . . . . . . . . . . . . 67

4 The MEMS-Based Multi-Band Six-Port Circuit 694.1 Introduction to Passive RF Multi-Port Interferometers . . . . . . . . . 694.2 Options for the Multi-Port Architecture . . . . . . . . . . . . . . . . . 70

4.2.1 The N-Port Interferometer . . . . . . . . . . . . . . . . . . . . . 704.2.2 Five-Port and Six-Port Interferometers . . . . . . . . . . . . . . 71

4.3 Design and Analysis of a 1.5 GHz Six-Port Interferometer (SP1500) . . 734.3.1 Theoretical Background of the Electromagnetic Simulations . . 744.3.2 Substrate and Microstrip Lines . . . . . . . . . . . . . . . . . . 744.3.3 Design and Simulation Results of the Power Divider . . . . . . . 764.3.4 Design and Simulation Results of the Quadrature Hybrid . . . . 774.3.5 Simulation and Measurement Results of SP1500 . . . . . . . . . 79

4.4 Analysis of the 2 GHz to 25 GHz Six-Port Interferometer (SP40) . . . . 854.4.1 Guidelines for Broadband Power Divider Design . . . . . . . . . 864.4.2 Measurement Results of a Broadband Power Divider . . . . . . 864.4.3 Guidelines for Broadband Quadrature Hybrid Design . . . . . . 874.4.4 Measurement Results of a Broadband Quadrature Hybrid . . . . 874.4.5 Measurement Results of the SP40 . . . . . . . . . . . . . . . . . 88

4.5 The Reconfigurable MEMS-Based Multi-Band Front-End . . . . . . . . 914.5.1 Targeted Applications of RF MEMS in Receiver Front-Ends . . 914.5.2 The MEMS-Based Reconfigurable Six-Port Front-End . . . . . . 934.5.3 Results of the MEMS-Based Reconfigurable Six-Port Front-End 944.5.4 The RF MEMS SPDT Antenna Switch . . . . . . . . . . . . . . 101

5 Performance of the Multi-Band Six-Port Receiver 1035.1 Simulation Environment . . . . . . . . . . . . . . . . . . . . . . . . . . 103

5.1.1 Functional Principle of the Simulation Program CppSim . . . . 1035.1.2 Simulation Set Up . . . . . . . . . . . . . . . . . . . . . . . . . 1045.1.3 Simulation Run . . . . . . . . . . . . . . . . . . . . . . . . . . . 106

5.2 Simulation of the Six-Port Receiver . . . . . . . . . . . . . . . . . . . . 1095.2.1 Influence of Channel Noise . . . . . . . . . . . . . . . . . . . . . 1095.2.2 Frequency Offset and Phase Noise Dependency . . . . . . . . . . 112

5.3 Characterization of the Schottky Diode Detectors . . . . . . . . . . . . 1135.4 Measurement Results of the Multi-Band Six-port Receiver . . . . . . . 115

ii

Contents

5.4.1 Measurement Set Up and Run . . . . . . . . . . . . . . . . . . . 1155.4.2 General Dependency of RF and LO Power on Reception . . . . 1175.4.3 General Noise Behavior . . . . . . . . . . . . . . . . . . . . . . . 1195.4.4 General Phase Behavior . . . . . . . . . . . . . . . . . . . . . . 1195.4.5 Influences of In-Band Interferers . . . . . . . . . . . . . . . . . . 1215.4.6 Frequency Dependent SER Performance of Multi-Band Receiver 121

5.5 Alternative Applications of the Multi-Port Principle . . . . . . . . . . . 132

6 Conclusion and Perspectives 1336.1 Summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1336.2 Main Achievements and Outlook . . . . . . . . . . . . . . . . . . . . . 135

Table of Figures 137

Abbreviations:

IQ in-phase/ quadratureRF radio frequencyMEMS micro electro-mechanical systemsSDR software defined radioSPDT single pole double throwSEM scanning electron microscopeDC direct currentGHz gigahertzSER symbol error rateSP1500 six-port interferometer with a center frequency of 1.5 GHzSP20 six-port interferometer for frequencies from 2 GHz through 25 GHzQPSK quadrature phase shift keying

iii

Contents

iv

1 Introduction

1.1 Motivation and State of the Art Technology

Today, the communications industry faces the challenge of integrating an increasingnumber of radio communications systems. Especially in automotive applications, avast amount of different receivers ranging from below one megahertz (amplitude mod-ulation (AM) radio) to electronic toll collect systems at 5.7 GHz or even radar systemsat 79 GHz. Future communications systems with higher data rates are envisaged toemploy the 17 GHz, 24 GHz, or 60 GHz bands. The state-of-the-art technology is theuse of hardware that is limited to cover one frequency band only. This becomes verycomplex, ineffective, and expensive as modern transceivers need to handle more andmore standards in a single, compact device.

Therefore, it is of great interest to design a single transceiver that can handle multi-ple frequency bands, understand different standards, and is easily reconfigurable andupgradeable. These requirements can be met by shifting traditional hardware com-ponents into the digital domain. This type of radio is called software defined radio(SDR) [1]. With the advances in complementary metal-oxide semiconductor (CMOS)technology, analog-to-digital converters (ADCs), programmable digital signal proces-sors (DSP), and high speed data transfer, the SDR principle becomes feasible from thedigital point of view and much effort is undertaken to implement SDR radios. How-ever, great effort also needs to be applied to the analog front-end, meaning anythingfrom the antenna(s) to the ADCs. An effective and well rounded principle needs to bediscovered and developed for universal frequency conversion processes over a large fre-quency range that allows high freedom and is independent of the baseband modulation.

The motivation of this work is the design of a broadband receiver for a frequencyrange from 1 GHz to 40 GHz with reduced hardware requirements. Hardware archi-tectures for SDR applications that cover such a large frequency range have not yetbeen reported. To achieve this goal, different SDR hardware architectures [2][3] havebeen reviewed and the most promising, the multi-port principle, was applied [4]. Thereported multi-port architectures differ in the number of output ports: Five-port re-ceivers [5][6][7] and six-port receivers [8][9][10][11][12][13].

1

1 Introduction

The six-port receiver under investigation in this work uses a local oscillator (LO) andbroadband matched power detectors for frequency conversion. The SDR principle isemployed for demodulation. To cover the aforementioned frequency range, the six-port architecture needs the ability to route signals to different antennas and to switchbetween different interferometer circuits. This feature was implemented with radiofrequency (RF) switches that consist of micro electro-mechanical systems (MEMS)[14]. As the realization of such new type of switches requires an enormous amountof effort, the RF MEMS switching structures were developed in a joint project withthe European Space Agency (ESA) called “Microwave Electrostatic Micro-MachinedDevices For On-Board ApplicationS” (MEMOS).

1.2 Contribution and Outline

This work is organized as a step by step approach from the theoretical background ofmulti-port receivers to the final performance evaluation of the designed reconfigurable,multi-band, multi-standard six-port receiver platform. The highlight lies in the devel-opment of the low loss, high power RF MEMS switches to be demonstrated in Chap.3 and their application in the six-port receiver context. To name a few, new scientificcontributions of the performed MEMS research are:

• Stable fabrication processes for new types of MEMS: an ohmic contact switch,an SPDT switch, and an RF cross. All devices can be fabricated on standard Siwafers with standard Si processes (CMOS compatibility).

• The increased handling and decreased insertion loss from the capacitive mem-brane switch, the toggle switch, and the SPDT switches make RF MEMS switchessuperior in their performance when compared to state of the art positive/ intrin-sic/ negative (PIN) diode switches.

• The MEMS RF cross offers low attenuation when signal lines need to lead overeach other.

• Detailed reliability and performance analysis of the toggle and capacitive shuntairbridge switch covering RF power, total number of switch cycles, direct current(DC) contact resistance, and temperature dependency.

Besides, there are several new achievements that are related to multi-port receivers.The most prominent are:

• A comprehensive and better to understand theoretical description that appliesthe concept of additive mixing to multi-port receivers and gives insight into thefrequency conversion processes

2

1.2 Contribution and Outline

• A six-port front-end that covers a very large bandwidth

• A better description of phase relations that further improve the understandingof multi-port theory

• Symbol error rate (SER) measurements at different signal-to-noise (SNR) ratios(more precisely Eb/N0) over a frequency range from 1.075 GHz to 40 GHz (atselected frequencies). SER measurements covering such a large frequency rangein a single receiver have not been published before.

The scientific content of this work starts off in the next chapter, “Theoretical Back-ground of Multi-Port Receivers” (Chap. 2), with a short introduction to SDRs in thecontext of multi-port receivers. From this, follows a description of the semiconductordiode detector as it is identified as the key component of multi-port receivers and theplace where the frequency conversion process takes place. A thorough elaboration andapplication of the diode theory is of great importance in understanding the functionalprinciple of multi-port receivers. The theory focuses on the multi-port principle in thecontext of communications receivers which is somehow detached from the original de-scription of six-port reflectometers found in the early work of Cohn [15], Engen [4][16],and Hoer [17][18]. This is necessary because in comparison to Engen’s work, we facea dynamic system which has different properties and requirements than quasi-staticsix-port reflectometers. This new approach to multi-port theory allows deep insightinto the functional principle and sheds light onto the “black box,” an expression forthe six-port interferometer found in a scientific article [19].

Operating the six-port receiver over a large frequency range requires a low loss, highpower switch in the analog front-end. For this reason, Chap. 3, “RF MEMS Switchesfor Signal Routing,” shows the possibility to realize such routing structures with thenew upcoming MEMS technology. The chapter begins with an introduction to RFMEMS with their typical applications and state-of-the-art technological possibilities.An overview illuminates the principle of the single pole double throw (SPDT) switch;namely that it is composed of two different RF MEMS switches, a capacitive shuntairbridge switch and an ohmic contact switch called a “toggle” switch [20]. In ad-dition, an RF cross is designed. To understand the design issues for all RF MEMSswitches, mechanical, electrostatic, and electromagnetic simulations are performed andwill be explained before presenting the simulation results achieved from the differentstructures. As the MEMS technology involves moving parts, it is highly sophisticatedand the design requires feedback from the advanced fabrication processes. Severalredesigns were necessary to achieve the measurement and reliability results that arepresented. Scattering (S) parameters and power handling are the main parametersthat were improved. Additional simulation and measurement results give insight intoswitching time, reliability, contact resistance, and temperature dependency.

3

1 Introduction

Once the results from the new RF MEMS switches are available, they can be used forthe design of the reconfigurable multi-band six-port circuit that is investigated in de-tail in Chap. 4, “The MEMS Based Multi-Band Six-port Circuit.” The chapter beginswith an introduction to various RF interferometers and design guidelines. To evaluatethe MEMS in the multi-port context, a 1.5 GHz six-port interferometer was designed,simulated, fabricated, measured, and thoroughly evaluated. It is organized in a wayto understand the function of all single components, three quadrature hybrids and apower divider, as well as their arrangement in the final interferometer. As multi-portreceivers are based on the superposition of electromagnetic waves under different phaseangles, this chapter aims to convey these phase relations in an accuracy and depththat has never been described before. Commercial components are used for the secondbroadband six-port interferometer, that covers a notable frequency range from 2 GHzto 25 GHz (experimental receiver measurements are performed up to 40 GHz)[21]. Fi-nally, S-parameter simulation results are presented for both six-port circuits includingthe RF MEMS switches and RF MEMS cross that serve to route the signal to the dif-ferent interferometers. For this purpose, real S-parameter measurement results fromthe RF MEMS components, as well as from the six-port interferometers, are includedin the system simulation.

In Chap. 5, “Performance of the Reconfigurable Six-port Receiver,” the multi-standardreceiver is evaluated over the designated frequency range from 1 GHz to 40 GHz. Thisevaluation is done by analyzing the SER in demodulated quadrature phase shift keying(QPSK) signals that is caused by noise, specified by the Eb/N0 ratio. Schottky diodedetectors (characterized in the beginning of the chapter) are applied to detect thepower of the signal. Simulations are performed that include influences of channelnoise and the phase noise of the LO on reception. For the time domain simulation ofthe SER, a new and rapid C++ based computer simulation program called CppSim(developed by M. Perrott from Massachusetts Institute of Technology (MIT)[22]) isused. Measurements at selected frequencies between 1 GHz and 40 GHz are performedand compared with their simulation results and theory.

Eventually, the work ends with a conclusion and outlook giving a short summary ofthe achieved results and discussing the maturation of this new technology.

4

2 Theoretical Background ofMulti-Port Receivers

This chapter will focus on the theoretical background and functional principle of themulti-port receiver. The beginning of the chapter will discuss various SDR archi-tectures and indicate why the six-port receiver is chosen to be the most promisingcandidate for a multi-band, multi-standard SDR platform for the frequency rangefrom 1 GHz through 40 GHz. The main focus in the multi-port theory is on the diodedetectors and the principle of additive mixing. This is essential in understanding thefrequency conversion processes which is the basis of multi-port theory.

2.1 The Software Defined Radio Concept

2.1.1 Introduction to Software Defined Radio

Radio communication is based on modifying the amplitude, the phase, and/or thefrequency of a carrier wave. In today’s radio transceivers, the frequency conversionfrom RF to baseband and vice versa is done with a hardware modulator that is solelydesigned for a single standard. During use, the radio receiver selects only the specificsignal type and filters out all other signals. Such transceivers are not flexible; systemsbecome complex and expensive when hardware for several standards has to be includedin one device. The solution is the SDR concept. SDRs distinguish themselves fromtheir conventional counterparts by shifting typical hardware related tasks into the dig-ital domain [1]. In other words, an SDR is a radio with a generic hardware based onanalog circuitry under a flexible software architecture. This includes reconfigurableradios, software-based radios, and SDRs based on digital signal processing technol-ogy. Although these concepts have been around for awhile, their practical designshave only now become feasible due to advances in many technologies such as: CMOS,silicon germanium (SiGe), MEMS, field-programmable gate arrays (FPGA), powerfuland cost-effective programmable DSPs, high-performance ADCs, and ultra-fast datatransfer interfaces [23].

SDRs have several advantages over today’s hard-wired radios. The most obvious ad-vantage is the flexibility under a multi-standard environment. SDRs can be repro-

5

2 Theoretical Background of Multi-Port Receivers

grammed and reconfigured on the fly to adopt to the different standards even indifferent frequency bands. Furthermore, SDRs can also be configured to handle mul-tiple communications protocols. An agile SDR can handle various popular standardsand protocols from a single design implementation such as: 802.11a, 802.11b, GlobalSystem for Mobile Communications (GSM), Code Division Multiple Access (CDMA),analog (FM) and Digital Audio and Video Broadcast (DAB, DVB), satellite radio(Satellite Digital Audio Radio Services – SDARS), Global Positioning System (GPS),and, in the future, the European Global Satellite Navigation System (GALILEO). Infact, some of these standards (i.e. frequency modulation (FM) radio and GSM) usesometimes slightly different frequencies and channel spacings in different countries.This becomes challenging and expensive for manufacturers to handle all the possiblevariants. In a global market and with increasing internationalization, there is a de-mand for inexpensive flexible hardware. For the automotive industry, it would be ofgreat ease to have one single platform that can be configured by software dependingupon which country it will be shipped to.

2.1.2 Software Defined Radio Architectures

With a generic hardware base, radio functionalities such as signal generation, waveformmodulation and demodulation, baseband digital signal processing, use of intermediatefrequency (IF), and use of multiple link-layer protocols that have traditionally beenhandled by hardware can now be defined by software. The focus of this work, however,is to provide the hardware front-end for multi-band, multi-frequency SDRs. The hard-ware modules for an SDR include antennas, analog front-ends (which convert the RFsignal into IF signals or baseband signals), and digital baseband modules. DifferentSDR receiver front-end architectures will be discussed in the following.

Fig. 2.1 shows idealized block diagrams of possible RF front-ends. The ideal SDRreceiver hardware includes direct sampling of the signal at the antenna, even withoutthe use of any filter as shown in Fig. 2.1(a). Filtering, frequency conversion to base-band, and demodulation are then performed with appropriate software algorithms inthe digital domain. The concept is called direct sampling. In this case, according tothe Nyquist Theorem, the sampling rate of the ADC ωs must be greater than or equalto twice the RF frequency ωRF . With the latest achievements in ADC technology, thiscan be done with great success for FM radio where the carrier frequency is around 100MHz [24].

However, for larger carrier frequencies at 1 GHz or 2 GHz, today’s mass market ADCsare not fast enough to meet the Nyquist criteria. In this case, one possible architec-ture is based on bandpass limited sub-sampling as illustrated in Fig. 2.1(b). Insteadof sampling the entire spectrum, only the modulated RF bandpass signal is sampled.This is an interesting approach that requires very sharp, high Q, RF filters and a

6

2.1 The Software Defined Radio Concept

RFADC

ωs

ωRF

(a) Direct digital(ωs ≥ 2 · ωRF )

RFADC

ωs

ωRF

BPF

(b) Bandpass-limited subsampling(ωs > 2 · bandwidth)

RF

ωs

ADC

LO

BPF

ωRF

(c) Heterodyne: IF/ low IF(ωLO < ωRF )

RFADC

LPF

ωRF

ωsLO

(d) Homodyne: zero IF/ direct conversion(ωLO = ωRF )

Fig. 2.1: Different RF front-ends for SDR platforms.

sampling rate greater than twice the filter bandwidth BBPF . For frequencies above 10GHz, such filters could become available in MEMS technology. Promising results canbe found in the article describing MEMS filters [14]. Another issue with sub-samplingarchitectures is the disadvantageous mapping of noise onto the baseband signal whosemagnitude depends on the RF to sampling rate ratio.

The remaining two architectures are heterodyne (Fig. 2.1(c)) and homodyne (Fig.2.1(d)). Sampling of the intermediate frequency (IF) in a heterodyne architecture isadvantageous, as the IF filters can be designed with sharp flanks. Filter banks fordifferent standards can be implemented at the IF level. Channel filtering, as wellas down conversion to baseband, can easily be done in the digital domain. The IFdepends on the ratio of the RF and LO frequency. As the LO frequency gets closerto RF, the IF becomes smaller and the signal can be filtered with a low pass filter(LPF). For IF greater than zero, the architecture is called low IF. If the RF equals theLO frequency, the architecture is called homodyne (this is also called zero-IF or directconversion). It should be mentioned that the ideal direct conversion requires hardwarebased complex down conversion as found in today’s in-phase quadrature (IQ) receivers(see Fig. 2.8). This will be discussed in detail in Chap. 2.5.

The more RF blocks introduced into the front-end, the less flexible it is. For thearchitectures in Fig. 2.1(b) to (d), low loss signal routing elements are needed toswitch to different antennas, bandpass filters, and down converter architectures. In areal architecture, high Q RF filters and low noise amplifiers (LNA) are additionallyneeded before down conversion. In this work, the main focus lies in the design of abroadband frequency down-converter that can handle a large frequency range. This

7


frequency down conversion process can be realized in multi-port technology coveringa much higher bandwidth and much higher maximum frequency when compared tostate-of-the art mixer concepts (i.e. multiplicative Gilbert cell mixers) [21][25][26][108].

2.2 The Theory of Diode Detectors

In a multi-port receiver, the extremely large bandwidth and the very high maximumfrequency is achieved by using semiconductor diodes in a power detector configuration.Cut-off frequencies of Schottky diodes can reach terahertz frequencies [27]. In fact,the diode based power detector can be regarded as a diode mixer with an LPF atthe output. Fig. 2.2(a) shows the process of conventional multiplicative mixing wheretransistors are used for frequency conversion (i.e. in a Gilbert cell) with subsequentlow pass filtering.

LO

RF

(a) Multiplicative mixing

RF

LO

Φi

(b) Additive mixing (multi-port down con-version)

Fig. 2.2: Multiplicative versus additive mixing. The additive mixing process is thebasis of the multi-port theory.

The new concept that is used in multi-port receivers is based on additive mixing asshown in Fig. 2.2(b). The issues with additive mixing will be thoroughly discussed inChap. 2.4.1. In a multi-port receiver (a six-port receiver is shown in Fig. 2.5), eachof the four output ports is connected to a power detector. The signal addition itselftakes place in the interferometer circuit which superposes the RF and LO signals underdifferent phase angles Φi. It is possible to make such interferometers very broadband[21]. In comparison to conventional IQ mixers with two output ports, the signals atthe output ports of the power detectors must be processed further. The basebandIQ signals are calculated from the four power readings after a calibration process.Before the multi-port theory is covered in detail, the functional principle of the diodedetectors (as they are one of the key elements in the architectures of the receiver) willbe described in the following.

8


2.2.1 Semiconductor Diode Circuit Model

In order to understand the diode power detector, it is necessary to have a circuitmodel for the Schottky diode. This model has to be valid for both the large signal,nonlinear case as well as for the small signal case [28]. Since the Schottky diode islargely immune to minority carrier effects, the junction capacitance

C(V ) =Cj0

√

1 − V/φbi

(2.1)

and diode current

Id(V ) = Is

(eαV − 1

), (2.2)

where α = e0V/nkBT , change almost instantaneously with junction voltage V (Cj0 isthe junction capacitance at zero bias, φbi is the built-in potential from Schottky con-tact, Is is the reverse saturation current, e0 is the charge of an electron, n is the diodeideality factor, kB is the Boltzmann constant (1.37×10−23 J/K), T is the absolutetemperature in Kelvin. Is is typically between 10−6 and 10−15 A, and at T=290 K,α ≈ 28 mV). Therefore, the DC expression for these quantities are valid to very highfrequencies in the hundreds of GHz. In the large signal diode model, it is assumedthat the capacitance and current are functions of the junction voltage alone. This isvalid up to at least 250 GHz [28].

C(V)

RS

gd(V)

Id(V)

V

(a) Simple equivalent circuit forSchottky diode

20

15

10

5

0

−5−0.2 −0.1 0 0.1 0.2 0.3 0.4 0.5 0.6

curr

ent

I d(V

) [m

A]

voltage V [V]

(i) (ii) (iii)

(iv)

(b) DC characteristics of Schottky diode withtypical regions (i) through (iv)

Fig. 2.3: Equivalent circuit of Schottky diode and its DC characteristics

A circuit model for the Schottky diode is shown in Fig. 2.3(a). It consists of a voltage-variable resistance (or conductance gd(V )) and capacitance for the junction C(V ), anda fixed series resistance RS. Other elements that describe packaging are not included.It is important to differentiate between large signal and small signal diode parame-ters. For large signal circuits such as the six-port receiver with only a large LO signal

9


applied, the junction current and capacitance have a non-linear dependence on theinstantaneous junction voltage (C(V ) and Id(V ) are given in Eq. 2.1 and 2.2, whereV represents the instantaneous voltage of the time varying voltage).

In the small signal case, it is assumed that the magnitude of the AC junction voltageis very small. There may also be a larger junction voltage component, such as a DCbias or a larger LO signal. If the alternating current (AC) voltage is small enough, thecapacitance and junction resistance may be treated as linear quantities, although theymay vary as the larger applied voltage is varied. The small signal junction conductancegd(V ) is the derivative of the diode current

gd(V ) =dId

dV= αIse

αV = α(Id(V ) + Is), (2.3)

which result in the junction conductance being proportional to its current. Is is verysmall compared to I(V ) for forward conduction and can be ignored.

A linear plot of the DC characteristics of a Schottky diode is shown in Fig. 2.3(b)(different regions are marked). For very small applied voltages V , the current responseof the diode can be approximated with its quadratic term from a Taylor expansionof Eqn. 2.2 (between (i) and (ii)). Higher order terms appear between (ii) and (iii).For higher voltages, the limit for the current is given by the series resistance RS

which leads to a linear dependence beyond point (iv). For power detection, the inputvoltage should stay in the quadratic region of the diode, where the output current Id

is proportional to the square of the input voltage and, therefore, proportional to theinput power:

Id ∝ V 2 ∝ Pin. (2.4)

2.2.2 Diode Detectors in Multi-Port Applications

The properties of a semiconductor diode (described in Chap. 2.2) are well suited formulti-port applications. What is needed is the quadratic relationship between RFinput power and baseband output voltages. Fig. 2.4(a) shows a simple power detectoras it is used in multi-port applications.

For broadband RF matching, the input impedance Z0 should equal the line impedance,which is 50 Ω in most applications. The input power generates an AC voltage Vd acrossthe diode. This AC voltage generates the diode current which is low pass filtered atthe output by a capacitance CLP . The load at the output is in the order of MΩ. Thedetector output voltage VRL is the voltage across the load resistance. In the picture ofa power detector, a single sine wave signal generates a DC offset voltage at the output

10


Vin

Vd

CLP RLZ0

(a) Simple power detector

100

10−2

10−4

10−6

20 0−20−40−60

de

tecto

r o

utp

ut

vo

lta

ge

[V

]

input power [dBm]

(i)

(ii)

(b) Characteristics of a simple power detector

Fig. 2.4: Characteristic response to RF power of a simple power detector.

port which, in the quadratic region of the diode, is proportional to the input power.Fig. 2.4(b) shows the characteristic input power to detector output power relation-ship. For most semiconductor diodes, the quadratic region goes up to approximately-20 dBm in a 50 Ω environment generating an output voltage in the order of mV [29].

For even larger input powers, the diode does not operate any longer in its quadraticregion and power is lost to higher order terms (transition region between (i) and (ii)).These higher order terms do not contribute to the DC offset and are filtered by CLP .Due to this, the detector output voltage does not increase linearly with input power.For even larger input voltages Vin, the diode operates in rectification mode where thedetector output voltage VRL is proportional to the amplitude of the RF input signal√

Pin (in other words, proportional to the square root of the input power):

VRL ∝ Vin ∝√

Pin. (2.5)

However, this model of the working principle of the diode detector in a multi-port ap-plication is not very accurate. In fact, in multi-port applications, there is not only onesingle sine wave signal to be detected, but the sum of the RF and LO signal. There-fore, the operation mode of the diode detectors in multi-port applications is rather amixing of two signals on a non-linearity. This non-linearity is given by the quadraticregion of the diode. The superposition, or the addition of the two signals RF andLO, is accomplished by the interferometer circuit. The mathematical background forthis additive mixing that takes place in multi-port receivers is thoroughly described inChap. 2.4.1

11


2.3 Simple Description of Multi-Port Receivers

Before proceeding to the mathematical background of the additive mixing and multi-port theory, the functional principle of the multi-port receiver will be explained. Wewill see that the diode detector output voltages, when operating the diode in itsquadratic region, have a linear dependence on the RF signal amplitude. In this case,the power of the LO signal has to be constant.

The six-port interferometer as shown in Fig. 2.5(a) superposes the RF and LO signalsunder different phase angles. Now, consider only the output voltage at port 5. Inthis case, the superposition of LO and RF signal is given by j/2 · (LO + RF ) (thesignal amplitudes are halved and the signals have a phase difference of 0). If theamplitudes of the RF and LO signals are equal, this superposition of the two wavesleads to twice the amplitude. Now, when sweeping the RF phase from 0 to 360,the detector output voltage produces one full circle sine function as plotted in Fig.2.5(b). With an initial phase offset that is given by the interferometer circuit, theoutput voltages at the other ports start with a different initial voltage but show theexact same behavior (sine function with a different phase offset). Using simple linearrelations, these detector output voltages can be used to calculate the amplitude andphase of any RF signal. How this can be done is described in the following Chapterin conjunction with a much more detailed mathematical approach of the functionalprinciple of multi-port receivers.

PD90o

90o

90o

RFjLO

22+

3

4

RF (2)

LO (1)

6

5

LOjRF

22+)(

2RFLO

j +

)(2

1RFLO −

PD

PD

PD

(a) Six-port interferometer circuit

1

0.8

0.6

0.4

0.2

0

360 270 180 90 0

dete

cto

r outp

ut

voltage [

V]

phase difference between LO and RF signal [degrees]

port 3port 4port 5port 6

(b) Output voltages of the detector

Fig. 2.5: Functional principle of the multi-port receiver (six-port circuit shown).When sweeping the RF phase from 0to 360, the output ports show thevoltage waveforms given in (b).

12

2.4 Mathematical Description of the Multi-Port Receiver

2.4 Mathematical Description of the Multi-Port

Receiver

2.4.1 Theory of Additive Mixing

The small signal diode mixer theory accurately describes the behavior of the basebandoutput voltages of the diode detectors depending on the magnitude of the LO and RFsignals. In the following, the small signal diode mixer theory from [30] is adapted tosuite its application in a multi-port environment. As we have seen earlier in Eqn. 2.2,the I-V characteristics of a diode can be written as

Id(V ) = Is(eαV − 1). (2.6)

This I-V response is plotted in Fig. 2.3(b). Now consider the total diode voltage toconsist of a small AC signal

V (t) = V0 + v(t) (2.7)

where V0 is the DC bias voltage and Id(V0) = I0 is the DC bias current (this DC biascan also result from a large AC signal). If we assume that v(t) represents only smalldeflections around a constant bias term, the expression for the total current at thispoint can be represented by a Taylor series as a function of the applied AC signalvoltage

Id(V ) = I0 + Gdv(t) +1

2G′

dv2(t) + . . . , (2.8)

where

Gd =dId(V )

dV

∣∣∣∣V =V0

= αIseαV0 = α(I0 + Is) (2.9)

is the dynamic conductance and

G′

d = αGd = α2IseαV0 = α2(I0 + Is) (2.10)

is the derivative of the dynamic conductance. The Taylor series in Eqn. 2.8 is thesmall signal approximation for a diode. The first two terms are of little interestas no frequency conversion occurs through them. The third term (containing v2(t))represents the square law response of the diode and is responsible for the dominantfrequency conversion terms. In the following, typical LO and RF signals (as used inmulti-port receivers), are applied to Eqn. 2.8 to derive the multi-port mixer theory.

Fig. 2.6 shows the process of additive mixing that is found in each arm of the multi-portreceiver. The superposition, or addition, of the LO input signal

vLO(t) = VLO cos(ωLOt + ϕLO) (2.11)

13


RF

LO [ ]2

LPFvLO(t)

vRF(t)

iIF(t)

Fig. 2.6: Theory of additive mixing: the RF and LO signals are added and thensquared. This leads to additional terms in the baseband.

and the modulated RF input signal

vRF (t) = vBP (t) (2.12)

= RevBB(t)ejωRF t

= |vBB(t)| cos(ωRF t + ϕRF + ϕBB(t))

= I(t) cos(ωRF t + ϕRF ) − Q(t) sin(ωRF t + ϕRF )

takes place in the interferometer circuit, where vBP (t) is the complex bandpass signal,vBB(t) is the complex baseband signal, VLO is the amplitude of the LO signal, ϕRF isthe initial phase of the RF signal, and ϕBB is the modulated phase (containing phaseinformation from the in-phase component I(t) and quadrature component Q(t)). Fornow, let us consider a modulated RF signal without additional phase shifts from theinterferometer as indicated in Eqn. 2.12 (the interferometer circuit will be discussedlater in more detail). This sum of the two signals

v(t) = vLO(t) + vRF (t) (2.13)

is then applied to the diode equation (Eqn. 2.8). Considering all terms up to v2(t) thisresults in a diode current

id(t) = I0 + Gd(vLO(t) + vRF (t))︸︷︷︸

(i)

+αGd

2(vLO(t) + vRF (t))2

︸︷︷︸

(ii)

. (2.14)

Now suppose I0 = 0 (no DC bias) and i(t) is low pass filtered at the output. Therefore,Gd = Is and the term (i) in Eqn. 2.14 is zero. The remaining term (ii),

id(t) =k

2(vLO(t) + vRF (t))2 (2.15)

=k

2v2

LO(t)︸︷︷︸

(i)

+k

2v2

RF (t)︸︷︷︸

(ii)

+ kvLO(t)vRF (t)︸︷︷︸

(ii)

,

where k = αIs, is now further evaluated to find the baseband signals. Putting the fullforms of Eqn. 2.11 and Eqn. 2.12 into Eqn. 2.15 with subsequent low pass filteringleads to:

14


(i):

LP

k

2v2

LO(t)

= LP

k

2V 2

LO cos2(ωLOt + ϕLO)

(2.16)

= LP

k

2V 2

LO · 1

2(1 + cos(2(ωLOt + ϕLO))

=k

4V 2

LO

(ii):

LP

k

2v2

RF (t)

= LP

k

2V 2

LO(I(t) cos(ωRF t + ϕRF )

−Q(t) sin(ωRF t + ϕRF ))2

=k

4

(I2(t) + Q2(t)

)

=k

4|vBB(t)|2

(2.17)

(iii):

LP kvLO(t)vRF (t) = LP

kVLO cos(ωLOt + ϕLO)(I(t) cos(ωRF t + ϕRF )

−Q(t) sin(ωRF t + ϕRF ))2

=k

2VLO (I(t) cos(∆ω + ∆ϕ) + Q(t) sin(∆ω + ∆ϕ))

(2.18)

where ∆ϕ = ϕLO − ϕRF and ∆ω = ωLO − ωRF . It is found from Eqn. 2.18 that for∆ω = 0 and ∆ϕ = 0 the quadrature component Q(t) = 0, while for ∆ω = 0 and∆ϕ = 90 the in-phase component I(t) = 0. However, the resulting expression for theentire baseband signal for additive mixing also contains the terms from Eqn. 2.16 and2.17 and can be written in the form:

iIF (t) =k

4V 2

LO +k

4

(I2(t) + Q2(t)

)

+k

2VLO (I(t) cos(∆ω + ∆ϕ) + Q(t) sin(∆ω + ∆ϕ)) .

(2.19)

This detector output current leads to a voltage across the load resistance RL as shownin Fig. 2.4 (referred to as the detector output voltage). The result states that for a con-stant LO signal, there is a linear dependence between the detector output voltage andI(t), Q(t), and I2(t)+Q2(t). Treating I2(t)+Q2(t) as a third unknown, at least threeindependent voltages at different phase shifts ∆ϕ are needed to linearly solve for thetwo unknown baseband signals I(t) and Q(t). This is in agreement with the mismatchthat is found in experimental measurement results of a four-port diode based receiverwith two output ports using QPSK calibration and 64QAM (quadrature amplitudemodulation) modulation [33].

15


2.4.2 The Multi-Port Theory

The multi-port theory includes the phase shifts from the interferometer circuit. Thisprocess is depicted in an abstract manner in Fig. 2.7

RF

LO

Φi

[ ]2

LPF

vRF(t)

vLO(t) gBB,i(t)

Fig. 2.7: Theory of additive mixing in multi-port receivers: in the interferometercircuit, the phase of the RF signal is shifted by an angle Φi.

Similar to the simple additive mixing process from Fig. 2.6, the LO and the phaseshifted RF signals are added, squared, and finally low pass filtered. With Eqn. 2.19,the baseband signal at the output of the low pass filter can be written as

gBB,i(t) = LP(vLO(t) + vRF,i(t))

2

=k

4

(

V 2LO + I2(t) + Q2(t)

)

+k

2VLO (I(t) cos(Θ(t) + Φi) + Q(t) sin(Θ(t) + Φi)) ,

(2.20)

where Θ(t) = ∆ω+∆ϕ. As mentioned earlier, this equation states a linear dependencebetween the detector output signals gBB,i(t) and the complex baseband signals y(t).Therefore, the multi-port equations can be written in the linear form

y(t) =

n∑

i=1

ci · gBB,i(t)

=k

4·

n∑

i=1

ci

(

V 2LO + I2(t) + Q2(t)

)

+kVLO

2·

n∑

i=1

ci (I(t) cos(Θ(t) + Φi) + Q(t) sin(Θ(t) + Φi)) ,

(2.21)

where ci = ai + j · bi are the constant complex calibration coefficients or, in terms ofthe in-phase and quadrature component, simply as:

I(t) =n∑

i=1

ai · gBB,i(t) (2.22)

Q(t) =

n∑

i=1

bi · gBB,i(t)

16


In matrix notation, the general expression for the multi-port receiver is:

y(t) =

c1

c2...cn

T

k

2VLO cos(Θ(t) + Φ1) −k

2VLO cos(Θ(t) + Φ1)

k

4k

2VLO cos(Θ(t) + Φ2) −k

2VLO cos(Θ(t) + Φ2)

k

4...

......

k

2VLO cos(Θ(t) + Φn) −k

2VLO cos(Θ(t) + Φn)

k

4

·

I(t)Q(t)

...I2(t) + Q2(t)

+k

4V 2

LO

11...1

(2.23)

2.4.3 Calibration Method and IQ Calculation

In an optimum six-port interferometer circuit, the phase shifts between the LO andRF signals at the four output ports are: Φi ∈ 0, 90, 180, 270. With a constantRF phase, Θ(t) = 0 at the input of the six-port receiver; this is true if no frequencyoffset and no modulation occurs. The six-port equations found from Eqn. 2.21 are:

y(t) =k

4·

4∑

i=1

ci

(

V 2LO + I2(t) + Q2(t)

)

︸︷︷︸

(i)

+kVLO

2·

4∑

i=1

ci (I(t) cos(Φi) + Q(t) sin(Φi))

︸︷︷︸

(ii)

.

(2.24)

The first term (i) can be eliminate by the requirement

4∑

i=1

ci ≡ 0. (2.25)

Putting the six-port phases Φi into the second term (ii) we obtain:

y(t) =k

2· VLO · (c1I(t) − c2Q(t) − c3I(t) + c4Q(t)) (2.26)

=k

2· VLO · ((c1 − c3)I(t) + (c4 − c2)Q(t) (2.27)

This system of linear equations can be solved for known signals. For best calibrationresults, the sent IQ signals should be equally distributed in the IQ space. In the case

17


of six-port calibration, the appropriate IQ signals for calibration are 90 apart: (1,1),(-1,1), (-1,-1), and (1,-1). For five-port calibration, the sent symbols are 120 apart:(0,

√2), (-1,-1), and (1,-1). Solving the corresponding systems of linear equations for

the unknown calibration coefficients, we obtain for six-port calibration:

ci =1

kVLO

[1 −j −1 j

](2.28)

and for five-port calibration with c4 = 0:

ci =1

kVLO

[

1 ej 23π ej 4

3π]. (2.29)

Once the calibration coefficients are known, the baseband IQ signals can be calculatedat each sampling instant from the voltage readings gBB,i of the power detectors withEqn. 2.22.

It has been shown how the calibration coefficients can be theoretically derived. How-ever, in a real multi-port application, the calibration is done by sending a sequence ofknown and suitable IQ values, storing the resulting voltages from the detector outputs,and solving Eqn. 2.22 for the unknow calibration coefficients. This is the experimentalmethod that is applied in Chap. 5.

The multi-port receiver can easily be calibrated by using its linear relations betweendetector output voltages and amplitude of the modulated RF input signal (see Eqn.2.19). More calibration methods that trace back to six-port reflectometer calibrationcan be found in the literature [34][35][36]. An interesting approach is the use of S-parameter measurements for calibration [37].

2.5 Frequency Conversion in Multi-Port Receivers

The theoretical results found in Chap. 2.4 are now used to depict the frequency con-version processes in multi-port receivers. To demonstrate this in a clear manner, themathematical formulation that includes convolution and Fourier transforms of complexsignals is avoided and the results are graphically explained in the frequency domain.A similar idea can also be found in the literature [31].

Complex and Real Frequency Conversion

Frequency conversion to an RF carrier is necessary in order to transmit the data (witha relatively low modulation frequency) over the air interface. At the receiver, this RF

18


signal needs to be converted to the baseband to retrieve the baseband data.

To illustrate the frequency conversion processes, it is important to differentiate betweencomplex and real down conversion. In general, the modulated RF signal

sRF (t) = SRF (t) sin(ωRF t + ϕ(t)) (2.30)

= I(t) cos(ωRF t) − Q(t) sin(ωRF t)

contains complex information and requires complex down conversion to maintain thecomplex baseband signals. During down conversion, the Fourier spectrum SRF (ω)of the RF signal sRF (t) is shifted by ωLO which results in a new spectrum SRF (ω −ωLO). From a mathematical point of view, this transformation can be achieved bymultiplication of the RF signal with the complex signal ejωLOt. However, the complexfrequency conversion process is not possible with only one mixer unit. Instead, onemust realize this complex mixing by two separate down conversion paths, one for thereal and the other for the imaginary part.

sRF(t)

sin(ωLOt)

cos(ωLOt)

exp(jωLOt)

LPF

LPF

(a) Complex

cos(ωLOt)

LPF

sRF(t)

(b) Real

Fig. 2.8: Principle of complex and real frequency conversion.

Fig. 2.8(a) shows the basic receiver principle that performs this complex down conver-sion. The RF signal sRF (t) is multiplied in one arm by cos(ωLOt), and in the other arm,by its 90 phase shifted counterpart sin(ωLOt). If 90 between the two LO signals areachieved, no IQ mismatch occurs – only a circular movement of the IQ constellationaround the origin due to an initial phase offset is possible. In fact, each arm carriesout a real down conversion as depicted in Fig. 2.8(b). A single real down conversioncannot obtain the entire complex baseband information.

The processes for real and complex down conversion in the frequency domain aredepicted in Fig. 2.9. It is indicated that in case of a complex down conversion, the

19


+ωLO

+ωRF−ωRF 0

0 +ωRF +2ωRF−ωRF−2ωRF

LPF

−ωLO

(a) Complex down conversion

+ωLO

+ωRF−ωRF 0

0 +ωRF +2ωRF−ωRF−2ωRF

−ωLO

LPF

(b) Real down conversion

Fig. 2.9: Spectral properties of complex and real frequency conversion. Real downconversion to baseband leads to an overlap and information loss.

convolution of the RF frequency ωRF with the LO frequency ωLO shifts the RF signalinto only one direction (Fig. 2.9(a)). When expanding the cosine function with

cos(ωLOt) = 1/2 ·(ejωLOt + e−jωLOt

), (2.31)

it can be seen that a real down conversion (as depicted in Fig. 2.9(b)) produces twofrequency shifts of the original spectrum SRF (ω): a positive and a negative shift, whichleads to an overlap in the baseband. From this baseband signal, it is not possible toretrieve the full complex signal and, therefore, information is lost. The resulting signalsBB,n(t) is a mixture of the I and Q component:

sBB,m(t) = LP(sRF (t) · cos(ωLOt + ϕLO,1)) (2.32)

= LP(I(t) cos(ωRF t) cos(ωLOt + ϕLO,1) − Q(t) sin(ωRF t) cos(ωLOt + ϕLO,1))

=1

2(I(t) cos(ωRF t) + Q(t) sin(ωRF t))

Therefore, a second path with another measurement is needed in order to separate thetwo components. This is usually done with another real down conversion using the90 phase shifted version of Eqn. 2.31: cos(ωLOt + ϕLO,2). The general expression forcomplex down conversion using multiplicative mixers can be written in the form:

[sBB,m(t)sBB,n(t)

]

=1

2

[cos(ϕLO,1) sin(ϕLO,1)cos(ϕLO,2) sin(ϕLO,2)

] [I(t)Q(t)

]

(2.33)

The in-phase and quadrature components can be calculated from the two measure-ments if the phase matrix is nonsingular. It is found from Eqn. 2.33 that the require-ment |ϕ1 − ϕ2| = 90 is not necessary to obtain I(t) and Q(t) as long as the phasematrix can be inverted.

20


In the case of a real RF bandpass signals, the two sidebands in Fig. 2.9(b) are sym-metric and this scheme of direct conversion can still be applied successfully if thephase between RF and LO is constant (phase synchronous, coherent). An additionalrequirement is that the phase is chosen carefully so that the two sidebands do notdestructively overlap.

+ωLO

+ωRF−ωRF 0

ωRF+ωLO−ωRF−ωLO ωRF−ωLO−ωRF+ωLO

−ωLO

BPF

(a) Complex down conversion

+ωLO

+ωRF−ωRF 0

ωRF+ωLO−ωRF−ωLO ωRF−ωLO−ωRF+ωLO

−ωLO

BPF

(b) Real down conversion

Fig. 2.10: For a sufficiently large frequency difference between LO and RF there isno overlap in the IF.

One solution to overcome the overlap problem is to down convert the RF signal toan IF signal. This is depicted in Fig. 2.10. In this case, it is not required to usecomplex down conversion as the spectra from the negative and positive frequencies(that are convoluted to IF frequency) do not overlap. The IF signal still contains theentire complex information. However, to demodulate this information it is necessaryto further down convert the IF signal. This can be done very elegantly in the digitaldomain after AD conversion of the IF signal. An appropriate receiver architecture isdepicted in Fig. 2.11

Frequency Conversion by Additive Mixing

The issue with additive mixing is that it is not a simple mathematical multiplicationof the two signals, RF and LO, but the baseband spectrum also contains other partsthat originate from Eq. 2.15 [32]. The baseband spectrum after a direct conversionadditive mixing process is depicted in Fig. 2.12.

It can be seen how the content from Eqn. 2.16, Eqn. 2.17, and Eqn. 2.18 are mappedinto the baseband: the desired complex baseband signals I(t) and Q(t) are influ-enced by the baseband interferers I2(t) + Q2(t) and a component from the LO signals2

LO(t). Depending on the type of modulation, I2(t) + Q2(t) is not always constant

21


ADC

I(t)

Q(t)

ωRF + ωLO−ωRF−ωLO ωRF−ωLO−ωRF+ωLO

cos(ωLOt)

cos(ω2t)

sin(ω2t)ωLO≠ωRF

ωRF

ωRF−ωLO+ ω20ωRF−ωLO+ ω2

BPF

LPF

LPF

BPF

Fig. 2.11: Principle of IF sampling. After a real down conversion to IF, the com-plex down conversion from IF to baseband can be achieved in the digitaldomain.

−ωRF +ωLO0

s2LO(t)

I(t),Q(t)

I2(t)+Q2(t)

Fig. 2.12: Baseband spectrum after direct frequency conversion as found in themulti-port receiver.

(√

I2(t) + Q2(t) is the amplitude of the baseband signal and is constant for QPSKsignaling). The broadening of its spectrum comes from real self-mixing of the I- andQ-component as indicated in Fig. 2.12. In order to remove this baseband interferer,I2(t) + Q2(t), a multi-port with additive mixers needs an additional port (altogether,three output ports).

The additive mixing process in multi-port receivers, which depends on the frequencydifference between the LO and RF signal, is depicted in Fig. 2.13. Overlapping spectralead to information loss that can be recovered with an additional arm (detector). Inthe case of Fig. 2.13(a), the IF spectra do not overlap and no information is lost withonly one output arm. An example where this principle can be used is a single Schottkydiode for down conversion of extremely high carrier frequencies to IF. The complexbaseband data can then be retrieved with a subsequent conventional IF to basebandstage (analog or digital). Fig. 2.13(b) and (c) again show the disadvantage of additivemixers in working with low- or zero-IF that requires an additional arm and digital

22


+ωLO

+ωRF−ωRF 0

−ωLO

ωRF − ωLO−ωRF+ωLO 0

s2LO(t)

I2(t)+Q2(t)

BPF

(a) Additive mixing with large IF

+ωLO

+ωRF−ωRF 0

−ωLO

ωRF−ωLO−ωRF+ωLO

s2LO(t)BPF

I2(t)+Q2(t)(ii)

(b) Additive mixing with low IF

+ωLO

+ωRF−ωRF 0

−ωLO

s2LO(t)

I2(t)+Q2(t)

0

(i)

(ii)

LPF

(c) Additive mixing with zero IF

Fig. 2.13: Spectrum of the additive mixing process in multi-port receivers for differ-ent LO and RF frequencies.

baseband algorithm to remove the interferer (i) and (ii).

23


24

3 RF MEMS For Signal Routing

Various RF MEMS are feasible today. Low loss routing structures are key elementsand their availability makes the design of inexpensive multi-band front-ends possible.Therefore, new types of RF MEMS elements have been designed for their special ap-plication as low loss RF routing elements in the receiving path and as receive/transmit(RX/TX) switches that can handle the high transmitter powers. We will see that theconcept of RF MEMS switches offers advantages compared to their diode based semi-conductor counterparts. The new structures are a shunt airbridge switch, an ohmiccontact switch (called “toggle” switch [38]), an SPDT switch [20][14], and an RFcross[40]. To gain a deeper insight into these new structures, their design, fabrication,simulation, and measurement results will be shown in detail in this chapter.

3.1 Motivation and Introduction to RF MEMS

The design of microsystems (especially at higher frequencies and higher powers) of-ten requires the integration of mechanical elements on an electronic circuit [41][42][43].Monolithic Microwave Integrated Circuits (MMICs) with MEMS elements on the sameSi chip have many advantages over the conventional method that use complex multi-chip packaging schemes. One advantage is the cost effective batch fabrication of alarge amount of systems on a chip. At the same time, this reduces the size, weightand complexity leading to better reliability. In addition, system performance can beincreased by several orders of magnitude because less electrical interconnect parasiticsare involved.

In today’s mass market communications transceivers, semiconductor pin diode switchesare used. They are either based on GaAs, AlGaAs, or Si. An advantage is their fasterswitching time which is in the order of nanoseconds. The disadvantages, however, aretheir greater insertion loss, their smaller power handling capability, their non-linearity,and their power consumption. RF MEMS switches promise superb power handling athigh frequencies, yet they do not suffer from non-linearities and they consume practi-cally no power. In addition, a main advantage is their compatibility and integrabilitywith Si batch fabrication. However, RF MEMS switches are mechanical elements andare limited in their switching time (usually in the order of microseconds) and needadvanced packaging solutions [44]. To learn about the performance of commercially

25


available SPDT pin diode switches, some products of main manufacturers will nowbe discussed. M/A-COM offers an AlGaAs pin diode switch for frequencies up to 50GHz with an extremely low insertion loss of only 0.7 dB at 50 GHz. The disadvantageis a similar insertion loss at lower frequencies, whereas RF MEMS switches have abarely perceptible insertion loss. In addition, maximum power handling is limited to23 dBm. HITTITE offers a GaAs pin diode switch for frequencies up to 20 GHz withan insertion loss of 1.7 dB. Switches with even higher insertion losses are availablefrom MITEQ (IL: 2.8 dB at 18 GHz), American Microwave Corporation (IL: 2.1 dBat 20 GHz), Sierra Microwave Technology (IL: 3.2 dB at 26.5 GHz), and TOKIMEC(IL: 3 dB at 18 GHz).

With the introduction of batch fabrication and advances in micro-machining tools,MEMS switches have overcome some major shortcomings in manufacturing – bothin cost reduction and in integration compatibility with electronics [45]. Since then,various papers have been published on RF MEMS switches. In general, they can bedifferentiated by their actuation mechanism and their structural designs. Actuationmechanisms include electromagnetic [46]-[49], magnetostatic [50], electrostatic [51],thermal-electric [52], and piezo-electric [53] actuation. The structural designs includerotating transmission line [54], surface micro-machined cantilever [55]-[64], multiple-supported or membrane based designs [55][57][65]-[70], bulk micro-machined or waferbonded designs [71]-[73], diamond cantilever and contact [74], polysilicon switch [75],mercury micro-drop contact [76][77], and bistable micro-relays [78][83]. All of thesestructures use vertical contacting. Besides, lateral contact switches have also beenstudied [80][83]. Though superior in their dynamic behavior to many of the verti-cal contacting switches, the contact mechanism of the lateral switches lags behinddue to the roughness in etched side-surfaces and contact materials [80]. All of thesedifferent designs have their advantages and disadvantages. The trade off is stronglyinfluenced by requirements of the targeted application. Today, cantilevered and multi-ple supported surface micro-machined electrostatic switches are by far the most widelystudied devices. Their design aims to optimize the parameters specified in Chap. 3.2.

All electrostatic switches have similar electromechanical behaviors. When a bias volt-age is applied between the contacts, charges distribute in such a way that an elec-trostatic force occurs between them. This force is independent of the polarity andit bends the cantilever, or membrane, down creating an opposing tensile force. At acertain threshold, the tensile force can no longer balance the electrostatic force andthe cantilever abruptly falls to the opposing electrode. When the magnitude of thevoltage is reduced, the cantilever jumps back into its initial position, but typically ata much lower voltage than the actuation voltage. This leads to a hysteresis character-istic which is typical for all electrostatically actuated MEMS switches. However, theirmain advantage is the zero static current in the actuation path that leads to extremelylow power consumption. Furthermore, their size and design makes them very suitablefor integration onto an Si MMIC chip.

26

3.1 Motivation and Introduction to RF MEMS

3.1.1 Typical Applications of RF MEMS

Until today, the major applications of RF MEMS switches included automatic testequipment [73], RF communications, radar systems as well as more general applica-tions such as automotive [2][81] and switching power supply. Electrical switches are anessential part of all communications devices and are widely used. They are currentlybased on solid state semiconductor devices. Mass market communications equipmentranges from below one megahertz (AM radio) to wireless local area networks (LAN)at 2.4 GHz (IEEE 802.11b and 802.11g) and frequency bands in the 5 GHz range(802.11a). However, to fulfill the demands for higher data rates in wireless systems,even higher frequency bands are considered. Interesting frequencies are located at 17GHz, and in the Industrial, Scientific and Medical (ISM) bands at 24 GHz and 60 GHz.

For some analog high power broadcasting applications - like AM, FM, and TV - therequirements on the switches might be less stringent and solid state devices can beused. However, for low power digital communications, a broadband low loss switch isof utmost importance when designing a multi-band, multi-standard transceiver archi-tecture with a frequency coverage from 1 MHz to 40 GHz and higher. In the front-endof digital receivers, the RF signal path must have only very low loss for not to degradethe SNR. Future transceivers that utilize extremely high carrier frequencies need to bedownward compatible to lower frequency standards in order to guarantee basic Qualityof Service (QoS) in cases where the air link is not as good. Therefore, broadband lowloss and high power devices for signal routing need to become available [82].

Besides the targeted application of the MEMS devices under investigation here (namelysignal routing and switching), there are several other applications where these switchescan be used. The applications cover the following fields:

• impedance matching networks

• filter tuning

• variable gain amplifier

• attenuator

• phase shifter and phase shifting networks

• variable capacitors (replacement for varactor diodes)

• capacitor banks

• time delay networks

• phased array antennas

• electrically configurable antennas

• redundant switching networks

27


3.2 Overview of the RF MEMS Under Investigation

In this section, the basic building elements for switching matrices are introduced.These elements include an RF cross and two basic switch types: ohmic and capaci-tive. The inherent design of a capacitive shunt airbridge switch makes it ideal for usetowards higher frequencies. The other switch type, an ohmic contact switch in serialconnection (called a “toggle” switch), shows better performance at lower frequencies.Both switches are combined in an SPDT switch to achieve higher isolation in thebandwidth from DC to 40 GHz and up [14]. The design of the toggle switch is suchthat it can be easily extended to make an SPDT switch. A special RF cross has beendesigned for large switching matrices where the RF lines cross each other.

To support operating frequencies ranging from the upper megahertz range to 50 GHz,MEMS switching matrices are advantageous when compared to transistor or diodeswitches. As mentioned earlier, MEMS switches distinguish themselves from theirsemiconductor counterpart by having extremely low insertion loss and attenuationover a large frequency range covering mm- and sub-mm-waves while capable of han-dling high RF power. Low losses in the RF front-end are of utmost importance sinceany loss strongly decreases the SNR.

Con, Coff

signal

ground

1 2

(a) Shunt airbridgeswitch

signal

ground

1 2

(b) Toggleswitch

ground

1

2

34

(c) RF cross

signal

ground

ground

C

C

21

3

(d) SPDTswitch

Fig. 3.1: Electrical models of the MEMS switches and the RF cross showing thefunctional principle.

Based on the two different switch types, the capacitive shunt airbridge switch (Fig.3.1(a)) and the toggle switch (Fig. 3.1(b)), higher order switches are used for routingthe signal to different ports. These are the SPDT switch in Fig. 3.1(d) and the a four

28

3.2 Overview of the RF MEMS Under Investigation

port switch [39]. For applications where the coplanar wave guides (CWG) cross eachother, an RF cross (which is basically a low loss airbridge as shown in Fig. 3.1(c)) wasdesigned.

The design of MEMS switches and RF cross are optimized for the following criteriabut show excellent performance beyond the values specified:

• frequency of operation: 1 GHz to 30 GHz

• insertion loss: <0.4 dB

• isolation: >50 dB

• return loss: >20 dB

• actuation voltage: <50 V

• power handling (to be maximized)

• operating ambient: -25C to +75C

• switching time (to be minimized)

• power consumption (to be minimized)

Concerning the multi-band RF front-end which is under investigation, high powerhandling capability is needed for the RX/TX switches, the antenna diplexers, andantenna selector switches.

toggle shunt airbridge

pull-in voltage applied not applied applied not appliedmembrane state activated relaxed activated relaxedelectrical state on off off onswitch state closed open open closedinsertion loss |S12| - - |S12|return loss |S11| - - |S11|isolation - |S12| |S12| -reflection - |S11| |S11| -

Tab. 3.1: Overview of switch states and S-parameter nomenclature. Notice thatwhen the membrane is “relaxed,” the shunt airbridge switch is “closed”or the electrical state is “on.” Loss is referred to the closed state of theswitch; reflection and isolation to the open state.

Tab. 3.1 gives an overview of the notation for the different switch states of the toggleand shunt airbridge switch that will be used in this work. Insertion and return loss arereferred to in the closed state of the switch, while reflection and isolation are referredto in the open state of the switch. The shunt airbridge switch is electrically closedwhen no voltage is applied, i.e. the membrane is relaxed. The toggle switch with its

29


ohmic contact is closed when the membrane is activated. The dB-values for insertionloss, return loss, isolation, and reflection that appear in the graphs have negative dBvalues which correspond directly to the S-parameter results. In the text, however,positive dB values are given.

3.3 Theoretical Background of the Simulations

The general objectives of the electromechanical design are low driving voltage, minimalswitching and release time, high reliability, and a long lifetime of the flexible parts.The latter mainly depends on the stress level inside the material. In the case of RFMEMS, additional electromagnetic simulations are essential when optimizing for scat-tering parameters. In the following, the theoretical background for these simulationtools are given.

The mechanical, electrostatic, fluid domain and transient response simulations havebeen carried in cooperation with J. Mehner from FEMWARE GmBH. The electro-magnetic domain simulations have been carried in cooperation with W. Simon fromIMST GmBH.

3.3.1 Mechanical Domain Simulations

Most micro-mechanical systems can be described by the theory of beams and plates[84]. The theory of beams covers flexible components that only vary in one direction.The theory of plates and shells is much more complex and needs to be applied in caseswhere a significant deformation in two directions takes place. Fortunately, the designof the toggle and the shunt airbridge switch mainly varies in the horizontal directionand the beam theory is sufficiently accurate for the analytical design.

The chosen analytical solutions methods are based on energy terms. Because energyis independent of the true physical background, it is considered key for coupled do-main simulations which are needed for electromechanical actuators. The Principleof Castigliano [85] is most practical for over-determined mechanical systems such asarrangements with multiple clamps. This is the case for the toggle switch. The methodis the inverse approach of the Lagrangian method, which assumes that the final defor-mation can be represented by a series of trail or shape functions. The latter method ismore suitable for the shunt airbridge switch. The Principle of Castigliano starts with agiven load situation for the electrostatic pressure, reacting spring forces, and moments[86]. It then computes the total mechanical strain energy W m which is stored in theflexible component. An iterative solution procedure is necessary since electrostatic

30


loads depend strongly on the deflection results. For bending beams we obtain

W m =

∫M2

b (x)

2EIdx (3.1)

where E is the Young’s modulus with material parameters, I is the area moment ofinertia, and Mb is the bending moment along the beam axis x which results fromexternal loads.The resulting deflection uA of characteristic points of the structure due to the appliedforce FA (auxiliary force) can be directly computed from the first derivative of themechanical energy

uA =∂W m

∂FA=

∫Mb(x)

EI· ∂Mb

∂FAdx. (3.2)

The other method, the Lagrangian method, is based on the superposition of shapefunctions. The final deformation state uD(x) along the beam axis is represented by aweighted combination of predefined shape or trail functions φi(x)

uD(x) =∑

qiφi(x) (3.3)

where qi are the unknown weights or generalized coordinates which describe how mucheach shape function contributes to the final deflection state. The shape functionscan be mathematical terms or eigenvectors of the linear mechanical system. Whenusing eigenvectors, the problem can be solved by superposition. It can be shown thatthe lowest eigenvectors, or modes, are sufficient to describe the static and dynamicbehavior with high accuracy [87].Rewriting the mechanical strain energy of Eqn. 3.1 as a function of n unknown weightswe obtain

W m(~q) =EI

2

∫ L

0

(n∑

i=1

qi∂2φ(x)

∂x2

)2

dx. (3.4)

Deriving this strain energy with respect to qi, the reacting spring forces of the ith

generalized coordinate can be calculated:

F mi (~q) =

∂W m

∂qi. (3.5)

The force balance equation (Eqn. 3.6) must be fulfilled for all n derivatives of theenergy function.

F mi (~q) = F el

i (V, ~q) i = 1..n, (3.6)

where F eli is the electrostatic force which acts on the ith mode (see electrostatic domain

simulations below for details). Dynamic properties such as the modal masses, Mi

31


(which correspond to each mode shape), and eigenfrequencies, Fi, can be obtainedfrom the kinetic energy equation at unit velocity:

W Ki (qi = 1) =

ρA

2

∫

φi(x)2dx =Mi

2(3.7)

Fi =1

2π

√

W mi (qi = 1)

W Ki (qi = 1)

. (3.8)

3.3.2 Electrostatic Domain Simulations

Remarkable force densities within small gaps can be achieved by electrostatic actu-ators. Such actuators are of high interest for MMIC as they can be readily manu-factured and integrated in standard semiconductor technologies. We will see that,unfortunately, such transducers require a rather large voltage when compared to lev-els used in standard electronic circuits. This is due to the fact that the electrostaticpressure, Pel, within small gaps is directly related to the local gap separation u:

Pel(V, u) =V 2ε0

2(d − u)2, (3.9)

where ε0 is the permittivity of air and d is the initial gap separation. One can seethat the electrostatic pressure goes to infinity as the gap separation approaches zero.Fringing fields can be neglected in most cases when the gap separation is much smallerthan the lateral dimensions. The electrostatic quantities can be described by modalshape functions φi(x) with respect to the generalized coordinates qi. Consequently,the capacitance function is given by

C(~q) =

∫ε0

d −∑

i qiφi(x)dA, (3.10)

the electrostatic field energy is given by

W el(V, ~q) =V 2C(~q)

2, (3.11)

and the force is given by

F el(V, ~q) =V 2

2· ∂C(~q)

∂qi. (3.12)

For a single shape function, the equilibrium of the stationary electromechanical systemof Eqn. 3.6 can be expressed by the voltage deflection relationship

V (q1) =

√

2 · ∂Em

∂qi

(∂C

∂qi

)−1

. (3.13)

32


When using more than one shape function to represent the deformation state, theresulting equation system has to be solved iteratively.

The optimum geometric dimensions and physical parameters can be found by analyz-ing the stationary behavior of the electromechanical system with analytical equations.After this analytical optimization process, a series of time consuming finite elementsimulations are carried out in order to assess the mechanical stress at different loadsituations, to compute the influence of initial warp, and to determine dynamic systemparameters such as damping ratios for transient analyzes [88].

3.3.3 Fluid Domain and Transient Response Simulations

The settling and release time of micro-mechanical structures after a voltage changedepends mainly on the viscous damping in the surrounding gas (e.g. air). Viscousdamping is caused by energy dissipation due to friction in the squeezed gas film betweenmembrane and fixed walls. Design parameters are damping coefficient Ci and dampingratios ζi characterizing damped vibrations from the physical point of view. Optimaldamping occurs if damping ratios are between 0.7 and 1.0 [89].Reynold’s squeeze film equation from lubricant theory can be used to compute thedamping parameter of mode shapes. The relation between the local beam velocityu(x) and the reacting pressure change p(x, y) in the squeezed gas film is give by

d3

12η

(∂2p(x, y)

∂x2+

∂2p(x, y)

∂y2

)

= u(x), (3.14)

where η is the dynamic viscosity of air and d the local gap separation. This partialdifferential equation must be solved for the wall velocities u(x) which correspond tothe modal shape functions φi(x). The results of the shunt airbridge and toggle switchwere obtained using the finite element tool ANSYS. The damping coefficients, Ci, canbe computed by scaling the pressure results pi(x, y) for each mode shape by φi(x) andintegrating them at the bottom face of the movable microstructure [90]:

Ci =

∫

φi(x)pi(x, y)dA. (3.15)

For the damping ratios ζi we obtain

ζi =Ci

2ωiMi

(3.16)

where ωi are the circular eigenfrequencies and Mi the modal masses.

33


In the case of optimal damping (0.7 ≤ ζ1 ≤ 1.0), the settling and release time after avoltage change take the duration of about one cycle [91]:

T1 =2π

ω1

=1

f1

(3.17)

Systems with a mechanical contact can be stimulated with voltages higher than neededto reach the stationary position. In this case, the movable beam hits the contact morequickly at the cost of a possible bouncing effect.

In order to simulate the transient response, the stationary force equilibrium of Eqn.3.6 must be extended by inertial and damping forces which leads to [92]

Miqi + Ci(~q)qi + F mi (~q) = F el

i (V, ~q). (3.18)

Movable microstructures are strongly non-linear. The mechanical stiffness, electro-static forces and damping coefficients vary with deflection and must be considered asfunctions of the generalized coordinates qi. Both the modal spring forces, F m

i , andthe electrostatic forces, F el

i , are available with analytical calculations. The dampingfunctions, Ci(~q), are derived from a series of finite element simulations which computethe damping coefficients at various deflection states in the operating range. Later, aregression algorithm (a least square fit) is applied in order to compute a mathemat-ical function (or response surface) which represents the damping data in Eqn. 3.18.The weights qi of Eqn. 3.18 are solved numerically by a Newmark time step integra-tion scheme. Afterwards, the deflection state of the entire structure can be computedaccording to Eqn. 3.3.

3.3.4 Electromagnetic Domain Simulations

To compute the scattering parameters, three-dimensional electromagnetic simulationswere performed with the three dimensional electromagnetic (EM) field simulator EM-PIRE developed at IMST GmbH. This simulator is a powerful tool for solving Maxwell’sequations based on the finite difference time domain (FDTD) method which includesall three-dimensional coupling effects. Using this method, the Maxwell equations arediscretized in time and space. This is accomplished by mapping the RF MEMS struc-ture under investigation onto a rectangular grid where the unknown field componentsare located in each cell. S-parameter simulations are then performed for the staticproblem. In the case for the switches, two independent S-parameter simulations wereperformed for the two states – one with an activated beam and one with a relaxedbeam.

The implemented FDTD method of the simulator solves an initial value problem byan efficient time stepping algorithm (the Yee’s leapfrog scheme). Any unknown field

34

3.4 Design, Layout, and Simulation Results

at a certain time is computed from the field values of the preceding time step. Forstability reasons, the size of the time steps is related to the size of the grid and cannotbe defined independently. Therefore, a suitably sized grid is needed for an efficientsimulation. Both switch types, capacitive shunt airbridge switch and toggle switch,utilize a thin 200 nm nitride isolation layer which is relatively small compared to theoverall dimensions of several millimeters of the whole switch structure. Extremely longsimulation times would be caused by the thin sheets. To overcome this problem, a newstability criteria for FDTD simulations with strongly graded meshes (e.g. a 200 nmnitride sheet next to a 3 µm metal sheet) is applied. During the optimization processof the switches, thicker equivalent sheets are used in the simulation. These thickersheets have a proportionally increased permittivity

εincr = ε · dinc

d0

(3.19)

where d0 is the original sheet thickness. The resulting capacitance of the switcheswithin this approximation is nearly correct. However, the original nitride film thick-ness was used in the final simulation of the structures before manufacturing. Theresults are presented later in this chapter.

The metal losses are included in the simulation by applying resistive sheets. Theequivalent sheet resistance Rsh at frequency f is given by

Rsh =1

2σa(3.20)

with the skin effect penetration depth

a =1√

π · f · µ0 · σ(3.21)

where σ is the intrinsic conductivity, and µr is the magnetic permeability. The sim-ulated losses are frequency dependent and, therefore, only accurate at a specifiedfrequency (20 GHz in all simulations). In reality, the losses below this frequency willbe smaller, and the losses above this frequency will be larger.


The design of RF MEMS switches is influenced by their mechanical as well as elec-tromagnetic properties. For both fields, the simulations described in section 3.3 wereperformed. To simulate the real behavior of the dynamic switch structures, combinedmechanical and electromagnetic simulation are required. However, such simulationsare very complex and not necessarily needed to design and understand the behavior ofthe final switch structure. Both switch types, shunt airbridge and toggle switch, are

35


digital two-state switches and are either “on” or “off” during the designated operation.The switching or activation procedure itself is certainly interesting to investigate anddynamic electromagnetic simulations might give more accurate results for activationtime, release time, activation voltage, and velocity of the membrane when hitting theelectrode. However, very accurate simulation results for these values are not of greatimportance when the main focus is on optimizing the scattering parameters for thetwo discrete states.

In the starting phase of the design, mechanical simulations were performed in orderto get enough information to understand mechanical properties of the targeted ma-terials and geometry. In a second phase, electromagnetic simulations are performedto achieve accurate results for the scattering parameters of the switch structure. Wewill see that the simulation results achieved not only show the correct tendency of thescattering parameters over the frequency, but also show accurate absolute results.

All RF structures were designed for manufacturing on 4 inch Si wafers with a resistivityof 4000 Ωcm. Typical dimensions of a 50 Ω CPW are 144 µm for the signal line widthand 78 µm for the gap between signal and ground lines.

3.4.1 Shunt Airbridge Switch

The geometry and the functional principle of the capacitive shunt airbridge switch isdepicted in Fig. 3.2 and Fig. 3.3. In a 50 Ω CPW environment, a flexible airbridgecalled “beam”, or “membrane,” is attached to the ground lines. To activate themembrane, a voltage of approximately 40 V is needed between the signal line andthe movable membrane (i.e. the DC lead). This results in an electrostatic force whichacts on the membrane above the ground electrode. In the ”activated” or “down” state,a nitride isolation layer on top of the signal line prevents both irreversible sticking anda dissipative current flow between signal and ground contacts.

Voltage Displacement Relationship of the Shunt Airbridge Switch

The mechanical simulation of the clamped flexible membrane is well described by theshape function technique explained in Chap. 3.3. The deformation state is rathersimple and represented by a single shape function φ(x). This shape function resultingfrom uniform pressure is given by

φ(x) =16

L4x4 − 32

L3x3 +

16

L2x2, (3.22)

where L is the length of the membrane.

36


membrane

isolation

signal line

gound line

port 2

port 1substrate surface

Fig. 3.2: Three-dimensional view ofthe shunt airbridge switch.

groundsubstrate

ground

movable membrane DC lead

signal line

isolation

(a) Membrane relaxed (Vact = 0 V )

groundsubstrate

ground

DC lead

signal line

(b) Membrane activated (Vact ≈ 40 V )

Fig. 3.3: Cross-sectional view of theshunt airbridge switch.

It was shown earlier that the equations of motion of electromechanical systems canbe established from the energy and capacitance functions. For the shunt airbridgeswitch, the bending energy from Eqn. 3.4 must be superimposed by a second term forthe stiffness change due to initial pre-stress, σu, as well as a third term for the effectsof stiffness at large deflections. The resulting strain energy is therefore given by

W m(q) =EI

2

∫ L

0

(∂2φ

∂x2

)2

dx +σuA

2

∫ L

0

√

1 +

(∂φ

∂x

)2

dx − L

+EA

2L

∫ L

0

√

1 +

(∂φ

∂x

)2

dx − L

2 (3.23)

and can be processed with MATLAB. The spring force, FS, can then be calculatedfrom the first derivative of Eqn. 3.23. The resulting shape function (mechanical de-formation) of the membrane of the shunt airbridge switch is depicted in Fig. 3.4. Theshading indicates the structural displacement uy(x).

More obvious than the force deflection relationship is the voltage deflection functiongiven by Eqn. 3.13 and shown in Fig. 3.5. The maximum of that function representsthe voltage which is necessary to activate the contact. It is remarkable that the mem-brane is not necessarily released if the voltage is lower than the pull-in voltage. As amatter of fact, the release voltage is usually much lower (hysteresis) and depends onthe thickness of the isolation layer. In the given example, the membrane is activatedin the maximum of the deflection function at 39 V and released at 14 V.

37


(a) Relaxed membrane

(b) Activated membrane

Fig. 3.4: Shape function simulation ofthe shunt airbridge switchmembrane.

0

5

10

15

20

25

30

35

40

0 0.2 0.4 0.6 0.8 1

voltage [V

]

deflection/gap ratio [1]

hysteresis release voltage

pull−in voltage

insula

tion layer

Fig. 3.5: Voltage deflection functionof a shunt airbridge switchshows hysteresis.

2.7

2

1

0 0 5 10 15 20 25 30

dis

pla

cem

ent

[µm

]

time [µs]

25 V30 V40 V

(a) Membrane activation

2.7

2

1

0

−1

−2

−2.7

0 20 40 60 80 100

dis

pla

cem

ent [µ

m]

time [µs]

(b) Membrane release

Fig. 3.6: Membrane displacement of the shunt airbridge switch as a function of time.

Switching and Release Time of the Shunt Airbridge Switch

The transient responses after different voltage jumps are depicted in Fig. 3.6(a). Forall voltages, the membrane moves smoothly against the contact. This is because bothelectrostatic and damping forces grow almost in the same order. However, to decreasethe settling time, higher driving voltages should be used. The displacement of 2.7 µmcorresponds to the distance between the membrane in the neutral position and theelectrode. Activation times depend on the activation voltage and vary from 9 µs to 24µs. After releasing the voltage, the membrane oscillates back into its initial position(Fig. 3.6(b)). The corresponding release time of 100 µs is larger than the activationtime due to the following reasons:

1. The relaxed state has a significantly lower damping ratio when compared to the

38


activated state where the membrane touches the electrode.

2. The stiffness and the eigenfrequency of the membrane is lower in the relaxedstate than in the activated state. This lower eigenfrequency leads to a largerperiod and longer cycles (see Eqn. 3.17).

Electromagnetic Simulation Results of the Shunt Airbridge Switch

To improve the RF scattering characteristics of the shunt airbridge switch, an LCmatching network was designed. This matching network compensates the capacitanceof the switch in a broad frequency range from DC to 30 GHz. The LC matchingnetwork consists of two serial inductances in a T configuration with the shunt capac-itance. The matching at the feeding port S11 as a function of the capacitance andinductance was investigated to improve the structure. It was found that a capacitanceof 100 fF can be compensated by a total inductance of 350 pH up to a frequency of34.5 GHz. For a lower frequency of 30 GHz, a maximum capacitance of 118 fF canbe compensated by an inductance of 450 pH yielding a matching of 20 dB. Higherinductances decrease matching significantly. Fig. 3.19 on page 52 shows the geometryand dimensions of the designed switch as it is manufactured. The inductive lines havea reduced width of 20 µm.

−50

−40

−30

−20

−10

0

0 5 10 15 20 25 30 35 40

0

−0.2

−0.4

−0.6

−0.8

−1

isola

tion, re

turn

loss [

dB

]

insert

ion loss,

reflection [

dB

]

frequency [GHz]

insertion lossreturn loss

isolationreflection

Fig. 3.7: Simulated S-parameter of the shunt airbridge switch. Notice that the in-creased capacitance of the membrane clearly increases the isolation above3 GHz.

The simulation result of this optimally designed shunt airbridge switch is shown in Fig.3.7. This switch is closed when no pull-in voltage is applied (i.e. when the membraneis relaxed (see Fig. 3.1 for overview of this notation)). In this state most of the powercan move from the input to the output port. Fig. 3.7 shows that the return loss isabove 30 dB for frequencies up to 25 GHz, and above 20 dB for frequencies up to 40

39


GHz. When the membrane is activated by a pull-in voltage, the switch opens andthe electrical state is “off”. In this state, less power can move from the input to theoutput port. Due to the inherent design of a shunt airbridge switch, the isolation ofthe switch is 0 dB at DC and increases almost linearly towards higher frequencies (upto approximately 30 dB at 40 GHz). The reflection (S11 in closed state) shows thatmore and more power gets reflected at higher frequencies.

3.4.2 Toggle Switch

The geometry and the functional principle of the toggle switch is depicted in Fig. 3.8and Fig. 3.9. The toggle switch can be regarded as a conventional air gap switch, orrelay, made for high frequencies. In a 50 Ω coplanar waveguide (CPW) environment,

cantilever

signal lineof CPW

flexiblemetalband

suspension

airbridge

ground lineof CPW

pushelectrode

pullelectrode

contact paddle

Fig. 3.8: Three-dimensional view ofthe toggle switch.

signalline

signalline

substratepull electrode

suspension

push electrode

(a) Contact closed (down-state) - voltage isapplied to pull electrode

signalline

signalline

substrate

movable cantilever

pull electrodepush electrode

suspension

(b) Contact open (up-state) - voltage is ap-plied to push electrode

Fig. 3.9: Cross-section view of the tog-gle switch.

a cantilever is embedded in the signal line and can open and close an ohmic contact inresponse to an external voltage. The metal cantilever is attached to the signal line onone side by a flexible metal band, and additionally, it is supported by a suspension.This flexible torsion spring is isolated and attached to the ground lines. In the closedstate, the signal is routed from the input port along the flexible metal band, acrossthe cantilever, and over the ohmic contact to the output port. The switch is closed byapplying a DC voltage to the pull electrode which activates the cantilever (Fig. 3.9(a)).In this state, the tip of the cantilever is in contact with the contact paddle. An iso-lation layer on top of the ground electrode prevents a dissipative current flow whenthe cantilever touches the electrode. When the external DC voltage is switched off,the cantilever is released and the contact is opened. To improve isolation in the openstate, the switch is designed in such a way that an external voltage can be applied tothe push electrode which increases the gap between the cantilever tip and the contact

40


paddle (Fig. 3.9(b)). The airbridge serves as a stop for the cantilever when it bends up.

Voltage Displacement Relationship of the Toggle Switch

For the design and the mechanical domain simulation, the cantilever can be repre-sented by a simplified three-dimensional model. One must take into account that boththe cantilever and flexible metal band bend significantly under an electrostatic loadand should be considered as flexible. The suspension springs are mapped by a springelement with two degrees of freedom which captures the transversal stiffness, Ctrans,and torsional stiffness, Crot. Furthermore, the electrostatic pressure is strongly depen-dent on the deflection and must be updated in each iteration cycle. To determine thedisplacements along the beam axis at various DC voltages, the Principle of Castigliano(see Chap. 3.3.1) is employed using the following iteration scheme [93]. Steps 1 to 5are repeated until convergence occurs:

1. Computation of the electrostatic pressure along the beam as a function of localdisplacement:

Pel(uy(x)) =ε0V

2

2(d − uy(x))2(3.24)

2. Evaluation of the lateral shear force Fq which depends on the pressure and thetransversal force of the suspension spring FS (unknown arbitrary parameter):

Fq(x, FS) =

∫ x

0

Pel(s)ds + FS (x > a) (3.25)

3. Evaluation of the bending moment Mb along the beam axis. The spring momentMS is an unknown parameter:

Mb(x, MS) =

∫ x

0

Fq(s)ds + MS (x > a) (3.26)

4. Calculation of the total mechanical strain energy. Ctrans and Crot are the transver-sal and torsional stiffness of the torsion spring:

WS(FS, MS) =1

2EI

∫

M2b (x)dx

︸︷︷︸

cantilever

+F 2

S

2Ctrans+

M2S

2Crot︸︷︷︸

torsion spring

(3.27)

5. Computation of the unknown spring force FS and moment for the current loadsituation MS:

∂WS(FS, MS)

∂FS= 0,

∂WS(FS, MS)

∂MS= 0 (3.28)

41


6. Computation of all displacements along the beam axis:

uy(x) = − 1

EI

∫ x

0

Mb(s − x)ds (3.29)

Convergence occurs if the structural displacement at step 6 is constant.

The simulations show that reasonably small voltages are required to pull the tip ofthe cantilever down to the contact paddle. Any further increase of the applied voltagewill lead to instability called “pull-in.” Pull-in occurs when the electrostatic forcesgrow faster than the spring forces with respect to the displacement. In this case, thecantilever snaps down to the pull electrode. This usually happens when the structureis displaced to about 33% to 43% of the initial gab.

In addition to the analytical model, the state of the art finite element tool ANSYS wasused to verify the stationary deformation state and to compute the stress distribution.The program allows simultaneous simulation of different physical models [94]. In themechanical region, the structure is described with approximately 3000 hexahedral solidelements; in the electrostatic domain, by about 800 transducer elements. These trans-ducer elements are based on quasi analytical descriptions of the capacitance-strokefunction within small gaps, and hold for systems where the fringing fields can be ne-glected. They compute the local electrostatic force density with regard to the appliedvoltage and deflection state, and they transfer the loads to the mechanical domain.The deviations between both simulations – analytical method and ANSYS – are below3%.

The geometry and dimensions of the simulated structure can be seen in the SEMmicrograph of the manufactured structure in Fig. 3.21(a) on page 54. The gap betweencantilever and electrode is approximately 2 µm. The contact closes when a pull voltageof about 5 V is applied to the pull electrode (Fig. 3.10(a)). A further increase of thepull voltage to 12 V raises the force between the tip of the cantilever and the contactpaddle and bends the center of the cantilever down towards the pull electrode (Fig.3.10(b)). A sudden snap, or pull-in, happens at 25 V (not shown). To avoid pull-induring operation, the voltage should be kept well away from the critical voltage andbelow 20 V. When applying a push voltage of 40 V to the push electrode, the tip ofthe cantilever rises about 0.6 µm above its neutral horizontal position (Fig. 3.10(c) -lift-up mode).

42


(a) Pull down mode (Vact = 5 V)

(b) Pull down mode (Vact = 12 V)

(c) Lift up mode (Vact = 40 V)

Fig. 3.10: Mechanical displacementsimulation of the tog-gle cantilever (shadingindicates height).

0.2

0

−0.2

−0.4

−0.6 0 50 100 150 200 250 300

dis

pla

ce

me

nt

[µm

]

time [µs]

cantilever tip

center of pull electrode

center of push electrode

(a) Pull down mode (Vact = 5 V)

0

−0.5

−0.8 0 50 100 150 200 250 300

dis

pla

ce

me

nt

[µm

]time [µs]

cantilever tip (hits paddle)

center of pull electrode


(b) Pull down mode (Vact = 12 V)

−0.2

0

0.2

0.4

0.6

0.8

0 50 100 150 200

dis

pla

cem

ent

[µm

]

time [µs]

cantilever tip


(c) Lift up mode (Vact = 40 V)

Fig. 3.11: Membrane displacement ofthe toggle cantilever as afunction of time.

Switching and Release Time of the Toggle Switch

Switching and release time of the toggle switch depend strongly on the damping ra-tios of the lowest eigenvectors. Numerical fluid flow simulations based on Reynold’ssqueeze film equation have been performed with the finite element tool ANSYS. Thesimulations show increased gas pressure at the center (underneath the cantilever whenpulling downwards) due to viscous friction. Therefore, perforation holes in the can-tilever are added to compensate for this effect (3.21). The transient response andcantilever displacement after a voltage jump is illustrated in Fig. 3.11. It can be seenthat an activation with 5 V in pull down mode takes approximately 300 µs (3.11(a)).As expected, activation with an increased voltage of 12 V takes less time – approxi-

43


mately 50 µs (3.11(b)). In this case, the center of the pull electrode bends downwards.In push mode, the cantilever tip bends upwards and reaches its maximum of 0.6 µmin approximately 100 µs when applying a voltage of 40 V.

In addition to these functional parameters, the simulations give insight into the frac-tural strength and fatigue behavior of the movable mechanical structures. As a rule ofthumb, the mechanical stress should be much lower than the yield stress of the involvedmaterials. The simulations state a peak stress of about 12 MPa in the silicon nitridesuspension spring. This value is significantly lower than the critical value reported ofabout 500 MPa [95]. Fig. 3.8 shows that curved clamps were used in the design inorder to minimize the stress concentration due to notching effects.

Electromagnetic Simulation Results of the Toggle Switch

The toggle switch has a parasitic capacitance due to the small distance of approxi-mately 2 µm between the cantilever and the grounded DC switching electrodes. AnLC matching network achieves a broadband compensation of this capacitance. Forthe regular toggle switch, a reduced signal line width of 32 µm with a length of 100µm for the input port and 120 µm for the output port gives optimum results. Thescaled toggle switch [96] (second design) has a compensation line width of 20 µm witha length of 120 µm on both sides.

The magnitude of the electric field of the regular toggle switch is depicted in Fig. 3.12.There is a 60 dB difference between maximum and minimum field values for boththe closed state (Fig. 3.12(a)) and the open state (Fig. 3.12(b)). In the closed state,the highest field values occur under the cantilever due to the small separation of thegrounded DC electrode. In the open state, the highest field values occur at the endof the inductive compensation line in the signal path where the flexible metal bandis attached (distinct peak in 3.12(b)). Lowest field values are found in the substratebetween the signal and the ground line.

S-parameter simulation results for regular and scaled toggle switches for activated(closed, on) and relaxed (open, off) states are shown in Fig. 3.13 (see Tab. 3.1 fordetails on notation). When activated, the regular toggle switch in Fig. 3.13(a) showsan insertion loss less than 0.25 dB and a return loss higher than 15 dB up to 40 GHz.This indicates good matching of the structure to the 50 Ω CWP environment. In thiscase, the simulation is neglecting metal losses. When relaxed or open, an isolationhigher than 19 dB up to 40 GHz is found. Up to 10 GHz the isolation is higher than23 dB. The reflection which is smaller than 0.1 dB indicates, that in this case, mostof the power is reflected at the input port.

44


(a) Contact closed (b) Contact open

Fig. 3.12: Magnitude of the electric field of the regular toggle switch at 20 GHz inboth switching states (height indicates the magnitude of the electric fieldstrength).

The improved smaller version of the toggle switch shows even better performance (Fig.3.13(b)). In the closed position, a return loss lower that 33 dB is found over the entirefrequency range from DC to 40 GHz. Insertion loss, including metal losses, is smallerthan 0.18 dB. In open position, an isolation greater than 18 dB up to 30 GHz isachieved, similar to the regular toggle switch. This is because the determining factoris the gap between the cantilever tip and the contact paddle which is the same in bothcases.

0

−20

−40

−60

−80 40 30 20 10 0

0

−0.1

−0.2

retu

rn loss,

isola

tion [

dB

]

insert

ion loss,

reflection [dB

]

frequency [GHz]

insertion loss (act)return loss (act)

isolation (rel)reflection (rel)

(a) Regular toggle switch

0

−20

−40

−60

−80 40 30 20 10 0

0

−0.1

−0.2

−0.3

−0.4

−0.5

retu

rn loss,

isola

tion [

dB

]

insert

ion loss,

reflection [dB

]

frequency [GHz]

insertion loss (act)return loss (atc)

isolation (rel)reflection (rel)

(b) Small toggle switch

Fig. 3.13: S-parameter simulation results for the two different designs of the toggleswitch neglecting metal losses.

45


3.4.3 Single Pole Double Throw (SPDT) Switch

The higher order SPDT switch is composed of the shunt airbridge switches and toggleswitches which are described in Chap. 3.4.1 and Chap. 3.4.2. The geometry and layoutcan be seen in Fig. 3.14. The functional principle of the RF MEMS SPDT switch isthat of a conventional SPDT switch. A signal at the input port (port 1) can eitherbe routed around a 90 degree bend to port 2 or can proceed straight to port 3 (Fig.3.14(a)). This is achieved by two toggle switches in the intersection at the center of thestructure that can alternately tap the input line (Fig. 3.14(b)). The shunt airbridgeswitches are even used to increase isolation at high frequencies [14] [39]. Two versionsof the SPDT switch are designed: one composed of two regular sized toggle switches(as shown in Fig. 3.14), and one composed of two small toggle switches (Fig. 3.23(d)).

port 1

port 2 port 3

(a) Overview

port 2port 3

airbridge

shunt airbridgeswitch

togglecantilever

togglesuspension

toggleelectrode

CPWsignal line

port 1

(b) Magnified view (note that ports are rotated)

Fig. 3.14: Three-dimensional view of the SPDT switch made of two toggle switchesand two shunt airbridge switches

.

Electromagnetic Simulation Results of the SPDT Switch

The small size of the scaled toggle switches allows for a compact 1 mm2 SPDT switch.The SPDT structure needs an adapted LC matching network (inductive lines) witha width of 25 µm and a length of 180 µm at the input port to compensate for theadditional capacitance of the crossing airbridge. At port 3, a matching line with alength of 70 µm is needed. With the use of the taper in the ground line, no matchingis required for port 2. The airbridges that cross over the toggle cantilever are used tostrongly suppress unwanted higher modes on the coplanar line if the signal is routedaround the 90 degree bend.

46


From the electromagnetic simulations, the current density distribution is obtained. Adifference of 50 dB is found between the lowest values behind the shunt airbridge atport 3 and the highest values on the cantilever tip, the small matching lines, and onthe ground signal airbridges. The cantilever of the open toggle switch still shows ahigh value due to induced current.

Scattering parameters of the SPDT switch are shown in Fig. 3.15. When the signal isrouted from port 1 to port 2, the insertion loss is smaller than 0.2 dB and the returnloss is above 30 dB for frequencies up to 35 GHz (Fig. 3.15(a)). The isolation to port3 in this routing state is greater than 38 dB. When routing the signal straight throughthe structure, from port 1 to port 3, the insertion loss is below 0.28 dB, and the returnloss is above 30 dB for frequencies up to 30 GHz (3.15(b)). The isolation to port3 in this routing state is above 40 dB up to 30 GHz. A reduced isolation occurs athigher frequencies due to substrate modes and can be improved by the use of a thinnersubstrate.

0

−20

−40

−60 30 20 10 0

0

−0.1

−0.2

−0.3

−0.4

−0.5

retu

rn loss,

isola

tion [dB

]

insert

ion loss [

dB

]

frequency [GHz]


isolation (port 3)

(a) Signal routed from port 1 to port 2

0

−20

−40

−60 30 20 10 0

0

−0.1

−0.2

−0.3

−0.4

−0.5

retu

rn loss,

isola

tion [dB

]

insert

ion loss [

dB

]

frequency [GHz]


isolation (port 2)

(b) Signal routed from port 1 to port 3

Fig. 3.15: S-parameter simulation results of the SPDT switch composed of smalltoggle switches for the two different routing states.

3.4.4 RF Cross

The RF cross depicted in Fig. 3.16 is designed to continuously route two signals acrossan intersection in a CPW environment without the need for switching. This RF crossis very useful when designing planar circuits without using multiple conductive lay-ers with via holes. EM field simulations with EMPIRE have been carried out duringthe design phase that supports all three-dimensional coupling effects between the twosignal lines [40]. The ground-to-ground spacing in the center of the cross is 110 µm.This relatively small width reduces the effective field and allows the use of a shorterairbridge. From port 2 to port 4, the signal is routed via an underpath that has a

47


metal upper shielding airbridge on top. From port 1 to port 3, the signal is routedalong a cross signal airbridge that runs on top of the crossing signal line with a lowershielding airbridge sandwiched in between.

port 1

underpath

uppershieldingairbridge

lowershieldingairbridge

crosssignalairbridge

ground

signalline

port 2 port 3

port 4

ground

signalline

(a) Three-dimensional view of the RF cross

ground

port 3

port 1

crosssignalairbridge

lowershieldingairbridge

signalroute

underpath

uppershieldingairbridge

port 2

port 4

(b) Electric field above cross-signal airbridge

Fig. 3.16: Three-dimensional view of the RF cross and electric field distributionabove the cross-signal airbridge.

Electromagnetic simulations reveal that the RF cross has a strong concentration ofthe scattering field in a small area above and below the underpath line when no lowershielding airbridge is applied. When the signal travels over the cross signal airbridgefrom port 1 to port 3, the electric field is located mainly between the cross signalairbridge, the underpath line, and at the edges of the airbridge. The scattered field iscomparably weak. The use of a grounded lower shielding airbridge prevents a directcoupling between the two lines. The outside coupling through air is minimized withthe use of upper shielding airbridges that connect the ground metalizations in a largearea of 600×600 µm around the cross. An overlap of 10 µm between the two shieldingairbridges prevents radiation leakage from the underpath to the cross signal airbridge.Because of the shieldings, the field of the underpath signal line is concentrated belowthe shielding bridges and in the substrate. The field from the crossing signal line isconcentrated mainly above the upper shielding bridge and in the air (Fig. 3.16(b)). Foroptimization of the RF performance, the small center conductor lines have been de-signed to create a microstrip mode against the shielding electrodes with an impedancenear 50 Ω. Short inductive lines are used for compensation outside the shielded area.

The S-parameter simulation results of the RF cross are shown in Fig. 3.17. Whenthe signal is routed over the airbridge (Fig. 3.17(a)), an insertion loss smaller than0.55 dB, and a return loss greater than 25 dB, is found for frequencies up to 40 GHz.The slightly higher insertion loss when the signal is routed along the underpath isdue to higher conductor losses in the 300 nm thin cross signal airbridge (Fig. 3.17(b)).However, the insertion loss is still below 0.65 dB up to 40 GHz. In this case, the return

48

3.5 Process Flow and Fabrication

0

−20

−40

−60 40 30 20 10 0

0

−0.2

−0.4

−0.6

−0.8

−1

retu

rn loss,

iso

latio

n [

dB

]

insert

ion

lo

ss [

dB

]

frequency [GHz]


isolation (port 2)

(a) Signal routed over airbridge (port 1 to 3)

0

−20

−40

−60 40 30 20 10 0

0

−0.2

−0.4

−0.6

−0.8

−1

retu

rn loss,

iso

latio

n [

dB

]

insert

ion

lo

ss [

dB

]

frequency [GHz]


isolation (port 3)

(b) Signal routed along underpath (port 2 to 4)

Fig. 3.17: S-parameter simulation results of the RF cross for the two different rout-ing states representing all symmetric routing combinations.

loss is above 20 dB for frequencies up to 40 GHz. Due to the use of the lower shieldingairbridge, the isolation between the two signal paths is above 40 dB for frequencies upto 40 GHz. The imbalance in the insertion loss can be compensated by increasing thelength of the underpath signal line.


All MEMS structures introduced in Chap. 3.4 are manufactured simultaneously on thesame silicon wafer during the fabrication process. Some steps apply only to certainstructures. A total of 42 different manufacturing steps using twelve different masksare needed for these designs. The process flow in Fig. 3.18 exemplarily shows themanufacturing of the toggle switch. It can be easily applied to the corresponding partsof the shunt airbridge switch and RF cross. A thorough description of the lithographysteps and the functional principle of the machines involved for metal sputtering, metalevaporation, dry chemical etching, and plasma enhanced chemical vapor deposition(PECVD) can be found in [97].

The RF MEMS switches and the RF cross are fabricated on oxidized, high-resistivity,525 µm thick, 4 inch diameter Si wafers with a resistivity greater than 4000 Ωcm.To further minimize substrate losses, a high quality SiO2 of 400 nm is deposited byPECVD and annealed at 1000C for 60 s in preparation for processing.

In the first process step, the lower electrode (underpass metalization) is defined by alift-off process (Fig. 3.18(a)). The two metals, 50 nm Ti and 250 nm Au, are depositedin a vacuum by high temperature evaporation. The structured photo resist (PR) used

49


substrate p-Si ρ > 4000 Ωcm

300 nm Au50 nm Ti

pull and push electrodes

(a) Lower electrodes (underpass metalization)


100 nm Si3N4

(b) Isolation layer for lower electrode

Au Au


2500 nm Au50 nm Ti

signal lineof CPW

(c) Definition of CPW lines

Au Au


1st sacrificial layer

(d) First sacrificial layer for cantilever-to-electrode distance definition

Au Au



torsion spring570 nm Si3N4

(e) Torsion spring formation

Au Au



400 nm

(f) Contact hole etching

Au


Au1st sacrificial layer

photoresist filling

(g) Filling of the contact hole

Au Au



cantilever

400 nm Au75 nm Ni

400 nm Au

(h) Cantilever formation

Au Au



2nd sacrificiallayer

(i) Second sacrificial layer for cantilever-to-airbridge distance definition

Au Au


cantileverairbridge

metalflexible

band

(j) Airbridges and the flexible metal band af-ter removal of both sacrificial layers in aCPD process

Fig. 3.18: Process flow for the fabrication of the toggle switch (cross-sectional view)– all other MEMS elements are fabricated by the same process.

in a lift-off process has an undercut profile that leads to sharp edges and vertical wallsin the metalization layer [97].

The 100 nm thick silicon nitride (Si3N4) passivation for lower electrodes and lower DCleads is deposited over the entire wafer by PECVD at 320C. Then, the photo resist

50


is structured and selectively etched in a LAM reactor by a CF4/CH3F plasma in a Heenvironment (Fig. 3.18(b)).

Signal and ground lines of the CPW lines have the same height of 2550 nm (50 nmTi and 2500 nm Au) and are defined by metal evaporation and a subsequent lift-offprocess (Fig. 3.18(c)).

The planar filling of the region between ground and signal lines with a sacrificial layeris a critical step. This first sacrificial layer defines the electrode-membrane distanceand needs to have a flat interface between the photo resist and the Au ground metal asidealized in Fig. 3.18(d). The overlapping photo resists on the ground pads is removedby a sophisticated, multiple step planarization process that was specially developedfor this problem.

Then, a second SiN isolation layer is deposited on top. This layer is 570 nm thick andforms the torsion spring for the toggle switch (Fig. 3.18(e)) as well as the isolation forthe contact leads. These DC leads run on top of the ground metal and contact themembranes of the shunt airbridge switch in SPDT designs. They are, in contrast tosingle shunt airbridge switches, isolated against the ground.

After this, the contact paddle for the toggle switch is defined by wet chemical etchingof 400 nm of gold at the area shown in Fig. 3.18(f). To make the contact paddle harderand, therefore, more robust, a 85 nm thick WTi layer is sputtered. The resulting holein the signal line needs to be filled again with photo resist to form an even togglecantilever which will be deposited on top.

Afterwards, the different cantilever metalization layers are sputtered. The cantilevermaterial consists of 400 nm of Au, 75 nm of Ni, and 400 nm of Au (Fig. 3.18(h)).The sandwich structure with the Ni layer is an elegant solution to control and adjustthe stress in the cantilever. The stress in the layer can be changed by thickness mod-ification and by variation of the sputtering process parameter (temperature and ionacceleration voltage). The final switch parameters (actuation voltage and switchingspeed) strongly depend not only on the geometry of the membrane, or cantilever, butalso on its metal and sputtering parameters which define tension and residual stress.While a change of these parameters for the membrane (which is fixed on two sides) isless critical, it becomes a more delicate matter for the toggle switch. This is mainlybecause one side of the toggle cantilever is not attached to anything and tends tostrongly bend up or downwards if the stress parameters are not well controlled.

In the last step, the flexible metal band, the upper stop bar for the toggle cantilever,

51


and the additional ground-to-ground airbridges are added. For this, an airbridge resistwith a height of 3 µm is patterned as a second sacrificial layer and annealed (Fig.3.18(h)). The 1200 nm Au is evaporated and structured by a lift-off process. Finally,the hard photo resist of both sacrificial layers is carefully removed with wed chemicaletching with H2O2:H2SO4 (1:4) and isopropanol. The critical point drying (CPD)technique is used to dry the finished wafer so as to avoid sticking of the membraneand cantilever.

3.6 SEM Micrographs and Experimental RF

Measurement Results

In this chapter, the experimental results of the fabricated RF MEMS, the toggle switch,the shunt airbridge switch, the SPDT switch, and the RF cross are presented. SEM mi-crographs were taken with a state of the art scanning electron microscope. To measurethe S-parameters, all RF MEMS were designed in CPW ground-signal-ground config-uration and were contacted in a Cascade probe station. The RF measurements werethen performed with a HP 8510C network analyzer. External voltages for actuationwere applied by Keithley 238 source measurement units. A standard line-reflect-match(LRM) method was used for calibration of the RF equipment.

3.6.1 Experimental Results of the Shunt Airbridge Switch

(a) 100 µm-wide membrane (b) 50 µm-wide membrane

Fig. 3.19: SEM micrograph of the shunt airbridge switches with perforation holes.

Fig. 3.19 shows the shunt airbridge switches in CWP design. The lighter part on theright and on the left side in both SEM micrographs is the ground metalization. The

52

3.6 SEM Micrographs and Experimental RF Measurement Results

darker part in between the signal and ground metal shows the silicon oxide of the wafersurface. The perforation holes in the membrane were applied for faster actuation asmentioned earlier. The switch in Fig. 3.19(a) has a 100 µm membrane which has ametal-to-metal contact with the ground line. Note that the SEM micrograph in Fig.3.19(a) shows a single pole single throw (SPST) version of the shunt airbridge switchwhich does not have a DC lead. Actuation of SPST shunt airbridge switches can bedone by applying a voltage between the signal and the ground line. The darker elec-trode below the membrane has a SiN cover. The 50 µm wide membrane of the shuntairbridge switch in Fig. 3.19(b) is isolated against the ground metal by SiN and hasa separate DC lead for actuation (right part of the SEM micrograph). The electrodebelow the membrane has the same width as the signal lines, but additional inductivelines (the thinner part in the signal line) are used to compensate for the capacitancein the non-actuated state.

−60

−50

−40

−30

−20

−10

0

0 5 10 15 20 25 30 35 40

0

−1

−2

−3

isola

tion,

retu

rn loss [

dB

]

insert

ion loss,

reflection [

dB

]

frequency [GHz]


isolationreflection

(a) 100 µm-wide membrane, Vact = 39 V

−60

−50

−40

−30

−20

−10

0

0 5 10 15 20 25 30 35 40

0

−1

−2

−3

isola

tion,

retu

rn loss [

dB

]

insert

ion loss,

reflection [

dB

]

frequency [GHz]


isolationreflection

(b) 50 µm-wide membrane, Vact = 27 V

Fig. 3.20: S-Parameter measurement results of the shunt airbridge switches depictedin Fig. 3.19.

S-parameter measurement results of the shunt airbridge switches from Fig. 3.19 aredepicted in Fig. 3.20. Both switches clearly show the simulated RF behavior. At 40GHz, there is a greater than 30 dB decrease in transmission between “on” and “off”state. This value is found from the difference between the isolation and insertion lossat 40 GHz. When the membrane is activated, isolation (S12) and reflection (S11) arethe parameters of interest; whereas in the relaxed state, insertion (S12) and return loss(S11) are given (see Fig. 3.1 for the RF notation of the shunt airbridge switch). In theclosed state (membrane relaxed), the 100 µm version of the shunt airbridge switch inFig. 3.20(a) has an insertion loss smaller than 0.4 dB and a return loss greater than 12dB up to 40 GHz. In the open state (membrane activated), the isolation is better than40 dB while the reflection of 0.5 dB indicates that most of the RF power is reflectedat higher frequencies. Comparison of the insertion loss with isolation over the wholefrequency range reveals that the capacitance change of the shunt airbridge switch has

53


no effect at frequencies below 1 GHz and increases to almost 40 dB at 40 GHz. Thesmaller 50 µm wide version of the shunt airbridge switch in Fig. 3.20(b) shows similarbehavior. It has a slightly better return loss of 18 dB at 40 GHz but an isolation ofonly 30 dB at 40 GHz. This is due to its smaller capacitance.

3.6.2 Experimental Results of the Toggle Switch

(a) Regular toggle switch (b) Small toggle switch with airbridge

(c) Cantilever tip and contactpad of (a)

(d) Cantilever tip and con-tact pad of (b)

(e) Inductive flexible metalband of (b)

Fig. 3.21: SEM micrographs of the two types of the toggle switches.

SEM micrographs of the fabricated toggle switches – the regular and the small version– are depicted in Fig. 3.21. The regular toggle switch in Fig. 3.21(a) has a slightlymodified suspension of Au-Ni-Au compound which extends from the cantilever to themounting poles. The cantilever has three perforation holes in the tip for faster actua-tion. The slightly darker part underneath the cantilever is the SiN cover on top of theelectrodes, where the larger is the pull electrode (towards the right) and the smalleris the push electrode (towards the left and on the back part of the cantilever). On theleft hand side, the flexible metal band that connects the cantilever to the signal linecan be seen. The DC leads for push and pull electrodes run underneath the brightground metal at the top of the photograph. The bright part on the bottom left is the

54


other ground line. The region of the contact, tip and pad, is magnified in Fig. 3.21(c).One can clearly see that the contact is open. The height of the cantilever above thepad is approximately 5 µm.

In the small toggle switch of Fig. 3.21(b), the RF signal travels from the lower rightcorner, over the 30 µm long flexible metal band, then over the cantilever, and up to theupper left corner. The airbridge that runs diagonally across the perforated cantileveris attached to the darker SiN isolations on top of the ground metal. The isolated DCleads are located under the cantilever and run under the ground metal. Their contourcan barely be seen. Fig. 3.21(d) and (e) are magnified views of the cantilever tip ontop of the contact pad and the flexible metal band, respectively.

−50

−40

−30

−20

−10

0

0 5 10 15 20 25 30 35 40

0

−1

−2

isola

tion,

retu

rn loss [

dB

]

insert

ion loss,

reflection

[d

B]

frequency [GHz]


isolationreflection

(a) Regular toggle switch

−50

−40

−30

−20

−10

0

0 10 20 30 40 50

0

−1

−2

isola

tion,

retu

rn loss [

dB

]

insert

ion loss,

reflection

[d

B]

frequency [GHz]


isolationreflection

(b) Small toggle switch

Fig. 3.22: S-parameter measurement results of the two different toggle switches de-picted in Fig. 3.21.

The S-parameter measurement results for both toggle sizes in the “on” and “off” statesare given in Fig. 3.22. The regular toggle switch in Fig. 3.22(a) can be activated witha voltage of 10 V. In this closed state, the insertion loss is below 0.8 dB up to 40 GHzand the return loss is greater than 22 dB up to 35 GHz. Due to the high residualstress in the cantilever, the contact opens when reducing the voltage to 0 V. Withan additional voltage at the push electrode, the distance between the cantilever andcontact paddle is increased. In this open position, the isolation decreases from above50 dB at 1 GHz down to 13 dB at 40 GHz. The reflection between 0.2 dB and 1.1 dBindicates that most of the power is reflected at the cantilever in the open state.

For the small toggle switch in Fig. 3.22(b), a voltage between 8 V and 15 V is neededfor actuation. In the closed state, the insertion loss increases almost linearly from 0.4dB at 1 GHz up to 1.2 dB at 50 GHz. The return loss of the small toggle switchdecreases slightly more when compared to the regular toggle switch, from above 30 dB

55


at 1 GHz down to 15 dB at 50 GHz. The cantilever opens when reducing the actuationvoltage to 0 V. In the open state, the isolation is comparable to that of the regulartoggle switch and decreases from above 50 dB at 1 GHz down to 16 dB at 50 GHz.The reflection indicates that there is slightly more scattering at frequencies over 30GHz with a maximum of 1.7 dB at 42 GHz. More measurement results of the toggleswitch can be found in the literature [98].

3.6.3 Experimental Results of the SPDT Switch

(a) SPDT switch composed of two regularsized toggle switches

(b) Magnified view of the center of the SPDTswitch from (a)

(c) Magnified view of the contact paddlesfrom (b) shows an open (right) and aclosed contact (top)

(d) SPDT switch made of two small toggleswitches with ground airbridges crossingover the cantilevers

Fig. 3.23: SEM micrograph of the SPDT switch.

Two different versions of the SPDT switches are fabricated. SEM micrographs ofdifferent views are shown in Fig. 3.23. The SPDT switch design in Fig. 3.23(a) is

56


derived from the single toggle switch. Two of these toggle switches can be seen in thecenter where the paths cross. The input port (port 1) is on the right. The signal cantravel straight to the left (port 2) or around 90 degrees (port 3) depending on whichtoggle cantilever is activated. Additional shunt airbridge switches can be seen at bothoutput ports. When activated, they further increase the isolation to that port. Onecan clearly see the SiN isolation under the shunt airbridge switches and the DC leadson top of the ground lines. A magnified view of the two toggle cantilevers can be seenin Fig. 3.23(b): port 1 is on the top, port 2 on the right, and port 3 on the bottom.The large pull electrodes can be seen under the front of both cantilevers. They havea SiN isolation cover, and their DC leads run underneath the ground lines. The pushelectrodes on the back of the cantilevers are smaller but constructed identically to thepull electrodes. More details of this structure can be seen by looking closely on thecontact region of another SPDT switch (Fig. 3.23(c)). The toggle switch at port 3 onthe right has an open contact, whereas the switch at port 2 is closed. In the SEMmicrograph, the contact is not closed by an activation voltage but due to a problemduring fabrication (final photo resist removal process). However, it nicely illustratesan open and closed contact under magnification. The airbridge that runs across theground metal is isolated by SiN and serves as an upper stop bar for the cantilever.

Fig. 3.23(d) shows an SDPT switch that is composed of two small toggle switches.The input port 1 is on the top left, port 2 on the upper right, and port 3 on thelower right. Similarities to the single small toggle switch in Fig. 3.21(b) can be seen.However, this version has no perforation holes. As for all small toggle switches, thesuspension of the cantilever is mounted on separate poles instead of onto the groundmetal. The upper stop bar for the cantilever has on ohmic contact with the groundbecause this suppresses higher order modes when the signal is routed around the 90degree bend.

S-parameter results of the small toggle SPDT switch are depicted in Fig. 3.24. Amisalignment in the SiN isolation layer occurred during the fabrication process. Thiscaused a short circuit in the DC leads of the electrodes for both the cantilever andthe membrane. In this case, application of a voltage to either one of the electrodesactivates the toggle cantilever and shunt airbridge switch simultaneously. The addi-tional activated shunt airbridge switch in the signal path increases the capacitanceand reduces the performance of the SPDT switch, especially at higher frequencies.Therefore, an adapted S-parameter simulation is performed in order to compare themeasured results with simulations.

Due to the usage of a frequency dependent loss model, the simulation results andthe corresponding measurement results in Fig. 3.24 show very good agreement. Fig.3.24(a) and (b) show the results for the signal routed from port 1 to port 2. Fig.3.24(c) and (d) show the results for the signal routed from port 1 to port 3. Even inthis imperfect configuration for both signal paths, the insertion loss is below 0.6 dB

57


0

−10

−20

−30

−40

−50 30 20 10 0

0

−0.5

−1

−1.5

−2

−2.5

retu

rn loss,

iso

latio

n [

dB

]

insert

ion

lo

ss [

dB

]

frequency [GHz]


isolation (port 3)

(a) Signal route from port 1 to port 2 (simulation)

0

−10

−20

−30

−40

−50 30 20 10 0

0

−0.5

−1

−1.5

−2

−2.5

retu

rn loss,

iso

latio

n [

dB

]

insert

ion

lo

ss [

dB

]

frequency [GHz]


isolation (port 3)

(b) Signal route from port 1 to port 2 (measure-ment)

0

−10

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−40

−50 30 20 10 0

0

−0.5

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−1.5

−2

−2.5

retu

rn loss,

isola

tion

[d

B]

insert

ion loss [

dB

]

frequency [GHz]


isolation (port 2)

(c) Signal route from port 1 to port 3 (simulation)

0

−10

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−40

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0

−0.5

−1

−1.5

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−2.5

retu

rn loss,

isola

tion

[d

B]

insert

ion loss [

dB

]

frequency [GHz]


isolation (port 2)

(d) Signal route from port 1 to port 3 (measure-ment)

Fig. 3.24: S-parameter measurement together with adapted simulation results forSPDT switch.

up to 30 GHz. The isolation to the non-switched port in both routing states is above20 dB at 30 GHz and above 30 dB at 10 GHz. If the shunt airbridge switch in theisolated path is closed, the isolation increases by the values measured from the singleshunt airbridge switch (Fig. 3.20)[14]. The return loss in the closed position is above17 dB up to 30 GHz for both routing states.

The S-parameter performance of the small SPDT toggle switch with regards to returnloss, isolation, and insertion loss is strongly related to the single (SPST) small toggleswitch. This is because the same dimensions and geometry of the small toggle switchwere used for both structures. The only modification in the SPDT is a slightly longerinductive matching line due to the additional airbridge that connects the ground met-alization.

58


A detailed discussion of the performance of the combination of the toggle switch withthe shunt airbridge switch can be found in the literature [99]. The results show betterisolation at higher frequencies when compared to a single toggle switch.

For the system level simulation of the multi-band reconfigurable six-port receiver, themeasurement results of the small SPDT switch depicted in Fig. 3.24 are used.

3.6.4 Experimental Results of the RF Cross

The main difficulties in the fabrication of the RF cross are the construction of the rel-atively large airbridge faces, especially the removal of the sacrificial photo resist layerunderneath them. Despite this fact, the SEM micrographs in Fig. 3.25 show nearlyperfect flat airbridge faces with only slight undulation. No switching or moving partsare involved in this structure.

(a) Upper shielding and ground lines. Theports are towards the corner of the pho-tograph

(b) Magnified view of the region where the sig-nal line goes onto the cross signal airbridge

Fig. 3.25: SEM micrograph of the RF cross.

The depicted RF cross in Fig. 3.25(a) corresponds to the schematic illustration of Fig.3.16: port 1 is on the lower left, port 2 on the upper left, port 3 on the upper right,and port 4 on the lower right. The signal from port 1 to port 3 travels underneath theupper shielding airbridge before it moves onto the cross signal airbridge in the center.It again travels underneath an upper shielding airbridge as it moves towards port 2.In the other signal path, the signal travels from port 2 to port 4 underneath a longerupper shielding airbridge, then underneath the cross signal airbridge in the center, andfinally underneath the upper shielding airbridge to the lower right corner before it getsto port 4. Fig. 3.16(b) shows a blow-up view of the region where the signal line leads

59


onto the cross signal airbridge. In the SEM micrograph, the lower shielding airbridgecan be seen underneath the cross signal airbridge. A small part of the signal line fromport 2 to port 4 appears at the top of the SEM micrograph running underneath thelower shielding airbridge. The large bright areas at the top left and at the top rightare the upper shielding airbridges. The bright areas on bottom left and bottom rightcorner of the picture are the ground metals.

The S-parameter measurement results of the RF cross given in Fig. 3.26 show goodagreement with the simulation results of Fig. 3.17. The signal route over the airbridgefrom port 1 to port 3 shows an insertion loss smaller than 1 dB for frequencies upto 30 GHz, and an insertion loss smaller than 1.5 dB for frequencies up to 40 GHz(Fig. 3.26(a)). The return loss is greater than 20 dB for frequencies up to 30 GHzand decreases towards 40 GHz. The isolation to port 2 is greater than 28 dB up to40 GHz. The signal path from port 2 to port 4 shows a slightly higher insertion loss,but still below 1.5 dB up to 37 GHz (Fig. 3.26(b)). The return loss is above 17 dB upto 30 GHz. The isolation between the signal lines is almost identical for both routes.For the signal route from port 2 to port 4, an isolation above 27 dB up to 40 GHz ismeasured at port 3.

0

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0

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−1

−1.5

−2

−2.5

retu

rn loss,

isola

tion [

dB

]

insert

ion loss [

dB

]

frequency [GHz]


isolation (port 2)

(a) Signal routed over airbridge (port 1 to 3)

0

−10

−20

−30

−40

−50 40 30 20 10 0

0

−0.5

−1

−1.5

−2

−2.5

retu

rn loss,

isola

tion [

dB

]

insert

ion loss [

dB

]

frequency [GHz]


isolation (port 3)

(b) Signal routed along underpath (port 2 to 4)

Fig. 3.26: S-parameter measurement result of the RF cross for the two differentsignal routes (representing the symmetric routing combinations).

For the system level simulation of the multi-band reconfigurable six-port receiver, themeasurement results of the RF cross depicted in Fig. 3.26 are used.

3.7 Additional Measurements and Reliability Results

In the following, additional measurements will be presented to characterize the perfor-mance of the MEMS in more detail. The measurements include switching time, switch

60


cycles, RF power performance, and contact resistance characterization

3.7.1 Switching Time Measurement Results

A laser Doppler vibrometer (Polytec SMV300) was used to measure the cantileverdisplacement and the switching time of the toggle and shunt airbridge switch. Themeasurement is based on detection of the Doppler shift of coherent laser light thatis scattered from a small area of the test object. The object reflects the light fromthe laser beam and the Doppler frequency shift is used to measure the velocity of thecantilever along the axis of the laser beam. A toggle switch with a gap of 10 µmbetween the cantilever tip and the contact paddle is used. The advantage of usingsuch a wide open structure is good optical control over the switch cycle.

A B

(a) Toggle in open state

A B

(b) Toggle in closed state

A

(c) Shunt airbridge switch

Fig. 3.27: SEM micrograph of the toggle and membrane switches. The laser marks(A and B) of the vibrometer for the displacement measurements areshown.

Fig. 3.27(a) and (b) indicate the laser marks of the measurement. Due to the gapof 10 µm, a pull-down voltage of 25 V is required. The measured switching time atpoint A for first contact (first minimum) is 170 µs. A quiet state is attained after300 µs (Fig. 3.28(a)). When reducing the pull-down voltage to 0 V, the cantileverjumps back to its initial position. The release time to the first maximum is 200 µs,and damped oscillations occur up to approximately 4 ms before reaching a quiet state(Fig. 3.28(a)).

The relatively large actuation voltage of 25 V leads to a distortion of the cantileversuch that, at point B, it touches the electrode upon activation (Fig. 3.29(a)). Bothactivation and release lead to damped oscillations. Because the cantilever is bent up,a displacement of approximately 4 µm at point B corresponds to a displacement of 10µm at point A. Both events need approximately 4 ms to quiet down (Fig. 3.29(b)).

Fig 3.30 shows the switching velocity of a membrane with gold metalization from theup-state to the down-state measured in the center of the membrane as indicated in

61


0

−5

−10

400 300 200 100 0

dis

pla

ce

me

nt

[µm

]

time [µs]

170 µs

(a) Toggle cantilever displacement at point A af-ter activation (0 V to 25 V)

5

0

−5

−10

4000 3000 2000 1000 0

dis

pla

ce

me

nt

[µm

]

time [µs]

200 µs

(b) Toggle cantilever displacement at point A af-ter release (25 V to 0 V)

Fig. 3.28: Displacement of the toggle cantilever during activation and release.(Laser position A is depicted in Fig. 3.27(a) and (b)).

0

−2

−4

−6

−8 4000 3000 2000 1000 0

dis

pla

cem

ent

[µm

]

time [µs]

(a) Toggle cantilever displacement at point B af-ter activation (0V to 25 V)

5

0

−5 4000 3000 2000 1000 0

dis

pla

cem

ent

[µm

]

time [µs]

(b) Toggle cantilever displacement at point B af-ter release (25 V)

Fig. 3.29: Displacement of the toggle cantilever during activation and release.(Laser position B is depicted in Fig. 3.27(a) and (b)).

Fig. 3.27(c). Switching time (neglecting the bounce) is approximately 11 µs. Thecorresponding release (excluding oscillations) is in the order of 17 µs. The releaseprocess is completely independent of the actuation voltage that was originally appliedin order to pull the switch down and, instead, is solely dominated by the tensile force ofthe membrane. This, of course, is only valid for an ideal dielectric layer which does notexhibit any charging effects [105]. The results are in agreement with the theoreticalcalculations and also with published data of similar switches [106].

In addition to DC switching, the RF switching behavior of the toggle switch is mea-sured with the setup shown in Fig. 3.31. A 1 GHz RF signal with constant power isapplied to one port of the toggle switch and measured by a peak power meter at the

62


20

15

10

5

0

−5

−10

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−20 40 20 11 0

ve

locity [

mm

/s]

time [µs]

(a) Membrane activation

20

15

10

5

0

−5

−10

−15

−20 60 40 17 0

ve

locity [

mm

/s]

time [µs]

(b) Membrane release

Fig. 3.30: Membrane velocities of the shunt airbridge switch during activation andrelease at point A indicated in Fig. 3.27(c).

RF source bias-T P PMbias-TP

DC PMP

pulse generator circuit

RF probes

toggle switch (DUT)

peak power meter

Fig. 3.31: The measurement setup for the RF switching time measurement.

other port. A rectangular DC voltage signal with a rise time of 5 µs is applied with atransistor circuit. The capacitance in the bias-T increases the signal rise time to about150 µs. The switching time measurement results of the single toggle switch for closingand for release are shown in Fig. 3.32. The distance between the cantilever tip andthe contact pad is approximately 3 µm in the open state. A voltage of 20 V is neededin the DC case to close the switch. During this switch time evaluation, slightly higherDC voltages between -25 V and +3 V were applied by the pulse generator circuit.The time to the first contact of the membrane is 12 µs and a stable state is achieveddirectly without any oscillations. If the switch is opened, a stable state is achieved forall switch cycles after 28 µs. These values are consistent with other measurements ontoggle switches.

3.7.2 RF Power Measurement

RF power measurements are performed on two fabricated shunt airbridge switches andtwo single toggle switches in order to investigate power handling capabilities. Powerhandling of the shunt airbridge switch is limited by either excessive heating due tohigh current densities on the transmission lines, or by the actuation of the membrane

63


0

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−30 100 50 12 0

0

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−40

−50

vo

lta

ge

[V

]

pow

er

[dB

m]

time [µs]

voltagepower

(a) Activating the toggle cantilever

0

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−30 100 50 28 0

0

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−50

vo

lta

ge

[V

]

pow

er

[dB

m]

time [µs]

voltagepower

(b) Releasing the toggle cantilever

Fig. 3.32: Switching times for RF power of the toggle switch.

due to a high average voltage between the signal line and membrane denoted as “self-biasing” [86]. The electrostatic force acting on the membrane can be derived fromeither negative or positive voltages. Since the relaxation time of free electrons inmetals is in the range of 10−14 s (Au: τAu = 2.9 ×10−14 s, Al: τAu = 0.8 ×10−14 s)which corresponds to frequencies of 100 THz, the electrostatic force instantaneouslyfollows the applied high frequency electromagnetic field. Therefore, the average voltagelevel of the rectified sine wave on the CPW line attracts the membrane. The RF poweris related to this average voltage by the following simple equation:

U0 =√

2 · Ueff =√

2RP =√

2 · 50Ω · 10W = 31V. (3.30)

It can be seen that in a 50 Ω CPW environment, a power of 10 W results in an aver-age voltage of 31 V. To allow higher power handling, the switch needs to be designedfor higher actuation voltages (i.e. 25 W requires an actuation voltage larger than 50 V).

Characterization of power handling requires relatively expensive measurement equip-ment, in particular a high power RF generator. This is why there is not much powermeasurement data available in the literature. For the power measurements, a HP83650 synthesizer and a type TWT 8010H15F00 amplifier was used. The cascadeprobe station was equipped with Picoprobe probes. The signal was attenuated with20 dB attenuators before being detected by a HP 438A power sensor connected to aHP 8487A power meter. For the shunt airbridge switch, an RF power up to 1 W at 30GHz is applied. Fig. 3.33(a) shows that self-actuation on a suboptimal switch occursat 0.96 W, whereas an optimized switch shows no self-actuation at 1 W. Higher powersat 30 GHz could not be applied due to limitations of the measurement equipment. Forthe toggle switch, an RF power up to 2 W was applied at frequencies up to 18 GHz.Fig. 3.33(b) shows that no power induced self actuation is found for the regular orfor the small toggle switch. However, the curves show the expected, and previously

64


0

−5

−10 30 20 10

Po

ut/ P

in [

dB

]

input power Pin [dBm]

non−optimizedoptimized

(a) Non-optimized (self actuation at 0.96 Wat 30 GHz) and optimized shunt airbridgeswitch (no self-actuation up to 1 W)

−15

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−25

−30

−35 25 20 15 10 5

isola

tio

n [

dB

]

frequency [GHz]

regular toggle cantileversmall toggle cantilever

(b) Regular and small toggle switch (2 W RFpower applied)

Fig. 3.33: Self actuation of the toggle and shunt airbridge switches due to RF power.

discussed, decrease in isolation at higher frequencies.

Detailed investigations on the toggle switch were performed by IMST GmbH. It wasdiscovered that switches with higher actuation voltages can handle higher RF powersbefore latching. This confirms Eqn. 3.30. The highest power handling capability wasfound from a toggle switch with an actuation voltage of 50 V. The cantilever could beswitched several times at a power of 2.51 W at 5 GHz with an on-state signal time of10 s. Four of six investigated switches could handle powers between 1 W and 2.5 Win the frequency range from 5 GHz and 15 GHz. The power measurement results arein good agreement with results reported in the literature [100][101][102][103].

3.7.3 Switch Cycle Measurement Results

Switch cycle measurements on a fabricated single toggle switch were performed toinvestigate degradation of the membrane, the contact paddle, and the mounting sus-pensions. The DC measurement set up depicted in Fig. 3.34 was used.

When the contact is closed by applying Vact to the DC lead, there will be a voltageV1 between the two probes on the signal line. To avoid degradation of the contactdue to a high current, the measurement current Imeas is limited 0.1 mA. A slightlyincreased actuation voltage of 25 V is applied for the switch cycle measurement. Themeasurement is automated using HP-VEE. The Keithley voltage source Vact is alter-nated between 0 V and 25 V with a period of 2 s. Unfortunately, the flexible metalband which connected the cantilever to the signal line became ripped off after 2.4×105

65


ImeasVact

V3V2

V1

V V

V

contacts of probes

additional contacts for voltage measurement

switch

Fig. 3.34: Experimental set up for switch cycle measurements to determine contactdegradation and contact resistance.

successful switching cycles. This indicates that the size of the flexible metal band,especially the area which connects it to the signal line, is too small.

A capacitance measurement is used to count the number of successful switch cycles ofthe shunt airbridge switch. With a distinct change in the capacitance, the membraneis successfully actuated more than 5×105 times. The measured capacitance in theactuated state is 7 pF, while in the relaxed or neutral position, the capacitance isat least a factor of 1000 below this value. Reported lifetimes for capacitive switchesfound in the literature range from 104 to 108 switch actuations, demonstrating anexponential relationship between lifetime and actuation voltage [105].

3.7.4 DC Contact Resistance

Three different test series are performed to extract the contact resistance of a singletoggle switch. Each tests includes 20 current-voltage measurements at the pointsindicated in Fig. 3.34. A linear fit was used to find the related resistance:

V1: measurement lines and probes + toggle cantilever + contact → 12.7 Ω

V2: measurement lines and probes + toggle cantilever → 3.9 Ω

V3: measurement lines and probes → 2.6 Ω

The resistance of the toggle cantilever is given by R2 - R3 = 1.3 Ω. The extractedpure contact resistance (RC) is given by R1 - R2 = 8.8 Ω. The equivalent insertionloss in a 50 Ω environment (ZL) at DC is:

S12 =2

2 + RC

ZL

= 0.36 dB. (3.31)

This value is slightly higher than the contact resistance of 2.1 Ω calculated fromthe insertion loss of 0.2 dB that was found from S-parameter measurements at lowfrequencies. The switch under investigation here is different from the one that wasused for S-parameter measurements. This reveals variations in the contact resistanceof different switches due to fabrication.

66


3.7.5 Temperature Dependency and Reliability

Temperature measurements of two other fabricated single toggle switches were per-formed to investigate temperature dependency of the actuation voltage and contactresistance using the setup from Fig. 3.34.

100

90

80

70

60

50 100 80 60 40 20

20

15

10

5

0

actu

atio

n v

olta

ge

V [

V]

co

nta

ct

resis

tan

ce

R [

Ω]

temperature [oC]

heating cyc. Vheating cyc. Rcooling cyc. Vcooling cyc. R

(a) Change in the resistance

40

35

30

25 80 70 60 50 40 30 20

25

20

15

10

5

0

actu

atio

n v

olta

ge

V [

V]

co

nta

ct

resis

tan

ce

R [

Ω]

temperature [oC]

heating cyc. Vheating cyc. R

(b) Keeping the resistance constant

Fig. 3.35: Temperature dependency of the actuation voltage of a toggle switch. Theresistance is dependent upon the force applied by the actuation voltage,even at constant temperature.

The results of a full heating and subsequent cooling cycle are shown in Fig. 3.35.The measurement points show the reversibility of the process. The wafer sits on athermo-chuck during this on-wafer measurement. Only higher temperatures up to100C are applied due to condensation at room temperature (20C). Two differenttoggle switches with respect to actuation voltage were chosen: one with a relativelyhigh actuation voltage of 83 V at room temperature (Fig. 3.35(a)), and one with anactuation voltage of 39 V at room temperature (Fig. 3.35(b)). Resistances found werein the range of 5 Ω to 16.5 Ω. This is in good agreement with the detailed contactresistance investigation above, even though the resistances here are not de-embeddedand include all resistances: the probes, the measurement lines, and the signal line.However, it shows that the contact resistance of a toggle switch is strongly dependenton temperature. While the cantilever metal gets softer at higher temperatures leadingto lower actuation voltages, higher temperatures also lead to higher resistances in themetal of the signal line. In addition, contact resistances vary greatly from switch toswitch.

For the toggle switch with a gold cantilever in Fig. 3.35(a), a linear fit results in agradient of -0.4 V/ C. The relatively large variation in the resistance – values between5 Ω and 13 Ω – come from the characteristic switching behavior at different temper-

67


atures. At each temperature under investigation, the actuation voltage which resultsin a clear drop in the voltage V1 (see Fig. 3.34) is shown. Higher actuation voltages atthe same temperature can always decrease the contact resistance by the higher forcethat is applied.

For the second toggle switch under investigation (depicted in Fig. 3.35(b)), the actu-ation voltage is chosen in such a way that it leads to the same contact resistance ofapproximately 16 Ω over the entire temperature range from 20C to 70C. Actuationvoltages for this switch vary from 26 V to 39 V resulting in a gradient of -0.3 V/ Con a linear fit. The lower actuation voltages of this switch come from a smaller stressgradient in the Au/Ni/Au-cantilever compound. Any gradient leads to a warping ofthe cantilever. The higher the stress gradient, the larger the distance from the can-tilever to the electrode, and therefore, the higher the actuation voltage. It is obviousthat this stress gradient also influences the temperature dependency of the actuationvoltage as the warping of the cantilever changes due to temperature. The stress gra-dient is dependent on sputtering process parameters and also on the used materials.The process parameters that control the stress gradient are: gas pressure, the sputtertime, and power. It is expected that the invar compound – a nickel-iron alloy with ≈35% Ni – can reduce this temperature dependency.

A dice was mounted onto a carrier to evaluate the temperature dependency of theactuation voltage of the shunt airbridge switch [86]. SPST shunt airbridge switchesare contacted with bond wires as depicted in Fig. 3.36(a). This allows measurementsin a low humidity chamber below room temperature. The results from 5 subsequentheating and cooling cycles are shown in Fig. 3.36. A linear dependency of the actuationon temperature can be seen in all cycles. The maximum actuation voltage of 44 V isfound at -30C, and the minimum actuation voltage of 14 V is found at +80C.

(a) dice on carrier

50

40

30

20

10

80 60 40 20 0−20

actu

ation v

oltage V

[V

]

temperature [oC]

heating cyc. 1heating cyc. 2heating cyc. 3heating cyc. 4heating cyc. 5

(b) Activation voltage vs. temperature

Fig. 3.36: Temp. dependency of the activation voltage of a shunt airbridge switch.

68

4 The MEMS-Based Multi-BandSix-Port Circuit

In this chapter it will be shown that recent developments in MEMS technology are verywell suited to design a multi-band, multi-standard transceiver. An application scenariowith the MEMS discussed in Chap. 3 will show their typical use in a multi-port RFreceiver front-end. In the receiving path, these MEMS SPDT switches are very wellsuited to serve as antenna diplexers which can accomplish a switching between the RXand TX path. Furthermore, these low loss MEMS routing elements allow switching theRF signal between two different six-port circuit, each operating at different frequencies– one around 1.5 GHz and the other from 2 GHz to 25 GHz. To understand thefunctional principle of RF multi-band interferometers in general, the design processand measurement results of the 1.5 GHz six-port interferometer (SP1500) will bediscussed in the beginning. Subsequent to the discussion of the performance of thesecond (broadband) six-port circuit, the complete RF MEMS-based re-configurablemulti-band receiver front-end is elaborated on.

4.1 Introduction to Passive RF Multi-Port

Interferometers

The first reported six-port circuits were used as alternative network analyzers for themeasurement of complex scattering parameters [15][34][17][16][18]. In such reflectome-ter applications the six-port circuit has a small isolation between the LO and the DUT(device under test) port. Whereas in a typical receiver application, the multi-port cir-cuit requires a high isolation between the RF and LO port. Besides low attenuation,one of the key requirement on the passive interferometer circuit of multi-port receiversis the phase difference between the LO and RF signal at the output port. Optimalphase differences are in multiples of 90 at the output ports. Depending on the circuitdesign, this can be achieved over a relatively large frequency range, and a bandwidth ofmany octaves is possible. However, modern broadcasting and high speed communica-tions bands start in the upper MHz range (FM radio at around 100 MHz) and extendto GHz frequencies (802.11a at around 5.35 GHz). Future high data rate standardswill make use of the industrial, scientific, and medical (ISM) bands at higher carrierfrequencies where extremely large bandwidths are available, extending the demand of

69

4 The MEMS-Based Multi-Band Six-Port Circuit

the front-end up to 24 GHz, 37 GHz, 60 GHz, or even higher (i.e. WIGWAM project[128] and WiMax [129]). Such large frequency differences cannot be covered by asingle passive circuit, and new systems have to be invented that utilize cost-effectivereconfigurable multi-standard front-ends which serve a large frequency range.

In the following, different multi-port realization schemes are sketched and discussed.A simple, broadband six-port interferometer consists of a power divider and three 90

hybrid couplers. This six-port interferometer is looked at in detail showing its designand simulation results. Measurement results of the same six-port architecture, thatis composed of composed of commercial blocks, are presented hereafter. This demon-strates the possibility to design a broadband system.

In general, various designs for the multi-port front-end are possible. The circuit shouldhave as little attenuation as possible and achieve multiples of 90 phase shifts atthe output ports. Additionally, the power from both, RF and LO port, should beequally distributed at the output ports. One distinctive feature is how the phaseshift is achieved. At higher frequencies (i.e. shorter wavelengths) simple delay linesare advantageous; whereas at lower frequencies (i.e. larger wavelengths), these delaylines get too long and cannot be practically implemented. Depending on the type ofsubstrate, the critical frequency for delay lines is around 2 GHz (i.e. a 90 phase shiftin a 50 Ω line requires a length of approximately 1.5 cm on a substrate with a dielectricconstant εr of 9.8). This fact makes the multi-port technology especially qualified forextremely high frequencies. Six-port circuits have been reported up to a frequencyof 94 GHz for radar applications [107]. For frequencies in the lower MHz range, itis advantageous to reduce size and cost by using lumped elements for the phase shift[108]. However, this reduces the bandwidth and increases the loss of the circuit. The1.5 GHz six-port demonstrator that is investigated in this chapter consists of delaybased couplers and a delay based power divider.

4.2 Options for the Multi-Port Architecture

4.2.1 The N-Port Interferometer

Fig. 4.1(a) shows the basic principle of multi-port circuits. The RF and LO signalsare split at the input ports, and the superposed signals are combined at output ports3 through 6. If the bandwidth is not an issue, this design can be used to combine RFand LO signal under any phase angle. Depending on the number of output ports, theRF and LO signal power, PRF and PLO, decreases due to the inherent splitting loss ofthe signal power (see Fig. 4.1(b)). This distribution or splitting loss AP is given by

AP [dB] = 10 · log10 Pn/PRF,LO (4.1)

70

4.2 Options for the Multi-Port Architecture

n-w

ay p

ower

div

ider

n-w

ay p

ower

div

ider

i=3

LO (i=1) RF (i=2)

n

i-2 λn-2

i=4

i=5

equal power combiner

(a) Simple multi-port design with two inputports (LO(1), RF(2)) and n output ports(3..n)

-3

-6

-9

-12

-15 20 15 10 6 5 4 3

split

ting loss [dB

]

total number of output ports [GHz]

(b) Theoretical loss due to power distributionas a function of the total number of outputports

Fig. 4.1: An n-port interferometer is a passive circuit that superposes an LO signalfrom port 1 and an RF signal from port 2 under different phase angles.

where Pn is the power at any output port n. For example, in a six-port circuit forexample, the best theoretically achievable attenuation is 6 dB. Note that in the graphs,the losses (as well as attenuation) are given by negative numbers. Also note, thatthe text uses the terminology from Chap. 3 with positive dB values for losses andattenuation. The voltage loss AV due to signal splitting is

AV [dB] = 20 · log10 Vn/VRF,LO. (4.2)

Interferometer circuits, based on the design of Fig. 4.1, that probe the superposedsignals on delay lines have a rather small bandwidth. To achieve a larger bandwidth,the designs need to utilize two arm – or even multiple arm – branch-line couplers (alsoreferred to as quadrature hybrids).

4.2.2 Five-Port and Six-Port Interferometers

Various designs for five- and six-port circuits can be found in the literature which rangefrom microstrips [12] to LC lumped element designs for different operating frequencies[108]. Simple structures with a sampled delay line are reported [109]. Others report ona six-port interferometer that is composed of four quadrature hybrids [110] (see 4.2(d)).

Fig. 4.2(c) shows the layout of a broadband six-port circuit made from three standardquadrature hybrid couplers and one power divider. One of the two requirements is thatpower is split equally to the output ports for signals that originate from both inputports. Uniform phase differences (multiples of 90) of the superpositions of the signalsfrom the LO and RF port are the second requirement. The latter can be achieved evenif the quadrature hybrids do not have a perfect 90 phase shift. This is because the

71


lines connecting the hybrids also account for a phase shift which can compensate forthe derivations of the quadrature hybrids. However, as mentioned earlier, using suchadditional delay lines for phase shifting reduces the bandwidth. An advantage of multi-port receiver is its calibration which can correct these imbalances up to a certain point.

90o

RFjLO

22+

3

4

RF (2)

LO (1)

5

LOjRF

22+

)(2

1RFLO +

(a) Five-Port Circuit consisting of 1 quadra-ture hybrid

90o90o

RFjLO

22+

3

4

RF (2)

LO (1)

6

5

LOjRF

22+)(

2RFLO

j +

)(2

1RFLO −

90o

(b) Six-Port Circuit consisting of 2 quadra-ture hybrids

90o

90o

90o

RFjLO

22+

3

4

RF (2)

LO (1)

6

5LO

jRF

22+)(

2RFLO

j +

)(2

1RFLO −

(c) Six-Port Circuit consisting of 3 quadraturehybrids

90o90o

RFjLO

22+

3

4

RF (2)

LO (1)

6

5

LOjRF

22+)(

2

1RFLO +−

)(2

RFLOj −

90o

LO (1)

90o

90o

(d) Six-Port Circuit consisting of 4 quadra-ture hybrids [110]

Fig. 4.2: Possible designs for five- and six-port interferometer circuits.

Fig. 4.2 shows various designs for five-port and six-port circuit interferometers. Inthe designs, the power dividers are subsequently replaced by quadrature hybrids. The90 phase shift of the quadrature hybrid needs to be adjusted with additional delaylines when using four quadrature hybrids (Fig. 4.2(d)). Using any three output portsof the circuit in Fig. 4.2(c) makes the same circuit a five-port interferometer. Simi-lar five-port designs can be made using directional couplers and phase shifters. Thewhole circuit can even be simplified by constructing a ring with five branches withthe appropriate isolation and transmission. Another intrinsic feature of the multi-portinterferometer is the permutability of the RF and the LO ports.

72

4.3 Design and Analysis of a 1.5 GHz Six-Port Interferometer (SP1500)

4.3 Design and Analysis of a 1.5 GHz Six-Port

Interferometer (SP1500)

In this section, the design, simulation, and measurement results of the six-port inter-ferometer circuit for frequencies around 1.5 GHz (the carrier frequency of the GPS) ispresented. Fig. 4.3 shows the fabricated passive six-port interferometer that will bediscussed in detail.

1

2

3

4

6

7

5

Fig. 4.3: Photograph of SP1500. The RF and LO input ports (1,2) and output ports(3,4,5,6) are marked.

This phase correlator superposes the input signals (the LO signal from port 1 andthe RF signal from port 2) creating four different phase combinations at output ports3 through 6. Every combination is a sum of the input signals with different phaseshifts (0, 90, 180, and 270). The simple design of Fig. 4.2(c) with three quadraturehybrids and one power divider has been chosen because the demand is relaxed on thebandwidth for the 1.5 GHz demonstrator. The phase correlator is implemented using amicrostrip technique. For its design, Advanced Design System (ADS) and Momentumfrom Agilent Technologies is used. Fig. 4.3 shows how the port numbers are assigned.The LO input signal is fed into port 1, and the RF input signal is fed into port 2. Theoutput ports are 3, 4, 5, and 6 whereas the unused port 7 (not shown) is terminatedwith an external 50 Ω resistor.

73


4.3.1 Theoretical Background of the Electromagnetic Simulations

Advanced Design System (ADS) Simulator

The computer-aided design (CAD) software ADS from Agilent Technology is used foranalytical simulation of the microstrip structures. The software is a very useful toolfor fast simulation, analysis, and optimization of planar structures. Microwave cir-cuits consisting of transmission lines, lumped elements, active devices, coupled lines,wave-guides, and other components can be analyzed. The design process begins withcalculations of the targeted impedances of the transmission lines and their correspond-ing lengths and widths on the substrate. The achieved results give first insight intothe S-parameters of the structure. However, for a more accurate and realistic result, asubsequent electromagnetic simulation that takes more parasitic effects into accounthas to follow.

Momentum Simulator

After the ADS simulation of the schematic, the circuit is converted into its physicallayout and simulated with the microwave mode of the Momentum simulator from Agi-lent Technology. This electromagnetic field simulation tool solves Maxwell’s equationsfor the 2.5-dimensional case based on the method of moments. It uses an adaptivefrequency sampling algorithm that allows fast and accurate simulation results. Thissimulator is a standard popular and wide-spread tool. It calculates S-parameters forany arbitrary planar geometrical pattern, and takes into account the electromagneticmechanisms such as:

1. Skin effect

2. Dielectric loss

3. Metalization loss

4. Dispersion

5. Radiation loss

4.3.2 Substrate and Microstrip Lines

The microstrip line is one of the most popular types of planar transmission lines,primarily because it can be fabricated by photolithographic processes and is easilyintegrated with other passive and active microwave devices. The geometry of a mi-crostrip line is shown in Fig. 4.4. A conductor of width W is printed on a thin,grounded dielectric substrate of thickness ds and relative permittivity εr; a sketch of

74


dssubstrate FR4

Cu metalization

W

dc

E-field

H-fieldCu

Fig. 4.4: Electric and magnetic field components in a microstrip line (geometricparameters are indicated.)

the field lines is shown in Fig. 4.4. Microstrip design keeps some of its field lines in thedielectric region (concentrated between the strip conductor and the ground plane) anda certain fraction in the air region above the substrate. For this reason, the microstripline cannot support a pure TEM wave since the phase velocity of TEM fields in the di-electric region is c/

√εr; while the phase velocity of transversal electromagnetic (TEM)

fields in the air region is c. Thus, a phase match at the dielectric-air interface wouldbe impossible to attain for a TEM-type [112].

In reality, the exact fields of a microstrip line constitute a hybrid transversal magnetic(TM)/ transversal electric (TE) wave, and require more advanced analysis techniques.However, in most practical applications, the dielectric substrate is very thin comparedto the wavelength (d << λ) and therefore, the fields are quasi-TEM. The phase velocityvp and propagation constant β can be expressed as

vp =c√εe

(4.3)

β = k0

√εe (4.4)

where εe is the effective dielectric constant of the microstrip line. Since some of thefield lines are in the dielectric region and some are in the air, the dielectric constantsatisfies the relation

1 < εe < εr (4.5)

and is dependent on the substrate thickness ds and conductor width W .

Using the ADS line calculation tool, the values for the microstrip line width and lengthat the desired frequency of 1.5 GHz are calculated with the given substrate thicknessH , the relative dielectric constant, the conductor thickness, and the dielectric losstangent. Tab. 4.1 shows the results for the different impedances of a regular 50 Ω line,a 50·

√2 Ω (70.71 Ω) line (for the Wilkinson power divider), and a 50/

√2 Ω (35.36 Ω)

line (for the quadrature hybrid).

75


application impedance Z 90 line length line width[Ω] [mm] [mm]

regular transmission line Z0=50 41.560 2.873

Wilkinson power divider Zpd=Z0 ·√

2=70.71 42.720 1.500

quadrature hybrid Zpd=Z0/√

2=35.36 40.504 4.923

Tab. 4.1: Widths and lengths of Cu microstrip lines on a FR4 substrate at 1.5GHz (substrate thickness ds=1500 µm, εr=4.3, line thickness 25 µm, Cuconductivity 5.96×107 Ω−1m−1, loss tangent 0.012).

4.3.3 Design and Simulation Results of the Power Divider

An important element in multi-port circuits is the power divider. The theory of theWilkinson power divider is based on even and odd mode analysis. In fact, this theoryshows how the the equal power split of -3 dB from port 1 to port 2 and from port 1 toport 3 can be achieved having all the ports matched [112] while the phases are equalat both output ports

LO (1)

23

100 Ωresistor

λ/4-line(70.71 Ω)

λ/4-line(70.71 Ω)

50 Ω -line

(a) Schematic

3 2

1

(b) Photograph

Fig. 4.5: The dimensions and geometry of the power divider.

Fig. 4.5 shows the layout of the designed, fabricated, and measured Wilkinson powerdivider in its simplest version with an equal amplitude, two way split, and a singlestage. The arms are quarter-wave transformers of impedance

√2 · Z0. As a power

splitter, the Wilkinson power divider works as follows (note that when reversed, theWilkinson power divider becomes a power combiner). When a signal enters port 1,it splits into equal amplitude and equal phase signals at output ports 2 and 3. Sinceeach end of the 100 Ω resistor between port 2 and port 3 is at the same potential,no current flows through it, and therefore, the resistor is decoupled from the input.The two output port terminations will add in parallel at the input, so they must eachbe transformed to 2 · Z0 each at the input port to combine to Z0. The quarter-wavetransformers in each leg accomplish this; without the quarter-wave transformers, the

76


combined impedance of the two outputs at port 1 would be Z0/2. The characteristicimpedance of the quarter-wave lines must be equal to

√2 · Z0 so that the input is

matched when ports 2 and 3 are terminated with Z0. The bandwidth of the Wilkinsonpower divider can be increased by adding more bows [111].

The S-parameter simulation results of the Wilkinson power divider are given in Fig.4.6. At the center frequency of 1.5 GHz, the return loss is above 50 dB and the isolationof port 2 and port 3 is larger than 40 dB. For the transmission, the theoretical powersplit of -3 dB is found at the center frequency which drops to -3.5 dB at 0.5 GHz and2.5 GHz. The phase shift in Fig. 4.6(b) is due to the delay caused by the λ/4-line plusan extra phase shift from the connecting 50 Ω lines.

-50

-40

-30

-20

-10

0

0.5 1 1.5 2 2.5

0

-2

-4

-6

-8

-10

S11,

S23

[d

B]

S12 [d

B]

frequency [GHz]

S12S11S23

(a) Transmission S12, return loss S11, and isola-tion S23

180

90

0

-90

-180

0.5 1 1.5 2 2.5

phase [d

eg

]

frequency [GHz]

S12

(b) Phase shift S12 from port 1 to port 2 (phaseshift S13 is identical)

Fig. 4.6: S-parameter simulation results of the 1.5 GHz power divider.

4.3.4 Design and Simulation Results of the Quadrature Hybrid

The quadrature hybrid, or branchline coupler (as shown in Fig. 4.7), is the simplesttype of quadrature coupler, since the circuitry is entirely planar. This element is theother basic building block in the design of broadband multi-port circuits. As for theWilkinson power divider, its bandwidth can be increased by adding more sections (orside arms) [111]. However, the tradeoff for bandwidth is an additional loss and alarger size of the overall structure. Using ideal transmission line impedances providesan equal power split of -3 dB at the center frequency.

The designed, simulated, fabricated, and measured ideal single box branchline coupleris shown in Fig. 4.7. Each transmission line is a quarter wavelength long. However,3/4, 5/4 or 7/4 wavelengths (etc.) can also be used on each arm if required by thecircuit layout (the tradeoff is a decreased bandwidth). A signal entering the top left

77


1

3

50 Ωline

2

4

λ/4-line(35.36 Ω)

λ/4-line(50 Ω)

(a) Schematic

1 2

34

(b) Photograph

Fig. 4.7: The dimensions and geometry of the quadrature hybrid.

port (port 1 in Fig. 4.7) is split into two quadrature signals on the right (port 2and port 3), with the remaining port 4 fully isolated from the input port at the centerfrequency. Remember that the lower output port (port 3) has the most negative trans-mission phase since it has the farthest path to travel. Using the ideal transmission lineimpedances shown above provides an equal power split of -3 dB at the center frequency.

-50

-40

-30

-20

-10

0

0.5 1 1.5 2 2.5

0

-3

-6

-9

S11,

S14 [

dB

]

S12,

S13 [

dB

]

frequency [GHz]

S12S13S11S14

(a) Transmission S12 and S13, return loss S11,and isolation S14

180

90

0

-90

-180

0.5 1 1.5 2 2.5

phase [deg]

frequency [GHz]

S12S13

S12-S13

(b) Absolute phases S12 and S13, and the phasedifference between port 2 to port 3 (S12-S13)

Fig. 4.8: S-parameter simulation results of the 1.5 GHz quadrature hybrid.

The S-parameter simulation results of this quadrature hybrid are given in Fig. 4.8.At the center frequency of 1.5 GHz, the return loss is above 30 dB and the isolationbetween port 1 and port 4 is larger than 50 dB. For the straight transmission from port1 to port 2 (S12), the theoretical power split of -3 dB is found at the center frequencywith a 1-dB bandwidth of approximately 500 MHz. The second transmission fromport 1 to port 3 (S13) has its -3 dB maximum at a slightly lower frequency and showsa slightly higher 1 dB bandwidth of approximately 700 MHz. The absolute phaseshifts (S12 and S13) in Fig. 4.8(b) are due to the delay caused by the λ/4-line plus anextra phase shift from the connecting 50 Ω lines. Their phase difference (S12 − S13) is

78


shown with a solid line and has its targeted 90 between 1.4 GHz and 1.75 GHz.

4.3.5 Simulation and Measurement Results of SP1500

After the single components (the quadrature hybrid and the power divider) are success-fully simulated and the targeted results are achieved, they can be arranged accordingto the different schemes shown in Fig. 4.2. Because of its simplicity, and the fact thatan additional delay line can be avoided, the arrangement in a six-port interferometer(SP1500) as depicted in Fig. 4.2(c) has been chosen for simulation and fabrication.

LO (1)

RF (2)7

3

45

6

26.416 mm

Fig. 4.9: Momentum layout with dimensions of SP1500.

Fig. 4.9 shows the geometry and dimensions of SP1500. The 35.36 Ω λ/4 arm of thequadrature hybrid is 26.416 mm long. It is interesting to note that the final relativephase differences at output ports 3 through 6 are independent of the length of the50 Ω transmission lines that connect the components. To achieve this, the power di-vider and the quadrature hybrid in the middle need to be centered and symmetric.The isolated port 7 is unused in all applications and terminated with an external 50Ω resistor. This six-port interferometer has been fabricated on FR4 substrate witha standard lithographic process. The accuracy of the transmission lines is ±10 µm.This uncertainty leads to only a minor imperceptible change in the S-parameters.

In the following, the simulated (dashed line) and measured (solid line) S-parameterresults will be presented. All measurement results show very good agreement withthe simulation results. The notation for the S-parameters is Sxy, where x refers toport 1 and y refers to port 2 of the two-port measurement (this is also valid for the

79


S-parameter simulation results). Transmission loss, return loss, and isolation will havenegative dB values in the graphs. In the text, they will be referred to as positivenumbers. This is the same notation that is used in Chap. 3.

Matching at Input Ports

Fig. 4.10 shows the simulated and measured return loss of SP1500. The return lossis of interest at the two input ports, the LO port 1 and the RF port 2. Fig. 4.10(a)shows the matching at the LO input port 1 of the power divider. The simulation andmeasurement results are in agreement and correspond to the simulation results of thesingle power divider from Fig. 4.6(a). The measured return loss S11 is above 20 dBbetween 1.45 GHz and 1.65 GHz, and above 10 dB between 1.3 GHz and 2 GHz. Fig.4.10(b) shows the matching at the RF input port 2 of the quadrature hybrid. Again,simulation and measurement results correspond to the simulation results of the singlequadrature hybrid from Fig. 4.8(a). However, the measured values are slightly shiftedtowards higher frequencies. The measured return loss S22 is above 20 dB between 1.45GHz and 1.65 GHz and above 10 dB between 1.3 GHz and 1.8 GHz. One can seethat the input port of the power divider (port 1) shows slightly better matching. In atypical receiver application it is advantageous to use the input port that has the lowerreturn loss as the RF port.

0

-10

-20

-30

-40

-50 0.5 1 1.5 2 2.5

S11 [

dB

]

frequency [GHz]

measuredsimulated

(a) At the LO input port 1 (power divider)

0

-10

-20

-30

-40

-50 0.5 1 1.5 2 2.5

S22 [

dB

]

frequency [GHz]

measuredsimulated

(b) At the RF input port 2 (quadrature hybrid)

Fig. 4.10: Simulated and measured return loss at the input ports of SP1500.

Isolation

The two input ports and the unused port 7 need to be well isolated so as to avoidwasting power. Fig. 4.11(a) shows an isolation of greater 20 dB from 1.4 GHz through

80


1.65 GHz between the LO port and the RF port. The isolation stays greater than 15dB for frequencies between 1 GHz and 2.5 GHz. Again, there is a good agreementbetween the simulation and the measurement. Due to the symmetry of the structure,the isolation from LO port to port 7 is identical. The isolation between the RF inputport and the unused port is mainly determined by the quadrature hybrid (see Fig.4.8a) – the isolation is larger than 20 dB between 1.4 GHz and 1.65 GHz, and largerthan 10 dB between 1.25 GHz and 1.75 GHz.

0

-10

-20

-30

-40

-50 0.5 1 1.5 2 2.5

S1

2 [

dB

]

frequency [GHz]

measuredsimulated

(a) From the LO port to the RF port

0

-10

-20

-30

-40

-50 0.5 1 1.5 2 2.5

S2

7 [

dB

]

frequency [GHz]

measuredsimulated

(b) From the RF port to port 7

Fig. 4.11: Simulated and measured isolation of SP1500.

Transmission

The purpose of the six-port circuit is to transfer power from the input to the outputport without any loss. However, due to the fact that power needs to be divided andtransfered equally to four different output ports, the signal is attenuated according tothe mechanisms described in Chap. 4.2.1.

Fig. 4.1 indicated earlier (when excluding other losses) that the signals from bothinput ports, LO and RF, are attenuated by 6 dB at the output ports 3 through 6.When looking at the simulation and measurement results of Fig. 4.12, one can seethat measurement results show an attenuation of approximately 7 dB. This 1 dBhigher loss is due to mismatch, substrate losses, and radiation. The transmission lossfrom the LO port to port 3 is 7 dB (Fig. 4.12(a)), and 7.5 dB from the RF port(Fig. 4.12(b)). As one can see, it is challenging to achieve a high bandwidth in thisparameter. For the LO signal, the transmission loss stays around 7 dB from 1.5 GHzto 1.8 GHz but drops abruptly outside of this range. For the RF signal, this behavioris even more exaggerated. Good transmission is found only at a 200 MHz bandwidtharound 1.5 GHz.

81


0

-3

-6

-9

-12

-15

-18 0.5 1 1.5 2 2.5

S1

3 [

dB

]

frequency [GHz]

measuredsimulated

(a) From the LO port to port 3

0

-3

-6

-9

-12

-15

-18 0.5 1 1.5 2 2.5

S2

3 [

dB

]

frequency [GHz]

measuredsimulated

(b) From the RF port to port 3

Fig. 4.12: Simulated and measured transmission of SP1500 from the LO and RFports to port 3.

0

-3

-6

-9

-12

-15

-18 0.5 1 1.5 2 2.5

S13,S

14,S

15,S

16 [

dB

]

frequency [GHz]

S13S14S15S16

(a) From the LO port 1

0

-3

-6

-9

-12

-15

-18 0.5 1 1.5 2 2.5

S23,S

24,S

25,S

26 [

dB

]

frequency [GHz]

S23S24S25S26

(b) From the RF port 2

Fig. 4.13: Measured transmission of SP1500 from the LO and RF ports to all outputports.

Fig. 4.13 shows the transmission from the LO input port and the RF input port to alloutput ports. Due to the symmetry of the structure, there are two pairs with equaltransmission originating from the LO input port (Fig. 4.13(a)). The outer transmis-sions S13 and S16 (as well as the inner transmissions S23 and S26) are equal. The innertransmissions show a slightly smaller frequency dependency with an attenuation above8 dB from 0.7 GHz to 2.2 GHz. As the structure is not symmtric for the signals fromthe RF port, there are four different transmission curves S2y(Fig. 4.13(b)). In addition,the frequency dependency is much stronger with an approximate transmission loss of7 dB at 1.55 GHz for all curves.

82


Phase Shifts

To achieve the best possible reception quality, the phase difference at the output portsis the second most important specification for a multi-port receiver. We have seen thatthe design goals is an uniform phase shift between LO and RF signals in multiples of90 at output ports 3 through 6.

180

0

-180

-360

-540

-720

0.5 1 1.5 2 2.5

ph

ase

[d

eg

]

frequency [GHz]

(a) From the LO port to port 3 (measured)

-180

-360

-540

-720 0.5 1 1.5 2 2.5

ph

ase

[d

eg

]

frequency [GHz]

S13 measuredS13 simulatedS23 measuredS23 simulated

(b) From the LO and RF port to port 3

Fig. 4.14: Simulated and measured absolute phase shifts between the LO and RFports and port 3. Note that the original phase transition in (a) from-180to +180has been removed for better illustration and easier expla-nation.

In all following phase relation graphs, the simulated and measured phases will bedrawn as explained in Fig. 4.14(a). For a clearer illustration, the phases that varybetween -180 and +180 are transformed to achieve a single continuous line. Fig.4.14(a) shows the absolute measured phase shift from the LO input port 1 to the out-put port 3. This includes all phase shifts: the signal leads as well as the transmissionlines between the hybrids and the power divider. Fig. 4.14(b) shows the simulatedand measured absolute phases between the LO port and port 3, and between the RFport and port 3 with the same scheme. Simulation and measurement show good agree-ment at the center frequency around 1.5 GHz, but vary greatly at other frequencies.However, the aimed 90 phase difference between the LO and RF signal at port 3 isachieved. We will see in Chap. 5 that an exact phase difference is not a prerequisitefor a functional multi-port interferometer. In fact, the intrinsic need for calibrationof multi-port interferometers removes this phase imbalance. However, for broadbandreception, the phase relations between the output ports must be maintained over thedesignated frequency range.

As discussed earlier, the symmetry of the six-port structure leads to two pairs of equalphase shifts for signals that originate from the LO port. These two pairs can be seen

83


in Fig. 4.15(a). At the center frequency of 1.5 GHz, the designated 90 phase shift isfulfilled well (as marked with the arrow). Furthermore, the parallel characteristics ofthe lines go from approximately 1.25 GHz to 1.75 GHz. The fact that the two pairs oflines do not transect each other or have equal values at a certain frequency will allowoperation over the whole frequency range from 0.5 GHz through 2.5 GHz. However,this requirement fails when looking at the phase relations for signals that originatefrom the RF port. Phase shifts at two different output ports become the same at1.075 GHz which results in the same phase information at these ports. They becomeindistinguishable, and it is not possible to calibrate the six-port at this frequency.Besides the phase perspective, the attenuation also drops steeply at this frequency.This is another reason for the six-port to fail operation at this frequency (see Fig.4.13).

-180

-360

-540

-720

-900 0.5 1 1.5 2 2.5

phase [

deg]

frequency [GHz]

S13S14S15S16

(a) From the LO port to all output ports

-180

-360

-540

-720

-900 0.5 1 1.5 2 2.5

phase [

deg]

frequency [GHz]

S23S24S25S26

(b) From the RF port to all output ports

Fig. 4.15: Measured phase shifts between LO and RF input ports and all outputports. The 90phase difference between port 3 and port 4 (as well as port6 and port 5) at the center frequency of 1.5 GHz is marked.

While Fig. 4.15 shows the absolute phase shifts at the output ports, Fig. 4.16 illus-trates the phase difference between the LO and RF signals at all output ports. Thisallows for a better understanding of how the six-port actually functions. We haveseen in Chap. 2 that the multi-port theory demands superpositions of the LO and RFsignals under different phase angles. When looking at Fig. 4.16(a), we can see howthe designated phase shifts in increments of 90 agree well at the center frequency of1.5 GHz. Fig. 4.16(b) demonstrates an even better picture of the phase relations asall phase differences are now referred to port 3 according to

∆ϕi =φ(S1i) − φ(S2i)

φ(S13) − φ(S23). (4.6)

It can now be clearly seen, how the phase differences are maintained over this frequency.Phase imbalances become stronger when moving away from the center frequency of

84

4.4 Analysis of the 2 GHz to 25 GHz Six-Port Interferometer (SP40)

1.5 GHz. As seen earlier, the six-port fails to operate at 1.075 GHz and 2.16 GHzwhen the phase information at the two different output ports becomes the same. Inthis case, the phase difference between the LO and RF signal becomes the same atport 4 and port 5 at 1.1 GHz, as well as at port 4 and port 6 at 2.2 GHz. However,the theory of multi-port receivers has shown that 3 different phases are enough forcalibration. In this case of critical frequencies, the six-port is reduced to a five-port.

180

0

-180

-360

0.5 1 1.5 2 2.5

ph

ase

[d

eg

]

frequency [GHz]


(a) Absolute difference

0

-90

-180

-270

0.5 1 1.5 2 2.5

ph

ase

[d

eg

]

frequency [GHz]


(b) Relative difference (normalized with respectto port 3)

Fig. 4.16: Measured phase differences between the LO and RF input signals at alloutput ports.

4.4 Analysis of the 2 GHz to 25 GHz Six-Port

Interferometer (SP40)

Wilkinson power divider quadrature hybrids power detectors

LO

RF

Fig. 4.17: Photograph of SP40 interferometer.

The second six-port interferometer (SP40) that allows an operation from 2 GHz to 25GHz is composed of commercial components. However, the basic structure is the oneof Fig. 4.2(c). This six-port (as illustrated in Fig. 4.17) consists of extreme broadband

85


components: one Wilkinson power divider and three quadrature hybrids. S-parametermeasurement results will be presented in this section.

4.4.1 Guidelines for Broadband Power Divider Design

A broadband multi-port receiver design requires single broadband components. Thiscan be achieved by additional λ/4-bows in the Wilkinson power divider, each corre-sponding to a different frequency range. Adding these bows increases the bandwidthat the cost of transmission loss. The additional loss is due to the longer lines andresistors that connect the bows between the path to port 2 and the path to port 3. Adetailed design description of a multiple bow power divider is given in elsewhere [111].

4.4.2 Measurement Results of a Broadband Power Divider

The bandwidth of a power divider is given by the frequency range of equal power split.In addition, the phase shift between the input port and the two output ports shouldbe equal and smooth over the frequency range. The best theoretical power split of anon-resistive power divider is -3 dB from the input port to both output ports (i.e. nopower is lost and the power is equally distributed at the output ports).

0

-10

-20

-30 5 10 15 20 25

0

-3

-6

-9

S11, S

23 [

dB

]

S12 [

dB

]

frequency [GHz]

S12S11S23

(a) Return loss, transmission, and isolation

180

90

0

-90

-180

25 20 15 10 5 1

phase [

deg]

frequency [GHz]

(b) Phase shift from port 1 to port 2 and 3

Fig. 4.18: S-parameter measurement result of the broadband (2 GHz through 25GHz) power divider.

Fig. 4.18 shows S-parameter measurement results of the MITEQ D0289 power dividerthat have an extremely broadband response. In Fig. 4.18(a), the transmission lossfrom the input port to the output port is between 3.5 dB, and 4 dB over the entirefrequency range from 2 GHz to 25 GHz. Due to symmetry, the path that leads to

86


port 2 equals the path that leads to port 3 (S12=S13). The return loss S11 is below 17dB from 10 GHz to 25 GHz and decreases towards smaller frequencies. However, it isstill above 10 dB even at smaller frequencies. The isolation between the two outputports, S23, is of minor importance for the six-port interferometer. The design of theinterferometer is such that the RF port is isolated against the LO port and only a smallamount of power will get from the RF port to the LO port. The isolation is above19 dB from 10 GHz to 25 GHz and decreases steeply towards smaller frequencies, upto 6 dB at 2 GHz. Fig. 4.18(b) shows the smooth run of the absolute phase from theinput port to both output ports 2 and 3.

4.4.3 Guidelines for Broadband Quadrature Hybrid Design

Similar to the design of the power divider in Chap. 4.4.1, the bandwidth of a quadraturehybrid is increased by adding λ/4-arms to the structure. A detailed design descriptionis given elsewhere [111].

4.4.4 Measurement Results of a Broadband Quadrature Hybrid

0

-10

-20

-30

-40

-50 25 20 15 10 5 1

0

-3

-6

-9

-12

S11,

S14 [

dB

]

S12,

S13 [

dB

]

frequency [GHz]

S12S13S11S14

(a) Return loss, transmission, and isolation

180

90

0

-90

-180

5 10 15 20 25

phase [

deg]

frequency [GHz]

S12S13

S12-S13

(b) Phase shift from port 1 to port 2 and 3, andphase difference between port 2 and port 3

Fig. 4.19: S-parameter measurement result of the broadband (2 GHz through 25GHz) quadrature hybrid .

Fig. 4.19 shows the measurement results of a MCLI HB23 quadrature hybrid. Thetransmission loss from port 1 to port 2 (S12) is between 3 dB to 5 dB from 3 GHzto 25 GHz (Fig. 4.19(a)). Almost equal power splitting is achieved at this frequencyrange. Towards 2 GHz, more power is transported to port 2 and less power gets toport 3 (S13). Good matching is achieved over the entire frequency range from 2 GHz to25 GHz; the return loss (S11) is above 18 dB. The isolation S14 is another importantparameter. It is above 23 dB from 9 GHz to 25 GHz and decreases towards lower

87


frequency. However, it stays above 14 dB over the entire frequency range. The phasebehavior is plotted in Fig. 4.19(b). A smooth run of the absolute phases from port1 to port 2 and 3 is achieved. The phase difference between the output port is veryclose to the theoretical 90 over the entire frequency range. This is an importantfeature when looking at the design of the six-port interferometer. The behavior ofthe phase is also very important for the understanding of the six-port and all othermulti-port interferometers. The single components do not need to hold the absolutephase constant, but they need to have a smooth and constant phase difference at theoutput ports. In the case of the hybrid, the phase difference between the output portsneeds to be constant.

4.4.5 Measurement Results of the SP40

Finally, the power divider and quadrature hybrid are connected with semi rigid linesand K-connectors and become a six-port interferometer (Fig. 4.2(c) and Fig. 4.17). Inthe following, the S-parameter measurement results will be presented.

Matching and Isolation

0

-10

-20

-30

-40

-50 5 10 15 20 25

S11,

S22 [

dB

]

frequency [GHz]

S11S22

(a) Return loss at port 1 and port 2

0

-10

-20

-30

-40

-50 5 10 15 20 25

S12,

S27 [

dB

]

frequency [GHz]

S12S27

(b) Isolation between port 2 and port 1, and port2 and port 7 (unused)

Fig. 4.20: Measured return loss at the input ports (1,2) and isolation of SP40.

First, good matching needs to be achieved at the LO and RF input ports. Fig. 4.20shows the return loss at the LO port (S11) and the RF port (S22). It can be seenthat the overall matching corresponds to that of the single components, the powerdivider, and the quadrature hybrid. The power divider is connected at the LO port.Comparing Fig. 4.18(a) with Fig. 4.20(a) shows stronger scattering due to reflections.This behavior is also found at the RF port where the quadrature hybrid is connectedas well as in the transmission that will follow in the next section. The return loss at the

88


LO port is above 10 dB from 2 GHz through 25 GHz while at the RF port, the returnloss is better over the entire frequency range (above 17 dB). An isolation greater than23 dB is found over the entire frequency range (Fig. 4.20(a)). The isolation betweenthe RF input port 2 and the unused isolated port 7 (S27) is important as all powerthat gets to port 7 is wasted in a 50 Ω termination. One can see that this isolationgets below 20 dB only at frequencies between 2 GHz and 9 GHz (down to 14 dB at 3GHz).

Transmission

Concerning power transfer, the transmission from the input ports to the output portsis the most important parameter. The transmission S1i should be kept close to thetheoretical 6 dB transmission loss that is the sum of the 3 dB transmission loss at thepower divider and the 3 dB loss at the quadrature hybrid. The interconnects shouldbe as short as possible to avoid additional loss.

0

-3

-6

-9

-12 5 10 15 20 25

S13,

S14,

S15,

S16 [

dB

]

frequency [GHz]

S13S14S15S16

(a) From the LO port to port 3 through 6

0

-3

-6

-9

-12 5 10 15 20 25

S23,

S24,

S25,

S26 [

dB

]

frequency [GHz]

S23S24S25S26

(b) From the RF port to port 3 through 6

Fig. 4.21: Measured transmission from the input ports (1,2) to the output ports (3through 6) of SP40.

Fig. 4.21(a) shows that a relatively good transmission between 6.5 dB and 8.5 dB fromthe LO input port to any output port is achieved from 3 GHz to 25 GHz. The splittingbehavior at lower frequencies is due to the quadrature hybrid (see Fig. 4.19(a)). Also,stronger scattering is found compared to the single components measurement.

The transmission from the RF port to any output port is similar to that from the LOport (Fig. 4.21(b)). Good power distribution is found at frequencies between 3 GHzand 25 GHz while frequencies below 3 GHz do not show equal power split. However,the transmission loss is between 6 dB and 8.5 dB over the frequency range of interest.

89


Phase Shifts

Besides equal power distribution at the output ports with low signal attenuation, thephase difference at the output ports must also support the large frequency range. For abroadband design, this phase difference between the RF and LO signal at the outputports needs to have parallel characteristics. Due to the extreme bandwidth from 2GHz through 25 GHz, a meaningful illustration of the phase relations is needed.

180

90

0

-90

-180

5 10 15 20 25

ph

ase

[d

eg

]

frequency [GHz]

S13S14S15S16

(a) From the LO port to port 3 through 6

0

-2500

-5000

5 10 15 20 25

ph

ase

[d

eg

]

frequency [GHz]

S23S24S25S26

(b) From the RF port to port 3 through 6

Fig. 4.22: Measured phase shifts from input ports (1,2) to the output ports 3 through6 of SP40. Due to its large bandwidth, the single lines can hardly bedistinguished, even in the non-zig-zag fashion (b).

Fig. 4.22(a) shows the absolute phases between LO input port 1 and the output ports3 through 6. In this illustration, the phase varies between +180 and -180. It can beseen that two equal pairs of lines are measured at frequencies up to approximately 15GHz . At higher frequencies, slightly different phases are measured at all output ports.

The absolute phases of the signals that originate from the RF input port in Fig. 4.22(b)are depicted with straight lines according to the scheme from Fig. 4.14. It can be seenthat the absolute phases show a smooth run over the entire frequency range while S23

and S25 lie on top of each other.

As discussed earlier, the absolute phases from the input ports to the output portsare of minor interest. The important parameter is the phase difference between theRF and LO signal at the output ports. These phase differences are depicted in Fig.4.23. In the illustration of the absolute phase differences at the output ports in Fig.4.23(a), the difference between the single lines is of interest. This difference is therelative phase difference. The theoretical value of a six-port interferometer is 90.This relative phase difference can be seen very well when normalized with respect toport 3. The results are shown in Fig. 4.23(b). It can be seen that the relative phases

90

4.5 The Reconfigurable MEMS-Based Multi-Band Front-End

have the theoretical 90 at 1 GHz. The theoretical 180 at port 4 is well maintainedover the entire frequency range. Also, the relative phases at port 5 and port 6 showa parallel run. The relative drop of the phase of the two pairs, port 3 and 4 andport 5 and 6, is due to the slightly different lengths of the semi rigid leads from thepower divider to the quadrature hybrid as depicted in Fig. 4.17. The different lengthsresult in a delay that is dependent on the frequency. We will see that the six-portinterferometer can still be calibrated over the entire frequency range.

0

-1000

-2000

-3000 25 20 15 10 5 1

ph

ase

[d

eg

]

frequency [GHz]


(a) Absolute phase difference

0

-90

-180

-270

25 20 15 10 5 1

ph

ase

[d

eg

]

frequency [GHz]


(b) Relative phase difference (normalized withrespect to port 3)

Fig. 4.23: Measured phase differences between input signals (LO and RF) at theoutput ports 3 through 6 of SP40. The designated 90phase differencebecomes better towards lower frequencies.

4.5 The Reconfigurable MEMS-Based Multi-Band

Front-End

This chapter gives a short introduction to targeted applications of different RF MEMSincluding the routing structures, and new elements such as resonators and low losstransmission lines.

4.5.1 Targeted Applications of RF MEMS in Receiver Front-Ends

Fig. 4.24 shows an application scenario of RF MEMS in reconfigurable and in regularreceiver front-ends. The illustration shows a six-port receiver including power detec-tors (PD), an ADC, a micro controller, and an FPGA. The use of RF MEMS alsoleads to great improvements in conventional RF front-ends. The evaluation in Chap.4.5.3 will focus on signal routing RF MEMS.

91


FPGAA/D

converter

microcontroller

data

I(t)

, Q(t

)

dem

odu-

latio

n

digi

tal

filte

ring

50 Ω

90o

90o

90o

powerdivider

VCO

six-port

LNA

PD

PD

PD

PD

power detectorPD

Fig. 4.24: Application scenario for different RF MEMS including switches, high-Qfilters, and low-loss transmission lines in a six-port receiver architecture.

The processes that are used to fabricate the RF MEMS switches and the RF crosscan be slightly modified to produce deeply etched structures (dry chemical etching ina LAM reactor). This leads to low loss transmission lines and cavity resonators. Adetailed description of their fabrication process and the measurement results is givenin [14]. These CPW transmission lines help to further decrease transmission loss andwould typically be applied in the relatively long lines of the interconnects of the singleelements in six-port interferometers. From an economic point of view, it is obviousthat a fabrication of a six-port interferometer on a Si wafer would only make sense forhigher frequencies due to the long λ/4 lines. Bandpass resonators or filters can be useddirectly behind the antenna or in large filter banks where the signal is switched withan SPDT or higher order switch. The depicted single shunt airbridge switch refersto the possibility of creating phase shifts with a switchable (digital) capacity. Thishas been carefully investigated in an application of a steerable antenna at 24 GHz [113].

The concept of using a MEMS capacitance for phase shifting has not been applied tothe broadband six-port front-end design. The reason for this is that the membranewould become too large to achieve a meaningful change of the capacitance at low fre-quencies around 2 GHz. Also, the use of SPDT structures in the quadrature hybriditself has been considered but not further investigated. One can think of two SPDTswitches in the quarter wavelength arms of the quadrature hybrid (or in the bows ofthe power divider) to switch the signal over an additional delay line with a length thatdepends on the operation frequency. However, the total number of SPDT switches

92


in such a structure (and its final size) is disadvantageous in comparison to simplyroute the signals through two separate six-port interferometers. Obviously, the twointerferometers can then be designed for different frequency ranges. This applicationis indicated with the MEMS at the lower right of Fig. 4.24 and will be thoroughly in-vestigated in the following. In addition to signal routing in the RF front-end, MEMSresonators are a promising alternative for high Q filters [14][131] and low loss trans-mission lines [132].

4.5.2 The MEMS-Based Reconfigurable Six-Port Front-End

SP1500 SP40

VCO1 VCO2

PA

D

PA

LNA LNA

ADCLNA

PD

ADAC

SPDT CROSS

MEMS

Fig. 4.25: The reconfigurable multi-band six-port receiver front-end under investi-gation.

Fig. 4.25 shows the reconfigurable multi-band six-port receiver front-end which willbe characterized in Chap. 4.5.3. SP1500 has a center frequency of 1.5 GHz; SP40covers the frequency range from 2 GHz to 25 GHz. RF MEMS SPDT switches areused to route the signals from the two six-port interferometers to one set of powerdetectors. In addition, an SPDT switch is applied as an RX/TX antenna switch.Behind each power detector, an LNA is used to amplify the signal before AD con-version. The algorithms for IQ data recovery that takes place in the digital domainare not included in this picture. Chap. 5 will consider all aspects of the digital domain.

The transmitter itself is not looked at in detail. A standard transmitter is depicted

93


with an digital-to-analog converter (DAC), a filter, an amplifier, and a mixer forup-conversion. The mixer is fed by a LO signal from a common voltage controlledoscillator (VCO). A power amplifier (PA) is used to amplify the signal before it getsto the air interface. The RF MEMS SPDT from Chap. 3.6.3 can be used directly asan RX/TX switch. However, the switch time is limited to 12 µs for activation and28 µs for release as discussed in Chap. 3.7.1. The block level design of the completeSP1500 receiver is equivalent to that of the SP40 receiver.

4.5.3 Results of the MEMS-Based Reconfigurable Six-PortFront-End

The system level simulation in this chapter is performed with Agilent ADS. Recorded S-parameter measurement results are used for the system level simulation. This includesboth six-ports, as well as the RF MEMS SPDT switches and the RF cross. Prior tothe simulation, the S-parameter measurement results from two-port measurements ofboth six-port interferometers as well as the RF MEMS SPDT switch and RF crossare converted to three-port (s3p) and six-port (s6p) files with MATLAB according tothe touchstone format. These blocks can then be inserted directly on system leveland simulated. The simulation considers all scattering effects of the six-ports andRF MEMS but neglects the connecting lines. This is a realistic assumption as themeasurement of all structures already includes some sort of connecting line. The designof the final front-end can be created in such a way as to avoid additional lines. Alsonote that, transitions from micro-strip to CPW are not included in the simulations.

The actual measured S-parameter results from Chap. 4.3.5 and Chap. 4.4.5 are usedfor both six-ports. The results from Chap. 3.6.3 and Chap. 3.6.4 are used for the RFMEMS SPDT and RF cross. The MEMS SPDT RX/TX switch between BPF andLNA will be discussed separately. The performance of this switch is the direct resultof the RF MEMS SPDT switch.

Signal Routed Through SP1500

Fig. 4.26 show the ADS block diagram including both six-ports, the RF MEMS SPDTswitches, and the RF cross. The position of the toggle switches in the SPDTs is suchthat the signal is routed through SP1500 to output ports 3 through 6. In the blockdiagram, the input ports of the broadband six-port (SP40) that covers the frequencyrange from 2 GHz to 25 GHz are marked 7 for LO and 8 for RF signal.

94


6

5

4

3

LO

RF LO

RF

sixport_20GHz_item

X2

SP40

78

sixport_1500MHz_item

X1

SP1500

1

2

S_Param

SP1

Step=5 MHz

Stop=2.50 GHz

Start=0.5 GHz

S-PARAMETERS

Term

Term8

Z=50 Ohm

Num=8

Term

Term7

Z=50 Ohm

Num=7

Term

Term2

Z=50 Ohm

Num=2

Term

Term1

Z=50 Ohm

Num=1

Term

Term3

Z=50 Ohm

Num=3

Term

Term4

Z=50 Ohm

Num=4

Term

Term5

Z=50 Ohm

Num=5

Term

Term6

Z=50 Ohm

Num=6

spdt13

X14

2

31

spdt13

X13

2

31

spdt13

X12

2

31

rfcross

X11

3

4

2

1

rfcross

X9

3

4

2

1

rfcross

X10

3

4

2

1

rfcross

X8

3

4

2

1

rfcross

X7

3

4

2

1

rfcross

X6

3

4

2

1

spdt13

X5

2

31

Fig. 4.26: The reconfigurable multi-band receiver front-end when the signals routedthrough SP1500.


Fig. 4.27)(a) shows the matching at the LO port and the RF port of SP1500. It canbe seen that the results are comparable to those from Fig. 4.10 that do not have theMEMS at the output ports. The return loss is larger than 30 dB at the center fre-quency. Fig. 4.27(b) shows all relevant isolations. It can be seen that both input portsof SP40 become well isolated by the use of the SPDTs. The isolation is greater than40 dB from 0.5 GHz to 2.5 GHz. Again, the isolation between the LO port and theRF port of SP1500 is similar to the results from the measurement without the MEMS.

The isolation between the input ports of SP40 and the output ports 3 through 6 canbe seen in Fig. 4.27. Fig. 4.27(a) shows the isolation of LO port 7 which is greaterthan 40 dB for frequencies up to 2.2 GHz. The isolation of the RF port 8 of SP40is comparable – it stays greater than 40 dB for all frequencies up to 2.2 GHz (Fig.4.27(b)).

95


0

-10

-20

-30

-40

-50 0.5 1 1.5 2 2.5

S11

, S

22

[d

B]

frequency [GHz]

S11S22

(a) Return loss at LO port 1 and RF port 2

0

-20

-40

-60

-80 0.5 1 1.5 2 2.5

S1

2,S

17

,S2

7,S

18

,S2

8 [

dB

]

frequency [GHz]

S12S17S18S27S28

(b) Isolation between port 1 and port 2, and iso-lation to the input ports of SP2

Fig. 4.27: Return loss at the input ports (1,2) and relevant isolation of the multi-band front-end when signals are routed through SP1500.

0

-20

-40

-60

-80 0.5 1 1.5 2 2.5

S73,S

74,S

75,S

76 [

dB

]

frequency [GHz]

S73S74S75S76

(a) Isolation of LO port 7

0

-20

-40

-60

-80 0.5 1 1.5 2 2.5

S83,S

84,S

85,S

86 [

dB

]

frequency [GHz]

S83S84S85S86

(b) Isolation of RF port 8

Fig. 4.28: Isolation of the input ports (7,8) of SP40 and output ports (3 through 6)when signals are routed through SP1500.

Transmission

Fig. 4.29 shows the transmission of the LO signal from input port 1 (Fig. 4.29(a))and the RF signal from input port 2 (Fig. 4.29(b)) of SP1500 to the output ports3 through 6. The transmission is insignificantly influenced by the application of theMEMS. The curves show a comparable behavior to the results from Fig. 4.13 with anadditional loss from the MEMS. However, it can be seen that the symmetry (two pairsof equal curves) is no longer well pronounced for the signals from the LO port. Thetransmission loss from the LO port is now between 6.5 dB and 8 dB, and between 7dB and 9 dB from the RF port.

96


0

-3

-6

-9

-12

-15

-18 0.5 1 1.5 2 2.5

S1

3,S

14

,S1

5,S

16

[d

B]

frequency [GHz]

S13S14S15S16

(a) From LO port 1

0

-3

-6

-9

-12

-15

-18

0.5 1 1.5 2 2.5

S2

3,S

24

,S2

5,S

26

[d

B]

frequency [GHz]

S23S24S25S26

(b) From RF port 2

Fig. 4.29: Transmission from the input ports (1,2) of SP1500 to the output ports(3 through 6) when signals are routed through SP1500.

Phase Relations

180

0

-180

-360

0.5 1 1.5 2 2.5

phase [

deg]

frequency [GHz]



0

-90

-180

-270

0.5 1 1.5 2 2.5

phase [

deg]

frequency [GHz]



Fig. 4.30: Phase differences from input ports (1,2) of SP1500 to output ports (3through 6) when signals are routed through SP1500. The designated 90

at the center frequency is marked.

The phase relations at output ports 3 through 6 are plotted in Fig. 4.30 for the casewhere the signals are routed through SP1500. Due to the application of the MEMSat the output ports, the signal travels over an additional delay line which changesthe absolute phase. However, the phase difference between the LO and RF signal atthe output port is the interesting parameter. In this application, both the LO andRF signal will travel exactly the same path over the MEMS field after leaving theinterferometer. In other words, adding any line behind the interferometer will notchange their phase difference. Of course, this argument only holds if the reflection at

97


the output port is small enough, and the reflected signal does not interfere with theincoming signals. The absolute and relative phase (Fig. 4.30(a) and (Fig. 4.30(b)) atthe output ports becomes only slightly modified by the use of the MEMS as the 90

phase difference is still well pronounced around 1.5 GHz.

Signal Routed Through 2 GHz to 25 GHz Six-Port Circuit

RF

RF

6

5

4

3

LO

LO

sixport_20GHz_item

X2

SP40

78

sixport_1500MHz_item

X1

SP1500

1

2

spdt12

X15

2

13

spdt12

X14

2

13

spdt12

X13

2

13

spdt12

X12

2

13

S_Param

SP1

Step=10 MHz

Stop=25.0 GHz

Start=0.5 GHz

S-PARAMETERS

Term

Term8

Z=50 Ohm

Num=8

Term

Term7

Z=50 Ohm

Num=7

Term

Term2

Z=50 Ohm

Num=2

Term

Term1

Z=50 Ohm

Num=1

Term

Term3

Z=50 Ohm

Num=3

Term

Term4

Z=50 Ohm

Num=4

Term

Term5

Z=50 Ohm

Num=5

Term

Term6

Z=50 Ohm

Num=6

rfcross

X11

3

4

2

1

rfcross

X9

3

4

2

1

rfcross

X10

3

4

2

1

rfcross

X8

3

4

2

1

rfcross

X7

3

4

2

1

rfcross

X6

3

4

2

1

Fig. 4.31: The reconfigurable multi-band receiver front-end when the signals routedthrough SP40.

Fig. 4.31 shows the ADS simulation platform including both six-ports, the RF MEMSSPDT switches, and the RF cross. The position of the toggle switches in the SPDTsis such that the signal is routed from LO and RF ports 7 and 8 of SP40 to the outputports 3 through 6.

98



Fig. 4.32(a) shows the insertion loss at the LO and RF input port 7 and 8 of SP40. Thescattering becomes more pronounced with deep peaks when compared to the versionwithout MEMS (Fig. 4.20). However, the return loss at the LO port (S77) is still above10 dB at lower frequencies and above 13 dB at higher frequencies. Important isolationvalues are given in Fig. 4.32(b). The isolation between the RF and LO input port ofSP40 stays above 20 dB over the entire frequency range. Apart from minor peaks thatcome down to 10 dB at around 8 GHz, 12 GHz, and 23 GHz, the isolation betweenRF and LO input port 7 and 8 of SP40 and the input RF port 2 of SP1500 stays wellabove 20 dB.

0

-10

-20

-30

-40

-50 5 10 15 20 25

S77,

S88 [

dB

]

frequency [GHz]

S77S88

(a) Return loss at LO port 7 and RF port 8

0

-20

-40

-60

-80 5 10 15 20 25

S78,S

71,S

72,S

81,S

82

[d

B]

frequency [GHz]

S78S71S72S81S82

(b) Isolation between port 7 and port 8, andtheir isolation to the input ports of SP1500

Fig. 4.32: Return loss at input ports (7,8) and relevant isolation of the multi-bandfront-end when signals are routed through SP40.

Fig. 4.33 shows the isolation between the input ports of SP1500 and the output portswhen the SPDTs connect SP2. It can be seen that apart from minor peaks at theaforementioned frequencies, the isolation also stays well above 20 dB.

Transmission

The transmission loss of the LO and RF signals that run through SP40 and the MEMSfield is depicted in Fig. 4.34. It can be seen that the use of the MEMS leads toan additional scattering of the transmission values as mentioned earlier. Also, theexpected additional attenuation due to the MEMS (approximately 3 dB) can be seenin both the LO (Fig. 4.34(a)) and the RF signals (Fig. 4.34(a)).

99


0

-20

-40

-60

-80 5 10 15 20 25

S1

3,S

14

,S1

5,S

16

[d

B]

frequency [GHz]

S13S14S15S16

(a) Isolation of the LO port 1

0

-20

-40

-60

-80 5 10 15 20 25

S2

3,S

24

,S2

5,S

26

[d

B]

frequency [GHz]

S23S24S25S26

(b) Isolation of the RF port 2

Fig. 4.33: Isolation of input ports (1,2) of SP1500 and output ports (3 through 6)when signals are routed through SP40.

0

-3

-6

-9

-12

-15

-18 5 10 15 20 25

S73,S

74,S

75,S

76 [dB

]

frequency [GHz]

S73S74S75S76

(a) Transmission from LO port 7

0

-3

-6

-9

-12

-15

-18 5 10 15 20 25

S83,S

84,S

85,S

86 [dB

]

frequency [GHz]

S83S84S85S86

(b) Transmission from RF port 8

Fig. 4.34: Transmission from the input ports (7,8) of SP40 to the output ports (3through 6) when signals are routed through SP40.

Phase Relations

Looking at the phase relations of the LO and RF signals when using the SP40 interfer-ometer (Fig. 4.35), it becomes clear that the six-port can be calibrated over the entirefrequency range and will function properly.

In the graph of the relative phase difference that is normalized with respect to port 3,one can see the characteristic two pairs of parallel lines as discussed with Fig. 4.23(b).The additional scattering due to reflections can still be seen but is small enough sothat it does not interfere with other lines. However, at lower frequencies where thescattering is relatively strong, a broadband operation might not be available.

100


0

-1000

-2000

-3000 25 20 15 10 5 1

ph

ase

[d

eg

]

frequency [GHz]



0

-90

-180

-270

25 20 15 10 5 1

ph

ase

[d

eg

]

frequency [GHz]



Fig. 4.35: Phase differences from input ports (7,8) of SP40 to the output ports (3through 6) when signals are routed through SP40.

4.5.4 The RF MEMS SPDT Antenna Switch

The RF MEMS SPDT switch is well described in Chap. 3. Its measurement results arepresented in Chap. 3.6.3. Its S-parameters are plotted in Fig. 3.24(b) and Fig. 3.24(d)for the two different routing states of the toggle switches. This SPDT switch canbe directly applied as an RX/TX switch. Compared to PIN diode switches, the RFMEMS SPDT switch is superior in power handling and decreased loss. A disadvantageis the slower switching time due to the movement of mechanical parts. The switchingtime for RF power is measured in Chap. 3.7.1 and is depicted in Fig. 3.32. The timefor activation is 12 µs and the time for release is 28 µs.

101


102

5 Performance of the Multi-BandSix-Port Receiver

In this chapter, the multi-band six-port receiver will be evaluated by appropriate SERmeasurements over the frequency range from 1 GHz to 40.0 GHz. The main focuslies in the empirical proof of the concept of the developed multi-port SDR receiver,especially its coverage of the large frequency range. Time domain simulation resultsat a frequency of 1.5 GHz will be presented in the beginning to explain the basicproperties of the receiver front-end. For these time domain simulation, a fast andflexible C++ based system simulation tool called “CppSim” was used that includesthe RF path as well as the phase noise of the local oscillator.

5.1 Simulation Environment

5.1.1 Functional Principle of the Simulation Program CppSim

The computer simulation program CppSim is very suitable for the time domain sim-ulation of the six-port receiver [22]. A short introduction in its functional principle[115] will be given next.

The ability to represent systems in an object oriented manner allows an elegant frame-work for their simulation. These facts make C++ the language of choice for maximalfreedom in the development of simulation code for the system under investigation.The CppSim package removes the drawbacks of plain C++ programming by supply-ing classes that allows for easy representation of system building blocks such as filters,amplifiers, voltage controlled oscillators (VCO), etc. It also supplies a netlist to C++conversion utility that enables automatic code generation from a graphical descriptionusing any of the mainstream schematic editor packages. Large systems can easily besimulated in this manner.

The simulation approach taken with CppSim is to represent blocks in the system basedupon input, state, and output relationships, similar to the mainstream Simulink fromMathWorks. However, unlike Simulink, these relationships do not need to be placed

103

5 Performance of the Multi-Band Six-Port Receiver

in state-space form and conditional loops, rather than vectorized expressions, are sup-ported. The blocks are represented in an object oriented manner. The simulation codecalculates the overall system behavior by computing the output of each block one ata time for each sample point in the simulation. This approach has the advantage ofallowing for fast computation, a straightforward description of blocks, and the abilityto easily support multi-rate operation of different blocks in the system. The primarydisadvantage of CppSim is that it is limited to architectural investigation only. How-ever, as we will see, the architectural investigation will model the transmitter andsix-port receiver behavior very well.

5.1.2 Simulation Set Up

The system level block diagram of the simulated transmitter and six-port receiveris depicted in Fig. 5.1. The blocks are executed in the order from “xi0” to “xi25”(after running the code in each block, its outputs are updated according to the newcalculated values which are then used as the inputs of the subsequent block). For allsimulations, a sample time of 5×10−11 s with a total number of 2×107 time steps isused. The RF frequency of the simulations is 1.5 GHz which is the center frequencyof the six-port described in Chap. 4.3. The symbol rate is 1 MHz, which results in adata bit rate of 2 MBit/s for a QPSK signal.

In the transmitting path, a voltage controlled oscillator block “vco” at 100 MHz servesas the clock signal for the phase locked loop (PLL) transmitter and the IQ samples forthe QPSK modulation. Its reference is set to zero phase noise as the focus of the simu-lation platform is in the receiver part. The PLL transmitter (“transmitter1500MHz”)itself is composed of a phase frequency detector (PFD), a charge pump, a loop filter,and a divider (not shown). This allows for a very accurate simulation of the PLLbehavior as described in [114]. However, in this simulation, the components in thetransmitter are supposed to be ideal and all phase noise is set to zero. In this way,the pure, noiseless RF signal has the form of a sine wave with four different phasesdepending on the QPSK state. Small spikes are generated in the PLL transmitter atthe point where the phase is shifted by modulation. These spikes can later be seen inthe time domain baseband signal. The signal amplitude behind the transmitter is 1V.

To introduce noise into the system, zero-mean additive white Gaussian noise (AWGN)is added to the signal in the PLL transmitter. This is done by adding AWGN tothe baseband IQ values that are used to control the phase of the PLL. The noise inthe amplitude is added to the amplitude of the RF signal that comes out of the PLLtransmitter. AWGN is needed for the symbol error rate (SER) simulation that is per-formed in Chap. 5.2.1. The magnitude of this noise is adjusted with the variance σ2

N .Before the signal gets to the receiver input, it is attenuated by 50 dB to have a signal

104

5.1Sim

ulation

Environ

men

t

amplitu

de

inth

equad

raticregion

ofth

edio

de.

out1_adc

out2_adc

out3_adc

out4_adc

val=0

out

xi0

constant

vco

vctrl

squareout

sineout

freq=100e6 Hz

kvco=1 Hz/V

xi1

xi2

probes

clk

I

fd = 100 [^-1*fc]

Q

xi3

transmitter1500MHz

q_in

i_in

RF_out

clk

var=awgn_channel_gl

noise

out

xi4

out in1

in2

add2xi5

a y

gain=gain_rf_gl [dB]

xi6

val=0

out

xi7

constant

var=phnoise_lo_gl

noise

out

xi8xi9

kvco = 30e6

0

90

180

270vco_sixport_phase

fc = 1500.000e6

phase

vctrl

a y

gain=gain_lo_gl [dB]

xi10

a y


xi11

a y


xi12

a y


xi13

outin1

in2

add2xi14

outin1

in2

add2xi15

outin1

in2

add2xi16

outin1

in2

add2xi17

xi18

in out

_sys_power_detector

xi19

in out

_sys_power_detector

xi20

in out

_sys_power_detector

xi21

in out

_sys_power_detector

a y

gain=gain_bb_gl

xi22

a y

gain=gain_bb_gl

xi23

a y

gain=gain_bb_gl

xi24

a y

gain=gain_bb_gl

xi25

Fig

.5.1

:Block

dia

gram

ofC

ppSim

six-port

receiversim

ula

tion.

105


The six-port receiver is realized by a VCO which produces sine wave outputs withphases 0, 90, 180, and 270 independent of the frequency. The magnitude of thephase noise is controlled by the parameter “phnoise lo gl”. The LO frequency of thereceiver runs at 1.5 GHz. Before the LO and RF signals are added, the LO signal isattenuated by 20 dB. The composed new signal (RF+LO) has the same frequency of1.5 GHz, but an amplitude that depends on the phase difference between the LO andRF signal. This is discussed in detail in Chap. 2 and Chap. 4.

In the quadratic region of the power detector, the voltage level at the output is pro-portional to the power of the input signal. This power detector is composed of the“diode schottky” block that applies the formula out = 10−4 · (e−in/0.025) − 1. Afterthis, the signal is low pass filtered by a standard RC filter with a corner frequency of2 MHz. Behind the power detectors (“sys power detector”), the signal is amplified by60 dB. The sampling of the baseband signal is realized by the “trigger” functionalityof CppSim which allows for storing the probed samples in an ASCII file. The triggerfrequency is set to 10 MHz which will result in 10 samples per symbol. This ADCconversion (as well as all amplification and attenuation blocks) do not introduce anynoise into the system.

5.1.3 Simulation Run

A typical simulation run with the subsequent six-port calibration and demodulationwill be discussed in the following. The simulation run takes about 300 s to cover thesystem time of 1 ms and produces 10,000 samples. With 10 samples/symbol, thisresults in a total of 1,000 symbols.

1

0

−1

10 9 8 7 6 5 4 3 2 1 0

in−

phase

relative time [µs]

1

0

−1

10 9 8 7 6 5 4 3 2 1 0

quadra

ture

relative time [µs]

Fig. 5.2: Snapshot of the sent in-phase(I) and quadrature (Q) se-quence (fsymb=1 MHz).

1.2

1.1

1

0.9 10 9 8 7 6 5 4 3 2 1 0

dete

cto

r outp

ut

voltage [

V]

relative time [µs]


Fig. 5.3: Voltages at the outputs of the4 baseband amplifiers shownoise at the edges.

106

5.1 Simulation Environment

Fig. 5.2 shows a snapshot of the sent time sequence of the in-phase (I) and quadrature(Q) components with a symbol rate of 1 MHz (no noise present). This IQ source isused in the PLL transmitter to produce the modulated RF signal. After the simula-tion of the various blocks (discussed in Chap. 5.1.2), the sampled output voltages ofthe four baseband amplifiers behind the diode detectors take the form shown in Fig.5.3. The spikes that were caused by the PLL transmitter can be seen at the edgesbetween two samples. According to the six-port theory, the I and Q component canbe calculated from these four voltages after calibration. The voltage samples that areused for the calculation of the I and Q component are taken at the position of 0.8 µsof the symbol duration.

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

pone

nt

(Q)

in−phase component (I)

(0)(1)

(2)(3)

Fig. 5.4: Symbols used for cali-bration.

1.2

1.1

1

0.9 240 200 160 120 80 40 0

dete

cto

r outp

ut

volta

ge

[V

]

relative time [µs]

(0) (3) (1) (2)

Fig. 5.5: Power levels of the calibration sequence(see Fig. 5.3 for the legend).

The QPSK symbols (1,1), (-1,1), (1,-1), and (-1,-1) are labeled with numbers from (0)to (3) as shown in Fig. 5.4. The detector output voltages Vi that correspond to theseIQ states are depicted in Fig. 5.5. For the calibration sequence in the left part of Fig.5.5, a symbol time of 40 µs (40 times the data symbol time) is used to achieve a goodcalibration by averaging over 400 samples. In this graph, the single sample pointsare not plotted to differentiate the line style for the four different output ports (seelegend in Fig. 5.3). In addition, these longer symbols can easily be found in the dataand recognized by the calibration algorithm. After the four different QPSK states ofthe calibration sequence, another (in this case unused) 10 µs long calibration sequencefollows before the actual random data samples start at 200 µs. (Note that not all ofthe 512 symbols of the sequence are shown in Fig. 5.5.)

We have seen in Chap. 2 that three output ports are actually sufficient for the calibra-tion of the multi-port receiver. Chap. 5.2.1 will present a comparison between six-portand five-port calibrations of the simulated output powers. The formulas used for the

107


calibration of the multi-port are

I(t) =n∑

i=1

ai · Vi (5.1)

Q(t) =

n∑

i=1

ai · Vi (5.2)

where n=4 for the six-port calibration and n=3 for the four-port calibration. Withthe four symbols and the 16 voltages from the four different ports, the above systemof linear equations can be solved by inversion of the voltage matrix and multiplicationby the sent IQ states. This will result in the calibration coefficients, ai and bi, thatcan be used to calculate the IQ states of the random symbols according to Eqn. 5.1.An algorithm implemented in MATLAB was used for the calculation of the calibrationcoefficients and for the subsequent calculation of the IQ values.

A 20 µs long section of the demodulated IQ values from the CppSim simulation isshown in Fig. 5.6. Some AWGN was added to the channel for illustrative purposes.Again, small spikes are formed at the edges (in between two different symbols) by thePLL of the transmitter and are not caused by the receiver. This time domain sequenceis composed of all power samples, not only the ones at the position of 0.8 µs that areassigned to the final IQ symbols.

1

0

−1

20 18 16 14 12 10 8 6 4 2 0

in−

phase

relative time [µs]

1

0

−1

20 18 16 14 12 10 8 6 4 2 0

quadra

ture

relative time [µs]

Fig. 5.6: Snapshot of the received in-phase (I)and quadrature (Q) sequence – thesymbol frequency is 1 MHz.

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)


Fig. 5.7: Demodulated I and Qsymbols show very lit-tle noise.

Fig. 5.7 shows the final demodulated constellation of symbols in the IQ space whenone out of ten samples of the power time sequence (the one at 0.8 µs) is assigned tothe demodulated IQ symbol. By using this one sample, the spikes that are caused bythe transmitter are not visible in the demodulated symbol. It can be seen that with

108

5.2 Simulation of the Six-Port Receiver

the low noise that was introduced, the measured points can be mapped to the correctsymbol without any error.


Some basic principles of the multi-port receiver can be demonstrated with the sim-ulation platform described in Chap. 5.1.2. In particular, these are: the influence ofAWGN in the signal path, the frequency offset between the LO and RF signal, andthe phase noise of the LO signal on reception quality.

5.2.1 Influence of Channel Noise

AWGN in the signal path causes a degradation of the symbol constellation and, there-fore, increases the SER. A symbol error occurs when a demodulated and demappedsymbol is different from the one sent – the symbol then appears in the wrong quadrant.The intention of adding noise is to find the SER that corresponds to a specified Eb/N0

ratio. The relationship between the Eb/N0 ratio and the signal-to-noise ratio (SNR)is given by

Eb

N0=

S

N+ log(

Brec

fsymb) (5.3)

where the noise N is given by the variance of the added noise signal σ2N , S is the signal

power, Brec is the receiver bandwidth (2 MHz), and fsymb is the symbol data rate (1MHz).

The SNR that belongs to a specific Eb/N0 ratio is calculated from the demodulatedsamples by applying a statistical approach. This statistical calculation is used becauseof the low numbers of samples available that would lead to a high uncertainty whensimply counting the wrong samples. In the statistical approach, a Gaussian distrib-ution of the demodulated symbols in the IQ space is assumed. For each noisy cloudof the demodulated symbols in the quadrants i=0..3 (see Fig. 5.4), the variance σ2

k,i

of the demodulated symbols xk,j of a quadrant is calculated in the I and Q direction(symbolized by the index k) according to:

σ2k,i =

1

n

n∑

j=1

(µk,i − xk,j)2, (5.4)

where

µk,i =1

n

n∑

j=1

xk,j (5.5)

109


is the mean value of the symbol cloud in the direction k. The SER for symbol i indirection k is then calculated by assuming the Gaussian distribution

G(xk,j, µk,i, σk,i) =1

σk,i

√2π

· e−

(xk,j−µk,i)2

2σ2k,i . (5.6)

This integral is numerically evaluated by adaptive Lobatto quadrature [116]. Eqn. 5.6is integrated for each symbol i and each direction k from −∞ or +∞ up to I=0 orQ=0, depending on the target quadrant.

The theoretical SER over the Eb/N0 ratio for QPSK transmission and coherent de-modulation can be calculated from the symbol error probability

PS = erfc(

√Eb

N0) − 1

4

[

erfc(

√Eb

N0)

]2

. (5.7)

Fig. 5.8(a) shows this theoretical curve together with the simulated results. Becausethe six-port receiver operates in its quadratic region where an optimum demodulationis guaranteed and no other inaccuracies are included in the simulation (ideal case),the simulated points are nearly identical to the theoretical values.

The noisy clouds in Fig. 5.8(b),(c), and (d) are samples that correspond to a singlepoint in the curve from Fig. 5.8(a). Fig. 5.8(b) shows that an Eb/N0 ratio of 19 dBleads to only minor noise in the IQ constellation. The calculated SER from the sim-ulated IQ symbols for this weak scattering is 1.3×10−30. By increasing the noise, thescattering of the demodulated IQ symbols becomes more pronounced with a SER of5.1×10−15 in the case of an Eb/N0 ratio of 15 dB, and a SER of 8.1×10−6 in the caseof an Eb/N0 ratio of 10 dB.

A noise evaluation for the five-port calibration is performed with the exact same col-lected voltage samples which were used for the six-port calibration in Fig. 5.8. This willhelp in understanding the influence of the additional output port 6. In this five-portcalibration, the power measurement value of output port 6 is neglected. Therefore, theremaining five-port has the output phases differences: 0, 90, and 180. Furthermore,only three IQ states are considered for the calibration (symbols (0), (1), and (3) arechosen).

The resulting dependency of AWGN in the signal path is depicted in Fig. 5.9. It isinteresting to see that, visually, the clouds show identical scattering when comparedto the six-port calibration. However, the cloud of symbol (2) is shifted away fromthe center (1,-1). The shift of this cloud is the reason for a slightly higher SERwhen compared to the simulation results of the six-port calibration in Fig. 5.8(a). Theshifting effect is caused by the disadvantageous calibration from only 3 known symbols

110


100

10−3

10−6

10−9

10−12

10−15

10−18

20 15 10 5 0

sym

bo

l e

rro

r ra

te (

SE

R)

Eb/N0 [dB]

simulationtheoretical limit

(a) SER versus Eb/N0

2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(b) Eb/N0 = 19 dB, SER = 1.3×10−30

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent (Q

)


(c) Eb/N0 = 15 dB, SER = 5.1×10−15

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent (Q

)


(d) Eb/N0 = 10 dB, SER = 8.1×10−6

Fig. 5.8: Influence of AWGN in the signal path for the six-port calibration and thecalculated SER.

not using the entire equally distributed phase spectrum with the phase differences of 0,120, and 240. This example of an imperfect calibration demonstrates the importanceand influence of the calibration. Therefore, it is of great importance for the multi-port receiver to have an accurate and reliable calibration process. Furthermore, thecalibration becomes more accurate by averaging many samples. There are severalways to achieve a good calibration; they are all done with appropriate algorithms inthe digital domain. A starting point is to include, after an initial rough guess, allcorrectly transmitted data points in the averaging process. This is another interestingfield in the study of multi-port receivers, but it is not the focus of this work.

111


100

10−3

10−6

10−9

10−12

10−15

10−18

20 15 10 5 0

sym

bo

l e

rro

r ra

te (

SE

R)

Eb/N0 [dB]

simulationtheoretical limit

(a) SER versus Eb/N0

2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(b) Eb/N0 = 19 dB, SER = 3.6×10−26

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent (Q

)


(c) Eb/N0 = 15 dB, SER = 2.6×10−13

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent (Q

)


(d) Eb/N0 = 10 dB, SER = 1.3×10−5

Fig. 5.9: Influence of AWGN in the signal path for the five-port calibration and thecalculated SER.

5.2.2 Frequency Offset and Phase Noise Dependency

Simulations with a frequency offset between the LO and RF signal, as well as increasedphase noise of the LO signal, are performed to investigate their dependencies in a six-port receiver. We have learned in Chap. 2 that, after calibration of the multi-portreceiver, any phase shift of either the LO or RF signal will directly translate to anequivalent phase shift in the IQ plane. In fact, this is a prerequisite for the properfunctioning of the receiver. Phase noise of a VCO is, in principle, a series of very smallphase shifts εi, where i refers to the LO or RF signal. Each of these small phase shiftstranslates to phase inaccuracies in the IQ plane and add up to move the IQ signalaround in a circle with the radius which equals the amplitude

√

I2 + Q2. Further-more, since in general the phase noise of the LO and RF signal is uncorrelated, bothphase errors add up to become εLO + εRF in the IQ plane.

112

5.3 Characterization of the Schottky Diode Detectors

Fig. 5.10(a) shows the implication of a frequency offset of 0.1 kHz between the LOand RF signal at a carrier frequency of 1.5 GHz. The angle that is indicated with thetwo arrows is 36, or 1/10 of the full circle. The phase offset of 0.1 kHz correspondsdirectly to 1/10 of the system time which is 1 ms. The small gaps in between the foursectors of the circle come from the 40 times longer calibration sequence that is alsoincluded in the demodulated data.

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

co

mp

on

en

t (Q

)


(a) Frequency offset 100 kHz

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

co

mp

on

en

t (Q

)


(b) Increased phase noise of LO

Fig. 5.10: Influence of the phase and frequency imbalances on the IQ constellationat 1.5 GHz.

The phase noise simulation in Fig. 5.10(b) shows the same impact as a frequency off-set. The demodulated symbols in the IQ plane bounce back and forth in small stepsin a circle around the origin. In addition to the small steps in the IQ plane, the entiresystem of the four QPSK points move around in the same circle. It can be proventhat – depending on the magnitude (standard deviation) of the phase noise σV CO –the symbols have, on average, completed a full circle in the IQ plane after 1/σV CO.

A frequency offset between the LO and RF signal can be corrected completely in theanalog front-end [117][118], with a feed back from the digital domain to the analogfront-end [119] or completely in the digital domain [120]

5.3 Characterization of the Schottky Diode Detectors

Before the multi-band measurement results of the six-port receivers are presented inChap. 5.4, it is of great importance to have a closer look at the Schottky power detec-

113


tors that are used behind the interferometer circuit for frequency conversion.

We have seen in Chap. 2 that, up to a certain input power, the detector output voltageVout is proportional to the input power PRF . This linear dependency is given in thequadratic region of the diode, where the exponential behavior is described accuratelyenough up to the square term of its power series. This square term then compensatesthe square dependency of the input power to yield a linear dependency with the outputvoltage.

0.1

1

10

100

1000

10 0−10−20−30

dete

cto

r volta

ge

Vo

ut [m

V]

power PRF [dBm]

measuredquadratic fit

Fig. 5.11: Characteristics of the Schottky power detectors type MCLI.

The measured power detector characteristics are illustrated in Fig. 5.11 in its typicaldouble logarithmic plot. One can see that for the MCLI power detectors, the linearrelation is valid from very small powers on up to approximately -15 dBm. This up-per limit defines the dynamic range of the diode when used for pure power detection.With increasing power, non-linear higher order terms are generated and become largerand degrade the power of the signal that is to be detected. However, the theory ofadditive mixing with diodes (multi-port theory) states, that with a large LO and asmall RF signal, the receiver still operates in a linear manner as long as the squarelaw approximation is valid.

The minimum detectable signal power, or sensitivity, defines the lower end of the dy-namic range. This parameter is more difficult to specify because it depends also onall the other components in the receiving path such as band pass filter and LNA. Thesix-port receiver under investigation here does not include these blocks and, therefore,the rather low sensitivity of this system does not reflect a general weakness of multi-port receivers.

The bandwidths of the MCLI power detectors has been determined experimentally

114

5.4 Measurement Results of the Multi-Band Six-port Receiver

with a modulated RF signal and a spectrum analyzer. The measured bandwidth isapproximately 80 MHz. The sensitivity can then be estimated by subtracting thethermal noise of the input resistor from -174 dBm/Hz [29]. This results in a noiseequivalent power of approximately -100 dBm/Hz at the input port which matchesexperimental investigations. However, more investigations with a complete receiverarchitecture are needed to specify the sensitivity in the receiver context, which is notthe main focus of this work. The main focus of this work is the demonstration andfeasibility of the six-port receiver for multiple standards at a large frequency range.

5.4 Measurement Results of the Multi-Band Six-port

Receiver

This chapter will present measurement results of the six-port receiver. The main ob-jective is to demonstrate the feasibility of the multi-band receiver for frequencies up to40.0 GHz. SER in dependency of the Eb/N0 ratio are performed at relevant frequencybands. In the beginning, the general behavior of the six-port receiver measurementset up is investigated and explained, following the identification of the influence ofin-channel interferers with different powers and at different offset frequencies.

5.4.1 Measurement Set Up and Run

file

SP40

SP1500

R&S SMR40

HP83623A

signal generator(IF)

RF source

MCLI ZD-5-Xpower detector




R&S SMU200

I Q

LO source

RF (2)

LO (1)

(3)

(4)

(5)

(6) PCI-DAS4020/12

ADC

common 10 MHz reference clock

data stored to file

calibration

demodulationI Q

six-port

IBMcompatible

microcomputer

filecomparison for SER evaluation

Fig. 5.12: Measurement Set Up for the Six-Port Receiver Characterization.

The measurement set up is depicted in Fig. 5.12. With this set up, the experimentaldata that are collected from the ADC and stored to a file can be directly processed

115


with the same MATLAB code that is applied to the simulated data from CppSim. Inparticular, the same calibration and data sequence is used.

The transmitter at the upper left corner of Fig. 5.12 is composed of a Rohde&Schwarz(R&S ) SMU200 vector signal generator. The IQ data sequences with different Eb/N0

ratios that are copied to the device are generated with R&S WinIQSIM. The generatoris limited to 3 GHz. Except for the RF frequency of 1.075 GHz, where the outputof the SMU200 is directly fed into the RF input port of the six-port, the SMU200generates an IF frequency of 333 MHz that is upconverted with a R&S SMR40 to theRF frequency (up to 40.0 GHz). The LO source is a HP83623A signal generator. Themaximum frequency of the generator is 20.0 GHz. Higher LO frequencies are gener-ated by using an external frequency doubler. Its conversion loss of approximately 10dB is compensated by changing the output power of the HP83623A signal generator.All signal generators as well as the ADC are fed by a common 10 MHz reference. Thisis useful since the frequency tracking and compensation of the IQ values is anothermajor topic and its impact is avoided. By this means, a direct comparison of theperformance at different frequencies can be carried out without having to deal withthe implications of the software correction algorithm or a hardware correction circuit.

The LO and RF signal are fed into input ports 1 and 2 of the six-port interferometercircuit. The switching between the two circuits – the designed interferometer SP1500for 1.5 GHz (see Chap. 4.3) and the hybrid broadband interferometer SP40 (see Chap.4.4) – is done manually be exchanging the circuits at their SMA connectors. At outputports 3 through 6 of the six-port circuit, the same MCLI ZD-5-X power detectors areused. Their inputs have a broadband ohmic 50 Ω matching. Their outputs have a highimpedance. The input ports of the Measurement Computing PCI-DAS 4020/12 ADChave an impedance of 1 MΩ. The signals are sampled at 10 MHz with a resolution of12 bit between -1 V and +1 V, i.e. a minimum voltage of 0.488 mV can be detected.After analog-to-digital conversion, the data is stored in an ASCII file that is processedoff line with the previously described MATLAB code to find the IQ symbols and thecalculate the corresponding SER. A six-port calibration is applied to all measurements.

The measurements of SP1500 and SP40 are performed with the following parameters:

• IF frequency: fIF = 333 MHz

• LO frequency: fLO = 1.5 GHz

• symbol rate: fsymb = 1 MHz

• sampling rate: fsample = 10 MHz

To avoid the use of additional baseband amplifiers, the output levels of the RF andLO signal generators need to be high enough to stay above the resolution and mini-mum detectable voltage of the ADC. With the relatively high RF power that is needed

116


to meet this requirement, the small signal approximation is not valid any longer. Infact, this non-linear operation mode of the diode mixers produces higher order termsthat take away energy and degrade the linear behavior. According to the non-linearadditive mixing theory, there is not only a non-linear relationship between RF ampli-tude and baseband amplitude, but also an additional intermodulation product thatstems from the modulated RF signal. The effects of a non-linear diode operationare basically the same as for a non-linear amplifier (compression point and interceptpoint). It was found experimentally that the six-port theory described in Chap. 2 isstill accurate enough with an RF power of approximately 0 dBm at the RF port of thesix-port receiver (a maximum power of -6 dB can then reach the output ports of thesix-port receiver which is approximately 100 mV at the 50 Ω termination). A strongnon-linear influence would degrade the signal constellation. Furthermore, the uniformdistribution of noise in the IQ constellation demonstrates that the large RF powerdoes not affect the experimental results. The focus is on an empirical verification ofthe bandwidth of the receiver. More investigations and a detailed analysis of noise insix-port receivers could be the scope of a future work.

5.4.2 General Dependency of RF and LO Power on Reception

The major limitation of the measurement set up is given by the resolution of theADC. Fig. 5.13 shows the IQ constellation of the demodulated measured signals forequal RF and LO powers from +4.1 dBm to -12.5 dBm. With the relatively largerRF power in Fig. 5.13(a), the mixing process also produces higher order terms causedby the non-linearity of the diode. However, it can be seen that the receiver can stillbe calibrated and a perfect IQ constellation can be found. A perfect IQ constellationwith minimum influences of the ADC quantization noise is still found for an RF powerof -6.1 dBm. An RF signal power of approximately 0 dBm has been chosen for theevaluation of the six-port receiver. At this power, the non-linearities have practicallyno influence on the IQ constellation. The measured IQ constellation for small Eb/N0

ratios are very smooth and uniform when the six-port receiver operates in its idealrange. This indicates the accurateness of this presumption.

By decreasing the input powers, the noise in the signal constellation increases andshows the discrete states caused by the quantization noise of the ADC. This is verystrongly pronounced at PRF = PLO = -12.5 dBm ( Fig. 5.13(d)). This quantizationnoise can be found in some of the BER measurements; this indicates an exceptionallyhigh attenuation caused by the six-port interferometer.

Fig. 5.14 shows measurement results with a low RF signal at different LO powers.One can see nearly the same IQ constellation with similar noise. This indicates that

117


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(a) PRF = PLO = +4.1 dBm

2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(b) PRF = PLO = -6.1 dBm

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)


(c) PRF = PLO = -8.8 dBm

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)


(d) PRF = PLO = -12.5 dBm

Fig. 5.13: Quantization noise of the ADC becomes visible when reducing RF andLO power (SP1500, fLO,RF = 1.5 GHz, Eb/N0 = 80 dB).

for very low RF powers, the noise is independent of the applied LO power. At therather large LO powers (or RF powers), the current that flows through the diode canno longer be neglected. In this region, the equation for the baseband signal (Eqn. 2.19on page 15) found in Chap. 2.4.1 is no longer valid. The high powers strongly influencethe matching of the diode detectors and lead to reflections. However for large powers,the maximum available gain (MAG) and the mixer conversion gain are similar andhave a constant value of approximately -5 dB. This is in good agreement with theconsiderations described in [130] and [28].

118


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(a) PRF = -15.3 dBm, PLO = 0.0 dBm

2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(b) PRF = -15.3 dBm, PLO=20.0 dBm

Fig. 5.14: IQ constellation with low RF and high LO power(SP1500, fLO,RF = 1.5 GHz, Eb/N0 = 80 dB).

5.4.3 General Noise Behavior

Noise is added to the system on the baseband level at the IQ signal generator. Becauseof the rather large RF and LO power, the noise that is introduced at the receiver canbe neglected due to the considerations explained earlier. Apart from minor deviationscaused by the six-port circuit, the IQ noise that is found in the demodulated signals inFig. 5.15 correspond directly to the Eb/N0 ratio. Perfect signal constellation is foundfor an Eb/N0 ratio of 50 dB (the SER is practically zero). Noise with an Eb/N0 ratioof 16 dB leads to a SER of 8.5×10−16. When further reducing the Eb/N0 ratio, thenoise significantly increases (the SER for Eb/N0 = 11 dB, the SER is 1.7×10−6, andfor an Eb/N0 ratio of 6 dB, the SER is 5.8×10−3 )

5.4.4 General Phase Behavior

The six-port simulations conducted in Chap. 5.2.2 have shown that both a frequencyoffset and a phase noise of the LO or RF signal lead to a circular movement of the IQvalues in the complex plane. The frequency of this circular movement equals exactlythe frequency offset. The phase noise behavior is slightly different – not in the circularmotion of the IQ signals but rather in their angular extension. The size of an arcsection is determined by the magnitude of the phase noise as well as the duration ofthe observation. A larger phase noise and a longer observation increase the length ofthe arc sections.

Fig. 5.16 shows the phase noise in the IQ constellation at an RF and LO frequency of24.0 GHz for a large Eb/N0 of 50 dB. Typically, the magnitude of the phase noise of

119


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(a) Eb/N0 = 50 dB

2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(b) Eb/N0 = 16 dB

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)


(c) Eb/N0 = 11 dB

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)


(d) Eb/N0 = 6 dB

Fig. 5.15: IQ signal constellation at different Eb/N0 ratios shows scattering depen-dency(SP1500, fLO,RF = 1.5 GHz, PRF = -0.16 dBm, PLO= -0.12 dBm).

local oscillators is larger for higher frequencies. The higher phase noise comes mainlyfrom the PLL that locks the RF onto a lower frequency reference signal. In additionto the relatively small steps that are caused by the high frequency component of thephase noise, there is circular movement of the IQ constellation. The frequency of thiscircular movement is related to the variance found from the signal spectrum. Becauseof the rather short recording time of the samples (1 ms), the noise that can be seen inFig. 5.16(a) is dominated by its high frequency component. Another indication is thatthe phase noise found in the IQ constellation is symmetric around the optimum IQcoordinates. Fig. 5.16(b) indicates that for smaller Eb/N0, not only is the amplitudenoise more pronounced, but the phase noise also leads to a larger deviation of thesignals from their optimum positions.

120


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(a) Eb/N0 = 50 dB

2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(b) Eb/N0 = 16 dB

Fig. 5.16: IQ signal constellation for two different Eb/N0 ratios shows phase noiseand scattering (SP40, fLO,RF = 24.0 GHz, PRF = -0.31 dBm, PLO= 0.0dBm).

5.4.5 Influences of In-Band Interferers

In-band interferers significantly influence the signal constellation and decrease theSER. This experiment has been conducted without any additional noise (Eb/N0 = 80dB).

Fig. 5.17(a) through (d) shows the resulting IQ constellations due to a sine waveinterferer (no modulation) at a frequency offset of 5.6 MHz away from the carrierfrequency of 1.5 GHz. The resulting circles around the optimum IQ positions increasewhen decreasing the ratio PRF /PINF between the RF power and the power of theinterferer PINF . At a PRF /PINF ratio of 6 dB (c) and -3 dBm (d), the resulting circlesshow accumulations at 5 different positions around the optimum IQ values. These 5different positions arise from the ratio of the symbol time to the offset frequency thatleads to exactly 5 different levels at the sampling instant.

5.4.6 Frequency Dependent SER Performance of Multi-Band

Receiver

One main difference between a full functional six-port receiver and the experimentalset up is depicted in Fig. 5.18. Fig. 5.18(a) shows the typical blocks in a receiverfront-end. Once the RF signal is coupled into the 50 Ω circuit, it needs to be amplifiedby an LNA as early as possible. Any loss before amplification directly degrades theSNR and Eb/N0 ratio. The high Q bandpass filter directly follows after this stage.With the (diode) mixer, the signal is down converted into IF or baseband. Then, the

121


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

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nt

(Q)


(a) PRF /PINF = 21 dBm

2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


(b) PRF /PINF = 15 dBm

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)


(c) PRF /PINF = 6 dBm

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)


(d) PRF /PINF = -3 dBm

Fig. 5.17: Influence of an in-band interferer with different power at a frequencyoffset of 5.6 MHz (SP1500, fLO,RF = 1.5 GHz, Eb/N0 = 80 dB).

signal is low-pass filtered at the IF or baseband level and is finally demodulated. Ina full receiver, the LNA and BPF have a noise factor and they additionally degradethe SNR from the antenna. In fact, the noise floor rises more and more as the signaltravels further through the various blocks in the receiver front-end. Therefore, BERperformance is a significant parameter for the evaluation of the receiver.

The idealized experimental six-port receiver set up is shown in Fig. 5.18(b). The noisewith an Eb/N0 ratio is added to the signal at the baseband level by the SMU200. Thesignal is attenuated by the six-port interferometer by at least 6 dB. Eventually, theRF signal with its Eb/N0 ratio reaches the diode of the detectors. This added noiseis not equivalent to the situation we have in a real six-port receiver where the noiseis added by the real physical behavior of the channel and the blocks in the front-end.What has been realize with this experimental set up is the application of an Eb/N0

122


Eb/N0DEMOD

LOLPFLNA BPF

SER

diode

(a) Typical blocks in a full six-port receiver front-end

Eb/N0DEMOD

LOLPF

SER

six-portinterferometer

diode

(b) Blocks included in the experimental six-port receiver set up (onlyone path shown)

Fig. 5.18: Typical blocks in a full six-port receiver vs. experimental set up.

ratio that is added onto a large RF signal. An equivalent signal would typically befound directly before the mixer stage in conventional receivers. Once there, the diodemixer does not add significant noise to the system at such large LO and RF powers.Therefore, with an optimum six-port interferometer (with optimum signal attenuationof 6 dB and uniform phase shifts), the measured resulting BER is very close to itstheoretical expectation.

With these considerations, we understand that the experimental set up analyzes themulti-band issue of the receiver with minimal influence from other typical receiverblocks. However, the noise of the RF and LO is still present and influences the BER,especially at higher frequencies. The other limiting factor is the resolution of the ADC(0.488 mV). When the signal attenuation by the six-port interferometer becomes toolarge, quantization noise can be seen in the IQ constellation.

In the following, the frequency dependent SER performance of the two different six-port interferometers will be presented. One set of diode detectors is used over theentire frequency range. The figures for each frequency will contain two graphs: onegraph that shows the SER over Eb/N0 ratio, and a second graph that exemplarilyshows the corresponding clouds in the IQ constellation for a certain Eb/N0 ratio. Theexperimental data that is collected and stored from each measurement is processedwith the MATLAB program as described earlier. A six-port calibration is applied. Inthe frequency range from 1.075 GHz to 40.0 GHz, significant frequencies are chosen:

SP1500: 1.075 GHz, 1.5 GHz, 2.16 GHz, 2.4 GHz

SP40: 2.4 GHz, 5.2 GHz, 10.0 GHz, 17.0 GHz, 20.0 GHz,24.0 GHz, 30.0 GHz, 37.0 GHz, 40.0 GHz

123


The 1.5 GHz Six-Port (SP1500)

Fig. 5.19 through Fig. 5.22 show the SER measurement results of the 1.5 GHz six-portinterferometer. The SP1500 performs very well at its design center frequency of 1.5GHz where the measurement results are very close to the theoretical expectations.The reason for their good matching has been discussed earlier: it is mainly due to thefact that the RF power with a certain Eb/N0 ratio is relatively high and the six-portreceiver itself does not add noise to it. The clouds in the IQ constellation are smoothand uniform which indicates that there is no distortion due to non-linearities or quan-tization noise.

100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bol err

or

rate

(S

ER

)

Eb/N0 [dB]

measurementtheory

(a) Symbol error rate vs. Eb/N0

2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)


(b) IQ constellation at Eb/N0= 8 dB

Fig. 5.19: Symbol error rate performance of SP1500 at fLO,RF = 1.5 GHz

(PRF = -0.16 dBm, PLO= -0.12 dBm).

The remaining 3 graphs for SP1500 are chosen according to critical frequencies thathave been found in the S-parameter measurements of Chap. 4.3. This will demon-strate the bandwidth limits of the interferometer circuit. The first critical frequencyis 1.075 GHz. We can see in the S-parameters of SP1500 (Fig. 4.13(b)) that, at thisfrequency, the transmission loss is below 10 dB from the RF port to ports 3,5, and6 and below -20 dB to port 4. In addition to the high signal attenuation, there is apole in the phase shift from the RF port to port 4 (Fig. 4.16). The IQ constellationat this frequency for an Eb/N0 ratio of 11 dB shows quantization noise. This indicatesthat the signal that comes to the diode detectors and finally to the ADC must be verysmall. Therefore, the SER performance is far below its optimum performance at 1.5GHz.

124


100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bo

l e

rro

r ra

te (

SE

R)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)




(PRF = -0.03 dBm, PLO= -0.12 dBm).

100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bol err

or

rate

(S

ER

)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)




(PRF = -0.31 dBm, PLO= -0.11 dBm).

Similar critical values are also found at 2.16 GHz where the phase information atports 4 and 6 is identical. This disadvantageous behavior of the interferometer circuitleads to a strong distortion of the IQ values and a rather bad SER performance. Atthe frequency of 2.4 GHz, the situation becomes better again as there is no phaseproblem. However, the high attenuation of the interferometer still has effects on signalconstellation and SER performance.

125


100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bo

l e

rro

r ra

te (

SE

R)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)




(PRF = -0.22 dBm, PLO= -0.19 dBm).

The 2 GHz through 25 GHz Six-Port (SP40)

The SER measurement results of SP40 are depicted in Fig. 5.24 through Fig. 5.33.Similar SER performance of the six-port SP40 is found at 5.2 GHz, 10 GHz, 17 GHz,20 GHz, and 24 GHz. When looking at the S-parameters, one can see that the atten-uation loss stays below 8 dB in this frequency range (Fig. 4.21(b)). The phase shiftsare very smooth and meet the 90 requirement very well (Fig. 4.23(b)). At higherfrequencies, the attenuation loss increases and, therefore, the SER performance de-creases. At 40 GHz, the attenuation is large enough to make ADC quantization noiseappear in the IQ constellation. The figures are listed below.

When using the SP40 at frequencies below 3 GHz, the attenuation from the inputto the output ports varies greatly and power is unevenly distributed (see Fig. 4.21(a)and (b)). The requirement for uniform phase shifts at the output ports is still fulfilled(Fig. 4.23(b)). The SER at 2.0 GHz is much higher compared to all other frequenciesdue to the disadvantageous attenuation.

Overview of Frequency Dependent BER Performance

One important result of the frequency dependent BER measurement is that both six-ports can be calibrated and give results both in and beyond their specified frequencyrange. All single SER over Eb/N0 graphs from Fig. 5.19 to Fig. 5.32 are included inan overview in Fig. 5.23 for SP1500 and SP40.

126


SP1500 shows its best performance at its designed center frequency of 1.5 GHz. Whenoperating in its critical frequencies, the six-port receiver can still function but the SERends up decreasing significantly.

The other six-port, SP40, shows a good SER performance for frequencies between 5.2GHz and 24 GHz which is predicted from S-parameter measurements. At lower andhigher frequencies, the SER is decreased; this is in agreement with the S-parametermeasurement results. However, the SP40 can be calibrated over the entire frequencyrange from 2 GHz to 40 GHz.

100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bol err

or

rate

(S

ER

)

Eb/N0 [dB]

1.5 GHz1.075 GHz2.16 GHz2.4 GHz

theory

(a) SP1500

100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bol err

or

rate

(S

ER

)

Eb/N0 [dB]

2.0 GHz2.4 GHz5.2 GHz10 GHz17 GHz20 GHz24 GHz30 GHz37 GHz

theory

(b) SP40

Fig. 5.23: Symbol error rate vs. Eb/N0 ratio for all frequencies.

100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bol err

or

rate

(S

ER

)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)




(PRF = -0.03 dBm, PLO= -0.14 dBm).

127


100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bo

l e

rro

r ra

te (

SE

R)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)




(PRF = -0.57 dBm, PLO= -0.20 dBm).

100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bol err

or

rate

(S

ER

)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)




(PRF = -0.04 dBm, PLO= -0.32 dBm).

128


100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bo

l e

rro

r ra

te (

SE

R)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)




(PRF = -0.36 dBm, PLO= -0.2 dBm).

100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bol err

or

rate

(S

ER

)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)




(PRF = -0.31 dBm, PLO= -0.06 dBm).

129


100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bo

l e

rro

r ra

te (

SE

R)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)




(PRF = -1.29 dBm, PLO= -1.14 dBm).

100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bol err

or

rate

(S

ER

)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)




(PRF = -0.31 dBm, PLO= 0.0 dBm).

130


100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bo

l e

rro

r ra

te (

SE

R)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)




(PRF = -1.38 dBm, PLO= -0.64 dBm).

100

10−3

10−6

10−9

10−12

15 10 5 0

sym

bol err

or

rate

(S

ER

)

Eb/N0 [dB]

measurementtheory


2

1

0

−1

−2 2 1 0−1−2

quadra

ture

com

ponent

(Q)




(PRF = -0.53 dBm, PLO= -4.6 dBm).

131


2

1

0

−1

−2 2 1 0−1−2

qu

ad

ratu

re c

om

po

ne

nt

(Q)


Fig. 5.33: IQ constellation of SP40 at fLO,RF = 40.0 GHz for Eb/N0= 16 dB(SER = 8.1 · 10−6, PRF = -2.88 dBm, PLO= -4.2 dBm).

5.5 Alternative Applications of the Multi-Port Principle

Apart from network analyzers (or reflectometers) and communications receivers, themulti-port principle offers several other application areas. One interesting and promis-ing approach is a smart antenna system that uses a total of four output ports to mea-sure the direction of arrival, as well as amplitude and phase of a mm-wave signal at thesame time [121]. Other applications are radar [122][123][127][125][124] and distancemeasurement devices [126]. In these fields, the six-port principle is very attractive asit is a cost-effective method for down conversion of mm-waves.

132

6 Conclusion and Perspectives

This thesis presents a novel receiver architecture that covers the very large frequencyrange from 1 GHz to 40 GHz and is suitable for multi-band, multi standard use. Thehardware design of the novel receiver becomes possible by combining RF MEMS withmulti-port technology. RF MEMS is an emerging technology. New suitable switches forhigh power and low loss applications that are required in the reconfigurable hardwarereceiver front-end are designed, fabricated, and thoroughly evaluated. The receiverconcept makes use of the SDR principle which shifts traditional hardware tasks (suchas demodulation) into the digital domain. The novel architecture is very flexible andnot limited to today’s standards. The combination of popular frequencies around 1GHz with frequencies at 17 GHz, 24 GHz, and 37 GHz will become very interestingin the future when new standards for these high carrier frequencies are established.Frequencies above 10 GHz are envisaged, as large bandwidths are available. A smartplatform that can handle standards in the entire frequency spectrum and is recon-figurable on the fly offers great advantages such as avoiding complex and expensivehardware.

The summary of this thesis is presented in the following. Furthermore, the maininnovations of this work and its contributions to the scientific community are listed.Finally, the thesis ends with possible extensions of this work and an outlook for futureinvestigations.

6.1 Summary

In this work, a solution has been presented that offers the ability to receive commu-nications standards over a frequency range from 1 GHz to 40 GHz. The thesis coversthe entire development process of the novel receiver front-end: from the theoreticalbackground of multi-port receivers and the specially developed RF MEMS switches tothe performance evaluation of the reconfigurable platform.

Chapter 1 outlines the motivation and need for a novel multi-band, multi-standardreceiver platform that can handle current and future communications standards. Themotivation is a cost-effective hardware solution that is reconfigurable on the fly andupgradeable. Software reconfigurability (implemented with SDR) and hardware re-

133


configurability (implemented with MEMS) is a prerequisite for the implementation ofthese demands.

Chapter 2 starts with a short introduction to SDR in the context of multi-port re-ceivers. Afterwards, the chapter covers a theoretical description of the semiconductordiode, the diode (power) detector, the theory of additive mixing, and the frequencyconversion processes in multi-port receivers. At the end of the chapter, a theoreti-cal basis of multi-port receivers is established that describes the diode detectors ina dynamic environment. This includes a mathematical formulation of the resultingbaseband signal behind the diode detectors that stems from the modulation of the RFsignal.

Chapter 3 presents the latest achievements in RF MEMS technology. New signalrouting structures that are needed in the multi-band receiver front-end are developedand characterized. The novel structures are a SPST toggle switch, a SPDT switch, andan RF cross. The chapter covers the theoretical background of the mechanical prop-erties of the movable parts, as well as the required mechanical and electro-magneticsimulations, the fabrication process, and the achieved measurement results.

Chapter 4 commences with the description of possible designs of multi-port inter-ferometers. The chosen six-port interferometer for the center frequency of 1.5 GHzconsists of three hybrid couplers and one power divider. Its design process is thor-oughly explained. Together with measurement results of the broadband 2 GHz through25 GHz six-port interferometer, this helps the reader to grasp the functional principleof the interferometer circuit. The second part of this chapter is the application of theMEMS in the interferometer circuit. Suitable system level simulations are performedthat include real S-parameter measurement results of the RF MEMS and both six-portinterferometers. It is found that the insertion of the RF MEMS into the RF lines ofthe interferometer leads to only minor degradation of the signal power and signal phase.

Chapter 5 rounds off the performance evaluation of the multi-band, multi-standardreceiver in the signaling context. Symbol error rate performance is chosen as a suitableevaluation method. The chapter starts with time domain simulations of the six-portreceiver that includes AWGN and phase noise. Finally, SER measurements at differ-ent noise levels (Eb/N0 ratios) are performed at selected frequencies between 1 GHzand 40 GHz. The measurement results verify the simulated, theoretical behavior.Performance limitations are indicated.

134

6.2 Main Achievements and Outlook

6.2 Main Achievements and Outlook

This section summarizes the main achievements and contributions of the work pre-sented in this thesis for further development of RF MEMS and multi-band SDRsbased on the multi-port principle.

RF MEMS:

• Stable fabrication processes for new types of RF MEMS: an ohmic contact switch(toggle switch), an SPDT switch, and an RF cross. All devices can be fabricatedon standard Si wafers with standard Si processes (CMOS compatibility).

• RF performance of a new capacitive membrane switch: up to 40 GHz, the switch-able attenuation is 35 dB with an insertion loss smaller than 0.4 dB. The powerhandling is at least 1 W at 30 GHz.

• RF performance of the toggle switch: up to 30 GHz, the insertion loss is below0.5 dB, the return loss is above 25 dB, and the isolation is above 14 dB. Thepower handling is at least 1 W at 30 GHz, 2 W at 18 GHz, and 2.5 W at 15GHz.

• RF performance of the SPDT switch: up to 30 GHz, the insertion loss is below0.5 dB, the return loss is above 22 dB, and the isolation is above 22 dB.

• RF performance of the RF cross: up to 30 GHz, the insertion loss of the twosignal routes are below 1 dB and 1.5 dB respectively.

• A detailed reliability and performance analysis of the toggle and capacitive shuntairbridge is given. This covers lifetime evaluation by switch cycles measurement,DC contact resistance, and temperature dependency.

• The measured RF performance of the MEMS is superior to conventional pindiode switches.

SDR multi-port receivers:

• A comprehensive and easier to understand mathematical description of multi-port receivers is given. The approach combines the theory of additive mixingwith multi-port theory and gives insight into the frequency conversion processes.It becomes clear why a multi-port requires at least three independent voltagemeasurements at the output ports.

• A new hardware for a six-port front-end that covers a very large bandwidth isdeveloped and evaluated. Advantages as well as disadvantages are discussed.

135


• The understanding of multi-port theory and the functional principle of the in-terferometer circuit was improved by giving a detailed description of the phaserelations in the interferometer circuit.

• A functional proof of the multi-band, multi standard receiver is given by SERmeasurements with different Eb/N0 ratios over a frequency range from 1 GHzto 40 GHz (at selected frequencies). SER measurements covering such a largefrequency range in one single SDR receiver have not been published before.

The work presented in this thesis can be extended in several ways. The fabricatedMEMS show promising results and a stable fabrication process was established in thiswork. However, the yield from a wafer needs to be improved in order to become eco-nomically interesting. Concerning their application in the reconfigurable front-end,the MEMS elements need to be put into a housing, or a six-port structure needs to beimplemented on a Si wafer. As for the multi-port receiver, a detailed noise evaluationof semiconductor diodes is still missing in the literature. To accomplish a reliablenoise evaluation, a complete six-port receiver that includes LNA, BPF, and basebandamplifiers needs to be fabricated. This will also allow the characterization of the sen-sitivity and the dynamic range.

New insights into multi-port receivers are presented within this thesis. The new multi-band, multi-standard receiver offers an interesting approach and is a promising candi-date for future mm-wave applications. The cut-off frequency of diodes is far above 100GHz. With the technology that is available today, this makes diodes the only possibil-ity for up and down conversion of such large frequencies. On the other hand, the diodesalso limit the use of the multi-port as a powerful communications receiver. Diodes havean increased conversion loss when compared to conventional double-balanced mixers(Gilbert cell). This reduces the capability of detecting very weak signals (i.e. decreasedsensitivity). The limitation is mainly given by the fact that, for a low conversion loss,diode detectors need a relatively large input signal. The demand for a large inputsignal, which is much larger than the signal input powers needed in transistor basedmixers, requires a higher amplification of the RF signal by an LNA. This increasedamplification can become very problematic at lower frequencies where frequency bandsare very close together and interferers are present. In such environments, the LNAcan easily be driven into its non-linear region by interferers. Low loss bandpass filterswith steep flanks that could be applied before the LNA stage are not available today.However, MEMS based resonators could help in this field as well. Today, almost nointerferers are expected at higher carrier frequencies, as there are no mass market ap-plications in the ISM bands. This area offers an interesting field for the application ofmulti-port receivers.

136

List of Figures

2.1 Different RF front-ends for SDR platforms . . . . . . . . . . . . . . . . 72.2 Multiplicative versus additive mixing . . . . . . . . . . . . . . . . . . . 82.3 Equivalent circuit of Schottky diode and its DC characteristics . . . . . 92.4 Simple power detector and its characteristics . . . . . . . . . . . . . . . 112.5 Functional principle of multi-port receiver . . . . . . . . . . . . . . . . 122.6 Process of additive mixing . . . . . . . . . . . . . . . . . . . . . . . . . 142.7 Process additive mixing in multi-port receivers . . . . . . . . . . . . . . 162.8 Comparison of complex and real frequency conversion . . . . . . . . . . 192.9 Spectral properties of complex and real frequency conversion . . . . . . 202.10 Spectrum after complex and real down conversion . . . . . . . . . . . . 212.11 Principle of IF sampling . . . . . . . . . . . . . . . . . . . . . . . . . . 222.12 Baseband spectrum after direct frequency conversion . . . . . . . . . . 222.13 Spectrum of the additive mixing process in multi-port receivers . . . . 23

3.1 Electrical models of the MEMS switches . . . . . . . . . . . . . . . . . 283.2 Three-dimensional view of the shunt airbridge switch . . . . . . . . . . 373.3 Cross-sectional view of the shunt airbridge switch . . . . . . . . . . . . 373.4 Shape function simulation of a shunt airbridge switch . . . . . . . . . . 383.5 Voltage deflection function of a shunt airbridge switch . . . . . . . . . . 383.6 Membrane displacement of the shunt airbridge switch . . . . . . . . . . 383.7 Simulated S-parameter of the shunt airbridge switch . . . . . . . . . . . 393.8 Three-dimensional view of the toggle switch . . . . . . . . . . . . . . . 403.9 Cross-section view of the toggle switch . . . . . . . . . . . . . . . . . . 403.10 Mechanical displacement simulation of the toggle cantilever . . . . . . . 433.11 Membrane displacement of the toggle cantilever . . . . . . . . . . . . . 433.12 Electric field simulation of the regular toggle switch . . . . . . . . . . . 453.13 S-parameter simulation results of the toggle switch . . . . . . . . . . . 453.14 Three-dimensional view of the SPDT switch . . . . . . . . . . . . . . . 463.15 S-parameter simulation of the SPDT switch . . . . . . . . . . . . . . . 473.16 Three-dimensional view of the RF cross with electric field . . . . . . . . 483.17 S-parameter simulation of the RF cross . . . . . . . . . . . . . . . . . . 493.18 Process flow for the fabrication of the MEMS . . . . . . . . . . . . . . 503.19 SEM micrograph of the shunt airbridge switches . . . . . . . . . . . . . 523.20 S-parameter measurement results of the shunt airbridge switches . . . . 53

137

List of Figures

3.21 SEM micrographs of different toggle switches . . . . . . . . . . . . . . . 543.22 Measurement results of different toggle switches . . . . . . . . . . . . . 553.23 SEM micrograph of the SPDT switches . . . . . . . . . . . . . . . . . . 563.24 Simulation and measurement results of the SPDT switch . . . . . . . . 583.25 SEM micrograph of the RF cross . . . . . . . . . . . . . . . . . . . . . 593.26 Measurement results of the RF cross . . . . . . . . . . . . . . . . . . . 603.27 SEM micrograph of the toggle switch . . . . . . . . . . . . . . . . . . . 613.28 Displacement of the toggle switch cantilever (at position A) . . . . . . 623.29 Displacement of the toggle switch cantilever (at position B) . . . . . . . 623.30 Membrane velocities of the shunt airbridge membrane . . . . . . . . . . 633.31 Measurement setup for the RF switching time measurement . . . . . . 633.32 Switching times for RF power of the toggle switch . . . . . . . . . . . . 643.33 Self actuation of the toggle and shunt airbridge switch due to RF power 653.34 Experimental set up for switch cycle measurement . . . . . . . . . . . . 663.35 Actuation voltage vs. temperature of a toggle switch . . . . . . . . . . 673.36 Activation voltage vs. temperature of a shunt airbridge switch . . . . . 68

4.1 S-parameter simulation results of the 1.5 GHz power divider . . . . . . 714.2 Possible designs for five- and six-port interferometer circuits . . . . . . 724.3 Photograph of SP1500 . . . . . . . . . . . . . . . . . . . . . . . . . . . 734.4 Electric and magnetic field components in a microstrip line . . . . . . . 754.5 The dimensions and geometry of the power divider . . . . . . . . . . . 764.6 S-parameter simulation results of the 1.5 GHz power divider . . . . . . 774.7 The dimensions and geometry of the quadrature hybrid . . . . . . . . . 784.8 S-parameter simulation results of the 1.5 GHz quadrature hybrid . . . 784.9 Momentum layout with dimensions of SP1500 . . . . . . . . . . . . . . 794.10 Simulated and measured return loss of SP1500 . . . . . . . . . . . . . . 804.11 Simulated and measured isolation of SP1500 . . . . . . . . . . . . . . . 814.12 Simulated and measured transmission of SP1500 to port 3 . . . . . . . 824.13 Measured transmission of SP1500 from LO and RF port all output ports 824.14 Absolute phase shifts between the LO and RF ports and port 3 . . . . 834.15 Measured phase shifts between the LO and RF ports and all output ports 844.16 Measured phase differences between the LO and RF input signals . . . 854.17 Photograph of SP40 . . . . . . . . . . . . . . . . . . . . . . . . . . . . 854.18 S-parameter measurement result of the broadband power divider . . . . 864.19 S-parameter measurement results of the broadband quadrature hybrid . 874.20 Return loss and isolation of SP40 . . . . . . . . . . . . . . . . . . . . . 884.21 Measured transmission of SP40 . . . . . . . . . . . . . . . . . . . . . . 894.22 Measured phase shifts of SP40 . . . . . . . . . . . . . . . . . . . . . . . 904.23 Measured phase differences between input and output ports SP40 . . . 914.24 Application scenario for different RF MEMS . . . . . . . . . . . . . . . 924.25 The reconfigurable multi-band six-port receiver front-end . . . . . . . . 934.26 The reconfig. multi-band receiver front-end (SP1500 contacted) . . . . . 954.27 Return loss and isolation of the multi-band front-end (SP1500 contacted) 96

138

List of Figures

4.28 Isolation of the input ports of SP40 (SP1500 contacted) . . . . . . . . . 964.29 Transmission of SP1500 (SP1500 contacted) . . . . . . . . . . . . . . . 974.30 Phase differences of SP1500 (SP1500 contacted) . . . . . . . . . . . . . 974.31 The reconfig. multi-band receiver front-end (SP40 contacted) . . . . . . 984.32 Return loss and isolation of the multi-band front-end (SP40 contacted) 994.33 Isolation of the input ports of SP1500 (SP40 contacted) . . . . . . . . . 1004.34 Transmission of SP40 (SP40 contacted) . . . . . . . . . . . . . . . . . . 1004.35 Phase differences of SP40 (SP40 contacted) . . . . . . . . . . . . . . . . 101

5.1 Block diagram of CppSim six-port receiver simulation . . . . . . . . . . 1055.2 Snapshot of the sent in-phase (I) and quadrature (Q) sequence . . . . . 1065.3 Voltages at the output of the power detectors . . . . . . . . . . . . . . 1065.4 Symbols used for calibration . . . . . . . . . . . . . . . . . . . . . . . . 1075.5 Power levels of the calibration sequence . . . . . . . . . . . . . . . . . . 1075.6 Snapshot of the received in-phase (I) and quadrature (Q) sequence . . . 1085.7 Demodulated I and Q symbols . . . . . . . . . . . . . . . . . . . . . . . 1085.8 Influence of AWGN in the signal path for the six-port calibration . . . 1115.9 Influence of AWGN in the signal path for the five-port calibration . . . 1125.10 Influence of phase and frequency imbalances on IQ constellation . . . . 1135.11 Characteristics of the Schottky power detectors . . . . . . . . . . . . . 1145.12 Measurement set up for the six-port receiver characterization . . . . . . 1155.13 Quantization noise of the ADC . . . . . . . . . . . . . . . . . . . . . . 1185.14 IQ constellation with high LO power . . . . . . . . . . . . . . . . . . . 1195.15 IQ signal constellation at different Eb/N0 ratios . . . . . . . . . . . . . 1205.16 IQ signal constellation with increased phase noise . . . . . . . . . . . . 1215.17 Influence of in-band interferer at a frequency offset of 5.6 MHz . . . . . 1225.18 Typical blocks in a real receiver vs. experimental six-port receiver set up1235.19 SER vs. Eb/N0 of SP1500 at 1.5 GHz . . . . . . . . . . . . . . . . . . . 1245.20 SER vs. Eb/N0 of SP1500 at 1.075 GHz . . . . . . . . . . . . . . . . . 1255.21 SER vs. Eb/N0 of SP1500 at 2.16 GHz . . . . . . . . . . . . . . . . . . 1255.22 SER vs. Eb/N0 of SP1500 at 2.4 GHz . . . . . . . . . . . . . . . . . . . 1265.23 SER vs. Eb/N0 ratio for all frequencies . . . . . . . . . . . . . . . . . . 1275.24 SER vs. Eb/N0 of SP40 at 2.0 GHz . . . . . . . . . . . . . . . . . . . . 1275.25 SER vs. Eb/N0 of SP40 at 2.4 GHz . . . . . . . . . . . . . . . . . . . . 1285.26 SER vs. Eb/N0 of SP40 at 5.2 GHz . . . . . . . . . . . . . . . . . . . . 1285.27 SER vs. Eb/N0 of SP40 at 10.0 GHz . . . . . . . . . . . . . . . . . . . 1295.28 SER vs. Eb/N0 of SP40 at 17.0 GHz . . . . . . . . . . . . . . . . . . . 1295.29 SER vs. Eb/N0 of SP40 at 20.0 GHz . . . . . . . . . . . . . . . . . . . 1305.30 SER vs. Eb/N0 of SP40 at 24.0 GHz . . . . . . . . . . . . . . . . . . . 1305.31 SER vs. Eb/N0 of SP40 at 30.0 GHz . . . . . . . . . . . . . . . . . . . 1315.32 SER vs. Eb/N0 of SP40 at 37.0 GHz . . . . . . . . . . . . . . . . . . . 1315.33 SER vs. Eb/N0 of SP40 at 40.0 GHz . . . . . . . . . . . . . . . . . . . 132

139

List of Figures

140

Abbreviations

AC alternating currentADC analog-digital converterADS Advanced Design System (Agilent)AM amplitude modulationANSYS finite element computer simulation programASCII American Standard Code for Information InterchangeAWGN additive white gaussian noiseBPF bandpass filterCAD computer-aided designCDMA code division multiple accessCMOS complementary metal-oxide-semiconductorCPD critical point dryingCPW coplanar waveguideCW constant waveDAB digital audio broadcastDAC digital-to-analog converterDC direct currentDSP digital signal processorsDUT device under testDVB digital video broadcastEM electromagneticEMPIRE fast three dimensional electromagnetic simulation programESA European Space AgencyFDTD finite difference time domainFM frequency modulationFPGA field-programmable gate arraysGALILEO european global satellite navigation systemGHZ gigahertzGPS Global Positioning SystemGSM Global system for mobile communicationsI in-phaseIF intermediate frequencyIQ in-phase/quadrature

141

List of Figures

ISM industrial, scientific, and medicalLAN local area networkLNA low noise amplifiersLO local oscillatorLPF low pass filterMAG maximum available gainMATLAB technical computing softwareMEMS micro electro-mechanical systemMIT Massachusetts Institute of TechnologyMMIC monolithic microwave integrated circuitPA power amplifierPC personal computerPCI Peripheral Component InterconnectPD power detectorsPECVD plasma enhanced chemical vapor depositionPFD phase frequency detectorPIN positive/intrinsic/negative (diode profile)PLL phase locked loopPR photo resistQ quadratureQAM quadrature amplitude modulationQPSK quadrature phase shift keyingRF radio frequencyRX/TX receive/transmitS ScatteringSDARS satellite digital audio radio servicesSDR software defined radioSEM scanning electron microscopeSER symbol error rateSNR signal to noise ratioSP1500 six-port interferometer (center frequency 1.5 GHz)SP40 six-port interferometer (2 -25 GHz)SPDT single pole double throwSPST single pole single throwTE transverse electricTEM transverse electromagnetic modeTM transverse magneticTV televisionVCO voltage controlled oscillator

142

Bibliography

[1] J. Mitola: “Software Radio Architecture,” Wiley & Sons, ISBN: 0-47138-492-5,2000

[2] J.-F. Luy, T. Mack: “Towards Software Configurable Millimeter Wave Architec-tures,” Proceedings of Radio and Wireless Conference (RAWCON), 2003

[3] J.-F. Luy, T. Muller, T. Mack, A. Terzis: “Configurable RF Receiver Architec-tures,” IEEE Microwave Magazine, Vol. 5, No. 1, pp. 75-82, March 2004

[4] G. F. Engen, C. A. Hoer: “Application of an Arbitrary 6-Port Junction to PowerMeasurement Problems,” IEEE Transactions on Instrumentation and Measure-ment, Vol. IM-21, pp. 470-474, November 1972

[5] J.-U. Jurgensen, D. Krupezevic, M. Ratni, Z. Wang: “Baseband Aspects of a Di-rect Conversion Receiver Concept Utilising Five-Port Technology,” Proceedingsof 2nd Karlsruhe Workshop on Software Radios (WSR), Karlsruhe, 2002

[6] M. Abe, N. Sasho, V. Brankovic, and D. Krupezevic: “Direct Conversion Re-ceiver MMIC Based on Six-Port Technology,” European Conference on WirelessTechnology (CNIT), Paris, October 2000

[7] F. R. Sousa, G. Neveux, B. G. Amante, B. Huyart: “Five-Port Junction: In theWay of General Public Applications,” Proceedings of 32nd European MicrowaveConference (EuMC), Milan, 2002

[8] J. Li, R. G. Bosisio, K. Wu: “A Six-Port Direct Digital Receiver,” Digest ofIEEE MTT Symposium, Vol. 3, pp. 1659-1662, San Diego, May 1994

[9] X. Xu, K. Wu, R. G. Bosisio: “Software Defined Radio Receiver Based on Six-Port Technology,” Digest of IEEE MTT-S, International Microwave Symposium,Vol. 2, Philadelphia, June 2003

[10] J. Li, R. G. Bosisio: “Computer and Measurement Simulation of a New DigitalReceiver Operating Directly at Millimeter-Wave Frequencies,” IEEE Transac-tions on Microwave Theory and Techniques, Vol. 43, No. 12, December 1995

143

Bibliography

[11] S. O. Tatu, E. Moldovan, K. Wu, R. G. Bosisio: “A New Direct Millimeter-WaveSix-Port Receiver,” IEEE Transactions on Microwave Theory and Techniques,Vol. 49, No. 12, December 2001

[12] S. O. Tatu, E. Moldovan, G. Brehm, R. G. Bosisio: “Ka-Band Direct DigitalReceiver,” IEEE Transactions on Microwave Theory and Techniques, Vol. 50,No. 11, November 2002

[13] S. O. Tatu, E. Moldovan, G. Brehm, K. Wu, R. G. Bosisio: “New Results onMMIC Six-Ports used in Ka Band Direct Conversion Receivers,” Digest of IEEEMTT-S, International Microwave Symposium, Philadelphia, June 2003

[14] T. Mack, K. M. Strohm, B. Schauwecker, J.-F. Luy: “RF MEMS Switchesand High Q Elements,” XXVII International Union of Radio Science (URSI),Maastricht, August 2002

[15] S. B. Cohn, N. P. Weinhouse: “An Automatic Microwave Phase-MeasurementSystem,” The Microwave Journal, February 1964

[16] G. F. Engen: “The Six-Port Reflectometer: An Alternative Network Analyzer,”IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-25, No. 12,pp. 1075-1080, December 1977

[17] C. A. Hoer, K. C. Roe: “Using an Arbitrary Six-Port Junction to Measure Com-plex Voltage Ratios,” IEEE Transactions on Microwave Theory and Techniques,Vol. MTT-23, No. 12, December 1975

[18] C. A. Hoer: “Performance of a Dual Six-Port Automatic Network Analyzer,”IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-27, No. 12,December 1979

[19] J. Hyyrylainen, L. Bogod, S. Kangasmaa, H.-O. Scheck, T. Ylamurto: “Six-PortConversion Receiver,” Proceedings of European Microwave Conference (EuMC),Jerusalem, 1997

[20] B. Schauwecker, K. M. Strohm, T. Mack, W. Simon, J.-F. Luy: “A Single-Pole-Double-Throw Switch Based on the Toggle Switch,” Electronics Letters, Vol. 39,No. 8, pp. 668-670, 2003

[21] T. Mack, A. Honold, J.-F. Luy: “An Extremely Broadband Software Config-urable Six-Port Receiver Platform,” Proceedings of European Microwave Con-ference (EuMC), Munich, October 2003

[22] M. H. Perrott: “CppSim Behavioral Simulator Package,” http://www-mtl.mit.edu/researchgroups/perrottgroup/tools.html

[23] R. Murty: “Software-Defined Reconfigurable Radios: Smart, Agile, Cognitive,and Interoperable,” Technology@Intel Magazine, July 2003

144

Bibliography

[24] T. Muller: “Direktabtastende Architekturen fur Hochfrequenzempfanger,” Dis-sertation, Technische Universitat Munchen (TUM), May 2002

[25] J. Hesselbarth, F. Wiedmann, B. Huyart: “Two New Six-Port ReflectometersCovering Very Large Bandwidths,” IEEE Transactions on Instrumentation andMeasurement, Vol. 46, No. 4, December 1997

[26] M. Ratni, D. Krupezevic, Z. Wang, J.-U. Jurgensen: “Broadband Digital DirectDown Conversion Receiver Suitable for Software Defined Radio,” 13th IEEE In-ternational Symposium on Personal, Indoor and Mobile Radio Communications,Vol. 1, pp 100-104, September 2002

[27] T. W. Crowe, R. J. Mattauch, H. P. Roser, W. L. Bishop, W. C. B. Peatman, X.Liu: “GaAs Schottky Diodes for THz Mixing Applications,” Proceedings IEEE,Vol. 80, No. 11, pp. 1827-1841, 1992

[28] S. A. Maas: “Microwave Mixers,” Artech House, Inc., ISBN: 0-89006-171-8, 1986

[29] Meinke, Gundlach: “Taschenbuch der Hochfrequenztechnik,” 5th Edition,Springer Verlag, ISBN: 3-540-54717-7, 1992

[30] D. M. Pozar: “Microwave and RF Design of Wireless Systems,” John Wiley &Sons, 1st edition, ISBN 0-471-32282-2, December 2000

[31] T. Hentschel: “The Six-Port as a Communication Receiver,” IEEE Transactionson Microwave Theory and Techniques, Vol. 53, No. 3, March 2005

[32] T. Mack, T. Eireiner: “Frequency Conversion Processes in SDR Front-Ends,”MTT-S International Microwave Symposium, Workshop: RF Aspects of Soft-ware Defined Radio, Long Beach, June 2005

[33] Y. Xu, R. G. Bosisio: “Four-Port Digital Millimetric Receiver,” Microwave andOptical Technology Letters, Vol. 22, No. 5, September 1999

[34] G. F. Engen: “Calibration of an Arbitrary Six-Port Junction for Measurement ofActive and Passive Circuit Parameters,” IEEE Transactions on Instrumentationand Measurement, Vol. IM-22, No. 4, December 1973

[35] G. F. Engen: “Calibrating the Six-Port Reflectometer by Means of Sliding Termi-nations,” IEEE Transactions on Microwave Theory and Techniques, Vol. MTT-26, No. 12, December 1978

[36] F. Wiedmann, B. Huyart, E. Bergeault, L. Jallet: “A New Robust Method forSix-Port Reflectometer Calibration,” IEEE Transactions on Instrumentation andMeasurement, Vol. 48, No. 5, December 1999

[37] M. Caron, F. Gagnon, N. Batani, D. J. Hindson, R. G. Bosisio: “ On Six-Port Junction Based K-Band Direct Receivers,” Second Ka-Band UtilizationConference, Florence, Italy, September 1996

145

Bibliography

[38] B. Schauwecker, K. M. Strohm, W. Simon, J. Mehner, J.-F. Luy: “A New Typeof High Bandwidth RF MEMS Switch Toggle Switch,” Journal of SemiconductorTechnology and Science, Special Issue on MEMS, Vol. 2, No. 4, pp. 237-245,December 2002

[39] W. Simon, B. Schauwecker, A. Lauer, A. Wien, I. Wolff: “EM Design of Broad-band RF Multiport Toggle Switches,” International Journal of RF and Mi-crowave Computer-Aided Engineering, Vol. 14, Issue 4 , pp. 329-337, June 2004

[40] W. Simon, A. Lauer, B. Schauwecker, A. Wien: “EM Design of an IsolatedCoplanar RF Cross for MEMS Switch Matrix Applications,” Silicon MonolithicIntegrated Circuits in RF Systems, pp. 162-165, Greinau, April 2003

[41] B. Schauwecker, K. M. Strohm, J.-F. Luy: “High Power RF-MEMS Switches,”IEEE MTT-S, International Microwave Symposium; Workshop notes: Advancesin RF MEMS: Components, Packaging Techniques, Reliability and Microphon-ics; Phoenics, May 2001

[42] W. Simon, B. Schauwecker, A. Lauer, A. Wien: “Designing a Novel RF MEMSSwitch for Broadband Power Applications,” Vol. 2, pp. 519-522, European Mi-crowave Conference (EuMC), Milan, 2002

[43] K. M. Strohm, B. Schauwecker, T. Mack, J.-F. Luy: “System Implications UsingRF MEMS,” Workshop WM1: Recent Developments in Si Micromachinig andMEMS for High Frequency Applications, Milan, November 2002

[44] A. Margomenos, D. Peroulis, K. J. Herrick, L. Katehi: “Silicon MicromachinedPackages for RF-MEMS Switches,” European Microwave Conference (EuMC),London 2001

[45] K. E. Petersen: “Micromechanical Membrane Switches on Silicon,” IBM Journalof Research and Development, Vol. 23, pp. 376-385, 1979

[46] H. Hosaka, H. Kuwano, K. Yanagisawa: “Electromagnetic Microrelays: Conceptsand Fundamental Characteristic,” Sensors Actuators A, Vol. 40, pp. 41-47, 1994

[47] W. P. Taylor, M. G. Allen, C. R. Dauwa1ter: “Batch Fabricated ElectromagneticMicrorelays,” Proceedings of 45th Relay Conference, 1997

[48] W. P. Taylor, M. G. Allen: “Integrated Magnetic Microrelays: Normally Open,Normally Closed, and Multi-Pole Devices,” Digest of International Conferenceon Solid-State Sensors and Actuators, pp. l149-1152, 1997

[49] H. A. C. Tilmans, E. Fullin, H. Ziad, M. D. J. Van de Peer, J. Kesters, E.Van Geffen, J. Bergqvist, M. Pantus, E. Beyne, K. Baert and F. Naso: “AFully-Packaged Electromagnetic Microrelay,” Digest of 12th IEEE InternationalConference on Micro Electro Mechanical Systems, pp. 25-30, 1999

146

Bibliography

[50] J. A. Wright, Y.-C. Tai: “Magnetostatic MEMS Relays for the Miniaturization ofBrushless DC Motor Controllers,” Digest of 12th IEEE International Conferenceon Micro Electro Mechanical Systems, pp. 594-599, 1999

[51] M.-A. Gretillat, F. Gretillat, N. F. de Rooij: “Micromechanical Relay with Elec-trostatic Actuation and Metallic Contacts,” Journal of Micromechanics and Mi-croengineering, Vol. 9, pp. 324-33l, 1999

[52] C. Sanders: “MCNC Thermally Actuated Microrelays,” News Release MEMSTechnology Application Center MCNC, November 1998

[53] M. Beck, M.M. Ahmed, C. J. Brierley, A. P. Needham, S. P. Marsh: “Mi-crowave Filters and Switches Produced using Micro-Machining Techniques,” In-ternational Microwave Symposium, Boston, 2000

[54] L. E. Larson, R. H. Hackett, R. F. Lohr: “Microactuators for GaAs-Based Mi-crowave Integrated Circuits,” Digest of 6th International Conference on Solid-State Sensors and Actuators, pp. 743-746, 1991

[55] J. J. Yao, M. F. Chang: “A Surface Micromachined Miniature Switch forTelecommunications Applications with Signal Frequencies from DC up to 4GHz,” Digest of 8th International Conference on Solid-State Sensors and Ac-tuators, pp. 384-387, 1995

[56] I. Schiele, J. Huber, C. Evers, B. Hillerich, F. Kozlowski: “MicromechanicalRelay with Electrostatic Actuation,” Digest of International Conference on Solid-State Sensors and Actuators, pp. 1165-1168, 1997

[57] H. J. De Los Santos, Y. H. Kao, A. L. Cajgoy, E. D. Ditmars: “Microwaveand Mechanical Considerations in the Design of MEMS Switches for AerospaceApplications,” Proceedings IEEE Aerospace Conference, Vol. 3, pp. 235-254,1997

[58] D. Hyman, A. Schmitz, B. Warneke, T. Y. Hsu, J. Lam, J. Brown, J. Schaffner,A. Walston, R.Y. Loo, G.L. Tangonan, M. Mehregany, J. Lee: “Surface Micro-machined RF MEMS Switches on GaAs Substrates” International Journal RFMicrowave CAE, Vol. 9, pp. 348361, August 1999

[59] D. Hyman, A. Schmitz, B. Warneke, T.Y. Hsu, J. Lam, J. Brown, J. Schaffner, A.Walston, R.Y. Loo, G.L. Tangonan, M. Mehregany, J. Lee: “GaAs-CompatibleSurface-Micromachined RF MEMS Switches,” Electronics Letters, Vol. 35, pp.224-226, 1999

[60] P. M. Zavracky, S. Majumder, N. E. McGruer: “Micromechanical Switches Fab-ricated using Nickel Surface Micromachining,” IEEE Journal of MEMS, Vol. 6,pp. 3-9, 1997

147

Bibliography

[61] S. Majumder, N. E. McGruer, P. M. Zavracky, G. G. Adams, R. H. Morrison,J. Krim: “Measurement and Modeling of Surface Micromachined, Electrostati-cally Actuated Microswitches,” Digest of International Conference on Solid-StateSensors and Actuators, pp. 1145-1148, 1997

[62] N. E. McGruer, P. M. Zavracky, S. Majumder, R. H. Morrison, G. G. Adams:“Electrostatically Actuated Microswitches; Scaling Properties,” Digest of SolidState Sensor and Actuator Workshop, pp. 132-5, 1998

[63] H. F. Schlaak, F. Amdt, M. Hanke: “Silicon-Microrelay – A Small Signal RelayWith Electrostatic Actuator,” Proceedings of 45th Relay Conference, pp. 10.1-10.7, 1997

[64] K. Suzuki, S. Chen, T. Marumoto, A. Ara, R. Iwata: “A Micromachined RFMicroswitch Applicable to Phased-Array Antennas,” Digest of IEEE MicrowaveTheory Techniques Symposium, pp. 1923-1926, 1999

[65] E. A. Sovero, R. Mihailovich, D. S. Deakin, J. A. Higgins, J. J. Yao, J. F.DeNatale, J. H. Hong: “Monolithic GaAs PHEMT MMICs Integrated with HighPerformance MEMS Microrelays,” Proceedings of IMOC, Rio de Janeiro, 1999

[66] J. B. Muldavin, G. M. Rebeiz: “30 GHz Tuned MEMS Switches,” Digest ofIEEE Microwave Theory and Techniques Symposium, pp. 1511-1514, 1999

[67] C. Goldsmith, T.-H. Lin, B. Powers, W.-R. Wu, B. Norvell: “MicromechanicalMembrane Switches for Microwave Applications,” Digest of IEEE MicrowaveTheory and Techniques Symposium, pp. 91-94, 1995

[68] C. Goldsmith, J. Randall, S. Eshelman, T.-H. Lin, D. Denniston, S. Chen, B.Norvell: “Characteristics of micromachined switches at microwave frequencies,”Digest of IEEE Microwave Theory and Techniques Symposium, pp. 1141, 1996

[69] Z. J. Yao, S. Chen, S. Eshelman, D. Denniston, C. Goldsmith: “MicromachinedLow-Loss Microwave Switches,” IEEE Journal of MEMS, Vol. 8, pp. 129-34,1999

[70] S. Pacheco, C. T. Nguyen, L. P. B. Katehi: “Micromachined Electrostatic K-band Switches,” Digest of IEEE MTT-S, International Microwave Symposium,pp. 1569-1572, 1998

[71] M. Sakata, Y. Komura, T. Seki, K. Kobayashi, K. Sano, S.Horiike: “Micro-machined Relay which Utilizes Single Crystal Silicon Electrostatic Actuator,”Digest of 12th IEEE International Conference on Micro Electro Mechanical Sys-tems, pp. 21-24, 1999

[72] K. M. Hiltmann, B. Schmidt, H. Sandmaier, W. Lang: “Development of Micro-machined Switches with Increased Reliability,” Digest of International Confer-ence on Solid-State Sensors and Actuators, pp. 1157-1160, 1997

148

Bibliography

[73] J. Drake, H. Jerman, B. Lutze, M. Stuber: “An Electrostatically ActuatedMicro-Relay,” Digest of 8th International Conference on Solid-State Sensors andActuators, pp. 380-383, 1995

[74] M. Admschik, S. Ertl, P. Schmid, P. Gluche, A. Floter, E. Kohn: “ElectrostaticDiamond Micro Switch,” Digest of 8th International Conference on Solid-SlateSensors and Actuators, pp. 1284-1287, 1999

[75] M.-A. Gretillat, P. Thiebaud, C. Linder, N. F. de Rooij: “Integrated CircuitCompatible Polysilicon Microrelays,” Journal of Micromechanics and Microengi-neering, Vol. 5, pp. 156-160, 1995

[76] J. Simon, S. Saffer, C. J. Kim: “A Micromechanical Relay with a Thermally-Driven Mercury Micro-Drop,” Proceedings of IEEE 9th Anniversary Interna-tional Workshop on Micro Electro Mechanical Systems, pp. 515-520, 1996

[77] S. Saffer, J. Simon, C. J. Kim: “Mercury-Contact Switching with Gap-ClosingMicrocantilever,” Proceedings of SPIE, Vol. 2882, pp. 204-208, Austin, October1996

[78] X. Q. Sun, K. R. Farmer, W. N. Carr: “A Bistable Microrelay based on Two-Segment Multimorph Cantilever Actuators“, Proceedings of IEEE 11th Anniver-sary International Workshop on Micro Electro Mechanical Systems, pp. 154-159,1998

[79] E. J. J. Kruglick, K. S. J. Pister: “Bistable MEMS Relays and Contact Charac-terization,” Digest of Solid State Sensor and Actuator Workshop, pp. 333-337,1998

[80] I. Schiele, B. Hillerich: “Comparison of Lateral and Vertical Switches for Appli-cation as Microrelays,” Journal of Micromechanics and Microengineering, Vol.9, 146-150, 1999

[81] B. Schauwecker, T. Mack, K. M. Strohm, W. Simon, J.-F. Luy: “RF MEMS forAdvanced Automotive Applications,” MSTNEWS Main Topic: MOEMS andRF MEMS, Vol. 04, pp. 37-38, September 2003

[82] B. Schauwecker, K. M. Strohm, W. Simon, J.-F. Luy: “RF-MEMS Componentsfor Broadband Applications,” IV. Topical Meeting on Silicon Monolithic Inte-grated Circuits in RF Systems, pp. 166169, Garmisch, April 2003

[83] E. J. J. Kruglick, K. S. J. Pister: “Lateral MEMS Microcontact Considerations,”IEEE Journal of MEMS, Vol. 8, pp. 264-271, 1999

[84] S. Timoshenko, S. Woinowskey-Kreiger: “Theory of Plates and Shells,” McGraw-Hill, New York, 1959

149

Bibliography

[85] C. Truesdell: “The Mechanical Foundations of Elasticity Theory,” Springer-Verlag, 1965

[86] B. Schauwecker, J. Mehner, K. Strohm, H. Haspeklo, J.-F. Luy: “Investigationson RF Shunt Airbridge Switches Among Different Environmental Conditions,”Sensors and Actuators A, Vol. 114, pp. 49-58, May 2004

[87] J. Mehner, F. Bennini, W. Dotzel: “CAD for Microelectromechanical Systems.System Design Automation - Fundamentals, Principles, Methods, Examples,”Kluwer Academic Publishers, pp. 111-132, 2000

[88] J. Mehner: “Entwurf in der Mikrosystemtechnik,” Dresden University Press,2000

[89] B. Schauwecker, J. Mehner, J.-F. Luy: “Modeling and Simulation Considera-tions For a New Micro-Electro-Mechanical Switch - Toggle Switch,” IV. TopicalMeeting on Silicon Monolithic Integrated Circuits in RF Systems, pp. 192-195,Garmisch, April 2003

[90] J. Mehner, S. Kurth, D. Billep, et al.: “Simulation of Gas Damping in Mi-crostructures with Nontrivial Geometries,” Proceedings of MEMS, pp. 172-177,Heidelberg, 1998

[91] S. D. Senuria: “Microsystem Design,” Kluwer Academic Publishers, 2001

[92] J. Mehner, L. Gabbay, S. D. Senturia: “Computer-Aided Generation of Nonlin-ear Reduced-Order Dynamic Macromodels: II. Stress-Stiffened Case,” Journalof Microelectromechanical Systems, 2000

[93] B. Schauwecker, K. M. Strohm, W. Simon, J. Mehner, J.-F. Luy: “Toggle-Switch– A New Type of RF MEMS Switch for Power Applications,” IMS 2002, Vol. 1,pp. 219-222, Seattle, June 2002

[94] D. Ostergaard, E. Antonova, J. Mehner, F. Vogel: “Dynamic Macromodeling ofElectrostatic Actuated MEMS Structures,” 20. International Congress on FEM-Technology, Friedrichshafen, Germany 2002

[95] J. A. King: “Materials Handbook of Hybrid Microelectronics,” Artech House,Boston, 1988

[96] B. Schauwecker, T. Mack, K. Strohm, W. Simon, J.-F. Luy: “A Very CompactRF MEMS Switch for Frequencies up to 50 GHz,” Proceedings of EuropeanMicrowave Week, Munich, Germany, October 2003

[97] T. Mack: “Si/SiGe Hetero-Feldeffekt-Transistoren fur Hochfrequenzanwendun-gen,” Diplomarbeit, Department of Semiconductor Physics, University of Ulm,November 2001

150

Bibliography

[98] B. Schauwecker, W. Simon, K. M. Strohm, T. Mack, J.-F. Luy: “New Measure-ment Results of the RF MEMS Toggle Switch,” Proceeding of the 4th RoundTable on Micro/Nano Technologies for Space, ESTEC, Noordwijk, May 2003

[99] B. Schauwecker, K. M. Strohm, T. Mack, W. Simon, J.-F. Luy: “Serial Com-bination of Ohmic and Capacitive RF MEMS Switches for Large BroadbandApplications,” Electronics Letters, Vol. 40, No. 1, pp. 44-46, 2004

[100] E. R. Brown: “RF-MEMS Switches for Reconfigurable Integrated Circuits,”IEEE Transactions on Microwave Theory and Technique, Vol. 46, No. 11, pp.1868-1880, November 1998

[101] J.H. Schaffner, R.Y. Loo, A.E. Schmitz, Tsung-Yuan Hsu, D.J. Hyman, A. Wal-ston, B.A. Warneke, G.L. Tangonan, S.W. Livingston: “RF MEMS switches forTunable Filters and Antennas,” Proceedings of 3rd International Conference onMicro Opto Electro Mechanical Systems (MOEMS 99), pp. 244, Mainz, 1999

[102] B. Pillans, J. Kleber, C. Goldsmith, M. Eberly: “RF Power Handling of Ca-pacitive RF MEMS Devices,” Digest of IEEE MTT-S, International MicrowaveSymposium, pp. 329-332, June 2002

[103] S.P. Pacheco, L.P.B. Katehi, and C.T.C. Nguyen: “Design of Low ActuationVoltage RF MEMS Switch” Digest of IEEE MTT-S, International MicrowaveSymposium, pp. 167170, Boston, June 2000

[104] N. Barker, G. Rebeiz: “Distributed MEMS True-Time Delay Phase Shiftersand Wide-Band Switches,” IEEE Transactions on Microwave Theory and Tech-niques, Vol. 46, No. 11, pp. 1881-1890, November 1998

[105] C. Goldsmith, J. Ehmke, A. Malczewski, B. Pillans, S. Eshelman, Z. Yao,J. Brank, M. Eberly: “Lifetime Characterization of Capacitive RF-MEMSSwitches,” Digest of IEEE MTT-S, International Microwave Symposium, pp.227-230, May 2001

[106] M. Ulm, T. Walter, J. Schier, R. Mueller-Fiedler, E. Kasper: “Microelectro-mechanical Capacitive RF Switches on High Resistivity Silicon Substrates,”VDE World Microtechnologies Congress (MICRO.tec), pp. 93-96, Hannover,September 2000

[107] E. Moldovan, S.-O. Tatu, T. Gaman, K. Wu, R. G. Bosisio: “A New 94-GHzSix-Port Collision-Avoidance Radar Sensor,” IEEE Transactions on MicrowaveTheory and Techniques, Vol. 52, No. 3, March 2004

[108] V. Bilik, V. Raffaj, J. Bezek: “A New Extremely Wideband Lumped Six-PortReflectometer,” Proceedings of 20th European Microwave Conference (EuMC),Vol. 2, pp. 1473, Budapest, 1990

151

Bibliography

[109] S. Ulker, R. M. Weikle: “A Millimeter-Wave Six-Port Reflectometer Based onthe Sampled-Transmission Line Architecture,” IEEE Microwave and WirelessComponents Letters, Vol. 11, No. 8, August 2001

[110] A. S. Mohra: “Six-Port Reflectometer Based on Four 0/180 Microstrip RingCouplers,” Microwave and Optical Technology Letters, Vol. 40, No. 2, September2004

[111] M. H. W. Hoffmann: “Hochfrequenztechnik,” Springer Verlag, ISBN: 3-540-61667-5, 1997

[112] D. M. Pozar: “Microwave Engineering,” 2nd edition, John Wiley&Sons Inc.,0-471-17096-8, 1998

[113] T. Wuchenauer: “Nachfuhrbare Antennen mit Micromechanischen Phasen-schiebern,” Diplomarbeit, Universitat Ulm, September 2002

[114] M. H. Perrott: “Fast and Accurate Behavioral Simulation of Fractional-N Fre-quency Synthesizers and other PLL/DLL Circuits,” Design Automation Confer-ence, June 2002

[115] M. H. Perrott: “CppSim Reference Manual,” http://www-mtl.mit.edu/ perrott

[116] W. Gander, W. Gautschi: “Adaptive Quadrature - Revisited,” BIT, Vol. 40, pp.84-101, 2000

[117] S. O. Tatu, E. Moldovan, K. Wu, R. G. Bosisio: “A Rapid Carrier RecoveryLoop for Direct Conversion Receivers,” Proceedings of IEEE Radio and WirelessConference (RAWCON), pp. 159-162, Boston, August 2003

[118] Y. Xu, R. G. Bosisio: “Analog Carrier Recovery Circuits for Six-Port DirectDigital Receivers,” Microwave and Optical Technology Letters, Vol. 31, No. 6,December 2001

[119] F. Rangel de Sousa, B. Huyart: “Carrier Recovery in Five-Port Receivers,”European Conference on Wireless Technology (ECWT), Munich, October 2003

[120] O. Mireux, J.-J. Brault, R. G. Bosisio: “Results on a Direct Digital ReceiverOperated with Fast Learning Networks,” Digest of IEEE MTT-S, InternationalMicrowave Symposium, Seattle, 2002

[121] T. Mack, T. Eireiner, T. Muller, J.-F. Luy: “A Digital mm-Wave Smart AntennaReceiver Based on Six-Port Technology for Near Range Radar Applications,”Proceedings of European Microwave Conference (EuMC), Amsterdam, October2004

[122] B. Huyart, J.-J Laurin, R. G. Bosisio, D. Roscoe: “A Direction-Finding AntennaSystem using an Integrated Six-Port Circuit,” IEEE Transactions on Antennasand Propagation, Vol. 43, No. 12, pp. 1508-1512, December 1995

152

Bibliography

[123] T.Yamaji, D. Kurose, O. Watanabe, S. Obayashi, T. Itakura: “A Four-InputBeam-Forming Downconverter for Adaptive Antennas,” IEEE Journal of Solid-State Circuits, Vol. 38, No. 10, October 2003

[124] C. G. Migelez, B. Huyart, E. Bergeault, L. P. Jallet: “A New Automobile RadarBased on the Six-Port Phase/ Frequency Discriminator,” IEEE Transactions onVehicular Technology, Vol. 49, No. 4, pp. 1416-1423, July 2000

[125] A. Stelzer, C. G. Diskus, R. Weigel, H. W. Thim: “Using a Six-Port Device inan FM-CW Radar,” Polytechnic International Press, pp. 323-326, ISBN 2-553-00935-6.

[126] A. Stelzer, C. G. Diskus, R. Weigel: “FM-CW Prazisionsradarsensor mitSix-Port Technologie,” Informationstagung Mikroelektronik, ME’01, Wien,Osterreichischer Verband fur Elektrotechnik, OVE-Schriftenreihe No. 26, pp.197-202, 2001

[127] C. G. Diskus, A. Stelzer, C. Gamsjager, K. Lubke, E. Kolmhofer, H. W. Thim:“35 GHz Six-Port Receiver For Radar Applications,” Proceedings of the SPIE -The International Society for Optical Engineering, Vol. 3752, pp 355-365, Den-ver, July 1999

[128] G. Fettweis, T. Hentschel, E. Zimmermann:“WIGWAM – A Wireless GigabitSystem with Advanced Multimedia Support,” presented a the VDE Congress,Berlin, October 2004

[129] http://www.wimax.com

[130] U. Tietze, Ch. Schenk: “Halblerter-Schaltungstechnik,” 12th edition, SpringerVerlag, ISBN 3-540-42849-6, 2002

[131] K. M. Strohm, F. J. Schmuckle, B. Schauwecker, J.-F. Luy, W. Heinrich: “SiliconMicromachined RF MEMS Resonators,” Digest of IEEE MTT-S, InternationalMicrowave Symposium, Vol. 2, pp. 1209-1212, Seattle, 2002

[132] K. M. Strohm, F. J. Schmuckle, B. Schauwecker, W. Heinrich, J.-F. Luy: “SiliconMicromachined CPW Transmission Lines,” Proceedings of European MicrowaveConference (EuMC), Vol. 2, pp. 895-898, Milan, 2002

153

Bibliography

154

Kurzfassung

Die Zielsetzung dieser Arbeit ist das Design und die Evaluierung eines universellen,rekonfigurierbaren Hochfrequenz- (HF) Empfangers fur mehrere Kommunikations-standards (Multi-Standard) bei unterschiedlichen Frequenzen (Multiband). DasFront-End basiert auf

”Software Defined Radio“ (SDR), was durch die Verlagerung

von traditionellen Hardwarestrukturen in den Digitalbereich eine signifikante Komple-xitatsreduzierung bedeutet. Zur Abwartsmischung dient das Multi-Tor-Prinzip. Diesstellt eine elegante Moglichkeit dar, Frequenzen uber einen sehr großen Frequenzbe-reich zu empfangen.

Die Original-Multi-Tor-Theorie bezieht sich ausschließlich auf alternative Netzwerk-Analysatoren. Seit den spaten neunziger Jahren wurde das Prinzip auf HF-Kommunikationsempfanger erweitert, jedoch wurde die zu Grunde liegende Theorienie ausreichend beschrieben. Deswegen soll mit dieser Arbeit auch eine exaktemathematische Beschreibung der Frequenzumsetzung und der Demodulationsprozessein Multi-Tor-Empfangern etabliert werden, die auf eine anschauliche Art erklartwerden. Mit dieser neuen detaillierten Multi-Tor-Theorie kann dann die Evaluierungdes realisierten Sechs-Tores erfolgen und mit konventionellen Empfangerarchitekturenverglichen werden.

Um gegenwartige und zukunftige Frequenzbander zwischen einem und 40 Gi-gahertz in einem einzigen Empfanger zu integrieren, muss die Hardware desMulti-Tor-Interferometers selbst rekonfigurierbar gewahlt werden. Diese Hardware-Rekonfigurierbarkeit im analogen Front-End setzt Strukturen voraus, die dasHF-Signal mit moglichst geringem Verlust umschalten konnen. Dies wird beispiels-weise im Umschalter zwischen Sende- und Empfangspfad, aber auch zur Auswahlverschiedener Sechs-Tor-Interferometer notwendig. Die Anforderung an solche Schaltersind einerseits moglichst geringe Signaldampfung, um das Signal-zu-Rausch-Verhaltnisnicht zu beeintrachtigen. Andererseits mussen die Schalter sehr hohe Sendeleistungenaushalten, wie sie in einem Umschalter zwischen Sende- und Empfangspfad auftreten.Diese beiden Bedingungen sind mit herkommlichen Schaltern auf Basis von Diodennicht ausreichend realisierbar. Die Arbeit zeigt, wie mit der neuartigen Technologieauf Basis mikro-elektromechanischer Strukturen (MEMS) genau solche Schalterumgesetzt werden konnen. Weil Schalter mit geringem Verlust und hoher Leistun-gresistenz das Schlusselelement in zukunftigen Multi-Standard- und Multiband-Empfangerarchitekturen darstellen, beschaftigt sich diese Arbeit ausfuhrlich mit demHerstellungsprozess und der Auswertung der Messwerte von HF-MEMS-Schaltern.Dies ist insbesondere ein neuartiger einpoliger Umschalter (SPDT) mit ohmschemKontakt. Derartige Schalter konnen einen Frequenzbereich von null bis 40 Gigahertzabdecken. Dabei bleibt die Einfugedampfung unterhalb von einem Dezibel, wahrend

HF-Leistungen von mehreren Watt angewendet werden konnen.

Sobald die Leistungsfahigkeit der neuartigen MEMS-Schalter bekannt und bewertetist, konnen sie im Sechs-Tor-Empfanger eingesetzt werden. Zur genauen Evaluie-rung des rekonfigurierbaren Sechs-Tor-Interferometers dienen Streuparameter, diehinsichtlich Phasenlage und Signaldampfung ausgewertet werden. Da dieser Punktsehr ausfuhrlich erfolgt, wird dadurch auch das Verstandnis uber die Funktionsweisedes Multi-Tor-Prinzipes verbessert, insbesondere hinsichtlich der großen Frequenzab-deckung des einzelnen Interferometers.

Letztlich muss das entwickelte Empfanger-Front-End auch den Anforderungenmoderner Kommunikationsstandards genugen. Zur ersten Abschatzung dienen hierRauschmessungen, bei denen die Symbolfehlerrate in Abhangigkeit vom Signal-zu-Rausch-Verhaltnis pro Bit aufgezeigt wird. Es zeigt sich anhand der durchgefuhrtenMessungen, dass der Empfanger in der Lage ist, samtliche Frequenzen zwischen einemund 40 Gigahertz mit guter Qualitat umzusetzen und Signale zu demodulieren.

Einleitung

Motivation und Stand der Technik

Die Kommunikationsindustrie steht heute der Herausforderung gegenuber, einewachsende Zahl von HF Kommunikationssystemen zu integrieren. Insbesondere imAutomobilbereich gibt es eine sehr große Zahl an unterschiedlichen Empfangern,angefangen bei Mittelwelle-Radio, uber Systeme zur elektronischen Mauterfassungbei 5,7 GHz, bis hin zu Radarsystemen bei 79 GHz. Zukunftige Kommunikations-standards mit deutlich hoheren Datenraten, die sich die hohen Bandbreiten in denFrequenzbandern bei 17 GHz, 24 GHz, und 60 GHz zu Nutze machen, werden folgen.Stand der Technik ist heute die Verwendung einer speziellen Empfangerhardware, dienur auf eine Frequenz angepasst ist. Mit derartigen Architekturen werden MultibandEmpfanger sehr komplex, ineffizient und teuer. Die Losung ist eine universelleHardware-Plattform fur samtliche Standards.

Aus diesem Grunde ist es von hochstem Interesse, eine Empfangerarchitektur zuentwickeln, die viele Frequenzbander und Standards bedienen kann. Diese Anfor-derung kann dadurch erfullt werden, dass traditionelle Hardware in den digitalenBereich verlagert wird. Diese Art von Empfanger nennt man Software DefinedRadio (SDR) [1]. Mit den Fortschritten in der CMOS Technologie, bei Analog-Digital-Wandlern (ADC), programmierbaren digitalen Signalprozessoren (DSP) undHochgeschwindigkeits-Datenleitungen ist die Umsetzung des SDR-Prinzips heute ausder Sicht der digitalen Datenverarbeitung moglich. Viel Forschungsarbeit wird aufdiesem Gebiet geleistet. Dennoch ist es unerlasslich, auch spezielle Hardware furSDR-Anwendungen zu entwickeln. Darunter fallt der gesamte Pfad von der Antennebis zum ADC. Ein effektives und zuverlassig funktionierendes Prinzip muss gefundenwerden, um Frequenzumsetzungen uber einen sehr großen Bereich durchfuhren zukonnen. Eine eigens dafur entwickelte, vereinfachte Hardware-Architektur muss dannflexibel sein und unabhangig von der Modulationsart funktionieren.

Die Motivation dieser Arbeit ist das Design eines breitbandigen Empfangers fur einenFrequenzbereich von 1 GHz bis 40 GHz mit reduziertem Hardwareaufwand. Solchebreitbandigen Kommunikationsempfanger tauchen bis heute nicht in der Literatur auf.Um dieses Ziel zu erreichen, werden zunachst alle moglichen Empfangerarchitekturenin Hinblick auf SDR-Anwendungen untersucht [2][3]. Es zeigt sich, dass das Multi-Tor-Prinzip [4] durch seine Breitbandigkeit und Einfachheit eine vielversprechende

Losung darstellt. Die in der Literatur aufgefuhrten Multi-Tore unterscheiden sichvor allem in der Anzahl der Ausgangstore. Zu finden sind Funf-Tore [5][6][7] undSechs-Tore [8][9][10][11][12][13].

Der Sechs-Tor-Empfanger, der in dieser Arbeit genauer untersucht wird, verwen-det ein Lokaloszillator (LO) und breitbandig angepasste Leistungsdetektoren zurFrequenzumsetzung. Zur Demodulation tritt das SDR-Prinzip in Erscheinung. Umjedoch den immensen Frequenzbereich abdecken zu konnen, muss die Hardware selbstMoglichkeiten bieten, Signale zwischen zwei verschiedenen Sechs-Tor-Interferometern,zwischen verschiedenen Antennen, und zwischen dem Sende- und Empfangspfadumzuschalten. Diese Besonderheit wurde mit HF-MEMS-Schaltern umgesetzt [14].Der Entwicklungsprozess dieser komplett neuartigen Schalter ist sehr komplex undwurde in Rahmen eines von der ESA beauftragten Forschungsprojekts durchgefuhrt.Die Projektbezeichnung lautet MEMOS:

”Microwave Electrostatic Micro-Machined

Devices For On-Board Applications“.

Wissenschaftlicher Beitrag der Arbeit und Ubersicht

Die Arbeit zeigt die schrittweise Entwicklung der Multi-Standard- und Multi-band Empfangerplattform, angefangen vom theoretischen Hintergrund des Multi-TorEmpfangers bis hin zur messtechnischen Evaluierung der Datenubertragung. Die Be-sonderheit liegt in der Entwicklung der dazu notwendigen verlustarmen HF-Schalter,die hohe Leistungen schalten konnen. Der Entwicklungsprozess ist ausfuhrlich in Ka-pitel 3 beschrieben. Der wissenschaftliche Mehrwert dieser Untersuchungen ist durchfolgende Punkte gegeben:

• Stabiler Herstellungsprozess neuartiger MEMS: ein ohmscher einpoliger Ein-/Ausschalter, ein einpoliger Umschalter und ein

”HF-Cross“. Diese Strukturen

konnen auf Standard Silizium-Wafern hergestellt werden (CMOS-Kompatibilitatist gewahrleistet)

• Durch ihre bessere Handhabung und die geringere Einfugedampfung sind dieMEMS-Schalter den konventionellen PIN-Dioden-Schaltern uberlegen.

• Neben S-Parameter Messungen bieten Leistungsmessung, Schaltzyklenmessung,Kontaktwiderstandsmessungen und Messung der Temperaturabhangigkeit einenEinblick in Performance und Zuverlassigkeitsuntersuchungen der Schalter.

Daneben werden im Hinblick auf Multi-Tor-Empfanger einige neue Erkenntnisse ge-wonnen. Darunter fallen:

• Eine umfassende und genaue mathematische Beschreibung der Multi-Tor-Theorie, die auf dem Prinzip der additiven Mischung aufbaut. Dies erlaubt, dieFrequenzumsetzungsprozesse in Multi-Tor-Empfangern besser zu verstehen.

• Ein neuartiges Sechs-Tor Front-End wird vorgestellt, das einen extrem großenFrequenzbereich abdeckt.

• Eine detaillierte Beschreibung der Phasenbeziehungen verbessert das Verstandnisdes Multi-Tor-Interferometers.

• Symbol-Fehlerraten-Messungen bei unterschiedlichen Signal-zu-Rausch-Verhalt-nissen werden uber den gesamten Frequenzbereich von einem bis 40 GHz durch-gefuhrt.

Der wissenschaftliche Inhalt beginnt im nachsten Kapitel”Theoretischer Hintergrund

von Multi-Tor-Empfangern” (Kapitel 2) mit einer kurzen Einfuhrung in SDR-Empfanger im Zusammenhang mit dem Multi-Tor-Prinzip. Es folgt eine Beschreibungdes verwendeten Halbleiter-Diodendetektors. Dies ist essentiell, um spater die Fre-quenzumsetzungsprozesse zu verstehen. Die Dioden sind letztlich die Abwartsmischer.Es erfolgt eine grundliche Ausarbeitung und Anwendung der Diodentheorie indieser Arbeit. Die Theorie fokussiert auf dem Multi-Tor-Prinzip im Kontext vonKommunikationsempfangern, dies weist nur noch wenige Gemeinsamkeiten mit derursprunglichen Anwendung als Netzwerkanalysator auf. Jedoch bilden die fruhenArbeiten zu Sechs-Tor-Reflektometern von Cohn [16], Engen [4][17] und Hoer [18][19]die Grundlage des heutigen Empfangers. Die heutige Empfangertheorie unterscheidetsich dadurch, dass sie ein moduliertes Signal und damit den dynamischen Misch-prozess beschreibt. Der neue Vorstoß gibt tiefe Einblicke in die Funktionsweisevon Multi-Tor-Empfangern und betrachtet die Frequenzumsetzung nicht als

”Black

Box“[20].

Um den Sechs-Tor-Empfanger uber einen großen Frequenzbereich zu betreiben,werden verlustarme HF-Schalter im analogen Front-End benotigt. In Kapitel 3,

”HF-MEMS-Schalter zur Signalleitung“, werden Moglichkeiten aufgezeigt, dieses

Problem mit Hilfe der MEMS-Technologie zu losen. Am Anfang des Kapitels stehteine Ubersicht uber den heutigen Stand dieser jungen Technologie. Das Prinzipdes einpoligen Umschalters (SPDT) wird erlautert. Dieser besteht aus einem neu-artigen ohmschen einpoligen Ein-/Ausschalter, genannt

”Toggle Schalter [21]”, in

Kombination mit einem kapazitiven einpoligen Ein-/Ausschalter, der erst bei hohenFrequenzen zur Wirkung kommt. Der Entwicklungsprozess wird von mechanischenund elektromagnetischen Simulationen begleitet. Design-Anderungen sind notwendig,um die MEMS auf ihre technologischen Fertigungsprozesse anzupassen und dieHF-Leistungsfahigkeit zu verbessern. Auf gute Streuparameter und Vertraglichkeitvon hoher HF-Leistung wurde dabei besonders geachtet. Zusatzliche Simulationenund Messungen geben Einblick in die Zuverlassigkeit der Schalter.

Die entwickelten verlustarmen Schalter werden im rekonfigurierbaren Sechs-Tor-Interferometer eingesetzt. Dies wird ausfuhrlich in Kapitel 4,

”Die Multi-Band

Sechs-Tor-Schaltung auf MEMS-Basis”, geschildert. Das Kapitel beginnt mit derEinfuhrung verschiedener HF-Interferometer und deren Designrichtlinien. Um die

MEMS im Kontext der Multi-Tor-Anwendungen zu beleuchten, wurde ein 1,5 GHzSechs-Tor-Interferometer, bestehend aus einem Leistungsteiler und drei 90 GradHybriden, konzipiert und vermessen. Da dies sehr ausfuhrlich erfolgt, soll der Leserauch an die Funktionsweise des Interferometers und die sich daraus ergebenden Pha-senbeziehungen herangefuhrt werden. Ein zweites Sechs-Tor wurde aus kommerziellenKomponenten aufgebaut, das eine bemerkenswerte Bandbreite von 2 bis 25 GHzabdeckt (der spatere Empfanger funktioniert sogar bei 40 GHz) [23]. Am Ende desKapitels wird die rekonfigurierbare Schaltung evaluiert. Dies gelingt mit Hilfe vonAgilent ADS, wobei die Messergebnisse der MEMS und der beiden Sechs-Tore in dieSimulation mit einfließen.

Im Kapitel 5,”Messergebnisse des rekonfigurierbaren Sechs-Tor-Empfangers”, wird

der Multi-Band-Empfanger in seinem vorgesehenen Frequenzbereich von einem bis40 GHz. evaluiert. Dies gelingt durch die Auswertung der Symbolfehlerraten inAbhangigkeit des Signal-zu-Rausch-Verhaltnisses von QPSK-modulierten Signalen.Schottky Dioden, die zunachst charakterisiert werden, dienen wie in der Theoriebeschrieben zur Abwartsmischung des modulierten Signals. Simulationen zur Unter-suchung von Kanalrauschen und Phasenrauschen des Lokaloszillators liegen in dentheoretischen Erwartungen. Die Simulationen werden im Zeitbereich durchgefuhrt,was eine erhohte Geschwindigkeitsanforderung an die Simulationsumgebung stellt.Ein neues auf C++ basierendes Computerprogramm (CppSim) [22], entwickeltvon Prof. Perrott am Massachusetts Institute of Technology (MIT), leistet bei denSystemsimulationen deutliche Geschwindigkeitsverbesserung. Es zeigt sich, dass dieMessungen den theoretischen und simulierten Ergebnissen entsprechen. Am Ende desKapitels wird auf weitere Anwendungsgebiete der Sechs-Tor-Technologie im BereichDistanzmessung, Radar und intelligente Antennen verwiesen.

Die Arbeit endet mit einer Zusammenfassung, der Angabe der erzielten Ergebnisse,sowie den erwarteten Aussichten der vorgestellten neuen Technologie.

Inhaltsverzeichnis

1 Einleitung 11.1 Motivation und Stand der Technik . . . . . . . . . . . . . . . . . . . . 11.2 Wissenschaftlicher Beitrag der Arbeit und Ubersicht . . . . . . . . . . . 2

2 Theorie von Multi-Tor-Empfangern 52.1 Das SDR-Konzept . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5

2.1.1 Einfuhrung in SDR . . . . . . . . . . . . . . . . . . . . . . . . . 52.1.2 SDR-Architekturen . . . . . . . . . . . . . . . . . . . . . . . . . 6

2.2 Theorie der Diodendetektoren . . . . . . . . . . . . . . . . . . . . . . . 82.2.1 Ersatzschaltbild der Halbleiterdiode . . . . . . . . . . . . . . . . 92.2.2 Diodendetektoren in Multi-Tor-Anwendungen . . . . . . . . . . 10

2.3 Einfache Beschreibung des Multi-Tor-Empfangers . . . . . . . . . . . . 122.4 Mathematische Beschreibung des Multi-Tor-Empfangers . . . . . . . . . 13

2.4.1 Theorie der Additiven Mischung . . . . . . . . . . . . . . . . . . 132.4.2 Die Multi-Tor-Theorie . . . . . . . . . . . . . . . . . . . . . . . 162.4.3 Kalibrierungsmethodik und IQ-Berechnung . . . . . . . . . . . . 17

2.5 Frequenzkonversionsprozesse in Multi-Tor-Empfangern . . . . . . . . . 18

3 Routing von HF-Signalen mittels MEMS 253.1 Motivation und Einfuhrung in HF-MEMS . . . . . . . . . . . . . . . . 25

3.1.1 Typische Anwendungen von HF-MEMS . . . . . . . . . . . . . . 273.2 Ubersicht der untersuchten HF-MEMS . . . . . . . . . . . . . . . . . . 283.3 Theoretischer Hintergrund der Simulationen . . . . . . . . . . . . . . . 30

3.3.1 Mechanische Simulationen . . . . . . . . . . . . . . . . . . . . . 303.3.2 Elektrostatische Simulationen . . . . . . . . . . . . . . . . . . . 323.3.3 Simulationen zum Einschwingverhalten . . . . . . . . . . . . . . 333.3.4 Elektromagnetische Simulationen . . . . . . . . . . . . . . . . . 34

3.4 Design-Phase mit Simulationsergebnissen . . . . . . . . . . . . . . . . . 353.4.1 Kapazitiver Bruckenschalter . . . . . . . . . . . . . . . . . . . . 363.4.2 Toggle-Schalter . . . . . . . . . . . . . . . . . . . . . . . . . . . 403.4.3 Einpoliger Umschalter (SPDT) . . . . . . . . . . . . . . . . . . 463.4.4 HF-Cross . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47

3.5 Prozessfolge zur Herstellung der MEMS . . . . . . . . . . . . . . . . . . 493.6 REM Aufnahmen und Ergebnisse aus experimentellen HF-Messungen . 52

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Inhaltsverzeichnis

3.6.1 Experimentelle Ergebnisse des kapazitiven Bruckenschalters . . 523.6.2 Experimentelle Ergebnisse des Toggle-Schalters . . . . . . . . . 543.6.3 Experimentelle Ergebnisse des SPDT-Schalters . . . . . . . . . . 563.6.4 Experimentelle Ergebnisse des HF-Cross . . . . . . . . . . . . . 59

3.7 Zusatzliche Messungen und Zuverlassigkeitsuntersuchungen . . . . . . . 603.7.1 Ergebnisse der Schaltzeitmessung . . . . . . . . . . . . . . . . . 613.7.2 Vertraglichkeit mit hoher HF-Leistung . . . . . . . . . . . . . . 633.7.3 Messung von Schaltzyklen . . . . . . . . . . . . . . . . . . . . . 653.7.4 Gleichstrom-Kontaktwiderstand . . . . . . . . . . . . . . . . . . 663.7.5 Temperaturabhangigkeit und Zuverlassigkeit . . . . . . . . . . . 67

4 Die MEMS-basierende Multiband Sechs-Tor-Schaltung 694.1 Einfuhrung in das Gebiet passiver Multi-Tor-Interferometer . . . . . . . 694.2 Mogliche Multi-Tor-Architekturen . . . . . . . . . . . . . . . . . . . . . 70

4.2.1 Das N-Tor-Interferometer . . . . . . . . . . . . . . . . . . . . . 704.2.2 Funf-Tor und Sechs-Tor-Interferometer . . . . . . . . . . . . . . 71

4.3 Design und Analyse des 1,5 GHz Sechs-Tor-Interferometers (SP1500) . 734.3.1 Theoretischer Hintergrund der elektromagnetischen Simulationen 744.3.2 Substrat und Mikrostreifenleitung . . . . . . . . . . . . . . . . . 744.3.3 Design und Simulationsergebnisse des Leistungsteilers . . . . . . 764.3.4 Design und Simulationsergebnisse des 90 Grad Hybrids . . . . . 774.3.5 Simulations- und Messergebnisse des SP1500 . . . . . . . . . . . 79

4.4 Analyse des Sechs-Tor-Interferometers von 2 bis 25 GHz (SP40) . . . . 854.4.1 Herstellung eines breitbandigen Wilkinson-Leistungsteilers . . . 864.4.2 Messergebnisse des breitbandigen Wilkinson-Leistungsteilers . . 864.4.3 Herstellung eines breitbandigen 90 Grad Hybrids . . . . . . . . 874.4.4 Messergebnisse des breitbandigen 90 Grad Hybrids . . . . . . . 874.4.5 Messergebnisse des SP40 . . . . . . . . . . . . . . . . . . . . . . 88

4.5 Das rekonfigurierbare MEMS-basierende Multiband-Front-End . . . . . 914.5.1 Mogliche Anwendungen von HF-MEMS in Empfanger-Front-Ends 914.5.2 Das MEMS-basierende rekonfigurierbare Sechs-Tor-Empfanger-

Front-End . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 934.5.3 Simulationsergebnisse des MEMS-basierenden rekonfigurierba-

ren Sechs-Tor Empfanger-Front-Ends . . . . . . . . . . . . . . . 944.5.4 Der HF-Umschalter zwischen Sende- und Empfangspfad . . . . 101

5 Messergebnisse des rekonfigurierbaren Sechs-Tor-Empfangers 1035.1 Simulationsumgebung . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103

5.1.1 Funktionsprinzip des Simulationsprogramms CppSim . . . . . . 1035.1.2 Aufbau der Simulationsplattform . . . . . . . . . . . . . . . . . 1045.1.3 Ablauf der Simulation . . . . . . . . . . . . . . . . . . . . . . . 106

5.2 Simulationsergebnisse des Sechs-Tor-Empfangers . . . . . . . . . . . . . 1095.2.1 Einflusse von Kanalrauschen . . . . . . . . . . . . . . . . . . . . 1095.2.2 Auswirkungen eines Frequenz-Offsets und Phasenrauschen . . . 112

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Inhaltsverzeichnis

5.3 Charakterisierung der Schottky-Diodendetektoren . . . . . . . . . . . . 1135.4 Messergebnisse des Sechs-Tor-Empfangers . . . . . . . . . . . . . . . . 115

5.4.1 Messaufbau und Messablauf . . . . . . . . . . . . . . . . . . . . 1155.4.2 Generelle Auswirkungen von HF- und LO-Leistung auf die

Empfangsqualitat . . . . . . . . . . . . . . . . . . . . . . . . . . 1175.4.3 Generelles Rauschverhalten . . . . . . . . . . . . . . . . . . . . 1195.4.4 Generelles Verhalten bei Phasenrauschen . . . . . . . . . . . . . 1195.4.5 Einflusse von Storern . . . . . . . . . . . . . . . . . . . . . . . . 1215.4.6 Frequenzabhangige Symbolfehlerraten-Untersuchungen des

Multiband-Empfangers . . . . . . . . . . . . . . . . . . . . . . . 1215.5 Alternative Applikationen des Multi-Tor Prinzips . . . . . . . . . . . . 132

6 Erzielte Ergebnisse und Perspektiven 1336.1 Zusammenfassung . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1336.2 Erzielte Ergebnisse und Ausblick . . . . . . . . . . . . . . . . . . . . . 135

Abbildungsverzeichnis 137

Abkurzungen:

IQ In-phase/ QuadratureHF HochfrequnzMEMS Mikro-Elektromechanische SystemeSDR Software Defined RadioSPDT Single Pole Double Throw (einpoliger Umschalter)REM RasterelektronenmikroskopDC GleichstromGHz GigahertzSER SymbolfehlerrateSP1500 Sechs-Tor-Interferometer mit Mittenfrequenz 1,5 GHzSP20 Sechs-Tor-Interferometer fur den Frequenzbereich von 2 GHz bis 25 GHzQPSK Quadratur-Phasen-Modulation (quadrature phase shift keying)

iii

Ergebnisse und Perspektiven

Mit dieser Dissertation wird eine neuartige Empfangerarchitektur vorgestellt, die eserlaubt, verschiedene Kommunikationsstandards im Frequenzbereich von 1 GHz bis40 GHz zu empfangen. Die Arbeit deckt den gesamten Entwicklungsprozess von dertheoretischen, mathematischen Beschreibung von Multi-Tor-Empfangern bis hin zuSymbolfehlerraten-Messungen an der rekonfigurierbaren Empfangerarchitektur ab.

Das dazu notwendig Hardware-Design wurde moglich, indem HF-MEMS-Schaltermit der sehr breitbandigen Multi-Tor-Technologie kombiniert wurden. Die MEMS-Technologie ist heute noch im Entwicklungsstadium. Neue geeignete Schalter furhohe Leistungen mit geringem Verlust wurden eigens dafur entwickelt, den gehobenenAnforderungen in dem rekonfigurierbaren Empfanger-Front-End zu genugen. DieEmpfangerarchitektur macht sich das SDR-Konzept zu Nutze, bei dem traditionelleHardware (z.B. zur Demodulation) in den digitalen Bereich verlagert wird. Dieneuartige Architektur ist sehr flexibel und nicht auf heutige Standards begrenzt. Umeine gewisse Servicequalitat gewahrleisten zu konnen, ist eine Abwartskompatibilitatzukunftiger Standards bei 17 GHz, 24 GHz oder 37 GHz zu den heutigen, weitverbreiteten Standards bei etwa zwei Gigahertz von entscheidender Bedeutung. DieUbertragung in Frequenzbandern oberhalb von 10 GHz bietet die Moglichkeit sehrhoher Datenraten. Eine universelle Empfangerplattform, die samtliche Standardsbedienen kann – und zwischen diesen auch sehr schnell umschaltbar ist – ist wegender damit verbundenen Komplexitats- und Kostenreduktion von hochstem Interesse.

Zusammenfassung

Nachstehend erfolgt eine Zusammenfassung der einzelnen Kapitel dieser Arbeit, sowieeine Auflistung der erzielten Ergebnisse und deren Mehrwert fur die Wissenschaft.Zum Schluss werden mogliche Anknupfungspunkte aufgefuhrt und der Ausblick dieserTechnologie diskutiert.

Kapitel 1 schildert die Notwendigkeit, eine universelle Empfangerplattform fur dengewaltigen Frequenzbereich von 1 GHz bis 40 GHz zu entwickeln. Die Motivationist eine kostengunstige Hardwarelosung, die flexibel und erweiterbar ist. Dazu sindSoftware- und Hardware-Konfigurierbarkeit ein Muss.

Kapitel 2 beginnt mit einer kurzen Einfuhrung in das SDR-Konzept im Zusam-menhang mit Multi-Tor-Empfangern. Daran anschließend erfolgt die theoretischeBeschreibung des Diodendetektors und der darin verwendeten Halbleiterdiode. Diesist Voraussetzung, um den Prozess der additiven Mischung und der Frequenz-Umsetzungsprozesse in Multi-Tor-Empfangern zu verstehen. Mit diesem theoretischenHintergrund kann dann auch die mathematische Beschreibung des an den Detekto-rausgangen anliegenden Basisbandsignals verstanden werden, wie es sich aus demmodulierten HF-Signal ableiten lasst.

In Kapitel 3 werden aktuelle Erkenntnisse im Bereich MEMS prasentiert. DieEntwicklung und die Evaluierung neuartiger Umschalter auf MEMS-Basis zurSignalleitung, die den Anforderungen in der Multi-Tor-Technologie entsprechen,werden detailliert aufgezeigt. Darunter sind neuartige Strukturen wie Toggle-Schalter,ein einpoliger Ein-/Ausschalter und ein daraus abgeleiteter einpoliger Umschal-ter, sowie ein

”HF Cross”. Der Theorieteil des Kapitels beschaftigt sich mit den

mechanischen Eigenschaften der bewegbaren Teile. Geeignete mechanische undelektromagnetische Simulationen dienen in der Designphase zur Verbesserung derspateren HF-Eigenschaften. Diese neuartigen Schalter wurden im Reinraum auf 4 ZollWafer hergestellt und anschließend messtechnisch genau charakterisiert.

Kapitel 4 knupft dann mit der Beschreibung des Designs des Multi-Tor-Interferometersan. Das designierte Sechs-Tor wurde fur die Mittenfrequenz von 1,5 GHz entwickelt. Esbesteht aus einem Leistungsteiler und drei 90-Grad-Hybriden in Mikrostreifentechnik.Der Designprozess ist ausfuhrlich beschrieben. Zusammen mit den Messergebnissendes zweiten, breitbandigen Sechs-Tores, das fur den Frequenzbereich von 2 bis 25GHz ausgelegt ist, wird dem Leser das Funktionsprinzip des rekonfigurierbarenEmpfangers klar. Der zweite Teil des Kapitels besteht in der Anwendung der MEMS-Schalter und des

”HF-Cross” zur Umschaltung zwischen den zwei Sechs-Toren. Dazu

werden geeignete Systemsimulationen durchgefuhrt, die die tatsachlich gemessenenS-Parameter verwenden. Es stellt sich heraus, dass die zusatzliche Signaldampfungund die Phasenanderungen verhaltnismaßig gering sind, was somit den reibungslosenEinsatz des Interferometers uber den gesamten Frequenzbereich ermoglicht.

Kapitel 5 schließt die Evaluierung des Multi-Standard- und Multiband-Empfangersmit geeigneten Symbolfehlerraten-Messungen ab. Zunachst werden Systemsimu-lationen des Empfangers im Zeitbereich durchgefuhrt und mit den theoretischenVorhersagen verglichen. Die Simulationen schließen weißes Gaußsches Rauschen undPhasenrauschen des Lokaloszillators mit ein. Am Ende des Kapitels werden gemesseneSymbolfehlerraten-Kurven in Abhangigkeit des Signal-zu-Rausch-Verhaltnisses proBit aufgezeigt. Um eine Aussage uber den Frequenzgang des Empfangers treffen zukonnen, wurden solche Symbolfehlerraten-Kurven bei allen relevanten Kommuni-kationsfrequenzen zwischen 1 GHz und 40 GHz aufgezeichnet. Die Messergebnissebestatigen die theoretischen Erwartungen. Des Weiteren wird auch der Grenzbereichdes Empfangers hierbei ersichtlich.

Erzielte Ergebnisse und Ausblick

Im Folgenden werden die wichtigsten Erfolge und deren wissenschaftlicher Beitrag zurVerbesserung des Kenntnisstandes auf dem Gebiet von HF-MEMS und breitbandigenSDR-Empfanger auf Multi-Tor Basis aufgelistet.

HF-MEMS

• Ein stabiler Herstellungsprozess fur neuartige HF-MEMS konnte aufgebaut wer-den. Mit diesem Herstellungsprozess lassen sich vor allem die kritischen, beweg-baren und freischwebenden Teile des kapazitiven Schalters, des Toggle-Schalters,des einpoligen Umschalters und des

”HF-Cross” zuverlassig wiederholen. Zudem

wurde auf CMOS-Kompatibilitat Wert gelegt.

• HF-Verhalten des neuen kapazitiven einpoligen Ein-/Ausschalters: bis 40 GHzerfolgt eine Signalabschwachung von 35 dB bei einer Einfugedampfung von 0,4dB.

• HF-Verhalten des neuartigen ohmschen einpoligen Ein-/Ausschalters (Toggle-Schalter): bis 30 GHz ist die Einfugedampfung unterhalb von 0,5 dB, die An-passung uber 25 dB und die Isolation uber 14 dB. Dabei halt der Schalter eineHF-Leistung von 1 W bei 30 GHz, 2 W bei 18 GHz, und 2,5 W bei 15 GHz aus.

• HF-Verhalten des neuartigen zweipoligen Umschalters: bis 30 GHz liegt dieEinfugedampfung unter 0,5 dB, die Anpassung uber 22 dB und die Isolationuber 22 dB.

• HF-Verhalten des neuartigen”HF-Cross”: bis 30 GHz liegt die Einfugedampfung

unter 1 dB (obere Leitung) bzw. 1,5 dB (untere Leitung).

Die gemessenen HF-Parameter sind denen von konventionellen Diodenschalternuberlegen. Sie haben sehr geringe Schaltspannungen und eine vernachlassigbareStromaufnahme. Die Einfugedampfung typischer GaAs PIN-Diodenschalter liegtdeutlich uber 1,5 dB bei 20 GHz. Zudem halten diese oftmals nur HF-Leistungenunterhalb von 200 mW stand und sind nicht CMOS kompatibel.

Eine detaillierte Analyse des Toggle- und des kapazitiven Schalters wurde durch-gefuhrt. Die Schalter zeigen erwartete Temperaturabhangigkeit, der Gleichstrom-Kontaktwiderstand stimmt mit den Berechnungen aus HF-Messungen uberein. Mit50.000 erfolgreich durchgefuhrten Schaltzyklen halt der Toggle-Schalter der erstenBewahrungsprobe fur den Alltagseinsatz stand. Verstarkungen der Struktur sindhier jedoch notwendig, was ein Ansatz fur weitere Arbeiten auf diesem Gebiet darstellt.

SDR-Multi-Tor-Empfanger

• Eine umfassende und einfach zu verstehende mathematische Beschreibung desMulti-Tor-Empfangers ist aufgefuhrt. Der Ansatz wendet die Theorie des additi-ven Mischens auf das Multi-Tor-Prinzip an und erlaubt Einsicht in die stattfin-denden Frequenzumsetzungsprozesse. Daraus wird dann auch klar, warum derMulti-Tor-Empfanger nicht - wie herkommliche IQ Empfanger - mit nur zweiAusgangen auskommt.

• Eine neue Hardwarebasis fur SDR-Empfanger von 1 GHz bis 40 GHz wurdeentwickelt und die Funktion messtechnisch nachgewiesen.

• Durch die detaillierten Messungen wurde auch das Verstandnis der Uberlage-rungsmechanismen im Multi-Tor-Interferometer verbessert.

• Symbolfehlerraten-Messungen in Abhangigkeit des Signal-zu-Rausch-Verhaltnis-ses pro Bit uber den Frequenzbereich von 1 GHz bis 40 GHz verifizieren die Funk-tion als Kommunikationsempfanger. Symbolfehlerraten-Messungen, die einensolch weiten Frequenzbereich abdecken, wurden bisher noch an keinem anderenKommunikationsempfanger aufgezeigt.

Die fur diese Dissertation durchgefuhrten Arbeiten konnen in verschiedenen Bereichenerweitert werden. So konnte zwar ein stabiler Herstellungsprozess fur die MEMSerzielt werden, jedoch ist der Ertrag pro Wafer noch gering. Fur eine Vermarktungdieser hervorragenden HF-Schalter ist jedoch auch ein wirtschaftlicher Herstellungs-prozess notwendig. Die Ausfuhrungen zum rekonfigurierbaren HF-Interferometersollen als Tauglichkeitsnachweis aufgefasst werden. Untersuchungen mit integriertenStrukturen, moglichst ein Interferometer auf Si Basis, mussen folgen. Eine genaueRauschuntersuchung, sowie Empfindlichkeits- und Dynamikbewertung der Dioden-detektoren steht aus. Dazu ist es unerlasslich, den kompletten HF-Pfad aufzubauen.Dies beinhaltet den LNA, einen Bandpassfilter, sowie Basisband-Verstarker.

Diese Dissertation erlaubt neue Einsichten in Multi-Tor-Empfanger. Der vorgestellteMulti-Standard- und Multiband-Empfanger ist eine interessante und vielversprechendeMethode zur Frequenzumsetzung bei der Kommunikation im Millimeterwellenbereich.Die Grenzfrequenz der Dioden liegt dabei bei mehreren hundert Gigahertz. Damitbieten Dioden heute die einzige Technologie, um Frequenzen im oberen Millimeter-wellenbereich umzusetzen. Auf der anderen Seite jedoch haben gerade die Diodenauch Nachteile in Multi-Tor-Empfangern. Dioden haben im Vergleich zu heuteerhaltlichen Transistoren-Mischern einen deutlich hoheren Konversionsverlust. Dieslimitiert die Detektion von sehr schwachen Signalen deutlich (sie haben eine geringeEmpfindlichkeit). Die Limitierung ergibt sich vor allem dadurch, dass die Dioden einehohe Vorverstarkung des Signals benotigen. Der dazu verwendete LNA kann dannjedoch sehr leicht in seinen nichtlinearen Bereich getrieben werden, weil Kanalstorerzu erwarten sind. Bandpass-Filter mit sehr hohen Gute- und geringen Dampfungs-werten sind notwendig, um diese Storer noch vor dem LNA abzutrennen. Solche

Filter gibt es heute nicht. Die MEMS-Technologie konnte aber auch hier wiederumhelfen. Erste Ansatze sind in dieser Arbeit erwahnt. Dabei zeigt sich der Engpassvon Kommunikationselektronik auch beim Multi-Tor: Die Leistungsfahigkeit heutigerEmpfanger hangt maßgeblich von der Gute der Bandpassfilter, der Linearitat derVorverstarker und nicht zuletzt vom Phasenrauschen des Lokaloszillators ab. Jedochsind heute noch keine großen Storeinflusse im Millimeterwellenbereich zu erwarten,weil es noch keine Massenmarkt-Anwendungen gibt. Deswegen ist es vielversprechend,Multi-Tor-Empfanger fur diesen Frequenzbereich auszulegen.

Lebenslauf

Torsten Mackgeboren am 13. Dezember 1972

in Ulm-Soflingen

06.92 Abitur am Hans und Sophie Scholl Gymnasium Ulm

08.92-11.93 Zivildienst

Studium Physik

01.94-07.94 Central Connecticut State University, New Britain, CT, USA

10.94-07.96 Universitat Ulm (Vordiplom Physik)

09.96-07.97 University of Leeds, England (Bachelor of Science)

09.97-02.02 Universitat Ulm (Physik Diplom)

Promotion Elektrotechnik

10.01-10.04 Doktorand bei der DaimlerChrysler Forschung in Ulm,Abteilung Mikrowellentechnik

11.03-05.04 Gaststudent am Massachusetts Institute of Technology (MIT),Research Laboratory of Electronics,Cambridge, USA

15.06.05 Antrag auf Zulassung zur Promotion amLehrstuhl fur technische Elektronikder Universitat Erlangen-Nurnberg

11.08.05 Tag der Promotion

seit 11.04 Entwicklungsingenieur bei der DaimlerChrysler AG

dissertation torsten mack

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