cmos device model objective –hand calculations for analog design –efficiently and accurately...
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CMOS Device Model
• Objective– Hand calculations for analog design– Efficiently and accurately simulation
• CMOS transistor models– Large signal model– Small signal model– Simulation model– Noise model
Large Signal Model• Nonlinear equations for solving dc values of
device currents given voltages• Level 1: Shichman-Hodges (VT, K', , and
NSUB)• Level 2: with second-order effects (varying
channel charge, short-channel, weak inversion, varying surface mobility, etc.)
• Level 3: Semi-empirical short-channel model• Level 4: BSIM models. Based on automatically
generated parameters from a process characterization. Good weak-strong inversion transition.
Transconductance when VDS is small
Transconductance when VDS is small
Transconductance when VDS is small
Effect of changing VDS for a large VGS
Effect of changing VDS for a given VGS
Effect of changing VDS for a given VGS
Effect of changing VDS for various VGS
VGS<=VT
Effect of changing VDS for various VGS
Effect of changing VDS for various VGS
MOST Regions of Operation
• Cut-off, or non-conducting: VGS <VT
– ID=0
• Conducting: VGS >=VT
– Saturation: VDS > VGS – VT
– Triode or linear or ohmic or non-saturation: VDS <= VGS – VT
)-)V - V((vL
WμC i DSV
DSTGSox
D 2
2
2
2) - V(v
L
WμC i TGS
oxD
With channel length modulation
)λV() - V(vL
WμC i DSTGS
oxD 1
22
) 22(0 | |φ - | |v| |φ V V fBSfTT
L
WK
L
WμC ox '
Capacitors Of The Mosfet
CBD and CBS include both the diffusion-bulk junction capacitance as well as the side wall junction capacitance. They are highly nonlinear in bias voltages.
C4 is the capacitance between the channel and the bulk. It is highly nonlinear and depends on the operation of the device. C4 is not measurable from terminals.
/2
Gate related capacitances
Small signal model
Typically: VDB, VSB are in such a way that there is a reversely biased pn junction.
Therefore: gbd ≈ gbs ≈ 0
In saturation:
But
In non-saturation region
High Frequency Figures of Merit T
• AC current source input to G• AC short S, D, B to gnd• Measure AC drain current output• Calculate current gain• Find frequency at which current gain = 1.
• Ignore rs and rd, Cbs, Cbd, gds, gbs, gbd all have zero voltage drop and hence zero current
• Vgs = Iin /jw(Cgs+Cgb+Cgd) ≈ Iin /jwCgs
• Io = − (gm − jw Cgd)Vgs ≈ − gmVgs
• |Io/Iin| ≈ gm/wCgs
• At T, current gain =1
• T ≈ gm/(Cgs+Cgd)≈ gm/Cgs
• or
• AC current source input to G• AC short S, D, B to gnd• Measure AC power into the gate• Assume complex conjugate load• Compute max power delivered by the transistor• Find maximum power gain• Find frequency at which power gain = 1.
High Frequency Figures of Merit max
• max: frequency at which power gain becomes 1
PL=
BSIM models• Non-uniform charge density• Band bending due to non-uniform gate voltage• Non-uniform threshold voltage
– Non-uniform channel doping, x, y, z– Short channel effects
• Charge sharing• Drain-induced barrier lowering (DIBL)
– Narrow channel effects– Temperature dependence
• Mobility change due to temp, field (x, y)• Source drain, gate, bulk resistances
“Short Channel” Effects
• VTH decreases for small L
– Large offset for diff pairs with small L
• Mobility reduction:– Velocity saturation
– Vertical field (small tox=6.5nm)
– Reduced gm: increases slower than root-ID
Threshold Voltage VTH
• Strong function of L– Use long channel for VTH matching
– But this increases cap and decreases speed
• Process variations– Run-to-run– How to characterize?– Slow/nominal/fast– Both worst-case & optimistic
Effect of Velocity Saturation
• Velocity ≈ mobility * field
• Field reaches maximum Emax
– (Vgs-Vt)/L reaches ESAT
• gm become saturated:– gm ≈ ½nCoxW*ESAT
• But Cgs still 2/3 WL Cox
• T ≈ gm/Cgs = ¾ nESAT /L
• No longer ~ 1/L^2
Threshold Reduction• When channel is short, effect of Vd extends to S• Cause barrier to drop, i.e. Vth to drop• Greatly affects sub-threshold current: 26 mV Vth
drop current * e• 100~200 mV Vth drop due to Vd not uncommon
100s or 1000 times current increase
• Use lower density active near gate but higher density for contacts
Other effects• Temperature variation• Normal-Field Mobility Degradation• Substrate current
– Very nonlinear in Vd
• Drain to source leakage current at Vgs=0– Big concern for static power
• Gate leakage currents– Hot electron– Tunneling – Very nonlineary
• Transit Time Effects
Consequences for Design • SPICE (HSPICE or Spectre)
– BSIM3, BSIM4 models– Accurate but inappropriate for hand analysis– Verification (& optimization)
• Design:– Small signal parameter design space:
• gm, CL (speed, noise)
• gm/ID, ID (power, output range, speed)
• Av0= gmro (gain)
– Device geometries from SPICE (table, graph);
– may require iteration (e.g. CGS)
Sweep V1Measure vgs
Intrinsic voltage gain of MOSFET
Intrinsic voltage gain = gm/go = vds/vgs for constant Id
Electronic Noise• Noise phenomena• Device noise models• Representation of noise (2-ports):
– Motivation– Output spectral density– Input equivalent spectral density– Noise figure– Sampling noise (“kT/C noise”)
• SNR versus Bits• Noise versus Power Dissipation
– Dynamic range– Minimum detectable signal
Noise in Devices and Circuits
•Noise is any unwanted excitation of a circuit, any input that is not an information-bearing signal.• External noise: Unintended coupling with other parts of the physical world; in principle, can be virtually eliminated by careful design.• Intrinsic noise: Unpredictable microscopic events inherent in the device/circuit; can be reduced, but never eliminated. •Noise is especially important to consider when designing low-power systems because the signal levels (typically voltages or currents) are small.
Noise vs random process variations
• random process variations– Variations from one device to another– For any device, it is fixed after fabrication
• Noise– Unpredictable variations during operation– Unknown after fabrication– Remains unknown after measurement during
operation– May change with environment
Time domain description of noise
What is signal and what is noise?
)()()( tntstx
srmsT
s PSrmsSdttsT
P )(,)(1
02
nrmsT
n PNrmsNdttnT
P )(,)(1
02
Signal and noise power:
Physical interpretation
power
If we apply a signal (or noise) as a voltage source across a one Ohm resistor, the power delivered by the source is equal to the signal power.
Signal power can be viewer as a measure of normalized power.
Signal to noise ratio
)(log20)(log10 1010rms
rms
n
s
N
S
P
PSNR
SNR = 0 dB when signal power = noise power
Absolute noise level in dB:w.r.t. 1 mW of signal power
)log(10dB30
mW1log10din
n
nmn
P
PP
B
SNR in bits• A sine wave with magnitude 1 has power
= 1/2.• Quantize it into N=2n equal levels between
-1 and 1 (with step size = 2/2n)• Quantization error uniformly distributed
between +–1/2n
• Noise (quantization error) power=1/3 (1/2n)2
• Signal to noise ratio = 1/2 ÷ 1/3 (1/2n)2 =1.5(1/2n)2 = 1.76 + 6.02n dB or n bits
-1<=C<=+1
C=0: n1 and n2 uncorrelatedC=1: perfectly correlated
Adding uncorrelated noises
Adding correlated noises
For independent noises
Frequency domain description of noise
T
TTn dttntn
TR )()(
2
1lim)(
))(()()( nnn RfSfPSD F
dffPSDP nn )(
dffPSDP nn )(0
Given n(t) stationary, its autocorrelation is:
The power spectral density of n(t) is:
For real signals, PSD is even. can use single sided spectrum: 2x positive side
↑ single sided PSD
)()( fXtx
dffXdttx
22)()(
)()( fPSDR xx
dffPSDRdttx xx
T
TT
)()0()(lim
2
Parseval’s Theorem:
If
If x(t) stationary,
Interpretation of PSD
PSDx(f)
Pxf1 = PSDx(f1)
Types of “Noise” • “man made”
– Interference– Supply noise– …– Use shielding, careful layout, isolation, …
• “intrinsic” noise– Associated with current conduction– “fundamental” –thermal noise– “manufacturing process related” – flicker noise
Thermal Noise • Due to thermal excitation of charge carriers in a
conductor. It has a white spectral density and is proportional to absolute temperature, not dependent on bias current.
• Random fluctuations of v(t) or i(t)• Independent of current flow• Characterization:
– Zero mean, Gaussian pdf– Power spectral density constant or “white” up to about
80THz
Thermal noise dominant in resisters
Example:R = 1kΩ, B = 1MHz, 4µV rms or 4nA rms
HW
Equivalently, we can model a real resistor with an ideal resistor in parallel with a current noise source. What rms value should the current source have?
Show that when two resistors are connected in series, we can model them as ideal series resistors in series with a single noise voltage source. What’s the rms value of the voltage source?
Show that two parallel resistors can be modeled as two ideal parallel resistors in parallel with a single noise current source. What’s the rms value of the current source?
Noise in Diodes Shot noise dominant– DC current is not continuous and smooth but
instead is a result of pulses of current caused by the individual flow of carriers.
It depends on bias, can be modeled as awhite noise source and typically larger than
thermal noise. − Zero mean – Gaussian pdf – Power spectral density flat – Proportional to current – Dependent on temperature
Example:ID= 1mA, B = 1MHz, 17nA rms
MOS Noise Model
Flicker noise
–Kf,NMOS 6 times larger than Kf,PMOS
–Strongly process dependent
−when referred to as drain current noise, it is inversely proportional to L2
BJT Noise
Sampling Noise • Commonly called “kT/C” noise
• Applications: ADC, SC circuits, …
von
R
C
Used:
Filtering of noise
H(s)x(t) y(t)
|H(f )|2 = H(s)|s=j2f H(s)|s=-j2f
Noise Calculations 1) Get small-signal model2) Set all inputs = 0 (linear superposition)
3) Pick output vo or io4) For each noise source vx, or ix Calculate Hx(s) = vo(s) / vx(s) (or … io, ix)5) Total noise at output is
6) Input Referred Noise: Fictitious noise source at input: 22
,2
, )(/ sAvv Toneffin
Example: CS Amplifier
CL
VDD
RL
Von=(inRL +inMOS)/goT
goT = 1/RL + sCL
LBnR RTki
L
142
mBnMOS gTki3
242
o=1/RLCL
Some integrals
HW
In the previous example, if the transistor is in triode, how would the solution change?
HW
If we include the flicker noise source, how would that affect the computation? What do you suggest we should modify?
HW
In the example, if RL is replaced by a PMOS transistor in saturation, how would the solution change? Assume appropriate bias levels.