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EVS28 International Electric Vehicle Symposium and Exhibition 1
EVS28
KINTEX, Korea, May 3-6, 2015
Boost Converter Selection and analysis for Automotive
Applications
Moataz Elsied1, 2, Amrane Oukaour1, Hamid Gualous1, Youssef salmanie1, Amr Amin2,
Radwan Hassan2 1 Lusac laboratoire –, BP78 Rue Louis Aragon (Cherbourg-France)
Université of Caen Basse- Normandie, Caen, France
Phone:0033 986 812685, e-mail: amrane.oukaour@unicaen.fr
2Electrical Power and Machines Department
Helwan University, Faculty of Engineering
Helwan, Cairo, Egypt
Phone: 0033 689 326 447, e-mail: moataz.elsied@unicaen.fr
Abstract
During the past decade, power electronics research has focused on the development of multi-phase dc/dc
power converters for electric vehicles (EVs) and DC micro-grids applications, and it is estimated that there
is still a huge potential of this field during the coming years. In this paper, Four-Phase Interleaved Boost
Converter (FP-IBC) that interfaces different power sources with the powertrain of hybrid electric vehicles
(HEVs) is introduced, discussed and analyzed. The proposed converter is compared with different topologies
such as conventional boost converter (BC), Multi-device boost converter (MDBC), and Two-phase
interleaved boost converter (TP-IBC) and Multi-device interleaved boost converter (MD-IBC). The
comparison is performed at different switching frequencies and power ratings to show the effect of these
variables on the converters losses. The comparison indicated that the FP-IBC is able to the size of passive
components with high efficiency compared with the other topologies of boost converters. The simulation
results and analysis proved that FP-IBC is more powerful than other dc/dc converter topologies in achieving
high performance and reliability for high-power rating dc/dc converters.
Keywords: electric vehicles (EVs), DC micro-grids, Four-Phase Interleaved Boost Converter (FP-IBC), small
signal model (SSM).
1 Introduction The growing number of human population yields
increased energy consumption and depletion of
finite resources such as, oil and gas. Therefore,
Electrical Vehicles (EVs) are seriously considered
as an alternative to fossil-fueled vehicles [1]-[2].
In EVs, fuel cells (FCs), super-capacitors (SCs),
and batteries are usually used as energy storage
devices. Combining such energy sources leads to a
FC/SC/battery hybrid power system (HPS) [3]-[4],
as it is shown in Figure. 1. Unlike single-sourced
systems, HPS has the potential to provide the load
with high quality, more reliable, and efficient
power. In these systems, FCs are emerging as a
promising supplementary power sources due to
their merits of cleanness, high efficiency, and high
reliability. Because of FC’s drawbacks such as long
startup time and slow dynamics, mismatch power
EVS28 International Electric Vehicle Symposium and Exhibition 2
between the load and FC should be managed by
an Energy Storage System (ESS) such as batteries
and SCs [5]-[6].
On the other hand, batteries are usually used as
storage devices for smoothing output power,
enhancing the peak power capacity, and
improving the system dynamic characteristics and
startup transitions. But, batteries suffer from some
serious drawbacks such as low life cycle, low
power density and long recharging time. To
overcome these disadvantages, SCs are used in
transition periods for their high power density to
deliver the required energy. However, their energy
density is low which limits their contribution
during extended transient states. Therefore, a
combination of batteries and SCs seems to be a
better choice as ESS working with FC as a
supplementary source, which not only provides a
higher power density but also increases the energy
storage capability of EVs [7].
In EVs, a dc/dc boost converter is a key element
to interface HPSs to the EV’s dc-bus. Various
dc/dc boost converters topologies have been
studied and analyzed for EV applications in [8]-
[11]. It is illustrated in [12] that Multi-Device
Interleaved Boost Converter (MD-IBC) is more
powerful and more efficient than other boost
converter topologies such as conventional Boost
Converter (BC), Two-Phase Interleaved Boost
Converter (TP-IBC), and Multi-Device Boost
Converter (MDBC).
In this paper, the proposed FP-IBC outperforms
MD-IBC and other topologies of boost converters
with its higher reliability, higher efficiency,
higher power and voltage ratings. The FP-IBC and
the other converters are modeled and simulated
using Matlab/Simulink. Simulation results are
analyzed and discussed. Moreover, the
converters’ power losses are evaluated
numerically to benchmark their efficient
performance. Simulation results along with
numeric comparison for efficiency have
demonstrated that the proposed converter is very
promising for EVs applications.
Figure 1: Block diagram for the FC/SC/battery hybrid
power system
2 Structure of the FP-IBC The structure of the FP-IBC is depicted in Figure.
2a. The proposed converter consists of four dc/dc
boost converter modules connected in parallel.
Figure. 2b shows the switching device gate signals
at D=0.5 where D is the duty cycle. The gate signals
are successively phase shifted by Ts /(n*m), where
Ts is the switching period, n is the number of phases,
and m is the number of parallel switches per phase.
For FP-IBC, m=1 and n=4. As such, the current
delivered by the electric source is shared equally
between each phase and has a ripple content of
period Ts/4. Similarly, the frequency of the output
voltage and the input current is n times higher than
the switching frequency fsw. As result, the size of
passive devices such as capacitor and inductor will
be reduced by n times compared to the conventional
BC. In addition, the system reliability and converter
power rating will be increased by using paralleling
phases. Moreover, the current sharing equally
between each phase will provide tight sizing of
power semiconductors, distribution of losses
between modules and size’s optimization of the
converter. These advantages are behind the use of
FP-IBC as a dc/dc converter for EV power systems
in particular for high power applications as opposed
to other converter topologies. The structures of
conventional BC, MDBC, TP-IBC, and MD-IBC
are described and discussed in details in [12].
(a)
(b)
Figure 2: (a) FP-IBC structure (b) The switching pattern
of FP-IBC
EVS28 International Electric Vehicle Symposium and Exhibition 3
3 FP-IBC Modeling The boost converter’s double pole and right half
plane (RHP) zero whose location in the s-plane is
dependent on the input voltage, output voltage,
load resistance, inductance, and output
capacitance. Therefore, appropriate selection of
the transfer function parameters and control law
are important for the system stability criteria to
achieve proper operation.
In this section, a generalized small signal model
(SSM) which is derived in [13]-[15] is applied to
the proposed converter to develop a transfer
function relating the duty cycle with the inductor
current and the output voltage. This model can
also be applied for all topologies of boost
converters which are discussed in this paper. The
transfer function between inductor current and
duty cycle (1) is a second order, two poles in the
left-half plane (LHP) and a LHP zero. The LHP
zero is written in (2).
𝐻𝑖(𝑠) =𝐼𝐿(𝑠)
𝑑(𝑠)=
𝑉𝑜 (𝑚+𝜎)
𝜎 𝑅𝑙+𝑛 𝑅𝑜(1−𝑚𝐷)2
(1+ 𝑠 𝜔𝑧𝑖⁄ )
(𝑠2
𝜔02+
2𝜉𝑠
𝜔0+1)
(1)
𝜔𝑧𝑖 = 1
𝐶( 𝑅𝑐+𝜎 𝑅𝑜
𝑚+𝜎) (2)
𝐻𝑣(𝑠) =𝑣𝑜(𝑠)
𝑑(𝑠)=
𝑉𝑜 [𝑛 𝑅𝑜(1−𝑚𝐷)2−𝑚 𝑅𝑙]
(1−𝐷) [𝑛 𝑅𝑜(1−𝑚𝐷)2+𝜎 𝑅𝑙] (1+ 𝑠 𝜔𝑧𝑣1⁄ )(1− 𝑠 𝜔𝑧𝑣2⁄ )
(𝑠2
𝜔02+
2𝜉𝑠
𝜔0+1)
(3)
𝜔𝑧𝑣1 = 1
𝐶 𝑅𝑐 (4)
𝜔𝑧𝑣2 = 𝑛 𝑅𝑜(1−𝑚𝐷)2−𝑚 𝑅𝑙
𝑚 𝐿 (5)
Formulation (3) consists of double pole, RHP zero
and LHP zero and this equation describes the
relation between output voltage and duty cycle.
The zero in the LHP is introduced in (4), and the
zero in RHP is given in (5).
𝜔0 = √𝜎 𝑅𝑙+ 𝑛 𝑅𝑜(1−𝑚𝐷)2
𝜎 𝐿 𝐶 (𝑅𝑜+𝑅𝑐) (6)
The double pole frequency ω0 presented in (6)
depends on the input voltage (𝑣𝑖𝑛) and the output
voltage (𝑣𝑜) as well as inductance (𝐿) and output
capacitance (𝐶). It is also important to note that
ω0 depends on the load resistance ( 𝑅𝑜 ), the
internal resistance of the inductor (𝑅𝑙 ) and the
internal resistance of the capacitor ( 𝑅𝑐 ). The
system damped ratio for both transfer functions is
given by,
𝜉 = 𝜎 𝐿+𝐶 [𝜎 𝑅𝑙 (𝑅𝑜+𝑅𝑐)+𝑛 𝑅𝑐 𝑅𝑜(1−𝑚𝐷)2]
2 √𝜎 𝐿 𝐶 (𝑅𝑜+𝑅𝑐)[𝜎 𝑅𝑙+ 𝑛 𝑅𝑜(1−𝑚𝐷)2] (7)
Where 𝐷 is the nominal duty ratio and its
expression is described by,
𝐷 = 𝑣𝑜−𝑣𝑖𝑛
𝑚 𝑣𝑜 𝜎 =
(1−𝑚 𝐷)
(1−𝐷) (8)
4 Closed Loop Control Design The boost converter feedback control is a nonlinear
function of the duty cycle, which makes the
controller design is more challenging from the
viewpoint of stability and bandwidth [16]-[17]. The
control design of different topologies of boost
converter consists of two control loops, inner and
outer loop, the inner loop is used for current control,
which is much faster than the voltage outer control
loop as it is shown in Figure .3. Where Cv(s) is the
voltage compensator and Ci(s) is current
compensator that assures cancellation of the static
error and high bandwidth. The resultant control
signal is (m1), while the duty cycle signal is (d).
The transfer function of the current and voltage
controllers are introduced in [15] and it is used in
this work to regulate the converter’s output voltage.
Compensators used in the inner and outer loops
introduce two poles and a zero. A pole at the origin
is considered as an integral action and provides a
very high gain at low frequencies. Moreover, the
pole-zero pairs (𝑝𝑖 ,𝑧𝑖) for current controller and
(𝑝𝑣 ,𝑧𝑣) for voltage controller aim to reduce the
phase shift between the frequency of the two plant
zeros and the frequency of two plant poles.
The transfer function (9) and (10) are used to design
the current and voltage controller and can be written
as the following:
𝐶𝑖(𝑠) = 𝑘𝑐𝑖 (𝑠+ 𝑧𝑖)
𝑠(𝑠+ 𝑝𝑖) (9)
𝐶𝑣(𝑠) = 𝑘𝑐𝑣 (𝑠+ 𝑧𝑣)
𝑠(𝑠+ 𝑝𝑣) (10)
𝑧𝑖 = 𝑤𝑐𝑛𝑖 √1−sin ∅𝑚𝑑𝑖
1+sin ∅𝑚𝑑𝑖 , 𝑝𝑖 =
𝑧𝑖
√1−sin ∅𝑚𝑑𝑖1+sin ∅𝑚𝑑𝑖
(11)
𝑧𝑣 = 𝑤𝑐𝑛𝑣 √1−sin ∅𝑚𝑑𝑣
1+sin ∅𝑚𝑑𝑣 , 𝑝𝑣 =
𝑧𝑣
√1−sin ∅𝑚𝑑𝑣1+sin ∅𝑚𝑑𝑣
(12)
Where 𝑤𝑐𝑛𝑖 and 𝑤𝑐𝑛𝑣 are the new crossover
frequency for the current loop and the voltage outer
loop, respectively. The compensator phase margin
of the current and the voltage controller are
abbreviated as ∅𝑚𝑑𝑖 and ∅𝑚𝑑𝑣, respectively. 𝑘𝑐𝑖 is
the current controller gain where 𝑘𝑣𝑖 is the voltage
EVS28 International Electric Vehicle Symposium and Exhibition 4
controller gain which can be calculated as the
following:
𝑘𝑐𝑖 = |1
Ti(S)| at 𝑤𝑐 = 𝑤𝑐𝑛𝑖 (13)
𝑘𝑣𝑖 = |1
Tv(S)| at 𝑤𝑐 = 𝑤𝑐𝑛𝑣 (14)
Where 𝑇𝑖(𝑠), 𝑇𝑣(𝑠) are the open loop transfer
functions for the inner and outer loop respectively.
𝑇𝑖(𝑠) = 𝐶𝑖(𝑠)𝐻𝑖(𝑠) for the inner loop
(15)
Tv(s) =Cv(s)Ci(s)Hv(s)
1+Ci(s)Hi(s) for the outer loop
(16)
Figure 3: Control design loop configuration
5 Power-Losses calculations
model This section aims to provide a mathematical tool
for the calculation of total power losses in boost
converter. The total power losses (𝑝𝑙𝑜𝑠𝑠_𝑡𝑜𝑡) in any
boost converter can be considered as the sum of
the inductor copper losses (𝑝𝑙), the inductor core
losses ( 𝑝𝑐𝑜𝑟𝑒 ), the capacitor losses ( 𝑝𝑐) , the
switching (𝑝𝑠) and the conduction (𝑝𝑐𝑜𝑛) losses of
the switching devices [18].
In this study, the Metal Oxide Semiconductor
Field Effect Transistor (MOSFET) and diode
represent the switching devices. The switching
losses of MOSFET ( 𝑝𝑠𝑀 ) and diode ( 𝑝𝑠𝐷 )
calculations are based on the MOSFET data sheet
parameters and can be obtained from the
following equations:
𝑝𝑠 = 𝑝𝑠_𝑀+𝑝𝑠_𝐷 (17)
𝑝𝑠_𝑀 = (𝐸𝑜𝑛_𝑀 + 𝐸𝑜𝑓𝑓_𝑀) 𝑓𝑠𝑤 (18)
𝑝𝑠_𝐷 = (𝐸𝑜𝑛_𝐷 + 𝐸𝑜𝑓𝑓_𝐷) 𝑓𝑠𝑤 ≈ 𝐸𝑜𝑛_𝐷 𝑓𝑠𝑤
(19)
𝐸𝑜𝑛_𝑀 = 𝑈𝑑𝑑 𝐼𝐷_𝑜𝑛 𝑡𝑟𝑖+𝑡𝑓𝑢
2 + 𝑄𝑟𝑟 (20)
𝐸𝑜𝑓𝑓_𝑀 = 𝑈𝑑𝑑 𝐼𝐷_𝑜𝑓𝑓 𝑡𝑟𝑢+𝑡𝑓𝑖
2 (21)
𝐸𝑜𝑛_𝐷 = 1
4𝑄𝑟𝑟 𝑈𝑑𝑟𝑟 (22)
Where 𝐸𝑜𝑛_𝑀 , 𝐸𝑜𝑛_𝐷 are the turn-on energy losses
of the power MOSFET and diode respectively,
𝐸𝑜𝑓𝑓_𝑀 , 𝐸𝑜𝑓𝑓_𝐷 are the turn-off energy loss, 𝑓𝑠𝑤 is
the switching frequency, 𝑈𝑑𝑑 is the open-circuit
voltage on the switching device, 𝑈𝑑𝑟𝑟 is the voltage
across the diode during reverse recovery, 𝑄𝑟𝑟 is the
reverse recovery charge, 𝑡𝑟𝑖 is switch rated rising
time, and 𝑡𝑓𝑖 is the rated falling time. 𝐼𝐷_𝑜𝑛 ,𝐼𝐷_𝑜𝑓𝑓,
𝑡𝑓𝑢 and 𝑡𝑟𝑢 analysis are listed in [19].
On the other hand, the conduction losses for
switching devices are calculated using the following
equations:
𝑝𝑐𝑜𝑛 = 𝑝𝑐_𝑀 + 𝑝𝑐_𝐷 (23)
𝑝𝑐_𝑀= 𝑅𝐷𝑆𝑜𝑛 𝐼𝑀_𝑟𝑚𝑠2 (24)
𝑝𝑐_𝐷= 𝑈𝐷𝑜 𝐼𝐷_𝑎𝑣 + 𝑅𝐷 𝐼𝐷_𝑟𝑚𝑠2 (25)
Where 𝑝𝑐_𝑀 , 𝑝𝑐_𝐷 are the conduction power losses
for the power MOSFET and diode as well as
𝐼𝑀_𝑟𝑚𝑠 , 𝐼𝐷_𝑟𝑚𝑠 are the root-mean square (RMS)
value of the drain current and diode current
respectively. 𝑅𝐷𝑆_𝑜𝑛, 𝑅𝐷 are the resistance of the
MOSFET and diode during conduction, 𝐼D_𝑎𝑣 is the
average value for the diode current and 𝑈𝐷𝑜 is the
diode on-state zero-current voltage.
The copper losses of the passive components are
approximately given by:
𝑝𝑐 + 𝑝𝑙 = 𝐼𝑐_𝑟𝑚𝑠2 𝑅𝑐 + 𝐼𝐿_𝑟𝑚𝑠
2 𝑅𝑙 (26)
Traditionally, inductor core loss has been divided
up into two components: hysteresis loss and eddy
current loss. According to the Steinmetz equation,
estimation of core losses based on the charts given
by the manufacturer (METGLAS, POWERLITE®
C-Cores) are calculated as follow:
𝑝𝑐𝑜𝑟𝑒 (W) = 𝑎. 𝑓𝑠𝑤𝑏. 𝐵𝑐 . 𝑤 (27)
𝐵 = 0.4𝜋.𝑁.𝛥𝐼 .10−4
𝐿𝑔 (28)
Where 𝑓𝑠𝑤 is the switching frequency in kHz, 𝐵 is
flux density in Tesla, while 𝑎 , 𝑏 , and 𝑐 are the
coefficients, which depend on the lamination
material, thickness, conductivity, as well as other
factors. It is assumed in this work as listed in
manufacture datasheet [20] that a=6.5, b= 1.5, and
c=1.74. 𝑤 is the weight of the core (kg), N is the
number of turns per coil, 𝛥𝐼 is the current ripple,
and 𝐿𝑔 is the length of the air-gab (cm).
Finally, the output power ( 𝑝𝑜 ) and converter
efficiency (𝜂) are shown the next equations:
EVS28 International Electric Vehicle Symposium and Exhibition 5
𝑝𝑜 = 𝑣𝑜 𝐼𝑜_𝑎𝑣𝑟 (29)
𝑝𝑙𝑜𝑠𝑠__𝑡𝑜𝑡 = 𝑝𝑐 + 𝑝𝑙 + 𝑝𝑐𝑜𝑟𝑒 + 𝑝𝑠 + 𝑝𝑐𝑜𝑛 (30)
𝜂 =𝑝𝑜
𝑝0+𝑝𝑝+ 𝑝𝑠+𝑝𝑐𝑜𝑛 (31)
Where 𝐼𝑜_𝑎𝑣𝑟 is the average output current and
𝐼𝑐_𝑟𝑚𝑠 , 𝐼𝐿_𝑟𝑚𝑠 are the RMS value of capacitor and
inductor current.
6 Simulation results and
discussion A comparative study is carried-out to evaluate the
proposed converter’s efficiency with regards to
other boost converter topologies. The system is
implemented using Matlab®/Simulink to
investigate the dynamic performance of the FP-
IBC compared to different topologies of boost
converter. Moreover, the efficiency of all studied
converters is investigated at different power rating
and switching frequencies. The converters’
parameters are summarized in Table. 1.
Simulation runs are carried-out using a battery
whose voltage is set to 100V as the converters
input voltage and resistive load (defined as
common load). The battery module comprises a
package with Ns cells that are connected in series
and Np batteries that are connected in parallel.
The parameters and characteristic of the battery
are introduced in Table. 2.
The output voltage reference is set to 200V. Using
(17)-(31), the switching and conduction losses are
calculated for each switch. The analytical
calculation is based on the datasheet of the
MOSFET module SKM121AR.
Figures.4, 5, 6, 7 and 8, respectively, show the
dynamic response of BC, MDBC, TP-IBC, MD-
IBC and FP-IBC during the step load variation. It
can be noticed from the figures that the input
current ripples frequency of FP-IBC is multiplied
by four compared to conventional BC and two
times compared to TP-IBC and MDBC. As a
result, the size of the passive components
(inductor, output capacitor, and EMI filter) is
reduced by 75% compared with conventional BC
and 50% compared with TP-IBC topology at the
same switching frequency. Moreover, the
dynamic response of the FP-IBC is faster than
other topologies and it has very small voltage
ringing at load step, because the interleaved
control between the power switches provides a
higher system bandwidth. It can be also noticed
from the figures that MD-IBC has approximately
the same dynamic performance as FP-IBC. For
this purpose, the current study should be
completed by the comparative efficiency study to
provide the efficient model for EVs application.
Table 1: dc/dc converter parameters
Items 𝐑𝐥(𝑚𝛺) 𝐑𝐜(𝑚𝛺) 𝐋(𝜇𝐻) 𝐂(𝜇𝐹) 𝒏 𝒎
Con-BC 64 0.50 600 800 1 1
MDBC 32 1.2 300 400 1 2
TP-IBC 32 1.2 300 400 2 1
MD-IBC 14 2 150 200 2 2
FP- IBC 14 2 150 200 4 1
Table 2: Battery module parameters
Capacity 65Ah
Initial SOC% 80%
Nominal voltage 19.2V
No of cell in series, Ns 5
No of parallel modules, Np 3
Figure 4: dynamic response of BC
EVS28 International Electric Vehicle Symposium and Exhibition 6
Figure 5: dynamic response of MDBC
Figure 6: dynamic response of TP-IBC
Figure 7: dynamic response of MD-IBC
Figure 8: dynamic response of FP-IBC
Figures. 9, 10, and 11 show the converters
efficiency versus the switching frequency range
from 5 to 40 kHz at power ratings of 5kW, 10kW,
and 20kW. It is worthy to note that for the switching
frequency range [10–20 kHz] and for the case of
power equal to 20kW, the efficiency decreases by
0.157 % for the FP-IBC and by 0.259 % for the
conventional BC (see Figure. 11).
For the change in the operating power rating range
from [10kW-20kW] at the same switching
frequency [20 kHz], the efficiency decreases by
0.1% for the FP-IBC and 9.69% for the
conventional BC (see Figures.10, 11).
Figure 9: Converters efficiency versus switching
frequency at 5kw.
EVS28 International Electric Vehicle Symposium and Exhibition 7
Figure 10: Converters efficiency versus switching
frequency at 10kw.
Figure 11: Converters efficiency versus switching
frequency at 20kw.
As it is shown in Figures. 9, 10, and 11, the
proposed FP-IBC converter has the highest
efficiency and hence, the lowest total power loss
compared to other boost converter topologies, in
particular at high power and high switching
frequencies.
Figure.12 presents the passive elements and
switching devices power losses of the FP-IBC and
other boost converter topologies at fsw =20kHz
and 20kW. This case is chosen to investigate the
worst case scenario in terms of power losses. On
the other hand, Figure. 13 shows the distribution
of all losses for each converter at the same
operating conditions.
As it is illustrated in Figures. 12 and 13, the
proposed converter is able to reduce the total
losses and especially the passive elements losses.
This comes from the reduction of its passive
components size by four times compared to
conventional BC and two times compared to TP-
IBC and MDBC. It is noteworthy that the
proposed converter (FP-IBC) efficiency
characteristics make it a good candidate for EVs,
particularly in high-power applications.
Figure 12: Converters MOSFET and Passive elements
power losses at 20kW.
Figure 13: Distributed power losses in (W) for each
converter
7 Conclusion In this research, an efficient FP-IBC has been
proposed for EVs applications. The performance of
the FP-IBC is compared against other boost
converter topologies and evaluated numerically
with power losses calculations. Simulation results
show that FP-IBC is the most efficient converter
among different boost converter topologies. FP-
IBC achieves this ranking because of the reduction
of its passive components size by four times
compared to conventional BC and two times
compared to TP-IBC and MDBC. Moreover, the
current and voltage ripples are also reduced by four
times compared to conventional BC and two times
compared to TP-IBC and MDBC. Simulation
results along with numeric comparison for
efficiency have demonstrated that the proposed
converter is very promising for EVs applications.
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Moataz Elsied was born in 1982. He
received the B.Sc. and M.Sc. in
Electrical Engineering from Helwan
University, Helwan, Egypt, in 2004,
2011. He is currently a PhD
candidate at LUSAC laboratory-
University of Caen Basse-Normadie,
Cherbourg, France. His research
interests include Energy
management systems, Power
electronic converters, and real time
control systems of smart grids.
A. Oukaour was born in Algeria
1963. He received Ph.D degree in
electrical engineering from Pierre &
Marie-Curie University (Paris IV),
France in 1993. From 1995 to 2003
he was an Associate Professor at the
Antilles and French Guyana
University. From 2003 to 2011 he
was an Associate Professor at the
Caen Basse Normandie University.
His current research interests are in
the field of Ageing State Diagnosis
of Power Sources (Batteries –
Supercapacitors – Full Cell …).
Hamid Gualous received the Ph.D.
degree in electronics from the
university Paris XI Orsay,France, in
1994. From 1996 to 2009, he was an
Associate Professor at the university
of Franche-Comte in FEMTO-ST
laboratory, France. His main
research activities are concerning
energy storage device
(SuperCapcitors and battery), hybrid
power sources, and energy
EVS28 International Electric Vehicle Symposium and Exhibition 9
management for vehicle
applications.
Radwan Hassan was born in Egypt,
1947. He received the B.Sc. and
M.Sc. in Electrical Engineering from
Helwan University, Helwan, Egypt,
in 1970, 1978. He received Ph.D
degree in electrical engineering in
1982 from (UMIST)- U.K. He is
currently a full professor in the
university of Helwan, Egypt. His
research interests include
Energy Conversion, Drives,
Control, Interfacing, and Industrial
applications
Amr Amin was born in Egypt, 1955.
He received the B.Sc. in Electrical
Engineering from Helwan
University, Helwan, Egypt and the
M.Sc. from New Mexico State
University, in 1978, 1982. He
received Ph.D degree in electrical
engineering from New Mexico State
University, Las Cruces, New
Mexico, U.S.A., 1986... He is
currently a full professor in the
University of Helwan, Egypt. His
research interests include Power
Electronics, Renewable Energy,
Electric Drives, Power Systems,
Computer-Based Controllers, and
Artificial Intelligence.
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