A Single-Phase Buck-Boost PFC Converter with Output-Voltage, Ripple-Reducing

Operation

SHIN-ICHI MOTEGI, and AKESHI MAEDATokyo Denki University, Japan

SUMMARY

This paper describes a new single-phase buck-boost

power-factor-correction (PFC) converter with output-volt-

age, ripple reducing operation. The converter consists of a

conventional buck-boost PFC converter and an additional

switch to obtain a freewheeling mode of the dc inductor

current, and is operated by two modulators. The first modu-

lator controls the buck-boost switch to obtain PFC. The

other modulator controls the square value of the instanta-

neous dc inductor current to perform the output-voltage-

ripple-reducing operation. In the two modulations, the time

integral value of the input and output currents in each

modulation period are controlled directly and indirectly,

respectively. Thus, modulation errors or undesirable distor-

tions of the input current and output voltage ripple are

eliminated even if the dc inductor current produces large

ripple in a low-frequency range. The theories and combina-

tion techniques for the two modulators, implementation,

and experimental results are described. © 1998 Scripta

Technica, Electr Eng Jpn, 126(2): 56�70, 1999

Key words: Buck-boost converters; high power fac-

tor output voltage ripple reduction; pulse space modulation;

input�output power balance.

1. Introduction

A single-phase high-power-factor converter is indis-

pensable for home air conditioners and medical treatment

equipment used in places where three-phase power supplies

cannot be obtained. However, its input power contains a

frequency component twice the source frequency, and

many ripples of dc-side output voltage and current occur

due to that component and this can be regarded as a difficult

point in the single-phase high-power-factor converter

which did not exist in the three-phase converter. Since these

output voltage ripples of the converter exert negative ef-

fects, for example, distorting the ac output voltage of the

inverter connected on the dc side, these ripples are reduced

usually by inserting a relatively large-capacity smoothing

capacitor at the output terminal of the converter. However,

the use of a large-capacity capacitor not only has the disad-

vantage that its cost, dimension, and weight will increase

but also the necessity of applying the electrolytic capacitor

which requires regular exchange. Accordingly, for the

aforementioned problems, methods for reducing the output

voltage ripples (such as the methods of providing additional

circuits to the main circuit [1�5] and the method of devising

input current reference [6]) have been reported centered

around the voltage-source converters. A method for dealing

with the same ripples on the side of the inverter connected

as a load but not for reducing the output voltage ripples on

the converter side has also been proposed [7].

On the other hand, the authors and others have re-

ported that the single-phase buck-boost converter can be

operated with a sufficiently high power factor by operating

it in current continuous mode [8]. Moreover, we have also

reported that by providing and properly controlling a re-

verse-blocking switch connected in parallel with the dc

inductor of the converter, the reduction of the output voltage

ripples is also possible in addition to the high-power-factor

operation on the input side [9]. In these high-power-factor

converters, the modulation (pulse space modulation) is

performed so that the area of the pulse current which is the

modulated wave is made agreeing with (proportional to) the

reference for every modulation period. The original form

of this pulse space modulation has been proposed in Ref.

10 using a single-edge carrier signal of single polarity; and

the results of the authors and others are that its concrete

function and theoretical basis are clarified and the practical

activities in various fields including the reducing control of

the output voltage ripples are demonstrated, such as the

application to various converters of not only single-phase

CCC0424-7760/99/020056-15

© 1998 Scripta Technica

Electrical Engineering in Japan, Vol. 126, No. 2, 1999Translated from Denki Gakkai Ronbunshi, Vol. 117-D, No. 7, July 1997, pp. 846�855

56

and buck-boost type [11, 12] but also three-phase and

voltage-source type [13, 14] by developing a simple gen-

eration method of single/double polarity double-edge car-

ries and so forth [9, 11]. Moreover, its usefulness has also

been confirmed in Refs. 15 to 20. This pulse space modu-

lation is a modulation scheme in which the distortion of the

input current due to the ripples will not occur by adopting

the improved carries while the modulated wave is section-

ally integrated and generated even in the case where the

ripples exist in the modulated wave such as the dc output

current of voltage of voltage-source converter or the dc

(inductor) current-source converter.

However, in the reducing control of the output volt-

age ripples proposed previously by us [9], imbalance of the

input and output powers occurs easily, and the unnecessary

rise (rise of power device current) and the deficiency (ob-

struction to output-voltage-ripple-reducing operation) of

the dc inductor current due to that imbalance remain as

problems. With respect to this, we have proposed a simple

method in which by properly controlling the instantaneous

power of the dc inductor or the square instantaneous value

of the current such that the low-frequency components of

the output power ripples will become zero, the reduction of

the output voltage ripples (low-frequency components) and

the balance implementation of the input and output powers

can be realized simultaneously [21]. However, since the

effect of the internal resistance of the dc inductor on the

instantaneous power is neglected even in this control, a

small low-frequency ripple voltage (double frequency of

source frequency) has remained in the output voltage. Ac-

cordingly, we will propose in this paper an improved out-

put-voltage-ripple-reducing control considering the

internal resistance of the dc inductor. By using this control,

it is possible to remove, almost completely, the ac compo-

nent of double frequency from the instantaneous value of

the output voltage; and, moreover, it is learned that the

capacitance of the dc filter capacitor can be reduced to about

1/500 compared to the case of not using this control. Thus,

the miniaturization of this capacitor and, moreover, the

maintenance-free implementation due to the application of

the film capacitor can be expected.

On the other hand, even if the proposed improved

control is completely implemented, the low-frequency rip-

ples will remain in the output power when there exists a

distortion in the source voltage. After studying the effect of

this source voltage distortion on the output voltage ripples,

no effect of the extent which might be an obstacle in

practice is seen, and so it is learned that the proposed control

can be sufficiently offered for practical use. We will also

touch on this point in the following.

2. Configuration and Operation of Main Circuit

2.1 Configuration of main circuit

The single-phase buck-boost converter shown in Fig.

1 has the distinctive feature that a switch for dc inductor

current freewheeling (SWF, DF) is newly added to the

conventional constituting elements of single-phase source,

single-phase diode bridge (D1 to D4), harmonic filter (Lf,

Cf), buck-boost switch SWF, buck-boost dc inductor Ld,

buck-boost dc diode DO, dc filter capacitor CO, and load

resistor RO.

2.2 Operation of main circuit

Depending on the operation scheme, the buck-boost

converter can be roughly divided into the current intermit-

tent scheme in which it is operated by intermitting the dc

inductor current for all modulation periods and the current

continuous scheme in which it is operated by always main-

taining the current. In the case of the current intermittent

scheme, there are advantages such as that the modulator is

simple and the inductance of the dc inductor can be made

smaller; however, since the inductor current becomes a

pulsed triangular wave of high crest value and the losses

due to the inductance and switching devices become large,

this scheme has the disadvantage that it is limited to small-

capacity use. In the current continuous scheme, on the other

hand, large-capacity implementation is possible compared

with the current intermittent scheme; however, since the

inductance current will pulsate largely, errors will occur in

the modulation of the input current when the inductance of

the dc inductor is not sufficiently large. In this case, suffi-

cient input current modulation cannot be realized unless the

pulsation of the inductor current itself is reduced or the

current is detected and performed with some measures in

the modulation method. Accordingly, in this converter, by

adopting the pulse space modulation which is a modulation

Fig. 1. Proposed single-phase buck-boost PFC

converter system.

57

method for solving the above-mentioned problems, the

increase of the conversion capacity is devised while the

modulation of the input current under pulsating current is

made a relatively simple one. Besides, the buck-boost con-

verter can step up (boost) the output voltage higher or step

down (buck) lower than that; and so various applications

reviving this distinctive feature can be expected.

2.3 Operating modes of main circuit operation

With respect to the dc inductor Ld, the operation of

this buck-boost converter can be divided into the following

four operating modes (circuit states) depending on its con-

nection state with the source and load.

· Mode I: Ld is connected to source side (SWS: ON,

SWF: OFF)

· Mode II: Ld is connected to load side (SWS: OFF,

SWF: OFF)

· Mode III: Ld is not connected to either side and the

inductor current freewheels (SWS: OFF, SWF:

ON)

· Mode IV: The current of Ld is zero (the dc inductor

is used under current intermittent operation; this

mode will be excluded in the proposed scheme)

Figure 2 shows the circuit states in these operating modes,

and the source diode bridge and so forth of Fig. 1 are

omitted.

From the above, since the mode in which the dc

inductor is connected to the source side is Mode I only, the

sinusoidal implementation of the input current, that is,

high-power-factor operation, can be realized by properly

controlling the period of this mode for every modulation

period as described later, by using the pulse space modula-

tion. Similarly, by properly controlling the period of Mode

II in which the dc inductor is connected to the load side, the

ripple-reducing operation of the load-side output voltage

can be realized. However, in order to properly control the

periods of these operating modes while satisfying the above

conditions (high-power-factor operation and ripple-reduc-

ing operation) simultaneously, it is necessary to provide the

mode in which the dc inductor is connected neither to the

input side nor the load side, namely, the freewheeling mode

(Mode III) of the dc inductor current proposed by us.

3. Principles of Control and Modulation

The controller/modulator of the proposed single-

phase buck-boost high-power-factor converter with output-

voltage-ripple-reducing operation consist of a modulator

(pulse space modulator) for making the input current a

sinusoidal wave in phase with the source voltage and an

output-voltage-ripple-reducing controller for reducing the

low-frequency ripples of the output voltage. In the follow-

ing, the principles of controller and modulator are ex-

plained by separating them into the pulse space modulator

and output-voltage-ripple-reducing controller.

3.1 Input-side high-power-factor control

In this high-power-factor converter, keeping the

miniaturization of the dc inductor in mind, the pulse space

modulation is adopted for avoiding the occurrence of the

input current distortion due to the large ripples of the

inductor current that occur at the time of miniaturization

(low-inductance implementation). Moreover, in addition to

coping with the measure by control and modulation

adopted in the proposed scheme, a method has also been

proposed in which an LC-parallel resonant circuit for filter-

ing the double-frequency component of the source fre-

quency which is the main component of the pulsating

current, is inserted in the main circuit [22]. In the latter, the

pulsation itself can be suppressed and the small-size light-

weight implementation of the equipment becomes possible

compared with the conventional scheme using a large-

capacity dc inductor; however, in the case of the proposed

scheme, since it is coped inside the controller/modulator,

further small-size light-weight implementation of the

equipment is possible even compared with the latter

method. In the following, we will introduce the details of

the principle of the pulse space modulation shown in Fig.

3.

The carrier signal nCAR used in this pulse space

modulation is generated by integrating the dc inductor

current signal iL (including ripples) detected via the current

detector while the integrator output is reset by the resetting

signal nRST for every constant (modulation) period TSW, and

becomes a sawtooth wave expressed asFig. 2. Operating modes.

58

On the other hand, as the area in each modulation

period of the buck-boost switch current iSW, that is, the

reference sB* of the sectionally integrated value sB, we use

the absolute value iB* of the input current reference iS

* in

phase with the source voltage multiplied by the modulation

period TSW:

Comparing these carrier signal nCAR and space refer-

ence sB* , the period of Mode I in which the dc inductor Ld

is connected to the source side is determined; and as shown

in the lower part of Fig. 3, Mode I is maintained from the

starting time t = tk of each modulation period to the time

t = tkg where the reference sB

* and the carrier signal nCAR

intersect. Therefore, the dc inductor current iL will flow to

the source side via the buck-boost switch SWS during this

period (TON; t = tk to t = tkg in Fig. 3). At this point, the area

sB of the switch current in each modulation period becomes

and agrees with the reference sB* . Besides, since the switch-

ing frequency component of the switch current iSW is fil-

tered by the harmonic filter (Lf, Cf), and its polarity is

reversed at every half period by the diode bridge and

becomes the input current iS, the average value in each

modulation period of the input current iS or the low-fre-

quency component of the input current iS will agree with

the input current reference iS*.

Moreover, the modulation period can be easily and

freely adjusted by changing the period of the resetting

signal nRST.

3.2 Output-voltage-ripple-reducing control

As described previously, as a scheme for removing

the low-frequency ripple of the output voltage in the single-

phase buck-boost high-power-factor converter, we have

proposed providing a freewheeling switch (SWF, DF) in

addition to the buck-boost switch SWS; however, in the

control scheme previously reported [9], since these two

switches are controlled by two independent current refer-

ences, there is the problem that imbalance between input

and output powers easily occurs. Accordingly, we have

proposed a method for reducing the output voltage ripples

by making the instantaneous value of the dc inductor power

and the ac component of the input power equal by properly

controlling the instantaneous value (LdiL2 /2) of the stored

energy of the dc inductor, that is, the square value of the

instantaneous inductor current (iL2), and thus indirectly per-

forming the control such that the low-frequency component

of the output power ripple will become zero (or the average

value in each modulation period is constant) [21]. In this

proposal, since the dc inductor current reference is com-

puted from the input current reference, the output-voltage-

ripple-reducing control becomes possible without the

occurrence of the imbalance between input and output

powers. However, since the consumed power of the internal

resistance of the dc inductor in that scheme is neglected, the

ripple component of that power is supplied from the load

side, and a pulsation of double source frequency will occur

in the instantaneous value of the output power; and so a

pulsating voltage of that frequency has barely remained in

the output voltage.

For improving this, we will propose in this paper an

improved output-voltage-ripple-reducing control in which

the instantaneous value of the pulsating power of the entire

dc inductor including the internal resistance of the dc

inductor is made equal to the ac component of the input

power. By using this control, the double-frequency ac com-

ponent can almost completely be removed from the instan-

taneous value of the output voltage. In the following, for

output-voltage-ripple reduction, the square-value reference

of the dc inductor current, (iL2)*, will be derived by separat-

ing it into the ac component (iL2)AC

* (abbreviated hereafter as

ac component) and the dc component (iL2)DC

* , excluding the

switching ripples.

Fig. 3. Principle of pulse space modulation.

(1)

(2)

(3)

59

This converter system is considered as ideal (exclud-

ing dc inductor Ld) and the losses of the switching devices

are neglected; and if pL is the instantaneous value of the

input power to the dc inductor Ld excluding the internal

resistance, pr is the instantaneous value of the consumed

power of the internal resistance rL of the dc inductor, and

pO is the instantaneous value of the output power, then the

instantaneous value ps of the input power can be expressed

as

Moreover, if the high-power-factor operation is carried out,

and the source voltage ns and current is are in-phase sinusoi-

dal waves [nS = Ö̀̀2Vs sin(wt), is = Ö̀̀2 Is sin(wt), w = 2pfs,fs is the source frequency), the instantaneous value ps of the

input power can be given as

Furthermore, the instantaneous value pO of the output

power, the instantaneous value pL of the dc inductor power,

and the instantaneous value pr of the consumed power of

the internal resistance can be expressed similarly by the sum

of the dc component and the ac component, respectively:

Now, from Eq. (4),

can be obtained and substituting this equation into Eqs. (5)

and (6) yields

Separating the above equation into the dc component

and the ac component yields

and the power flows of Fig. 4 can be obtained. However,

the upper part of the figure is the case where the output-

voltage-ripple-reducing control is not performed and the

lower part is the case where the same control is performed.

Now, in Eq. (10), if

the ac component pO-AC of the output power becomes zero

as shown in the lower part of Fig. 4; and if the load is a

(quasi-) constant-power load or a (quasi-) constant-resis-

tance load, the output voltage ripple will disappear.

However, if the respective instantaneous values of the

voltage (excluding the voltage of the resistance component)

and current are nL and iL, the instantaneous value pL of the

dc inductor power becomes

On the other hand, the instantaneous value pr of the

consumed power of the internal resistance rL of the dc

inductor can be given by

Here, iL2 will be called the square instantaneous value

of the dc inductor current (moreover, the square root of the

above-mentioned square instantaneous value will be called

the �nonsquare instantaneous value� in the following to

distinguish it from the square current). Separating it into the

ac component (iL2)AC and the dc component (iL

2)DC and

letting

Eq. (12) becomes

(10)

(4)

(5)

(6)

(7)

(8)

(9)

(11)

(12)

(13)

(14)

Fig. 4. Power flow on buck-boost converter.

60

Separating the above equation into the ac component

and the dc component yields

However, the dc component pL in the above equation is zero

in the steady state. Similarly, by substituting Eq. (14) into

Eq. (13) and separating it into the ac component and dc

component, the consumed power of the internal resistance

of the dc inductor becomes

Hence, when the ac component of the instantaneous

value of the output power, pO-AC = 0, is satisfied, the

following equation holds by substituting Eqs. (16) and (19)

into Eq. (11):

Hence, rewriting the left-hand side of the above equa-

tion using Eq. (5), we have

and the square current equivalent circuit of Fig. 5 can be

obtained. Here, the left-hand side of the above equation is

equivalent to the source voltage n of angular frequency 2wand the right-hand side is equivalent to the series circuit of

the inductor L with an inductance Ld/2 and the resistor R

with a resistance rL. Moreover, the ac component (iL2)AC of

the dc inductor square current is equivalent to the current i

flowing in the equivalent circuit. Therefore, solving for the

ac component (iL2)AC of the square current from the equiva-

lent circuit, we obtain

where

Here, ZL is the equivalent impedance of the equivalent

inductance (Ld/2) and resistance rL of the equivalent circuit

of Fig. 5 at angular frequency 2w, and d, q are the phase

angles of the equivalent impedance.

Equation (23) is the condition for obtaining pO-AC =

0, that is, making the ac component of the output voltage

nO zero; thus, it is seen that if the control is performed such

that (iL2)AC satisfies Eq. (23), the ripples of the output

voltage nO will disappear. Moreover, ZL and the phase angle

q on the right-hand side of the above equation are the

inherent constants of the dc inductor and will not be affected

by the exchange power, and so it is possible to set them

inside the controller/modulator beforehand in the design

stage of the dc inductor. Besides, since the output voltage

ripples will increase when errors occur in these setting

values, it is necessary to study the effects of the setting

errors on the proposed control; however, we intend to report

our findings elsewhere.

Now, from Eqs. (14) and (23), the reference (iL2)* of

the square instantaneous value of the dc inductor current for

reducing the output voltage ripples becomes

where

Moreover, from the above equation, the reference iL* of the

nonsquare instantaneous value of the dc inductor current

can be expressed as

where

(18)

(19)

(20)

(21)

(22)

(16)

(17)

(15)

(23)

(23a)

(24)

Fig. 5. Equivalent circuit to solve square current

equation.

(24a)

(25)

(25a)

61

However, the dc component (iL2)DC of the above ref-

erence is arbitrary in the range where there is no hindrance

to the output-voltage-ripple-reducing control; however,

when the loss due to the internal resistance rL of the dc

inductor is taken into account, it is desirable to make it the

minimum (optimal) value [let this be the dc component

reference (iL2)DC

* ] on satisfying the above-mentioned condi-

tion. In the following, we will derive the average value of

the square instantaneous value reference (iL2)* of the dc

inductor current during the source period, namely, the dc

component reference (iL2)DC

* .

At point A of the main circuit shown in Fig. 1, the

current equation is

Here, the low-frequency components (including the dc

component, abbreviated as low-frequency component in

the following) excluding switching ripples of the respective

currents will be expressed by adding �#� to the symbol of

the respective instantaneous values, as �iL#�; and when Eq.

(26) is written as the current equation in the low-frequency

region, it becomes

In the above equation, the low-frequency component iSW# of

the buck-boost switch current is modulated by the pulse

space modulation such that it is equal to the input current

reference iB* , and the low-frequency component iL

# of the dc

inductor current is controlled by the proposed ripple-reduc-

ing control such that it is equal to the dc inductor current

reference iL* expressed by Eq. (25). Moreover, by perform-

ing the ripple-reducing control, the low-frequency compo-

nent iD# of the buck-boost diode current is equal to the

average value IO of the output current and becomes constant

and smooth. Therefore, rewriting Eq. (27) from the above,

we obtain

Here, Fig. 6(a) plots Eq. (28) over a half period of the

source, where from the bottom are shown the input current

reference iB* , this reference added to the average value IO of

the output current (iB + IO), and the nonsquare instantane-

ous value reference iL* of the dc inductor current. Moreover,

Fig. 6(b) shows the square currents when the respective

waveforms of Fig. 6(a) are squared; Figs. 6(c) and 6(d)

show the enlargements near wt = p / 4 in Fig. 6(b).

Now, as described in the preceding section, the dc

inductor current in this buck-boost converter flows to the

source side in Mode I and to the load side in Mode II and

becomes the input current iB and the output current IO,

respectively. Therefore, in order to perform the high-power-

factor operation on the source side or the output-voltage-

ripple-reducing operation on the load side,

must be satisfied in all modulation periods. Namely, on

satisfying the above equation, the dc component reference

(iL2)DC

* can be derived under the condition that the nonsquare

instantaneous value reference iL* of the dc inductor current

and the sum of the input current reference and the average

value of the output current, (iB* + IO ), come in contact at one

point. However, since the control equation will become

complicated in the derivation of the dc component reference

based on this condition, it is not practical in performing the

output-voltage-ripple-reducing control. Accordingly, we

will derive the dc component reference (iL2)DC

* under the

simple condition that the nonsquare reference iL* (minimum

value) and the sum of the input current reference and the

average value of the output current, (IB* + IO), agree at the

phase where the nonsquare instantaneous value reference

iL* of the dc inductor current is a minimum value as shown

in Fig. 6(a) (when the source voltage is sinusoidal and the

internal resistance of the dc inductor is neglected:

wt = p / 4). Therefore, letting the minimum value of the

nonsquare instantaneous value reference iL* of the dc induc-

tor current be (iL*)-MIN and substituting the phase p/4 into

Eq. (29),

(26)

(27)

(28)

Fig. 6. Model waveforms in buck-boost converter.

(29)

62

can be obtained. Here, squaring both sides of Eq. (30) yields

Moreover, since ((iL*)

wMIN)2 = (iL2)wMIN

* , the above equation

is

Furthermore, from Eq. (24), the minimum value (iL2)

wMIN* of

the square instantaneous value reference of the dc inductor

current can be expressed as

Here, substituting Eq. (33) into the left-hand side of Eq. (32)

and rearranging for (iL2)DC,

can be obtained. However, if we let (iL2)DC given by the

above equation be the dc component reference (iL2)DC

* of

(iL2)*, the period

will be formed near wt = p /4 as shown in Fig. 6(c), and it

is necessary to avoid this because the output-voltage-ripple-

reducing operation will be hindered a little. As shown in the

following equation, we have multiplied the entire right-

hand side of Eq. (34) by a constant KDC of over 1 (setting

in such a way that the output voltage ripple will not occur

at rated load) and let it be the dc component reference

(iL2)DC

* , and the output-voltage-ripple-reducing operation is

carried out by letting this reference have a margin as shown

in Fig. 6(d).

This is the dc component reference (iL2)DC

* of (iL2)* for

performing the output-voltage-ripple reduction while mini-

mizing the loss due to the internal resistance rL of the dc

inductor.

By the above theoretical analysis, the average value

of the optimal (minimum) square instantaneous value ref-

erence (iL2)* of the dc inductor current in the source period,

namely, the dc component reference (iL2)DC

* , can be ob-

tained; and from this and Eq. (24), the reference (iL2)* of the

square instantaneous value of the dc inductor current for

ripple reduction can be obtained (see Fig. 6). Therefore, if

the control is performed such that the square value iL2 of the

dc inductor current agrees with the reference (iL2)*, the

output voltage ripples can be eliminated theoretically.

3.3 Controller/modulator

Figure 7(a) shows the pulse space modulator and the

output-voltage-ripple-reducing controller obtained as dis-

cussed above; Fig. 7(b) shows the model waveforms and

Fig. 7(c) the computing unit of the average value of the

output current.

As shown inside the broken line in the upper part of

Fig. 7(a), the pulse space modulator for performing the

sinusoidal implementation of the input current is composed

of an absolute-value circuit (ABS) for converting the input

current reference iS* into the bridge current reference iB

* , an

amplifier for converting the reference iB* into the pulse-

space reference sB* of the input current by multiplying it

with the modulation period TSW, a reset integrator (Reset-

ting-Integrator) for generating the carrier signal nCAR by

integrating the dc inductor current while resetting the out-

put for every modulation period, and a comparator (Comp.

1) for generating the gate signal nGS of the buck-boost

switch SWS by comparing the carrier signal and the pulse-

space reference sB* ; and the high-power-factor operation

becomes possible relatively simply.

On the other hand, as shown inside the broken line in

the lower part of Fig. 7(a), the output-voltage-ripple-reduc-

ing controller is composed of a square computing circuit

(SQR1) for squaring the instantaneous value iL of the dc

inductor, a multiplier for obtaining the computed value

pS@ (the computed value will be expressed by adding �@�

in the following) of the input power instantaneous value by

multiplying the sinusoidal input current reference

iS* = Ö̀̀2 IS

* sin (wt) (for adjusting the amplitude IS* for the

purpose of output voltage control) which is in phase with

the voltage nS, a band-pass filter (BPF) for filtering the

double-frequency components of source frequency and ob-

taining the ac component pS-AC@ from the computed value

pS@, a phase shifter (Phase-shifter) for obtaining the ac

component reference (iL2)AC* (sine wave) of the square in-

stantaneous value of the dc inductor current by phase-shift-

ing pS-AC@ (cosine wave) by �90 + q°, an ac-component

eliminator for obtaining the (dc component) average value

pS@ by eliminating the ac component from the computed

value pS@ of the input power instantaneous value, a square

computing circuit (SQR2) for squaring the sum of the input

current amplitude reference IS* and the output current aver-

age value IO, a comparator (Comp. 2) and a D Flip-Flop for

(30)

(31)

(32)

(33)

(34)

(35)

(36)

63

obtaining the switching pattern nGF of the freewheeling

switch SWF shown in the model waveforms of Fig. 7(b)

from the pulse patterns (nF, nGS) which are obtained from

the comparator output and the pulse space modulator; and

the reduction of the output voltage ripples becomes possible

with a relatively simple configuration.

Now, in the proposed controller/modulator, the out-

put current iO is detected from the main circuit, and by

passing through the low-pass filter (LPF2), the output cur-

rent average value IO is obtained; however, in the practical

device, computation of the output current average value is

possible by the circuit shown in Fig. 7(c). Besides, although

neglected in the experiment, it is necessary in the practical

device to add the low-pass filter (LPF1) and the error

amplifying unit (PI) shown in Fig. 7(a) for the purpose of

the output voltage control.

Moreover, in our previous proposal [21], since the

switching pattern of the freewheeling switch SWF is set as

nF, nF has chattered and generated unnecessary switching

in the SWF when the slope of the square reference (iL2)* of

the dc inductor current is small or negative. Accordingly, in

the proposed controller/modulator, this is avoided by letting

SWF be �ON� during the period from the rise of the nF pulse

to the end of each modulation period as shown in the model

waveforms of Fig. 7(b). Moreover, in Ref. 21, since the ac

component reference (iL2)AC

* of the square instantaneous

value of the dc inductor current is computed from the input

current reference only, the variation of the source voltage

(effective value) VS is not considered, and so the output-

voltage-ripple reduction is not sufficient on that point. With

respect to this, in the proposed controller/modulator, since

the abovementioned reference (iL2)AC

* is obtained from the

computed value of the input power instantaneous value

which is obtained by the multiplication of the source volt-

age and the input current reference, there is theoretically no

increase of output voltage ripple due to the variation of the

source voltage (effective value).

Now, in the analysis of this section, the source voltage

waveform is treated as a sine wave; however, since the

actual commercial source voltage contains the harmonic

components to some extent, it is necessary to study these

harmonics and this detailed study will be carried out in

section 4.

4. Experimental Results

For confirming the appropriateness of the theory

discussed in section 3, Figs. 8 to 12 show the various

experimental results obtained in a prototype system using

the buck-boost high-power-factor converter shown in Fig.

1 and the controller/modulator of Fig. 7. Moreover, Tables

1 and 2 show, respectively, the constants and ratings of the

main circuit in the prototype system as well as the major

measured values for the various experiments. In the follow-

ing experiments, to verify the effect of the proposed ripple-

reducing control, a sinusoidal wave is used in the source

Fig. 7. A proposed controller/modulator for buck-boost PFC converter with output voltage-ripple-reducing operation.

64

voltage and the converter is operated with the same main

circuit constants, and the case of applying and not applying

the control are compared. However, in practice, since the

distortion of the source voltage actually exists, it is neces-

sary to evaluate the effect of this distortion on the proposed

control. Therefore, an experiment applying the proposed

control is also performed using a distorted source voltage.

In the following, the output-voltage-ripple factor x is

defined by the following equation using the effective value

VO-AC of the low-frequency component (measured up to 2

kHz) of the output voltage (ac component) and the output

voltage average value VO-AVG:

x = Effective value VO-AC of output-voltage ac component

Output-voltage average value VO-AVG

´ 100%

(37)

Moreover, in the following experiments, the constant

KDC is set such that the output voltage ripples will not occur

at rated load shown in Table 2.

4.1 Input-side high-power-factor operation

Figure 8 shows the experimental waveforms and

spectra in various parts when the output-voltage-ripple-

reducing control is carried out. It is seen from Fig. 8(a) that

since the input current iS has drawn a sinusoidal wave in

phase with the source voltage nS, the pulse space modulator

has operated according to the principle and the low-distor-

tion operation of the input current has been realized. More-

over, as shown in Fig. 8(b), the low-order harmonic

components are very small and the measured value of the

Table 2. Major measured data

Ripple

reducing

control

dc filter

capacitor

CO (mF)

Source voltage Input current

Input

power

PS (kW)

Total

power

factor

TPF (%)

Inductor

current

average

value

iL-AVG (A)

Waveform VS (VRMS) THDv (%) IS (ARMS) THDi (%)

with

1007.8

sine wave 100.1 0.39 15.1 0.95 1.51 99.5 38.0

without sine wave 100.2 0.96 14.1 1.16 1.41 99.6 24.9

with distorted

wave

99.7 1.82 15.0 0.91 1.49 99.6 37.6

Table 1. Power circuit constants and ratings

dc inductor Ld: 7.75 mH (50 Hz)

internal resistance of dc

inductor

rL: 0.066 W

inductor of harmonic filter Lf: 0.6 mH

capacitor of harmonic filter Cf: 9.4 mF

capacitor of dc filter CO: 1007.8 mF

diode bridge (for general

commutation)

D1 to D4: 800 V, 100 A

self-excited switches SWS, SWF: IGBT 600 V,

150 A

high-speed diodes DO, DF: FRD 600 V, 150 A

(trr = 300 ns)

modulation (carrier)

frequency

fCAR: 10 kHz

dc output voltageOutput current

IO (AAVG)

Output power

PO (kW)

Conversion

efficiency

h (%)

Reference

diagramVO (VAVG) VO-AC (VRMS)

Ripple factor x(%)

100.3 0.148 0.147 12.6 1.26 83.7 Fig. 8

100.2 11.63 11.61 12.5 1.25 88.8 Fig. 9

99.8 0.209 0.210 12.4 1.24 83.1 Fig. 12

65

distortion factor THDi of the input current iS is 0.95%

(measured within 1 kHz). It is seen that the switching

frequency component (about 10 kHz) has also been sup-

pressed sufficiently by using the harmonic filter (Lf, Cf) as

shown in Fig. 8(c). From these experimental waveforms

and spectra, it can be observed that the input power factor

TPF is very high, although its measured value is 99.5%

(measured within 20 kHz); thus, a good value is obtained.

On the other hand, as shown in Fig. 8(d), the dc inductor

current iL has pulsated largely, but the effect of this pulsa-

tion is not seen in the input current iS due to the adoption

of the pulse space modulation. Moreover, it is seen from

Figs. 8(d) and 8(h) that the change (envelope of the apexes

of the sawtooth wave signal) of the maximum value of the

carrier signal nCAR for every modulation period agrees well

with the change of iL and so the carrier is generated accord-

ing to the principle.

Thus, it can be confirmed that pulse space modulation

has worked according to the principle and the modulation

of the input current without causing the distortion of the

input current due to the pulsation of the dc inductor current

is possible.

Fig. 8. Experimental results on sinusoidal source voltage (with output-voltage-ripple-reducing operation).

Fig. 9. Experimental results on sinusoidal source voltage (without output-voltage-ripple-reducing operation).

66

4.2 Output-voltage-ripple-reducing operation

Figure 9 shows the experimental waveforms and

spectra in various parts when the output-voltage-ripple-

reducing control is not used. As shown in Figs. 9(a) and

9(b), the high-power-factor operation on the source side has

been performed; however, it is seen from the output voltage

waveforms and the spectra of the ac components shown in

Figs. 9(c) and 9(d) that large ripples of double frequency of

the source frequency are contained in the output voltage nO.

The ripple factor x in this case is 11.61% from Eq. (37).

On the other hand, as shown in Fig. 8(e), the output

voltage ripples are not completely eliminated in the output

voltage waveform because of the control error of the out-

put-voltage-ripple-reducing controller; however, a good

output voltage waveform with very few ripples is obtained

at an output voltage ripple factor x = 0.147%. Moreover,

the ripple-reducing effect of the output voltage can also be

confirmed from the spectra of Figs. 9(d) and 8(f).

Furthermore, it is seen from Fig. 8(g) that the square

instantaneous value reference (iL2)* of the dc inductor cur-

rent and the square iL2 of the actual current have overlapped

and both agree well to the extent that they are hard to

distinguish.

Besides, as shown in Table 2, the efficiency has

dropped somewhat because of performing the output-volt-

age-ripple-reducing operation; however, this is due to the

fact that the average value iL-AVG of the dc inductor current

has increased due to the application of the output-voltage-

ripple-reducing control (see Table 2) and so the copper loss

and the losses due to the switching devices have increased.

Thus, it can be confirmed that the proposed ripple-re-

ducing control has worked effectively. Moreover, the low-

frequency output voltage ripples caused by the tracking

errors of the output-voltage-ripple-reducing controller and

the high-frequency output voltage ripples caused by the

switching can be reduced by raising the switching (modu-

lation) frequency.

4.3 Static characteristics

Figure 10 shows the experimental results, namely, the

static characteristics of the proposed converter, when the

output voltage average value vO is a parameter and the

amplitude of the input current reference iS* (or its effective

value IS*) is changed.

Figure 10(a) shows a good result with a total input

power factor TPF of over 99% in the wide range of output

power pO between 0.3 and 1.3 kW. Moreover, due to the

disappearance of the narrow-width pulse caused by the

decrease of the fundamental component, the total distortion

factor THDi of the input current has increased in the very

light load region where the distortion of the input current

increases; however, a good result of below 5% has been

obtained at an output power pO of over 0.2 kW.

As shown in Fig. 10(a), although the output voltage

ripple factor x has become somewhat larger because the

output current becomes larger at the output voltage of 60 V,

a good result of below 1% has been obtained in all meas-

urement regions.

As shown in Fig. 10(b), the efficiency h has dropped,

for the respective output voltages, as the output power

becomes larger, namely, as the average value iL-AVG of the

dc inductor current becomes larger. This is because the

copper loss in the inductor and the losses due to the switch-

ing devices become larger accompanying the increase of

the dc inductor current.

It has become clear from the above experimental

results that the high-power-factor and low-distortion-factor

operations as well as the output voltage of small ripples can

Fig. 10. Static characteristics.

Fig. 11. dc filter capacitance versus output voltage

ripple factor.

67

be obtained in the proposed converter with output-voltage-

ripple-reducing operation, which can withstand practical

use in the output voltage range from buck to boost (between

60 and 180 V in the experiments) and the considerably wide

range of output power (between 0.3 and 1.3 kW in the

experiments).

4.4 Reduction of the capacitance of dc filter

capacitor

For evaluating the ripple-reducing function of the

proposed converter with output-voltage-ripple-reducing

operation, we have carried out experiments by changing the

capacitance of the dc filter capacitor. In this case, the source

voltage is a sinusoidal wave, the output voltage nO is 100

VAVG, and the output power pO is 1.25 kW.

Figure 11 shows the output voltage ripple factor xwith respect to the capacitance CO of the dc filter capacitor

for the cases of performing and not performing the ripple-

reducing control. As seen from the fact that the ripple factor

in the case when the ripple-reducing control is used with a

capacitance of about CO = 40 mF and the ripple factor in the

case when the ripple-reducing control is not used with a

capacitance of about CO = 20,000 mF are about the same, it

is possible to reduce the capacitance of the dc filter capaci-

tor from 20,000 mF to 40 mF by using the output-voltage-

ripple-reducing control, and so the reduction factor of the

capacitance in this case is 1/500. Therefore, the capacitance

of the dc filter capacitor can be greatly reduced and the

small-size light-weight implementation of the dc filter ca-

pacitor and, moreover, the maintenance-free implementa-

tion without using chemical capacitors can be expected.

4.5 Effect of source voltage distortion in

output-voltage-ripple-reducing control

From the above experimental results, the aspects of

the high input power factor, the output voltage ripple reduc-

tion, and the possibility of the reduction of the capacitance

of the dc filter capacitor have become clear. However, since

the actual commercial source voltage has a certain degree

of harmonic distortion, it is necessary to study the effect of

this harmonic distortion on the output voltage ripples. Here,

we will study the effect of the ripple-reducing control under

the source voltage with distortion.

When the proposed converter with output-voltage-

ripple-reducing control, shown in Fig. 12(a), is operated

with a source voltage of total distortion factor THDn =

1.82% (see Fig. 12(b) for spectrum), the output voltage

waveform is shown in Fig. 12(c) and the spectrum of the ac

component of the output voltage is shown in Fig. 12(d).

From the spectrum of Fig. 12(d), the ripple of the double

frequency of the source frequency has been reduced, al-

though the increase of the output voltage ripples of 4, 6, and

8 times the source frequency is seen compared with the case

of the source voltage being a sinusoidal wave [Fig. 8(g)].

This is due to the fact that the frequency components of 4,

6, and 8 times the source frequency are contained in the

input power pS due to the low-order harmonic components

contained in the source voltage as we have assumed that the

source voltage is a sinusoidal wave in the proposed ripple-

reducing controller. However, the output-voltage ripple fac-

tor x is 0.210% and so a sufficient ripple-reducing effect

has been obtained even compared with the case of sinusoi-

dal voltage (x = 0.147%; see Fig. 8).

Thus, it is learned that output voltage ripple reduction

can be sufficiently performed by the proposed control even

in the case of practical source voltage waveform with a

certain degree of distortion.

5. Conclusions

By incorporating the pulse space modulation and the

(square) instantaneous value control of the dc inductor

current into the single-phase buck-boost high-power-factor

converter with a freewheeling mode of dc inductor current,

Fig. 12. Experimental results on distorted source voltage (with output-voltage-ripple-reducing operation).

68

we have studied the converter in which the high-input-

power-factor characteristics and the output-voltage-ripple-

reducing function have been realized; and its fundamental

functions are verified by a series of experimental results.

The proposed output-voltage-ripple-reducing control

method is a method for realizing the output voltage ripple

reduction and the balance of the input and output powers

simultaneously, by properly controlling the instantaneous

value of the stored energy of the inductor including the

internal resistance of the dc inductor, that is, the square

instantaneous value (iL2) of the current, and thus indirectly

performing the control such that the low-frequency compo-

nents of the output power instantaneous value become

constant. By doing so, elimination of the ripples has be-

come possible theoretically. Moreover, from the analysis of

section 3, the optimal (minimum) control of the dc inductor

current has been made clear and the suppression of the loss

due to the internal resistance of the inductor has been made

possible.

Moreover, the proposed converter can greatly sup-

press the low-frequency ripples of the dc output voltage;

and since the capacitance of the dc filter capacitor can be

reduced to about 1/500, compared with the case of not using

the proposed control, the small-size implementation of the

capacitor and, moreover, the maintenance-free implemen-

tation without using chemical capacitors can be expected.

Furthermore, in the theoretical analysis of this paper,

the source voltage waveform is treated as a sinusoidal wave;

however, it is learned from the experimental results that

even in the case where the voltage waveform has a certain

degree of distortion, the dc voltage-ripple-reducing func-

tion will operate without greatly being hindered.

On the other hand, the following may be cited as

future subjects and we intend to report on them in other

publications.

· Study of the capacity (maximum stored energy)

accompanying the increase and decrease of the dc

inductor current due to the application of the out-

put-voltage-ripple-reducing control; as well as the

optimization of the inductance of the inductor and

the evaluation of the dimension, weight, and cost

of the inductor.

· Study of the effect of the setting errors of the

constants of the dc inductor on the output-voltage-

ripple-reducing control.

· Study of transient responses.

· Application to the case where the dc load which

generates the ripple power is connected.

· Application to three-phase high-power-factor

converter.

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AUTHORS (from left to right)

Shin-ichi Motegi (student member) completed the master�s course in applied electronic engineering at the Science and

Engineering Research Division of the Graduate School of Tokyo Denki University in 1996. He then enrolled in the second half

of the doctoral course in applied system engineering. He is engaged in research on harmonics-reduced type converters. He is a

member of IEICE.

Akeshi Maeda (member) completed the doctoral course at the Engineering Research Division of the Graduate School of

Tokyo Denki University in 1965. He has a D.Eng. degree. After serving as a lecturer and an associate professor, he became a

professor in 1975 in the Department of Electrical Engineering of the Faculty of Engineering of the same university, and in 1977

a professor in the Department of Management Engineering of the Faculty of Science and Engineering. He has been a professor

in the Department of Applied Electronic Engineering of the Faculty of Science and Engineering since 1986. In 1970, he was a

researcher at AEG-TELEFUNKEN in Germany. He was a dean of the Faculty of Science and Engineering of Tokyo Denki

University from 1989 to 1993. He was an officer (in charge of general affairs) of the Industrial Applications Section of the IEE

Japan from 1991 to 1992, and an executive chairman of the 1993 National Conference of the Industrial Applications Section

of the IEE Japan. He is engaged in research and education on power electronics. He is a Senior Member of IEEE and the Japan

Society of Power Electronics.

70