5g mimo conformal microstrip antenna designdownloads.hindawi.com/journals/wcmc/2017/7616825.pdf ·...

Research Article5G MIMO Conformal Microstrip Antenna Design

Qian Wang1 Ning Mu1 LingLi Wang1 Safieddin Safavi-Naeini2 and JingPing Liu1

1School of Electronic and Optical Engineering Nanjing University of Science and Technology Nanjing Jiangsu 210094 China2Electrical and Computer Engineering Department University of Waterloo Waterloo ON Canada N2L 3G1

Correspondence should be addressed to JingPing Liu liujingpin2002aliyuncom

Received 14 June 2017 Revised 29 October 2017 Accepted 23 November 2017 Published 17 December 2017

Academic Editor Pai-Yen Chen

Copyright copy 2017 Qian Wang et al This is an open access article distributed under the Creative Commons Attribution Licensewhich permits unrestricted use distribution and reproduction in any medium provided the original work is properly cited

With the development of wireless communication technology 5G will develop into a new generation of wireless mobilecommunication systems MIMO (multiple-input multiple-output) technology is expected to be one of the key technologies inthe field of 5G wireless communications In this paper 4 pairs of microstrip MIMO conformal antennas of 35GHz have beendesigned Eight-element microstrip Taylor antenna array with series-feeding not only achieves the deviation of the main lobe ofthe pattern but also increases the bandwidth of the antenna array and reduces sidelobe MIMO antennas have been fabricated andmeasured Measurement results match the simulation results wellThe return loss of the antenna at 35GHz is better than 20 dB thefirst sidelobe level is minus16 dB and the angle between the main lobe and the plane of array is 60∘

1 Introduction

Multiple-input multiple-output (MIMO) technology is orig-inated from wireless communication antenna diversity tech-nology and intelligent antenna technology It is a combinationof multiple-input single-output (MISO) and single-inputmultiple-output (SIMO) and therefore has the advantagesand characteristics of the two [1 2] The MIMO system isequipped with multiple antennas at the transmitter and thereceiver It can improve the quality of wireless communica-tion and the rate of data exponentially without increasingthe bandwidth and transmitted power [3 4] Multiantennasystem is an important part of MIMO technology MIMOwireless system is not only affected by the multipath char-acteristics of the wireless communication channel but alsodepends on the design and layout of themultiantenna systemThe research of MIMOmultiantenna design mainly includesthe form of antenna element the layout of multiple antennasand the mutual coupling analysis At present the research ofMIMOmultiantenna is focused on the exploration of low costand high performance designs of antenna and layout [5ndash7]

Antennas are usually placed on the surface of the carrierto achieve the desired electromagnetic performance To thisend the conformal antenna was designed [8] Conformalantenna can be designed on the surface of the carrier which

will not damage the mechanical structure of the carrier andcan save space [9ndash11] It can be placed anywhere on the sur-face of the carrier Conformal antennas are usuallymicrostripantenna stripline antenna or crack antenna The microstripantenna has many advantages such as low profile small sizelight weight and ease to integrate with other carriers It istherefore more suitable for conformal antenna [12 13] Inaddition the millimeter-wave band has attracted a lot ofattention due to the advent of 5G technology and its inherentcharacteristics such as short wavelength wide frequencyband and propagation characteristics in the fog snow anddust environment [14ndash16]Therefore there has been extensiveresearch on millimeter-wave microstrip antenna

This paper presents the design of a MIMO conformalantenna for 5G The frequency is 35GHz the carrier ofconformal is a cylinder and the angle between the mainlobe of pattern and the carrier axis is 60∘ The sidelobecharacteristics of the antenna significantly affect the inter-ference of the system and the suppression of the clutter Theantenna designed in this paper requires the first sidelobe levelto be about minus18 dB In view of this characteristic a series-fed standing wave antenna array with Taylor distribution isdesigned Considering the influence of coupling 4 pairs ofantennas are designed The results of the research are wellsuited for the 5G MIMO communication

HindawiWireless Communications and Mobile ComputingVolume 2017 Article ID 7616825 11 pageshttpsdoiorg10115520177616825

2 Wireless Communications and Mobile Computing

2 5G MIMO Conformal Antenna Design

21 Radiation Elements Design The first part of the MIMOconformal antenna design is the radiation elements Thedesign uses microstrip patch antenna as radiation elementsThere are two main steps in the rectangular microstripantennas design The first step is theoretical analysis and thesecond step is simulation and optimization

Firstly we choose dielectric substrate For microstrip cir-cuit the loss of the microstrip is very large in the millimeter-wave band The loss can be divided into dielectric lossconductor loss and radiation loss [17] Substrates with lowloss tangent dielectric are usually chosen to reduce the dielec-tric loss When the dielectric constant is low the total lossof the microstrip would not change with the characteristicimpedance On the contrary when the substrate has highdielectric constant the loss of the microstrip will changerapidly with the characteristic impedance Thicker substratewill increase radiation losses and the surface wave is moreserious A smaller height is more effective in suppressing thehigher mode and reducing the radiation loss Additionallythe thinner substrate with good flexibility is good for confor-mal antenna [18] Taking these factors into consideration weuse RTduroid5880 (120576119903 = 22 tan 120575 = 00009) as substrateand the height of the dielectric substrate is 05mm

Next the width 119882 of the patch elements is determinedDirectivity factor of microstrip antenna radiation resistanceand other characteristics will vary with the change of 119882These characteristics directly affect the frequency bandwidthand radiation efficiency of the antenna In order to get thedesired frequency bandwidth and radiation efficiency thechoice of 119882 is particularly important The size of the widthshould meet the following requirement [18]

119882 le 1198882119891119903 (

120576119903 + 12 )minus12 (1)

where 119888 is speed of light and 119891119903 is the resonant frequency Inthis design 119891119903 is 35GHz

The next step is to determine the length 119871 of the patchelements The size of unit length 119871 is determined by theeffective dielectric constant and the operating frequencyEffective dielectric constant of substrate is defined as 120576119890which is given by the following [18]

120576119890 = 12 [120576119903 + 1 + (120576119903 minus 1) (1 +

12ℎ119882 )minus12] (2)

The length 119871 of the rectangular microstrip patch antennais approximately 1205821198922 and is given by the following [18]

Δ119897 = 0412(120576119890 + 03) (119882ℎ + 0264)(120576119890 minus 0258) (119882ℎ + 08)ℎ

119871 = 1198882119891119903radic120576119890 minus 2Δ119897

(3)

From (1)ndash(3)119882 is chosen to be 338mm and 119871 is 28mmThere are three main methods to feed the rectangular

microstrip patch Microstrip line feeding is usually used to

6m

m

6mm

245mm

332

mm

157mm

046

mm

Figure 1 The model of rectangular microstrip patch antenna

minus50

minus40

minus30

minus20

minus10

0

S11

(dB)

340 345 350 355 360 365335f (GHz)

Figure 2 119878-parameter of rectangular microstrip patch antenna

design array elements coaxial feeding is usually used forsingle microstrip antenna and the electromagnetic couplingfeeding is usually used in the microstrip antenna of doublestructure [18] Microstrip patch element designed here isa radiation element in the antenna array so microstrip ischosen as feeding method

The rectangular microstrip antenna element is shownin Figure 1 119882 and 119871 of the patch element are adjustedduring the simulation process The size of 119871 mainly affectsthe resonant frequency and 119882 mainly affects impedancematching Through simulation and optimization we get that119882 = 332mm and 119871 = 245mm the width of feed line119908 = 046mm and the center of the feed line ismidpoint of119882

All simulations are performed using HFSS in the follow-ing Figure 2 is 119878-parameter of rectangular microstrip patchantenna It can be seen from the figure that return loss hasreached minus49 dB at 35GHzThe relative bandwidth for |11987811| ltminus10 dB can be found from the figure to be 66

In this design the line width of the microstrip line is046mm and the characteristic impedance is 50Ω There-fore the input impedance of the patch element needs to beclose to 50Ω Figure 3 is the input impedance of rectan-gular microstrip patch antenna From the figure the input

Wireless Communications and Mobile Computing 3

340 345 350 355 360 365335f (GHz)

45

50

55

60

65

Z(o

hm)

Figure 3 The input impedance of rectangular microstrip patchantenna

Gai

n (d

B)

f (GHz)150100500minus50minus100minus150

E

H

minus40

minus30

minus20

minus10

0

10

Figure 4 The gain of rectangular microstrip patch antenna

impedance is about 5034Ω which matches well to thecharacteristic impedance of the microstrip line

Figure 4 is the gain of rectangular microstrip patchantenna which shows that the maximum gain of the patchelement is 793 dB the 3 dB lobe-width of 119864-Plane is 76∘ andthe 3 dB lobe-width of119867-plane is 73∘The pattern of the patchis consistent with the theory and the main lobe-width of 119864-plane is slightly larger than the119867-plane

The frequency band width of the microstrip antenna isnot enough for the whole systemTherefore in the followingwe will analyze the array antenna tomeet the requirements ofthe frequency bandwidth

22 Theoretical Analysis of Series-Fed Array The second partof MIMO conformal antenna design is the microstrip series-fed array To get high gain low sidelobe beam scanning andbeam control we need to use the discrete radiating elementto form the array according to the appropriate excitation anddistance In this paper the requirements of the microstrip

Figure 5The radiating element connected with a fine line to realizethe feeding

array are as follows the gain is 10 dB the angle between themain lobe and plane of array is not less than 10 (|11987811| ltminus10 dB) and the first sidelobe level is about minus18 dB Thedesign of microstrip is divided into three steps The first stepis to select the feed method of the linear array the secondstep is to realize the offset of the main lobe and third step isto reduce the first sidelobe level

For the antenna array the feeding method can be formedwith parallel feed and series feed or the combination of thetwo [18 19] In this paper we use series feed as shown inFigure 5 The radiating elements are connected through amicrostrip line and the end is open circuitThe first radiatingelement is fed by a coaxial line In order to avoid the influenceof the microstrip on the antenna radiation it is necessary tomake it as thin as possible

The second step is to realize the offset of the mainlobe of the antenna pattern There are typically series-fedtraveling-wave array and the series-fed standing wave arrayFor series-fed traveling-wave array the distance betweenradiating elements can be adjusted to achieve the offset ofthe main lobe However in the design of standing wave arraywith series-feeding as long as the input impedance of the lastradiating element of the array is designed to be consistentwith the characteristic impedance of themicrostrip line it canalso play a role in impedance matching to achieve the effectof traveling-wave array This design uses a series of standingwave array as shown in Figure 5 The advantage of this arrayis that it does not require the addition of terminal load Inthe design of the radiating element the input impedanceof the radiating element is designed near the characteristicimpedance of themicrostripTherefore the radiating elementcan be regarded as thematched loadWe can change the phaserelationship between elements by adjusting the distancebetween them in order to realize the arbitrary beam directionand achieve the effect of the main lobe [20]

For the design of a series-fed traveling-wave array assum-ing that the main lobe angle from the end fire direction is 120579the relationship between the main lobe direction angle andthe radiating element spacing is as follows

cos 120579 = 120582120582119892 minus

120582119878 (4)

where 119878 is the distance between the radiating elements and 120582119892is the effective wavelength in themediumWhen the distance119878 lt 120582119892 the main lobe biases feed otherwise it biases loadElement spacing 119878 is an important parameter influencing theradiation characteristics of an antenna array In order to avoidgrating lobes radiating element spacing 119878 needed to meet thefollowing formula

119878 lt 12058201 + 1003816100381610038161003816cos 1205791198981003816100381610038161003816 (5)


Table 1 Normalized current value of eight-element Taylor array

Unit number 1 2 3 4 5 6 7 8Normalized current 06 063 083 1 1 083 063 06

Table 2 11987812 of eight-element Taylor array

Unit number 2 3 411987812 minus31792 minus28252 minus22721

Formulas (4) and (5) are for traveling wave In this paperthe design of the standing wave array can also use these twoformulas The distance between the radiating elements of thestanding wave array obtained by the above two formulas is441mm

The third step is to reduce the sidelobe level The antennadesign is based on 8-element linear array Because of the needto achieve the main lobe deviation the distance between theradiating elements is consistent The sidelobe amplitude canbe reduced by controlling the currentThe current amplitudedistribution design is based on the Taylor distribution [2122] In the comprehensive design of the Taylor it is necessaryto determine the ratio 119877 of the main lobe level to thesidelobe level The value of 119860 is calculated by 119877 Under theguarantee of 119899 ge 21198602 + 12 selecting appropriate 119899 (thevalue of 119899 increases the value of 120590 decreases and the lobe-width narrows down) The value of 119899 should not be toolarge otherwise the amplitude distribution of the currentwill change dramatically After selecting 119899 beam broadeningfactor 120590 and current amplitude distribution of each radiatingelement can be calculated Ratio of the main lobe level to thesidelobe level is 119877 = minus18 dB 119899 = 4 The normalized currentvalues of all levels are shown in Table 1

There are two methods to change the current amplitudedistributionThe first one is 1205824 impedance transformer andthe second is the patch width distribution method Due tothe relatively small spacing of the radiating elements the1205824 section cannot be added so the patch width distributionmethod is used to change the current amplitude distributionIn fact the change of current amplitude distribution can berealized by changing the radiation admittance of the elementFirstly 11987812 of eight-element Taylor array at all levels shouldbe calculated according to the current distribution Feedingin this paper is from the center to both ends of the arrayTherefore according to the symmetry only half of the arrayneeds to be considered where calculating 11987812 The calculatedresults are shown in Table 2 According to these values thewidth of each radiating element can be adjusted and theappropriate size can be found to satisfy the current amplitudedistribution through simulation and optimization

23 Simulation and Analysis of Microstrip Series-Fed LinearArray The model of rectangular array with uniform distri-bution of one-end feeding is shown in Figure 6 119883119874119884 planeis the plane of the array The rectangular microstrip patcheswith the same shape are used to design the array elementThespacing between patches is the same First we adjust the unitspacing to meet the main direction deflection of the beams

of the microstrip series-fed linear array Then the array isconnected to the external 50 ohm coaxial line Because theimpedance of the whole array is not matched to the 50 ohmcoaxial line an impedance transforming section should beadded at the front of the array to match the impedance Thelength of the section is 1205821198924 After optimization the distancebetween the radiating element and the radiating element is419mm

The 119878-parameter of a rectangular array with uniformdistribution of one-end feeding is shown in Figure 7 It can beseen from the figure that at the center frequency 35GHz thereturn loss is minus277 dBThe relative bandwidth |11987811| lt minus10 dBcan be found from the figure to be 2614

The impedance of a rectangular array with uniformdistribution of one-end feeding is shown in Figure 8 It canbe seen that the antenna is well matched at 488Ω

The119864-plane gain is shown in Figure 9Themaximumgainis 1379 dB The first sidelobe level is minus132 dB The main lobedeflection offset is achieved on119864-plane the angle is about 60∘and the 3 dB lobe-width in the 119864-plane is 168∘

The first sidelobe level is higher which cannot meet thedesign requirementsWe need to find the appropriate spacingbetween elements tomeet themain beamdeflectionThen thecurrent distribution is designed to reduce the sidelobe level

In this paper the Taylor current distribution is used toreduce the sidelobe level Taylor distribution is usually used inthe form of intermediate feedThe spacing between radiationunits usually takes one wavelength As the design needs toachieve main beam offset the spacing is no longer a wave-length The feed position is required to be transferred to thecenter of the array and the form needs to be adjusted For anarray which makes main beam deviation through changingthe spacing between the radiation units we in fact change thecurrent phase difference between the radiating elements andthen the main beam is offset Considering the current phasedifference for the whole array add serpentine at one side ofthe array to adjust the phase difference and at another side ofthe serpentine the phase difference of 180 degrees should beadded at the beginning of the unit Then adjust the currentphase difference between the left and right arrays

From the simulation results we know that the unitspacing which can satisfy the main beam offset is 419mmThe next step is to transfer the feed position to the centerof the array and add serpentine at one side of the arraydetermine the length of serpentine through the simulationBefore the design of Taylor matrix we need to design auniformly distributed rectangular array with intermediatefeed to decide the length of serpentine The design is shownin Figure 10119883119874119884 plane is the model plane

The rectangular microstrip patches with the same shapeare used to design the array element The spacing betweenantennas is the sameThe feed structure of this array is in themiddle of the array and is connected to a 50Ω coaxial lineThe design process is similar to that of a rectangular arraywith uniform distribution at one end of the feed The designis divided into two parts As can be seen from Figure 10 theleft end of the feed is the same as the uniform distribution ofone-end feeding In the right end serpentine lines are addedto realize phase array progression so as to realize the beam


03m

m

03 mm

12m

m

24 mm

02m

m

33m

m

14 mm

19mm 229 mm

Figure 6 Model of rectangular array with uniform distribution of one-end feeding

minus30

minus25

minus20

minus15

minus10

minus5

S11

(dB)

32 34 36 38 4030f (GHz)

Figure 7 119878-parameter diagram of a rectangular array with uniformdistribution of one-end feeding

30

40

50

60

70

80

90

100

Z(o

hm)

34 36 38 4032f (GHz)

Figure 8 Impedance of a rectangular array with uniform distribu-tion of one-end feeding

deviation After the simulation the length of the serpentineis found to be 651mm

The 119878-parameter of a rectangular array with uniformdistribution and intermediate feeding is shown in Figure 11It can be seen from the plot that at the center frequencyof 35GHz the return loss is up to minus3187 dB The relativebandwidth |11987811| lt minus10 dB can be found to be 217

The impedance is shown in Figure 12 It can be seen fromthe figure that the antenna is well matched at 515Ω The gaingraph of 119864-plane is shown in Figure 13 It can be seen fromthe figure that themaximumgain is 1336 dB the first sidelobelevel is minus137 dB the main beam deflection offset is achieved

Gai

n (d

B)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

05

101520

1401201000 20 40 60 80 160 180minus40minus60 minus20minus80

Theta (deg)

Figure 9 The gain of 119864-plane of a rectangular array with uniformdistribution of one-end feeding

on 119864-plane the angle is about 62∘ and the 3 dB lobe-width inthe 119864-plane is 18∘

The designed array has satisfied the requirement of themain beam offset but the sidelobe level is still too high Thefeeding position is in the middle of the array which satisfiesthe design of Taylor distribution The design of Taylor distri-bution is carried out based on this array The model diagramis shown in Figure 14119883119874119884 plane is the plane of the array

The form of the array is the same as that of the middlefeed and there is a difference in the size of the array The sizeof the array element is designed according to Taylor currentdistribution regulation 11987812 of the elements at all levels whichcan be obtained from Table 2 By adjusting the radiation edgesize of each element the radiation admittance of each elementcan be changed and the corresponding value of the radiationedge size can be obtained

After simulation and optimization from the feed point tothe right the sizes of radiation side are1198821 = 37mm1198822 =34mm1198823 = 41mm1198824 = 32mm

The 119878-parameter diagram of a rectangular array with Tay-lor distribution and intermediate feed is shown in Figure 15It can be seen from the figure that at the center frequencythe return loss is very high (212 dB at 35GHz) The relativebandwidth |11987811| lt minus10 dB can be found from the picture tobe 116

The impedance is shown in Figure 16 It can be seen thatthe antenna is well matched at 501Ω The gain graph of 119864-plane is shown in Figure 17 It can be seen from the figure


03

mm

03

mm

19 mm 19 mm

02

mm

229 mm

33

mm

03 mm

Figure 10 A uniformly distributed rectangular array with intermediate feeding

30 31 32 33 34 35 36 37 38 39 40 4129f (GHz)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

0

S11

(dB)

Figure 11 119878-parameter of a rectangular array with uniform distri-bution with intermediate feeding

30 31 32 33 34 35 36 37 38 39 4029f (GHz)

20

40

60

80

100

120

140

160

Z(o

hm)

Figure 12 Impedance of a rectangular array with uniform distribu-tion with intermediate feeding

that the maximum gain is 139 dB the first sidelobe level isminus156 dB the main beam deflection offset is achieved on 119864-plane the angle is about 60∘ and the 3 dB lobe-width in the119864-plane is 20∘

Three kinds of arrays are given in this designThe first twoarrays actually provide reference for the Taylor distributionmatrix The first array provides an appropriate spacing of the

Gai

n (d

B)

minus60 minus40 minus20minus80 20 40 60 80 100 120 140 160 1800Theta (deg)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

05

101520

Figure 13 The gain of 119864-plane of a rectangular array with uniformdistribution with intermediate feeding

radiating elements The second one determines the lengthof the serpentine The final array form is based on the twoarrays to adjust the radiation side of each radiating element torealize the current redistribution In the array of rectangularpatch the gain is higher in the form of uniform distributionwith intermediate feedThe lowest sidelobe level is the Taylordistribution with intermediate feed to reduce the sidelobeThe narrowest beam and the best matched impedance are theuniform distribution with one end of the feed It can be seenthat the reduction of the first sidelobe level is at the expenseof width of the main lobe

3 Design of Conformal Arrays

The conformal array is designed in the third part of theMIMO conformal antenna and the conformal carrier is thecylinder with a diameter of 60mm [23]

The center frequency of the design is 35GHz and thedielectric substrate with relative dielectric constant 120576119903 = 22is selected Thickness of the substrate is 05mm Accordingto the design of the radiation unit the size of the microstrippatch antenna is only about 3mm The curvature of 60mmcylindrical diameter is smaller than the microstrip patchantenna So the antenna can be regarded as a planar antennaand analyzed by the theory of planar antenna The designneeds to achieve a specific beam direction which is 60∘ tothe conformal vector axis and can be realized by conformal


03 mm

03

mm

03

mm

03

mm

41

mm 19 mm19 mm

34

mm

37

mm

02 mm

32

mm

285 mm

Figure 14 A rectangular array with Taylor distribution with intermediate feed

minus25

minus20

minus15

minus10

minus5

0

S11

(dB)

32 33 34 35 36 37 38 3931f (GHz)

Figure 15 119878-parameter of a rectangular array with Taylor distribu-tion with intermediate feeding

32 33 34 35 36 37 38 39 4031f (GHz)

20

40

60

80

100

120

140

160

180

Z(o

hm)

Figure 16 Impedance of a rectangular arraywithTaylor distributionwith intermediate feeding

array According to the analysis of the series-feed array themicrostrip patch antenna can be composed of a series-feedarray to achieve such a beam direction It can be realizedby adjusting the spacing between the elements The lowsidelobe can be realized by the Taylor synthesis method Thedistribution current of the antenna array is tapered to reducethe sidelobe level

Gai

n


Theta (deg)

minus30

minus25

minus20

minus15

minus10

minus5

0

5

10

15

20

Figure 17 The gain of 119864-plane of a rectangular array with Taylordistribution with intermediate feeding

Figure 18 The microstrip antenna conformal array

The microstrip antenna conformal array is shown inFigure 18

Through simulation and optimization the radiation char-acteristic of microstrip antenna array is obtained The 119878-parameter is shown in Figure 19 The return loss of theconformal array is down to 14 dB at 35GHz The relativebandwidth of |11987811| lt minus10 dB can be calculated to be 118

The input impedance of the array is shown in Figure 20It can be seen that the whole antenna is matched to 45Ω The


32 33 34 35 36 37 38 39 4031f (GHz)

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)

Figure 19 119878-parameter of conformal microstrip antenna array

20

40

60

80

100

120

140

160

180

200

Z(o

hm)

32 33 34 35 36 37 38 39 40 41 4231f (GHz)

Figure 20 Input impedance of conformal microstrip antenna array

gain of 119864-plane microstrip antenna conformal array is shownin Figure 21 It can be seen that themaximumgain of the arrayis 133 dB and the first sidelobe level is minus16 dB The 119864-planerealized the main beam offset The offset angle is about 62∘The 3 dB beam width of 119864 plane is 182∘

We fabricated a pair of 8-element series-fed conformalantenna array as shown in Figure 22 The analysis and testresults are compared119878-parameters are shown in Figure 23 It can be seenfrom the results of the 119878-parameters of the antenna thatthe resonance point of the measurement and simulation isconsistent The measurement results of the antenna relativebandwidth is about 11 which is slightly less than thesimulation results

The comparison of measurement results and simulationresults of normalized pattern of119864 plane is shown in Figure 24It can be seen that the radiation plot of 119864-plane achieves themain beam offset The angle is about 62∘ and this meets therequirementsThe first sidelobe level rose to minus14 dBThe 3 dBlobe-width of 119864-plane is about 17∘ The measurement resultsare in good agreement with the simulation results

Gai

n (d

B)

minus60 minus40 minus20 0 20 40 60 80 100 120 140 160 180minus80

Theta (deg)

minus30

minus20

minus10

0

10

20

Figure 21 Gain of 119864-plane of conformal microstrip antenna array

Figure 22 Eight-element series-fed conformal antenna array

The gain measurement is performed by comparing to astandard horn antenna The gain of the antenna is 122 dB

4 Coupling Analysis of 5G MIMOConformal Antenna

It is also very important to decide the number of the anten-nas in the design of MIMO conformal antenna Antennacoupling has significant impact on radiation pattern Whenusing multiple antennas the cross coupling between theantennas should be discussed and the coupling needs to beminimized The main factor that affects the coupling is thedistance between antennas The closer the distance betweenantennas the stronger the couplingTherefore we should findthe appropriate antenna spacing to meet the performancerequirement of the antenna coupling At the same timethis determines the maximum number of antennas [24] Forconformal system the diameter of the conformal carrier is60mm The carrier space is limited so the number of RFcircuits is limited At the same time too many RF circuitswill increase the cost of the system Therefore the numberof antennas needs to be determined by considering thesefactors

As we know the energy of array antenna can be coupledby space wave or surface wave when the coupling level isgreater than minus20 dB the performance of the antenna willbe greatly affected [25] In this design each antenna is used


SimulatedMeasured

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)

32 34 36 38 40 4230f (GHz)

Figure 23 Comparison of 119878-parameters and simulation results ofthe 8-element series-feed conformal antenna array

SimulatedMeasured

Gai

n (d

B)

minus50

minus40

minus30

minus20

minus10

0

0 50 100 150minus50

Theta (deg)

Figure 24 Comparison of the measurement and simulation resultsof the 119864 surface orientation of the 8-element series-fed conformalantenna array

as an independent antenna so the requirement of couplingbetween the antennas is low and the coupling betweenantennas isminus40 dBThrough simulation and optimization wefind thatwhen the number of the antennas is eight it canmeetthe requirement of the coupling of antenna to be less thanminus40 dB Eight antenna arrays can be placed on the conformalcarrier Actually we cannot put so many antennas First of allto consider the cost of the RF circuit eight arrays of antennasrequire eight RF links Secondly to consider the volume ofthe conformal carrier not more than four radio frequencylinks can be placed in this limited spaceTherefore this designuses four arrays Combined with the spatial symmetry of theantenna the four pairs of antennas are distributed with equaldistance in the conformal cylindrical carrier The simulationmodel is shown in Figure 25

Figure 25 MIMO radar conformal antenna

32 33 34 35 36 37 38 39 40 4131f (GHz)

minus140

minus120

minus100

minus80

minus60

minus40

minus20

0

S(d

B)

S11

S12

S13

S14

Figure 26 119878-parameters of MIMO conformal antenna 1

The 119878-parameters of the first antenna in the MIMO radarconformal antenna are shown in Figure 26 It can be seenfrom the figure that the antennarsquos reflection parameters areminus215 dB Considering the coupling of antennas it is clear thatthe cross coupling satisfies the requirements of minus40 dB

The radiation plot of the first antenna in theMIMO radarconformal antenna is shown in Figure 27 It can be seen fromthe figure that by considering the coupling effects the firstsidelobe level has been significantly improved to minus115 dBThis is consistent with the theoretical analysis

The fabricated conformal antenna with 4 arrays of anten-nas is shown in Figure 28


20

minus80

minus60

minus40

minus20 40 60

minus100 0 80 100

120

140

160

180

200

minus120

Theta (deg)

Gai

n (d

B)

minus40

minus30

minus20

minus10

0

10

20

Figure 27 Gain of 119864-plane MIMO conformal antenna 1

Figure 28 Planar expansion of four pairs conformal microstripantenna

5 Conclusion

In this paper in order tomeet the requirements of the antennasystem an 8-cell series-fedmicrostrip standing wave antennaarray by traveling-wave theory has been designed in Ka band(35GHz) The deflection of the main lobe and the planeof array is realized by adjusting the spacing between theelements At the same time the Taylor distribution is used forthe antenna synthesis and the first sidelobe level is reducedby controlling the current amplitude of the unit NextMIMOconformal antenna at 35GHz is designed The bandwidth ofantenna is greater than 10 the gain is greater than 10 dB andthe first sidelobe level is reduced tominus16 dBThe angle betweenthe main lobe and the carrier axis is 60∘ The measurementresults agree well with the simulation results which meetthe requirements of the system to the antenna performanceConsidering the cost of system space limitation and antennacoupling we design four 8-cell series-fedmicrostrip standingwave antenna arrays The four antenna arrays are evenly dis-tributed on the conformal of carrier and the cross couplingof the antenna is lower than minus40 dBConflicts of Interest

The authors declare that they have no conflicts of interest

References

[1] YaoHuan Gong ldquoMultiple input multiple out of smart antennatechnologyrdquo ZTE Technology vol 6 pp 19ndash21 2002

[2] Li Gang Ren and Mei Song ldquoMIMO technology in mobilecommunication [J]rdquoModern telecommunication technology vol1 pp 42ndash45 2004

[3] G J Foschini ldquoLayered space-time architecutre of wirelesscommunication in a fding environment when using multi-element antennasrdquo Bell Labs Technical Journal vol l no 2 pp41ndash59 1996

[4] G J Foschini ldquoOn limits ofwireless communications in a fadingenvironment when using multiple antennasrdquo Wireless PersonalCommunications vol 6 no 3 pp 311ndash335 1998

[5] WeiHong Xiao Multi Antenna Design for MIMO Mobile Com-munication System Xidian University Xirsquoan China 2006

[6] YanJie Zhang Research on Coupling Characteristics of MIMOAntena Xidian University Xirsquoan China 2012

[7] Z Yang H Yang and H Cui ldquoA compact MIMO antennawith inverted C-shaped ground branches for mobile terminalsrdquoInternational Journal of Antennas and Propagation vol 2016Article ID 3080563 2016

[8] Kyungjung Kim Sarkar M C Wicks et al ldquoDOA EstimationUtilizing Directive Elements on a Conformal Surfacerdquo in RadarConference 2003 Proceeding of the 2003 IEEE pp 91ndash96Huntsville Alabama 2003

[9] R KHerseyW LMelvin and J HMcClellan ldquoClutter-limiteddetection performance of multi-channel conformal arraysrdquoSignal Processing vol 84 no 9 pp 1481ndash1500 2004

[10] Z Li X Kang J Su Q Guo Y L Yang and J Wang ldquoClutterLimited Detection Performance of Multi-channel ConformalArraysrdquo International Journal of Antennas and Propagation vol2016 Article ID 9812642 2016

[11] T E Morton and K M Pasala ldquoPattern synthesis of conformalarrays for airborne vehiclesrdquo in Proceedings of the 2004 IEEEAerospace Conference Proceedings pp 1030ndash1038 Big SkyMon-tana USA March 2004

[12] R K Mishra and A Patnaik ldquoDesign of circular microstripantanna using neural networksrdquo IETE Journal of Research vol44 no 1-2 pp 35ndash39 2015

[13] A Sayed R Ghonam andA Zekry ldquoDesign of a Compact DualBand Microstrip Antenna for Ku-Band Applicationsrdquo Interna-tional Journal of Computer Applications vol 115 no 13 pp 699ndash702 2015

[14] DongLiang Zhao ldquoThoughts on the development of 5G mobilecommunicationrdquo Information Communication vol 9 2015

[15] HongJie Yao ldquoThe key technology and process of 5G mobilecommunicationrdquo Communication World vol 6 2015

[16] Warren LStutzman and Gary A thiele Encyclopedia of RF andMicrowave Engineering Peoplersquos Posts and TelecommunicationsPublishing House China 2nd edition 2006

[17] YuFang Tang Theoretical study and engineering application ofmicrostrip line loss Nanjing University of Science and Technol-ogy 2009

[18] Jun zhang and KeCheng Liu Microstrip antenna theory andEngineering National Defense Industry Press China 1988

[19] ShuJie Li Research on Microstrip Planar Array Antenna in KuBand Xidian University Xirsquoan China 2006

[20] DaJunWu Design And Research of Millimeter Wave CylindricalConformal Microstrip Antenna Nanjing University of Scienceand Technology Nanjing China 2007

[21] XingJian Kang Principle And Design of Antenna NationalDefense Industry Press China 1995

[22] ChuFang Xie Modern Antenna Theory Chengdu Telecommu-nication Engineering Institute Press Sichuan China 1987

[23] Song Zhu ldquoDevelopment of conformal antenna and its appli-cation in electronic warfarerdquo Journal of the Chinese Academy ofElectronic Science vol 12 2007


[24] ChangSheng Shi ldquoIsolation of antennardquo Electronic Science andTechnology Revie vol 11 pp 16ndash18 1997

[25] QiaoLong Lan Research on Millimeter Wave Microstrip An-tenna University of Electronic Science and technology Cheng-du China 2006

RoboticsJournal of

Hindawi Publishing Corporationhttpwwwhindawicom Volume 2014


Active and Passive Electronic Components

Control Scienceand Engineering

Journal of


International Journal of

RotatingMachinery


Hindawi Publishing Corporation httpwwwhindawicom

Journal of

Volume 201

Submit your manuscripts athttpswwwhindawicom

VLSI Design



Shock and Vibration


Civil EngineeringAdvances in

Acoustics and VibrationAdvances in



Electrical and Computer Engineering

Journal of

Advances inOptoElectronics


Volume 2014

The Scientific World JournalHindawi Publishing Corporation httpwwwhindawicom Volume 2014

SensorsJournal of


Modelling amp Simulation in EngineeringHindawi Publishing Corporation httpwwwhindawicom Volume 2014


Chemical EngineeringInternational Journal of Antennas and

Propagation




Navigation and Observation



DistributedSensor Networks



2 5G MIMO Conformal Antenna Design

21 Radiation Elements Design The first part of the MIMOconformal antenna design is the radiation elements Thedesign uses microstrip patch antenna as radiation elementsThere are two main steps in the rectangular microstripantennas design The first step is theoretical analysis and thesecond step is simulation and optimization

Firstly we choose dielectric substrate For microstrip cir-cuit the loss of the microstrip is very large in the millimeter-wave band The loss can be divided into dielectric lossconductor loss and radiation loss [17] Substrates with lowloss tangent dielectric are usually chosen to reduce the dielec-tric loss When the dielectric constant is low the total lossof the microstrip would not change with the characteristicimpedance On the contrary when the substrate has highdielectric constant the loss of the microstrip will changerapidly with the characteristic impedance Thicker substratewill increase radiation losses and the surface wave is moreserious A smaller height is more effective in suppressing thehigher mode and reducing the radiation loss Additionallythe thinner substrate with good flexibility is good for confor-mal antenna [18] Taking these factors into consideration weuse RTduroid5880 (120576119903 = 22 tan 120575 = 00009) as substrateand the height of the dielectric substrate is 05mm

Next the width 119882 of the patch elements is determinedDirectivity factor of microstrip antenna radiation resistanceand other characteristics will vary with the change of 119882These characteristics directly affect the frequency bandwidthand radiation efficiency of the antenna In order to get thedesired frequency bandwidth and radiation efficiency thechoice of 119882 is particularly important The size of the widthshould meet the following requirement [18]

119882 le 1198882119891119903 (

120576119903 + 12 )minus12 (1)

where 119888 is speed of light and 119891119903 is the resonant frequency Inthis design 119891119903 is 35GHz

The next step is to determine the length 119871 of the patchelements The size of unit length 119871 is determined by theeffective dielectric constant and the operating frequencyEffective dielectric constant of substrate is defined as 120576119890which is given by the following [18]

120576119890 = 12 [120576119903 + 1 + (120576119903 minus 1) (1 +

12ℎ119882 )minus12] (2)

The length 119871 of the rectangular microstrip patch antennais approximately 1205821198922 and is given by the following [18]

Δ119897 = 0412(120576119890 + 03) (119882ℎ + 0264)(120576119890 minus 0258) (119882ℎ + 08)ℎ

119871 = 1198882119891119903radic120576119890 minus 2Δ119897

(3)

From (1)ndash(3)119882 is chosen to be 338mm and 119871 is 28mmThere are three main methods to feed the rectangular

microstrip patch Microstrip line feeding is usually used to

6m

m

6mm

245mm

332

mm

157mm

046

mm

Figure 1 The model of rectangular microstrip patch antenna

minus50

minus40

minus30

minus20

minus10

0

S11

(dB)

340 345 350 355 360 365335f (GHz)

Figure 2 119878-parameter of rectangular microstrip patch antenna

design array elements coaxial feeding is usually used forsingle microstrip antenna and the electromagnetic couplingfeeding is usually used in the microstrip antenna of doublestructure [18] Microstrip patch element designed here isa radiation element in the antenna array so microstrip ischosen as feeding method

The rectangular microstrip antenna element is shownin Figure 1 119882 and 119871 of the patch element are adjustedduring the simulation process The size of 119871 mainly affectsthe resonant frequency and 119882 mainly affects impedancematching Through simulation and optimization we get that119882 = 332mm and 119871 = 245mm the width of feed line119908 = 046mm and the center of the feed line ismidpoint of119882

All simulations are performed using HFSS in the follow-ing Figure 2 is 119878-parameter of rectangular microstrip patchantenna It can be seen from the figure that return loss hasreached minus49 dB at 35GHzThe relative bandwidth for |11987811| ltminus10 dB can be found from the figure to be 66

In this design the line width of the microstrip line is046mm and the characteristic impedance is 50Ω There-fore the input impedance of the patch element needs to beclose to 50Ω Figure 3 is the input impedance of rectan-gular microstrip patch antenna From the figure the input


340 345 350 355 360 365335f (GHz)

45

50

55

60

65

Z(o

hm)


Gai

n (d

B)


E

H

minus40

minus30

minus20

minus10

0

10











cos 120579 = 120582120582119892 minus

120582119878 (4)


119878 lt 12058201 + 1003816100381610038161003816cos 1205791198981003816100381610038161003816 (5)



















03m

m

03 mm

12m

m

24 mm

02m

m

33m

m

14 mm

19mm 229 mm


minus30

minus25

minus20

minus15

minus10

minus5

S11

(dB)

32 34 36 38 4030f (GHz)


30

40

50

60

70

80

90

100

Z(o

hm)

34 36 38 4032f (GHz)





Gai

n (d

B)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

05

101520


Theta (deg)









03

mm

03

mm

19 mm 19 mm

02

mm

229 mm

33

mm

03 mm


30 31 32 33 34 35 36 37 38 39 40 4129f (GHz)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

0

S11

(dB)


30 31 32 33 34 35 36 37 38 39 4029f (GHz)

20

40

60

80

100

120

140

160

Z(o

hm)




Gai

n (d

B)


minus35

minus30

minus25

minus20

minus15

minus10

minus5

05

101520







03 mm

03

mm

03

mm

03

mm

41

mm 19 mm19 mm

34

mm

37

mm

02 mm

32

mm

285 mm


minus25

minus20

minus15

minus10

minus5

0

S11

(dB)

32 33 34 35 36 37 38 3931f (GHz)


32 33 34 35 36 37 38 39 4031f (GHz)

20

40

60

80

100

120

140

160

180

Z(o

hm)



Gai

n


Theta (deg)

minus30

minus25

minus20

minus15

minus10

minus5

0

5

10

15

20







32 33 34 35 36 37 38 39 4031f (GHz)

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)


20

40

60

80

100

120

140

160

180

200

Z(o

hm)

32 33 34 35 36 37 38 39 40 41 4231f (GHz)





Gai

n (d

B)


Theta (deg)

minus30

minus20

minus10

0

10

20








SimulatedMeasured

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)

32 34 36 38 40 4230f (GHz)


SimulatedMeasured

Gai

n (d

B)

minus50

minus40

minus30

minus20

minus10

0

0 50 100 150minus50

Theta (deg)




32 33 34 35 36 37 38 39 40 4131f (GHz)

minus140

minus120

minus100

minus80

minus60

minus40

minus20

0

S(d

B)

S11

S12

S13

S14






20

minus80

minus60

minus40

minus20 40 60

minus100 0 80 100

120

140

160

180

200

minus120

Theta (deg)

Gai

n (d

B)

minus40

minus30

minus20

minus10

0

10

20



5 Conclusion



References



























RoboticsJournal of





Journal of



RotatingMachinery



Journal of

Volume 201


VLSI Design



Shock and Vibration







Journal of



Volume 2014


SensorsJournal of





Propagation










340 345 350 355 360 365335f (GHz)

45

50

55

60

65

Z(o

hm)


Gai

n (d

B)


E

H

minus40

minus30

minus20

minus10

0

10











cos 120579 = 120582120582119892 minus

120582119878 (4)


119878 lt 12058201 + 1003816100381610038161003816cos 1205791198981003816100381610038161003816 (5)



















03m

m

03 mm

12m

m

24 mm

02m

m

33m

m

14 mm

19mm 229 mm


minus30

minus25

minus20

minus15

minus10

minus5

S11

(dB)

32 34 36 38 4030f (GHz)


30

40

50

60

70

80

90

100

Z(o

hm)

34 36 38 4032f (GHz)





Gai

n (d

B)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

05

101520


Theta (deg)









03

mm

03

mm

19 mm 19 mm

02

mm

229 mm

33

mm

03 mm


30 31 32 33 34 35 36 37 38 39 40 4129f (GHz)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

0

S11

(dB)


30 31 32 33 34 35 36 37 38 39 4029f (GHz)

20

40

60

80

100

120

140

160

Z(o

hm)




Gai

n (d

B)


minus35

minus30

minus25

minus20

minus15

minus10

minus5

05

101520







03 mm

03

mm

03

mm

03

mm

41

mm 19 mm19 mm

34

mm

37

mm

02 mm

32

mm

285 mm


minus25

minus20

minus15

minus10

minus5

0

S11

(dB)

32 33 34 35 36 37 38 3931f (GHz)


32 33 34 35 36 37 38 39 4031f (GHz)

20

40

60

80

100

120

140

160

180

Z(o

hm)



Gai

n


Theta (deg)

minus30

minus25

minus20

minus15

minus10

minus5

0

5

10

15

20







32 33 34 35 36 37 38 39 4031f (GHz)

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)


20

40

60

80

100

120

140

160

180

200

Z(o

hm)

32 33 34 35 36 37 38 39 40 41 4231f (GHz)





Gai

n (d

B)


Theta (deg)

minus30

minus20

minus10

0

10

20








SimulatedMeasured

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)

32 34 36 38 40 4230f (GHz)


SimulatedMeasured

Gai

n (d

B)

minus50

minus40

minus30

minus20

minus10

0

0 50 100 150minus50

Theta (deg)




32 33 34 35 36 37 38 39 40 4131f (GHz)

minus140

minus120

minus100

minus80

minus60

minus40

minus20

0

S(d

B)

S11

S12

S13

S14






20

minus80

minus60

minus40

minus20 40 60

minus100 0 80 100

120

140

160

180

200

minus120

Theta (deg)

Gai

n (d

B)

minus40

minus30

minus20

minus10

0

10

20



5 Conclusion



References



























RoboticsJournal of





Journal of



RotatingMachinery



Journal of

Volume 201


VLSI Design



Shock and Vibration







Journal of



Volume 2014


SensorsJournal of





Propagation



























03m

m

03 mm

12m

m

24 mm

02m

m

33m

m

14 mm

19mm 229 mm


minus30

minus25

minus20

minus15

minus10

minus5

S11

(dB)

32 34 36 38 4030f (GHz)


30

40

50

60

70

80

90

100

Z(o

hm)

34 36 38 4032f (GHz)





Gai

n (d

B)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

05

101520


Theta (deg)









03

mm

03

mm

19 mm 19 mm

02

mm

229 mm

33

mm

03 mm


30 31 32 33 34 35 36 37 38 39 40 4129f (GHz)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

0

S11

(dB)


30 31 32 33 34 35 36 37 38 39 4029f (GHz)

20

40

60

80

100

120

140

160

Z(o

hm)




Gai

n (d

B)


minus35

minus30

minus25

minus20

minus15

minus10

minus5

05

101520







03 mm

03

mm

03

mm

03

mm

41

mm 19 mm19 mm

34

mm

37

mm

02 mm

32

mm

285 mm


minus25

minus20

minus15

minus10

minus5

0

S11

(dB)

32 33 34 35 36 37 38 3931f (GHz)


32 33 34 35 36 37 38 39 4031f (GHz)

20

40

60

80

100

120

140

160

180

Z(o

hm)



Gai

n


Theta (deg)

minus30

minus25

minus20

minus15

minus10

minus5

0

5

10

15

20







32 33 34 35 36 37 38 39 4031f (GHz)

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)


20

40

60

80

100

120

140

160

180

200

Z(o

hm)

32 33 34 35 36 37 38 39 40 41 4231f (GHz)





Gai

n (d

B)


Theta (deg)

minus30

minus20

minus10

0

10

20








SimulatedMeasured

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)

32 34 36 38 40 4230f (GHz)


SimulatedMeasured

Gai

n (d

B)

minus50

minus40

minus30

minus20

minus10

0

0 50 100 150minus50

Theta (deg)




32 33 34 35 36 37 38 39 40 4131f (GHz)

minus140

minus120

minus100

minus80

minus60

minus40

minus20

0

S(d

B)

S11

S12

S13

S14






20

minus80

minus60

minus40

minus20 40 60

minus100 0 80 100

120

140

160

180

200

minus120

Theta (deg)

Gai

n (d

B)

minus40

minus30

minus20

minus10

0

10

20



5 Conclusion



References



























RoboticsJournal of





Journal of



RotatingMachinery



Journal of

Volume 201


VLSI Design



Shock and Vibration







Journal of



Volume 2014


SensorsJournal of





Propagation










03

mm

03

mm

19 mm 19 mm

02

mm

229 mm

33

mm

03 mm


30 31 32 33 34 35 36 37 38 39 40 4129f (GHz)

minus35

minus30

minus25

minus20

minus15

minus10

minus5

0

S11

(dB)


30 31 32 33 34 35 36 37 38 39 4029f (GHz)

20

40

60

80

100

120

140

160

Z(o

hm)




Gai

n (d

B)


minus35

minus30

minus25

minus20

minus15

minus10

minus5

05

101520







03 mm

03

mm

03

mm

03

mm

41

mm 19 mm19 mm

34

mm

37

mm

02 mm

32

mm

285 mm


minus25

minus20

minus15

minus10

minus5

0

S11

(dB)

32 33 34 35 36 37 38 3931f (GHz)


32 33 34 35 36 37 38 39 4031f (GHz)

20

40

60

80

100

120

140

160

180

Z(o

hm)



Gai

n


Theta (deg)

minus30

minus25

minus20

minus15

minus10

minus5

0

5

10

15

20







32 33 34 35 36 37 38 39 4031f (GHz)

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)


20

40

60

80

100

120

140

160

180

200

Z(o

hm)

32 33 34 35 36 37 38 39 40 41 4231f (GHz)





Gai

n (d

B)


Theta (deg)

minus30

minus20

minus10

0

10

20








SimulatedMeasured

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)

32 34 36 38 40 4230f (GHz)


SimulatedMeasured

Gai

n (d

B)

minus50

minus40

minus30

minus20

minus10

0

0 50 100 150minus50

Theta (deg)




32 33 34 35 36 37 38 39 40 4131f (GHz)

minus140

minus120

minus100

minus80

minus60

minus40

minus20

0

S(d

B)

S11

S12

S13

S14






20

minus80

minus60

minus40

minus20 40 60

minus100 0 80 100

120

140

160

180

200

minus120

Theta (deg)

Gai

n (d

B)

minus40

minus30

minus20

minus10

0

10

20



5 Conclusion



References



























RoboticsJournal of





Journal of



RotatingMachinery



Journal of

Volume 201


VLSI Design



Shock and Vibration







Journal of



Volume 2014


SensorsJournal of





Propagation










03 mm

03

mm

03

mm

03

mm

41

mm 19 mm19 mm

34

mm

37

mm

02 mm

32

mm

285 mm


minus25

minus20

minus15

minus10

minus5

0

S11

(dB)

32 33 34 35 36 37 38 3931f (GHz)


32 33 34 35 36 37 38 39 4031f (GHz)

20

40

60

80

100

120

140

160

180

Z(o

hm)



Gai

n


Theta (deg)

minus30

minus25

minus20

minus15

minus10

minus5

0

5

10

15

20







32 33 34 35 36 37 38 39 4031f (GHz)

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)


20

40

60

80

100

120

140

160

180

200

Z(o

hm)

32 33 34 35 36 37 38 39 40 41 4231f (GHz)





Gai

n (d

B)


Theta (deg)

minus30

minus20

minus10

0

10

20








SimulatedMeasured

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)

32 34 36 38 40 4230f (GHz)


SimulatedMeasured

Gai

n (d

B)

minus50

minus40

minus30

minus20

minus10

0

0 50 100 150minus50

Theta (deg)




32 33 34 35 36 37 38 39 40 4131f (GHz)

minus140

minus120

minus100

minus80

minus60

minus40

minus20

0

S(d

B)

S11

S12

S13

S14






20

minus80

minus60

minus40

minus20 40 60

minus100 0 80 100

120

140

160

180

200

minus120

Theta (deg)

Gai

n (d

B)

minus40

minus30

minus20

minus10

0

10

20



5 Conclusion



References



























RoboticsJournal of





Journal of



RotatingMachinery



Journal of

Volume 201


VLSI Design



Shock and Vibration







Journal of



Volume 2014


SensorsJournal of





Propagation










32 33 34 35 36 37 38 39 4031f (GHz)

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)


20

40

60

80

100

120

140

160

180

200

Z(o

hm)

32 33 34 35 36 37 38 39 40 41 4231f (GHz)





Gai

n (d

B)


Theta (deg)

minus30

minus20

minus10

0

10

20








SimulatedMeasured

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)

32 34 36 38 40 4230f (GHz)


SimulatedMeasured

Gai

n (d

B)

minus50

minus40

minus30

minus20

minus10

0

0 50 100 150minus50

Theta (deg)




32 33 34 35 36 37 38 39 40 4131f (GHz)

minus140

minus120

minus100

minus80

minus60

minus40

minus20

0

S(d

B)

S11

S12

S13

S14






20

minus80

minus60

minus40

minus20 40 60

minus100 0 80 100

120

140

160

180

200

minus120

Theta (deg)

Gai

n (d

B)

minus40

minus30

minus20

minus10

0

10

20



5 Conclusion



References



























RoboticsJournal of





Journal of



RotatingMachinery



Journal of

Volume 201


VLSI Design



Shock and Vibration







Journal of



Volume 2014


SensorsJournal of





Propagation










SimulatedMeasured

minus18

minus16

minus14

minus12

minus10

minus8

minus6

minus4

minus2

S11

(dB)

32 34 36 38 40 4230f (GHz)


SimulatedMeasured

Gai

n (d

B)

minus50

minus40

minus30

minus20

minus10

0

0 50 100 150minus50

Theta (deg)




32 33 34 35 36 37 38 39 40 4131f (GHz)

minus140

minus120

minus100

minus80

minus60

minus40

minus20

0

S(d

B)

S11

S12

S13

S14






20

minus80

minus60

minus40

minus20 40 60

minus100 0 80 100

120

140

160

180

200

minus120

Theta (deg)

Gai

n (d

B)

minus40

minus30

minus20

minus10

0

10

20



5 Conclusion



References



























RoboticsJournal of





Journal of



RotatingMachinery



Journal of

Volume 201


VLSI Design



Shock and Vibration







Journal of



Volume 2014


SensorsJournal of





Propagation










20

minus80

minus60

minus40

minus20 40 60

minus100 0 80 100

120

140

160

180

200

minus120

Theta (deg)

Gai

n (d

B)

minus40

minus30

minus20

minus10

0

10

20



5 Conclusion



References



























RoboticsJournal of





Journal of



RotatingMachinery



Journal of

Volume 201


VLSI Design



Shock and Vibration







Journal of



Volume 2014


SensorsJournal of





Propagation












RoboticsJournal of





Journal of



RotatingMachinery



Journal of

Volume 201


VLSI Design



Shock and Vibration







Journal of



Volume 2014


SensorsJournal of





Propagation









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